JPH0723070A - Demodulator for multi-carrier modulation signal - Google Patents

Abstract

PURPOSE:To reduce the hardware scale of a demodulator which demodulates a multi-carrier modulation signal. CONSTITUTION:This device is constituted of an end batch detecting means 21 which commonly detects multi-carrier modulation signals Sm from a transmission side altogether for all channels by a common local frequency signal Lc, and synchronization detecting means N 22-1, 22-2,...22-N which operate synchronization detection based on each difference frequency between the frequency of each carrier wave of the multi-carrier modulation signal and the frequency of the common local frequency signal Lc, and reproduce original data.

Description

Detailed Description of the Invention

[0001]

FIELD OF THE INVENTION This invention relates to demodulators for multi-carrier modulated signals. In a wireless communication system such as digital multiplex wireless, a technique for compensating for channel distortion due to fading in a wireless transmission space is indispensable, and various proposals such as space diversity and transversal equalizer have already been made. .

On the other hand, a multi-carrier system has been proposed and put into practical use as a technique for improving the resistance against such channel distortion and further improving the line quality.
Until this multi-carrier method was proposed, the single-carrier method was widely adopted.

[0003]

2. Description of the Related Art FIG. 5 is a diagram for explaining a multi-carrier system in comparison with a single-carrier system. In this figure, (A) shows the frequency spectrum under the single carrier system, where a single carrier (frequency f ₀ ) is
This is the spectrum when modulated with x bit / s data to be transmitted.

On the other hand, FIG. 5B shows a frequency spectrum under the multi-carrier system, which is the premise of the present invention, where N (N is an integer of 2 or more, but N = 3 in the figure). The original data (each x / 3 bit)
/ S) is a spectrum in which carrier waves (frequency f ₁ , f ₂ and f ₃ ) divided into N (N = 3) are respectively modulated one to one.

The greatest advantage of adopting the multi-carrier system is that the resistance to the above-mentioned transmission line distortion is improved and the line quality is greatly improved. In FIG. 5, it is assumed that the frequency characteristic is inclined (represented by a chain line) due to the influence of fading. Assuming that the band-to-band deviation due to this slope appears as y dB under the single carrier system (A), each carrier (f ₁ , f in the multi carrier system (B) has the same slope as this. _In each band having the center frequency of ₂ and f ₃ ), it appears only as a deviation of y / 3 dB.
Conversely, in the multi-carrier method (B), the resistance to fading is tripled as compared with the single-carrier method (A).

FIG. 6 is a diagram showing a conventional example of a communication system based on the multicarrier system. In the figure, the left side is the transmission side and the right side is the reception side with the wireless transmission space 4 shown in the center as the center, and the present invention mainly refers to the demodulator 10 for a multicarrier modulation signal on the reception side. First, in the multicarrier modulator 1 on the transmission side, x [bit is divided into N (N is an integer of 2 or more) pieces.
/ S] original data D1, D2 ... DN, each of which is divided into N carrier waves C1, C2 ... CN are modulated one-to-one by modulation units 1-1, 1-2 ... 1-N, respectively. The multi-carrier modulated signal S _m obtained by combining these in the hybrid circuit 2 is transmitted from the transmitting-side antenna 3, received by the receiving-side antenna 13, and separated by the hybrid circuit 12 to obtain the multi-carrier modulated signal. The demodulator 10 for carrier modulation signal demodulates and reproduces the original data.

The demodulator 10 for multi-carrier modulation signal is
It is comprised from the demodulation unit 11-1, 11-2 ... 11-N corresponding to each said modulation unit 1-1, 1-2 ... 1-N. In the figure, the frequency spectrum of the multicarrier modulation signal S _m is shown near the wireless transmission space 4,
This corresponds to FIG. 5 (B). Further, in the figure, reference numerals 14-1, 14-2 ... 14-N are band pass filter (F) for extracting the modulated signals (# 1, # 2 ... #N) of each carrier.

Thus, by synthesizing the outputs of all demodulation units (11), the original data D1, D2 ... DN can be reproduced at the output of the demodulator 10 for multicarrier modulation.

[0009]

According to the conventional demodulator 10 for a multi-carrier modulation signal shown in FIG. 6, the resistance to fading is improved, but the hardware scale required under the single carrier system is reduced. Carrier wave C
The hardware scale is N times as large as the number N of 1, C2 ... CN.

Therefore, in view of the above problems, it is an object of the present invention to propose a demodulator for a multi-carrier modulation signal which can reduce the amount of hardware as compared with the conventional one.

[0011]

FIG. 1 is a block diagram showing the principle configuration of the present invention. In this figure, reference numeral 21
22-N is a collective detection means, and 22-1 is a synchronous detection means. That is, in the demodulator 10 for multicarrier modulation signal according to the present invention, the collective detection means 21 collectively detects the received multicarrier modulation signal S _m by the common local frequency signal L _C to obtain N synchronous detection signals. The means 22-1, 22-2, ... 22-N include the collective detection means 21.
From the N carrier waves C1,
Synchronous detection is performed by each of N difference frequencies between each frequency of C2 ... CN and the frequency of the common local frequency signal L _C , and N synchronous detection means 22-1, 22-2, ...
The original data D1, D2 ... DN is reproduced from -N.

[0012]

The detector required for detection, which is the first stage of processing on the receiving side, conventionally required N detectors corresponding to the respective carriers C1, C2 ... CN. This can be realized by the collective detection means 21, and the amount of hardware in this portion can be reduced to 1 / N.

[0013]

DESCRIPTION OF THE PREFERRED EMBODIMENTS Before explaining the embodiments of the present invention, the principle of the present invention will be explained in detail. Note that this description will be made by taking an example of the case where a 4-phase phase modulation (quadrature modulation) signal is received under the 3 multi-carrier method, which is the most practical condition. However, even when using four or more types of carrier waves, when receiving a two-phase phase-modulated signal, or when eight-phase or one-phase
It can also be applied to the case of receiving a phase modulated signal of 6 phases or the like.

FIG. 2 is a frequency spectrum diagram for explaining the principle of the present invention. This figure shows the frequency spectrum exhibited by the multi-carrier modulation signal S _m under the 3-multi-carrier method, and the first carrier C1 (angular frequency ω ₁ ).
The first modulated signal centered at is # 1, and similarly the second modulated signal is
The first and second modulated signals having the center frequencies of the third and third carriers C2 and C3 (angular frequencies ω ₂ and ω ₃ ) are shown as # 2 and # 3, respectively.

In FIG. 2, the angular frequency of the common local frequency signal L _C given to the collective detection means 21 shown in FIG. 1 is shown as ω _C. Furthermore, in FIG. 6, each modulation unit (1-1, 1-2 and 1-3) on the transmission side.
Quadrature modulation data output from I, i (In-p
hase) ch (Channel) data I and Q
Let (quadrature-phase) ch data Q be I ₁ , Q ₁ ; I ₂ , Q ₂ ; I ₃ , Q _{3, respectively,} and let time be t.

Then, the first, second and third modulated signals (# 1, # 2 and # 3) are expressed by the following equations (1),
It can be represented by (2) and (3). _{# 1 = I 1 sinω 1 t} + Q 1 cosω 1 t (1) # 2 = I 2 sinω 2 t + Q 2 cosω 2 t (2) # 3 = I 3 sinω 3 t + Q 3 cosω 3 t (3) After all, from the sender The received multi-carrier signal transmitted through the wireless transmission space 4 and received by the receiving side is expressed by the following equation (4).
Can be expressed as However, it is represented as a received signal IF after being converted to a low frequency by a down converter.

IF = # 1 + # 2 + # 3 (4) When this received signal IF is applied to the collective detection means 21 of FIG. 1, it is collectively detected by the common local frequency signal L _C here. Thus, the quadrature demodulation signals I _DET and Q _DET quadrature detected at the angular frequency ω _C are I _DET = IF × sin
from _{ω C t, Q DET = IF} × cosω C t, is represented by the following formulas (5) and (6).

I _DET = (# 1 ＋ # 2 ＋ # 3) sin ω _C t ＝ I ₁ sin ω ₁ tsin ω _C t ＋ Q ₁ cos ω ₁ tsin ω _C t ＋ I ₂ sin ω ₂ tsin ω _C t ＋ Q ₂ cos ω _{2 tsin ω C t + I 3 sin} ω 3 sinω C t + Q 3 cos ω 3 tsin ω C t = 0.5 _{[{I 1 (-cos (ω 1} + ω C) t + cos (ω 1 -ω C) t) + Q 1 × ( sin (ω ₁ + ω _C ) t−sin (ω ₁ −ω _C ) t)} + {I ₂ (−cos (ω ₂ + ω _C ) t + cos (ω ₁ −ω _C ) t) + Q ₂ (sin (ω ₁ + Ω _C ) t-sin (ω ₁ −ω _C ) t)} + {I ₃ (−cos (ω ₃ + ω _C ) t + cos (ω ₃ −ω _C ) t) + Q ₃ (sin (ω ₃ + ω _C ) t −sin (ω ₃ −ω _C ) t)}] (5) Q _DET ＝ (# 1 ＋ # 2 ＋ # 3) cos ω _C t ＝ I ₁ sin ω ₁ tcos ω _C t ＋ Q ₁ cos ω ₁ tcos ω _C t ＋ I ₂ sin ω ₂ tcos ω _C t ＋ Q ₂ cos ω ₂ tcos ω _C t ＋ I ₃ sin ω ₃ tcos ω _C t ＋ Q ₃ cos ω ₃ tcos ω _C t = 0.5 [{I ₁ (sin (ω ₁ + ω _C ) t + sin (ω ₁ −ω _C ) t) ＋ Q ₁ (cos (ω ₁ ＋ ω _C ) t ＋ cos (ω ₁ −ω _C ) t) ＋＋ {I ₂ (sin (ω ₂ + ω _C ) t + sin (ω ₂ −ω _{C ) t) + Q 2 (cos} (ω 2 + ω C) t + cos (ω 2 - _{C) t)} + {I} 3 (sin (ω 3 + ω C) t + sin (ω 3 -ω C) t) + Q 3 (cos (ω 3 + ω C) t + cos (ω 3 -ω C) t)} ] ... (6) From the above equation (5) representing I _DET , the high frequency component (ω _N + ω
Next, the low-pass filtered demodulation signal I _{det from} which _C ) (N = 1, 2, and 3) has been removed is generated. Similarly, the low-pass filtered demodulation signal Q _det from which the high frequency component (ω _N + ω _C ) (N = 1, 2 and 3) has been removed is next generated from the above equation (6) representing Q _DET . The low-pass filtered demodulated signals I _det and Q _det thus generated can be expressed by the following equations (7) and (8), respectively.

Q _det = 0.5 [{I ₁ sin (ω ₁ −ω _C ) t + Q ₁ cos (ω ₁ −ω _C ) t} + {I ₂ sin (ω ₂ −ω _C ) t + Q ₂ cos (ω ₂ − ω _C ) t} + {I ₃ sin (ω ₃ − ω _C ) t + Q ₃ cos (ω ₃ − ω _C ) t}]… (7) I _det = 0.5 [{I ₁ cos (ω ₁ − ω _C ) t−Q ₁ sin (ω ₁ −ω _C ) t} + {I ₂ cos (ω ₂ −ω _C ) t−Q ₂ sin (ω ₂ −ω _C ) t} + {I ₃ cos (ω ₃ −ω _C ) t−Q ₃ sin (ω ₃ −ω _C ) t}] (8) The angular frequencies appearing here are ω ₁ −ω _C , ω ₂ −ω _C and ω ₃ −ω _C , and Carriers C1, C2 and C of
It deviates from each angular frequency (ω ₁ , ω ₂ and ω ₃ ) of ₃ by ω _C. Therefore, the low-pass filtered demodulated signals I _det and Q
First, second and third synchronous detection means 22 for _det
-1, 22, 2-2 and 22-3 perform synchronous detection to decode data I ₁ , Q ₁ ; I ₂ , Q ₂ ; I ₃ , Q ₃ and obtain corresponding original data D 1, D 2 and D 3 respectively. Reproduce. In addition, these synchronous detection means (22) are an infinite phase shifter (EPS: Endless Phase) mentioned later.
It is preferable to configure the shifter).

Synchronous detection means (EPS) 22-1, 22
Channel-corresponding orthogonal decoded data I ₁ ′, Q ₁ ′; I ₂ ′, Q ₂ ′; I ₃ ′, Q ₃ ′ are output from 2 and 22-3. Here, the channel-corresponding orthogonal decoded data of these three channels are collectively referred to as I _N ′ and Q _N ′. N
Are N = 1, 2 and 3. _{Then, I N '= I det cosω} N' t + Q det sinω N 't (9) Q N' = I det sinω N 't-Q det cosω N' t (10) becomes operational row in the synchronous detection means Be seen. First, in order to obtain the channel-corresponding decoded data I ₁ ′ and Q ₁ ′ in the first channel (C1), the angular frequency ω _N ′ in the above equations (9) and (10) is used.
Substitute for ω _N ′ = ω _N −ω _C (11). In this case, since N = 1, ω _N ′ is ω
Substitute ₁ − ω _C.

Then, regarding the I _ch data I ₁ ′, the above equation (9) is developed as the following equation (12). I ₁ ′ = cos (ω ₁ −ω _C ) [{I ₁ cos (ω ₁ −ω _C ) t−Q ₁ sin (ω ₁ −ω _C ) t} + {I ₂ cos (ω ₂ −ω _C ) t−Q ₂ sin (ω ₂ −ω _C ) t} + {I ₃ cos (ω ₃ −ω _C ) t−Q ₃ sin (ω ₃ −ω _C ) t} + sin (ω ₁ −ω _C ) [{ I ₁ sin (ω ₁ −ω _C ) t + Q ₁ cos (ω ₁ −ω _C ) t} + {I ₂ sin (ω ₂ −ω _C ) t + Q ₂ cos (ω ₂ −ω _C ) t} + {I ₃ sin (ω ₃ −ω _C ) t + Q ₃ cos (ω ₃ −ω _C ) t}] (12) Further, regarding the Q _ch data Q ₁ ′, the above equation (10) is expanded as the following equation (13). Q ₁ ′ = sin (ω ₁ −ω _C ) t [{I ₁ cos (ω ₁ −ω _C ) t−Q ₁ sin (ω ₁ −ω _C ) t} + {I ₂ cos (ω ₂ − ω _C ) t−Q ₂ sin (ω ₁ −ω _C ) t} + {I ₃ cos (ω ₃ −ω _C ) t−Q ₃ sin (ω ₃ −ω _C ) t} −cos (ω ₁ −ω _C ) t [{I ₁ sin (ω ₁ −ω _C ) t ＋ Q ₁ cos (ω ₁ −ω _C ) t} ＋ {I ₂ sin (ω ₂ −ω _C ) t ＋ Q ₂ cos (ω ₂ −ω _C ) t } + {I ₃ sin (ω ₃ −ω _C ) t + Q ₃ cos (ω ₃ −ω _C ) t}] (13) By the way, the above equation (12) is further It is developed as the following expression (14).

= 0.5 [{I ₁ (cos 2 (ω ₁ −ω _C ) t + cos (0)) −Q ₁ (sin 2 (ω ₁ −ω _C ) t + sin (0))} + {I ₂ (cos (ω ₁ + Ω ₂ −2ω _C ) t + cos (ω ₁ −ω ₂ ) t) −Q ₂ (sin (ω ₁ + ω ₂ −2ω _C ) t + sin (ω ₁ −ω ₂ ) t} + {I ₃ (cos (ω ₁ + ω ₃ − 2ω _C ) t + cos (ω ₁ −ω ₃ ) t) −Q ₃ (sin (ω ₁ + ω ₃ −2ω _C ) t + sin (ω ₁ −ω ₃ ) t} + 0.5 [{I ₁ (−cos 2 (ω ₁₋ ω _C ) t ＋ cos (0)) ＋ Q ₁ (sin2 (ω ₁ −ω _C ) t ＋ sin (0)} ＋ {I ₂ (−cos (ω ₁ ＋ ω ₂ −2 ω _C ) t + cos (ω ₁ −ω ₂ ) t) + Q ₂ (sin (ω ₁ + ω ₂ − 2ω _C ) t + sin (ω ₁ − ω ₂ ) t} + {I ₃ (−cos (ω ₁ + ω ₃ − 2ω _C ) t + cos (ω ₁ − ω ₃ ) t ) + Q ₃ (sin (ω ₁ + ω ₃ −2ω _C ) t + sin (ω ₁ −ω ₃ ) t} = 0.5 [2I ₁ + 2I ₂ cos (ω ₁ −ω ₂ ) t + 2I ₃ cos (ω ₁ −ω ₃ ) t] = I ₁ + I ₂ cos (ω ₁ −ω ₂ ) t + I ₃ cos (ω ₁ −ω ₃ ) t (14) As is clear from the final equation of the equation (14), I _ch original data I ₁ appears as a direct current component . Circuit on the signal component of the frequency (ω ₁ -ω _2), be removed by a filter or the like signal component of the frequency (ω ₁ -ω _3), original data I ₁ of the I _ch seeking is decoded.

Similarly, the above equation (13) is further transformed into the following equation (1)
It is developed like 5). = 0.5 [{I ₁ (sin2 (ω ₁ −ω _C ) t + sin (0)) −Q ₁ (−cos 2 (ω ₁ −ω _C ) t + cos (0))} + {I ₂ (sin (ω ₁ + ω ₂ −2ω _C ) t + sin (ω ₁ −ω ₂ ) t −Q ₂ (−cos (ω ₁ + ω ₂ −2ω _C ) t + cos (ω ₁ −ω ₂ ) t} ＋ {I ₃ (sin (ω ₁ + ω ₃ − 2ω _C ) t + sin (ω ₁ −ω ₃ ) t −Q ₃ (−cos (ω ₁ + ω ₂ −2ω _C ) t + cos (ω ₁ −ω ₃ ) t} − 0.5 [{I ₁ (sin 2 (ω ₁ −ω _C ) t−sin (0)) ＋ Q ₁ (cos2 (ω ₁ −ω _C ) t ＋ cos (0)) ＋＋ {I ₂ (sin (ω ₁ ＋ ω ₂ −2ω _C ) t ＋ sin (ω ₁ −ω ₂ ) t + Q ₂ (cos (ω ₁ + ω ₂ −2ω _C ) t + cos (ω ₁ −ω ₂ ) t} + {I ₃ (sin (ω ₁ + ω ₃ −2ω _C ) t + cos (ω ₁ −ω ₃ ) t + Q ₃ ( cos (ω ₁ + ω ₂ −2ω _C ) t + cos (ω ₁ −ω ₃ ) t} = − {Q ₁ + Q ₂ cos (ω ₁ −ω ₂ ) t + Q ₃ cos (ω ₁ −ω ₃ ) t}… ( 15) As is clear from the final expression of the above expression (15), the original data Q _{1 of} Q _ch appears as a direct current component.On the circuit, the signal component of the frequency (ω ₁ −ω ₂ ) and the frequency component (ω ₁ − be removed by a filter or the like signal components of omega _3), is the raw data to Q ₁ Q _ch seeking, is decoded as a -Q _{1.-Q 1} may be returned to Q ₁ through an inverter.

The above description of the equation (10) has been made with respect to the first channel (C1), but the same applies to the remaining second and third channels (12) to (8).
The equation (15) is applied to obtain I ₂ , Q ₂ and I ₃ , Q ₃ . FIG. 3 is a diagram showing an embodiment according to the present invention. In this figure, the collective detection means 21 shown in FIG. 1 and the N synchronous detection means 22-1, 22-2 ...
They are located in the parts with the same reference numbers.

First, the collective detection means 21 is realized by a general quadrature detector composed of two hybrid circuits 41 and 42 and two mixers 43 and 44, as shown in the figure. The hybrid circuit 42 demultiplexes the common local frequency signal L _C into two signals of 0 ° and 90 °. These two signals are further applied to the quadrature detection mixers 43 and 44, respectively.

Collective detection means 21 and N synchronous detection means 2
2-1, 22-2, ... Between the carrier waves (C1, C2
Frequency (omega ₁ of ...), inserts the omega ₂ ...) and a common local frequency signal collectively low-pass filter means 51 for extracting a signal component having a difference frequency between the L _C of the frequency (omega _C). The collective low-pass filtering means 51 can be configured by a general low-pass filter.

By the collective low-pass filtering means 51, the high-frequency component (ω _N + ω) appearing in the above-mentioned equations (5) and (6).
_C ) (N = 1, 2 ...) Is removed, and the aforementioned low-pass filtered demodulated signals I _det and Q _det are obtained. Here, the analog signal is converted into a digital signal. The analog / digital converter (A / D) 52 does this. CK is a sampling clock. However, the positions of these A / Ds are not limited to those shown in the figure.

N synchronous detection means (22-shown in FIG. 1)
, 22-2 ... 22-N) are respectively composed of the infinite phase shifters (45-1, 45-2 ... 45-N) described above. Each of the infinite phase shifters 45-1, 45-2, ..., 45-N commonly receives the output subjected to the collective quadrature detection by the collective detection means 21 via the collective low-pass filtering means 51, and the collective quadrature. Sinω is added to the in-phase component and the quadrature component of the detected output.
Perform a complex operation that multiplies t and cosωt, where
The angular frequency ω has a value proportional to each of the N difference frequencies. This complex operation is described in (9) above.
It is as described in the equation, the equation (10), the equation (12), and the equation (13). Therefore, the angular frequency ω is the same as ω _N ′ (ω ₁ ′, ω ₂ ′ ...
ω _N ′), that is, ω _N ′ = ω _N −ω _C (N = 1,
2 ...).

FIG. 4 is a diagram showing a specific example of an infinite phase shifter. In the figure, the infinite phase shifter 45 is composed of four multipliers 4
6, and an adder 47 and a subtractor 48. The calculation is as shown in the equations (9) and (10). Ω in the figure corresponds to ω _N ′ in these equations (9) and (10) (N = 1, 2).
…).

Returning to FIG. 3, each infinite phase shifter 45-1, 4
N individual low-pass filtering means 55-1 and 55-2 for extracting corresponding original data from the outputs of 5-2 ... 45-N.
55-N is provided, and a data synthesizing means 56 for reproducing the original data by sequentially extracting the outputs from the individual low-pass filtering means is provided. The data synthesizing means 56 is composed of a normal discriminator and a parallel / serial converter.

The above individual low-pass filtering means 55-1 and 55-
2 ... 55-N can be configured by a general low-pass filter, and the frequency (ω ₁ −ω ₂ ) and the frequency (ω ₁ −ω ₃ ) of the equations (14) and (15) can be selected. It serves to remove the signal component and retrieve the original data I ₁ and Q ₁ . In the embodiment of FIG. 3, N individual low-pass filtering means 5 are provided.
5-1, 55-2 ... 55-N, in response to the outputs from N, respectively, N carrier recovery units (CR) 61-1 and 61-, which output N kinds of sin ωt and cos ωt, respectively.
2 ... 61-N. The N species sinωt and cosωt referred to herein, the (11) is that of formula omega 'sin .omega _N expressed in' _N t and cos .omega _N 't, N
Is N = 1, 2, ...

As the carrier wave reproducing section (61), for example, a well-known inverse modulation type carrier wave reproducing circuit can be used, and the VCO at the final stage of the inverse modulation type carrier wave reproducing circuit.
It is necessary to decompose the output of (1) into a sin component and a cos component and apply it to the infinite phase shifter. Therefore, it is conceivable to use the ROM as the simplest method. This ROM is V
A pair of sin component data and cos component data corresponding to each output of CO is stored in advance, and the output of VCO is used as an address input to output the corresponding sin component data and cos component data.

In the embodiment shown in FIG. 3, N individual low-pass filtering means 55-1, 55-2, ... 55-N and N carrier wave recovery units 61-1, 61-2, ... 61-N are provided. Between the transversal equalization units (T / EQL) 71-1 and 71-2, respectively.
Insert 71-N. In principle, it is not always necessary to provide a transversal equalization unit, but waveform equalization by the transversal equalization unit enables faster and more accurate carrier wave reproduction and data decoding.

Further, the transversal equalizer 71-
Clock recovery units (BTR) 72-1, 72-2, ... For defining the respective operation timings of 1, 71-2 ... 71-N.
72-N to the individual low-pass filtering means 55-1, 55-2 ... 5
Provided on each output side of 5-N. It is necessary to use a clock recovery unit to perform sampling when the eye pattern is most open.

In the embodiment shown in FIG. 3, the analog / digital converter (A / D) 52 is provided between the collective detecting means 21 and the synchronous detecting means 22-1, 22-2 ... 22-N.
It may be inserted in the signal line corresponding to the position indicated by A / D.1 or the position indicated by A / D.2 in FIG. As is clear from FIG. 3, the synchronous detection means 22-1, 22-2 ... 22-
N has hardware corresponding to each channel (carrier), and the amount of hardware is considerably large. Therefore, the hardware part corresponding to these N channels should be as LS as possible.
It is desirable to have a design suitable for I conversion. Then, at least in the preceding stage of the hardware part corresponding to the N channel, it is necessary to be converted into a digital signal. From this, these synchronous detection means (22)
Analog / digital converter 52 from each input to the transmitting side
It is desirable to provide (for example, A / D-1).

[0036]

As described above, according to the present invention, all channels (carriers C1, C2 ... C) are provided in the first stage on the receiving side.
Since a single quadrature detector shared by N) is provided, it is not necessary to provide a detector (N detectors) individually for each channel, and the amount of hardware is reduced. The hardware subsequent to this single quadrature detector corresponds to each channel, but the hardware corresponding to each channel (EPS, CR, BT
According to the current technology, R, T, EQL, etc.) are extremely easy to be integrated into an LSI, and their ratio to the total hardware scale is small.

[Brief description of drawings]

FIG. 1 is a block diagram showing a principle configuration of the present invention.

FIG. 2 is a frequency spectrum diagram for explaining the principle of the present invention.

FIG. 3 is a diagram showing an embodiment according to the present invention.

FIG. 4 is a diagram showing a specific example of an infinite phase shifter.

FIG. 5 is a diagram illustrating a multi-carrier method in comparison with a single-carrier method.

FIG. 6 is a diagram showing a conventional example of a communication system based on a multicarrier system.

[Explanation of symbols]

10 ... Demodulator for multicarrier modulation signal 21 ... Collective detection means 22-1, 22-2 to 22-N ... Synchronous detection means 31 ... Quadrature detector 45-1, 45-2 to 45-N ... Infinite phase shifter 51 ... Collective low-pass filtering means 52 ... Analog / digital converters 55-1, 55-2 to 55-N ... Individual low-pass filtering means 56 ... Data synthesizing means 61-1, 61-2 to 61-N ... Carrier wave recovery unit 71-1, 71-2 to 71-N ... Transversal equalization unit 72-1, 72-2 to 72-N ... Clock recovery unit

Claims (10)

[Claims] 1. N carrier waves (C) each having N (N is an integer of 2 or more) original data (D1, D2 ... DN).
, C2 ... CN) are respectively modulated one-to-one, and a multi-carrier modulated signal (S _m ) obtained by combining them is transmitted from the transmitting side, and the multi-carrier modulated signal received at the receiving side is demodulated. reproducing the original data Te in a multi-carrier modulated signal demodulator (10), and said multicarrier modulated signal common local frequency signal received (L _C) bulk detection means for detecting at once by (21), wherein While receiving the output from the collective detection means in common, N
N synchronous detection means (22-1, 22-2 ...) Which respectively perform synchronous detection by N difference frequencies between the frequencies of the carrier waves and the frequency of the common local frequency signal.
22-N), wherein the original data is reproduced by the N synchronous detection means. 2. The frequency of each carrier and the common local frequency signal are provided between the collective detection means (21) and the N synchronous detection means (22-1, 22-2 ... 22-N). The demodulator for a multi-carrier modulated signal according to claim 1, wherein a collective low-pass filter means (51) for extracting a signal component having a frequency difference from the frequency is inserted. 3. N synchronous detection means (22-1, 2-2, 1)
2-2 ... 22-N) are infinite phase shifters (45-
45-2 ... 45-N), The demodulator for multicarrier modulation signals according to claim 2. 4. Each of the infinite phase shifters (45-1, 45-2)
45-N) commonly receives the outputs which have been subjected to the collective quadrature detection by the collective detection means (21) via the collective low-pass filtering means (51), and has the same phase of the outputs subjected to the collective quadrature detection. Sin ωt and cos ωt for the component and the orthogonal component
Is performed, and the angular frequency ω is
The demodulator for a multicarrier modulation signal according to claim 3, wherein the demodulator has a value that is proportional to each of the N difference frequencies. 5. The infinite phase shifters (45-1, 45-2)
45-N), the N individual low-pass filtering means (55-1, 55-) for extracting the corresponding original data from the output of
55-N) and data synthesizing means (56) for retrieving the original data by sequentially extracting the outputs from the individual low-pass filtering means. Demodulator. 6. N individual low-pass filtering means (55-
, 55-2 ... 55-N) in response to the respective outputs from the N carrier recovery sections (61-1, 61-2 ...) respectively outputting the N kinds of sin ωt and cos ωt.
61-N), the demodulator for a multicarrier modulation signal according to claim 5. 7. N individual low-pass filtering means (55-
, 55-2 ... 55-N) and the N carrier recovery sections (61-1, 61-2 ... 61-N) between the transversal equalization sections (71-1, 71-2), respectively. ... 71-
The demodulator for a multicarrier modulation signal according to claim 6, wherein N) is inserted. 8. N transversal equalizers (7)
7-1, 71-2 ... 71-N) clock recovery units (72-1, 72-2 ... 7) for defining respective operation timings
2-N) to the N individual low-pass filtering means (55-1,
The demodulator for a multicarrier modulation signal according to claim 7, which is provided on each output side of 55-2 ... 55-N). 9. The demodulator for a multi-carrier modulation signal according to claim 1, further comprising an analog / digital conversion section (52) for converting an analog signal into a digital signal. 10. The N synchronous detection means (22-1,
22-2. 22-N) The demodulator for a multi-carrier modulation signal according to claim 9, wherein the analog / digital conversion unit (52) is provided on the transmission side from each input.