JP5658497B2 - Semiconductor light-emitting element lighting device and lighting fixture using the same - Google Patents

Description

  The present invention relates to a lighting device for a semiconductor light emitting element for dimming and lighting a semiconductor light emitting element such as a light emitting diode (LED), and a lighting fixture using the same.

  Patent Document 1 (Japanese Patent Application Laid-Open No. 2005-294063) discloses an LED lighting device as shown in FIG. The LED lighting device includes a switching element Q1 connected in series to a DC power source 2 and controlled to be turned on / off at a high frequency; connected to the switching element Q1 in series, and a current flows from the DC power source 2 when the switching element Q1 is turned on. An inductance element L1; a regenerative diode D1 that discharges energy stored in the inductance element L1 to the LED 4 when the switching element Q1 is turned on; and a current detection resistor that detects a current flowing through the switching element Q1 R1; when the current value detected by the current detection resistor R1 reaches a predetermined value (ON voltage of the transistor Tr1), when the switching element Q1 is turned off and the energy emission of the inductance element L1 is completed (diode D2) Turns off And a control means for turning on the switching element Q1 when).

  However, since the LED lighting device of Patent Document 1 does not have a dimming function, it cannot be used as an LED dimming lighting device.

  Patent Document 2 (Japanese Patent Publication No. 2003-522393) proposes a concept of PWM dimming an LED illumination module at 100 Hz or 120 Hz synchronized with a commercial power frequency (50/60 Hz). No means were disclosed.

JP-A-2005-294063 Japanese translation of PCT publication No. 2003-522393

  According to the present invention, when the current flowing through the inductance element reaches a predetermined value when the switching element is turned on, the switching element is controlled to be off, and when the switching element is turned off, the current discharged from the inductance element to the semiconductor light emitting element through the regenerative diode is zero. Accordingly, an object of the present invention is to propose a circuit means for accurately dimming and lighting a semiconductor light emitting element with a simple configuration in a lighting device for a semiconductor light emitting element that controls the switching element to be turned on.

In order to solve the above problems, another invention of the present application is, as shown in FIG. 1, a switching element Q1 connected in series to a DC power source and controlled to be turned on and off at a high frequency; and connected in series to the switching element Q1. An inductor L1 through which a current flows from the DC power source when the switching element Q1 is turned on; and a regeneration that discharges energy stored in the inductor L1 to the semiconductor light emitting element 4 when the switching element Q1 is turned off. A diode D1, a current detection means (resistor R1) for detecting a current flowing through the switching element Q1, and when the current value detected by the current detection means reaches a predetermined value, the switching element Q1 is turned off and the inductor When the energy release of L1 is completed, the switching element Q1 In a semiconductor light emitting device lighting device comprising a control circuit 5 to be turned on, the control circuit 5 adjusts the semiconductor light emitting element 4 by intermittently blocking the operation of detecting the completion of energy emission of the inductor L1 at a low frequency. It is characterized by light.

In order to solve the same problem, the invention of claim 1 includes a switching element Q1 connected in series to a DC power source and controlled to be turned on and off at a high frequency, as shown in FIG. 1; and connected in series to the switching element Q1. An inductor L1 through which a current flows from the DC power source when the switching element Q1 is turned on; and a regenerative diode that discharges energy stored in the inductor L1 to the semiconductor light emitting element 4 when the switching element Q1 is turned off. D1 and; current detecting means for detecting a current flowing through the switching element Q1 (the resistor R1); when the detected current value by said current detecting means reaches a predetermined value, the conjunction turns off the switching element Q1 When the energy release of the inductor L1 is completed, the switching element Q1 is turned on. In a lighting device for a semiconductor light emitting device comprising a control circuit 5 for turning on the semiconductor light emitting device, the semiconductor light emitting device 4 is formed by intermittently forming a state where the current value detected by the current detecting means reaches a predetermined value at a low frequency. It is characterized by dimming.

The invention of billed claim 1, by reducing the previous SL predetermined value below intermittently zero at low frequencies, the state where the current value detected by said current detecting means reaches a predetermined value at a low frequency It is characterized by being formed intermittently.

In order to solve the same problem, the invention of claim 2 includes a switching element Q1 connected in series to a DC power source and controlled to be turned on and off at a high frequency, as shown in FIG. 1; and connected in series to the switching element Q1. An inductor L1 through which a current flows from the DC power source when the switching element Q1 is turned on; and a regenerative diode that discharges energy stored in the inductor L1 to the semiconductor light emitting element 4 when the switching element Q1 is turned off. D1; current detection means (resistor R1) for detecting the current flowing through the switching element Q1; when the current value detected by the current detection means reaches a predetermined value, the switching element Q1 is turned off and the inductor L1 When the energy release of the switching element Q1 is completed. In a lighting device for a semiconductor light emitting device comprising a control circuit 5 for turning on the semiconductor light emitting device, the semiconductor light emitting device 4 is formed by intermittently forming a state where the current value detected by the current detecting means reaches a predetermined value at a low frequency. It is characterized by dimming. The invention of billed second aspect, by pre-SL current the larger current value than a predetermined value to a current value detected by the detecting means is intermittently superimposed at a low frequency, is detected by the current detecting means A state in which the current value reaches a predetermined value is intermittently formed at a low frequency.

According to a third aspect of the present invention, in the lighting device for a semiconductor light emitting device according to the first or second aspect, the operation of intermittently blocking the operation of detecting the completion of the energy emission of the inductor L1 at a low frequency is simultaneously performed. It is characterized by.

According to a fourth aspect of the present invention, in the lighting device for a semiconductor light emitting element according to any one of the first to third aspects, the control electrode of the switching element Q1 is short-circuited in synchronization with the low frequency.

According to a fifth aspect of the present invention, in the lighting device for a semiconductor light emitting element according to any one of the first to fourth aspects, the frequency of the low frequency is 100 Hz to 2 kHz and is switched in synchronization with an electronic shutter of a video camera. It is characterized by.

A sixth aspect of the present invention is the lighting device for a semiconductor light emitting element according to any one of the first to fourth aspects, wherein the low frequency frequency is set to an integral multiple of the reciprocal of the shutter speed of the video camera. .

The invention of claim 7 is a luminaire comprising a lighting device of the semiconductor light-emitting device according to any one of claims 1 to 6, a semiconductor light-emitting element current is supplied from the lighting device (Fig. 6) .

  According to the present invention, when the current flowing through the switching element reaches a predetermined value, the switching element is controlled to be turned off. After the switching element is turned off, the switching element is turned on when the discharge of the energy accumulated in the inductance element is completed. In a lighting device for a semiconductor light-emitting element comprising a control means for on-control, the control means intermittently prevents the operation of detecting the completion of energy release of the inductance element at a low frequency, or the current flowing through the switching element By forming the detection value at a predetermined value intermittently at a low frequency, the current flowing through the semiconductor light-emitting element can be accurately adjusted with a simple configuration, and a semiconductor capable of high-precision light control A lighting device for a light emitting element can be realized at low cost.

It is a circuit diagram of the lighting device of Embodiment 1 of the present invention. It is the circuit diagram which simplified and showed the internal structure of the control integrated circuit used for the lighting device of Embodiment 1 of this invention. It is a block circuit diagram which shows the whole structure of the LED light control lighting apparatus using the lighting device of Embodiment 1 of this invention. It is a circuit diagram of the lighting device of Embodiment 2 of the present invention. It is a circuit diagram of various switching power supply circuits to which the present invention can be applied. It is sectional drawing which shows schematic structure of the lighting fixture of Embodiment 5 of this invention. It is a circuit diagram of a conventional example.

(Embodiment 1)
FIG. 1 is a circuit diagram of a lighting device according to Embodiment 1 of the present invention. The lighting device includes a power connector CON1 and an output connector CON2. A commercial AC power supply (100 V, 50/60 Hz) is connected to the power connector CON1. A semiconductor light emitting element 4 such as a light emitting diode (LED) is connected to the output connector CON2. The semiconductor light emitting element 4 may be an LED module in which a plurality of LEDs are connected in series, in parallel, or in series-parallel.

  A DC power supply circuit 2b is connected to the power supply connector CON1 via a current fuse FUSE and a filter circuit 2a. The filter circuit 2a includes a surge voltage absorbing element ZNR, filter capacitors Ca and Cb, and a common mode choke coil LF. Although the DC power supply circuit 2b is shown here as a rectifying / smoothing circuit including a full-wave rectifier DB and a smoothing capacitor C0, it may be a power factor correction circuit using a boost chopper circuit.

  A step-down chopper circuit 3 is connected to the output terminal of the DC power supply circuit 2b. The step-down chopper circuit 3 includes an inductor L1 connected in series to the semiconductor light emitting element 4 that is lit by a direct current, and is connected in series between the series circuit of the inductor L1 and the semiconductor light emitting element 4 and the output of the DC power supply circuit 2b. A switching element Q1 connected to the semiconductor element, and a parallel circuit connected to a series circuit of the inductor L1 and the semiconductor light emitting element 4, and discharging energy stored in the inductor L1 to the semiconductor light emitting element 4 when the switching element Q1 is turned off. And a regenerative diode D1 connected thereto. Further, an output capacitor C2 is connected in parallel with the semiconductor light emitting element 4. The output capacitor C2 is set to have a capacitance so that a pulsating component due to the on / off of the switching element Q1 is smoothed and a smoothed DC current flows through the semiconductor light emitting element 4.

  The switching element Q1 is driven on and off at a high frequency by the control circuit 5. The control circuit 5 includes a control integrated circuit 50 and its peripheral circuits. Here, L6562 manufactured by STMicroelectronics is used as the control integrated circuit 50. This chip (L6562) is originally a control IC for a PFC circuit (a step-up chopper circuit for power factor correction control), and includes an extra component for controlling the step-down chopper circuit, such as a multiplier circuit. Yes. On the other hand, in order to control the average value of the input current to be similar to the envelope of the input voltage, the function of controlling the peak value of the input current and the zero cross control function are provided in one chip. The function is diverted to control the step-down chopper circuit.

  FIG. 2 shows a simplified internal configuration of the control integrated circuit 50 used in this embodiment. Pin 1 (INV) is an inverting input terminal of a built-in error amplifier (error amplifier) EA, Pin 2 (COMP) is an output terminal of error amplifier EA, Pin 3 (MULT) is an input terminal of multiplier circuit 52, 4 Pin (CS) is a chopper current detection terminal, Pin 5 (ZCD) is a zero cross detection terminal, Pin 6 (GND) is a ground terminal, Pin 7 (GD) is a gate drive terminal, Pin 8 (Vcc) is Power supply terminal.

  When a control power supply voltage equal to or higher than a predetermined voltage is supplied between the power supply terminal Vcc and the ground terminal GND, the control power supply 51 generates the reference voltages Vref1 and Vref2 and enables each circuit in the integrated circuit to operate. When power is turned on by the starter 53, a start pulse is supplied to the set input terminal S of the flip-flop FF1, and the Q output of the flip-flop FF1 becomes High level. As a result, the 7th pin (gate drive terminal GD) becomes High level via the drive circuit 54.

  When the 7th pin (gate drive terminal GD) becomes High level, the gate drive voltage divided by the resistors R21 and R20 in FIG. 1 is applied between the gate and source of the switching element Q1 made of MOSFET. Since the resistor R1 is a small resistor for current detection, it hardly affects the drive voltage between the gate and the source.

  When the switching element Q1 is turned on, a current flows from the positive electrode of the capacitor C0 to the negative electrode of the capacitor C0 via the output capacitor C2, the inductor L1, the switching element Q1, and the resistor R1. At this time, the chopper current i flowing through the inductor L1 is a current that rises substantially linearly unless the inductor L1 is magnetically saturated. This current is detected by the resistor R1 and input to the fourth pin (CS) of the control integrated circuit 50.

  The fourth pin (CS) of the control integrated circuit 50 is a chopper current detection terminal, and the voltage is applied to the + input terminal of the comparator CP1 through a 40 KΩ and 5 pF noise filter inside the IC. A reference voltage is applied to the negative input terminal of the comparator CP1. This reference voltage is determined by the applied voltage V1 at the first pin (INV) and the applied voltage V3 at the third pin (MULTI).

  When the voltage at the chopper current detection terminal CS exceeds the reference voltage, the output of the comparator CP1 becomes high level, and a reset signal is input to the reset input terminal R of the flip-flop FF1. As a result, the Q output of the flip-flop FF1 becomes low level. At this time, since the drive circuit 54 operates so as to draw current from the 7th pin (gate drive terminal GD), the diode D22 of FIG. 1 is turned on, and the gate-source charge of the switching element Q1 is changed via the resistor R22. As a result, the switching element Q1 made of MOSFET is quickly turned off.

  When the switching element Q1 is turned off, the electromagnetic energy stored in the inductor L1 is released to the output capacitor C2 via the regenerative diode D1. At this time, since the voltage across the inductor L1 is clamped to the voltage Vc2 of the output capacitor C2, the current i of the inductor L1 decreases with a substantially constant slope (di / dt≈−Vc2 / L1).

  When the voltage Vc2 of the capacitor C2 is high, the current i of the inductor L1 is rapidly attenuated. When the voltage Vc2 of the capacitor C2 is low, the current i of the inductor L1 is slowly attenuated. Therefore, even if the peak value of the current flowing through the inductor L1 is constant, the time until the current i of the inductor L1 disappears changes. The required time is shorter as the voltage Vc2 of the capacitor C2 is higher and longer as the voltage Vc2 is lower.

  During the period when the current i flows through the inductor L1, a voltage corresponding to the slope of the current i of the inductor L1 is generated in the secondary winding n2 of the inductor L1. This voltage disappears when the current i of the inductor L1 finishes flowing. The timing is detected by the fifth pin (zero cross detection terminal ZCD).

  A negative input terminal of a comparator CP2 for zero cross detection is connected to the fifth pin (zero cross detection terminal ZCD) of the control integrated circuit 50. A reference voltage Vref2 for zero cross detection is applied to the + input terminal of the comparator CP2. When the voltage of the secondary winding n2 applied to the fifth pin (zero cross detection terminal ZCD) disappears, the output of the comparator CP2 becomes high level, and the set pulse is applied to the set input terminal S of the flip-flop FF1 via the OR gate. Is supplied, and the Q output of the flip-flop FF1 becomes High level. As a result, the 7th pin (gate drive terminal GD) becomes High level via the drive circuit 54. Thereafter, the same operation is repeated.

  In this way, a DC voltage obtained by stepping down the output voltage of the capacitor C0 is obtained at the output capacitor C2. This DC voltage is supplied to the semiconductor light emitting element 4 via the output connector CON2. When a light emitting diode (LED) is used as the semiconductor light emitting element 4, when the forward voltage of the LED is Vf and the number of series is n, the voltage Vc2 of the output capacitor C2 is clamped to approximately n × Vf.

  When the number n of LEDs is large, the voltage Vc2 of the output capacitor C2 is high, so that the voltage difference (Vdc−Vc2) from the voltage Vdc of the capacitor C0 is small. For this reason, the voltage shared by the inductor L1 when the switching element Q1 is on is small, and the rising speed di / dt = (Vdc−Vc2) / L1 of the current i flowing through the inductor L1 is slow. As a result, the time until the current i flowing through the inductor L1 reaches a predetermined peak value becomes longer, and the on-time of the switching element Q1 becomes longer.

  When the switching element Q1 is off, the back electromotive force generated at both ends of the inductor L1 is clamped to the voltage Vc2 (= n × Vf) of the capacitor C2. For this reason, when the number n of LEDs in series is large, the voltage applied to the inductor L1 is large when the switching element Q1 is turned off, and the decay rate di / dt = −Vc2 / L1 of the current i flowing through the inductor L1 becomes fast. As a result, the time until the current i flowing through the inductor L1 becomes zero becomes short, and the OFF time of the switching element Q1 becomes short.

  When the number n of LEDs in series is small, the on-time of the switching element Q1 becomes short and the off-time becomes long, contrary to the above description. That is, when the number n of LEDs in series is small, the voltage Vc2 of the output capacitor C2 is low, and the voltage difference (Vdc−Vc2) from the voltage Vdc of the capacitor C0 is large. For this reason, the voltage shared by the inductor L1 when the switching element Q1 is turned on is large, and the rising speed di / dt = (Vdc−Vc2) / L1 of the current i flowing through the inductor L1 is increased. As a result, the time until the current i flowing through the inductor L1 reaches a predetermined peak value is shortened, and the ON time of the switching element Q1 is shortened.

  When the switching element Q1 is off, the back electromotive force generated at both ends of the inductor L1 is clamped to the voltage Vc2 (= n × Vf) of the capacitor C2. For this reason, when the number n of LEDs in series is small, the voltage applied to the inductor L1 when the switching element Q1 is off is small, and the decay rate di / dt = −Vc2 / L1 of the current i flowing through the inductor L1 is slow. As a result, the time until the current i flowing through the inductor L1 becomes zero becomes long, and the OFF time of the switching element Q1 becomes long.

  Thus, according to the lighting device of the present embodiment, when the number n of LEDs in series increases, the ON time of the switching element Q1 automatically increases and the OFF time decreases, and when the number n of LEDs in series decreases, The on time of the switching element Q1 is automatically shortened and the off time is lengthened automatically. Accordingly, the constant current characteristic can be maintained regardless of the number n of LEDs in series.

  Although the detailed configuration of the control power supply circuit 10 is not limited, a smoothing capacitor C3 and a Zener diode ZD1 for regulating the voltage are provided here. In the simplest example, a charging current may be supplied from the positive electrode of the capacitor C0 to the positive electrode of the capacitor C3 via a high resistance. As a more efficient power supply means, a configuration in which the capacitor C3 is charged from the secondary winding n2 of the inductor L1 in a steady state may be adopted.

  Further, in this embodiment, the timing at which the current flowing through the inductor L1 becomes substantially zero is detected by detecting the voltage disappearance timing of the secondary winding n2 of the inductor L1. The specific means may be changed as long as it is a means capable of detecting the timing at which the regenerative current disappears, such as detecting an increase in the reverse voltage of the diode D1 or detecting a drop in the voltage across the switching element Q1. Absent.

  According to the configuration of the present embodiment, the average value of the chopper current hardly changes even when the load is different. Therefore, the effective value of the output current supplied to the load after the pulsating component of the chopper current is smoothed by the output capacitor C2 is substantially constant regardless of the load.

  Therefore, by intermittently stopping the high-frequency chopper operation in accordance with the low-frequency PWM signal, an output current corresponding to the duty of the PWM signal can be supplied to the semiconductor light-emitting element 4, and high-precision light control is achieved. It becomes possible.

  Therefore, in the embodiment of FIG. 1, the switching element Q2 is connected between the control electrode of the switching element Q1 and the ground, and the gate voltage V2 of the switching element Q2 is controlled according to the low-frequency PWM signal, or The applied voltage V1 of the first pin (INV), the applied voltage V3 of the third pin (MULTI), the applied voltage V4 of the fourth pin (CS), or the fifth pin (ZCD) of the control integrated circuit 50 One of the applied voltages V5 is controlled in accordance with the low frequency PWM signal. Two or more of these means may be implemented in combination, or any one of them may be selected and implemented.

Hereinafter, each means will be described individually.
First, the case where the switching element Q2 is controlled to be turned on / off according to the low frequency PWM signal will be described. The low-frequency PWM signal is, for example, a rectangular wave voltage signal of 1 kHz, and is a dimming signal such that the dimming output becomes larger as the Low level period in one cycle is longer. This type of PWM signal is widely used in the field of dimming / lighting devices for fluorescent lamps. As shown in FIG. 3, the PWM signal is supplied from the dimming signal line via the connector CON3 of the lighting device 1, and the rectifier circuit 5a. The signal is input to the control circuit 5 through the insulation circuit 5b and the waveform shaping circuit 5c.

  Here, the low-frequency PWM signal is used as the gate voltage V2 of the switching element Q2, and when the gate voltage V2 is at a high level, the switching element Q2 is turned on to short-circuit between the control electrode of the switching element Q1 and the ground. When the gate voltage V2 is at the low level, the switching element Q2 is turned off (high impedance state), and is in the same state as not being connected.

  While the switching element Q2 is on, the connection point between the resistor R21 and the switching element Q2 is always at the low level. Therefore, even if the 7th pin (gate drive terminal GD) of the control integrated circuit 50 is switched to High / Low at a high frequency, the gate drive output is consumed by the resistor R21, and the switching element Q1 is turned off. Maintained. At this time, the operation of the IC may be stopped by short-circuiting the fifth pin of the control integrated circuit 50 to the ground as in the second embodiment described later. In any case, the chopper operation is stopped.

  When the switching element Q2 is turned off, the switching element Q1 is turned on / off in response to the switching of the seventh pin (gate drive terminal GD) of the control integrated circuit 50 to High / Low at a high frequency. Chopper operation.

  Therefore, the ratio between the chopper operation period and the chopper operation stop period coincides with the ratio between the low level period and the high level period of the PWM signal. Since a constant current is supplied in the chopper operation period and the current supply is stopped in the chopper operation stop period, a current corresponding to the ratio of the Low level period to one cycle of the PWM signal is eventually supplied to the semiconductor light emitting element 4. become. Thereby, high-precision light control is possible.

  Next, a case where the applied voltage V1 of the first pin (INV) of the control integrated circuit 50 is controlled according to the low frequency PWM signal will be described. The peak value of the current flowing through the switching element Q1 is controlled so as to decrease as the applied voltage V1 of the first pin (INV) increases. Therefore, for example, when the low frequency PWM signal is at the high level, the applied voltage V1 of the first pin (INV) is set high, and when the low frequency PWM signal is at the low level, the first pin (INV) is set. The applied voltage V1 of the applied voltage V1 is set low. When the applied voltage V1 is high, the peak value of the current flowing through the switching element Q1 is controlled to be low, and when the applied voltage V1 is low, the peak value of the current flowing through the switching element Q1 is controlled to be high. Dimming can be achieved by changing.

  Next, the case where the applied voltage V3 of the third pin (MULT) of the control integrated circuit 50 is controlled according to the low frequency PWM signal will be described. The peak value of the current flowing through the switching element Q1 is controlled so as to increase as the applied voltage V3 of the third pin (MULT) increases. Therefore, for example, when the low-frequency PWM signal is at a high level, the applied voltage V3 of the third pin (MULT) is set low, and when the low-frequency PWM signal is at a low level, the pin 3 (MULTI) The applied voltage V3 is set high. When the applied voltage V3 is high, the peak value of the current flowing through the switching element Q1 is controlled to be high, and when the applied voltage V3 is low, the peak value of the current flowing through the switching element Q1 is controlled to be low. Dimming can be achieved by changing.

  Next, the case where the applied voltage V4 of the fourth pin (CS) of the control integrated circuit 50 is controlled according to the low frequency PWM signal will be described. As described above, when the voltage V4 applied to the fourth pin (CS) becomes higher than the internal reference voltage (the voltage applied to the negative input terminal of the comparator CP1), the switching element Q1 is controlled to be turned off. Therefore, for example, when the low-frequency PWM signal is at a high level, control is performed so that the applied voltage V4 of the fourth pin (CS) reaches the reference voltage earlier. As a concrete means, the detection voltage of the 4th pin (CS) is increased in a pseudo manner by superimposing a direct current on the resistor R1 via the diode D4, and the applied voltage V4 of the 4th pin (CS). Is controlled so as to reach the reference voltage earlier. In this case, the peak value of the current flowing through the switching element Q1 is lower than usual. Further, when the low-frequency PWM signal is at the low level, the normal operation is restored by removing the superimposed current through the diode D4. With this control, when the low frequency PWM signal is at a high level, the peak value of the current flowing through the switching element Q1 is lower than usual, and when the low frequency PWM signal is at a low level, the switching element Q1. The peak value of the current flowing through the current returns to the normal value. Thus, dimming can be performed according to the ratio of the low-frequency PWM signal during the low level to the high level.

  As an extreme case, when the low frequency PWM signal is at a high level, the applied voltage V4 of the 4th pin (CS) is always higher than the reference voltage (the applied voltage of the input terminal of the comparator CP1). You may control to. In that case, since the ON time of the switching element Q1 becomes substantially zero, the chopper operation is substantially stopped.

  Next, a case where the applied voltage V5 of the fifth pin (ZCD) of the control integrated circuit 50 is controlled according to the low frequency PWM signal will be described. As described above, when the applied voltage V5 of the fifth pin (ZCD) falls, the switching element Q1 is turned on again. Therefore, for example, when the low-frequency PWM signal is at a high level, control is performed so that the applied voltage V5 of the fifth pin (ZCD) does not fall. As a specific means, a DC voltage is superimposed via a diode D5. When the low-frequency PWM signal is at the low level, the superimposed voltage via the diode D5 is removed to restore the normal operation. With this control, when the low frequency PWM signal is at a high level, the switching element Q1 is not turned on. When the low frequency PWM signal is at a low level, the switching element Q1 is on / off controlled. Thus, dimming can be performed according to the ratio of the low-frequency PWM signal during the low level to the high level.

  As another means, as described in the second embodiment (FIG. 4) to be described later, the fifth pin (ZCD) of the control integrated circuit 50 is short-circuited to the ground in synchronization with the PWM signal. You may control to stop operation | movement. As described above, when L6562 manufactured by ST Microelectronics is used as the control integrated circuit 50, the disable circuit 55 is connected to the fifth pin (ZCD) as the zero cross detection terminal as shown in FIG. The operation of the IC can be stopped by short-circuiting the fifth pin to the ground. Therefore, when the low frequency PWM signal is at the high level, the fifth pin (ZCD) is short-circuited to the ground to stop the operation of the IC, and when the low frequency PWM signal is at the low level, the fifth pin (ZCD) is stopped. ) To return to normal operation. Thus, dimming can be performed according to the ratio of the low-frequency PWM signal during the low level to the high level.

  The whole structure of the LED dimming / lighting device 1 incorporating the lighting device of FIG. 1 is shown in FIG. The power supply circuit 2 includes the above-described filter circuit 2a and DC power supply circuit 2b. Capacitors Cc and Cd are capacitors for connecting the circuit ground (the negative electrode of the capacitor C0) to the appliance chassis at a high frequency. CON1 is a power supply connector connected to the commercial AC power supply Vs, CON2 is an output connector connected to the semiconductor light emitting element 4 via the lead wire 44, and CON3 is a connector for connecting a dimming signal line. The dimming signal line is supplied with a dimming signal composed of a rectangular wave voltage signal having a variable duty with a frequency of 1 kHz and an amplitude of 10 V, for example.

  The rectifier circuit 5a connected to the connector CON3 is a circuit for making the wiring of the dimming signal line non-polar, and operates normally even if the dimming signal line is reversely connected. That is, the input dimming signal is full-wave rectified by the full-wave rectifier DB1, and a rectangular wave voltage signal is obtained at both ends of the Zener diode ZD via the impedance element Z1 such as a resistor. The insulating circuit 5b includes a photocoupler PC1 and transmits a rectangular wave voltage signal while insulating the dimming signal line from the lighting device. The waveform shaping circuit 5c is a circuit that shapes the signal output from the photocoupler PC1 of the insulating circuit 5b and outputs it as a clear PWM signal having a high level and a low level. Since the waveform of the rectangular wave voltage signal transmitted over a long distance via the dimming signal line is distorted, a waveform shaping circuit 5c is provided.

  In the conventional inverter-type fluorescent lamp dimming / lighting device, a low-pass filter circuit such as a CR integration circuit (smoothing circuit) is further provided after the waveform shaping circuit 5c to generate an analog dimming voltage. In this embodiment, the PWM signal after waveform shaping is directly input to the control circuit 5 (see FIG. 1). The control circuit 5 stops the chopper operation of the step-down chopper circuit 3 while the PWM signal is at a high level, and permits the chopper operation of the step-down chopper circuit 3 while the PWM signal is at a low level. The magnitude of the direct current supplied to the semiconductor light emitting element 4 after being smoothed by the output capacitor C2 is adjusted. That is, the control circuit 5 and the step-down chopper circuit 3 function as a low-pass filter circuit that smoothes the PWM signal.

  When the dimming signal line is disconnected or the connector CON3 is disconnected, the PWM signal is always at the low level. In this case, the chopper operation of the step-down chopper circuit 3 is always permitted, so that the semiconductor light emission The element 4 is fully lit.

(Embodiment 2)
FIG. 4 is a circuit diagram of the lighting device according to the second embodiment of the present invention. In the present embodiment, when the PWM signal is at the high level in the basic configuration of FIG. 1, the gate voltage V2 of the switching element Q2 is set to the high level, and the applied voltage V1 at the first pin and the applied voltage V5 at the fifth pin. Are controlled to be at a low level (short-circuited to the ground).

  As described above, when L6562 manufactured by ST Microelectronics is used as the control integrated circuit 50, the disable circuit 55 is connected to the fifth pin (ZCD) as the zero cross detection terminal as shown in FIG. The operation of the IC can be stopped by short-circuiting the fifth pin to the ground. As described in the first embodiment, the switching element Q1 can be maintained in the off state only by turning on the switching element Q2. However, the current consumption can be reduced by further stopping the operation of the IC.

  In the following, the circuit configuration of FIG. 4 will be described, but the same reference numerals are given to the portions having the same functions as those of the circuit configuration of FIG.

  The AC input terminal of the full-wave rectifier DB is connected to the power connector CON1 through the filter circuit 2a and the current fuse FUSE. The configuration of the filter circuit 2a is the same as that in FIG.

  The DC output terminal of the full-wave rectifier DB is connected to the smoothing capacitor C0 via the positive temperature coefficient thermistor PTC. The positive characteristic thermistor PTC is a thermistor whose resistance value increases as the temperature increases.

  The smoothing capacitor C0 has a capacity of about several tens of μF, for example. A capacitor Co connected in parallel to the smoothing capacitor C0 is a small-capacitance capacitor for high-frequency bypass.

  Immediately after the power is turned on, the DC output terminal of the full-wave rectifier DB is short-circuited by the smoothing capacitor C0 before charging, so-called inrush current flows. This inrush current is limited by the positive temperature coefficient thermistor PTC.

  On the other hand, after the smoothing capacitor C0 is charged, current limitation by the positive temperature coefficient thermistor PTC is not necessary. In addition, useless power consumption occurs.

  Therefore, a reverse blocking three-terminal thyristor (SCR) Q14 is connected in parallel with the positive temperature coefficient thermistor PTC, and the thyristor Q14 is controlled to turn on when charging of the smoothing capacitor C0 is completed after the power is turned on.

  In order to generate the gate voltage of the thyristor Q14, the anodes of the diodes D11 and D12 are connected to the AC input terminals of the full-wave rectifier DB, and the cathodes of the diodes D11 and D12 are connected to the resistors R54, R55, R56, and R57. Is connected to the negative electrode of the DC output terminal of the full-wave rectifier DB.

  An electrolytic capacitor C57 for timer is connected in parallel to the resistor R57. The capacity of the electrolytic capacitor C57 defines a delay time from when the power is turned on until the thyristor Q14 is turned on.

  When the voltage of the electrolytic capacitor C57 rises, the gate voltage is supplied to the thyristor Q14 via the parallel circuit of the diodes D13 and D14 and the resistor R58. The capacitor C58 connected in parallel between the gate and cathode of the thyristor Q14 is for preventing malfunction.

  The circuit including the resistors R54 to R58, the capacitors C57 and C58, the diodes D11 to D14, the thyristor Q14, and the positive characteristic thermistor PTC constitutes an inrush current prevention circuit 2c.

  The diodes D11 and D12 of the inrush current prevention circuit 2c are also used as a rectifier of the power interruption detection circuit 2d. The power interruption detection circuit 2d includes a series circuit of resistors R51, R52, and R53, a capacitor C53 connected in parallel to the resistor R53, and a transistor Q13 that is forward-biased by the voltage of the capacitor C53. A series circuit of resistors R51 to R53 is connected between the cathodes of the diodes D11 and D12 and the negative electrode of the smoothing capacitor C0. If the AC power supply is energized, a current flows through the series circuit of the resistors R51 to R53, the capacitor C53 is charged, and the transistor Q13 is turned on. When the AC power supply is cut off, the current through the series circuit of the resistors R51 to R53 is cut off immediately. Then, the electric charge of the capacitor C53 is discharged through the resistor R53, and the forward bias of the transistor Q13 is lost, so that the transistor Q13 is turned off.

  Note that the capacitance of the capacitor C53 is set so that the transistor Q13 is turned off when the power-off state continues for several cycles of the AC power supply, and the AC power supply is momentarily interrupted or phase-controlled. In this case, or near the zero cross of the AC power supply, the transistor Q13 does not turn off.

  On the other hand, the current flowing from the smoothing capacitor C0 through the series circuit of the resistors R15 to R17 is not cut off while the charge of the smoothing capacitor C0 remains even if the AC power supply is cut off. In the present embodiment, when the transistor Q13 of the power interruption detection circuit 2d is turned off, the transistor Q11 is forward-biased by the current supplied to the resistor R14 through the series circuit of the resistors R15 to R17, and the control integrated circuit 50 The first pin (INV) is short-circuited to the ground. At this time, the fifth pin (ZCD) is also short-circuited to the ground via the diode D8. This prevents the light output from flickering when the power is turned off.

  The voltage of the smoothing capacitor C0 is a voltage (about 140V) near the peak value of the commercial AC power supply voltage (100V, 50/60 Hz). A charging current is supplied from the smoothing capacitor C0 to the capacitor C3 for supplying the control power supply voltage Vcc via the step-down resistors R31 to R34.

  When the voltage of the capacitor C3 rises above the operable voltage of the control integrated circuit 50, the on / off operation of the switching element Q1 is started, and a high-frequency triangular wave current flows through the inductor L1. Wave voltage is generated. A current flows through the diode D10, the capacitor C10, and the resistor R10 by the voltage generated in the secondary winding of the inductor L1 when the switching element Q1 is turned on, and the capacitor C10 is charged. When the switching element Q1 is turned off, a voltage having a reverse polarity is generated in the secondary winding of the inductor L1, so that a charging current is supplied to the capacitor C3 via the diode D3 and the resistor R10 by adding this voltage and the charging voltage of the capacitor C10. Flows. As a result, the voltage of the capacitor C3 tries to rise further, but since the Zener diode ZD1 is connected in parallel, it is clamped by the Zener voltage to generate a constant control power supply voltage Vcc.

  Note that the voltage of the capacitor C3 for supplying the control power supply voltage Vcc is about ten and several volts. A capacitor C11 connected in parallel with the capacitor C3 is a small-capacitance capacitor for bypassing the high-frequency component of the charging current via the diode D3.

  The control power supply voltage Vcc is divided by resistors R11, R12, and R13 and applied to the first pin (INV) of the control integrated circuit 50. As described above, this voltage is used to define the peak value of the current flowing through the switching element Q1.

  In the present embodiment, the second pin (COMP) and the third pin (MULT) of the control integrated circuit 50 are short-circuited. The detection voltage of the current detection resistor R1 is input to the fourth pin (CS) via a series circuit of resistors R41 and R42. A variable resistor VR1 for adjusting current detection sensitivity is connected between the connection point of the resistors R41 and R42 and the ground. When the resistance value of the variable resistor VR1 is lowered, the detection voltage of the current detection resistor R1 is divided by the resistor R41 and the variable resistor VR1 and input to the fourth pin (CS), so that the current detection sensitivity can be lowered. The peak value of the current flowing through the switching element Q1 can be increased.

  A DC voltage is superimposed on the variable resistor VR1 from the seventh pin (GD) that supplies the gate drive voltage of the switching element Q1 via the diode D7, the resistor R43, and the variable resistor VR2. When the resistance value of the variable resistor VR2 is lowered, the superimposed DC voltage increases, so that the voltage at the 4th pin (CS) increases and the peak value of the current flowing through the switching element Q1 can be lowered.

  By adjusting these two variable resistors VR1 and VR2, the peak value of the current flowing through the switching element Q1 can be set appropriately. Here, in terms of the upper limit value, it is appropriate that the inductor L1 is not magnetically saturated and that the maximum peak current of the switching element Q1 is not exceeded. It is appropriate that the operating frequency of the switching element Q1 is not too high.

  The fifth pin (ZCD) is connected to a low-pass filter circuit composed of a resistor R5 and a capacitor C5. Further, it is connected to the first pin (INV) via the diode D8, and when the first pin (INV) is short-circuited to the ground by the transistor Q11 or Q12, the potential of the fifth pin (ZCD) is also set to the ground potential. I try to drop it.

  As described above, the transistor Q11 is turned on when the power supply is detected. The transistor Q12 connected in parallel with the transistor Q11 is turned on when the PWM signal is at a high level.

  In this embodiment, the PWM signal is a 1 kHz rectangular wave voltage signal. When the PWM signal is at a high level, a current flows through the diode D9 and the resistors R24 and R23, the voltage across the resistor R23 rises, and the switching is performed by the MOSFET. When the gate voltage of the element Q2 exceeds the threshold voltage, the switching element Q2 is turned on. Also, current flows through the diode D9 and the resistors R25 and R26, the voltage across the resistor R26 increases, and the transistor Q12 is turned on. When the PWM signal is at a low level, both the switching element Q2 and the transistor Q12 are turned off.

  Next, the no-load detection circuit 6 will be described. The no-load detection circuit 6 includes a Zener diode ZD6, resistors R61 to R64, and transistors Q61 and Q62. As shown in FIG. 3, the semiconductor light emitting element 4 is connected to the output connector CON2 via a lead wire 44. The semiconductor light emitting element 4 includes a series circuit of a plurality of LEDs 4a to 4d. Assuming that the number in series is n and the forward voltage is Vf, the voltage of the output connector CON2 when the load is connected is generally clamped to n × Vf. The Zener voltage of the Zener diode ZD6 is set higher than this n × Vf with a little margin.

  If the lead wire 44 is disconnected or disconnected, or if any one of the plurality of LEDs is disconnected or if a contact failure occurs in the output connector CON2, the voltage of the output connector CON2 is clamped to n × Vf. It will not be done. When the on / off operation of the switching element Q1 continues in this state, the voltage of the capacitor C2 increases. Eventually, when the voltage of the capacitor C2 exceeds the zener voltage of the zener diode ZD6, current flows through the resistors R61 and R62, the transistor Q61 is turned on, current flows through the resistors R63 and R64, and the transistor Q62 is turned on. Turn on. As a result, a current flows through the transistor Q62 and the resistor R65, a voltage is generated across the resistor R65, and the voltage at the first pin (INV) of the control integrated circuit 50 increases through the diode D6. Thereby, since the current peak value of the switching element Q1 is controlled to be low, the voltage rise of the capacitor C2 is suppressed.

(Embodiment 3)
In each of the above-described embodiments, the circuit example in which the switching element Q1 of the step-down chopper circuit 3 is disposed on the low potential side has been described. However, as illustrated in FIG. 5A, the switching element Q1 of the step-down chopper circuit 3a includes Needless to say, the present invention can be applied even in the case of being arranged on the high potential side.

  In each of the above-described embodiments, the separately-excited control circuit using the control integrated circuit 50 shown in FIG. 2 is exemplified as the control circuit. However, the scope of application of the present invention is not limited to this, and The present invention is also applicable to an LED lighting device using an excitation type control circuit.

  For example, in the configuration in which the switching element Q1 arranged on the high potential side is turned on / off by a self-excited control circuit as in the conventional example of FIG. 7, the diode D2 for energizing the regenerative current is an n-channel MOSFET (FIG. If a low-frequency PWM signal is supplied to the gate electrode instead of the switching element Q2 of FIG. 1, the same state as the regenerative current flows (or the resistor R1) while the n-channel MOSFET is conducting. Therefore, the dimming control according to the present invention is possible. Further, during normal oscillation operation, the reverse diode between the drain and source of the n-channel MOSFET can also be used as the diode D2 for energizing the regenerative current.

  In the conventional example of FIG. 7, when the DC power supply 2 is turned on, the switching element Q1 is turned on by the starting resistor R3. When the switching element Q1 is turned on, current flows from the DC power source 2 through the path of the switching element Q1, the current detection resistor R1, the inductor L1, and the capacitor C2, and electromagnetic energy is accumulated in the inductor L1. At this time, the feedback voltage from the secondary winding n2 of the inductor L1 is supplied to the control terminal of the switching element Q1 through the parallel circuit of the resistor R2 and the capacitor C1, and the switching element Q1 continues to be on.

  When the current flowing through the switching element Q1 reaches a predetermined value, the transistor Tr1 is turned on by the current detection resistor R1, and the switching element Q1 is turned off. Then, an electromotive voltage in the reverse direction is generated in the secondary winding n2 of the inductor L1, and a current flows through the parallel circuit of the resistor R2 and the capacitor C1 via the diode D2, thereby controlling the switching element Q1. The charge accumulated in the terminal is pulled out, and the charge in the capacitor C1 is discharged.

  Further, the current due to the electromagnetic energy accumulated in the inductor L1 flows in the closed circuit of the inductor L1, the capacitor C2, and the diode D1, so that the capacitor C2 is charged and the electromagnetic energy of the inductor L1 is consumed. When the electromagnetic energy of the inductor L1 is consumed, the diode D2 is turned off, and the switching element Q1 is turned on again by the starting resistor R3. Thereafter, the same operation is repeated, and when the charging voltage of the capacitor C2 rises to the above-mentioned n × Vf, the light emitting diode 4 is turned on.

  In the conventional example shown in FIG. 7, if the diode D2 for energizing the regenerative current is replaced with an n-channel MOSFET and a low-frequency PWM signal is supplied to the gate electrode, the above self-oscillation operation is performed with the High / Low PWM signal. Accordingly, the oscillation can be stopped / restarted, and the light can be adjusted according to the off-duty of the PWM signal. Further, the diode D2 for energizing the regenerative current can also be used as an n-channel MOSFET as a PWM switch, and the number of parts can be reduced.

  Note that the present invention can also be applied to various switching power supply circuits as shown in FIGS. FIG. 5B shows an example of a step-up chopper circuit 3b, FIG. 5C shows an example of a flyback converter circuit 3c, and FIG. 5D shows an example of a step-up / step-down chopper circuit 3d. These are examples, and a peak current detection operation for controlling the switching element Q1 to be turned off when the current flowing through the inductance element (inductor L1 or transformer T1) reaches a predetermined value when the switching element Q1 is turned on, and an inductance when the switching element Q1 is turned off. The present invention can be applied to any switching power supply circuit that uses a zero-cross detection operation for controlling the switching element Q1 to be turned on when the current discharged from the element through the regenerative diode D1 becomes substantially zero.

(Embodiment 4)
In the above embodiment, the case where a 1 kHz rectangular wave voltage signal is used as the PWM signal is illustrated, but the present invention is not limited to this. For example, a voltage signal whose waveform has been shaped after full-wave rectification of the phase-controlled AC voltage may be used as the low-frequency PWM signal.

  The frequency of the PWM signal is preferably set in the range of 100 Hz to 2 kHz. When the frequency of the PWM signal is lower than 100 Hz, flickering of the optical output is perceived by human eyes. On the contrary, when the frequency of the PWM signal is higher than 2 kHz, when the dimming is deepened, the length of the oscillation period during which the switching element Q1 is turned on / off in one cycle of the PWM signal is shortened. It becomes impossible to finely control the number of on pulses of the switching element Q1 included in the above, and the number of on pulses varies discretely, so that the light control resolution is lowered.

  When the frequency of the PWM signal is an audible frequency (especially an unpleasant frequency such as 1 kHz), a high noise noise is generated from the inductor L1 for chopper. It is good to take measures.

  By the way, in each above-mentioned embodiment, since the output capacitor C2 is provided in parallel with the semiconductor light emitting element 4, even if the high frequency switching operation of the switching element Q1 is intermittently stopped at a low frequency according to the PWM signal, the semiconductor The direct current flowing through the light emitting element 4 is a smoothed current with less low frequency ripple. That is, the semiconductor light emitting element 4 is continuously lit by a smooth DC current corresponding to the off-duty (ratio of the low level period in one cycle) of the PWM signal. However, the output capacitor C2 is not essential, and when the output capacitor C2 is eliminated or the capacity is designed to be relatively small, the semiconductor light emitting element 4 is intermittently lit at a low frequency (that is, at a high speed that is not visible). (Flashing lighting).

  In this way, when a lighting device whose light output fluctuates at a low frequency is used for indoor lighting or outdoor night lighting, when a monitoring video camera is installed within the illumination range, the PWM signal is a shutter of the video camera. It is preferable to set an integral multiple of the reciprocal of the speed. For example, if the shutter speed of the video camera is 1/60 seconds, the frequency of the PWM signal is set to one of 60 Hz, 120 Hz, 180 Hz, 240 Hz, 300 Hz,. If the shutter speed of the video camera is 1/100 second, the frequency of the PWM signal is set to any one of 100 Hz, 200 Hz, 300 Hz, 400 Hz,. Furthermore, if the shutter speed of the video camera is 1/120 seconds, the frequency of the PWM signal is set to any one of 120 Hz, 240 Hz, 360 Hz, 480 Hz,. If the shutter speed of the video camera is 1/180 seconds, the frequency of the PWM signal is set to any one of 180 Hz, 360 Hz, 540 Hz, 720 Hz,. Furthermore, if the shutter speed of the video camera is 1/240 seconds, the frequency of the PWM signal is set to 240 Hz, 480 Hz, 720 Hz, 960 Hz,. With this setting, even if the optical output fluctuates at a low frequency, it is possible to prevent the video camera image from flickering.

  When the lighting device of the present invention is a light source device attached to a video camera, the PWM signal is preferably switched in synchronization with the electronic shutter of the video camera. For example, if a synchronization signal of a video camera is input to a lighting device and the semiconductor light emitting element is turned on only during the exposure period in synchronization with the timing of the electronic shutter of the video camera, it is not necessary to consume unnecessary lighting power. It becomes power saving. As is well known, a CCD type video camera has a charge accumulation period and a charge transfer period. In the charge accumulation period, the photocurrent of a photodiode serving as a pixel is accumulated as a charge for each pixel. Since the photocurrent of the photodiode which is a pixel is not accumulated during this period, the illumination may be turned off during this period. Thereby, when the video camera with LED illumination is battery-driven, the battery life can be extended.

(Embodiment 5)
FIG. 6 shows a schematic configuration of a separate power source type LED lighting apparatus using the LED lighting device of the present invention. In the power remote installations LED lighting apparatus has a built-in dimmer lighting device 1 of the power supply unit to another case the instrument housing 42 of the LED module 40. By doing so, the LED module 40 can be thinned, and the dimming / lighting device 1 as a separate power supply unit can be installed regardless of the location.

  The instrument housing 42 is made of a metal cylinder that is open at the lower end, and the lower end open portion is covered with a light diffusion plate 43. The LED module 40 is disposed so as to face the light diffusion plate 43. Reference numeral 41 denotes an LED mounting board on which the LEDs 4a to 4d of the LED module 40 are mounted. The appliance housing 42 is embedded in the ceiling 100, and is wired from the dimming / lighting device 1 as a power supply unit arranged behind the ceiling via a lead wire 44 and a connector 45.

  A circuit as shown in FIG. 3 is accommodated in the dimming / lighting device 1 as a power supply unit. A series circuit (LED module 40) of the LEDs 4a to 4d corresponds to the semiconductor light emitting element 4 described above.

  In the present embodiment, the dimming / lighting device 1 as a power supply unit is exemplified as a separate power supply type LED lighting device housed in a housing different from the LED module 40, but the power supply unit is mounted in the same housing as the LED module 40. You may use the lighting device of this invention for the stored power supply integrated LED lighting fixture.

  The lighting device of the present invention is not limited to a lighting fixture, and may be used as various light sources, for example, a backlight of a liquid crystal display, a light source of a copying machine, a scanner, a projector, or the like.

  In the description of each embodiment described above, a light emitting diode is exemplified as the semiconductor light emitting element 4, but the present invention is not limited to this, and may be, for example, an organic EL element or a semiconductor laser element.

Q1 Switching element L1 Inductor D1 Diode R1 Current detection resistor 4 Semiconductor light emitting element 5 Control circuit

Claims (7)

A switching element connected in series to a DC power source and controlled to be turned on and off at a high frequency; an inductance element connected in series with the switching element and through which a current flows from the DC power source when the switching element is turned on; and when the switching element is turned on releasing the energy stored in the inductance element to the semiconductor light emitting element during off of the switching element regenerative diode and; current detecting a current flowing through the switching element detecting means and; detected electric current values by the current detecting means In a lighting device for a semiconductor light emitting device comprising: a control means for turning off the switching element and turning on the switching element when energy emission of the inductance element is completed when the value reaches a predetermined value;
Before SL current value detected by the current detecting means is intermittently formed to be dim the semiconductor light emitting element by Rukoto at a low frequency state has reached a predetermined value,
By reducing the predetermined value below intermittently zero at low frequencies, the current value detected by said current detecting means is characterized that you intermittently formed in a low frequency state has reached a predetermined value Lighting device for semiconductor light emitting element. A switching element connected in series to a DC power source and controlled to be turned on and off at a high frequency; an inductance element connected in series with the switching element and through which a current flows from the DC power source when the switching element is turned on; and when the switching element is turned on A regenerative diode that discharges energy stored in the inductance element to the semiconductor light emitting element when the switching element is turned off; current detection means that detects a current flowing through the switching element; and a current value detected by the current detection means is predetermined. When the value is reached, in a lighting device for a semiconductor light emitting device comprising a control means for turning off the switching element and turning on the switching element when energy emission of the inductance element is completed,
Dimming the semiconductor light emitting element by intermittently forming a state where the current value detected by the current detection means has reached a predetermined value at a low frequency ,
By intermittently superimposing a current value larger than the predetermined value at a low frequency on the current value detected by the current detection means, the state where the current value detected by the current detection means reaches a predetermined value is reduced. lighting device of the semiconductor light emitting element characterized that you intermittently formed in frequency. Oite to claim 1 or 2, before Symbol lighting device of the semiconductor light-emitting element at the same time carried to said Rukoto the operation of an operation of detecting the completion of the energy released intermittently blocked by the low-frequency inductance element. Oite to any one of claims 1 to 3, before Symbol lighting device of the semiconductor light emitting device characterized by in synchronism with the low-frequency short the control electrode of the switching element. In any one of claims 1-4, before Symbol frequency of the low frequency is at 100Hz above 2kHz or less, the lighting device of the semiconductor light emitting device characterized by switching in synchronization with the electronic shutter of the video camera. In any one of claims 1-4, wherein the frequency of the low-frequency lighting device of the semiconductor light emitting element characterized that you set to an integral multiple of the reciprocal of the shutter speed of the video camera. Luminaires and lighting equipment of semiconductors light emitting device according to any one of claims 1 to 6, a current from the lighting device includes a semiconductor light emitting element to be supplied.

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