JP4533582B2 - A CMOS compatible 3D image sensing system using quantum efficiency modulation - Google Patents

Abstract

A preferably CMOS-implementable method and system measures distance and/or brightness by illuminating a target with emitted optical energy having a modulated periodic waveform whose high frequency component may be idealized as S>1< = cos(W.t). A fraction of the emitted optical energy is reflected by a target and detected with at least one in a plurality of semiconductor photodetectors. Photodetector quantum efficiency is modulated to process detected signals to yield data proportional to the distance z separating the target and photodetector. Detection includes measuring phase change between the emitted optical energy and the reflected fraction thereof. Quantum efficiency can be modulated with fixed or variable phase methods and may be enhanced using enhanced photocharge collection, differential modulation, and spatial and temporal multiplexing. System power requirements may be reduced with inductors that resonate with photodetector capacitance at the operating frequency. The system includes on-chip photodetectors, associated electronics, and processing.

Description

Detailed Description of the Invention

(Relationship with prior application)
Claiming priority from applicant's co-pending US provisional patent application serial number 60 / 254,873, filed December 11, 2000, "CMOS 3D multi-pixel sensor using photodiode quantum efficiency modulation" Yes. The applicant refers to the application and incorporates it here. Applicant also refers to co-pending US utility model application serial number 09 / 876,373, filed June 6, 2001, “CMOS-compatible 3D image sensing using reduced peak energy”. , Take in here.

  The present invention relates generally to a rangefinder type image sensor, and more particularly to an image sensor that can be implemented on a single integrated circuit using CMOS fabrication. More particularly, it relates to reducing power consumption of a system using such a sensor.

  Electronic circuits that measure the distance from a circuit to an object are known in the art and can be illustrated in the system 10 of FIG. The generalized system of FIG. 1 uses the imaging circuitry within the system 10 and uses an approximate distance to the object 20 (eg, Z1, Z2) whose uppermost portion is shown farther than its lowermost portion. , Z3). In general, the system 10 includes a light source 30 that collects the light output of the light source 30 by a lens 40 and directs it to the object being imaged, here the object 20. Other prior art systems do not provide a light source 30, but instead rely on ambient light reflected by the object and actually require ambient light.

  The surface portion of the object 20 reflects light from the light source 30 at various ratios, and the lens 50 collects this light. This return light is incident on various detector elements 60 such as photodiodes in an array on an integrated circuit (IC) 70, for example. Element 60 provides an interpretation of the brightness of an object (eg, 10) in this scene and estimates distance data therefrom. Depending on the application, device 60 may be charge coupled (CCD) or an array of CMOS devices.

  In general, the CCD is set in the form of a so-called bucket relay in which the charge detected by the first CCD is coupled in series to the adjacent CCD and the output is then coupled to the third CCD. This bucket relay setting normally prevents the fabrication of the processing circuit configuration on the same IC including the CCD array. In addition, CCDs provide serial readout rather than random readout. For example, when a CCD rangefinder system is utilized in a digital zoom lens application, an array is needed to access the relevant data, even though most of the relevant data is provided by only a few CCDs in the array. The whole needs to be read and the process is time consuming. For still images and some moving image applications, CCD based systems may still be practical.

  As described, the upper part of the object 20 is intentionally shown farther than the lower part. That is, the distance Z3> Z2> Z1. In a rangefinder autofocus camera environment, an approximate average distance from the camera (eg, from Z = 0) to the object 10 is obtained from the element 60 by examining the relative brightness data obtained from the object. Depending on the application, for example in range finding binoculars, the field of view is very narrow and all the objects in focus are at substantially the same distance. But usually, a lightness-based system doesn't work well. For example, in FIG. 1, the top of the object 20 is shown darker than its lower part, possibly farther away from its lower part. However, in the real world, the farther away parts of an object may be more glossy or shining (ie, reflecting more light energy) than closer but darker parts. In complex scenes, it is very difficult to approximate the focal length to an object or subject standing in the background using changes in brightness to distinguish the subject from the background. In such various applications, circuit 80, circuit 90, and circuit 100 in system 10 of FIG. 1 assist in this signal processing. As described, when IC 70 includes CCD 60, other circuits such as circuit 80, circuit 90, and circuit 100 are fabricated off-chip.

  Since the reflectance of the object is unknown, the reflected brightness data unfortunately does not provide a truly accurate interpretation of distance. Thus, the shiny surface of a remote object surface may reflect as much (possibly more) light than the matte surface of a nearby object.

  Other focus systems are known in the art. Infrared (IR) autofocus systems used in cameras and binoculars produce a single distance that is the average or shortest distance to all targets in the field of view. In other camera autofocus systems, it is often necessary to mechanically focus the lens on the subject in order to measure distance. These prior art focusing systems cannot at the same time measure the distances of all objects in the field of view, at best, with the lens set to one object in the field of view.

  In general, by reproducing and approximating the original brightness in a scene, the human visual system can understand what objects were present in the scene and estimate their relative positions binocularly become. For a binocular image as projected on an ordinary television screen, the human brain uses past experience to determine the apparent size, distance and shape of the object. A specialized computer program can also estimate the distance of an object under special conditions.

  The binocular image allows the observer to more accurately determine the distance of the object. However, it is difficult for a computer program to determine the distance of an object from a binocular image. Errors often occur and essential signal processing requires specialized hardware and computation. Binocular images are at best an indirect way of creating a three-dimensional image suitable for direct computer use.

  In many applications it is necessary to obtain a three-dimensional interpretation of the scene directly. However, in practice, it is difficult to accurately extract distance and velocity data along the visual axis from the brightness measurement. Nevertheless, accurate tracking of distance and speed is required in many applications. For example, an assembly line welding robot must measure the exact distance and speed of objects to be welded. The required distance measurement may be incorrect due to changing lighting conditions and other drawbacks mentioned above. Systems that can directly capture 3D images can benefit such applications.

  Although specialized three-dimensional imaging systems exist in the field of nuclear magnetic resonance and scanning lasers, such systems require considerable equipment costs. In addition, these systems are terribly disturbing and are dedicated to specific tasks, such as projecting internal organs.

  In other applications, a scanning laser range finding system raster scans an image using a mirror that refracts the laser beam also in the x-axis plane and possibly the y-axis plane. The angle of curvature of each mirror is used to determine the coordinates of the image pixel to be sampled. In such a system, it is necessary to accurately detect the angle of each mirror in order to know which pixel is being viewed. Of course, the need to provide precision moving mechanical parts makes such a range finding system larger, more complex and costly. Furthermore, since such a system samples each pixel in series, the number of complete image frames that can be sampled per unit time is limited. The term “pixel” shall refer to an output result produced by one or more detectors in an array of detectors.

In summary, there is a need for a system that can perform direct three-dimensional imaging, preferably using CMOS fabrication technology on a single IC, requiring fewer individual parts and requiring no moving parts. Optionally, the system can output data from the detector non-serially or randomly. More preferably, such a system can also use an inexpensive illuminant that requires a relatively low peak light output, but provides high sensitivity.
The present invention provides such a system.

Summary of invention

  The present invention provides a system for measuring distance and speed data in real time using time-of-flight (TOF) data rather than relying on brightness data. The system is CMOS compatible and provides such 3D imaging without the need for moving parts. The system can be fabricated on a single IC that includes both a CMOS compatible pixel detector that detects photon light energy and associated processing circuitry.

  In Applicant's US Pat. No. 6,323,942B1 (2001), “CMOS-compatible three-dimensional image sensor IC”, a microprocessor on a CMOS-compatible IC preferably triggers an LED or laser light source continuously, A point on the surface of the object to be imaged at least partially reflects the light output pulse. To obtain a good image resolution, eg cm, a large but short pulse of light energy is required. For example, a peak pulse energy of 10 W, a pulse width of about 15 ns, and a repetition frequency of about 3 KHZ. The average energy of Applicant's previous system is only 1 mW, whereas the requirement of 10 W peak power essentially dictates the use of a relatively expensive laser diode as the preferred energy source. Each pixel detector of the detector array has associated electronics that measure the time of flight from the transmission of the light energy pulse to the detection of the return signal. The invention requires a high bandwidth pixel detector amplifier for the transmission of high peak power narrow energy pulses.

  The referenced Applicant's co-pending US patent application discloses a system that transmits a periodic signal with a small average power and a high frequency component with a small peak power, such as several tens of milliwatts rather than several watts. In order to simplify the analysis, a periodic signal of light energy having an ideal sinusoidal waveform such as cos (ω · t) is assumed, but this is also assumed here. By emitting a periodic signal having such a low peak power and a high frequency component, an inexpensive light source and a simple narrow bandwidth pixel detector can be used. For an operating (radiant energy modulation) frequency of about 200 MHz, the bandwidth may be on the order of several hundred KHz. Good resolution accuracy can also be obtained using low peak power light emitters, which have a higher effective repetition rate than the output from narrow pulse high peak power light emitters.

In such a system, and in the present invention, the energy emitted from the light source is approximately S ₁ = K · cos (ω · t), K is the amplitude coefficient, ω = 2πf, and the frequency f is probably 200 MHZ. Suppose that Further, assume that the distance separating the light energy emitter and the target object is z. In order to express mathematically easily, K = 1, but a coefficient of 1 or less or 1 or more may be used. The term “approximately” is used with the recognition that it is difficult to generate a complete sinusoidal waveform. Because of the time of flight required for the energy to traverse the distance z, the phase S ₂ = A · cos (ω · t + φ) between the transmitted energy and the energy detected by the photodetectors in the array There is a shift φ. The coefficient A represents the brightness of the detected reflected signal and can be measured separately using the same feedback signal received by the pixel detector.

The phase shift φ by time of flight is:
Φ = 2 · ω · z / C = 2 · (2 · π · f) · z / C
It becomes. Here, C is the speed of light of 300,000 Km / sec. Thus, the distance z from the energy emitter (and from the detector array) is determined as follows:
z = Φ · C / 2 · ω = ΦC / {2 · (2 · π · f)}

The distance z is obtained modulo 2πC / (2 · ω) = C / (2 · f). If necessary, f ₁ , f ₂ , f ₃ ,. . . . Z can be measured using C / (2 · f ₁ ), C / (2 · f ₂ ), and C / (2 · f ₃ ) as modulo using several different modulation frequencies of light radiation energy such as Good. Using a plurality of different modulation frequencies has the effect of reducing aliasing. If f ₁ , f ₂ , and f ₃ are integers, aliasing is reduced to the least common multiple of f ₁ , f ₂ , and f ₃ expressed as LCM (f ₁ , f ₂ , f ₃ ). When f ₁ , f ₂ , and f ₃ are not integers, it is preferable to model into fractions that can be expressed as a ₁ / D, a ₂ / D, and a ₃ / D. Here, _i of a i is an integer, and D = (GCD) represents the greatest common divisor of a ₁ , a ₂ , and a ₃ . From the above, the distance z can be determined by using LCM (a ₁ , a ₂ , a ₃ ) / D as modulo. A similar analytical approach is implemented in the various embodiments of the invention described herein below.

The phase Φ and the distance z mix the signal S ₂ = A · cos (ω · t + Φ) detected by each pixel detector and the signal S ₁ = cos (ω · t) that drives the light energy emitter (or It can be measured by receiving homodyne. The mixed result S ₁ · S ₂ is 0.5 · A · {cos (2 · ω · t + Φ) + cos (Φ)}, and the time average value is 0.5 · A · cos (Φ). If necessary, the amplitude of the detected feedback signal, that is, the luminance A may be separately measured from the output of each pixel detector.

  To perform homodyne measurement of phase Φ and distance z, each pixel detector in the detector array amplifies the signal detected by the associated pixel detector, a low noise amplifier, a variable phase delay unit, a mixer, a low pass filter And dedicated electronics including an integrator. The mixer mixes the sinusoidal signal transmitted from the output of the low noise amplifier with a variable phase delayed signal. The output of the mixer is processed by a low-pass filter, integrated, and fed back to the control phase shift of the variable phase delay unit. In equilibrium, the output of each integrator is the time of flight TOF between the associated pixel detector and a point on the target object that is a distance z or the phase Ψ (Ψ = Φ ± π / 2) related to the distance z. And). The analog phase information is immediately digitized and the microprocessor on the chip can then calculate the value of z from each pixel detector to a point on the associated target object. If necessary, the microprocessor can also calculate dz / dt (and / or dx / dt, dy / dt).

  However, in the co-pending patent provisional application referred to by the applicant, a periodic signal having a low peak power and a high frequency component is used, and phase delay is used to make TOF, dz / dt (and / or dx / dt, dy / The detection sensitivity of such systems measuring dt) and other information is considerably increased. More specifically, there is a description of an improved mixer, in which mixing is performed, for example, by using the gate of a MOS transistor or by changing the reverse bias of the photodetector, the quantum efficiency (QE) of the photodetector in the detector array. ) As a result of modulation. Such mixing effects include improved high frequency sensitivity, improved detection signal / noise ratio, smaller form factor, reduced power consumption, and reduced manufacturing costs.

  In the present invention, several examples of QE modulation are described. Conceptually, these embodiments can be divided into two general categories. The first category relates to variable phase delay approaches (those not similar to those described in Applicant's co-pending application serial number 09 / 876,373), but dedicated electronic mixers (eg Gilbert cell etc.) ) Is replaced by QE modulation. The second category mixes with fixed phase using QE modulation and involves the implementation of various spatial and temporal multiplexing approaches. Both methods modulate the QE of a photodiode implemented in MOS by changing the reverse bias of the photodiode or by changing the gate voltage after applying the gate voltage to a photodiode with a photogate implemented in MOS. effective. Either method may use single-ended or double-ended differential signal processing. The differential QE modulation has an effect of enabling a quicker QE modulation, and provides a differential output that sufficiently eliminates a common mode effect caused by ambient light or a dark current of a photodiode. In general, both method categories are advantageous for storing photodetector signal charge in a photodiode capacitor. If necessary, the charge accumulated when QE modulation is stopped may be periodically checked. Such a signal storage approach is preferred over methods that attempt to directly measure high frequency and small amplitude photocurrents.

  Using variable phase delay (category 1), the photocurrent from each QE modulated pixel photodiode (or photogate photodiode) does not need to exhibit wide bandwidth, high frequency response or high closed loop gain Coupled as an input to a relatively high input impedance amplifier. The output of the amplifier is directly applied to a low pass filter (LPF), and the output of this LPF drives the integrator. The output of the integrator is coupled to control a variable phase delay (VPD) that controls the QE modulation signal that drives the photodetector diode. The VPD is also driven by a signal from a periodic signal generator that controls the light energy emitter. There may or may not be a DC offset associated with the output signal from the pixel photodiode detector and the homodyne drive signal. Assuming no offset, the LPF output is zero in steady state. Assuming that there is an appropriate DC offset, the LPF output will have a minimum or maximum value in steady state. Although this method may be implemented single-ended, it is preferred to implement it double-ended using a complementary approach, where positive and negative signals are obtained from photodiodes that are QE modulated and out of phase.

  When using a fixed phase delay (category 2), each photodetector is QE modulated using a fixed homodyne signal. In category 2, different groups or banks of photodiode detectors can be defined in the array in a non-localized manner. For example, the first bank of photodiode detectors is QE modulated with a fixed 0 degree phase shift, the second bank is QE modulated with a fixed 90 degree phase shift, and the third bank is fixed with a 180 degree phase shift. QE modulation and the fourth bank can be QE modulated with a fixed 270 degree phase shift. Within each pixel there may be a photodiode detector corresponding to each of the four banks. The phase information and the luminance information of the target object can be measured by examining the output value of each bank in one pixel. This fixed delay approach simplifies the electronic circuitry associated with each pixel, reduces power consumption, and also reduces the required IC chip area, for temporal and spatial multiplexing. Various technologies can be applied.

  In various embodiments of the present invention, the measurement information on the chip may be a random output rather than in a sequential order, immediately tracking objects that require a three-dimensional image and on-chip signal processing of other information. It can be carried out. The overall system is small, robust, requires relatively few off-chip individual components, and exhibits improved detection signal characteristics. The circuit configuration on the chip can thus simultaneously measure the distance and speed of all points on the object or the distance and speed of all objects in one scene using TOF data.

  Other features and advantages of the present invention are set forth in the following detailed description of the preferred embodiment and are shown in conjunction with the associated drawings.

Detailed Description of the Preferred Embodiment

  The present invention advantageously transmits and detects periodic light energy with high frequency components and relies on the phase shift between the transmitted waveform and the detected waveform to identify the time of flight and thus the distance data z. Although a pulse-type periodic waveform is used, the present invention describes sinusoidal radiation and detection because the waveform is mathematically easy to analyze. However, it is understood that a periodic pulse waveform having a high frequency component including an incomplete sine waveform can be mathematically expressed as a group of complete sine waveforms having various coefficients and frequency multiples. Such waveform transmission and detection has the advantage of being able to utilize a relatively inexpensive light emitter with low peak power and allowing the use of a relatively narrow bandwidth amplifier. This is the applicant's reference US Pat. No. 6,323,942B1 (2001), which describes that a series of pulses with narrow pulse width and low duty cycle are emitted by a very high peak power emitter. ).

  FIG. 2A depicts the high frequency component of a typical and ideal periodic light energy signal emitted in the present invention, here represented as cos (ωt). The period T of the waveform is represented by T = 2 · π / ω. The signal is depicted as if it were AC coupled in the sense that there is no offset of any magnitude. As will be described below, the operating frequency of the transmitted signal is preferably in the range of several hundred MHz, and the average and peak transmission powers are relatively weak, for example, about 50 mW or less.

  A portion of the transmitted energy reaches the target object and at least a portion is reflected and detected in the direction of the device of the present invention. FIG. 2B depicts the returned waveform of the transmitted waveform and is expressed as A · cos (ωt + φ). Where A is the attenuation coefficient and φ is the phase shift resulting from the time of flight (TOF) of energy taken to traverse the distance from the device of the invention to the target object. Knowing the TOF is equivalent to knowing the distance z from a point on the target object, such as the target 20, to the receiving pixel detector in the array of detectors in the system according to the invention.

  FIG. 2C is similar to that shown in FIG. 2B, except that there is a DC offset in the present invention. The waveform shown in FIG. 2B can be expressed as 1 + A · cos (ωt + φ). As will be described below, DC offset is desirable in some embodiments for biasing the photodiode, but does not significantly affect the underlying mathematical calculations. Again, it is assumed that the period T of the waveform in FIG. 2C is T = 2 · π / ω, as shown in FIGS. 2A and 2B.

2D and 2E are useful in understanding the concept of duty cycle used here. For a pulse-type periodic signal as shown in FIG. 2D, the duty cycle d is defined by a time ratio T _H / T, where T _H is the time when the signal is above a given threshold V _H and T Is the period of the signal. The threshold level V _H is usually the average of the maximum signal level and the minimum signal level. In the context of the present invention, the above definition is similar except that T _H represents the time during which the photodiode detector 240-X is modulated, and T is the light emitter as shown in FIG. 2E. This is a repetition period in which the 220 modulation is turned on and off. In the context of the present invention, T _H / T can be reduced if the peak power emission of the light energy emitter 220 is appropriately adjusted to keep the average power constant. As described, if the light energy emitted by the light emitter 220 is periodic, they need not be like a square wave or a square wave. A waveform as shown in FIG. 2E can be emitted and detected. However, it is assumed that the above definition of duty cycle can be applied to a waveform as shown in FIG. 2E.

  Identifying the repetition rate of the transmitted periodic light energy signal includes consideration of the transmitted waveform and duty cycle, the desired accuracy in resolving the distance z, and the peak power requirements of the light energy emitter. With trade-offs. For example, assuming that the detected phase shift information is converted from analog to digital by 8 bits, a periodic transmission signal whose high-frequency component is several hundred MHz, such as 200 MHz, has a resolution of a distance z of about cm. Match. In practice, assuming a continuous sinusoidal waveform, the peak power required from the light energy emitter is about 10 mW. Of course, if the duty cycle of the transmission waveform is reduced to 1%, the peak power of the optical energy light emitter must be increased to about 500 mW. It is understood that the ability to use low peak power light emitters is one of the factors that show the difference between the present invention and applicant's previously referenced US Pat. No. 6,323,942B1 (2001). it can.

  Next, processing and use of the phase shift information of the present invention will be described with reference to FIG. FIG. 3 is a block diagram depicting the present invention 200, which is a three-dimensional imaging system that is preferably fabricated on a single IC 210. FIG. System 200 does not require moving parts and may require relatively few off-chip components. FIG. 3 is an excerpt from the applicant's co-generic utility model application as a reference and may be used to illustrate the present invention, although the circuit details of the various elements of FIG. 3 are different. Is possible. In summary, in various embodiments of the present invention, each photodetector 240-x in array 230 preferably has an associated electronics 250-x that implements QE modulation within the photodetector. Whichever technique is used, variable phase delay or fixed phase delay, in the present invention, the distance z is given by z = φ · C / 2 · ω = φ · C / {2 · (2 · π · f)}. To decide.

  The system 200 includes a light emitter such as a low peak power laser diode or a low peak power LED that can output a peak power of about 50 mW when driven at a repetition rate of several hundred MHz, for example, The duty cycle is close to 100%. At present, useful light emitters are made from materials such as AlGaAs, whose band gap energy is significantly different from silicon, which is the preferred material for CMOS IC 210. Therefore, in FIG. 3, the light emitter 220 is depicted as being out of the chip of 210, but the phantom line surrounding the light emitter 220 instead has a light emitter 220 made from a CMOS compatible material on the IC 210. Indicates that it may be produced.

  The light source 220 is preferably a low peak power LED or laser that emits energy at a wavelength of 800 nm if possible, although other wavelengths are possible. At wavelengths below 800 nm, it becomes radiated visible light and laser fabrication becomes more difficult. At wavelengths above 900 nm, the efficiency of CMOS / silicon photodiodes decreases rapidly. In any case, 1100 nm is the highest wavelength for an element manufactured on a silicon substrate such as the IC 210. By using emitted light having a specific wavelength and filtering incident light of different wavelengths, the system 200 operates with or without ambient light. The ability of system 200 to function in the dark is useful for security and military type imaging applications. The lens 290 mounted off-chip preferably filters the incident light on the sensor array 230 so that the pixel detector 240x receives light from one particular point in the field of view (eg, a point on the surface of the object). Condensed to

  Due to the characteristics of light wave transmission, ordinary lenses 290 can be used to focus light onto the sensor array. If the lens (290 ') needs to collect the light energy transmitted from the light emitter 220, a single lens is used for 290 and 290' if a mirror type arrangement is used. A typical LED or laser diode light emitter 220 probably has a shunt capacity on the order of 100 pF. Therefore, when driving the light emitter 220, a small inductance (possibly on the order of a few nH) is placed in parallel with the capacitance so that the coupled inductance-capacitance will resonate at the periodic frequency of the light emitter, usually several hundred MHz. It is advantageous. Instead, the inductance (again a few nH) can be coupled in series with the light emitter and its parasitic capacitance. Such inductance can be made using a bonding wire to the light emitter, if desired.

  On top of the CMOS compatible IC 210 is an oscillator 225 driver, an array 230 (possibly consisting of a 100 × 100 (or more) pixel detector 240 and a 100 × 100 (or more) associated electronics processing circuit 250), a micro A processor or microcontroller unit 260, preferably a random access memory or RAM and a read only memory or ROM) 270, for example an 8-bit A / F of phase information φ detected by various pixel detectors in the array 230 Preferably, various computational and input / output (I / O) circuitry 280 including an analog / digital (A / D) conversion unit that provides D conversion is fabricated. Depending on the implementation, as part of each electronic processing circuit 250, an (A / D) converter function is provided on a single chip, or a dedicated A / D converter is installed. Preferably, the I / O circuit 280 can also provide a signal that controls the frequency of the oscillator 225 that drives the energy light emitter 220.

  The DATA output lines shown in FIG. 3 represent some or all of the information calculated by the present invention using phase shift information from the various image detectors 240 in the array 230. Preferably, the microprocessor 260 examines successive frames stored in the RAM 270 to identify objects in the view scene. The microprocessor 260 can then calculate the distance z and calculate the object velocities dz / dt, dx / dt, dy / dt. In addition, the circuitry on the chip associated with the microprocessor 260 can provide a desired image of, for example, a user's finger if the application using the system 200 detects a user interface with a virtual input device. Can be programmed to recognize shapes. In such an application, data provided by the microprocessor 260 is converted into keystroke information. Part or all of this information (shown as DATA in FIG. 3) can also be transferred from this IC to an external computer for further processing, for example via a universal serial bus. If the microprocessor 260 has sufficient computing power, additional on-chip processing may be performed. It should also be noted that the output from the array of CMOS compatible detectors 240 can be accessed in a random manner, so that TOF data can be output in any order, if desired.

As another function, the microprocessor 260 operating through the interface circuit 280 periodically oscillates the driver 225 at a desired frequency and a desired duty cycle, for example, f ₁ = 200 MHz. In response to a signal from the oscillator driver 225, the laser diode or LED 220 emits light energy at a desired frequency, duty cycle, eg, f ₁ = 200 MHz. Again, sine or cosine waveforms are assumed for ease of mathematical explanation, but periodic waveforms with similar duty cycle, repetition rate, and peak power, such as square waves, may be used. Good. As described, the present invention has an effect that the average power and the peak power are considerably low, for example, 10 mW. As a result, the cost of the LED emitter 220 is much higher than that of the high peak power laser diode in the applicant's earlier invention described in US Pat. No. 6,323,942B1 (2001). About 30 cents.

The light energy in which the periodic high-frequency component is ideally expressed by S ₁ = cos (ωt) is collected by the lens 290 ′ on the target object 20 at a distance z away. At least a portion of the light energy falling on the target 20 is reflected toward the system 200 and detected by one or more pixel detectors 240 in the array 230. Because of the distance z that separates the target point on the object 20 from any pixel detector 240 in the system 200, and more particularly the array 230, the phase is proportional to the time of flight or the distance z that the detected light energy is. Delay by the amount φ. Incident light energy detected by different pixel detectors 240 may have different phases φ because of different flight times or distances z. In various diagrams including FIG. 3, the incident light energy is represented by S ₂ = A · cos (ωt + φ), and is actually an AC component of a feedback signal including a DC component, for example. However, the DC component is not relatively important and is not shown.

  As described below, in conjunction with the microprocessor 260 and the software installed in the memory 270 executed by the microprocessor, examining and determining the relative phase delay is the electronics 250 associated with each pixel detector 240 in the array 230. It is a function. In applications where the system 200 imagines a data entry mechanism, such as a virtual keyboard, the microprocessor 260 may determine which of several virtual keys or which area of a virtual device, such as a virtual keyboard, is the user's. Process enough detection data to identify whether it was touched with a finger or stylus. Therefore, the DATA output from the system 200 contains various information. Such information includes information such as distance z, velocity dz / dt (and / or dx / dt, dy / dt) of the object 20, and identification of virtual keys touched by the user's hand or stylus, for example. This includes, but is not limited to, object identification.

  The preferred IC 210 also includes a microprocessor or microcontroller unit 260, a memory 270 (preferably including random access memory or RAM and read only memory or ROM), and various computational, input / output (I / O) circuits 280. . For example, the output from the I / O circuit 280 can control the frequency of the oscillator 225 that drives the energy light emitter 220. As another function, the controller unit 260 can calculate the distance z to the object and the velocity (dz / dt, dy / dt, dx / dt) of the object. The DATA output lines shown in FIG. 3 represent some or all of such information calculated by the present invention using phase shift information from various pixel detectors 240.

  The two-dimensional array 230 of pixel sensing detectors is preferably fabricated using standard commercial silicon technology. This has the advantage that a single IC 210 can be fabricated that includes not only the circuits 225, 260, 270, 280, but also various pixel detectors 240 and their associated circuitry 250, and preferably also includes the energy emitter 220. Of course, if such circuits and components can be fabricated on the same IC as the pixel detector array, the signal path can be shortened and the processing time and delay time can be reduced. In FIG. 3, the system 200 may include condenser lenses 290 and / or 290 ′, which are assumed to be fabricated outside the IC chip 210.

  Each pixel detector 240 outputs a current that is equal to the parallel combination of current source, ideal diode, shunt impedance, and noise current source, and is proportional to the amount of incident photon light energy falling on it. In order to realize an array of CMOS pixel diodes or photogate detector elements, it is preferable to utilize CMOS fabrication. Typical photodiode fabrication techniques include well diffusion, substrate diffusion, well to substrate junction and photogate structures. A well-to-substrate photodiode is more sensitive to infrared (IR) and exhibits less capacitance and is therefore preferred over a substrate-diffused photodiode.

  As described, FIG. 4 represents an embodiment described in Applicants' co-pending utility model application. FIG. 4 shows a portion of IC 210 and a portion of array 230, depicting 240-1 to 240-x pixel detectors and 250'-1 to 250'-x typical diode-related electronics. In various figures, including FIG. 4, lens 290 is not shown for clarity. FIG. 4 is not directly related to the present invention, but is included to evaluate and better understand the benefits that the present invention provides. In the description that follows, FIGS. 9A-9C are for QE modulation techniques in Category 1 VPD, FIGS. 10A-10C are for Category 2 fixed phase modulation techniques, and the remaining figures are The features of various technologies are illustrated.

  Although the actual array includes hundreds or thousands of more pixel detectors and associated electronics circuits, FIG. 4 is shown with only two pixel detectors 240 and two associated electronics circuits 250 'for clarity of illustration. Is illustrated. As described, instead of implementing the omnibus A / D function on the IC chip 210 as needed, a dedicated A / D converter is installed as part of each electronics circuit 250'-1 to 250'-x. be able to.

Next, detection of incident light energy by the pixel detector 240-1 will be considered. Assuming that a low power LED or laser diode or the like 220 emits light radiation having an ideal high frequency component S ₁ = cos (ω · t), on the surface of the target 20 (distance away z) The portion of such emitted light reflected at the point is given by S ₂ = A · cos (ω · t + φ). When this incident radiation is received, the pixel detector 240-1 outputs a signal amplified by the low noise amplifier 300. A typical amplifier 300 has a closed loop gain on the order of 12 dB.

  As described, the periodic radiation from the light source 220 is preferably sine or sinusoidal with a high frequency component of several hundred MHz. In spite of this high light modulation frequency, all frequencies of interest are close to this modulation frequency, so it is sufficient for amplifier 300 to have a bandwidth of 100 KHz or so, perhaps as low as tens of KHz. Installing hundreds of thousands of low noise, relatively low bandwidth amplifiers 300 on IC 210 is easier than installing wide bandwidth amplifiers that transmit narrow pulses as in Applicants' original invention, It is understood that this is an economic effort. Thus, in FIG. 4, array 230 functions with a relatively small bandwidth amplifier 300, with the output of each amplifier coupled directly to the first input of the associated mixer 310. The second input of the associated mixer 310 is a signal having a frequency similar to the frequency at the first input. If each amplifier 300 and the related mixer 310 are realized as a single unit, the entire unit has a bandwidth of about several tens of KHz and a high frequency response of about several tens of KHz is sufficient.

As shown in FIG. 4, when the detection signal and the transmission signal are compared, there is a phase shift φ related to the TOF and the distance z. Each circuit 250 ′ -x couples the output of low noise amplifier 300 associated with the first input of mixer 310. In the earlier invention of the applicant described in FIG. 4, the mixer 310 can be realized as a Gilbert cell, a multiplier, or the like.
In essence, each mixer 310 performs homodyne detection of the amplified detection output signal S ₂ from the associated pixel detector 240 with the generator 225 signal S ₁ . Assuming that the emitted light energy has an ideal high-frequency component expressed as a sine wave or cosine wave, the product S ₁ · S ₂ of the mixer output is 0.5 · A · {cos (2 · ω · t + φ) + Cos (φ)}, and the average value is 0.5 · A · cos (φ). The amplitude of the detected feedback signal, that is, the luminance A, may be measured separately from each pixel detector output as necessary. In practice, the 8-bit analog to digital resolution of A · cos (φ) results in a resolution of about 1 centimeter in the z measurement.

Each mixer 310 has a second input coupled to the output of a variable phase delay (VPD) unit 320. The VPD unit 320 can be implemented in various ways. For example, there is a method of using a series-coupled inverter in which the operating power supply voltage changes so as to increase or decrease the speed at which each inverter sends a signal. The first input to each VPD unit 320 is made by the signal generator 225 and S ₁ = cos (ωt), which is the addition or subtraction of the signal coefficients. Assume that VPD 320 adds a variable time delay ψ to the cos (ωt) signal produced by generator 225. Therefore, the mixer 310 mixes the amplified cos (ω · t + φ) signal output from the amplifier 300 and the cos (ω · t + ψ) signal output from the VPD 320. Then, the mixer 310 outputs a signal including 0.5 · A · {cos (φ−ψ) + cos (2 · ω · t + φ + ψ)}. The output of mixer 310 is coupled to the input of low pass filter 340. The filter 340 preferably has a bandwidth of about 100 Hz to several KHz. Thus, the output from the filter 340 becomes a low frequency signal proportional to 0.5 · A · cos (φ−ψ). This low frequency signal is input to the integrator 330 here. The output of the integrator 330 is φ _x for the pixel detector 240-x.

The VDP 320 is driven by two signals having a phase difference of (φ−ψ) but having the same modulation frequency as that emitted by the light emitter 220. It should be noted that the output polarity of integrator 330 changes when the phase shift is ψ = φ ± 90 degrees. In the structure shown in FIG. 4, the phase shift ψ _x = φ _x ± 90 degrees associated with the feedback signal detected by each pixel detector 240-x is obtained from the integrator 330-x of the pixel detector.
The phase shift φ according to the flight time can be obtained as follows.
φ = 2 · ω · z / C = 2 · (2 · π · f) · z / C
Here, C is the speed of light of 300,000 Km / sec. Accordingly, the distance z from the energy emitter 220 to the pixel detector 240-x in the array 230 is determined as follows.
z = φ · C / 2 · ω = φ C / {2 · (2 · π · f)}

The distance z is known as modulo 2πC / (2 · ω) = C / (2 · f). By using several different modulation frequencies such as f ₁ , f ₂ , f ₃ ..., the distance z is C / (2 · f ₁ ), C / (2 · f ₂ ), C / (2 · f ₃₎ makes it possible to determine the modulo the like, further, avoid or aliasing, it can be at least reduced. For example, the microprocessor 260 can instruct the generator 225 to output a sine drive signal of a selected frequency, eg, f ₁ , f ₂ , f ₃ . For example, assuming that f ₁ , f ₂ , and f ₃ are integers such as i = integer, aliasing is the least common multiple of f ₁ , f ₂ , and f ₃ expressed by LCM (f ₁ , f ₂ , f ₃ ). Reduced to When f ₁ , f ₂ , and f ₃ are not integers, it is preferably modeled as a fraction that can be expressed as a ₁ / D, a ₂ / D, or a ₃ / D. Here, a _i is an integer i and D = GCD (a ₁ , a ₂ , a ₃ ), where GCD indicates the greatest common divisor. Therefore, the distance z is determined by using LCM (a ₁ , a ₂ , a ₃ ) / D as modulo.

The two input signals to each mixer 310 are 90 degrees out of phase with each other, for example, in a phase where either ψ _x = φ _x +90 degrees or ψ _x = φ _x −90 degrees is selected depending on the circuit implementation. Sometimes the closed loop feedback circuit configuration of FIG. 4 reaches a stable point. In a proper 90 degree out-of-phase steady state, the output signal from each low pass filter 340 is ideally zero. For example, if the output signal of the low-pass filter 340 signal is positive, the output signal from the associated integrator 330 will further add a phase shift that drives the low-pass filter output to return to the zero state.

When the feedback system is in a steady state, the pixel detector electronics 250 '-x in the array 230, [psi as _X = _{φ X} ± 90 °, [psi _1, [psi _2, various phase angles such ψ ₃ ··· ψ _N I will provide a. Preferably, the phase angle is converted from an analog format to a digital format, for example using an analog / digital conversion function associated with electronics 280. If necessary, the electronics 250'-x can mix signals with a constant phase value for all pixels. The microprocessor 260 can execute software for calculating the distance z (and / or other information) using the mathematical relationship described above, for example, software installed in or mountable on the memory 270. effective. If necessary, the macro processor 260 may use discrete frequencies such as f ₁ , f ₂ , f ₃ ... To improve system performance by reducing or eliminating aliasing errors. The generator 225 can also be commanded to output.

  With continued reference to FIG. 4, various implementations can be used to produce the phase angle ψ = φ ± 90 degrees. Assume that a given application needs to acquire an image at a frame rate of 30 frames / second. In such an application, it is sufficient to sample the phase angle ψ during A / D conversion with a sample rate of about 30 ms. This sample rate is equivalent to the relatively low bandwidth that is inherently present in electronics 250'-x, as shown in FIG. In practice, the system 200 can provide a distance z resolution of about 1 cm, and in practical applications, the range of z is on the order of 100 m or less.

  Note that the distance z is determined from the TOF information obtained from the phase delay ψ, but the relative luminance of the signal returning from the target object 20 can also provide useful information. The amplitude coefficient “A” on the feedback signal indicates relative luminance. The feedback structure of FIG. 4 attempts to achieve a minimum output signal from the low pass filter 340, but with a slight change, the maximum low pass filter output signal may be used instead, and the output signal will exhibit a luminance coefficient A. . Such a structure can be implemented using signals that are 90 degrees out of phase with the output from the VPD 320 to modulate another copy of the output of the low noise amplifier 300. The average amplitude of the signal thus modulated is proportional to the factor A in the incident detected feedback signal.

  Having completed the applicant's description of the prior invention, various embodiments of the present invention will now be described with reference primarily to FIGS. 9A-9C (Category 1) and FIG. 10 (Category 2). In the present invention, a dedicated electronics mixer (as used in the prior invention as shown here in FIG. 4) is not used, but instead a quantum efficiency (QE) modulation technique is used. These QE modulation techniques have the effect of being able to accumulate detected signal charge and are preferred over methods that attempt to directly measure small amplitude signals at high frequencies generated by the detected photocurrent.

Before classifying the QE modulator circuit topography according to the present invention, it is useful to describe the behavior of the MOS diode and how the MOS diode quantum efficiency varies with bias potential and / or photogate potential. . FIGS. 5A and 5B depict a portion of IC 210, here depicting a single photodiode detector 240, shown fabricated on a p-doped substrate 410. FIG. The photodiode 240 has a depletion layer 420 having a width W and an n region 430 having a low doping concentration and an n region 440 having a higher doping concentration thereon. (Here, the terms depletion layer and depletion region are used interchangeably.) The n ⁺ doped region 440 acts as a photodiode anode and the connection to it is shown as 450. A p ⁺ doped region 460 formed in the upper region of the substrate 520 serves as the cathode of the photodiode and the connection thereto is shown as 470. A depletion region 480 having a depletion width of W exists between the − region 430 and the p substrate region 410. (It is understood that the doping polarity described here may be reversed and the structure can be fabricated on an n substrate material instead of on the described p substrate material.)

The width W of the depletion region 480 changes or modulates when the reverse bias voltage applied between the photodiode anode 450 and the photodiode cathode 470 is changed. This bias potential is represented as Vr1 in FIG. 5A and Vr2 in FIG. 5B. 5A and 5B, Vr2> Vr1, and as a result, the width W of the depletion region increases.
For example, photons representing incident light energy, such as energy reflected from the target 20, fall on the photodiodes 240-x in the array 230. For example, see FIG. 3 in the figure. Photons can create electron-hole pairs in the depletion and quasi-neutral regions of these photodiodes. These electron-hole pairs have a relatively long lifetime before recombination. Photons that generate hole pairs in the depletion region have the effect of generating much higher photocurrent per photon than photons that generate electron-hole pairs in the quasi-neutral region of the substrate. This is because electron-hole pairs generated in the depletion region are quickly swept by the electric field and contribute significantly to the resulting photocurrent. In contrast, electron-hole pairs generated in the quasi-neutral region remain there for a while and are likely to recombine without any substantial contribution to the photocurrent. Increasing the width W of the depletion layer provides a wider region where hole pairs are generated and quickly swept to contribute to the photocurrent, thereby increasing the quantum efficiency of the photodiode.

One skilled in the relevant arts will recognize that the width W of the depletion layer is expressed as:
W = [2ε · (φ ₀ + V _R −V _B )] ^0.5 {[qN _A · (1 + N _A / N _D )]} ^−0.5 + [qN _D · (1 + N _D / N _A )] ^−0.5 }
Here, (V _R −V _B ) is a reverse bias of the photodiode 240, N _A and N _d are doping concentrations into the n region and the q region of the diode, respectively, and ψ ₀ = V _T In (N _A N _D / N _i ² ) where V _T = kT / q = 26 mV and n = 1.5 · 10 ¹⁰ cm ⁻³ .

In quantum efficiency (QE) modulation according to the present invention, it is recognized from the above equation that the photodiode depletion width W can be modulated by various reverse biases applied between the anode and cathode of the photodiode. This makes it possible to change the quantum efficiency (QE) of the photodiode, resulting in improved detection sensitivity of the entire system. Table 1 shows typical data for individual PIN photodiodes exposed to a fixed level of light and is a measurement of the photodiode current as the reverse bias voltage applied to the photodiode is varied. Photodiode data implemented in CMOS may of course differ from the data in Table 1.

It should be noted in Table 1 that for a typical PIN photodiode, the magnitude of the photodiode current (eg photocurrent) changes by a factor of four as the reverse bias varies between 0.5 VDC and 2 VDC. That is.
Photodiode reverse bias modulation is a mechanism that changes the QE to improve the detection sensitivity of the photodiodes in the array. However, a more efficient QE modulation detection implementation uses a photogate structure. In such embodiments, the photogate is preferably implemented as a photogate MOS photodiode that modulates the QE by changing the potential applied to the gate of the photodiode structure.

  6A and 6B, it is assumed that the substrate 410 is made of p-type material, and the MOS type source region and drain region of S and D, respectively, are made of n-doped material. However, as described above, the doping polarity may of course be reversed. Further, it is assumed that the source S and the drain D are coupled to each other as shown in FIG. 6A. When the voltage S1 (t) applied to the gate G is high, assuming that it is again an n-channel device, the device 240-x is depleted and then inverted. In this configuration, it is assumed that the gate G and the underlying thin oxide (TOX) are substantially transparent to the incident photon energy S2 (t). This condition is satisfied when the polysilicon material used to make the gate G is not polycide.

  Referring to FIGS. 6A and 6B, the gate structure G is substantially transparent to the incident light energy denoted S2 (t). The structure shown in FIG. 6A includes both a source region and a drain region represented by S and D. In contrast, the structure of FIG. 6B has no drain structure to improve quantum efficiency modulation. In FIG. 6A, since the source region and the drain region are coupled to each other, the device 240x can operate without the drain region as shown in FIG. 6B. In order to realize the IC 70 as described, it is preferable to use a MOS manufacturing process. The present invention is thereby implemented. In many MOS fabrication processes, the drain region of element 240x may be omitted as shown in FIG. 6B. By effectively omitting the drain region, the relative change in the collection efficiency of the device between the low sensitivity operating state and the high sensitivity operating state is increased. As described below, a change in the bias of the optically transparent gate potential changes the shape of the depletion layer. A layer 480 substantially confined to the source region is present when the gate bias is low, and its depletion layer region 480 'extends substantially below the gate region when the gate bias is high.

For example, photocharges such as EH1 and EH2 are generated in the substrate under the gate region in response to photon energy S2 (t). In the absence of a channel in the gate region, most of the photocharge is lost and only the source and drain regions collect the photocharge. However, if the region under the gate is inverted and / or depleted, the generated photocharge is captured and swept into the source and drain regions. This effectively increases the efficiency of the photon collection structure 240-x. The increase in collection efficiency is roughly proportional to the ratio between the area under the gate G and the areas of the source and drain regions, that is, the areas of S and D. If the photogate diode 240x is of an appropriate size, this ratio is 10: 1 or higher. This efficiency suddenly increases, and when the voltage S1 (t) exceeds the threshold, the efficiency increases rapidly. When the channel portion is not doped and the substrate doping is 10 ¹⁷ or more, the threshold is about 0V, the photogate photodetector 240x is in the low sensitivity mode with a gate voltage of about −0.1V, and the gate voltage is about +0. When set to 1, it is in high sensitivity mode. It can be seen that relatively small changes in the gate voltage result in significant changes in device sensitivity.

Figure 6C shows the photogate photodiode 240X, that is approximately equal circuit between the capacitor C ₀ conventional MOS photodiode than bound to D1. Of course, the voltage level of the MOS photodiode may be different from the voltage level of the photogate photodiode. Accordingly, the terms photodiode, photodetector, pixel detector 240x are understood to include a photogate photodiode as described above in connection with FIGS. 6A-6C. Similarly, it can be appreciated that various circuits and analyzes of QE modulation for the conventional MOS photogate described herein can be used for the photogate photodiode 240x as described above. For ease of illustration, most of the embodiments herein are illustrated with respect to a MOS type photodiode detector rather than a photogate detector, but either type of detector can be used.

7A and 7B depict an equivalent circuit of the photodiode detector 240 shown in D1, including parasitic shunt capacitor C _1. FIG. 7A may be referred to as an illustration of high side QE modulation where the modulation signal is coupled through capacitor C ₀ . In FIG. 7B, the modulated signals are combined via capacitor C1, and the figure illustrates low-side QE modulation. In FIG. 7B, the capacitor C ₀ is typically installed in an amplifier (not shown) in the electronics associated with the pixel detector D1.

In the right part of FIG. 7A, an excitation source V2 is coupled to a light emitter L1, such as a laser diode or LED, to cause photoelectron emission from L1 that is proportional to V2. In the left part of FIG. 7A, photodiode D1 receives such photon energy from L1, and photocurrent I1 is induced accordingly. Since photodiode D1 (eg, photodiode 240-x in array 230) is reverse biased, it can be seen that bias source V1 includes a voltage offset. Alternatively, prior to detecting the incident signal, it is possible to precharge the node N _d of photodiode during initialization. 7A and 7B, it can be seen that V2 is similar to the periodic waveform generator 225 and L1 is similar to the light energy emitter 220. (See figures in other figures)

7A and 7B, the bias voltage of the photodiode, and thus the QE of the photodiode, is modulated by the bias source V1. In FIG. 7A, the reverse bias voltage is given by Vd1 = V1 · (C ₀ ) / (C ₀ + C ₁ ), where C ₀ is coupled in series between V1 and D1. From Table 1 and FIGS. 5A and 5B, a large amplitude V1 represents a large reverse bias that can advantageously increase the width W of the photodiode depletion layer region. This in turn increases the sensitivity of the photodiode D1 (or 240) so that the photodiode current I1 corresponds to the photon energy incident from L1 (or the photon energy reflected from the target object 20 and incident). To increase.

When the excitation source V ₂ and the bias source V ₁ operate at the same frequency (ω), the total charge provided by the current source I _{1 for} each cycle is, for example, V ₁ (ωt) when V ₁ and V ₂ are in phase. ) And V ₂ (ωt) are maximized at the same time. This is a result of the maximum sensitivity of the photodiode when the incident photon energy is at its maximum intensity, that is, when the luminance is highest. Conversely, if the D1 sensitivity is minimized when the incident photon energy is maximized, the amount of charge per cycle delivered by I ₁ is minimized.

The change in the amount of charge ΔQ _N on the photodiode node N _d after a given number of cycles is the amount of charge delivered by I ₁ during that cycle. The change in ΔQ _N is determined by measuring the difference in voltage ΔV _D on node N _d before and after capacitors C ₀ and C ₁ are discharged by photocurrent I1. Usually, the photocurrent I ₁ is very small and difficult to measure directly. However, the accumulation effect after many cycles results in a measurable voltage change ΔV _D.

If the anode and cathode terminals of the photodiode are each set to the arbitrary voltage of FIG. 5B, the upper lead of C ₀ can be at ground potential as shown in FIG. 7B. As will be described below with respect to some embodiments, node _Nd is typically coupled to the amplifier input and a shunt capacitor is also coupled to the same input node. The effect of the configuration of FIG. 7B is that the parasitic shunt capacitance of the amplifier can be used as C ₁ instead of an additional capacitor or a dedicated shunt capacitor. By doing so, the number of parts can be reduced and the area required to implement the present invention on an IC chip can be reduced. In addition, this configuration has low noise and is not sensitive to variations in production technology.

  When photon energy falls on the photodiode, there is a time delay between the arrival of incident photon energy and the collection of emitted electrons. This time delay increases greatly as the wavelength of light energy becomes longer, and is about several ns for a waveform of about 850 nm. Thus, the light energy emitter 225 is selected to emit a short wavelength so that the photodiodes 240-x in the array 230 react quickly and can be QE modulated at a high frequency ω.

  Of course, it is desirable that the photodiodes used in the various embodiments of the present invention not only be effectively detected, but also quickly detected. Utilizing a light emitter 220 capable of transmitting light energy at relatively short wavelengths promotes detector efficiency, but making such light emitters is more expensive than light emitters providing long wavelength energy. . For example, a relatively inexpensive laser diode can be used as the light emitter 220 for transmitting energy having a wavelength of about 850 nm. Such a light emitter is relatively inexpensive, but long wavelengths penetrate deep into the structure of the pixel detector, for example 7 μm, resulting in a loss of quantum efficiency and a slow reaction.

  Referring now to the exemplary CMOS structure of FIG. 7, most of the incident photon energy reflected by the target object 20 creates electron-hole pairs (EHx) deep in the epitaxial region 410 of the pixel detector 240. In addition, since the electron-hole pair (EHx ′) is created also in the region 412 deeper in the structure, the quantum efficiency is lowered. Unfortunately, many of these deeply emitted electrons cannot reach the surface area of the photodetector and are collected in the surface area, so these electrons cannot contribute to the photodiode detection signal current. . Furthermore, using a longer wavelength causes an undesirable time delay before the signal current is generated. This delay, typically a few ns, occurs because the diffusion effect is superior to the drift effect in collecting deeply emitted electrons to contribute to the photodiode current.

  If the electrons associated with EHx and EHx 'are somehow moved closer to the surface area of the photodiode structure, the drift effect is superior to the diffusion effect, and the detected current can be seen earlier. Since the doping of the epitaxial layer 410 is very low, it is possible to move electrons made deep in the epitaxial layer using a relatively small current.

Referring to FIG. 7C, the epitaxial layer 410 is typically approximately 7 μm thick, and its dopant concentration is approximately N _A = 10 ¹⁵ / cm ³ , and the underlying high dopant concentration substrate region 412 is approximately several hundred μm. The dopant concentration is about N _A = 10 ¹⁸ / cm ³ . A structure such as that shown in FIG. 7C is readily available from a number of vendors.

  In FIG. 7C, an n-well region 430 and a P ++ region 460 are created in the epitaxial layer 410. N + region 440 is formed with n well region 430. As described below, collection leads 445 and collection leads 447 are provided to move the deeply emitted charge, preferably moved upwards and collected by n-well 430. (The dopant polarity described can be reversed, for example, using an n-type substrate instead, and the dopant level and structure thickness can be varied.)

  Next, a method will be described in which the charge related to EHx is moved upward so that once the charge approaches the n-well, the n-well 430 can finally collect the charge by the diffusion effect. The goal is to encourage deeply emitted electrons sufficiently slowly to be collected by lead 445 associated with the n-well rather than lead 447 associated with the P ++ region. Although the method described below can successfully collect electrons associated with the electron-hole pair EHx, it cannot reach deeper in the structure and collect electrons associated with EHx '. Such movement is indicated by the L-shaped chain line in FIG. 7C. Retrieving EHx 'electrons also requires an unacceptably large current due to the high dopant level associated with layer 412.

Now consider the magnitude of the current required to move the electrons in accordance with the present invention. When viewed from above, the structure shown in FIG. 7C is a square size of 1μmX1μm is, assume that the area is represented by A _s. If the thickness of the region 410 is 7 μm, the resulting volume is 7 × 10 ⁻¹² / cm ³ . The required charge that must be removed from such a volume is 10 ¹⁵ X10 ^-8 X7X10 ^-4 X1.6X10 ^-19 A _s = 1.12X10 ^-15 A _s , where 1.6X10 ^-19 is the electron The charge associated with one. If the goal is to remove such charges within 1 ns, for example, the required current is about 1.12 μA. Although this current is not very small, it is practical to provide this current for every square micron associated with the photodetector array 230. In an array of size 1 mm × 1 mm modulated at 200 MHz, the total current is about 200 mA to move the electrons upward by 7 μm. It can be seen that due to the high dopant level associated with substrate region 412, it is not possible to attempt to recover electrons from EHx ′ using this method.

  Thus, one approach to somehow move deeply emitted electrons upward from the layer 410 for collection is to sweep substantially all holes down about 7 μm. Since the mobility of electrons and holes is fairly close, the electrons thus freed can be moved up by at least 7 microns and reach sufficiently close to the n-well region 430 to favor the effects of the depletion region there. Can be received. The effect of the depletion region facilitates such deeply emitted electrons being collected at the top of the structure.

  By passing a pulse current preferably under the n-well region 430, the holes can be moved downward by about 7 μm while moving the electrons upward by the same distance due to the high mobility. As described, once an electron comes close enough to be affected by the electric field created by the depletion region in the n-well region, the chance of collecting the electron is significantly increased.

  In one embodiment, ohmic contacts 460 are formed on the substrate outside the n-well region 430 and are used to facilitate bringing electrons near the depletion layer. This approach works well because the dopant concentration of the epitaxial layer 410 is relatively low and the charge magnitude required to sweep the electrons upward by about 7 μm is acceptable. Since there are too many holes in the high dopant concentration region at the top of structure 210, there is no reason to recommend moving the electrons upwards by more than 7 μm. If desired, an AC coupling approach using a capacitor structure rather than an ohmic contact may be used instead.

  Next, detector structures using doping gradients in various types of epitaxial regions will be described. Although the structure shown in FIG. 7D is similar to the structure of FIG. 7C, the depth of the structure 240 'of FIG. 7D may be greater than about 7 μm. In FIG. 7D, the epitaxial layer 410 'preferably defines different dopant concentrations ranging from a relatively high concentration (p1) to a lower concentration (p3). The transition of the dopant concentration may be continuous, or may be stepwise, for example, forming separate epitaxial layers with each associated dopant concentration.

  The person skilled in the art knows that there is an electric field associated with the boundary of each doping region. In structure 240 ', where the dopant concentration decreases as it approaches the upper surface of the structure, the direction of the electric field can be defined downward. Electrons in the EHx 'near the upper surface of the region 412 move upward through the interface existing between the region 412 and p1 by the electric field of the interface. Since these electrons do not move down through the interface, it is very likely that they can be induced to move up quickly (due to diffusion effects) near the next epitaxial doping interface (p1, p2). Are induced by the electric field existing at p1 and p2 so as to enter the next dopant region, here p2 again. Once in its (lower dopant concentration) epitaxial region (here p2), the electrons also pass through the p1 and p2 interfaces and do not move down, move up and are affected by the next dopant region (p3) The possibility becomes very high and from there it is guided to enter the area further.

  Of course, the same phenomenon as described above initially works for electrons from EHx pairs that are initially open somewhere in the epitaxial region. It will also be appreciated that less than two or more than four dopant concentrations can be defined within the epitaxial region.

  Therefore, the drift current phenomenon associated with the electric field in various interfaces or boundary regions such as p1, p2, p3 made of epitaxial layers is p1, p2,. . . Through each of the interface areas to guide the electrons to move up quickly.

  As described above, the step-doped epitaxial region functions as a “staging” region or “holding” region for electrons that have arrived close enough to enter the region. However, if a continuous dopant gradient is defined throughout the epitaxial region 410 ', there is no "holding time" within a region (since there is no separate epitaxial region). The effect is to capture the electrons freed for collection by the n-well 430 more quickly and sweep them upward.

  The following sections describe differential QE modulation and the effects it can produce. Again, QE modulation including differential QE modulation may be performed using a conventional MOS type photodiode detector and / or photogate detector.

Referring again to FIGS. 5A and 5B, the incident photon energy generates an electron-hole pair including an electron-hole pair EH ₁ to be generated at an arbitrary position “X” in the illustrated substrate of the photodiode. Suppose that In FIG. 5A, the position X is a quasi-neutral region and not a depletion region (a portion indicated by hatching). In the present invention, it is desirable at this point that the modulation reduces QE and abandon as many electron-hole pairs as possible, including EH ₁ . If the photodiode QE is immediately increased, for example, by increasing the reverse bias of the photodiode, the width W of the depletion region can be increased to cover the position X (see FIG. 5B).

In FIG. 5B, EH ₁ still remains at position X in the depletion region at this point, and EH ₁ now contributes greatly to the photocurrent. On the other hand, the increased depletion region in FIG. 5B can increase photon detection sensitivity. However, electron-hole pairs generated when photons enter when QE is low (FIG. 5A) can contribute to the total photocurrent when QE is high (FIG. 5B). For example, the contribution will be made at another time. An undesirable result is that the QE cannot be changed effectively at high modulation rates. However, it is preferable that only photons that enter during high QE always contribute to the photocurrent.

  It is desirable to achieve faster photodiode QE modulation by removing the time delay effect described above. It is further desirable to eliminate the photodiode output signal common mode effect caused by ambient light and so-called photodiode dark current. Overall, it can be seen that QE modulation is basically modulating the size of the collection target for electrons within the photodiode structure. Without other collection targets, even small targets, most electrons will eventually be collected for a relatively long lifetime. Therefore, with respect to the number of electrons, QE modulation is considerably less than changes in the target area.

  The various elements of the present invention that use differential QE modulation techniques that increase or decrease the size of the acquisition target while increasing or decreasing the size of the alternative adjacent target are now described. The effect is to provide a large alternative target for electrons and holes while reducing the target area of a given photodiode. This allows electrons to be collected on the alternate target long before their lifetime expires, removing them from circulation for the reduced target and increasing QE.

  The present invention recognizes that during QE modulation, some regions within the photodiode, typically regions with lower dopant concentration, become quasi-neutral or depleted. Yes. If these regions can be minimized, the photodiode can be QE-modulated more clearly. As described later with respect to FIGS. 8A and 8B, such enhanced QE modulation is facilitated using a differential modulation approach. 8A and 8B represent “snapshots” of two adjacent photodiodes separated by 180 degrees, designated A and B. FIG. Preferably, in array 230, adjacent photodiodes A and B are sufficiently close to each other and have a small surface area, and each always receives substantially the same amount of incident photon energy. In the photodiode group A or bank A and the photodiode group B or bank B, the QE of the photodiode A reaches the maximum when the QE of the photodiodes is 180 degrees out of phase, that is, when the QE of the photodiode B is minimum, The bias modulation is performed so that the reverse is also true.

  It should be noted in FIGS. 8A and 8B that since the quasi-neutral region 500 between adjacent photodiodes A and B is always quite small, the number of electron-hole pairs created there is also quite small. This is advantageous because it is the quasi-neutral region near the depletion region that reduces QE modulation. In FIG. 8B, when the QE of the photodiode B increases, the photocurrent of the adjacent photodiode B is changed to the electron-hole pair in the quasi-neutral region 500 between the photodiodes A and B. You may sweep in. Since the quasi-neutral region 500 is small, there is little deterioration in QE modulation by the region 500, which is advantageous.

  8A and 8B, assume that photodiode A and photodiode B are reverse-biased at 0 VDC and 2 VDC, respectively. For example, assuming that A and B are manufactured by an appropriate CMOS 0.25 μm process, the photodiode B converts up to 30% more photon energy than the normal photodiode A. The QE of photodiode A increases rapidly from 0 VDC to a slight increase in reverse bias, while photodiode B reverse-biased at about 1 VDC is hardly affected by small changes in reverse bias. Therefore, it is advantageous for maximum QE modulation that the reverse bias of photodiode A is as small as possible. This bias method corresponds to a MOS transistor in which a channel is formed in the quasi-neutral region 500 between the photodiode A and the photodiode B. Although there is no MOS transistor gate structure, it can be assumed to exist for some voltage value in the sub-threshold region under high source-drain voltage.

  During the time frame shown in FIG. 8A, photodiode A is weakly reverse biased. As a result, there can be a significant amount of leakage current between photodiode A and photodiode B, which is the result of the MOS transistor having photodiode A as the source and photodiode B as the drain shown in FIGS. 8A and 8B. Matches subthreshold leakage. Such leakage current forms a polysilicon gate G ′ that is assumed to be transparent to the light energy of interest, at least in the region between photodiode A and photodiode B, and below gate G ′. This can be reduced by forming a thin oxide (TOX) insulating layer. When such a gate is fabricated, the subthreshold leakage current can be controlled by controlling the gate voltage. For an undoped channel, a gate voltage of approximately −0.4 VDC is usually sufficient to significantly reduce leakage current.

  FIG. 8C is a plan view of a portion of the array 230 depicting columns and rows of photodiodes referred to herein as photodiode A or photodiode B. FIG. As indicated by the different diagonal lines, the QE modulation nodes of all photodiodes A are coupled in parallel, and the QE modulation nodes of all photodiodes B are coupled in parallel. In essence, FIG. 8C can be seen as a plan view of one large photodiode A and one large photodiode B. FIG. In the differential QE modulation mode of the present invention, all photodiodes A can be modulated with a phase of 180 degrees from the signal that modulates all photodiodes B. Since there is only a very small quasi-neutral region between photodiode A and photodiode B, the QE of each of the A and B class photodiodes can be clearly modulated. It is only the quasi-neutral region that is substantially in the bottom region of each photodiode that significantly blurs QE modulation at high modulation frequencies.

  Having described an overview of the concepts underlying QE modulation, the various configurations of systems using such techniques are described next. In the first category of embodiments, the present invention uses a variable phase delay (VPD) technique in which a dedicated electronic mixer (eg, Gilbert cell) is replaced with QE modulation. The system topography depicting the first category is shown primarily in FIGS. 9A-9C. In the second category of embodiments, fixed phase delays are mixed using QE modulation to implement various spatial and temporal multiplexing approaches. The system topography depicting the second category is mainly shown in FIG.

  Conveniently, in both categories of embodiments, the photodiodes implemented in MOS can be modified by changing the reverse bias of the photodiode or by changing the gate voltage by adding a photogate to the MOS mounted photodiode. QE can be modulated. Either method may use single-ended or double-ended differential signal processing. Differential QE modulation has the effect of enabling rapid QE modulation and provides a differential output that significantly eliminates common mode effects due to ambient light and photodiode dark current. Either method category is advantageous because it stores the photodetector signal charge in the photodiode capacitor. In each category, the charge may be periodically checked when QE modulation is stopped. Such a signal storage approach is preferred over methods that directly measure high frequency small signal photocurrents.

  Next, FIGS. 9A-9C will be described with respect to a variable phase delay (VPD) QE modulation embodiment of the present invention, referred to as a category 1 embodiment. Using VPD technology, the photocurrent from each of the QE-modulated pixel photodiodes (or photogate photodiodes) does not need to exhibit a wide bandwidth, high frequency, or high closed loop gain, associated with a relatively high input Coupled as an input to an impedance amplifier. The amplifier output is fed directly to a low pass filter (LPF). The output of this low pass filter drives the integrator. The integrator output is coupled to control the variable phase delay (VPD). The variable phase delay (VPD) controls the QE modulation signal that drives the photodetector diode. The VPD is also driven by a signal from a periodic signal generator that controls the light energy transmitter. There may or may not be a DC offset associated with the output signal from the pixel photodiode detector and a DC offset associated with the homodyne drive signal. Assuming no offset, the LPF output is zero in steady state. Assuming that there is an appropriate offset, the LPF output will have a minimum or maximum value in the steady state. Although this method may be implemented single-ended, it is preferably implemented double-ended using a complementary approach where positive and negative signals are generated from photodiodes that are QE modulated out of phase.

  For the sake of clarity, the biasing of the photodiode (or photogate) detector is not explicitly shown. One skilled in the art will recognize that biasing can be accomplished by simply coupling resistors from a reference source to various photodetectors for single-ended QE modulation and differential mode QE modulation. In the case of differential QE modulation, it is further preferred to add feedback to the common mode biasing reference to ensure that the sum of the two signals being compared remains within the desired dynamic range.

  Next, a category 1 variable phase delay (VPD) embodiment will be described with reference to FIG. 9A. FIG. 9A depicts a portion of IC 210, array 230, pixel detectors 240-1 through 240-x, and associated typical electronics 250'-1 through 250'-x for each diode. Elements in FIG. 9A that have the same reference numerals as those in the previous figure may be the same, but not necessarily the same. For example, the variable phase delay unit 320 or filter 340 of FIG. 9A may be the same as the components of FIG. 4, but need not be the same. Each of the pixel diodes 250-x in FIG. 9A has an associated electronic unit denoted 250-x (corresponding to the notation 250'-x in FIG. 4). Again, for ease of illustration, only an electronic circuit associated with two of the thousands of pixel diodes 240 is depicted. Again, if necessary, a dedicated A / D converter is provided as part of each of the electronics 250'-1 to 250'-x rather than implementing the omnibus A / D function on the IC chip 210. You can also.

  Comparing the configuration of FIG. 4 with the configuration shown in FIG. 9A, FIG. 4 provides a dedicated electronics mixer 310 for each pixel diode, whereas electronics 250-x in FIG. The specified mixer is not included. Instead, according to the present invention, the configuration of FIG. 9A uses QE modulation, derives the phase difference between the transmitted signal and the received signal in the data, and derives the TOF. 9A and the other QE modulation embodiments described herein have the advantage that no mixer is used and there is no sufficiently amplified signal that the mixer requires as input for mixing.

  In FIG. 9A, the detected waveform signal photodiode 240-x in the array 230 includes a DC offset of the form .1 + A · cos (ω · t + Φ) as shown in FIG. 2C. The minimum value of this 1 + A · cos (ω · t + Φ) signal is preferably 0 VDC, and the maximum value is preferably +3 VDC. As noted above with respect to FIG. 2C, the change in notation to include any DC offset does not affect the mathematical analysis involved.

In Figure 9A, for each electronic system 250-x in the array 230, the output signal from the variable phase delay (VPD) 320 is coupled to node N _d of photodiode 240-x associated via capacitor C ₀ ing. When the modulated signal coupled with C ₀ is in phase with the detected optical energy such that S ₂ = A · cos (ωt + Φ), the signal generated at both ends of the input impedance R _i of the amplifier 400 becomes the maximum value. R _i is large, for example> 1 GΩ, and the amplitude of the signal voltage across R _i increases slowly over many periods of the periodic signal cos (ωt). The feedback path within each electronics 250-x includes a low pass filter 340 and an integrator 330, and the resulting feedback minimizes the amplitude of the input of amplifier 400, such as the voltage across R _i . When the signal S ₂ = A · cos (ωt + Φ) received by the photodiode 240-x is 180 degrees out of phase with the modulation signal cos (ωt + Ψ), the amplitudes at both ends of Ri are minimized. As shown in FIG. 5, for each electronics 250-x, the resulting phase value Ψ _x can be read as a voltage signal at the output terminal of each integrator 330.

  Thus, electronics 250-x in FIG. 9A works somewhat like electronics 250′-x in FIG. 4 to examine the incident periodic photon energy and measure the distance z from the system to target object 20. Create a phase output signal that can. In FIG. 9A, the amplifier outputs are each routed directly to the input of the low pass filter 340 so that no high frequency response for the amplifier 400 is required. Furthermore, the voltage signal across each amplifier input impedance Ri can increase over a number of periodic cycles. Therefore, the signal finally detected is relatively large, and is preferably several mV or several tens mV, for example. As a result, unlike the amplifier 300 of FIG. 4, in the embodiment of FIG. 9A, the amplifier 400 need not be a very high gain, very low noise, high frequency device. As a result, amplifier 400 can be implemented within a small area of the IC chip and consumes less current, but can provide better distance z resolution than the complex configuration of FIG.

Turning now to FIG. 9B, another Category 1 VPD embodiment is depicted. In Figure 9B, using the output from the complementary 180 degrees phase shifted VPD320, one of which VPD output is coupled to the photodiode D, i.e. 240-x associated via capacitor C _0. Complementary VPD output is here similar via the same capacitor C ₁₀ is coupled between the photodiode represented by D '. Thus, the photodiode 240-x is QE modulated by one VPD output, while the diode D ′ is QE modulated by 180 degrees out of phase with another VPD output. In essence, the QE modulation nodes of the various photodiodes are coupled in parallel such that groups of photodiodes are QE modulated in parallel. Each of the photodiodes 240-x and D 'discharges and generates a common mode signal that requires the reverse bias to each photodiode to be periodically refreshed to a predetermined level. In addition, the configuration of FIG. 9B uses a differential input to amplifier 400 ′, and the effect of ambient light falling on photodiode 240-x of array 230 is minimal. Another advantage provided by the configuration of FIG. 9B is a differential configuration that allows the photodiode D ′ associated with the photodiode 240-x to be rapidly QE modulated with respect to the set of diodes without significant delay. It can be implemented. Thus, for each photodiode 240-x in array 230, a photodiode D ′ having substantially the same characteristics is coupled to the inverting input (in the configuration of FIG. 9B) of each amplifier 400 ′.

  Next, FIG. 9C shows an example of QE modulation of VPD using a differential comparator and a digital integrator. Again, the QE modulation nodes for the various photodiodes are coupled in parallel so that the photodiodes are QE modulated in parallel. In FIG. 9C, the amplifier 400 ′ and the generally analog integrator 330 in FIG. 9B are replaced with a differential comparator 510 and a digital integrator 520. At regular intervals, the microprocessor 260 (see FIG. 3) instructs the energy transmitter 220 to stop emitting or shut down, and both outputs of the VPD 320 are set to a constant voltage. Each differential comparator 510 then compares the differential signal shown at its input node. Each digital integrator 520 then reads the result of this comparison (C), increasing its digital output by a small amount when C = 1, and decreasing its digital output by a small amount when C = 0. If necessary, comparator 510 may be shut down during QE modulation of a photodiode that does not require voltage comparison.

  Consider the following example, with continued reference to FIG. 9C. In the steady state, the output from the digital comparator 510 toggles between “0” and “1”. The output from the digital integrator 520 toggles between two values, for example 5 and 6. The VPD unit 320 creates a delay while toggling between 5 and 6 (in this example). The photodetector D is subsequently modulated with a signal that toggles between cos (ωt + 5) and cos (ωt + 6). In the above example, if the values 5 and 6 are substantially close, the photodiode D appears to be modulated by cos (ωt + 5.5) in the equilibrium state.

  Next, a so-called category 2 embodiment using fixed phase QE modulation will be described mainly with reference to FIG. In the Category 2 embodiment, a fixed phase signal is used to QE modulate each photodetector. Different groups or banks of photodiode detectors can be defined in a non-local manner within the array. For example, the first bank of photodiode detectors is QE modulated with a fixed 0 degree phase shift, the second bank is QE modulated with a fixed 90 degree phase shift, and the third bank is fixed with a 180 degree phase shift. QE-modulate the fourth bank and QE-modulate the fourth bank with a fixed 270 degree phase shift. Within each pixel is a photodiode detector corresponding to all four banks. The phase information and the luminance information of the target object can be determined by examining the output value of each bank in the pixel. This fixed delay approach simplifies the electronic circuitry associated with each pixel, reduces power consumption, reduces the required IC chip area, and provides a variety of temporal and spatial multiplexing options. Technology is also possible.

  Various aspects of Category 2 QE modulation are described, including single-ended or differential spatial and temporal multiplexing, and mappings that do not correspond to a real photodiode to pixel one-to-one. Furthermore, in the category 2 embodiment, an inductor is used, and power consumption can be reduced by detuning or compensating for the loss due to the capacitor capacity.

  Category 2 fixed phase delay QE modulation is described next with reference to FIG. The effect of this configuration is that the electronics 250-x can be simplified to some extent, and brightness measurements can be output, as in other embodiments of QE modulation. In FIG. 10, the photodiode 240-x and the photodiode D ′ of the array 230 are modulated by the fixed phase modulator 530. The output of the fixed phase modulator 530 can select a phase of 0 degree or 90 degrees by, for example, the microcontroller 260 (see FIG. 3). Software that can be included in the memory 270 preferably corrects for (fixed) modulation phase differences between pixel photodiodes due to delays in the path to the pixels. The modulation signal and its complementary signal may be provided to the array 230, and the complementary signal includes a 180 degree delay unit 540 coupled to a single output of the fixed phase delay unit 530 within each pixel electronics 250-x. It may be played back by

In FIG. 10, the system 200 (see FIG. 3) is allowed to operate for a number of periods (core frequency ω), after which the laser or other photon energy transmitter 220 is shut down. When the oscillator 220 is shut down, the diode modulated voltage signal and its complementary signal are set to a constant amplitude. In the following description, a so-called “cos (ωt) +1” analysis is used. Assuming that the QE modulation is somewhat linear, the photodiode (D) signal (B {cos (ωt + Φ) +1) is multiplied by the modulation signal (cos (ωt) +1) and then integrated, the result is B (0. 5 {cos (Φ)} + 1) The result of multiplying the photodiode (D ′) signal (B {cos (ωt + Φ) +1} by the modulation signal (cos (ωt + 180 °) +1)) is B (−0. 5 {cos (Φ)} + 1) By subtracting the two equations, the signal V ₀ = B · cos (Φ) is obtained at the output of the differential amplifier 400 ′, where B is a luminance coefficient. Therefore, a new measurement is performed with a modulation phase shifted by 90 degrees from the original modulation signal, and the result at the output of the differential amplifier 400 ′ is V ₉₀ = B · sin (Ψ), 0 degrees and 90 degrees. With a degree measurement, the angle Ψ can be obtained as follows:
tan (Ψ) = V ₉₀ / V ₀

The luminance B is obtained from the following formula.
B = √ (V ₀ ² + V ₉₀ ² )
Conveniently and in contrast to the embodiment described earlier in the text, the configuration of FIG. 10 does not require an integrator in each electronics 240-x, thus simplifying the system design.
A further advantage of the configuration of FIG. 10 is that a matched impedance inductor can be used to reduce system operating power. For example, assume that the photodiodes 240-x are each about 15 μm square and have a capacity (C) of about 10FF. Also assume that the modulation frequency f = ω / (2π), f is approximately 1 GHz, and the system 200 is operating on a 3 VDC power supply source such as a battery power source. The power consumption for each photodiode pixel is proportional to C · V ² · f and is approximately 8 μW. In the array 230 of 200 pixels × 200 pixels, the power consumption is about 0.32 W.

Since the power consumption is directly proportional to the capacity C, the power consumption can be reduced by reducing the effective capacity. This desirable result can be achieved by coupling a tuning inductor (L _p ) in parallel with the capacitance of the photodiode. However, if a tuning inductor L _p is installed in each pixel as shown in FIG. 11, each inductor L _p is about 100 μH to resonate at 1 GHz, which is too large to be mounted in each pixel photodiode. is there.

In contrast to the embodiment of QE modulation by VPD of FIG. 9C, the embodiment of FIG. 10 is common for each bank of photodiodes coupled in parallel similar to photodiode A and photodiode B of FIG. 8C. All pixels are modulated using the modulation signal. The effect of this configuration is that all photodiodes in the bank of photodiodes coupled in parallel are driven in parallel. The various parasitic shunt capacitances for each of the photodiodes coupled in parallel are themselves coupled in parallel. As a result, one (or relatively few) inductors may be coupled to all photodiodes in one parallel bank to achieve resonance at the desired frequency. In the above example of a 200 × 200 array, 100 μH is required for each pixel photodiode. For example, when 200 × 200 photodiodes are coupled in parallel, the value of L _p is reduced to 100 μH / (200 · 200) or 0.25 nH, which is a very realistic inductance size to manufacture. Furthermore, the size of the array may actually be larger than 200 × 200, in which case the number of photodiodes increases and the overall capacitance of the photodiodes increases, and one inductor L _p required to resonate at the desired QE modulation frequency. The size is further reduced. Such an inductance can be fabricated on the IC chip 210 or mounted outside the chip. In the above example, the effective capacitance of the 200 × 200 photodiode in which one inductor L _p of about 0.25 nH level in FIG. 11B is coupled in parallel is detuned, but in FIG. 11A, each photodiode has a separate large inductance. Requires an inductor.

  The fixed phase delay (category 2) configuration of FIG. 10 is intended as a typical example. In practice, various so-called spatial and temporal multiplexing techniques are used. Different spatial topologies (of which the differential QE variation shown in FIG. 8C is an example) can be used for different groups or banks of photodetectors in the array that can be modulated with fixed phase as a group. Spatial topology can increase the efficiency of signal detection because the photon energy increases the collection efficiency of the charges released into the photodetector. Temporal topology refers to modulating the same bank of photodetectors with different fixed modulation phases at different times. Some spatial topologies allow spatial multiplexing, which can include sharing a photodetector across multiple pixels, for example, reusing the same photodetector for different pixels. Temporal topology allows multiplexing in time, thereby facilitating pipelining. The present invention can be implemented in any or all of these aspects utilizing various pixel bank topologies and various time-phase topologies.

  The spatial multiplexing technique implemented in FIG. 8D is that shown in the typical example of FIG. 10, where the photodetector topology is that of FIG. 8C, with a time topology of 0 ° -180 °, 90 ° -270 °. Is used. Furthermore, the exemplary configuration of FIG. 10 can be used to support not only temporal multiplexing and pipelining, but also spatial multiplexing of photodiodes.

Next, another spatial topology embodiment of the present invention will be described with reference to FIG. 12A. The spatial multiplexing embodiment of FIG. 12A operates in principle similar to the 0 degree-180 degree-90 degree-270 degree time division topology embodiment of FIG. However, the difference is that, for example, the four photodetectors d ₁ or 240- (x), d ₂ or 240- (x + 1), d ₃ or 240- (x + 2) and d ₄ or 240- shown in the plan view of FIG. 12A. Using (x + 3), the measured value is obtained simultaneously at time τ ₁ .
As before, ΔV _d = [ΔV _d1 (τ ₁ ) −ΔV _d2 (τ ₁ )] / [ΔV _d3 (τ ₁ ) −ΔV _d4 (τ ₁ )] = tan (Φ).

  Turning now to FIG. 12B, it can be seen that the photodetector can be shared by different pixels throughout the photodetector array. In FIG. 12B, the four photodetectors shown in FIG. 12A are depicted with diagonal lines so that their dual roles can be seen. For example, the photodiodes d1-d2-d3-d4 can be said to form a cluster of four photodetectors within one pixel of the array 230. However, photodiodes d1 and d3 are also part of a cluster of photodiodes consisting of photodiodes d1, d5, d3, d6, and so on. It should be noted that individual photodiodes can play multiple roles in different clusters, but require additional IC chip portions to implement the spatially multiplexed embodiment as shown. This is therefore to facilitate efficient utilization of IC chip area. If necessary, further data measurements can be obtained by reusing part of the spatial measurements.

Although this may not be the most efficient embodiment, an embodiment of 0-120 degrees-240 degrees time division QE modulation may be implemented if desired. Such an embodiment uses two measurements taken in time frame τ ₁ and time frame τ ₂ from the array of pixels shown in FIG. 8C. For the first measurement at time τ ₁ , the photodetector bank (bank A) consisting of photodetector A operates with a sinusoidal waveform of S1 (t) at a phase of 0 degrees, while the adjacent light The photodetector bank (bank B) consisting of detector B is 120 degrees out of phase due to S2 (t). For the second measurement at time τ ₂ , the phase of bank B is shifted by 120 degrees and the phase of bank A is shifted by 240 degrees. The sum of the phase differences is determined as follows:

ΔV _d = [ΔV _d2 (τ ₂ ) −ΔV _d1 (τ ₂ )] / ΔV _d1 (τ ₁ )
Where at time τ ₁ ,
ΔV _d1 = A [1 + cos (ωt)] cos (ωt + Φ)
ΔV _d1 = A cos (ωt + Φ) +0.5 A {cos (Φ) + cos (2ωt + Φ)}
Also at time τ ₂
ΔV _d1 = A [1 + cos (ωt−120)] cos (ωt + Φ)
ΔV _d1 = A cos (ωt + Φ) +0.5 A {cos (Φ + 120) + cos (2ωt + Φ−120)}
ΔV _d2 = A [1 + cos (ωt−240)] cos (ωt + Φ)
ΔV _d2 = Acos (ωt + Φ) +0.5 A [cos (Φ−120) + cos (2ωt + Φ + 120)]
And

So after filtering,
ΔV _d = [cos (Φ−120) −cos (Φ + 120)] / cos (Φ)
ΔV _d = 2sin (Φ) sin (120) / cos (Φ)
ΔV _d = K ₁ sin (Φ) / cos (Φ)
ΔV _d = K ₁ tan (Φ). Here, K ₁ = √3.
Referring now to FIG. 12C, an example of 0 degree-120 degree-240 degree modulation (spatial multiplexing) is shown. This spatial multiplexing embodiment uses the three detectors d ₁ , d _2, and d ₃ to measure the measured values at the time τ _{1 at} the same time, except for the above 0 ° -120 ° -240 °. This is the same as the embodiment of time division multiplexing.

As above,
ΔV _d = [ΔV _d3 (τ ₁ ) −ΔV _d2 (τ ₁ )] / ΔV _d1 (τ ₁ ) = K ₁ tan (Φ), and K ₁ = √3.
As can be seen from what has been described with respect to FIG. 12B, the photodetector of FIG. 12C may be shared across different pixels in the photodetector array 230.
Referring again to FIG. 8C, it can be seen that each photodetector in bank A can be shared across, for example, four pixels, top, bottom, left, and right. For example, in the second row of photodetectors, the first photodetector A may be associated with each of the four adjacent photodetectors B.

  To enable the spatial multiplexing of the present invention, the raw data is obtained single-ended from each photodetector rather than first generating a differential signal between banks of photodiode detectors to obtain differential data. It turns out that it is advantageous. Nevertheless, QE modulation is preferably performed differentially, that is, the banks of detectors are modulated with different phases. Such single-ended raw data provides greater flexibility when processing the data than when only differential data is available, eg, adding or subtracting data from adjacent photodetectors (eg, (Probably digitally), in that it is preferred. FIG. 13A shows typical differential signal processing of the photodetector output, while FIG. 13B shows single-ended signal processing.

The concept of pipelining for the embodiment as shown in FIG. 10 will now be described. As used herein, pipelining refers to reducing the latency when obtaining pixel measurements in successive frames of acquired data.
Measurements within a frame of acquired data can be interlaced to increase the overall measurement throughput as follows.
0 ° -180 ° measurement: ΔV _d (τ ₁ )
Measurement at 90 degrees to 270 degrees: ΔV _d (τ ₂ ) → ΔV _d (τ ₂ )] / ΔV _d (τ ₁ ) = tan (Φ)
Measurement from 0 degree to 180 degrees: ΔV _d (τ ₃ ) → ΔV _d (τ ₂ )] / ΔV _d (τ ₃ ) = tan (Φ)
Measurement at 0 degree to 270 degrees: ΔV _d (τ ₄ ) → ΔV _d (τ ₄ )] / ΔV _d (τ ₃ ) = tan (Φ), etc.

In this way, a continuous pipeline of measurement information can be calculated with a latency of one measurement while effectively doubling the calculation speed. In fact, one of the effects of the time division multiplexing QE modulation embodiment described above is that the frame rate of data acquisition can be significantly increased. As stated, the on-chip CPU system 260 can be used to perform the information processing steps described herein, and the on-chip electronics 250-x can be used to implement the various forms of QE modulation and signals that have been described. Processing can be performed.
Referring again to FIG. 8A, each of two adjacent photodetectors 240- (x) (ie, detector “A”) and photodetectors 240- (x + 1) (ie, detector “B”) is planar. It is assumed that they have substantially the same area when viewed in the figure. The following describes reducing the adverse effects of non-uniform light intensity falling on these photodetectors, including the effects associated with differences in the effective area of actual photodetectors, and are used with these photodetectors. This is a technique for reducing 1 / f noise related to the gain of an amplifier.

Referring to FIGS. 3 and 8A, it is assumed that photon energy returned from the target object 20 falls on the photodetector A and the photodetector B, and these two photodetectors output different signals, for example, different amplitudes. . The detected output signal may be different for several reasons. The amount of light falling on the photodetector A may be different from the amount of light falling on the photodetector B. The effective detection area of the light detector A and the effective detection area of the light detector B may differ due to component mismatch, or the light detector A is simply made better and exhibits better detection characteristics. It may be.
Referring again to FIG. 10, for simplicity of explanation, using “1 + cos” analysis, the incident photon energy signal that the photodetector A sees is A ′ {cos (ωt + Φ) +1}, and the incident photon energy. Let B ′ {cos (ωt + Φ) +1} be what the photodetector B sees in the signal. If A ′ = B ′, the light quantity is uniform, but in other cases, the light quantity is not uniform. However, in the more general case, A ′ and B ′ are not the same.

In FIG. 10, A ′ {cos (ωt + Φ) +1}, which is an energy signal seen by the photodetector A, is multiplied by {cos (ωt) +1} to obtain A ′ (0.5 cos (Φ) +1) after accumulation. Get. From now on, this equation is shown as an equation {1}. Similarly, B ′ {cos (ωt + Φ) +1}, which is the energy signal seen by the photodetector B, is multiplied by {cos (ωt + 180 °) +1} to obtain B ′ (− 0.5 cos (Φ) +1 after accumulation. ) From now on, this expression will be expressed as Expression {2}. Assuming that A ′ = B ′, it is easy to obtain A ′ cos (Φ) as described above in the text. The problem is that A 'and B' are not identical.
In the description of FIG. 10, the goal was to obtain Kb {cos (Φ)} and Kb {sin (Φ)} using Kb as a luminance coefficient. In the case of non-uniform light quantity, A ′ (cos (ωt + Φ) +1) is multiplied by {cos (ωt + 180 °) +1} and integrated to obtain A ′ (− 0.5 cos (Φ) +1). From now on, this equation is shown as equation {3}. Further, in the present invention, B ′ {cos (ωt + Φ) +1} is multiplied by {cos (ωt) +1} to obtain B ′ (0.5 cos (Φ) +1). From now on, this equation is expressed as equation {4}.

At this point, the present invention executes (expression {1} −expression {2} −expression {3} −expression {4}) to calculate (A ′ + B ′) {cos (Φ)}. Execute. Similarly, as described above with respect to FIG. 10, the same operation can be performed to arrive at a similar (A ′ + B ′) {sin (Φ)}.
Therefore, one calculation may be executed based on (Expression {1} −Expression {2}), and the same calculation may be executed based on (Expression {3} −Expression {4}). Referring now to FIGS. 8A, 10, 14A, and 14B, the procedure can be performed generally as follows.
(1) At time 0 <t <t1, for example, in the modulation of 0 degree and 180 degrees, the detector D, that is, 240- (x) is biased with the signal S1 = 1 + cos (ωt), and the detector 240- (x + 1) Is biased with signal S2 = 1 + cos (ωt + 180 °),
(2) store signals output from these two detectors during time 0 <t <t1 and at time t = t1, and store or sample the differential signal in digital or analog form;
(3) During time t1 <t <t2, the detector 240- (x) is biased with the signal S1 = 1 + cos (ωt + 180 °), and the detector 240- (x + 1) is biased with the signal S2 = 1 + cos (ωt). Apply
(4) Accumulate the output signals from the two detectors, and store or sample the differential signal in digital or analog form at the end of the accumulation at time t = t2,
(5) A difference signal is calculated for the sampled or stored analog and / or digital signal.

14A and 14B depict exemplary techniques for subtraction of signals in the analog and digital domains, respectively. The analog or digital “shared” component 700 can be placed outside the photodiode pixel detector, such as using one shared component for each row in the column-row array of pixel detectors. The sample and hold (S / H) unit in the pixel holds both measurements at all times during the readout operation, which is repeated independently for each column of pixels. Instead, the average value may be calculated within the pixel block, or even analog-digital conversion (ADC) may be performed.
In FIG. 14A, shared circuitry 700 includes an analog adder 710. The analog output of analog adder 710 is digitized by analog-to-digital converter 720. In FIG. 14B, the shared circuitry is essentially a digital adder 730 whose input is negated. The output from the adder 730 is input to the register 740. The output of register 740 is fed back to the input of the adder. A / D converter 720 provides a digital input to the adder. In FIG. 14B, the average calculation is done in the digital domain, and analog-to-digital conversion can be shared across all columns of pixels, which holds the signal until it sends the accumulated voltage signal to the ADC for conversion. This means that S / H is required for each pixel. Therefore, the calculation of the average value of the signal in the digital domain embodiment of FIG. 14B requires twice as many A / D conversions as the analog domain embodiment of FIG. 14A. It will be appreciated that similar approaches can be used in the various other modulation schemes described, including time division multiplexing and spatial multiplexing.

  In various embodiments described herein, the movement of individual objects within the contours of the detected image may be determined by identifying, for example, contour movements between frames of data obtained by the microprocessor 260. Can be calculated. All pixel detectors in the contour can use a uniform speed, which is the speed of the contour. Since the object can be identified using the outline of the object, the processor 260 on the chip can be used to track the object. In such a case, if necessary, the IC chip 210 can export one value (DATA) that represents a change in the position of the entire object 20 regardless of where the object 20 moves. Therefore, instead of exporting the pixels of the entire frame from the IC chip at the frame rate, one vector representing the change in the position of the object may be sent. As a result, the input / output of the IC chip is considerably reduced, and the amount of data processing required outside the chip can be greatly reduced. It can be seen that the microprocessor 260 on the chip can supervise the spatial and / or temporal topology sequencing and can also optimize the spatial and / or temporal multiplexing.

  In other applications, the system 200 may be used to recognize an object that is a virtual input device, such as a keyboard whose virtual keys are pressed by a user's finger. For example, a virtual input device is implemented in the copending US application serial number 09 / 502,499 filed February 11, 2000, “Method and Apparatus for Entering Data Using a Virtual Input Device”. Therefore, a three-dimensional range finding TOF system is used. When the user's hand or stylus “presses” a virtual key or an area of such a device, the system using the TOF measurement determines which key or which area is “pressed”. The system can then output the equivalent keystroke information to an associated device that receives input data from the virtual input device and user interaction, such as a PDA. The present invention can also be used for such applications, in which case DATA in FIG. 3 represents keystroke identification information processed on the chip by microprocessor 260.

  As described, the microprocessor 260, possibly running software associated with the memory 270, can control the modulation of the generator 225 and detection by the various electronic circuits 250. If necessary, the detection signal may be processed using special image processing software. Since system 200 is preferably battery powered due to its low power consumption, once such software determines that sufficient image resolution has been obtained, it selectively stops operating power to various portions of array 230. be able to. Furthermore, if sufficient photon energy reaches array 230 to ensure proper detection, the shape of the signal output by transmitter 220 can be changed. For example, the peak power and / or duty cycle of the transmitter energy can be reduced, and thus the overall power consumption of the system 200 can be reduced. The design trade-off by changing the shape of the light energy output signal is related to the accuracy of the z resolution, user safety, and the processing capability of the transmitter 220.

  In summary, there is an effect that the entire system can be operated with a small battery because the peak power and average power from the transmitter 220 are preferably in the range of tens of mW. Nevertheless, the distance resolution is in the cm range and the signal / noise ratio is acceptable. While various embodiments have been described with respect to obtaining information proportional to the distance z, it will be appreciated that the present invention may be implemented to obtain information related only to the brightness of the target object, if desired. In such applications, the present invention can be used as a rather good filter that substantially reduces the impact of ambient light on luminance information. On the other hand, obtaining z information relates to modulating the energy source with a modulation frequency greater than 100 MHz, and applications directed to obtain luminance information will probably cause the energy source to be turned off at a fairly low rate, such as 50 Khz. Modulate.

  Modifications and changes may be made to the disclosed embodiments without departing from the scope and spirit of the invention as defined by the claims set forth herein.

1 is a diagram illustrating a conventional range finding system based on lightness according to the prior art. FIG. 2 illustrates a transmitted periodic signal having a high frequency component transmitted according to the present invention, where it is an ideal cosine waveform. 2B illustrates a feedback waveform having a phase delay of the signal transmitted in FIG. 2A as used by the present invention. A feedback waveform similar to that shown in FIG. 2B is illustrated, but with a DC offset level as used in the present invention. Fig. 2 illustrates a pulse-type periodic waveform of radiant light energy as currently emitted by the system according to Applicant's prior invention in the form of U.S. Patent No. 6,323,942B1. 2 illustrates a non-pulse type waveform of emitted light energy according to the present invention. 2 is a block diagram of a preferred embodiment of the present invention. 2 is a block diagram showing two pixel detectors and their associated electronics according to the applicant's original utility model application. FIG. 3 is a cross-sectional perspective view of a photodetector diode according to the present invention, showing reverse bias voltage modulation of the depletion layer width for implementing QE modulation. FIG. 3 is a cross-sectional perspective view of a photodetector diode according to the present invention, showing reverse bias voltage modulation of the depletion layer width for implementing QE modulation. 1 shows a photogate photodiode that can be QE modulated by changing the gate voltage according to the present invention. 1 shows a photogate photodiode that can be QE modulated by changing the gate voltage according to the present invention. It shows that the MOS type photodiode coupled in series with the capacitor according to the present invention is almost equivalent to the photogate photodiode as shown in FIG. 6A. 5A and 5B are equivalent circuit and voltage bias configurations of the exemplary photodiodes of FIGS. 5A and 5B in accordance with the present invention, showing higher QE modulation and lower QE modulation, respectively. 5A and 5B are equivalent circuit and voltage bias configurations of the exemplary photodiodes of FIGS. 5A and 5B in accordance with the present invention, showing higher QE modulation and lower QE modulation, respectively. FIG. 3 is a cross-section of a typical photodetector structure according to the present invention, illustrating how the charge generated by photon energy is recovered using current. FIG. 2 is a cross-section of a typical photodetector structure according to the present invention, showing a continuous or discontinuous change in the dopant concentration of the epitaxial layer and illustrating how the charge generated by the photon energy is recovered by the current. ing. FIG. 5 is a side cross-sectional view of two adjacent photodiodes with low leakage gates that are 180 degree phase shifted and QE modulated in accordance with the present invention. FIG. 5 is a side cross-sectional view of two adjacent photodiodes with low leakage gates that are 180 degree phase shifted and QE modulated in accordance with the present invention. In a plan view of an array of photodiodes according to the invention, the modulation nodes of a bank of photodiodes are alternately and parallel coupled in parallel to the remaining bank of photodiodes for QE modulation. 2 is a block diagram illustrating two photodetectors and their associated electronics in an embodiment of the single-ended variable phase delay (VPD) QE modulation of the present invention. FIG. 4 is a block diagram of a VPD embodiment showing two pixels, consisting of four photodetectors whose photodiodes are QE differentially modulated and their associated electronics, in accordance with the present invention. FIG. 4 is a block diagram of a VPD embodiment showing two pixels, consisting of four photodetectors whose photodiodes are QE differentially modulated and their associated simple electronics including a digital integrator, in accordance with the present invention. FIG. 4 is a block diagram showing two pixels, consisting of four photodiodes and their associated electronics where a selectable fixed phase QE modulation of the photodiodes is used according to the present invention. FIG. 11 illustrates the use of a tuned inductor in a photodiode of the configuration of FIG. 10 to reduce power consumption according to the present invention. FIG. 11 illustrates the use of a tuned inductor in a photodiode of the configuration of FIG. 10 to reduce power consumption according to the present invention. FIG. 6 is a plan view of an embodiment of 0-90 degrees-180 degrees-270 degrees spatial multiplexing QE modulation according to the present invention, showing four adjacent photodiodes. FIG. 12D illustrates sharing a photodetector among different pixels for the embodiment of spatial multiplexing QE modulation of FIG. 12A in accordance with the present invention. FIG. 4 illustrates an embodiment of 0-120 degrees-240 degrees spatial multiplexing QE modulation according to the present invention, showing three photodetectors. FIG. 6 illustrates differential signal processing and single-ended signal processing of photodetector output according to the present invention. FIG. 6 illustrates differential signal processing and single-ended signal processing of photodetector output according to the present invention. FIG. 4 illustrates a circuit configuration for reducing the effects of non-uniform illumination and 1 / f noise on a photodetector according to the present invention. FIG. 4 illustrates a circuit configuration for reducing the effects of non-uniform illumination and 1 / f noise on a photodetector according to the present invention.

Claims (50)

A method for measuring a distance z between at least one photodetector and one target, comprising:
(A) irradiating the target with light energy having a modulated periodic waveform including a high-frequency component S ₁ (ω · t);
(B) differentially detecting a portion of the light energy reflected from the target with a pair of semiconductor photodetectors formed adjacently on one integrated circuit;
(C) the adjacent by applying to each of the high-frequency component S ₁ (ω · t) have a same modulation frequency as and voltages of opposite phases to each other of the pair of semiconductor photodetectors created by the semiconductor optical Modulating the quantum efficiency of the semiconductor photodetector by changing the width of the depletion region of the detector, between the light energy emitted in step (a) and the signal detected in step (b) Measuring a phase change and outputting data proportional to the distance z from the phase change;
And the quantum efficiency is the number of electron-hole pairs that contributed to the photocurrent per number of incident photons. A method further comprising a plurality of semiconductor photodetectors fabricated on a single integrated circuit chip, comprising:
The method of claim 1, wherein the integrated circuit chip includes circuitry for performing steps (b) and (c).   The semiconductor photodetector is in any form of (i) a photodiode detector, (ii) a MOS element with a bias gate, and (iii) a MOS element with a photogate. Item 2. The method according to Item 1.   Step (c) is coupled to the source of the modulated periodic waveform and includes using a variable phase delay operating in a closed loop, wherein the phase delay of the variable phase delay is step (b) A method according to claim 1, characterized in that it indicates the phase delay of the detected signal.   The method of claim 1, wherein step (c) uses at least one fixed phase delay.   The method of claim 1, wherein step (c) changes the reverse bias of the semiconductor photodetector.   The method of claim 1, wherein the semiconductor photodetector comprises a photogate detector, and step (c) changes the gate potential of the photogate detector. Defining a bank of the semiconductor photodetectors;
Increasing the efficiency of the quantum efficiency modulation by modulating the bank of semiconductor photodetectors with different phases;
The method of claim 1 further comprising: The semiconductor photodetector is formed on a semiconductor substrate;
Step (c) comprises generating a current in the substrate to facilitate collection of photocharges released into the substrate by the reflected light energy;
The method of claim 1, wherein quantum efficiency modulation is enhanced. The semiconductor photodetector is formed on a semiconductor substrate including an epitaxial region;
Step (c), wherein the epitaxial region comprises (i) a plurality of layers each having a different doping concentration, and the uppermost layer of the plurality of layers is doped at a lower concentration than the lower layer of the plurality of layers. And (ii) at least one selected from the following: (ii) the epitaxial region defines a layer with a dopant gradient such that the doping concentration is higher in the lower portion of the region than in the upper portion thereof The method of claim 1, comprising using a substrate having features. A method further comprising coupling an inductor to detune at least a portion of a capacitance coupled to a voltage node of the semiconductor photodetector that controls its quantum efficiency modulation,
The method of claim 1, wherein power loss of the capacity is reduced. Defining at least a first bank of the semiconductor photodetector and a second bank of the semiconductor photodetector, each bank being quantum efficiency modulated with a constant phase;
Defining at least one pixel comprising one semiconductor photodetector from the first bank and one semiconductor photodetector from the second bank;
Further comprising:
The method of claim 1, wherein step (c) comprises processing the output from one of the semiconductor photodetectors used for one or more of the pixels. Determining a distance z over a plurality of time frames, comprising:
Step (c)
Quantitatively modulating the semiconductor photodetector with at least a first phase shift on a frame-by-frame basis, obtaining information from the semiconductor photodetector during the first phase shift;
Further including
Information obtained during the first phase shift from the semiconductor photodetector is used for at least two of the time frames,
The method of claim 1, wherein: Digitally converting the analog output from each of the semiconductor photodetectors;
The method of claim 1 further comprising:   Step (a) includes at least one of (i) emitting light energy having the frequency ω of at least 100 MHz; and (ii) emitting light energy having a wavelength of about 850 nm. The method of claim 1, characterized in that:   The method of claim 1, further comprising providing an integrated circuit comprising an electronic circuitry that performs at least one of step (b) and step (c). By measuring the amplitude of the portion of the emitted light energy reflected from the target,
(A) irradiating the target with light energy having a modulated periodic waveform including a high-frequency component S ₁ (ω · t);
(B) providing a pair of semiconductor photodetectors fabricated adjacent on a single integrated circuit for detecting the portion of light energy reflected from the target;
(C) differentially detecting the portion of the light energy reflected from the target with a pair of adjacent semiconductor photodetectors created ;
; (D) adjacent a voltage and 90 ° different voltage from the voltage having the same modulation frequency as the high frequency components S ₁ (ω · t) to one of the pair of semiconductor photodetectors alternating manner is applied, and The other of the pair of adjacent semiconductor photodetectors has the same modulation frequency as the high-frequency component S ₁ (ω · t), and has a voltage opposite in phase to the voltage and a voltage different from the voltage different from the voltage by 90 °. Are applied alternately, and a pair of adjacent semiconductor photodetectors are applied with voltages having opposite phases to each other at the same time, thereby changing the width of the depletion region of the semiconductor photodetector. Modulating the quantum efficiency of the photodetector, obtaining signal components corresponding to the voltages of the signal detected in step (c), and outputting data proportional to the amplitude from these signal components;
And the quantum efficiency is the number of electron-hole pairs that contributed to the photocurrent per number of incident photons.   The method according to claim 17, wherein the frequency ω is at least 100 MHz. A system for measuring a distance z between at least one semiconductor photodetector and one target,
A light energy source that emits a modulated periodic waveform having a high frequency component S ₁ (ω · t);
A pair of semiconductor photodetectors created and arranged adjacent to each other on an integrated circuit so as to differentially detect a portion of the light energy reflected from the target;
The high-frequency component S ₁ (ω · t) applying a perforated with and voltages of opposite phases to each other by the same modulation frequency and to the semiconductor photodetector to each of the pair of semiconductor photodetectors positioned created the adjacent Means for modulating the quantum efficiency of the semiconductor photodetector by changing the width of the depletion region of
Measure the phase change between the light energy emitted from the light energy source and the signal detected by at least some of the semiconductor photodetectors, and from this phase change, data proportional to the distance z A circuit for outputting
And the quantum efficiency is the number of electron-hole pairs that contributed to the photocurrent per number of incident photons.   The system of claim 19, wherein the plurality of semiconductor photodetectors and the modulating means are fabricated on a single integrated circuit chip.   The plurality of semiconductor photodetectors may be at least one of (i) a photodiode detector, (ii) a MOS element with a bias gate, and (iii) a MOS element with a photogate. The system of claim 19.   A variable phase delay circuit coupled to the modulated periodic waveform source and operating in a closed loop, wherein the phase delay of the variable phase delay is detected by a semiconductor photodetector for emitted light energy. 20. The system of claim 19, further comprising the variable phase delay circuit indicating a phase delay of the received signal.   The circuit for measuring a phase change between light energy emitted from the light energy source and a signal detected by at least some of the semiconductor photodetectors uses at least one fixed phase delay. 20. The system according to 19.   The system of claim 19, wherein the modulating means changes a reverse bias of the semiconductor photodetector.   20. The system of claim 19, wherein the semiconductor photodetector includes a photogate detector, and the modulating means changes a gate potential of the photogate detector. A circuit configuration for measuring a phase change between the light energy emitted from the light energy supply source and the signal detected by the semiconductor photodetector;
A bank of said semiconductor photodetectors;
A system further comprising:
20. The system of claim 19, wherein the modulating means modulates the bank of the semiconductor photodetector with different phases. The semiconductor photodetector is formed on a semiconductor substrate;
A system further comprising means for generating a current in the substrate to facilitate collection of photocharges emitted into the substrate by the reflected light energy;
20. The system of claim 19, wherein quantum efficiency modulation is enhanced.   The semiconductor photodetector is formed on a semiconductor substrate including an epitaxial region, and the epitaxial region of the substrate comprises (i) a plurality of layers each having a different doping concentration. The uppermost layer is doped at a lower concentration than the lower layer of the plurality of layers; and (ii) the epitaxial region has a dopant gradient such that the doping concentration is higher in the lower portion of the region than in the upper portion thereof. 20. The system of claim 19, wherein the system has at least one feature selected from a feature defining a layer. A system further comprising an inductor coupled to detune at least a portion of a capacitance coupled to a voltage node of the semiconductor photodetector that controls the quantum efficiency modulation;
20. The system of claim 19, wherein power loss of the capacity is reduced. A first bank of the semiconductor photodetectors;
A second bank of said semiconductor photodetectors;
The modulating means is quantum efficiency modulating the first bank and the second bank with a constant phase;
At least one pixel comprising one semiconductor photodetector from the first bank and one semiconductor photodetector from the second bank;
A system further comprising:
20. The system of claim 19, wherein the circuit processes the output from one of the semiconductor photodetectors used for two or more of the pixels. The system determines the distance z over a plurality of time frames;
On a frame-by-frame basis, the quantum efficiency modulation means modulates the semiconductor photodetector with at least a first phase shift, obtaining information from the semiconductor photodetector during the first phase shift,
Information obtained during the first phase shift from the semiconductor photodetector is used for at least two of the time frames,
The system of claim 19.   The system of claim 19 further comprising means for digitally converting the analog output from each of the semiconductor photodetectors.   The system of claim 19, wherein the frequency ω is at least 100 MHz. An integrated circuit, implemented in CMOS, that measures the distance z between a source of light energy controlled by the system and a target, the integrated circuit being modulated with a high frequency component S ₁ (ω · t) A generator that can be coupled to a source of light energy that emits a periodic waveform;
A pair of semiconductor photodetectors created and arranged adjacent to each other on an integrated circuit so as to differentially detect a portion of the light energy reflected from the target;
The high-frequency component S ₁ (ω · t) applying a perforated with and voltages of opposite phases to each other by the same modulation frequency and to the semiconductor photodetector to each of the pair of semiconductor photodetectors positioned created the adjacent Means for modulating the quantum efficiency of the semiconductor photodetector by changing the width of the depletion region of
Measure the phase change between the light energy emitted from the source of light energy and the signal detected by at least some of the semiconductor photodetectors, and from this phase change is proportional to the distance z A circuit for outputting data;
An integrated circuit implementable in the CMOS, wherein the quantum efficiency is the number of electron-hole pairs that contributed to the photocurrent per number of incident photons.   The plurality of semiconductor detectors may be at least one of (i) a photodiode detector, (ii) a MOS element with a bias gate, and (iii) a MOS element with a photogate. 35. An integrated circuit according to claim 34.   A variable phase delay circuit coupled to the modulated periodic waveform source and operating in a closed loop, wherein the phase delay of the variable phase delay is detected by a semiconductor photodetector for emitted light energy. 35. The integrated circuit of claim 34, further comprising the variable phase delay circuit that exhibits a phase delay of the received signal.   35. The circuit for measuring a phase change between light energy emitted from the light energy source and signals detected by at least some of the semiconductor photodetectors uses at least one fixed phase delay. The integrated circuit according to 1.   35. The integrated circuit of claim 34, wherein the modulating means changes a reverse bias of the semiconductor photodetector.   35. The integrated circuit of claim 34, wherein the semiconductor photodetector includes a photogate detector, and the modulating means changes a gate potential of the photogate detector. An integrated circuit further comprising a bank of said semiconductor photodetectors,
35. The integrated circuit of claim 34, wherein the modulating means modulates the bank of the semiconductor photodetector with different phases. An integrated circuit further comprising means for generating a current in the substrate to facilitate the collection of photocharges released into the substrate by the reflected light energy;
35. The integrated circuit of claim 34, wherein quantum efficiency modulation is enhanced. An integrated circuit further comprising a bias circuit for generating a current in the substrate to facilitate collection of photocharges emitted into the substrate by the reflected light energy;
35. The integrated circuit of claim 34, wherein quantum efficiency modulation is enhanced.   The semiconductor photodetector is formed on a semiconductor substrate including an epitaxial region, and the epitaxial region of the substrate comprises (i) a plurality of layers each having a different doping concentration, and the plurality of layers And (ii) a dopant gradient such that the epitaxial region has a doping concentration higher in the lower portion of the region than in the upper portion thereof. 35. The integrated circuit of claim 34, wherein the integrated circuit has at least one feature selected from a feature defining a certain layer. An integrated circuit further comprising an inductor coupled to detune at least a portion of a capacitance coupled to a voltage node of the semiconductor photodetector that controls its quantum efficiency modulation;
35. The integrated circuit of claim 34, wherein power loss of the capacitance is reduced. A first bank of the semiconductor photodetectors;
A second bank of said semiconductor photodetectors;
The modulating means is quantum efficiency modulating the first bank and the second bank with a constant phase;
At least one pixel comprising one semiconductor photodetector from the first bank and one semiconductor photodetector from the second bank;
An integrated circuit further comprising:
35. The integrated circuit of claim 34, wherein the circuit processes the output from one of the semiconductor photodetectors used for two or more of the pixels. The system measures the distance z over a plurality of time frames;
On a frame-by-frame basis, the quantum efficiency modulation means modulates the semiconductor photodetector with at least a first phase shift, obtaining information from the semiconductor photodetector during the first phase shift,
Information obtained during the first phase shift from the semiconductor photodetector is used for at least two of the time frames,
35. The integrated circuit of claim 34.   35. The integrated circuit of claim 34, further comprising a microprocessor that controls at least the operation of the modulating means.   35. The integrated circuit of claim 34, further comprising means for digitally converting an analog output from each of the semiconductor photodetectors.   35. The integrated circuit of claim 34, wherein the frequency [omega] is at least 100 MHz.   35. The integrated circuit of claim 34, wherein the emitted light energy has a wavelength of about 850 nm.

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