JP3863356B2 - Antenna shared equipment - Google Patents

Description

但し、Ｃ（ハイブリッド回路の結合係数）＝１／２^1/2（＝０．７０７）、θ＝９０°である。
【００１２】
また、測定端子以外の端子を無反射終端器で終端し、端子（Ｔ₁−Ｔ₃）、（Ｔ₃−Ｔ₁）、（Ｔ₂−Ｔ₄）、（Ｔ₄−Ｔ₂）間の結合係数を測定すると、下記（４）式を得ることができる。
【数４】

但し、Ｃ（ハイブリッド回路の結合係数）＝１／２^1/2（＝０．７０７）、θ＝９０°である。
また、測定端子以外の端子を無反射終端器で終端し、端子（Ｔ₁−Ｔ₄）、（Ｔ₄−Ｔ₁）、（Ｔ₂−Ｔ₃）、（Ｔ₃−Ｔ₂）間の結合係数を測定すると、下記（５）式を得ることができる。
【数５】
Ｓ₁₄＝Ｓ₄₁＝Ｓ₂₃＝Ｓ₃₂＝０ ・・・・・・・・・・・・・・・ （５）
前記（２）〜（５）の関係式を用いて、図３に示すハイブリッド回路の〔Ｓ〕マトリクスを求めると、下記（６）式のように表される。
【００１３】
【数６】

図３に示すハイブリッド回路の端子Ｔ₁に、入力電圧（Ｅin）を印加したときに、端子Ｔ₂、Ｔ₃、Ｔ₄から出力される出力電圧（Ｅ₁₁，Ｅ₁₂，Ｅ₁₃，Ｅ₁₄）を、前記（６）式を用いて求めると、下記（７）式のようになる。
【００１４】
【数７】

前記（７）式から分かるように、端子Ｔ₂には、Ｅin／２^1/2（Ｅ₁₂＝Ｅin／２^1/2）の電圧が、端子Ｔ₃には、−ｊＥin／２^1/2（Ｅ₁₃＝−ｊＥin／２^1/2）の電圧が得られる。
【００１５】
図４は、本実施の形態の定インピーダンス帯域通過フィルタにおいて、ハイブリッド回路（Ｈ１）の端子（Ｔ₁₁）に、Ｅin1の電圧が入力された場合における、各部の出力電圧を説明するための図である。
図４において、ＢＰＦ（Ｂ１ａ，Ｂ１ｂ）の入力電圧反射係数を、それぞれΓ₁₁、Γ₂₁とすると、ハイブリッド回路（Ｈ１）の端子（Ｔ₁₂，Ｔ₁₃）における反射電圧（Ｅ_{Γ 11}，Ｅ_{Γ 12}）は、下記（８）式で表される。
【数８】
Ｅ_{Γ 11}＝Γ₁₁Ｅin1／２^1/2
Ｅ_{Γ 12}＝−ｊΓ₂₁Ｅin1／２^1/2 ・・・・・・・・・・・・・・ （８）
前記（６）式、（８）式を用いて、ハイブリッド回路の端子（Ｔ₁₁，Ｔ₁₂，Ｔ₁₃，Ｔ₁₄）における出力電圧（Ｅ₁₁，Ｅ₁₂，Ｅ₁₃，Ｅ₁₄）を求めると、下記（９）式のようになる。
【００１６】
【数９】

前記（９）式から分かるように、ＢＰＦ（Ｂ１ａ，Ｂ１ｂ）の入力電圧反射係数が互いに等しく、Γ（＝Γ₁₁＝Γ₂₁）であれば、ハイブリッド回路（Ｈ１）の第１の端子Ｔ₁₁には、ＢＰＦ（Ｂ１ａ，Ｂ１ｂ）で反射された反射電圧が出力されず、ハイブリッド回路（Ｈ１）の第４の端子Ｔ₁₄にのみ、ＢＰＦ（Ｂ１ａ，Ｂ１ｂ）で反射された反射電圧の合成電圧が出力され、無反射終端器Ｒに吸収される。
【００１７】
図４において、端子（Ｔ₁₂，Ｔ₁₃）からそれぞれ出力されるＥ₁₂（＝Ｅin1／２^1/2）、Ｅ₁₃（＝−ｊＥin1／２^1/2）の電圧は、ＢＰＦ（Ｂ１ａ，Ｂ１ｂ）に入力され、ＢＰＦ（Ｂ１ａ，Ｂ１ｂ）から、ＢＰＦ（Ｂ１ａ，Ｂ１ｂ）の電圧伝達関数（電圧伝達特性）（Ｌ₁₁，Ｌ₁₂）で定まる電圧が出力される。
ＢＰＦ（Ｂ１ａ，Ｂ１ｂ）の出力電圧は、下記（１０）式のように表される。
【数１０】
Ｅ₂₁＝Ｌ₁₁Ｅin1／２^1/2
Ｅ₂₄＝−ｊＬ₂₁Ｅin1／２^1/2 ・・・・・・・・・・・・・・・ （１０）
本実施の形態において、ハイブリッド回路（Ｈ２）の第１の端子（Ｔ₂₁）および第４の端子（Ｔ₂₄）に、前記（１０）式に示す電圧を印加したとき、ハイブリッド回路（Ｈ２）の各端子からの出力電圧は、下記（１１）式のように表される。
【００１８】
【数１１】

また、Ｌは、ＢＰＦ（Ｂ１ａ，Ｂ１ｂ）の電圧伝達係数で下記（１２）で表される。
【数１２】

なお、この（１２）式の各数値については後述する。
前述の（１２）式から分かるように、ＢＰＦ（Ｂ１ａ，Ｂ１ｂ）の電圧伝達係数にバラツキがあると、ハイブリッド回路（Ｈ２）の端子（Ｔ₂₂）に、不平衡出力が生じ、アイソーレーション特性が劣化する。
【００１９】
図５は、本実施の形態の定インピーダンス帯域通過フィルタにおいて、ハイブリッド回路（Ｈ２）の端子（Ｔ₂₂）に、Ｅin2の電圧が入力された場合における、各部の出力電圧を説明するための図である。
図５に示すハイブリッド回路（Ｈ２）の端子（Ｔ₂₂）に、Ｅin2の電圧を印加した場合、端子（Ｔ₂₁）の出力電圧（Ｅ₂₁）、端子（Ｔ₂₄）の出力電圧（Ｅ₂₄）は、下記（１３）式のように表される。
【数１３】

端子（Ｔ₂₁）の出力電圧がＢＰＦ（Ｂ１ａ）に印加され、ＢＰＦ（Ｂ１ａ）で反射された反射電圧（Ｅ_{Γ 21}）と、端子（Ｔ₂₄）の出力電圧がＢＰＦ（Ｂ１ｂ）に印加され、ＢＰＦ（Ｂ１ｂ）で反射された反射電圧（Ｅ_{Γ 24}）は、下記（１４）式のように表される。
【数１４】

前述の（１４）式に示す反射電圧により、ハイブリッド回路（Ｈ２）の各端子に出力される出力電圧は、下記（１５）式のように表される。
【００２０】
【数１５】

前述の（１５）式から分かるように、ハイブリッド回路（Ｈ２）の端子（Ｔ₂₃）にのみ、−ｊΓＥin2の電圧が出力される。
【００２１】
図５に示すハイブリッド回路（Ｈ２）の端子（Ｔ₂₂）に、Ｅin2の電圧が印加された場合に、図５に示すハイブリッド回路（Ｈ１）の端子（Ｔ₁₂）、端子（Ｔ₁₃）に印加される電圧は、下記（１６）式のように表される。
【数１６】
Ｅ₁₂＝Ｌ₁₂Ｅin2／２^1/2
Ｅ₁₃＝−ｊＬ₂₂Ｅin2／２^1/2 ・・・・・・・・・・・・・・ （１６）
前述の（１６）式に示す出力電圧により、ハイブリッド回路（Ｈ１）の各端子に出力される出力電圧は、下記（１７）式のように表される。
【００２２】
【数１７】

【００２３】
前述の（１６）式、（１７）式から分かるように、ハイブリッド回路（Ｈ２）の端子（Ｔ₂₂）に、Ｅin2の電圧を加えた場合に、ハイブリッド回路（Ｈ２）の端子（Ｔ₂₃）に、ＢＰＦ（Ｂ１ａ，Ｂ１ｂ）の反射電圧による電圧が出力され、ハイブリッド回路（Ｈ１）の端子（Ｔ₁₄）に、ＢＰＦ（Ｂ１ａ，Ｂ１ｂ）を通過した電圧による電圧が出力される。
本実施の形態において、ハイブリッド回路（Ｈ１）の端子（Ｔ₁₁）にＥin1の電圧を、ハイブリッド回路（Ｈ２）の端子（Ｔ₂₂）にＥin2の電圧を印加した場合に、ハイブリッド回路（Ｈ１）の端子（Ｔ₁₄）に出力される電圧（Ｅout1）と、ハイブリッド回路（Ｈ２）の端子（Ｔ₂₃）に出力される電圧（Ｅout2）は、下記（１８）式のように表される。
【数１８】

【００２４】
以下、本実施の形態のＢＰＦ（Ｂ１ａ，Ｂ１ｂ）について説明する。
４次有極形（楕円関数形）帯域通過フィルタで、回路次数（ｎ）が偶数の場合の伝送特性（Ｌ）は、下記（１９）式のように表される。
【００２５】
【数１９】

したがって、本実施の形態のＢＰＦ（Ｂ１ａ，Ｂ１ｂ）として使用される、４次有極形（楕円関数形）帯域通過フィルタの伝送特性（Ｌ）は、前述の（１２）式のように表される。
【００２６】
４次有極形帯域通過フィルタの入力電圧の反射損失（Ｌ_Γ）は、下記（２０）式のように表される。
【数２０】
Ｌ_Γ（ｄＢ）＝１０×ｌｏｇ（（Ｓ＋１）^２／４Ｓ） ・・・・・・ （２０）
なお、この（２０）式は、ＢＰＦの通過帯域内の許容リップルを表す。
例えば、Ｓ＝１．４の時は、Ｌ_Γ≒０．１２ｄＢ、Ｓ＝１．５の時は、Ｌ_Γ≒０．１８ｄＢ、Ｓ＝１．６の時は、Ｌ_Γ≒０．２４ｄＢとなる。
また、減衰域の補償減衰量（Ａmin）と、減衰域の反射係数（Γmin）との間には、下記（２１）式に示す関係がある。
【数２１】
Ａmin（ｄＢ）＝１０×ｌｏｇ（１／（１−（Γmin）^２））
（Γmin）^２＝１−１０^{−Ａｍｉｎ／１０}
・・・・・・・・・・・・・・・ （２１）
また、減衰域の反射係数（Γmin）が（２１）式の場合の減衰域の反射損失（Ｌ_Γｍａｘ）は、下記（２２）式で表される。
【数２２】
Ｌ_Γｍａｘ（ｄＢ）＝１０×ｌｏｇ（Γmin）^２
・・・・・・・・・・・・・・・ （２２）
【００２７】
例えば、図６に示すように、Ｓ＝１．４、Ａmin＝１２ｄＢのとき、Ｌ_Γｍａｘ≒０．２８ｄＢ、Ｓ＝１．５、Ａmin＝１３．６ｄＢのとき、Ｌ_Γｍａｘ≒０．１９４ｄＢ、Ｓ＝１．６、Ａmin＝１４．８ｄＢのとき、Ｌ_Γｍａｘ≒０．１４６ｄＢとなる。
なお、減衰域の補償減衰量（Ａmin）は、前述の（１９）式で表される帯域通過フィルタの伝送特性（Ｌ）から求めることができる。
前述の（２０）式と、（２２）式をグラフ化したものが、図７に示すグラフである。
この図７に示すグラフに示すように、電圧定在波比（ＶＳＷＲ）が１．５の時に、４次有極形帯域通過フィルタの入力電圧の反射損失（Ｌ_Γ）と、減衰域での反射損失（Ｌ_Γｍａｘ）とを加算した値が最小となる。
即ち、ＢＰＦ（Ｂ１ａ，Ｂ２ｂ）として、許容通過周波数帯域幅（Ｂ_ｗｒ）が５．６ＭＨｚ、減衰極の周波数（ｆ_∞）が、ｆ_ｏ±３．３５ＭＨｚの４次有極形帯域通過フィルタを使用する場合には、電圧定在波比（ＶＳＷＲ）は、１．５が最適値となる。
なお、ＢＰＦ（Ｂ１ａ，Ｂ２ｂ）として、許容通過周波数帯域幅（Ｂ_ｗｒ）が５．６ＭＨｚ、減衰極の周波数（ｆ_∞）が、ｆ_ｏ±（３．３５±０．１ＭＨｚ）の４次有極形帯域通過フィルタを使用する場合には、電圧定在波比（ＶＳＷＲ）は、１．５±０．１が最適値となる。
【００２８】
一般に、帯域通過フィルタでは、使用される共振器の無負荷Ｑ（Ｑu）が低いと、帯域通過フィルタにおける通過帯域および反射帯域内において、理想的な場合よりも減衰量が大きくなり、湾曲部が大きくなる。
そのため、本願実施の形態では、円形導波管共振器、あるいはデュアルモード誘電体共振器を用いて、無負荷Ｑ（Ｑu）の問題を解決している。
ＢＰＦ（Ｂ１ａ，Ｂ２ｂ）として、デュアルモード誘電体共振器形帯域通過フィルタを用い、Ｑu≒３８０００、ｎ＝４、Ｂ_wr≒５．６ＭＨｚ、ｆ_o＝６００ＭＨｚ、ｆ_∞≒ｆ_o±３．３５ＭＨｚ（５９６．６５ＭＨｚ、６０３．３５ＭＨｚ）として、Ｓ＝１．４、Ｓ＝１．５、Ｓ＝１．６の場合について、計算機で計算した特性を、図８ないし図１０に示す。
図８は、Ｓ＝１．４の時の計算結果、図９は、Ｓ＝１．５の時の計算結果、図１０は、Ｓ＝１．６の時の計算結果である。
図８ないし図１０の各図において、Ａ１は減衰特性、Ａ２は減衰特性（Ａ１）の通過帯域領域を拡大したもの、Ｂは反射減衰量特性、Ｃは群遅延時間特性である。
【００２９】
実際に、ｎ＝４、Ｂ_wr≒５．６ＭＨｚ、ｆ_o＝４８５ＭＨｚ、ｆ_∞≒ｆ_o±３．３５ＭＨｚ（５８１．６５ＭＨｚ、４８８．３５ＭＨｚ）の、デュアルモード誘電体共振器形帯域通過フィルタを試作し、１．４〜１．６内で、Ｓの値を適宜変更した場合の特性例を、図１１ないし図１５に示す。
図１１、図１２は、通過帯域内の減衰特性の一例を示すグラフであり、各図において、横軸は周波数（ＭＨｚ）で、メモリ間隔は１ＭＨｚ、縦軸は減衰量（ｄＢ）で、メモリ間隔は１ｄＢである。
図１１において、周波数が４８５ＭＨｚ（図１１のグラフの１の点）のときの減衰量は、−０．２２９７ｄＢ、周波数が４８２．２ＭＨｚ（図１１のグラフの２の点）のときの減衰量は、−０．３６８９ｄＢ、周波数が４８７．８ＭＨｚ（図１１のグラフの３の点）のときの減衰量は、−０．２３６９ｄＢである。
図１２は、通過帯域内の振幅偏差の一例を示すグラフであり、図１２において、周波数が４８２．２ＭＨｚ（図１１のグラフの２の点）のときの減衰量と、減衰量が最も少ない点（図１２のグラフの１の点）との減衰量との差は、−０．２９８０ｄＢ、周波数が４８７．８ＭＨｚ（図１１のグラフの３の点）のときの減衰量と、減衰量が最も少ない点（図１２のグラフの１の点）との減衰量との差は、−０．１９１１ｄＢである。
この図１１、図１２から分かるように、本実施の形態のＢＰＦ（Ｂ１ａ，Ｂ１ｂ）では、通過帯域内の損失とリップル、および反射帯域内の反射損失とリップルとを小さくでき、かつ、通過帯域内の振幅偏差を少なく（図１２の場合には、０．３ｄＢ以内）することができる。
【００３０】
図１３は、減衰特性の一例を示すグラフであり、同図において、横軸は周波数（ＭＨｚ）で、メモリ間隔は３ＭＨｚ、縦軸は減衰量（ｄＢ）で、メモリ間隔は５ｄＢである。
図１３において、周波数が４８５ＭＨｚ（図１３のグラフの１の点）のときの減衰量は、−０．２２８０ｄＢ、周波数が４８２．２ＭＨｚ（図１３のグラフの２の点）のときの減衰量は、−０．３６９６ｄＢ、周波数が４８１．６６２ＭＨｚ（図１３のグラフの４の点）のときの減衰量は、−２６．９４７ｄＢ、周波数が４８７．８ＭＨｚ（図１３のグラフの３の点）のときの減衰量は、−０．２３１８ｄＢであり、周波数が４８８．３３７ＭＨｚ（図１３のグラフの５の点）のときの減衰量は、−１７．２３１ｄＢである。
図１４は、反射減衰量特性の一例を示すグラフであり、同図において、横軸は周波数（ＭＨｚ）で、メモリ間隔は３ＭＨｚ、縦軸は減衰量（ｄＢ）で、メモリ間隔は１ｄＢである。
図１４において、周波数が４８１ＭＨｚ（図１４のグラフの２の点）のときの減衰量は、−０．２４７８ｄＢ、周波数が４８１．６６２ＭＨｚ（図１４のグラフの４の点）のときの減衰量は、−０．２３３５ｄＢ、周波数が４８８．３３７ＭＨｚ（図１４のグラフの５の点）のときの減衰量は、−０．２７２９ｄＢ、周波数が４８９．２３９ＭＨｚ（図１４のグラフの１の点）のときの減衰量は、−０．３１１１ｄＢであり、周波数が４９１ＭＨｚ（図１４のグラフの３の点）のときの減衰量は、−０．１８２９ｄＢである。
【００３１】
図１５は、減衰特性と反射減衰量特性の一例を示す図であり、同図において、横軸は周波数（ＭＨｚ）で、メモリ間隔は３ＭＨｚ、縦軸は減衰量（ｄＢ）で、メモリ間隔は５ｄＢである。
図１５に示す減衰特性において、周波数が４８５ＭＨｚ（図１５のＡのグラフの１の点）のときの減衰量は、−０．２７１９ｄＢ、周波数が４８２．２ＭＨｚ（図１５のＡのグラフの２の点）のときの減衰量は、−０．４０９６ｄＢ、周波数が４８１．６６２ＭＨｚ（図１５のＡのグラフの４の点）のときの減衰量は、−２７．０５６ｄＢ、周波数が４８７．８ＭＨｚ（図１５のＡのグラフの３の点）のときの減衰量は、−０．２９５８ｄＢ、周波数が４８８．３３７ＭＨｚ（図１５のＡのグラフの５の点）のときの減衰量は、−１７．０６ｄＢである。
また、図１５に示す反射減衰特性において、周波数が４８５ＭＨｚ（図１５のＢのグラフの１の点）のときの減衰量は、−１４．１０３ｄＢ、周波数が４８２．２ＭＨｚ（図１５のＢのグラフの２の点）のときの減衰量は、−２２．２６１ｄＢ、周波数が４８１．６６２ＭＨｚ（図１５のＢのグラフの４の点）のときの減衰量は、−０．２４０９ｄＢ、周波数が４８７．８ＭＨｚ（図１５のＢのグラフの３の点）のときの減衰量は、−１９．７７５ｄＢ、周波数が４８８．３３７ＭＨｚ（図１５のＢのグラフの５の点）のときの減衰量は、−０．２６１９ｄＢである。
【００３２】
以下、本実施の形態のＢＰＦ（Ｂ１ａ，Ｂ１ｂ）として使用される帯域通過フィルタの例について説明する。
図１６は、本実施の形態のＢＰＦ（Ｂ１ａ，Ｂ１ｂ）として使用可能なデュアルモード誘電体共振器形ＢＰＦの一例の概略構成を示す図であり、また、図１７（ａ）は、図１６に示すＡ−Ａ’切断線に沿った断面を示す要部断面図、図１７（ｂ）は、図１６に示すＢ−Ｂ’切断線に沿った断面を示す要部断面図、図１７（ｃ）は、図１６に示すＣ−Ｃ’切断線に沿った断面を示す要部断面図である。図１６、図１７において、１はＴＥ₁₁カットオフ導波管より成る外部導体、２は内部隔壁（結合アイリス）、３は入力（または出力）端子、４は出力（または入力）端子、５ａ，５ｂはＶモード周波数調整ネジ、６ａ，６ｂはＨモード周波数調整ネジ、７ａ，７ｂは結合調整ネジ、１２ａは内部隔壁２に形成された容量性結合窓、１２ｂは内部隔壁２に形成された誘導性結合窓、１４，１５は入出力結合プローブ、２０ａ，２０ｂはデュアルモード誘電体共振素子である。
【００３３】
本実施の形態では、デュアルモード誘電体共振器形ＢＰＦとして、図１８に示すデュアルモード誘電体共振素子を有するものも使用可能である。
図１８は、本実施の形態のＢＰＦ（Ｂ１ａ，Ｂ１ｂ）として使用可能なデュアルモード誘電体共振器形ＢＰＦの他の例の、デュアルモード誘電体共振素子の断面構造を示す断面図である。
なお、この図は、図１６に示すＡ−Ａ’切断線に沿った断面を示す。
図１８に示すデュアルモード誘電体共振素子は、デュアルモード誘電体共振素子２０ａのＥ₁の電界方向の両端部を、Ｅ₁の電界方向対して直角に切断して、デュアルモード誘電体共振素子２０ａのＥ₁の電界方向の両端部に、Ｅ₁の電界方向に対して直角なカット面を形成したことを特徴とする。
即ち、図１８に示すデュアルモード誘電体共振素子２０ａは、円筒状の外部導体１の軸方向に直交する面で切断した断面の外形形状が、互いに対向する２つの円弧状部分（１０ａ，１０ｂ）と、互いに対向する直線部分（１１ａ，１１ｂ）とを有し、この円弧状部分（１０ａ，１０ｂ）と直線部分（１１ａ，１１ｂ）とは、連続していることを特徴とする。
【００３４】
図１８に示すデュアルモード誘電体共振素子を有するデュアルモード誘電体共振器形ＢＰＦでは、誘電体共振素子２０ａの電気長は、Ｅ₁の電界方向では短くなり、Ｅ₂の電界方向では少し短くなるか、あるいは、ほとんど短くならない。
このため、図１８に示すデュアルモード誘電体共振素子を有するデュアルモード誘電体共振器形ＢＰＦでは、第１の誘電体共振器のＨモード共振回路の容量の容量値を、図１６、図１７に示すＢＰＦよりも大きくすることが可能となる。
この結果として、周波数調整ネジ（６ａ）の挿入長を、従来よりも小さくすることが可能となり、高電力のＯＦＤＭ波（直交周波数分割多重（ＯＦＤＭ；Orthogonal Frequency Divison Multiplex）変調方式の変調波）が印加しても、この部分で絶縁破壊を起こすことがなくなるので、耐電圧（耐電力）特性を向上させることが可能となる。
なお、前述の説明では、デュアルモード誘電体共振素子２０ａについて説明したが、デュアルモード誘電体共振素子２０ｂも、同様な構成とされる。
【００３５】
図１９は、本実施の形態のＢＰＦ（Ｂ１ａ，Ｂ１ｂ）として使用可能な円形導波管形ＢＰＦの一例の概略構成を示す図であり、また、図２０（ａ）は、図１９に示すＡ−Ａ’切断線に沿った断面を示す要部断面図、図２０（ｂ）は、図１９に示すＢ−Ｂ’切断線に沿った断面を示す要部断面図である。
同図において、１はＴＥ₁₁カットオフ導波管より成る外部導体、２は内部隔壁（結合アイリス）、３は入力（または出力）端子、４は出力（または入力）端子、５ａ，５ｂはＶモード周波数調整ネジ、６ａ，６ｂはＨモード周波数調整ネジ、７ａ，７ｂは結合調整ネジ、１２ａは内部隔壁２に形成された容量性結合窓、１２ｂは内部隔壁２に形成された誘導性結合窓である。
図２１は、本実施の形態のＢＰＦ（Ｂ１ａ，Ｂ１ｂ）として使用可能な矩形導波管形ＢＰＦの一例の概略構成を示す図であり、同図（ａ）は、上平面図、同図（ｂ）は、同図（ａ）に示すＡ−Ａ’切断線に沿った断面を示す要部断面図、同図（ｃ）は、同図（ａ）に示すＢ−Ｂ’切断線に沿った断面を示す要部断面図である。
図２１において、１１は外部導体、２は隔壁、３は入力（または出力）端子、４は出力（または入力）端子、８は入力（または出力）結合ループ素子、９は出力（または入力）結合ループ素子、２３は容量結合窓、２４は段間結合孔、２５ａ〜２５ｄは周波数調整ネジである。
【００３６】
図２２は、本実施の形態のＢＰＦ（Ｂ１ａ，Ｂ１ｂ）として使用可能なＴＭ_{01 δ}モード誘電体共振器形ＢＰＦの一例の概略構成を示す図であり、誘電体共振素子の軸長方向と直交する方向に沿った断面構造を示す要部断面図である。
図２３（ａ）は、図２２に示すＢ−Ｂ’切断線に沿った断面を示す要部断面図、図２３（ｂ）は、図２２に示すＡ−Ａ’切断線に沿った断面を示す要部断面図である。
図２２、図２３において、１１は外部導体、２は隔壁、３は入力（または出力）端子、４は出力（または入力）端子、８は入力（または出力）結合ループ素子、９は出力（または入力）結合ループ素子、１６はＳ字状の結合ループ素子、２４は段間結合孔、４０ａ〜４０ｄはＴＭ_{01 δ}モード誘電体共振器、４１ａ〜４１ｄは誘電体共振素子、５１ａ〜５１ｃは段間磁界結合調整素子、５６は結合調整ネジである。
図２３に示すように、１番目の誘電体共振素子（４１ａ）と４番目の誘電体共振素子（４１ｄ）との間が、誘電体共振素子の軸長方向に設けられるとともに、隔壁２を貫通する部分を境にして、隔壁２の誘電体共振素子の軸長方向の上下異なる位置で、両端部が前記隔壁２に電気的、機械的に接続されるＳ字状の結合ループ素子１６により副結合されているので、この間の副結合は、容量性結合となる。
【００３７】
図２４は、本実施の形態のＢＰＦ（Ｂ１ａ，Ｂ１ｂ）として使用可能なＴＥ_{01 δ}モード誘電体共振器形ＢＰＦの一例の概略構成を示す図であり、同図（ａ）は、上平面図、同図（ｂ）は、同図（ａ）に示すＡ−Ａ’切断線に沿った断面を示す要部断面図、同図（ｃ）は、同図（ａ）に示すＢ−Ｂ’切断線に沿った断面を示す要部断面図である。
図２４において、１１は外部導体、２は隔壁、３は入力（または出力）端子、４は出力（または入力）端子、８は入力（または出力）結合ループ素子、９は出力（または入力）結合ループ素子、１６はＳ字状の結合ループ素子、２４は段間結合孔、２５ａ〜２５ｄは周波数調整ネジ、４２ａ〜４２ｄは誘電体共振素子、４３ａ〜４３ｃは支持部材、５６ａ〜５６ｃは段間結合調整ネジである。
図２４に示すように、１番目の誘電体共振素子（４２ａ）と４番目の誘電体共振素子（４２ｄ）との間が、誘電体共振素子の軸長方向と直交する方向に設けられるとともに、隔壁２を貫通する部分を境にして、隔壁２の誘電体共振素子の軸長方向と直交する方向の上下異なる位置で、両端部が前記隔壁２に電気的、機械的に接続されるＳ字状の結合ループ素子１６により副結合されているので、この間の副結合は、容量性結合となる。
【００３８】
図２５は、本実施の形態のＢＰＦ（Ｂ１ａ，Ｂ１ｂ）として使用可能な、同軸共振器を用いた有極形インターディジタル形ＢＰＦの一例の概略構成を示す図であり、同図（ａ）は、上平面図、同図（ｂ）は、同図（ａ）に示すＡ−Ａ’切断線に沿った断面を示す要部断面図、同図（ｃ）は、同図（ａ）に示すＢ−Ｂ’切断線に沿った断面を示す要部断面図である。
図２５において、１１は外部導体、２は隔壁、３は入力（または出力）端子、４は出力（または入力）端子、８は入力（または出力）結合ループ素子、９は出力（または入力）結合ループ素子、１６はＳ字状の結合ループ素子、３０ａ〜３０ｄは共振棒である。
この図２５においても、誘電体共振素子の軸長方向に設けられるとともに、隔壁２を貫通する部分を境にして、隔壁２の誘電体共振素子の軸長方向の上下異なる位置で、両端部が前記隔壁２に電気的、機械的に接続されるＳ字状の結合ループ素子１６の副結合回路は、容量性結合となる。
【００３９】
以上説明したように、本実施の形態の空中線共用装置によれば、隣接するチャネルの送信波を合成できるともに、アンテナ（ＡＮＴ）に供給される各チャネル送信波の挿入損失とリップルとを低減し、かつ、その通過帯域内振幅偏差を少なくすることが可能となる。
例えば、デジタル方式のテレビジョン放送の場合には、ビット・エラー・レート特性の制約から、例えば、０．５ｄＢ以内の通過帯域内振幅偏差が求められるが、本実施の形態の空中線共用装置では、この０．５ｄＢ以内の通過帯域内振幅偏差が容易に実現可能となる。
以上、本発明者によってなされた発明を、前記実施の形態に基づき具体的に説明したが、本発明は、前記実施の形態に限定されるものではなく、その要旨を逸脱しない範囲において種々変更可能であることは勿論である。
【００４０】
【発明の効果】
本願において開示される発明のうち代表的なものによって得られる効果を簡単に説明すれば、下記の通りである。
本発明によれば、隣接チャネルの送信波を空中線に供給する空中線共用装置において、各送信波の挿入損失とリップルとを低減するとともに、各送信波の通過帯域内振幅偏差を少なくすることが可能となる。
【図面の簡単な説明】
【図１】本発明の実施の形態の空中線共用装置の概略構成を示すブロック図である。
【図２】本実施の形態の空中線共用装置の伝送特性の一例を示す図である。
【図３】λ／４・３ｄＢ結合器で構成されるハイブリッド回路を説明するための図である。
【図４】本発明の実施の形態の定インピーダンス帯域通過フィルタにおいて、ハイブリッド回路（Ｈ１）の端子（Ｔ₁₁）に、Ｅin1の電圧が入力された場合における、各部の出力電圧を説明するための図である。
【図５】本発明の実施の形態の定インピーダンス帯域通過フィルタにおいて、ハイブリッド回路（Ｈ２）の端子（Ｔ₂₂）に、Ｅin2の電圧が入力された場合における、各部の出力電圧を説明するための図である。
【図６】帯域通過フィルタの電圧定在波比（ＶＳＷＲ）と、減衰域の補償減衰量（Ａmin）との関係を説明するための図である。
【図７】帯域通過フィルタの電圧定在波比（ＶＳＷＲ）と、通過帯域内の反射損、および減衰域内での反射損との関係を示すグラフである。
【図８】本実施の形態の帯域通過フィルタ（Ｂ１ａ，Ｂ２ｂ）として、デュアルモード誘電体共振器形帯域通過フィルタを用いて、Ｓ＝１．４の場合について、計算機で計算した特性を示すグラフである。
【図９】本実施の形態の帯域通過フィルタ（Ｂ１ａ，Ｂ２ｂ）として、デュアルモード誘電体共振器形帯域通過フィルタを用いて、Ｓ＝１．５の場合について、計算機で計算した特性を示すグラフである。
【図１０】本実施の形態の帯域通過フィルタ（Ｂ１ａ，Ｂ２ｂ）として、デュアルモード誘電体共振器形帯域通過フィルタを用いて、Ｓ＝１．６の場合について、計算機で計算した特性を示すグラフである。
【図１１】実際に、デュアルモード誘電体共振器形帯域通過フィルタを試作し、１．４〜１．６内で、Ｓの値を適宜変更した場合の特性例の一例を示すグラフである。
【図１２】実際に、デュアルモード誘電体共振器形帯域通過フィルタを試作し、１．４〜１．６内で、Ｓの値を適宜変更した場合の特性例の他の例を示すグラフである。
【図１３】実際に、デュアルモード誘電体共振器形帯域通過フィルタを試作し、１．４〜１．６内で、Ｓの値を適宜変更した場合の特性例の他の例を示すグラフである。
【図１４】実際に、デュアルモード誘電体共振器形帯域通過フィルタを試作し、１．４〜１．６内で、Ｓの値を適宜変更した場合の特性例の他の例を示すグラフである。
【図１５】実際に、デュアルモード誘電体共振器形帯域通過フィルタを試作し、１．４〜１．６内で、Ｓの値を適宜変更した場合の特性例の他の例を示すグラフである。
【図１６】本発明の実施の形態の帯域通過フィルタ（Ｂ１ａ，Ｂ１ｂ）として使用可能なデュアルモード誘電体共振器形帯域通過フィルタの一例の概略構成を示す図である。
【図１７】図１６に示す各切断線に沿った断面を示す要部断面図である。
【図１８】本発明の実施の形態の帯域通過フィルタ（Ｂ１ａ，Ｂ１ｂ）として使用可能なデュアルモード誘電体共振器形帯域通過フィルタの他の例の、デュアルモード誘電体共振素子の断面構造を示す断面図である。
【図１９】本発明の実施の形態の帯域通過フィルタ（Ｂ１ａ，Ｂ１ｂ）として使用可能な円形導波管形帯域通過フィルタの一例の概略構成を示す図である。
【図２０】図１９に示す各切断線に沿った断面を示す要部断面図である。
【図２１】本発明の実施の形態の帯域通過フィルタ（Ｂ１ａ，Ｂ１ｂ）として使用可能な矩形導波管形帯域通過フィルタの一例の概略構成を示す図である。
【図２２】本発明の実施の形態の帯域通過フィルタ（Ｂ１ａ，Ｂ１ｂ）として使用可能なＴＭ_{01 δ}モード誘電体共振器形帯域通過フィルタの一例の概略構成を示す図である。
【図２３】図２２に示す各切断線に沿った断面を示す要部断面図である。
【図２４】本発明の実施の形態の帯域通過フィルタ（Ｂ１ａ，Ｂ１ｂ）として使用可能なＴＥ_{01 δ}モード誘電体共振器形帯域通過フィルタの一例の概略構成を示す図である。
【図２５】本発明の実施の形態の帯域通過フィルタ（Ｂ１ａ，Ｂ１ｂ）として使用可能な、同軸共振器を用いた有極形インターディジタル形帯域通過フィルタの一例の概略構成を示す図である。
【図２６】従来のアナログ方式のテレビジョン放送における使用チャネル配列の一例を示す図である。
【図２７】使用チャネルが図２６に示す配列の場合の、従来の空中線共用装置の概略構成を示すブロック図である。
【図２８】図２７に示す従来の空中線共用装置の伝送特性を示すグラフである。
【図２９】隣接チャネルを使用するテレビジョン放送における使用チャネル配列の一例を示す図である。
【符号の説明】
１，１１…外部導体、２…隔壁、３…入力（または出力）端子、４…出力（または入力）端子、５ａ，５ｂ，６ａ，６ｂ，２５ａ〜２５ｄ…周波数調整ネジ、７ａ，７ｂ，５６，５６ａ〜５６ｃ…結合調整ネジ、８…入力（または出力）結合ループ素子、９…出力（または入力）結合ループ素子、１２ａ，１２ｂ，２３…結合窓、１４，１５…入出力結合プローブ、１６…Ｓ字状の結合ループ素子、２０ａ，２０ｂ，４１ａ〜４１ｄ，４２ａ〜４２ｄ…誘電体共振素子、２４…段間結合孔、３０ａ〜３０ｄ…共振棒、４０ａ〜４０ｄ…ＴＭ_{01 δ}モード誘電体共振器、４３ａ〜４３ｃ…誘支持部材、５１ａ〜５１ｃ…段間磁界結合調整素子ＡＮＴ…アンテナ、Ｒ…無反射終端器、Ｈ１，Ｈ２，Ｈ１３ａ，Ｈ１３ｂ，Ｈ１４ａ，Ｈ１４ｂ，Ｈ１５ａ，Ｈ１５ｂ，Ｈ２２ａ，Ｈ２２ｂ，Ｈ２３ａ，Ｈ２３ｂ…ハイブリッド回路、Ｂ１ａ，Ｂ１ｂ，Ｂ１３ａ，Ｂ１３ｂ，Ｂ１４ａ，Ｂ１４ｂ，Ｂ１５ａ，Ｂ１５ｂ，Ｂ２２ａ，Ｂ２２ｂ，Ｂ２３ａ，Ｂ２３ｂ…帯域通過フィルタ。[0001]
BACKGROUND OF THE INVENTION
The present invention relates to an antenna sharing device, and more particularly to a technique that is effective when a transmitting antenna in a television broadcasting mechanism of VHF and UHF bands is shared by a plurality of transmitters.
[0002]
[Prior art]
FIG. 26 is a diagram showing an example of a channel arrangement used in a conventional analog television broadcast, and FIG. 27 is a schematic configuration of a conventional antenna sharing apparatus when the channel used is the arrangement shown in FIG. FIG.
As shown in FIG. 27, in the conventional antenna sharing apparatus, the second terminals (T13) of the adjacent first hybrid circuits (H13a, H15a, H23a)._{twenty two}) And the third terminal (T_{twenty three}Are connected to each other, and the second terminals (T) of the first hybrid circuit (H13a) at both ends are connected to each other._{twenty two}) Is connected to the non-reflection terminator (R) at the third terminal (T) of the first hybrid circuit (H23a) at both ends._{twenty three}) Is connected to an antenna (ANT).
[0003]
Further, the first terminal (T of each first hybrid circuit (H13a, H15a, H23a)_{twenty one}) And the second terminal (T) of each second hybrid circuit (H13b, H15b, H23b)₁₂) Are connected to band-pass filters (hereinafter simply referred to as BPF) (B13a, B15a, B23a) that transmit transmission waves of predetermined channels, respectively, and the first hybrid circuits (H13a, H15a, H23a) fourth terminal (T_{twenty four}) And the third terminal (T of each second hybrid circuit (H13b, H15b, H23b)₁₃) Are connected to BPFs (B13b, B15b, B23b) that allow transmission waves of predetermined channels to pass therethrough.
Furthermore, the first terminal (T) of each second hybrid circuit (H13b, H15b, H23b).₁₁) Is connected to a transmitter that outputs a transmission wave of each channel, and the fourth terminal (T) of each second hybrid circuit (H13b, H15b, H23b).₁₄) Are connected to the anti-reflection terminator (R), respectively.
[0004]
Here, each of the first and second hybrid circuits and each BPF constitute a constant impedance band-pass filter that passes the transmission wave of each channel.
That is, the first hybrid circuit (H13a), the second hybrid circuit (H13b), and the BPF (B13a, B13b) are connected to the first hybrid circuit ( H15a), the second hybrid circuit (H15b), and the BPF (B15a, B15b) include a constant-impedance band-pass filter that transmits a 15-channel transmission wave, and the first hybrid circuit (H23a), the second The hybrid circuit (H23b) and the BPF (B23a, B23b) constitute a constant impedance band-pass filter that passes 23-channel transmission waves.
[0005]
Hereinafter, the operation of the antenna sharing apparatus shown in FIG. 27 will be briefly described with an example of 13-channel transmission waves.
In the antenna sharing apparatus shown in FIG. 27, the first terminal (T of the second hybrid circuit (H13b)₁₁The 13-channel transmission waves input from) pass through the BPF (B13a, B13b) and are input to the first hybrid circuit (H13a) and the third terminal (T13) of the first hybrid circuit (H13a)._{twenty three}) Is output.
The 13-channel transmission wave output from the first hybrid circuit (H13a) is input to the first hybrid circuit (H15a), totally reflected by the BPF (B15a, B15b), and the first hybrid circuit (H15a). ) Third terminal (T_{twenty three}) Is output.
Thereafter, similarly, the antenna reaches the antenna (ANT) and is radiated from the antenna (ANT).
[0006]
[Problems to be solved by the invention]
The design conditions of the BPF (B13a, B13b, B15a, B15b, B23a, B23b) used in FIG. 27 are loose, and the transmission characteristics of the antenna shared device shown in FIG. The band is wide and the falling characteristics are gradual.
On the other hand, in recent years, in addition to conventional analog television broadcasting, digital television broadcasting is about to be started due to technological progress.
When transmitting a digital television broadcast in addition to an analog television broadcast, for example, a television using an adjacent channel as shown in FIG. John Broadcast is needed.
[0007]
However, when the above-described antenna sharing device shown in FIG. 27 is used for the purpose of synthesizing the transmission waves of adjacent channels, the antenna sharing device shown in FIG. There is a problem that the insertion loss of the transmission wave supplied to ANT) and the in-band amplitude deviation increase.
The present invention has been made to solve the above-described problems of the prior art, and an object of the present invention is to reduce insertion loss of each transmission wave in an antenna sharing apparatus that supplies transmission waves of adjacent channels to the antenna. In addition, another object is to provide a technique that can reduce the in-band amplitude deviation of each transmission wave.
The above and other objects and novel features of the present invention will become apparent from the description of this specification and the accompanying drawings.
[0008]
[Means for Solving the Problems]
In order to synthesize the transmission waves of adjacent channels, the BPF in the constant impedance band pass filter has a steeper falling characteristic than the conventional one shown in FIG. 27, and the transmission wave supplied to the antenna (ANT). What has a small insertion loss and a small in-band amplitude deviation is required.
The present inventor has found that a quaternary polarized bandpass filter satisfying the following expression (1) is optimal as a bandpass filter for such applications.
[Expression 1]
B_wr= 5.6 ± 0.1 MHz
f_∽= F_o± (3.35 ± 0.1MHz)
VSWR = 1.5 ± 0.1 (1)
Where f_oIs the center frequency of the fourth-order polarized bandpass filter, f_∽Is the frequency of the attenuation pole, B_wrIs an allowable pass frequency bandwidth, and VSWR is a voltage standing wave ratio in the pass band.
[0009]
The invention of the present application has been made based on the above knowledge, and the outline of typical ones of the inventions disclosed in the present application will be briefly described as follows.
That is, the present invention is an antenna sharing apparatus including a plurality of cascaded constant impedance bandpass filters, wherein the first and second bandpass filters in each of the constant impedance bandpass filters are (1 It is a quaternary polarized band-pass filter that satisfies the above formula.
In a preferred embodiment of the present invention, the fourth-order polarized bandpass filter is TM_{01 δ}Mode dielectric resonator type bandpass filter, TE_{01 δ}Mode dielectric resonator type bandpass filter, dual mode dielectric resonator type bandpass filter, circular waveguide type bandpass filter, rectangular waveguide type bandpass filter, or interdigital type using coaxial resonator One of the bandpass filters.
[0010]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.
Note that components having the same function are denoted by the same reference symbols throughout the drawings for describing the embodiment, and the repetitive description thereof will be omitted.
FIG. 1 is a block diagram showing a schematic configuration of an antenna sharing device according to an embodiment of the present invention.
The antenna sharing apparatus according to the present embodiment has a second terminal (T) of the first hybrid circuit (H14b)._{twenty two}) And the first terminal (T) of the second hybrid circuit (H14b, H15b, H22b)₁₁) Is a transmission wave of an adjacent channel shown in FIG. 29, and BPF (B14a, B14b, B15a, B15b, B22a, B22b) satisfies the above-described expression (1) 4 Since it is a second-polarized bandpass filter, it is different from the antenna shared device shown in FIG. 27, and detailed description thereof is omitted.
FIG. 2 is a diagram illustrating an example of transmission characteristics of the antenna sharing device of the present embodiment.
As shown in FIG. 2, the transmission characteristics of the antenna shared device of the present embodiment are narrower in pass band and sharper in falling characteristics than those shown in FIG.
[0011]
Hereinafter, the antenna sharing apparatus of the present embodiment will be described.
FIG. 3 is a diagram for explaining a hybrid circuit including a λ / 4 · 3 dB coupler.
Note that the hybrid circuit of this embodiment includes the second terminal (T₂) And the third terminal (T_ThreeAny hybrid circuit can be used as long as the outputs have a phase difference of 90 ° from each other and the amplitudes are substantially equal.
In the figure, assuming that each terminal other than the measurement terminal is terminated with a non-reflection terminator, the voltage reflection coefficient at each input terminal is zero.
[Expression 2]
S₁₁= S_{twenty two}= S₃₃= S₄₄= 0 (2)
It becomes.
Terminals other than the measurement terminal are terminated with a non-reflection terminator and the terminal (T₁-T₂), (T₂-T₁), (T_Three-T_Four), (T_Four-T_Three), The following equation (3) can be obtained.
[Equation 3]

However, C (coupling coefficient of hybrid circuit) = 1/2^1/2(= 0.707) and θ = 90 °.
[0012]
Terminals other than the measurement terminal are terminated with a non-reflective terminator, and the terminal (T₁-T_Three), (T_Three-T₁), (T₂-T_Four), (T_Four-T₂), The following equation (4) can be obtained.
[Expression 4]

However, C (coupling coefficient of hybrid circuit) = 1/2^1/2(= 0.707) and θ = 90 °.
Terminals other than the measurement terminal are terminated with a non-reflective terminator,₁-T_Four), (T_Four-T₁), (T₂-T_Three), (T_Three-T₂), The following equation (5) can be obtained.
[Equation 5]
S₁₄= S₄₁= S_{twenty three}= S₃₂= 0 (5)
When the [S] matrix of the hybrid circuit shown in FIG. 3 is obtained using the relational expressions (2) to (5), the following expression (6) is obtained.
[0013]
[Formula 6]

Terminal T of the hybrid circuit shown in FIG.₁When the input voltage (Ein) is applied to the terminal T₂, T_Three, T_FourOutput voltage (E₁₁, E₁₂, E₁₃, E₁₄) Using the above equation (6), the following equation (7) is obtained.
[0014]
[Expression 7]

As can be seen from the equation (7), the terminal T₂Ein / 2^1/2(E₁₂= Ein / 2^1/2) Voltage is applied to the terminal T_Three-JEin / 2^1/2(E₁₃= -JEin / 2^1/2) Is obtained.
[0015]
FIG. 4 shows a terminal (T) of the hybrid circuit (H1) in the constant impedance bandpass filter of the present embodiment.₁₁) Is a diagram for explaining the output voltage of each part when the voltage of Ein1 is input.
In FIG. 4, the input voltage reflection coefficients of BPF (B1a, B1b) are₁₁, Γ_{twenty one}Then, the terminal (T of the hybrid circuit (H1)₁₂, T₁₃) Reflected voltage (E_{Γ 11}, E_{Γ 12}) Is represented by the following equation (8).
[Equation 8]
E_{Γ 11}= Γ₁₁Ein1 / 2^1/2
E_{Γ 12}= -JΓ_{twenty one}Ein1 / 2^1/2      (8)
The terminal (T) of the hybrid circuit is obtained by using the expressions (6) and (8).₁₁, T₁₂, T₁₃, T₁₄) Output voltage (E₁₁, E₁₂, E₁₃, E₁₄), The following equation (9) is obtained.
[0016]
[Equation 9]

As can be seen from the above equation (9), the input voltage reflection coefficients of BPF (B1a, B1b) are equal to each other, and Γ (= Γ₁₁= Γ_{twenty one}), The first terminal T of the hybrid circuit (H1)₁₁Does not output the reflected voltage reflected by the BPF (B1a, B1b), and the fourth terminal T of the hybrid circuit (H1).₁₄Only, the combined voltage of the reflected voltage reflected by the BPF (B1a, B1b) is output and absorbed by the non-reflecting terminator R.
[0017]
In FIG. 4, the terminal (T₁₂, T₁₃) Output from each₁₂(= Ein1 / 2^1/2), E₁₃(= -JEin1 / 2^1/2) Is input to the BPF (B1a, B1b), and from the BPF (B1a, B1b) to the voltage transfer function (voltage transfer characteristic) of the BPF (B1a, B1b) (L₁₁, L₁₂) Is output.
The output voltage of BPF (B1a, B1b) is expressed by the following equation (10).
[Expression 10]
E_{twenty one}= L₁₁Ein1 / 2^1/2
E_{twenty four}= -JL_{twenty one}Ein1 / 2^1/2   (10)
In the present embodiment, the first terminal (T of the hybrid circuit (H2)_{twenty one}) And the fourth terminal (T_{twenty four}), When the voltage shown in the expression (10) is applied, the output voltage from each terminal of the hybrid circuit (H2) is expressed as the following expression (11).
[0018]
## EQU11 ##

L is a voltage transfer coefficient of BPF (B1a, B1b) and is expressed by the following (12).
[Expression 12]

Each numerical value of the equation (12) will be described later.
As can be seen from the above equation (12), if there is a variation in the voltage transfer coefficient of BPF (B1a, B1b), the terminal (T2) of the hybrid circuit (H2)_{twenty two}), An unbalanced output occurs, and the isolation characteristics deteriorate.
[0019]
FIG. 5 shows a terminal (T) of the hybrid circuit (H2) in the constant impedance bandpass filter of the present embodiment._{twenty two}) Is a diagram for explaining the output voltage of each part when the voltage of Ein2 is input.
The terminal (T of the hybrid circuit (H2) shown in FIG._{twenty two}) To the terminal (T_{twenty one}) Output voltage (E_{twenty one}), Terminal (T_{twenty four}) Output voltage (E_{twenty four}) Is expressed by the following equation (13).
[Formula 13]

Terminal (T_{twenty one}) Is applied to the BPF (B1a) and reflected by the BPF (B1a) (E_{Γ twenty one}) And terminal (T_{twenty four}) Is applied to the BPF (B1b) and reflected by the BPF (B1b) (E_{Γ twenty four}) Is expressed by the following equation (14).
[Expression 14]

The output voltage output to each terminal of the hybrid circuit (H2) by the reflected voltage shown in the above equation (14) is expressed by the following equation (15).
[0020]
[Expression 15]

As can be seen from the above equation (15), the terminal (T of the hybrid circuit (H2)_{twenty three}) Is output only at −jΓEin2.
[0021]
The terminal (T of the hybrid circuit (H2) shown in FIG._{twenty two}) To the terminal (T) of the hybrid circuit (H1) shown in FIG.₁₂), Terminal (T₁₃) Is expressed as the following equation (16).
[Expression 16]
E₁₂= L₁₂Ein2 / 2^1/2
E₁₃= -JL_{twenty two}Ein2 / 2^1/2   ... (16)
The output voltage output to each terminal of the hybrid circuit (H1) by the output voltage shown in the above equation (16) is expressed as the following equation (17).
[0022]
[Expression 17]

[0023]
As can be seen from the aforementioned equations (16) and (17), the terminal (T of the hybrid circuit (H2)_{twenty two}) To the terminal (T2) of the hybrid circuit (H2) when the voltage Ein2 is applied._{twenty three}), A voltage based on the reflected voltage of BPF (B1a, B1b) is output to the terminal (T of the hybrid circuit (H1).₁₄), A voltage based on the voltage passing through the BPF (B1a, B1b) is output.
In the present embodiment, the terminal (T of the hybrid circuit (H1)₁₁) The voltage of Ein1 to the terminal of the hybrid circuit (H2) (T_{twenty two}) To the terminal of the hybrid circuit (H1) (T₁₄) Output to the terminal (T2) of the hybrid circuit (H2)_{twenty three}) Is output as shown in the following equation (18).
[Formula 18]

[0024]
Hereinafter, the BPF (B1a, B1b) of the present embodiment will be described.
The transmission characteristic (L) when the circuit order (n) is an even number in a quaternary polarized (elliptic function) band-pass filter is expressed by the following equation (19).
[0025]
[Equation 19]

Therefore, the transmission characteristic (L) of the fourth-order polar (elliptic function type) bandpass filter used as the BPF (B1a, B1b) of the present embodiment is expressed as the above-described equation (12). The
[0026]
  Return loss of input voltage (L_Γ) Is expressed by the following equation (20).
[Expression 20]
  L_Γ(DB) = 10 × log ((S + 1)²/ 4S) (20)
  In addition, this (20) Formula represents the allowable ripple in the passband of BPF.
  For example, when S = 1.4, L_Γ≒ 0.12dB, S = 1.5, L_ΓWhen ≒ 0.18dB and S = 1.6, L_Γ≈0.24 dB.
  Further, there is a relationship expressed by the following equation (21) between the compensation attenuation amount (Amin) in the attenuation region and the reflection coefficient (Γmin) in the attenuation region.
[Expression 21]
  Amin (dB) = 10 × log (1 / (1− (Γmin)²))
  (Γmin)²= 1-10^{-Amin / 10}
                                (21)
  Further, when the reflection coefficient (Γmin) of the attenuation region is the expression (21)Reflection loss in the attenuation region(L_Γmax) Is represented by the following equation (22).
[Expression 22]
  L_Γmax(DB) = 10 × log (Γmin)²
                                (22)
[0027]
  For example, as shown in FIG. 6, when S = 1.4 and Amin = 12 dB, L_Γmax≒ 0.28 dB, S = 1.5, Amin = 13.6 dB, L_Γmax≒ 0.194 dB, S = 1.6, Amin = 14.8 dB, L_Γmax≈0.146 dB.
  The compensation attenuation amount (Amin) in the attenuation region is the transmission characteristic (L) of the band-pass filter expressed by the above equation (19).Can be requested from.
  FIG. 7 is a graph obtained by graphing the above equations (20) and (22).
  As shown in the graph of FIG. 7, when the voltage standing wave ratio (VSWR) is 1.5, the reflection loss (L_Γ) And reflection in the attenuation rangeloss(L_Γmax) Is added to the minimum value.
  That is, as the BPF (B1a, B2b), the allowable pass frequency bandwidth (B_wr) Is 5.6 MHz, the frequency of the attenuation pole (f_∞) But f_oWhen a ± 3.35 MHz fourth-order polarized bandpass filter is used, 1.5 is the optimum voltage standing wave ratio (VSWR).
  As the BPF (B1a, B2b), the allowable pass frequency bandwidth (B_wr) Is 5.6 MHz, the frequency of the attenuation pole (f_∞) But f_oWhen using a ± (3.35 ± 0.1 MHz) fourth-order polarized bandpass filter, the optimum voltage standing wave ratio (VSWR) is 1.5 ± 0.1.
[0028]
In general, in a band-pass filter, when the unloaded Q (Qu) of the resonator used is low, the amount of attenuation is larger than in an ideal case in the pass band and the reflection band of the band-pass filter, and the curved portion is reduced. growing.
Therefore, in the present embodiment, the problem of no load Q (Qu) is solved by using a circular waveguide resonator or a dual mode dielectric resonator.
As a BPF (B1a, B2b), a dual-mode dielectric resonator type bandpass filter is used, and Qu≈38000, n = 4, B_wr≒ 5.6MHz, f_o= 600 MHz, f_∞≒ f_oThe characteristics calculated by the computer in the case of S = 1.4, S = 1.5, S = 1.6 as ± 3.35 MHz (596.65 MHz, 603.35 MHz) are shown in FIGS. .
8 shows a calculation result when S = 1.4, FIG. 9 shows a calculation result when S = 1.5, and FIG. 10 shows a calculation result when S = 1.6.
8 to 10, A1 is an attenuation characteristic, A2 is an enlarged passband region of the attenuation characteristic (A1), B is a return loss characteristic, and C is a group delay time characteristic.
[0029]
Actually, n = 4, B_wr≒ 5.6MHz, f_o= 485 MHz, f_∞≒ f_oExample of characteristics when a dual mode dielectric resonator type bandpass filter of ± 3.35 MHz (581.65 MHz, 488.35 MHz) is prototyped and the value of S is appropriately changed within 1.4 to 1.6 Is shown in FIGS.
11 and 12 are graphs showing examples of attenuation characteristics in the passband. In each figure, the horizontal axis is frequency (MHz), the memory interval is 1 MHz, and the vertical axis is attenuation (dB). The interval is 1 dB.
In FIG. 11, the attenuation when the frequency is 485 MHz (point 1 in the graph of FIG. 11) is −0.2297 dB, and the attenuation when the frequency is 482.2 MHz (point 2 in the graph of FIG. 11). , −0.3689 dB, and the amount of attenuation when the frequency is 487.8 MHz (point 3 in the graph of FIG. 11) is −0.2369 dB.
FIG. 12 is a graph showing an example of the amplitude deviation in the passband. In FIG. 12, the amount of attenuation when the frequency is 482.2 MHz (point 2 in the graph of FIG. 11) and the amount of attenuation are the smallest. (The point 1 in the graph of FIG. 12) and the difference from the attenuation amount are −0.2980 dB and the frequency is 487.8 MHz (the point 3 in the graph of FIG. 11). The difference between the small amount (point 1 in the graph of FIG. 12) and the amount of attenuation is −0.1911 dB.
As can be seen from FIGS. 11 and 12, in the BPF (B1a, B1b) of the present embodiment, the loss and ripple in the passband and the reflection loss and ripple in the reflection band can be reduced, and the passband Can be reduced (in the case of FIG. 12, within 0.3 dB).
[0030]
FIG. 13 is a graph showing an example of attenuation characteristics. In the figure, the horizontal axis is frequency (MHz), the memory interval is 3 MHz, the vertical axis is attenuation (dB), and the memory interval is 5 dB.
In FIG. 13, the attenuation when the frequency is 485 MHz (point 1 in the graph of FIG. 13) is −0.2280 dB, and the attenuation when the frequency is 482.2 MHz (point 2 in the graph of FIG. 13). When the frequency is 481.662 MHz (point 4 in the graph of FIG. 13), the attenuation is −26.947 dB and the frequency is 487.8 MHz (point 3 of the graph in FIG. 13). The attenuation is −0.2318 dB, and the attenuation when the frequency is 488.337 MHz (point 5 in the graph of FIG. 13) is −17.231 dB.
FIG. 14 is a graph showing an example of the return loss characteristics, in which the horizontal axis is frequency (MHz), the memory interval is 3 MHz, the vertical axis is attenuation (dB), and the memory interval is 1 dB. .
In FIG. 14, the attenuation when the frequency is 481 MHz (point 2 in the graph of FIG. 14) is −0.2478 dB, and the attenuation when the frequency is 481.662 MHz (point 4 in the graph of FIG. 14) is When the frequency is 488.337 MHz (point 5 in the graph of FIG. 14), the attenuation is −0.2729 dB and the frequency is 489.239 MHz (point 1 in the graph of FIG. 14). Is −0.3111 dB, and when the frequency is 491 MHz (point 3 in the graph of FIG. 14), the attenuation is −0.1829 dB.
[0031]
FIG. 15 is a diagram illustrating an example of attenuation characteristics and reflection attenuation amount characteristics, in which the horizontal axis is frequency (MHz), the memory interval is 3 MHz, the vertical axis is attenuation (dB), and the memory interval is 5 dB.
In the attenuation characteristic shown in FIG. 15, when the frequency is 485 MHz (point 1 in the graph of A in FIG. 15), the attenuation is −0.2719 dB, and the frequency is 482.2 MHz (of 2 in the graph of A in FIG. 15). In the case of (point), the attenuation is −0.4096 dB, and the frequency is 481.662 MHz (point 4 in the graph of FIG. 15A), the attenuation is −27.056 dB, and the frequency is 487.8 MHz (FIG. When the frequency is 488.337 MHz (point 5 in the graph of A in FIG. 15), the attenuation is −17.06 dB. It is.
Further, in the reflection attenuation characteristics shown in FIG. 15, when the frequency is 485 MHz (point 1 in the graph of B in FIG. 15), the attenuation is −14.103 dB, and the frequency is 482.2 MHz (the graph of B in FIG. 15). When the frequency is 481.662 MHz (point 4 in the graph of B in FIG. 15), the attenuation is -0.2409 dB and the frequency is 487. The amount of attenuation at 8 MHz (point 3 in the graph of B in FIG. 15) is −19.775 dB, and the amount of attenuation when the frequency is 488.337 MHz (point 5 in the graph of B in FIG. 15) is − 0.2619 dB.
[0032]
Hereinafter, an example of a band pass filter used as the BPF (B1a, B1b) of the present embodiment will be described.
FIG. 16 is a diagram showing a schematic configuration of an example of a dual mode dielectric resonator type BPF that can be used as the BPF (B1a, B1b) of the present embodiment, and FIG. 17 (a) is shown in FIG. FIG. 17B is a cross-sectional view of the main part showing a cross section taken along the line AA ′ shown in FIG. 17, and FIG. 17B is a cross-sectional view of the main part showing a cross section taken along the line BB ′ shown in FIG. ] Is principal part sectional drawing which shows the cross section along CC 'cut line shown in FIG. 16 and 17, 1 is TE.₁₁An outer conductor made of a cut-off waveguide, 2 is an inner partition (coupled iris), 3 is an input (or output) terminal, 4 is an output (or input) terminal, 5a and 5b are V-mode frequency adjusting screws, 6a and 6b Is an H-mode frequency adjusting screw, 7a and 7b are coupling adjusting screws, 12a is a capacitive coupling window formed in the inner partition 2, 12b is an inductive coupling window formed in the inner partition 2, and 14 and 15 are input / output couplings. The probes 20a and 20b are dual mode dielectric resonant elements.
[0033]
In the present embodiment, a dual mode dielectric resonator type BPF having the dual mode dielectric resonator shown in FIG. 18 can be used.
FIG. 18 is a cross-sectional view showing a cross-sectional structure of a dual mode dielectric resonator element as another example of a dual mode dielectric resonator type BPF that can be used as the BPF (B1a, B1b) of the present embodiment.
This figure shows a cross section taken along the line A-A 'shown in FIG.
The dual mode dielectric resonator shown in FIG. 18 has an E of the dual mode dielectric resonator 20a.₁Both ends of the electric field direction of E₁Is cut at right angles to the electric field direction of the dual-mode dielectric resonant element 20a.₁E at both ends of the electric field direction₁A cut surface perpendicular to the electric field direction is formed.
That is, the dual-mode dielectric resonator element 20a shown in FIG. 18 has two arc-shaped portions (10a, 10b) in which the outer shape of a cross section taken along a plane orthogonal to the axial direction of the cylindrical outer conductor 1 is opposed to each other. And linear portions (11a, 11b) facing each other, and the arc-shaped portions (10a, 10b) and the linear portions (11a, 11b) are continuous.
[0034]
In the dual mode dielectric resonator type BPF having the dual mode dielectric resonator shown in FIG. 18, the electrical length of the dielectric resonator 20a is E₁Becomes shorter in the electric field direction of E₂In the direction of the electric field, it is slightly shorter or hardly shortened.
For this reason, in the dual mode dielectric resonator type BPF having the dual mode dielectric resonator shown in FIG. 18, the capacitance value of the capacitance of the H mode resonance circuit of the first dielectric resonator is shown in FIGS. It becomes possible to make it larger than the BPF shown.
As a result, the insertion length of the frequency adjusting screw (6a) can be made smaller than before, and high-power OFDM waves (orthogonal frequency division multiplex (OFDM) modulation modulation waves) can be obtained. Even if it is applied, dielectric breakdown does not occur at this portion, so that it is possible to improve withstand voltage (power resistance) characteristics.
In the above description, the dual mode dielectric resonator 20a has been described. However, the dual mode dielectric resonator 20b has the same configuration.
[0035]
FIG. 19 is a diagram showing a schematic configuration of an example of a circular waveguide BPF that can be used as the BPF (B1a, B1b) of the present embodiment, and FIG. FIG. 20B is a main part sectional view showing a section along the line BB ′ shown in FIG. 19.
In the figure, 1 is TE.₁₁An outer conductor made of a cut-off waveguide, 2 is an inner partition (coupled iris), 3 is an input (or output) terminal, 4 is an output (or input) terminal, 5a and 5b are V-mode frequency adjusting screws, 6a and 6b Is an H-mode frequency adjusting screw, 7a and 7b are coupling adjusting screws, 12a is a capacitive coupling window formed in the inner partition wall 2, and 12b is an inductive coupling window formed in the inner partition wall 2.
FIG. 21 is a diagram showing a schematic configuration of an example of a rectangular waveguide type BPF that can be used as the BPF (B1a, B1b) of the present embodiment. FIG. 21 (a) is an upper plan view, FIG. b) is a cross-sectional view of the main part showing a cross section taken along the line AA ′ shown in FIG. 4A, and FIG. 4C is taken along the line BB ′ shown in FIG. FIG.
In FIG. 21, 11 is an external conductor, 2 is a partition, 3 is an input (or output) terminal, 4 is an output (or input) terminal, 8 is an input (or output) coupling loop element, and 9 is an output (or input) coupling. The loop element, 23 is a capacitive coupling window, 24 is an interstage coupling hole, and 25a to 25d are frequency adjusting screws.
[0036]
FIG. 22 shows a TM that can be used as the BPF (B1a, B1b) of the present embodiment._{01 δ}It is a figure which shows schematic structure of an example of a mode dielectric resonator type | mold BPF, and is principal part sectional drawing which shows the cross-sectional structure along the direction orthogonal to the axial length direction of a dielectric resonance element.
23A is a cross-sectional view of the main part showing a cross section taken along the line BB ′ shown in FIG. 22, and FIG. 23B is a cross section taken along the line AA ′ shown in FIG. It is a principal part sectional view shown.
22 and 23, 11 is an external conductor, 2 is a partition, 3 is an input (or output) terminal, 4 is an output (or input) terminal, 8 is an input (or output) coupling loop element, and 9 is an output (or) Input) Coupling loop element, 16 is an S-shaped coupling loop element, 24 is an interstage coupling hole, and 40a to 40d are TM_{01 δ}Mode dielectric resonators, 41a to 41d are dielectric resonator elements, 51a to 51c are interstage magnetic field coupling adjusting elements, and 56 is a coupling adjusting screw.
As shown in FIG. 23, the space between the first dielectric resonator element (41a) and the fourth dielectric resonator element (41d) is provided in the axial length direction of the dielectric resonator element and penetrates the partition wall 2. With the S-shaped coupled loop element 16 whose both ends are electrically and mechanically connected to the partition wall 2 at different positions in the axial length direction of the dielectric resonance element of the partition wall 2 from the boundary portion. Since they are coupled, the sub-coupling between them becomes capacitive coupling.
[0037]
FIG. 24 shows TE usable as the BPF (B1a, B1b) of the present embodiment._{01 δ}It is a figure which shows schematic structure of an example of a mode dielectric resonator type | mold BPF, The figure (a) is an upper top view, The figure (b) is an AA 'cut line shown to the figure (a). The principal part sectional view which shows the section which followed, and the figure (c) are principal part sectional views which show the cross section along the BB 'cutting line shown to the figure (a).
In FIG. 24, 11 is an external conductor, 2 is a partition, 3 is an input (or output) terminal, 4 is an output (or input) terminal, 8 is an input (or output) coupling loop element, and 9 is an output (or input) coupling. Loop element, 16 is an S-shaped coupling loop element, 24 is an interstage coupling hole, 25a to 25d are frequency adjusting screws, 42a to 42d are dielectric resonance elements, 43a to 43c are support members, and 56a to 56c are interstages It is a coupling adjustment screw.
As shown in FIG. 24, between the first dielectric resonator element (42a) and the fourth dielectric resonator element (42d) is provided in a direction orthogonal to the axial length direction of the dielectric resonator element, S-shaped in which both end portions are electrically and mechanically connected to the partition wall 2 at different positions in the direction perpendicular to the axial length direction of the dielectric resonance element of the partition wall 2 with a portion penetrating the partition wall 2 as a boundary. Since the sub-coupled loop elements 16 are sub-coupled, the sub-coupling between them is capacitive coupling.
[0038]
FIG. 25 is a diagram showing a schematic configuration of an example of a polarized interdigital BPF using a coaxial resonator that can be used as the BPF (B1a, B1b) of the present embodiment. FIG. 4B is a cross-sectional view of the main part showing a cross section taken along the line AA ′ shown in FIG. 1A, and FIG. It is principal part sectional drawing which shows the cross section along a BB 'cutting line.
In FIG. 25, 11 is an external conductor, 2 is a partition, 3 is an input (or output) terminal, 4 is an output (or input) terminal, 8 is an input (or output) coupling loop element, and 9 is an output (or input) coupling. The loop element, 16 is an S-shaped coupling loop element, and 30a to 30d are resonance bars.
Also in FIG. 25, both end portions are provided in the axial length direction of the dielectric resonance element, and at both ends thereof at different positions in the axial length direction of the dielectric resonance element of the partition wall 2 with the portion penetrating the partition wall 2 as a boundary. The sub coupling circuit of the S-shaped coupling loop element 16 electrically and mechanically connected to the partition wall 2 is capacitive coupling.
[0039]
As described above, according to the antenna sharing apparatus of the present embodiment, it is possible to synthesize the transmission waves of adjacent channels and reduce the insertion loss and ripple of each channel transmission wave supplied to the antenna (ANT). In addition, the amplitude deviation in the pass band can be reduced.
For example, in the case of digital television broadcasting, for example, an amplitude deviation within a passband of 0.5 dB or less is obtained due to restrictions on bit error rate characteristics. In the antenna sharing device of the present embodiment, The amplitude deviation in the passband within 0.5 dB can be easily realized.
Although the invention made by the present inventor has been specifically described based on the above-described embodiment, the present invention is not limited to the above-described embodiment, and various modifications can be made without departing from the scope of the invention. Of course.
[0040]
【The invention's effect】
The effects obtained by the representative ones of the inventions disclosed in the present application will be briefly described as follows.
According to the present invention, it is possible to reduce the insertion loss and ripple of each transmission wave and reduce the amplitude deviation in the passband of each transmission wave in the antenna sharing device that supplies the transmission wave of the adjacent channel to the antenna. It becomes.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a schematic configuration of an antenna sharing device according to an embodiment of the present invention.
FIG. 2 is a diagram showing an example of transmission characteristics of the antenna sharing device of the present embodiment.
FIG. 3 is a diagram for explaining a hybrid circuit including a λ / 4 · 3 dB coupler.
FIG. 4 shows a terminal (T) of a hybrid circuit (H1) in the constant impedance bandpass filter according to the embodiment of the present invention.₁₁) Is a diagram for explaining the output voltage of each part when the voltage of Ein1 is input.
FIG. 5 shows a terminal (T) of the hybrid circuit (H2) in the constant impedance band-pass filter according to the embodiment of the present invention._{twenty two}) Is a diagram for explaining the output voltage of each part when the voltage of Ein2 is input.
FIG. 6 is a diagram for explaining a relationship between a voltage standing wave ratio (VSWR) of a band-pass filter and a compensation attenuation amount (Amin) in an attenuation region.
FIG. 7 is a graph showing a relationship between a voltage standing wave ratio (VSWR) of a band pass filter, a reflection loss in a pass band, and a reflection loss in an attenuation region.
FIG. 8 is a graph showing characteristics calculated by a computer when S = 1.4 using a dual-mode dielectric resonator type band-pass filter as the band-pass filter (B1a, B2b) of the present embodiment; It is.
FIG. 9 is a graph showing characteristics calculated by a computer when S = 1.5 using a dual-mode dielectric resonator type band-pass filter as the band-pass filter (B1a, B2b) of the present embodiment; It is.
FIG. 10 is a graph showing characteristics calculated by a computer when S = 1.6 using dual-mode dielectric resonator type band-pass filters as band-pass filters (B1a, B2b) of the present embodiment; It is.
FIG. 11 is a graph showing an example of a characteristic example when a dual-mode dielectric resonator type band-pass filter is actually manufactured and the value of S is appropriately changed within 1.4 to 1.6.
FIG. 12 is a graph showing another example of a characteristic example when a dual-mode dielectric resonator type bandpass filter is actually manufactured and the value of S is appropriately changed within 1.4 to 1.6. is there.
FIG. 13 is a graph showing another example of a characteristic example when a dual-mode dielectric resonator type band-pass filter is actually manufactured and the value of S is appropriately changed within 1.4 to 1.6. is there.
FIG. 14 is a graph showing another example of a characteristic example when a dual-mode dielectric resonator type band-pass filter is actually prototyped and the value of S is appropriately changed within 1.4 to 1.6. is there.
FIG. 15 is a graph showing another example of a characteristic example when a dual-mode dielectric resonator type band-pass filter is actually manufactured and the value of S is appropriately changed within 1.4 to 1.6. is there.
FIG. 16 is a diagram showing a schematic configuration of an example of a dual-mode dielectric resonator type band-pass filter that can be used as the band-pass filters (B1a, B1b) according to the embodiment of the present invention.
17 is a cross-sectional view of a principal part showing a cross section taken along each cutting line shown in FIG. 16;
18 shows a cross-sectional structure of a dual-mode dielectric resonator element as another example of a dual-mode dielectric resonator type band-pass filter that can be used as the band-pass filters (B1a, B1b) according to the embodiment of the present invention. FIG. It is sectional drawing.
FIG. 19 is a diagram showing a schematic configuration of an example of a circular waveguide band-pass filter that can be used as the band-pass filters (B1a, B1b) according to the embodiment of the present invention.
20 is a main-portion cross-sectional view showing a cross section taken along each cutting line shown in FIG. 19;
FIG. 21 is a diagram showing a schematic configuration of an example of a rectangular waveguide bandpass filter that can be used as the bandpass filters (B1a, B1b) according to the embodiment of the present invention.
FIG. 22 shows a TM that can be used as a bandpass filter (B1a, B1b) according to an embodiment of the present invention._{01 δ}It is a figure which shows schematic structure of an example of a mode dielectric resonator type | mold band pass filter.
23 is a cross-sectional view of a principal part showing a cross section taken along each cutting line shown in FIG. 22;
FIG. 24 shows a TE that can be used as a bandpass filter (B1a, B1b) according to an embodiment of the present invention._{01 δ}It is a figure which shows schematic structure of an example of a mode dielectric resonator type | mold band pass filter.
FIG. 25 is a diagram showing a schematic configuration of an example of a polarized interdigital bandpass filter using a coaxial resonator that can be used as the bandpass filters (B1a, B1b) according to the embodiment of the present invention.
FIG. 26 is a diagram illustrating an example of a channel arrangement used in a conventional analog television broadcast.
FIG. 27 is a block diagram showing a schematic configuration of a conventional antenna sharing apparatus in the case where the use channel is the arrangement shown in FIG.
FIG. 28 is a graph showing transmission characteristics of the conventional antenna sharing apparatus shown in FIG. 27;
[Fig. 29] Fig. 29 is a diagram illustrating an example of a used channel arrangement in television broadcasting using adjacent channels.
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 1,11 ... External conductor, 2 ... Bulkhead, 3 ... Input (or output) terminal, 4 ... Output (or input) terminal, 5a, 5b, 6a, 6b, 25a-25d ... Frequency adjustment screw, 7a, 7b, 56 , 56a to 56c ... coupling adjustment screw, 8 ... input (or output) coupling loop element, 9 ... output (or input) coupling loop element, 12a, 12b, 23 ... coupling window, 14, 15 ... input / output coupling probe, 16 ... S-shaped coupling loop element, 20a, 20b, 41a to 41d, 42a to 42d ... Dielectric resonant element, 24 ... Interstage coupling hole, 30a to 30d ... Resonant rod, 40a to 40d ... TM_{01 δ}Mode dielectric resonators, 43a to 43c ... inductive support members, 51a to 51c ... interstage magnetic field coupling adjusting element ANT ... antenna, R ... non-reflection termination, H1, H2, H13a, H13b, H14a, H14b, H15a, H15b , H22a, H22b, H23a, H23b... Hybrid circuit, B1a, B1b, B13a, B13b, B14a, B14b, B15a, B15b, B22a, B22b, B23a, B23b.

Claims (2)

It has a plurality of constant impedance band pass filters connected in cascade,
Each of the constant-impedance bandpass filters includes first and second hybrid circuits in which the outputs of the second and third terminals have a phase difference of 90 ° from each other, and have electrical characteristics that are substantially equal in amplitude,
A first bandpass filter connected between a first terminal of the first hybrid circuit and a second terminal of the second hybrid circuit;
A second bandpass filter connected between a fourth terminal of the first hybrid circuit and a third terminal of the second hybrid circuit;
Each of the constant impedance bandpass filters is an antenna sharing device in which the second terminal and the third terminal of the first hybrid circuit of adjacent constant impedance bandpass filters are connected to each other and cascaded,
The bandpass filter of each constant impedance bandpass filter is a quaternary polarized dual mode dielectric resonator type bandpass filter,
The dual mode dielectric resonator type bandpass filter includes a cylindrical outer conductor,
Two dual-mode dielectric resonant elements provided in the arc-shaped outer conductor,
Each of the dual mode dielectric resonator elements has an outer shape of a cross section cut by a plane orthogonal to the axial direction of the cylindrical outer conductor, two arc-shaped portions facing each other, and straight portions facing each other. And
Each arc-shaped portion and each linear portion are continuous,
In the dual mode dielectric resonator type bandpass filter, the center frequency of the filter is f _o , the allowable pass frequency bandwidth is B _wr , the frequency of the attenuation pole is f _∽ , and the voltage standing wave ratio in the pass band is VSWR. An antenna sharing device that satisfies the following formula.
B _wr = 5.6 ± 0.1 MHz
_{f ∽ = f o ± (3.35} ± 0.1MHz)
VSWR = 1.5 ± 0.1   2. The antenna sharing device according to claim 1, wherein the dual mode dielectric resonator type band pass filter has an amplitude deviation within 0.5 dB within a pass band. 3.

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