JP2000125559A - Voltage-resonance type switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To miniaturize and make light the circuit of a voltage-resonance-type switching power supply circuit and to reduce cost. SOLUTION: A parallel resonance capacitor Cr is connected between the collector of a switching element Q1 and an Ei line, thus reducing a resonance voltage Vcr and hence forming a parallel resonance capacitor Cr from two series connections. Also, by connecting a secondary side parallel resonance capacitor C2, in parallel with a secondary side rectification diode, radiation noise generated at the inverse recovery time of a rectifying diode is absorbed.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a voltage resonance type switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a so-called soft switching power supply circuit, a switching power supply circuit having a voltage resonance type switching converter is known. The voltage resonance type switching converter has a low noise because a smooth waveform is obtained for the switching output voltage pulse and the switching output current flowing into the insulating converter transformer, and can be configured with a relatively small number of parts.

FIG. 7 is a circuit diagram showing an example of a voltage resonance type switching power supply circuit which can be constructed based on the invention proposed by the present applicant. In the switching power supply circuit shown in this figure, for example, a commercial AC power supply in Japan or the United States is a so-called AC 100 V system, and corresponds to a condition where the maximum load power is 150 W or more.

The switching power supply circuit shown in FIG. 1 is provided with a so-called voltage doubler rectification circuit comprising rectification diodes Di1, Di2 and smoothing capacitors Ci1, Ci2 as a rectification and smoothing circuit for rectifying and smoothing the AC power supply AC. Can be In this voltage doubler rectifier circuit, assuming that a DC input voltage corresponding to, for example, one time the peak value of the AC input voltage VAC is Ei, a DC input voltage 2Ei that is approximately twice that of the DC input voltage Ei.
Generate For example, if the AC input voltage VAC is 144V, the DC input voltage 2Ei is about 400V. As described above, the reason why the voltage doubler rectifier circuit is employed as the rectifier / smoothing circuit is that the AC input voltage is 100 V AC as described above.
The system is designed to cope with a relatively heavy load condition in which the maximum load power is 150 W or more. That is, by increasing the DC input voltage to twice the normal value, the amount of current flowing into the subsequent switching converter is suppressed, and the reliability of the components forming the switching power supply circuit is ensured. . In the voltage doubler rectifier circuit shown in this figure, an inrush current limiting resistor Ri is inserted in the rectified current path, for example, so as to suppress the inrush current flowing into the smoothing capacitor when the power is turned on. I have.

The voltage-resonant type switching converter shown in FIG. 1 has a self-excited type provided with a single switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT;
Junction type transistor). The base of the switching element Q1 is connected to the positive side of the smoothing capacitor Ci1 (rectified smoothed voltage 2Ei) via a starting resistor RS so that a base current at the time of starting can be obtained from the rectified smoothing line. A self-oscillation resonance circuit including a damping resistor RB, an inductor LB, a resonance capacitor CB, and a drive winding NB is connected in series between the base of the switching element Q1 and the temporary ground. In this case, the drive winding NB is connected to an insulation converter transformer PIT (Power Isol
Transformer), so that the required inductance for setting the switching frequency can be obtained together with the inductor LB in the resonance circuit for self-excited oscillation. A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a damper current flowing when the switching element Q1 is off. , The collector of switching element Q1 is connected to one end of primary winding N1 of insulating converter transformer PIT, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Is a voltage based on its own capacitance and a combined inductance (L1 + LR) obtained by connecting a primary winding N1 of an isolated converter transformer PIT described later and a controlled winding NR of a quadrature control transformer PRT (Power Regulating Transformer) in series. A parallel resonant circuit of the resonant converter is formed. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage Vd across the resonance capacitor Cr is obtained by the action of the parallel resonance circuit.
cr is actually a sinusoidal pulse waveform so that a voltage resonance type operation can be obtained.

In practice, the parallel resonance capacitor Cr is actually composed of two capacitors CrA, made of a polypropylene film capacitor having a low loss with respect to a high-frequency current.
The CrBs are connected in series and are configured as a set of package components. The reason for this will be described later.

The insulated converter transformer PIT is for transmitting the switching output of the switching element Q1 to the secondary side. In this case, the insulated converter transformer PIT is used.
One end of the primary winding N1 of T is connected to the collector of the switching element Q1, and the other end is connected in series with the controlled winding NR of the orthogonal control transformer PRT as shown in the figure.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, a parallel resonance circuit is formed by connecting the secondary side parallel resonance capacitor C2 to the secondary winding N2 in parallel. With this parallel resonance circuit,
The alternating voltage excited in the secondary winding N2 is a resonance voltage,
This resonance voltage is supplied to two sets of half-wave rectifier circuits, a half-wave rectifier circuit including a rectifier diode DO1 and a smoothing capacitor CO2, and a half-wave rectifier circuit including a rectifier diode DO2 and a smoothing capacitor CO2. Then, the DC output voltages EO1 and EO2 are obtained by these two sets of half-wave rectifier circuits, respectively. At this time, since the resonance circuit is formed by providing the secondary side parallel resonance capacitor C2 as described above, the resonance operation causes the amount of current flowing to the secondary side to increase. As a result, the maximum load power that can be handled is increased. The rectifier diodes DO1 and DO2 forming the half-wave rectifier circuit use a high-speed rectifier to rectify the alternating voltage in the switching cycle.

The control circuit 1 compares, for example, a DC voltage output on the secondary side with a reference voltage and outputs a DC current corresponding to the error.
The control winding N of the orthogonal control transformer PRT is used as the control current.
This is the error amplifier that supplies C. In this case, the DC voltage output EO1 is input to the control circuit 1 as the detection voltage, and the DC output voltage EO2 is input as the operation power supply.

The orthogonal control transformer PRT has a control winding N
The controlled winding NR is wound so that its winding direction is orthogonal to C, and a gap G is formed in a required portion of the core magnetic leg to form a saturable reactor. It is configured as follows.

For example, when the DC output voltage EO2 on the secondary side fluctuates with the fluctuation of the AC input voltage VAC or the minimum load power, the control circuit 1 changes the control current flowing through the control winding NC within a required range. Let it. Thus, the orthogonal control transformer PRT operates so that the inductance LR of the controlled winding NR changes within a predetermined range.

Since the controlled winding NR forms a parallel resonance circuit for obtaining a voltage-resonant type switching operation as described above, the parallel resonance circuit NR is controlled with respect to a fixed switching frequency. The resonance condition of the circuit is changed. At both ends of the parallel connection circuit of the switching element Q1 and the parallel resonance capacitor Cr, a sinusoidal resonance pulse is generated by the action of the parallel resonance circuit corresponding to the OFF period of the switching element Q1, but the resonance condition of the parallel resonance circuit , The width of the resonance pulse is variably controlled. That is, PWM (P
(ulse Width Moduration) control operation is obtained. The PWM control of the width of the resonance pulse means the control of the off period of the switching element Q1, which in other words means variably controlling the on period of the switching element Q1 under the condition of a fixed switching frequency. . By variably controlling the ON period of the switching element Q1 in this manner, the switching output transmitted from the primary winding N1 forming the parallel resonance circuit to the secondary side changes, and the DC output voltage ( The output levels of EO1, EO2) are also changed. As a result, the secondary DC voltage (EO1, EO2) can be made constant. Note that such a constant voltage control method is hereinafter referred to as an inductance control method.

If the power supply circuit having the configuration shown in FIG. 7 is adapted to an AC 200 V area such as Europe, for example, a rectifier circuit for generating a rectified smoothed voltage may be replaced with a bridge voltage rectifier circuit instead of a doubler rectifier circuit. A full-wave rectifier circuit including a rectifier circuit may be provided to generate a rectified smoothed voltage corresponding to an equal multiple of the AC input voltage VAC.

FIG. 8 shows the switching cycle of the power supply circuit having the configuration shown in FIG. 7 when the load is short-circuited at an AC input voltage VAC = 144 V, a rectified smoothed voltage (DC input voltage) Ei = 400 V. 5 shows the operation waveforms of the main parts in FIG.

Under this condition, as the primary-side parallel resonance circuit, the action of the capacitance of the parallel resonance capacitor Cr, the inductance L1 of the primary winding N1, and the inductance LR of the controlled winding NR whose variable is in a controlled state is performed. Thereby, a stable voltage resonance type switching operation can be obtained. For this reason, as shown in FIG. 8A, a sinusoidal wave is applied to both ends of the parallel connection circuit of the switching element Q1 // parallel resonance capacitor Cr when the switching element Q1 is off, as shown in FIG. Is obtained as the resonance voltage Vcr. This resonance voltage Vcr is
The peak level is about 1800V. Also, during the period when the switching element Q1 is off, the resonance current Icr flows through the parallel resonance capacitor Cr as shown in FIG. 8B. On the secondary side, the voltage (resonance voltage) VC2 across the secondary side parallel resonance capacitor C2 is, as shown in FIG. 8 (c), the DC output voltage during the period when the rectifier diodes DO1, DO2 are conducting. During the period in which the rectifier diodes DO1 and DO2 are non-conductive, a sine wave-shaped 2 is generated by the resonance operation of the secondary side parallel resonance circuit.
The waveform has a peak level of 85V. Therefore, if the DC output voltage EO is about 135 V, the peak to peak level of the resonance voltage VC2 is 420 Vp-p.
About. As shown in FIG. 8D, the resonance current IC2 flowing through the secondary parallel resonance capacitor C2 changes from a positive level to a negative level by a sinusoidal curve during a period in which the rectifier diodes DO1 and DO2 are non-conductive. An inverted waveform is obtained.

[0017]

As can be seen from the waveform shown in FIG. 8A, a maximum voltage of 1800 V is applied to the switching element Q1 and the parallel resonance capacitor Cr. For this reason, it is necessary to use a withstand voltage product of 1800 V for the switching element Q1 and the parallel resonance capacitor Cr. However, regarding the parallel resonance capacitor Cr,
In order to give low-loss characteristics to high-frequency resonance currents, for example, a so-called polypropylene film capacitor is used in which, for example, the derivative is a polypropylene film, the electrodes are metallized polyester films, and the exterior is made of flame-retardant epoxy resin. I am trying to do it. This polypropylene film capacitor is currently at the limit where a withstand voltage of 1600 V can be manufactured. Therefore, as shown in FIG. 7, for example, as an actual parallel resonance capacitor Cr, two polypropylene film capacitors (CrA, CrB) of 0.02 μF / 1000 V are connected in series, and these two capacitors are made of plastic. 0.01μF / 2 by molding with epoxy resin in the case
It is configured as a single 000 V capacitor. In this case, for example, in order to obtain the capacitance of 0.01 μF required as the parallel resonance capacitor Cr, two capacitors which are increased in size according to the doubled capacitance are required. The size of the capacitor Cr is increased. For example, the parallel resonance capacitor C configured as described above
The external dimension of r is about 13 × 22 × 32 mm.

As described with reference to FIG. 7, on the secondary side, the high-frequency alternating voltage due to the switching cycle is changed to a high withstand voltage, high-speed recovery type rectifier diode (DO).
Although a DC output voltage is obtained by rectification by a half-wave rectifier circuit provided with (1, DO2), unnecessary radiation noise is generated during the reverse recovery time of the rectifier diode. For this reason,
In practice, a rectifier diode (D
O1, DO2), a small-capacity ceramic capacitor C
By connecting p and Cp in parallel, the radiation noise is absorbed.

As for the switching power supply circuit, it is preferable to reduce the number of component elements and reduce the size of the elements themselves, thereby promoting reduction in size and weight. As described above, the circuit configuration shown in FIG. In the above, there is a limit in reducing the size and weight of the circuit board due to the increase in the size of the primary-side parallel resonance capacitor Cr and the addition of capacitors Cp and Cp on the secondary side.

[0020]

SUMMARY OF THE INVENTION In view of the above-mentioned problems, the present invention provides a rectifying / smoothing in which a commercial AC power is input and rectified and smoothed to generate a rectified smoothed voltage and output as a DC input voltage. Means, switching means for intermittently outputting a DC input voltage to output to the primary winding of the insulating converter transformer, at least an inductance component including the primary winding of the insulating converter transformer, and an output terminal of the switching means. A primary-side resonance circuit formed by the capacitance of a resonance capacitor inserted between the positive lines of the DC input voltage to make the operation of the switching means a voltage resonance type; a secondary-side rectifier diode element and a secondary-side smoothing capacitor , The secondary DC output voltage from the alternating voltage obtained in the secondary winding of the isolated converter transformer A DC output voltage generating means for generating for,
A secondary resonance circuit formed on the secondary side by an inductance component of a secondary winding of the insulating converter transformer and a capacitance of a secondary parallel resonance capacitor connected in parallel to the secondary rectifier diode element; Orthogonal control in which a controlled winding provided so as to function as an inductance component of a primary side resonance circuit and a control winding whose winding direction is orthogonal to the controlled winding are wound. A constant current control for the secondary-side DC output voltage by providing a variable control current to the control winding according to the level of the DC output voltage and changing the inductance of the controlled winding. A voltage-resonant switching power supply circuit is configured to include the configured constant voltage control means.

According to the above configuration, in the voltage resonance type converter, the resonance capacitor of the primary side resonance circuit for making the switching operation a voltage resonance type is provided between the positive line of the DC input voltage and the switching output terminal of the switching means. As a result, the reference potential of the resonance voltage generated at both ends of the resonance capacitor due to the operation of the primary side resonance circuit is shifted by the DC input voltage level.

[0022]

DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 1 is a circuit diagram showing a configuration of a power supply circuit according to an embodiment of the present invention. In FIG. The description of is omitted. In the power supply circuit shown in this figure, the parallel resonance capacitor Cr on the primary side is
It is provided so as to be inserted between the positive electrode (rectified smoothing voltage (DC input voltage) line) of the smoothing capacitor Ci1 and the collector of the switching element. In such a connection form, one end of the parallel resonance capacitor Cr can be regarded as being connected to the primary side ground through the series connection of the smoothing capacitors Ci1 to Ci2. And the combined inductance (L1 + LR) obtained by connecting the primary winding N1 and the controlled winding NR in series to form a parallel resonance circuit of the voltage resonance type converter.

On the secondary side, the secondary parallel resonance capacitor C2 is connected in parallel with the rectifier diode DO1.

FIG. 2 shows an operation waveform diagram of a main part according to the switching cycle of the power supply circuit having such a configuration. FIG.
(A) is the collector-emitter voltage VCP of the switching element Q1, FIG. 2 (b) is the resonance voltage Vcr obtained across the parallel resonance capacitor Cr, and FIG. 2 (c) is the secondary-side parallel resonance capacitor C2. The figure shows the resonance voltage VC2. In this case, the resonance voltage Vcr shown in FIG. 2B has a waveform that becomes a sinusoidal voltage resonance pulse when the switching element Q1 is off due to the action of the primary side parallel resonance circuit formed as described above. In addition, the resonance voltage VC2 shown in FIG. 2C is such that the secondary-side parallel resonance capacitor C2 becomes the smoothing capacitor CO during the period when the rectifier diode DO1 is turned off.
1 is connected to the secondary side ground through the secondary winding N2.
Is a waveform in which a sinusoidal voltage resonance pulse is obtained by the secondary side parallel resonance circuit formed together with the inductance L2 of FIG.

The collector-emitter voltage VCP of the switching element Q1 shown in FIG. 2A is the same as the resonance voltage Vcr shown in FIG.
In the off state, a peak level of 1800 V appears. On the other hand, the resonance voltage Vcr shown in FIG.
Is offset from the zero point of the reference potential at the level of 2Ei of the rectified smoothed voltage. Therefore, the peak level of the resonance voltage Vcr when the switching element Q1 is off is 14
It drops to 00V (= 1800V-400V). The offset of the reference potential 0 point is an operation obtained when one end of the parallel resonance capacitor Cr is connected to the positive electrode side (the positive electrode terminal of the smoothing capacitor Ci1) of the series connection of the smoothing capacitors Ci1 and Ci2.

For this reason, in the present embodiment, the parallel resonance capacitor Cr only needs to withstand a voltage of 1400 V. As described above, the withstand voltage of 1600 V is the manufacturing limit of the polypropylene film capacitor. In the present embodiment, the parallel resonance capacitor Cr actually mounted on the circuit board is, for example, Cr = 0.01 μF / If a material having a characteristic of 1600 V is selected, the withstand voltage is sufficient, and only one polypropylene film capacitor is required.
At this time, the capacitance of the parallel resonance capacitor Cr becomes
By setting to 0.01 μF as described above, for example, FIG.
Compared with the case where the polypropylene film capacitors C2A and C2B provided in the circuit shown in FIG.
r is considerably small. In particular,
0.01μF / 1600V (DC) parallel resonance capacitor C
The external dimension of r is 6 × 15 × 26.5 mm, which is about 1/4 the size of the parallel resonance capacitor Cr shown in FIG.

Also, by connecting the secondary side parallel resonance capacitor C2 in parallel with the rectifier diode DO1,
The resonance voltage VC2 of the secondary parallel resonance capacitor C2 is shown in FIG.
As shown in (c), the level during which the rectifier diode DO1 is turned on is substantially zero level, and the peak level of 285V is obtained during the period when the rectifier diode DO1 is turned off. Therefore, the Peak to Peak level is approximately 285 Vp-p. Compared to the circuit shown in FIG. For example, a polypropylene capacitor is used as the secondary side parallel resonance capacitor C2. However, at present, even if the resonance voltage VC2 becomes 285 Vp-p, a polypropylene capacitor having a withstand voltage of 400 V (DC) should be selected. Therefore, it is not possible to reduce the size much. However, for example, if a smaller polypropylene capacitor having a withstand voltage of about 300 V (DC) can be obtained, the size of the secondary-side parallel resonance capacitor C2 can be reduced. However, in the present embodiment, since the secondary parallel resonance capacitor C2 is connected in parallel to the rectifier diode DO1, the secondary parallel resonance capacitor C2 is generated during the reverse recovery time of the secondary rectifier diode. It will also work to absorb radiation noise. Therefore, in the present embodiment, the ceramic capacitor Cp shown in FIG. 7 can be omitted.

FIG. 3 is a circuit diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention. The same parts as those in FIG.

In the power supply circuit shown in FIG. 3, a full-wave rectification circuit including a bridge rectification circuit Di and a smoothing capacitor Ci is provided as a rectification and smoothing circuit for receiving an AC input voltage VAC to obtain an AC input voltage. AC input voltage V
A rectified and smoothed voltage Ei corresponding to a level of one time of AC is generated. That is, the present embodiment does not include a voltage doubler rectifier circuit as in the related art and the configuration of FIG. In this specification, a full-wave rectifier circuit that generates a rectified and smoothed voltage Ei corresponding to one time the level of the AC input voltage VAC is also referred to as a “single-voltage rectifier circuit”.

Further, in the insulation converter transformer PIT of the power supply circuit shown in FIG. 3, in addition to the primary winding N1, the secondary winding N2, and the driving winding NB, the winding N3 is wound so as to wind up the primary winding N1. Be provided. An end of the winding N3 is connected to a smoothing capacitor Ci for generating a boost voltage described later.
Connected to B positive electrode. The negative electrode of the smoothing capacitor CiB is connected to the positive electrode (Ei line) of the smoothing capacitor Ci.
Further, in the power supply circuit shown in this figure, a boost diode DB is provided. This boost diode DB
Has an anode connected to a connection point (Ei line) between a negative electrode of the smoothing capacitor CiB and a positive electrode of the smoothing capacitor Ci, and a cathode connected to the primary winding through a series connection of a controlled winding NR of the orthogonal control transformer PRT. It is connected to the connection point between N1 and winding N3. According to such a connection form, the switching output voltage obtained at the winding N3 is rectified by the boost diode DB and the smoothed capacitor CiB
As a result, a boost circuit that generates a boost voltage VB is formed across the smoothing capacitor CiB. However, as described above, the controlled winding NR is inserted in series in this boost circuit.

By providing the boost circuit, the voltage resonance type switching converter including the switching element Q1 operates the boost smoothed voltage EB obtained by superimposing the boosted voltage VB on the rectified smoothed voltage Ei. Switching is performed as (DC input voltage). In other words, when the boost circuit operates, the apparent DC input voltage level to be supplied to the voltage resonance type switching converter increases, and thus, for example, in a configuration in which the DC input voltage is obtained by voltage doubling rectification operation It is possible to increase the maximum load power that can be handled to the same extent as.

In the configuration having the boost circuit as described above, the boost smoothing voltage EB obtained by superimposing the boost voltage VB on the rectified smoothing voltage Ei is applied to both ends of the smoothing capacitor CiB and the smoothing capacitor Ci connected in series. As a result, the boost smoothing voltage EB is

(Equation 1) Can be represented by Then, the relationship of L3 = L1 is obtained as the inductance of the winding N3 and the primary winding N1, and the rectified smoothing voltage Ei, the drop voltage VF of the boost diode DB, and the saturation voltage V (SAT) of the switching element Q1 are obtained. Assuming that the relationship of Ei≫VF, V (SAT) holds, the boost smoothed voltage EB is calculated by the above (Equation 1)
On the basis of the,

(Equation 2) Will be indicated by In this case, in the power supply circuit shown in FIG. 3, for example, the inductance LR of the controlled winding NR of the orthogonal transformer PRT is set to 0.1 × L1 to 1.2 × L1.
, The boost smoothing voltage EB is substantially changed from Ei to 2Ei (Ei is the smoothing capacitor Ci).
(Corresponding to the level of the rectified and smoothed voltage obtained at both ends).

When the boost circuit is provided in this manner, the primary-side parallel resonance capacitor Cr of the present embodiment is provided.
Is connected between the positive electrode of the smoothing capacitor CiB of the boost circuit (the line of the boost smoothing voltage EB) and the collector of the switching element Q1. Even with such a connection form, the resonance voltage Vcr obtained at both ends of the parallel resonance capacitor Cr has a reference potential shifted by the boost smoothed voltage EB, so that a voltage lower than 1400 V can be obtained. It can be a book capacitor. That is, the configuration of the present embodiment can also reduce the size of the parallel resonance capacitor Cr.

On the secondary side of the insulating converter transformer PIT shown in FIG. 3, the secondary winding N2 is provided with a center tap, and the rectifier diodes DO1, DO2, DO3,
By connecting DO4 and the smoothing capacitors CO1 and CO2 as shown in the figure, two sets of [rectifying diodes DO1, DO2, smoothing capacitor CO1] and [rectifying diodes DO3, DO4, smoothing capacitor CO2] are provided. Is provided. A full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] generates a DC output voltage EO1, and generates a [rectifier diode DO3, DO4, smoothing capacitor CO1].
O2] generates a DC output voltage EO2. This secondary-side output voltage is supplied to a subsequent load (not shown).

Insulated converter transformer P of the present embodiment
As shown in FIG. 4, the IT is provided with an EE-type core in which E-type cores CR1 and CR2 made of, for example, a ferrite material are combined so that their magnetic legs face each other. , The primary winding N1 using the bobbin B
(And N3, NB) and the secondary winding N2 are wound separately. In the present embodiment, a gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained.

In the insulating converter transformer PIT, the primary winding N1 is determined by the relationship between the polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the connection of the rectifier diodes DO (DO1, DO2 / DO3, DO4). The mutual inductance M between the inductance L1 of the secondary winding N2 and the inductance L2 of the secondary winding N2 may be + M or -M. For example, when the connection configuration shown in FIG. 5A is employed, the mutual inductance is + M, and when the connection configuration shown in FIG. 5B is employed, the mutual inductance is -M.
If this is made to correspond to the operation on the secondary side of the present embodiment described above, for example, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, the operation in which the rectified current flows through the rectifier diodes DO1 and DO3. Can be regarded as a + M operation mode (forward mode). Conversely, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, the rectification current flowing through the rectification diodes DO2 and DO4 is the -M operation mode ( Flyback method). That is, in the present embodiment, each time the alternating voltage obtained in the secondary winding becomes positive / negative, the mutual inductance operates in the + M / -M mode.

When the full-wave rectifier circuit is provided on the secondary side as described above, the secondary-side parallel resonant capacitor according to the present embodiment is the same as the rectifier circuit for generating the DC output voltage E01. , The secondary-side parallel resonant capacitors C2A and C2B are connected to the rectifier diodes DO1 and DO2, respectively.
Are connected in parallel. Thus, when the rectifier diode DO1 is off, a parallel resonance circuit is formed by the secondary side parallel resonance capacitor C2A and the secondary winding N2 via the smoothing capacitor CO1, and when the rectifier diode DO2 is off, the parallel resonance circuit is provided via the smoothing capacitor CO2. Thus, a parallel resonance circuit is formed by the secondary side parallel resonance capacitor C2B and the secondary winding N2. With such a connection form, similarly to the previous embodiment, the secondary-side parallel resonance capacitor C2
The generation of radiation noise during the reverse recovery time of the rectifier diode on the secondary side is suppressed by A and C2B, so that there is no need to provide a ceramic capacitor Cp.

When the secondary side parallel resonance circuit is formed by providing the secondary side parallel resonance capacitors C2A and C2B as in the power supply circuit shown in FIG. As described above, the maximum load power increases as compared with the case where the secondary side parallel resonance capacitor is not provided. In addition to this,
When a full-wave rectifier circuit is connected to the secondary-side parallel resonance circuit as in the circuit shown in this figure, the rectification current is alternately changed in both the operation modes in which the mutual inductance is + M / -M, as described above. Is made to flow. In other words, since the rectified output is obtained during the period in which the alternating voltage is both the positive electrode and the negative electrode, the power supplied to the load increases accordingly, and the rate of increase of the maximum load power also improves.

In this case, the control circuit 1 supplies a control current of a level corresponding to the fluctuation of the DC output voltage EO1 to the control winding NC of the orthogonal transformer PRT so that the DC output voltage EO1 is constant. Thus, the operation is performed to vary the inductance LR of the controlled winding NR. Thus, the constant voltage control by the above-described inductance control method is performed. In this case, the boosted smoothing voltage EB represented by (Equation 1) by varying the inductance LR.
Is controlled to be constant, but by this operation, the DC output voltage on the secondary side is operated to be constant.

For example, in the configuration of the soft switching power supply circuit previously proposed by the present applicant, for example, the AC input voltage A
C100V system and maximum load power 150W ~ 160W
In the case of adopting the configuration corresponding to the above, a rectified smoothed voltage corresponding to almost twice the level of the AC input voltage is obtained by the voltage doubler rectifier circuit. Thereby, the DC voltage level input to the switching converter is increased to cope with the condition of relatively high load.

On the other hand, in the power supply circuit shown in FIG. 3, by increasing the maximum load power as described above, the rectifying and smoothing circuit for generating the DC input voltage (rectified and smoothed voltage) is a voltage doubler rectifier. It is no longer necessary to cover the load power by taking As a result, as shown in FIG. 3, for example, a configuration of a normal equal-voltage rectifier circuit using a bridge rectifier circuit can be adopted. Thereby, for example, in the power supply circuit shown in FIG. 3, the rectified smoothed voltage Ei at the time of the AC input voltage VAC = 144 V becomes about 200 V. Here, the collector-emitter voltage VCP of the switching element Q1 is generated when the switching element Q1 is turned off by the action of the primary-side parallel resonance circuit on the rectified smoothed voltage Ei, but in the circuit of FIG. As described above, the rectified smoothed voltage Ei is reduced to about 1/2 of the voltage doubled rectification.
However, in the configuration shown in FIG. 3, since the boost voltage VB is superimposed on the rectified smoothed voltage Ei to generate the boost smoothed voltage EB, the resonance voltage Vcr is changed to the boost smoothed voltage EB.
Of the resonance voltage Vc
r can be suppressed to about 1200V. Therefore, in the circuit shown in FIG. 3, a 1200V withstand voltage product may be selected for the switching element Q1.

Further, as described above, the full-wave rectifier circuit is provided on the secondary side, and the rectified current flows during both the positive and negative periods of the alternating voltage of the secondary winding N2A, so that the secondary side resonance The voltage V2 is suppressed to the same level as the rectified smoothed voltage Ei in both the positive and negative periods.
As a result, as the rectifier diodes (DO1 to D04) forming the secondary-side full-wave rectifier circuit, a withstand voltage product substantially corresponding to the level of the rectified smoothed voltage Ei can be selected.

As described above, in the circuit shown in FIG. 3, the size of the parallel resonance capacitor Cr is reduced, and the switching element Q
Since a low-withstand voltage product can be used for the rectifier diode forming the full-wave rectifier circuit on the primary and secondary sides, the element is less expensive. Therefore, for example, the switching element Q1 and a rectifying diode forming a full-wave rectifier circuit on the secondary side have improved characteristics (for the switching element Q1, the saturation voltage VCE (SAT ), Accumulation time tSTG, fall time t
f, good characteristics such as current amplification factor hFE, and forward voltage drop VF and reverse recovery time t for a rectifier diode.
rr and the like) can be selected, and the power loss can be reduced accordingly. That is, power conversion efficiency can be improved at a lower cost or at substantially the same cost as compared with a configuration in which a DC input voltage is generated by a voltage doubler rectifier circuit. Further, by improving the power conversion efficiency, for example, a radiator plate or the like which is necessary for radiating heat of the rectifier diode of the voltage doubler rectifier circuit becomes unnecessary.

If a high switching frequency of, for example, about 100 KHz is set as the configuration of the voltage resonance type converter, the above-mentioned various components can be reduced in size and weight. Here, if the winding N3 is selected such that the boost voltage VB is optimized according to the maximum load power to be actually handled, further reduction in size and weight of various components can be realized. Furthermore, from the viewpoint of reducing the size and weight of the power supply circuit, in the configuration including the voltage doubler rectifier circuit for generating the DC input voltage, two sets of rectifier diodes and a smoothing capacitor were required, respectively. In the circuit shown in FIG. 3, for example, a full-wave rectifier circuit using a normal bridge rectifier circuit can be used, so that a set of a block-type smoothing capacitor and a bridge rectifier diode can be employed. In addition, miniaturization of parts can be achieved. That is, in the circuit shown in FIG. 3, miniaturization of various components including the insulating converter transformer PIT and the orthogonal control transformer PRT is achieved. Experiments have also shown that the power conversion efficiency can be improved.

FIG. 6 is a circuit diagram showing a configuration of a power supply circuit according to a third embodiment of the present invention. The power supply circuit shown in this figure has a configuration in which a commercial AC power supply is an AC 100 V system and can cope with a relatively heavy load having a maximum load power of 160 W or more. 1 and 3 are denoted by the same reference numerals, and description thereof is omitted.

In the case of this embodiment, a bridge rectifier circuit Di and one smoothing capacitor C are used as rectifier circuits for inputting the AC input voltage VAC to obtain the rectified smoothed voltage Ei.
i. As described below, the load power is covered by increasing the switching output by the push-pull switching operation by the two switching elements, so that the DC input voltage is particularly 2Ei. Level is not required.

Further, a center tap is provided on the primary winding N1, so that the primary winding N1A (inductance L1A), N
1B (inductance L1B). This center tap terminal is connected to the line of the rectified and smoothed voltage Ei through the series connection of the controlled winding NR of the orthogonal control transformer PRT.

The voltage resonance type converter shown in FIG.
A separately-excited configuration having stone switching elements Q1 and Q2 is employed. In this case, the switching element Q1 has M
OS-FET is adopted. In this case, the drain of the switching element Q1 is connected to the end of the primary winding N1A, and the source is grounded to the primary side ground. Further, a damper diode DD1 forming a current path during the switching-off period is connected in parallel between the drain and the source of the switching element Q. The drain of the switching element Q2 is connected to the end of the primary winding N1B, and the source is grounded to the primary side ground. Further, a damper diode DD2 is also connected in parallel between the drain and source of the switching element Q.

The primary side parallel resonance capacitors of the present embodiment include parallel resonance capacitors Cr1 and Cr2 corresponding to the switching elements Q1 and Q2, respectively. The parallel resonance capacitor Cr1 is connected so as to be inserted between the collector of the switching element Q1 and the positive electrode (rectified and smoothed voltage Ei line) of the smoothing capacitor Ci, so that the capacitance of the parallel resonance capacitor Cr1 and the inductance L1A are provided. , The switching element Q1
Is formed as a voltage resonance type switching operation. Similarly, the parallel resonance capacitor Cr2
Is connected so as to be inserted between the collector of the switching element Q2 and the positive electrode (rectified and smoothed voltage Ei line) of the smoothing capacitor Ci, so that the capacitance of the parallel resonance capacitor Cr2 and the inductances L1B and LR are reduced. The combined inductance forms a parallel resonance circuit that makes the switching operation of the switching element Q2 a voltage resonance type. In the conventional configuration,
The parallel resonance capacitors Cr1 and Cr2 are connected in parallel between the collector and the emitter of the switching elements Q1 and Q2, respectively. The switching element Q1,
The switching output point of Q2 will be the center tap of primary winding N1. That is, in the present embodiment,
Two switching circuit systems for driving the switching elements Q1 and Q2, respectively, are provided.

The oscillating drive circuit 2 is started by a starting resistor RS. For example, the oscillating drive circuit 2 oscillates and outputs a frequency signal corresponding to, for example, a switching frequency fs = 100 KHz. Voltage) to generate the switching element Q
1, Apply to the gate of Q2. At this time, the switching drive signals applied to the gates of the switching elements Q1 and Q2 are signals of opposite polarities inverted from each other.

Here, assuming that both switching elements Q1 and Q2 perform a switching operation, switching element Q1
1, Q2 is, for example, the switching frequency fs = 100 KH
At z, the switching operation is performed at the timing of turning on / off alternately. That is, the switching element Q
When both Q1 and Q2 perform switching, a switching operation by a so-called push-pull method (push-pull operation) is performed.
More power can be supplied to the load side than in the case of a so-called single-ended switching operation (single-ended operation). Thus, the present embodiment corresponds to a heavy load condition with a maximum load power of 160 W or more.

The control circuit 1 shown in this figure compares the DC voltage output (EO1) on the secondary side with the reference voltage, similarly to the control circuit 1 previously shown in FIGS. 1 and 3, for example. An error amplifier is configured to supply a DC current corresponding to the error as a control current to the control winding NC of the orthogonal control transformer PRT.

For example, when the DC output voltage EO2 on the secondary side fluctuates due to the fluctuation of the AC input voltage VAC or the minimum load power, the control circuit 1 controls the control current flowing through the control winding NC to a predetermined range, for example. Change. As a result, the inductance LR of the controlled winding NR is changed within a predetermined range.

In this embodiment, as described above, the controlled winding NR is inserted so as to form a parallel resonance circuit on the switching element Q1 side and a parallel resonance circuit on the switching element Q2 side. . Therefore, when the inductance LR changes, the resonance condition of the resonance circuit changes with respect to the fixed switching frequency during the switching operation of the switching elements Q1 and Q2. The output level of the secondary-side DC output voltage (EO1, EO2) changes in accordance with the change in the resonance condition, whereby the secondary-side DC voltage (EO1, EO2) is made constant. Will be. That is, also in this case, the constant voltage control operation by the inductance control method is performed.

According to the configuration shown in FIG. 6, as in the first and second embodiments, the size of the primary-side parallel resonance capacitors Cr1, Cr2 is reduced, and the radiation noise on the secondary side is absorbed. The ceramic capacitor Cp is omitted.

It should be noted that the present invention is not limited to the configurations shown in the respective drawings as the first to third embodiments, but can be modified. For example, in the configurations shown in FIGS. 3 and 6, for example, the secondary side rectifier circuit may be configured to include a voltage doubler full-wave rectifier circuit. In this case, for example, If a DC output voltage of the same level as that of the full-wave rectifier circuit is to be obtained, the number of turns of the secondary winding N2 can be reduced to half. As for the switching elements, not only bipolar transistors but also other types of switching elements such as MOS-FET transistors may be employed.

[0057]

As described above, in the voltage resonance type converter of the present invention, the resonance capacitor of the primary side resonance circuit for making the switching operation a voltage resonance type is provided by connecting the positive line of the DC input voltage and the switching output of the switching means. Insert between terminals. by this,
The reference potential of the resonance voltage generated at both ends of the resonance capacitor due to the action of the primary side resonance circuit is shifted by the DC input voltage level, and the level of the resonance voltage pulse generated at both ends of the resonance capacitor when the switching element is turned off is reduced accordingly. It is possible to lower it. As a result, in an actual circuit, it is possible to keep it within the withstand voltage of a capacitor (for example, a polypropylene film capacitor), which is a limit of manufacture, and it is not necessary to form a resonance capacitor by connecting two or more capacitors in series. That is, as the resonance capacitor, it is possible to reduce the number of series-connected capacitors to, for example, a single capacitor, and accordingly, it is possible to reduce the size and weight of the resonance capacitor and to reduce the cost.

A secondary parallel resonance capacitor forming a secondary parallel resonance circuit together with an inductance component of the secondary winding is connected in parallel to a secondary rectifier diode forming a secondary rectifier circuit. By connecting, it also operates to eliminate the radiated noise generated during the reverse recovery time of the secondary rectifier diode.Conventionally, ceramic connected in parallel with the secondary rectifier diode to suppress radiation noise Capacitors can be omitted. As described above, the present invention has an effect that the size and weight of the circuit of the voltage resonance type switching power supply circuit and the cost reduction can be promoted.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration example of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part of the power supply circuit of the embodiment.

FIG. 3 is a circuit diagram illustrating a configuration example of a power supply circuit according to a second embodiment of the present invention.

FIG. 4 is a cross-sectional view illustrating a configuration of an insulating converter transformer.

FIG. 5 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 6 is a circuit diagram illustrating a configuration example of a power supply circuit according to a third embodiment of the present invention.

FIG. 7 is a circuit diagram showing a configuration example of a power supply circuit as a conventional example.

8 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

[Explanation of symbols]

1 control circuit, 2 oscillation circuit, 3 drive circuit, C
i, CiB smoothing capacitor, Cr, Cr1, Cr2 parallel resonance capacitor, C2, C2A, C2B (secondary side) parallel resonance capacitor, Di bridge rectifier circuit, DO1, DO
2, DO3, DO4 rectifier diode, PIT isolation converter transformer, N1, N1A, N1B primary winding, N2 secondary winding, PRT orthogonal control transformer, NC control winding,
NR controlled winding, Q1, Q2 switching element, DB
Boost diode, N3 (for boost) winding,

Claims (5)

[Claims] A rectifying / smoothing means for generating a rectified / smoothed voltage by inputting a commercial AC power supply and performing rectification / smoothing, and outputting the rectified / smoothed voltage as a DC input voltage; A switching unit configured to output to a winding; an inductance component including at least a primary winding of the insulating converter transformer; and an insertion component inserted between an output terminal of the switching unit and a positive line of the DC input voltage. By providing a primary-side resonance circuit formed by the capacitance of the resonance capacitor to make the operation of the switching means a voltage resonance type, a secondary-side rectifier diode element, and a secondary-side smoothing capacitor, DC output voltage generating means for generating a secondary DC output voltage from an alternating voltage obtained at the next winding A secondary resonance circuit formed on the secondary side by an inductance component of a secondary winding of the insulating converter transformer and a capacitance of a secondary parallel resonance capacitor; and functioning as an inductance component of the primary resonance circuit. And a control winding in which a controlled winding and a control winding whose winding direction is orthogonal to the controlled winding are wound, and the DC output voltage of By flowing a variable control current through the control winding according to the level and changing the inductance of the controlled winding,
And a constant voltage control means configured to perform constant voltage control on the secondary-side DC output voltage. 2. The voltage resonance according to claim 1, wherein the secondary-side parallel resonance capacitor forming the secondary-side resonance circuit is connected in parallel to a secondary-side rectifier diode element. Type switching power supply circuit. 3. A boost rectified smoothed voltage is obtained by superimposing a boost voltage generated using a switching output of the switching means on the rectified smoothed voltage, and the boost rectified smoothed voltage is used as a DC input voltage. A boost which is configured to be supplied to switching means and has a configuration in which the controlled rectifying smoothed voltage can be controlled to be constant by a change in inductance of the controlled winding by including the controlled winding. The voltage resonance type switching power supply circuit according to claim 1, further comprising: 4. The switching means is provided in two sets,
2. The voltage resonance type switching power supply circuit according to claim 1, wherein the voltage resonance type switching power supply circuit is configured to output a DC input voltage intermittently by a push-pull operation. 5. The voltage resonance type switching power supply circuit according to claim 1, wherein said DC output voltage generation means is formed with a full-wave rectifier circuit.