CN101887095B - Method for testing radiated noises of digital gravity metering device - Google Patents

Abstract

The invention discloses a method for testing radiated noises of a digital gravity metering device, which comprises the following steps of: calculating corresponding transfer impedance according to a high-frequency current clamp structure; putting a measuring loop in the high-frequency current clamp on a power cord of the digital gravity metering device so as to extract a first noise voltage and a second noise voltage in the power cord; performing a vector operation and a vector difference operation on the first noise voltage and the second noise voltage by using a noise separation network so as to obtain a difference mode conducted noise voltage and a common mode conducted noise voltage; performing operation on the difference mode conducted noise voltage, the common mode conducted noise voltage and the transfer impedance of the high-frequency current clamp to obtain a difference mode conducted noise current and a common mode conducted noise current; and acquiring an electric field which emits the radiation noises in a free space according to a common mode noise current formula. In the method, the common mode noise current of a single-chip digital gravity metering device is measured by using the high-frequency current clamp and the noise separation network, and the radiated noises are predicted through the common mode noise current.

Description

A kind of radiation noise test method for digital gravity metering device

Technical field

The present invention relates to a kind of radiation noise test method for digital gravity metering device, belong to electromagnetic compatibility test and power line communication field.

Background technology

Electronic equipment, must be by corresponding standard testing before carrying out marketing, and wherein one is exactly the Radiative EMI standard testing.And based on the radiated noise of single-chip microcomputer digital gravity metering device mainly because its transmission cable radiation causes.And the existing cost that adopts microwave dark room to measure electromagnetic radiation is higher, therefore how for the characteristics of single-chip microcomputer digital gravity metering device, developing a kind of short-cut method of radiation noise test method for digital gravity metering device is the problem that the present invention studies.

Summary of the invention

The object of the invention is to provide a kind of radiation noise test method for digital gravity metering device, the method is first measured the common mode noise current of single-chip microcomputer digital gravity metering device by high-frequency current clamp and noise separating network, then estimates radiated noise by this common mode noise current.

For achieving the above object, the technical solution used in the present invention is: a kind of radiation noise test method for digital gravity metering device comprises the following steps:

Step 1: the performance of test conducted noise separated network common mode insertion loss and differential mode rejection ratio in the radiated noise frequency range meets and measures requirement in the radiated noise frequency range to determine this conducted noise separated network;

Step 2: according to the high-frequency current clamp Structure Calculation, draw corresponding transfer impedance Z (ω), the specific formula for calculation of this transfer impedance Z (ω) is:

$Z\left(\mathrm{\ω}\right)=\frac{R_2}{R_2+\mathrm{j\ω}{L}_2}\·\mathrm{j\ωM}$

In formula, R ₂For frequency spectrograph receives impedance, L ₂For the high-frequency current clamp coefficient of self-induction, M is the coefficient of mutual inductance between test circuit and high-frequency current clamp, and ω is angular frequency, and j is the imaginary part of symbol;

Step 3: measure ring by one in high-frequency current clamp and be placed on the live wire and ground wire of digital gravity metering device power lead, with the first noise voltage (V of line over the ground of live wire in power lead in the extraction digital gravity metering device ₁); Another measurement ring in high-frequency current clamp is placed on the center line and ground wire of digital gravity metering device power lead, with the second noise voltage (V of center line to ground wire in power lead in the extraction digital gravity metering device ₂);

Step 4: by described the first noise voltage (V ₁), the second noise voltage (V ₂) in conjunction with the transfer impedance Z (ω) of described high-frequency current clamp, carry out computing, obtain differential mode noise electric current and common mode noise current;

Concrete formula is as follows:

$\left|V_1\right|=\frac{\mathrm{j\ωM}\·R_2}{R_2+\mathrm{j\ω}{L}_2}(I_{\mathrm{CM}}+I_{\mathrm{DM}})$

$\left|V_2\right|=\frac{\mathrm{j\ωM}\·R_2}{R_2+\mathrm{j\ω}{L}_2}(I_{\mathrm{CM}}-I_{\mathrm{DM}})$

In formula, R ₂For frequency spectrograph receives impedance, L ₂For the high-frequency current clamp coefficient of self-induction, M is the coefficient of mutual inductance between test circuit and high-frequency current clamp, and ω is angular frequency (rad/s), and j is imaginary part unit, I _CMFor common mode noise current (A), I _DMFor differential mode noise electric current (A);

Step 5: pass through following formula according to common mode noise current:

$E_{\mathrm{\θ}}\≈j\frac{l{Z}_0{I}_{\mathrm{CM}}{\mathrm{\β}}_0\sin \mathrm{\θ}}{4\mathrm{\πr}}{e}^{-j{\mathrm{\β}}_{0}r}$

, obtain and launch the radiated noise electric field to free space, in formula, Z ₀For the free space wave impedance, unit is Ω; L is conductor length, and unit is m; I _CMFor common mode noise current, unit is A; R is measuring distance, and unit is m; β ₀Be 2 π/λ, wherein λ is the correlated frequency signal wavelength, and unit is m; E _θFor the radiated noise electric field, unit is dBuV/m; J is the imaginary part sign; θ is for measuring vector angle;

Described high-frequency current clamp is connected to noise separating network, and this noise separating network is connected to described frequency spectrograph.

Related content in technique scheme is explained as follows:

1, in such scheme, described common mode insertion loss and differential mode rejection ratio performance are that the conducted noise separated network is the performance more than 30MHz in frequency of operation.

2, in such scheme, described R ₂For receiver internal impedance (Ω), L ₂For sense (H) in high-frequency current clamp, M is the volume mutual inductance (H) between circuit and high-frequency current clamp, and ω is angular frequency (rad/s), and j is imaginary part unit.

Because technique scheme is used, the present invention compared with prior art has following advantages:

1, the present invention to be to measure the conducted noise in the single-chip microcomputer digital gravity metering device, and is isolated into common-mode noise and differential mode noise, and adopts corresponding squelch measure to make electric line communication system reach standard-required at conduction portion.

2, the present invention estimates for radiated noise according to common mode noise current, can save great number measurement expense such as utilizing microwave dark room.

3, the present invention adopts separated network to extract common-mode noise, and designs corresponding wave filter and suppress for its common-mode noise, thereby reaches the purpose that suppresses single-chip microcomputer digital gravity metering device radiated noise.Make the inhibition of radiation EMI noise more effective.

4, the present invention uses high-frequency current clamp can measure little electric current to the microampere magnitude.

5, measure traverse line of the present invention between balance to guarantee to optimize the integrality of signal.

The accompanying drawing explanation

The measurement system diagram that Fig. 1 high-frequency current clamp and noise separating network form;

Fig. 2 high-frequency current clamp circuit topological structure figure;

Fig. 3 RF current probe is estimated principle wherein, and I1, I2...In are every segment corresponding current, and E1, E2...En are every section electric field that electric current is corresponding;

Fig. 4, based on single-chip microcomputer digital gravity metering device circuit diagram, amplifies position for causing the radiated noise reason;

Fig. 5 high-frequency current clamp transfer impedance test pattern;

Fig. 6 noise separating network high frequency characteristics figure;

Fig. 7 radiated noise estimation results.

Embodiment

The invention will be further described below in conjunction with drawings and Examples:

Embodiment: a kind of radiation noise test method for digital gravity metering device comprises the following steps:

Step 1: the performance of test conducted noise separated network common mode insertion loss and differential mode rejection ratio in the radiated noise frequency range meets and measures requirement in the radiated noise frequency range to determine this conducted noise separated network;

Step 2: according to the high-frequency current clamp Structure Calculation, draw corresponding transfer impedance Z (ω), the specific formula for calculation of this transfer impedance Z (ω) is:

$Z\left(\mathrm{\ω}\right)=\frac{R_2}{R_2+\mathrm{j\ω}{L}_2}\·\mathrm{j\ωM}$

In formula, R ₂For frequency spectrograph receives impedance, L ₂For the high-frequency current clamp coefficient of self-induction, M is the coefficient of mutual inductance between test circuit and high-frequency current clamp, and ω is angular frequency, and j is the imaginary part of symbol;

Step 3: measure ring by one in high-frequency current clamp and be placed on the live wire and ground wire of digital gravity metering device power lead, with the first noise voltage (V of line over the ground of live wire in power lead in the extraction digital gravity metering device ₁); Another measurement ring in high-frequency current clamp is placed on the center line and ground wire of digital gravity metering device power lead, with the second noise voltage (V of center line to ground wire in power lead in the extraction digital gravity metering device ₂);

Step 4: by noise separating network by live wire noise voltage (V _L), center line noise voltage (V _N) carry out vector and phasor difference computing, obtain differential mode conducted noise voltage (V _DM) and common mode conducted noise voltage (V _CM), with differential mode conducted noise in the discrete conductive noise and common mode conducted noise;

Concrete formula is as follows:

$\left|V_{\mathrm{DM}}\right|=\left|\frac{V_1-V_2}{2}\right|$

$\left|V_{\mathrm{CM}}\right|=\left|\frac{V_1+V_2}{2}\right|$

In formula, V ₁Be the first noise voltage, V ₂Be the second noise voltage, V _DMFor differential mode conducted noise voltage, V _CMFor common mode conducted noise voltage.

Step 5: just described the first noise voltage (V ₁), the second noise voltage (V ₂) in conjunction with the transfer impedance Z (ω) of described high-frequency current clamp, carry out computing, obtain differential mode noise electric current and common mode noise current;

Concrete formula is as follows:

$\left|V_1\right|=\frac{\mathrm{j\ωM}\·R_2}{R_2+\mathrm{j\ω}{L}_2}(I_{\mathrm{CM}}+I_{\mathrm{DM}})$

$\left|V_2\right|=\frac{\mathrm{j\ωM}\·R_2}{R_2+\mathrm{j\ω}{L}_2}(I_{\mathrm{CM}}-I_{\mathrm{DM}})$

In formula, in formula, R ₂For frequency spectrograph receives impedance, L ₂For the high-frequency current clamp coefficient of self-induction, M is the coefficient of mutual inductance between test circuit and high-frequency current clamp, and ω is angular frequency (rad/s), and j is imaginary part unit, I _CMFor common mode noise current (A), I _DMFor differential mode noise electric current (A);

Step 6: pass through formula according to following common mode noise current:

$E_{\mathrm{\θ}}\≈j\frac{l{Z}_0{I}_{\mathrm{CM}}{\mathrm{\β}}_0\sin \mathrm{\θ}}{4\mathrm{\πr}}{e}^{-j{\mathrm{\β}}_{0}r}$

, obtain and launch the radiated noise electric field to free space, in formula, Z ₀For the free space wave impedance, unit is Ω; L is conductor length, and unit is m; I _CMFor common mode noise current, unit is A; R is measuring distance, and unit is m; β ₀Be 2 π/λ, wherein λ is the correlated frequency signal wavelength, and unit is m; E _θFor the radiated noise electric field, unit is dBuV/m; J is the imaginary part sign; θ is for measuring vector angle.

Described high-frequency current clamp is connected to noise separating network, and this noise separating network is connected to described frequency spectrograph.

Described common mode insertion loss and differential mode rejection ratio performance are that the conducted noise separated network is the performance more than 30MHz in frequency of operation.

Described R ₂For receiver internal impedance (Ω), L ₂For sense (H) in high-frequency current clamp, M is the volume mutual inductance (H) between circuit and high-frequency current clamp, and ω is angular frequency (rad/s), and j is imaginary part unit.

The present embodiment foregoing specific explanations is as follows.

High-frequency current clamp and noise separating network extract for the common mode noise current of single-chip microcomputer gravity metering device, and then by common mode noise current, carry out the new test macro of the radiated noise in anticipator circuit.

Utilize the test macro of noise separating network and high-frequency current clamp composition circuit theory such as Fig. 1 for metering circuit electromagnetic interference noise electric current.Wherein, high-frequency current clamp utilizes the mutual inductance effect that actual electromagnetic interference noise in circuit is sensed in current clamp, and is received by spectrum analyzer.The equivalent model of high-frequency current clamp metering circuit as shown in Figure 2.According to Fig. 2 circuit, transfer impedance Z (ω) defines suc as formula shown in (1):

$Z\left(\mathrm{\ω}\right)=\frac{V_2\left(\mathrm{\ω}\right)}{I_1\left(\mathrm{\ω}\right)}---\left(1\right)$

Wherein, V ₂The magnitude of voltage (ω) measured for high-frequency current clamp, I ₁(ω) be current value in circuit.Due to the mutual inductance effect of high-frequency current clamp, be not difficult to draw that the electromagnetic interference (EMI) electric current in circuit produces in current probe induction electromotive force is j ω MI ₁(ω), because the internal impedance of surveying instrument is 50 Ω, identical with matched impedance again.Current probe measure circuital current in the voltage of gained and circuit relation suc as formula shown in (2):

$V_2=\frac{R_2}{R_2+\mathrm{j\ω}{L}_2}\·\mathrm{j\ωM}{I}_1\left(\mathrm{\ω}\right)---\left(2\right)$

Wherein, R ₂Be 50 Ω loaded impedances, j ω MI ₁The magnitude of voltage (ω) obtained by mutual inductance effect induction for voltage probe, L ₁Self-induction for current probe.In circuit, because the matched impedance resistance is identical with the surveying instrument internal impedance, so can utilize the magnitude of voltage measured on matched impedance to come electric current in counting circuit suc as formula shown in (3):

V ₃(ω)＝R ₁·I ₁(ω)＝50·I ₁(ω) (3)

Wherein, R ₁Be 50 Ω loaded impedances.So (2) formula can be converted to:

$V_2=\frac{R_2\mathrm{j\ωM}}{50(R_2+\mathrm{j\ω}{M}_5)}\·{\stackrel{\·}{V}}_3\left(\mathrm{\ω}\right)---\left(4\right)$

Utilize noise separating network, can extract high-frequency current clamp and measure the common mode component in the overall noise electric current.Single-chip microcomputer gravity metering device line noise voltage V _L, V _NBe defined as overall noise, can obtain differential mode noise in circuit and common-mode noise suc as formula shown in (5) (6) by both being carried out to vector and phasor difference:

$\left|V_{\mathrm{DM}}\right|=\left|\frac{V_L-V_N}{2}\right|=50\left|i_{\mathrm{DM}}\right|---\left(5\right)$

$\left|V_{\mathrm{CM}}\right|=\left|\frac{V_L+V_N}{2}\right|=50\left|i_{\mathrm{CM}}\right|---\left(6\right)$

For the conducted noise electric current, according to the electromagnetic compatibility theory, also it can be divided into to differential mode noise electric current and common mode noise current.The differential mode noise electric current results between live wire and center line, and it defines suc as formula shown in (7):

$I_{\mathrm{dm}}=\frac{I_L-I_N}{2}---\left(7\right)$

Common mode noise current is current noise sum on live wire and ground wire and zero line and ground wire, and it defines suc as formula shown in (8):

I _cm＝I _L+I _N (8)

According to the common mode current shown in Fig. 1, flow to as can be known, because the impedance of reference ground is less than the inductive impedance of safety line value, so common mode current finally flows to reference ground after by live wire and zero line.Therefore, common mode current is the main cause that causes Radiative EMI.In the design of single-chip microcomputer gravity metering device, due to the existence of circuit intermediate ring road, cause existing in circuit the common mode current of this kind form, thereby cause the increase of its radiated noise.

As can be known by above analysis, the relation of common mode in the voltage measured and circuit (CM)/differential mode (DM) electric current can be meaned by formula (9):

$V_2=\frac{R_2}{R_2+\mathrm{j\ω}{L}_2}\·\mathrm{j\ωM}{I}_1\left(\mathrm{\ω}\right)$

$=\mathrm{j\ωM}\frac{(V_{\mathrm{CM}}+V_{\mathrm{DM}})}{50+\mathrm{j\ω}{L}_1}\·\frac{R_2}{R_2+\mathrm{j\ω}{L}_2}$

$=\mathrm{j\ωM}\frac{(V_{\mathrm{CM}}+V_{\mathrm{DM}})\·50}{(50+\mathrm{j\ω}{L}_1)(50+\mathrm{j\ω}{L}_2)}---\left(9\right)$

$=\frac{\mathrm{j\&omega;M}\&CenterDot;50}{50+\mathrm{j\&omega;}{L}_2}(\frac{{\mathrm{<figure-callout\; id="50"\; label="V\; CM"\; filenames="HSA00000185914100041.png"\; state="\{\{state\}\}">V</figure-callout>}}_{\mathrm{CM}}}{50+\mathrm{j\&omega;}{L}_1}+\frac{{\mathrm{<figure-callout\; id="50"\; label="V\; DM"\; filenames="HSA00000185914100041.png"\; state="\{\{state\}\}">V</figure-callout>}}_{\mathrm{DM}}}{50+\mathrm{j\&omega;}{L}_1})$

$=\frac{\mathrm{j\&omega;M}\&CenterDot;50}{50+\mathrm{j\&omega;}{L}_2}(I_{\mathrm{CM}}+I_{\mathrm{DM}})$

The induced voltage that current probe can be measured by this formula and the common mode noise current in circuit and differential mode noise electric current interrelate, and by noise separating network, can obtain the common mode noise current of single-chip microcomputer gravity metering device, thereby estimate its radiated noise.Its concrete implementation step is as follows:

1, at first, the transfer impedance for current probe in frequency domain is tested, and considers that the equivalent electrical circuit of current probe is comprising self-induction and two aspects of mutual inductance, so can write out current probe partial circuit equation, is not difficult to derive V ₂(ω) and V ₁Relation (ω) is as shown in (4) formula.

2, for the performance of conducted noise separated network in the radiation frequency range, test, in the radiation frequency range, whether can meet the measurement requirement to determine its performance.

3, set up single-chip microcomputer numeral gravity metering device common mode noise current extraction system, this system is formed (seeing Fig. 1) by high-frequency current clamp, digital gravity metering device, spectrum analyzer and noise separating network.High-frequency current clamp 1 is clipped on live wire and ground wire, with the flow through electromagnetic interference noise electric current I L of live wire and ground wire of measurement; Simultaneously another high-frequency current clamp 2 is clipped on center line and ground wire, with the flow through electromagnetic interference noise electric current I N of center line and ground wire of measurement.By the output terminal access noise separated network of two high-frequency current clamp, in order to the common-mode noise in extraction system.And common-mode noise is converted into to common mode noise current, can estimate according to the common mode noise current of extracting the radiated noise of this system.The electromagnetic radiation of single-chip microcomputer digital gravity metering device is mainly derived from the common mode radiation on circuit, and common mode radiation derives from high frequency common mode current, common mode current is launched radiation field along transmission line to free space, and the field intensity computing formula is suc as formula shown in (10):

$E_{\mathrm{\&theta;}}\&ap;j\frac{l{Z}_0{I}_{\mathrm{CM}}{\mathrm{\&beta;}}_0\sin \mathrm{\&theta;}}{4\mathrm{\&pi;r}}{e}^{-j{\mathrm{\&beta;}}_{0}r}---\left(10\right)$

Wherein: Zo is free space wave impedance (Ω), and l is conductor length (m), I _CMFor common mode noise current (A), r is measuring distance (m), and β o=2 π/λ, λ are correlated frequency signal wavelength (m).

Along with the increase of frequency, the physical length of wire will be comparable with the physical dimension of wavelength, along the uniformity no longer of the distribution of current on wire.Wire evenly can be divided into to N segment for this reason, adopt the radio frequency high-frequency current clamp in the centre position of every segment, to measure its electric current (seeing Fig. 3) separately, establish and be respectively I1, I2...In, can adopt formula (10) to calculate for each section lead.

Adopt formula (10) to calculate the radiation field of each segment.Due to open testing field (OATS), normally measure the standard place of electromagnetic radiation, so need, consider the ground return effect, total equivalent radiated power field is calculated and is converted into suc as formula shown in (11).

$\left|E_c\right|\&ap;\frac{2\mathrm{\&pi;f}\&times;{10}^{-7}F(I_1+I_2+...+I_n)l}{3\sqrt{r^2+{(H-0.8)}^2}}---\left(11\right)$

Wherein, | EC| is radiated electric field, and 1 is every segment antenna eliminator length, and f means test frequency, and r means open testing field standard testing distance, and H means the test antenna height, and F is for calculating the modifying factor under the test environment of open testing field.

4, because the digital gravity metering device transmission cable is longer, therefore can utilize the test macro that high-frequency current clamp and noise separating network form to extract for the common mode noise current of zones of different on transmission cable, with the optimum position of determining that braking measure loads.

5, adopt ferrite bead to suppress for the radiated noise in single-chip microcomputer numeral gravity metering device, by relatively loading before and after ferrite bead on cable the attenuation degree of radiated noise to check the effect of method in literary composition.

The method of testing that the present invention adopts, can estimate for the radiation EMI noise on all kinds of single-chip microcomputer numeral gravity metering device transmission cables.Thereby adopt high-frequency current clamp and noise separating network to extract for the radiation EMI noise and estimate for common mode noise current on cable, the method is simple to operate and test is accurate.By this method of testing, can test the radiation EMI noise of single-chip microcomputer numeral gravity metering device transmission line, further for suppressing this radiation EMI noise, provide theoretical foundation.

As shown in Figure 4, it is due to due in circuit, the 12MHz clock circuit is coupled in transmission line that its radiated noise produces reason to its circuit.Set up test macro as shown in Figure 1, this test macro is comprised of high-frequency current clamp, noise separating network, spectrum analyzer and digital gravity metering device etc.

Concrete method of testing:

In time domain, its experiment connects as shown in Figure 5, this experiment utilizes signal generator in required measurement frequency range, to export the sinusoidal signal of different frequency in a port of dual trace oscilloscope, simultaneously high-frequency current clamp is carried on transmission line, the value measured is input to the another one port of dual trace oscilloscope, according to 3 formulas, can obtain on different frequency point the transfer impedance of this current probe.

The noise separating network adopted is measured in the performance of radiation frequency range, can be applied in the middle of estimating of radiated noise to determine this noise separating network.The good and bad leading indicator of judgement common-mode noise separated network performance be its common mode insertion loss (CMIL) and its definition of differential mode rejection ratio (DMRR) suc as formula shown in (12):

$\mathrm{CMIL}=20\lg \left(\frac{V_{\mathrm{OC}}}{V_{\mathrm{CM}}}\right)$ $\mathrm{DMRR}=20\lg \left(\frac{V_{\mathrm{OC}}}{V_{\mathrm{DM}}}\right)---\left(12\right)$

Wherein, V _CMFor common mode input, V _OCFor common mode output; V _DMFor difference-mode input, V _ODFor differential mode output.In the common mode separated network, the ideal value of common mode insertion loss should be 0, to guarantee that the loss of its common mode noise signal is less in the common-mode noise transmission; In like manner as can be known, its differential mode rejection ratio should be large as far as possible, to guarantee its differential mode noise, can not be coupled in the transmission of common mode noise signal, so the ideal value of its differential mode rejection ratio should be-∞.By test, can obtain its common mode insertion loss and differential mode rejection ratio test result as shown in Figure 6.This result shows that this separated network can apply to estimating of radiation EMI frequency range.

At the test noise separated network on the basis of radiation frequency range performance, can the connecting circuit system.By the digital gravity metering device energising, and high-frequency current clamp 1 is clipped on live wire and ground wire; Another high-frequency current clamp is clipped on center line and ground wire, to measure live wire noise and the center line noise V on cable _L, V _NIts frequency that exceeds standard is 36MHz (as shown in Figure 7).The output port of high-frequency current clamp is linked in noise separating network, by the noise inputs spectrum analyzer after separating to analyze the common mode noise current of single-chip microcomputer gravity metering device.

High-frequency current clamp is placed in to different places on the digital gravity metering device cable, to determine the position of radiation maximum on cable.In this case, be divided into nearly power end, mid point and nearly equipment end three places are detected, and its testing result is as shown in table 1:

Table 1 digital gravity metering device Conducted EMI noise separation result

Filter location	Nearly power end	Mid point	Nearly equipment end
				The radiated noise amplitude	42dBuV	42.9dBuV	48dBuV

Shown in table, its inhibition site should be at nearly equipment end place.

Above-described embodiment only is explanation technical conceive of the present invention and characteristics, and its purpose is to allow the person skilled in the art can understand content of the present invention and implement according to this, can not limit the scope of the invention with this.All equivalences that Spirit Essence is done according to the present invention change or modify, within all should being encompassed in protection scope of the present invention.

Claims (3)

1. radiation noise test method for digital gravity metering device is characterized in that: comprise the following steps:

Step 1: the performance of test conducted noise separated network common mode insertion loss and differential mode rejection ratio in the radiated noise frequency range meets and measures requirement in the radiated noise frequency range to determine this conducted noise separated network;

Step 2: according to the high-frequency current clamp Structure Calculation, draw corresponding transfer impedance Z (ω), the specific formula for calculation of this transfer impedance Z (ω) is:

$Z\left(\mathrm{\&omega;}\right)=\frac{R_2}{R_2+\mathrm{j\&omega;}{L}_2}\&CenterDot;\mathrm{j\&omega;M}$

In formula, R ₂For frequency spectrograph receives impedance, L ₂For the high-frequency current clamp coefficient of self-induction, M is the coefficient of mutual inductance between test circuit and high-frequency current clamp, and ω is angular frequency, and j is the imaginary part of symbol;

Step 3: measure ring by one in high-frequency current clamp and be placed on the live wire and ground wire of digital gravity metering device power lead, with the first noise voltage (V of line over the ground of live wire in power lead in the extraction digital gravity metering device ₁); Another measurement ring in high-frequency current clamp is placed on the center line and ground wire of digital gravity metering device power lead, with the second noise voltage (V of center line to ground wire in power lead in the extraction digital gravity metering device ₂);

Step 4: by described the first noise voltage (V ₁), the second noise voltage (V ₂) in conjunction with the transfer impedance Z (ω) of described high-frequency current clamp, carry out computing, obtain differential mode noise electric current and common mode noise current;

Concrete formula is as follows:

$\left|V_1\right|=\frac{\mathrm{j\&omega;M}\&CenterDot;R_2}{R_2+\mathrm{j\&omega;}{L}_2}(I_{\mathrm{CM}}+I_{\mathrm{DM}})$

$\left|V_2\right|=\frac{\mathrm{j\&omega;M}\&CenterDot;R_2}{R_2+\mathrm{j\&omega;}{L}_2}(I_{\mathrm{CM}}-I_{\mathrm{DM}})$

In formula, R ₂For frequency spectrograph receives impedance, L ₂For the high-frequency current clamp coefficient of self-induction, M is the coefficient of mutual inductance between test circuit and high-frequency current clamp, and ω is angular frequency (rad/s), and j is imaginary part unit, I _CMFor common mode noise current (A), I _DMFor differential mode noise electric current (A);

Step 5: pass through following formula according to common mode noise current:

$E_{\mathrm{\&theta;}}\&ap;j\frac{l{Z}_0{I}_{\mathrm{CM}}{\mathrm{\&beta;}}_0\sin \mathrm{\&theta;}}{4\mathrm{\&pi;r}}{e}^{-j{\mathrm{\&beta;}}_{0}r}$

, obtain and launch the radiated noise electric field to free space, in formula, Z ₀For the free space wave impedance, unit is Ω; L is conductor length, and unit is m; I _CMFor common mode noise current, unit is A; R is measuring distance, and unit is m; β ₀Be 2 π/λ, wherein λ is the correlated frequency signal wavelength, and unit is m; E _θFor the radiated noise electric field, unit is dBuV/m; J is the imaginary part sign; θ is for measuring vector angle;

Described high-frequency current clamp is connected to noise separating network, and this noise separating network is connected to described frequency spectrograph.

2. method according to claim 1 is characterized in that: also comprise: by noise separating network by the first noise voltage (V ₁), the second noise voltage (V ₂) carry out vector and phasor difference computing, obtain differential mode conducted noise voltage (V _DM) and common mode conducted noise voltage (V _CM), with differential mode conducted noise in the discrete conductive noise and common mode conducted noise;

Concrete formula is as follows:

$\left|V_{\mathrm{DM}}\right|=\left|\frac{V_1-V_2}{2}\right|$

$\left|V_{\mathrm{CM}}\right|=\left|\frac{V_1+V_2}{2}\right|$

In formula, V ₁Be the first noise voltage, V ₂Be the second noise voltage, V _DMFor differential mode conducted noise voltage, V _CMFor common mode conducted noise voltage.

3. method according to claim 1, it is characterized in that: described common mode insertion loss and differential mode rejection ratio performance are that the conducted noise separated network is the performance more than 30MHz in frequency of operation.

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