CN101887095A - Method for testing radiated noises of digital gravity metering device - Google Patents

Abstract

The invention discloses a method for testing radiated noises of a digital gravity metering device, which comprises the following steps of: calculating corresponding transfer impedance according to a high-frequency current clamp structure; putting a measuring loop in the high-frequency current clamp on a power cord of the digital gravity metering device so as to extract a first noise voltage and a second noise voltage in the power cord; performing a vector operation and a vector difference operation on the first noise voltage and the second noise voltage by using a noise separation network so as to obtain a difference mode conducted noise voltage and a common mode conducted noise voltage; performing operation on the difference mode conducted noise voltage, the common mode conducted noise voltage and the transfer impedance of the high-frequency current clamp to obtain a difference mode conducted noise current and a common mode conducted noise current; and acquiring an electric field which emits the radiation noises in a free space according to a common mode noise current formula. In the method, the common mode noise current of a single-chip digital gravity metering device is measured by using the high-frequency current clamp and the noise separation network, and the radiated noises are predicted through the common mode noise current.

Description

A kind of radiation noise test method that is used for digital gravimetric equipment

Technical field

The present invention relates to a kind of radiation noise test method that is used for digital gravimetric equipment, belong to electromagnetic compatibility test and power line communication field.

Background technology

Electronic equipment must be tested by corresponding standard before carrying out marketing, and wherein one is exactly the test of electromagnetic radiation interference standard.And mainly cause owing to its transmission cable radiation based on the radiated noise of the digital gravimetric equipment of single-chip microcomputer.And the existing cost that adopts microwave dark room to measure electromagnetic radiation is higher, therefore how at the characteristics of the digital gravimetric equipment of single-chip microcomputer, developing a kind of radiation noise test method short-cut method that is used for digital gravimetric equipment is the problem that the present invention studies.

Summary of the invention

The object of the invention provides a kind of radiation noise test method that is used for digital gravimetric equipment, this method is measured the common mode noise current of the digital gravimetric equipment of single-chip microcomputer earlier by high-frequency current clamp and noise separating network, estimates radiated noise by this common mode noise current again.

For achieving the above object, the technical solution used in the present invention is: a kind of radiation noise test method that is used for digital gravimetric equipment,

May further comprise the steps:

Step 1: test conducted noise separated network common mode in the radiated noise frequency range is inserted the performance of loss and differential mode rejection ratio, to determine that this conducted noise separated network satisfies measurement requirement in the radiated noise frequency range;

Step 2: draw corresponding transfer impedance Z (ω) according to the high-frequency current clamp Structure Calculation, the concrete computing formula of this transfer impedance Z (ω) is:

$Z\left(\mathrm{\&omega;}\right)=\frac{R_2}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">R</figure-callout>}}_2+\mathrm{j\&omega;}{\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">L</figure-callout>}}_2}\&CenterDot;\mathrm{j\&omega;M}$

In the formula, R ₂Be receiver internal impedance (Ω), L ₂Be sense (H) in the high-frequency current clamp, M is the volume mutual inductance (H) between circuit and the high-frequency current clamp, and ω is angular frequency (rad/s), and j is an imaginary part unit;

Step 3 a: measurement ring in the high-frequency current clamp is placed on the live wire and ground wire of digital gravimetric device power supply (DPS) line, to extract in the digital gravimetric equipment in the power lead live wire first noise voltage (V of line over the ground ₁); Another measurement ring in the high-frequency current clamp is placed on the center line and ground wire of digital gravimetric device power supply (DPS) line, to extract in the digital gravimetric equipment in the power lead center line the second noise voltage (V of ground wire ₂);

Step 4: by noise separating network with live wire noise voltage (V _L), center line noise voltage (V _N) carry out vector and with the phasor difference computing, obtain differential mode conducted noise voltage (V _DM) and common mode conducted noise voltage (V _CM), with differential mode conducted noise in the discrete conductive noise and common mode conducted noise;

Concrete formula is as follows:

$\left|V_{\mathrm{DM}}\right|=\left|\frac{V_L-{\mathrm{<figure-callout\; id="2"\; label="V\; N"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">V</figure-callout>}}_N}{2}\right|$

$\left|V_{\mathrm{CM}}\right|=\left|\frac{V_L+{\mathrm{<figure-callout\; id="2"\; label="V\; N"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">V</figure-callout>}}_N}{2}\right|$

In the formula, V _LBe live wire noise voltage, V _NBe center line noise voltage, V _DMBe differential mode conducted noise voltage, V _CMBe common mode conducted noise voltage.

Step 5: with the described first noise voltage (V ₁), the second noise voltage (V ₂) carry out computing in conjunction with the transfer impedance Z (ω) of described high-frequency current clamp, obtain differential mode conducted noise electric current and common mode conducted noise electric current;

Concrete formula is as follows:

$\left|{\mathrm{<figure-callout\; id="2"\; label="V"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">V</figure-callout>}}_2\right|=\frac{\mathrm{j\&omega;M}\&CenterDot;R_2}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">R</figure-callout>}}_2+\mathrm{j\&omega;}{L}_2}(I_{\mathrm{CM}}-I_{\mathrm{DM}})$

$\left|V_1\right|=\frac{\mathrm{j\&omega;M}\&CenterDot;R_2}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">R</figure-callout>}}_2+\mathrm{j\&omega;}{L}_2}(I_{\mathrm{CM}}+I_{\mathrm{DM}})$

In the formula, R ₂Be receiver internal impedance (Ω), L ₂Be sense (H) in the high-frequency current clamp, M is the volume mutual inductance (H) between circuit and the high-frequency current clamp, and ω is angular frequency (rad/s), and j is an imaginary part unit, I _CMBe common mode noise current (A), I _DMBe differential mode noise electric current (A);

Step 6: pass through formula according to following common mode noise current:

$E_{\mathrm{\&theta;}}\&ap;j\frac{{\mathrm{lZ}}_{0}I{\mathrm{\&beta;}}_0\sin \mathrm{\&theta;}}{4\mathrm{\&pi;r}}{e}^{-j{\mathrm{\&beta;}}_{0}r}---\left(3\right)$

, obtain to launch the radiated noise electric field to free space, in the formula, Z ₀Be the free space wave impedance, unit is Ω; L is a conductor length, and unit is m; I is an electric current, and unit is A; R is a measuring distance, and unit is m; β ₀Be 2 π/λ, wherein λ is the correlated frequency signal wavelength, and unit is m; E _θBe the radiated noise electric field, unit is dBuV/m; J is the imaginary part sign; θ is for measuring vector angle.

Related content in the technique scheme is explained as follows:

1, in the such scheme, it be the conducted noise separated network in frequency of operation is performance more than the 30MHz that described common mode is inserted loss and differential mode rejection ratio performance.

2, in the such scheme, described R ₂Be receiver internal impedance (Ω), L ₂Be sense (H) in the high-frequency current clamp, M is the volume mutual inductance (H) between circuit and the high-frequency current clamp, and ω is angular frequency (rad/s), and j is an imaginary part unit.

Because the technique scheme utilization, the present invention compared with prior art has following advantage:

1, the present invention to be measuring the conducted noise in the digital gravimetric equipment of single-chip microcomputer, and it is separated into common-mode noise and differential mode noise, and adopts corresponding squelch measure to make electric line communication system reach standard-required at conduction portion.

2, the present invention estimates for radiated noise according to common mode current, can save great number measurement expense such as utilizing microwave dark room.

3, the present invention adopts separated network to extract common-mode noise, and designs corresponding wave filter and suppress for its common-mode noise, thereby reaches the purpose that suppresses the digital gravimetric radiation of equipment of single-chip microcomputer noise.Make more effective to the radiation EMI Noise Suppression.

4, the present invention uses high-frequency current clamp can measure little electric current to the microampere magnitude.

5, the present invention measure lead between balance to guarantee to optimize the integrality of signal.

Description of drawings

The measurement system diagram that Fig. 1 high-frequency current clamp and noise separating network are formed;

Fig. 2 high-frequency current clamp circuit topological structure figure;

Fig. 3 radio-frequency current probe is estimated principle wherein, and I1, I2...In are every segment corresponding current, and E1, E2...En are the electric field of every section electric current correspondence;

Fig. 4 amplifies the position for causing the radiated noise reason based on the digital gravimetric circuitry of single-chip microcomputer figure;

Fig. 5 high-frequency current clamp transfer impedance test pattern;

Fig. 6 noise separating network high frequency characteristics figure;

Fig. 7 radiated noise estimation results.

Embodiment

Below in conjunction with drawings and Examples the present invention is further described:

Embodiment: a kind of radiation noise test method that is used for digital gravimetric equipment may further comprise the steps:

Step 1: test conducted noise separated network common mode in the radiated noise frequency range is inserted the performance of loss and differential mode rejection ratio, to determine that this conducted noise separated network satisfies measurement requirement in the radiated noise frequency range;

Step 2: draw corresponding transfer impedance Z (ω) according to the high-frequency current clamp Structure Calculation, the concrete computing formula of this transfer impedance Z (ω) is:

$Z\left(\mathrm{\&omega;}\right)=\frac{R_2}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">R</figure-callout>}}_2+\mathrm{j\&omega;}{\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">L</figure-callout>}}_2}\&CenterDot;\mathrm{j\&omega;M}$

In the formula, R ₂For, L ₂For, M is that ω is that j is;

Step 3 a: measurement ring in the high-frequency current clamp is placed on the live wire and ground wire of digital gravimetric device power supply (DPS) line, to extract in the digital gravimetric equipment in the power lead live wire first noise voltage (V of line over the ground ₁); Another measurement ring in the high-frequency current clamp is placed on the center line and ground wire of digital gravimetric device power supply (DPS) line, to extract in the digital gravimetric equipment in the power lead center line the second noise voltage (V of ground wire ₂);

Step 4: by noise separating network with live wire noise voltage (V _L), center line noise voltage (V _N) carry out vector and with the phasor difference computing, obtain differential mode conducted noise voltage (V _DM) and common mode conducted noise voltage (V _CM), with differential mode conducted noise in the discrete conductive noise and common mode conducted noise;

Concrete formula is as follows:

$\left|V_{\mathrm{DM}}\right|=\left|\frac{V_L-{\mathrm{<figure-callout\; id="2"\; label="V\; N"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">V</figure-callout>}}_N}{2}\right|$

$\left|V_{\mathrm{CM}}\right|=\left|\frac{V_L+{\mathrm{<figure-callout\; id="2"\; label="V\; N"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">V</figure-callout>}}_N}{2}\right|$

In the formula, V _LBe live wire noise voltage, V _NBe center line noise voltage, V _DMBe differential mode conducted noise voltage, V _CMBe common mode conducted noise voltage.

Step 5: the just described first noise voltage (V ₁), the second noise voltage (V ₂) carry out computing in conjunction with the transfer impedance Z (ω) of described high-frequency current clamp, obtain differential mode conducted noise electric current and common mode conducted noise electric current;

Concrete formula is as follows:

$\left|V_1\right|=\frac{\mathrm{j\&omega;M}\&CenterDot;R_2}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">R</figure-callout>}}_2+\mathrm{j\&omega;}{L}_2}(I_{\mathrm{CM}}+I_{\mathrm{DM}})$

$\left|{\mathrm{<figure-callout\; id="2"\; label="V"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">V</figure-callout>}}_2\right|=\frac{\mathrm{j\&omega;M}\&CenterDot;R_2}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">R</figure-callout>}}_2+\mathrm{j\&omega;}{L}_2}(I_{\mathrm{CM}}-I_{\mathrm{DM}})$

In the formula, R ₂Be receiver internal impedance (Ω), L ₂Be sense (H) in the high-frequency current clamp, M is the volume mutual inductance (H) between circuit and the high-frequency current clamp, and ω is angular frequency (rad/s), and j is an imaginary part unit, I _CMBe common mode noise current (A), I _DMBe differential mode noise electric current (A);

Step 6: pass through formula according to following common mode noise current:

$E_{\mathrm{\&theta;}}\&ap;j\frac{{\mathrm{lZ}}_{0}I{\mathrm{\&beta;}}_0\sin \mathrm{\&theta;}}{4\mathrm{\&pi;r}}{e}^{-j{\mathrm{\&beta;}}_{0}r}$

, obtain to launch the radiated noise electric field to free space, in the formula, Z ₀Be the free space wave impedance, unit is Ω; L is a conductor length, and unit is m; I is an electric current, and unit is A; R is a measuring distance, and unit is m; β ₀Be 2 π/λ, wherein λ is the correlated frequency signal wavelength, and unit is m; E _θBe the radiated noise electric field, unit is dBuV/m; J is the imaginary part sign; θ is for measuring vector angle.

It is the conducted noise separated network in frequency of operation is performance more than the 30MHz that described common mode is inserted loss and differential mode rejection ratio performance.

Described R ₂Be receiver internal impedance (Ω), L ₂Be sense (H) in the high-frequency current clamp, M is the volume mutual inductance (H) between circuit and the high-frequency current clamp, and ω is angular frequency (rad/s), and j is an imaginary part unit.

Present embodiment foregoing specific explanations is as follows.

High-frequency current clamp and noise separating network extract for the common mode noise current of single-chip microcomputer gravimetric equipment, and then come the new test macro of the radiated noise in the anticipator circuit by common mode noise current.

The test macro that utilizes noise separating network and high-frequency current clamp to form is used for circuit theory such as Fig. 1 of metering circuit electromagnetic interference noise electric current.Wherein, high-frequency current clamp utilizes the mutual inductance effect that actual electromagnetic interference noise in the circuit is sensed in the current clamp, and is received by spectrum analyzer.The equivalent model of high-frequency current clamp metering circuit as shown in Figure 2.According to Fig. 2 circuit, transfer impedance Z (ω) defines as the formula (1):

$Z\left(\mathrm{\&omega;}\right)=\frac{V_2\left(\mathrm{\&omega;}\right)}{I_1\left(\mathrm{\&omega;}\right)}---\left(1\right)$

Wherein, V ₂(ω) magnitude of voltage that measures for high-frequency current clamp, I ₁(ω) be current value in the circuit.Because the mutual inductance effect of high-frequency current clamp, being not difficult to draw the induction electromotive force that the electromagnetic interference (EMI) electric current in the circuit produces in current probe is j ω MI ₁(ω), because the internal impedance of surveying instrument is 50 Ω, identical again with matched impedance.Then current probe measure circuital current in the voltage of gained and the circuit relation as the formula (2):

${\mathrm{<figure-callout\; id="2"\; label="V"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">V</figure-callout>}}_2=\frac{R_2}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">R</figure-callout>}}_2+\mathrm{j\&omega;}{\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">L</figure-callout>}}_2}\&CenterDot;\mathrm{j\&omega;M}{I}_1\left(\mathrm{\&omega;}\right)---\left(2\right)$

Wherein, R ₂Be 50 Ω loaded impedances, j ω MI ₁(ω) magnitude of voltage that obtains by mutual inductance effect induction for voltage probe, L ₁Self-induction for current probe.In circuit because the matched impedance resistance is identical with the surveying instrument internal impedance, so can utilize the magnitude of voltage that measures on the matched impedance come in the counting circuit electric current as the formula (3):

V ₃(ω)＝R ₁·I ₁(ω)＝50·I ₁(ω)????(3)

Wherein, R ₁Be 50 Ω loaded impedances.So (2) formula can be converted to:

${\mathrm{<figure-callout\; id="2"\; label="V"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">V</figure-callout>}}_2=\frac{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">R</figure-callout>}}_2\mathrm{j\&omega;M}}{50({\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">R</figure-callout>}}_2+\mathrm{j\&omega;M})}\&CenterDot;{\stackrel{\&CenterDot;}{V}}_3\left(\mathrm{\&omega;}\right)---\left(4\right)$

Utilize noise separating network, can extract high-frequency current clamp and measure common mode component in the overall noise electric current.Single-chip microcomputer gravimetric device transmission line noise voltage V _L, V _NBe defined as overall noise, by both being carried out vector and can obtaining differential mode noise in the circuit and common-mode noise suc as formula shown in (5) (6) with phasor difference:

$\left|V_{\mathrm{DM}}\right|=\left|\frac{V_L-{\mathrm{<figure-callout\; id="2"\; label="V\; N"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">V</figure-callout>}}_N}{2}\right|=50\left|i_{\mathrm{DM}}\right|---\left(5\right)$

$\left|V_{\mathrm{CM}}\right|=\left|\frac{V_L+{\mathrm{<figure-callout\; id="2"\; label="V\; N"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">V</figure-callout>}}_N}{2}\right|=50\left|i_{\mathrm{CM}}\right|---\left(6\right)$

For the conducted noise electric current,, also it can be divided into differential mode conducted noise electric current and common mode conducted noise electric current according to the electromagnetic compatibility theory.The differential mode noise electric current results between live wire and the center line, and it defines as the formula (7):

$I_{\mathrm{dm}}=\frac{I_L-I_N}{2}---\left(7\right)$

Common mode noise current is a current noise sum on live wire and ground wire and zero line and the ground wire, and it defines as the formula (8):

I _cm＝I _L+I _N????(8)

Flow to as can be known according to the common mode current shown in Fig. 1, because the impedance of reference ground is littler than the inductive impedance value of safety line, so common mode current finally flows to reference ground after by live wire and zero line.Therefore, common mode current is the main cause that causes electromagnetic radiation to disturb.In single-chip microcomputer gravimetric device design,, thereby cause the increase of its radiated noise owing to the existence of circuit intermediate ring road causes existing in the circuit common mode current of this kind form.

By above analysis as can be known, the relation of common mode (CM) in voltage that measures and the circuit/differential mode (DM) electric current can be represented by formula (9):

${\mathrm{<figure-callout\; id="2"\; label="V"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">V</figure-callout>}}_2=\frac{R_2}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">R</figure-callout>}}_2+\mathrm{j\&omega;}{\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">L</figure-callout>}}_2}\&CenterDot;\mathrm{j\&omega;M}{I}_1\left(\mathrm{\&omega;}\right)$

$=\mathrm{j\&omega;M}\frac{(V_{\mathrm{CM}}+V_{\mathrm{DM}})}{50+\mathrm{j\&omega;}{L}_1}\&CenterDot;\frac{R_2}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">R</figure-callout>}}_2+\mathrm{j\&omega;}{\mathrm{<figure-callout\; id="2"\; label="L"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">L</figure-callout>}}_2}$

$=\mathrm{j\&omega;M}\frac{(V_{\mathrm{CM}}+V_{\mathrm{DM}})\&CenterDot;50}{(50+\mathrm{j\&omega;}{L}_1)(50+\mathrm{j\&omega;}{L}_2)}---\left(9\right)$

$=\frac{\mathrm{j\&omega;M}\&CenterDot;50}{50+\mathrm{j\&omega;}{L}_2}(\frac{{\mathrm{<figure-callout\; id="50"\; label="V\; CM"\; filenames="HSA00000185914100041.png"\; state="\{\{state\}\}">V</figure-callout>}}_{\mathrm{CM}}}{50+\mathrm{j\&omega;}{L}_1}+\frac{{\mathrm{<figure-callout\; id="50"\; label="V\; DM"\; filenames="HSA00000185914100041.png"\; state="\{\{state\}\}">V</figure-callout>}}_{\mathrm{DM}}}{50+\mathrm{j\&omega;}{L}_1})$

$=\frac{\mathrm{j\&omega;M}\&CenterDot;50}{50+\mathrm{j\&omega;}{L}_2}(I_{\mathrm{CM}}+I_{\mathrm{DM}})$

Induced voltage that current probe can be measured by this formula and the common mode noise current in the circuit and differential mode noise electric current interrelate, and by noise separating network, can obtain the common mode noise current of single-chip microcomputer gravimetric equipment, thereby estimate its radiated noise.Its concrete implementation step is as follows:

1, at first, the transfer impedance for current probe in frequency domain is tested, and considers that the equivalent electrical circuit of current probe is comprising self-induction and two aspects of mutual inductance, so can write out current probe partial circuit equation, is not difficult to derive V ₂(ω) and V ₁Relation (ω) is shown in (4) formula.

2, test for the performance of conducted noise separated network in the radiation frequency range, to determine whether its performance can satisfy measurement requirement in the radiation frequency range.

3, set up single-chip microcomputer numeral gravimetric equipment common mode noise current extraction system, this system is by high-frequency current clamp, digital gravimetric equipment, spectrum analyzer and (see figure 1) that noise separating network constitutes.High-frequency current clamp 1 is clipped on live wire and the ground wire, with the flow through electromagnetic interference noise electric current I L of live wire and ground wire of measurement; Simultaneously another high-frequency current clamp 2 is clipped on center line and the ground wire, with the flow through electromagnetic interference noise electric current I N of center line and ground wire of measurement.The output terminal of two high-frequency current clamp is inserted noise separating network, in order to the common-mode noise in the extraction system.And common-mode noise is converted into common mode noise current, can estimate the radiated noise of this system according to the common mode noise current of extracting.The electromagnetic radiation of the digital gravimetric equipment of single-chip microcomputer is mainly derived from the common mode radiation on the circuit, and common mode radiation derives from high frequency common mode current, and common mode current is launched radiation field along transmission line to free space, the field intensity computing formula as the formula (10):

$E_{\mathrm{\&theta;}}\&ap;j\frac{{\mathrm{lZ}}_{0}I{\mathrm{\&beta;}}_0\sin \mathrm{\&theta;}}{4\mathrm{\&pi;r}}{e}^{-j{\mathrm{\&beta;}}_{0}r}---\left(10\right)$

Wherein: Zo is free space wave impedance (Ω), and l is conductor length (m), and I is electric current (A), and r is measuring distance (m), and β o=2 π/λ, λ are correlated frequency signal wavelength (m).

Along with the increase of frequency, the physical length of lead will be comparable with the physical dimension of wavelength, then along the uniformity no longer of the distribution of current on the lead.Lead evenly can be divided into N segment for this reason, adopt the radio frequency high-frequency current clamp to measure its electric current (see figure 3) separately in the centre position of every segment, establish and be respectively I1, I2...In can adopt formula (10) to calculate for each section lead.

Adopt formula (10) to calculate the radiation field of each segment.Because the standard place of electromagnetic radiation is normally measured at open testing field (OATS), so need to consider the ground return effect, then total equivalent computation of radiation field is converted into as the formula (11).

$\left|E_c\right|\&ap;\frac{2\mathrm{\&pi;f}\&times;{10}^{-7}F(I_1+I_2+...+I_n)l}{3\sqrt{{\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="HSA00000185914100022.png"\; state="\{\{state\}\}">r</figure-callout>}}^2+{(H-0.8)}^2}}---\left(11\right)$

Wherein, | EC| is a radiated electric field, and 1 is every segment antenna eliminator length, and f represents test frequency, and r represents open testing field standard testing distance, and H represents the test antenna height, and F is for calculating the modifying factor under the test environment of open testing field.

4, because digital gravimetric device transmission cable is longer, therefore the test macro that can utilize high-frequency current clamp and noise separating network to form extracts for the common mode noise current of zones of different on the transmission cable, with the optimum position of determining that braking measure loads.

5, adopt ferrite bead to suppress, by relatively loading before and after the ferrite bead on the cable attenuation degree of radiated noise to check the effect of method in the literary composition for the radiated noise in the single-chip microcomputer numeral gravimetric equipment.

The method of testing that the present invention adopts can be estimated at the radiation EMI noise on all kinds of single-chip microcomputer numeral gravimetric device transmission cables.Estimate thereby adopt high-frequency current clamp and noise separating network to extract for the radiation EMI noise for common mode noise current on the cable, this method is simple to operate and test is accurate.By this method of testing, can test the radiation EMI noise of single-chip microcomputer numeral gravimetric device transmission line, further provide theoretical foundation for suppressing this radiation EMI noise.

Its circuit as shown in Figure 4, it is because due to the 12MHz clock circuit is coupled in the transmission line in the circuit that its radiated noise produces reason.Set up test macro as shown in Figure 1, this test macro is made up of high-frequency current clamp, noise separating network, spectrum analyzer and digital gravimetric equipment etc.

Concrete method of testing:

In time domain, its experiment connects as shown in Figure 5, this experiment utilizes signal generator to export the sinusoidal signal of different frequency in required measurement frequency range in a port of dual trace oscilloscope, simultaneously high-frequency current clamp is carried on the transmission line, the value that measures is input to the another one port of dual trace oscilloscope, according to 3 formulas, can obtain on different frequency point the transfer impedance of this current probe.

The noise separating network that is adopted is measured in the performance of radiation frequency range, can be applied in the middle of the estimating of radiated noise to determine this noise separating network.Judge that the good and bad leading indicator of common-mode noise separated network performance is that its common mode is inserted loss (CMIL) and its definition of differential mode rejection ratio (DMRR) as the formula (12):

$\mathrm{CMIL}=20\lg \left(\frac{V_{\mathrm{OC}}}{V_{\mathrm{CM}}}\right)$ $\mathrm{DMRR}=20\lg \left(\frac{V_{\mathrm{OC}}}{V_{\mathrm{DM}}}\right)---\left(12\right)$

Wherein, V _CMBe common mode input, V _OCBe common mode output; V _DMBe difference-mode input, V _ODBe differential mode output.In the common mode separated network, the ideal value that common mode is inserted loss should be 0, to guarantee that the loss of its common mode noise signal is less in the common-mode noise transmission; In like manner as can be known, its differential mode rejection ratio should be big as far as possible, can not be coupled in the transmission of common mode noise signal to guarantee its differential mode noise, so the ideal value of its differential mode rejection ratio should be-∞.By test, can obtain its common mode insertion loss and differential mode rejection ratio test result as shown in Figure 6.This result shows that this separated network can apply to estimating of radiation EMI frequency range.

At the test noise separated network on the basis of radiation frequency range performance, can the connecting circuit system.Digital gravimetric equipment is switched on, and high-frequency current clamp 1 is clipped on live wire and the ground wire; Another high-frequency current clamp is clipped on center line and the ground wire, to measure live wire noise and the center line noise V on the cable _L, V _NIts frequency that exceeds standard is 36MHz (as shown in Figure 7).The output port of high-frequency current clamp is linked in the noise separating network, with in the noise input spectrum analyser after separating to analyze the common mode noise current of single-chip microcomputer gravimetric equipment.

High-frequency current clamp is placed in different places on the digital gravimetric equipment cable, to determine the position of radiation maximum on the cable.Be divided into nearly power end in this case, mid point and nearly equipment end three places are detected, and its testing result is as shown in table 1:

The digital gravimetric equipment of table 1 Conducted EMI noise separation result

Filter location	Nearly power end	Mid point	Nearly equipment end
				The radiated noise amplitude	??42dBuV	??42.9dBuV	??48dBuV

Shown in table, it suppresses the position should be at nearly equipment end place.

The foregoing description only is explanation technical conceive of the present invention and characteristics, and its purpose is to allow the personage who is familiar with this technology can understand content of the present invention and enforcement according to this, can not limit protection scope of the present invention with this.All equivalences that spirit is done according to the present invention change or modify, and all should be encompassed within protection scope of the present invention.

Claims (4)

1. radiation noise test method that is used for digital gravimetric equipment is characterized in that: may further comprise the steps:

Step 1: test conducted noise separated network common mode in the radiated noise frequency range is inserted the performance of loss and differential mode rejection ratio, to determine that this conducted noise separated network satisfies measurement requirement in the radiated noise frequency range;

Step 2: draw corresponding transfer impedance Z (ω) according to the high-frequency current clamp Structure Calculation, the concrete computing formula of this transfer impedance Z (ω) is:

$Z\left(\mathrm{\&omega;}\right)=\frac{R_2}{R_2+\mathrm{j\&omega;}{L}_2}\&CenterDot;\mathrm{j\&omega;M}$

In the formula, R ₂For frequency spectrograph receives impedance, L ₂Be the high-frequency current clamp coefficient of self-induction, M is the coefficient of mutual inductance between test circuit and the high-frequency current clamp, and ω is an angular frequency, and j is the imaginary part of symbol;

Step 3 a: measurement ring in the high-frequency current clamp is placed on the live wire and ground wire of digital gravimetric device power supply (DPS) line, to extract in the digital gravimetric equipment in the power lead live wire first noise voltage (V of line over the ground ₁); Another measurement ring in the high-frequency current clamp is placed on the center line and ground wire of digital gravimetric device power supply (DPS) line, to extract in the digital gravimetric equipment in the power lead center line the second noise voltage (V of ground wire ₂);

Step 4: with the described first noise voltage (V ₁), the second noise voltage (V ₂) carry out computing in conjunction with the transfer impedance Z (ω) of described high-frequency current clamp, obtain differential mode conducted noise electric current and common mode conducted noise electric current;

Concrete formula is as follows:

$\left|V_1\right|=\frac{\mathrm{j\&omega;M}\&CenterDot;R_2}{R_2+\mathrm{j\&omega;}{L}_2}(I_{\mathrm{CM}}+I_{\mathrm{DM}})$

$\left|V_2\right|=\frac{\mathrm{j\&omega;M}\&CenterDot;R_2}{R_2+\mathrm{j\&omega;}{L}_2}(I_{\mathrm{CM}}-I_{\mathrm{DM}})$

In the formula, R ₂Be receiver internal impedance (Ω), L ₂Be sense (H) in the high-frequency current clamp, M is the volume mutual inductance (H) between circuit and the high-frequency current clamp, and ω is angular frequency (rad/s), and j is an imaginary part unit, I _CMBe common mode noise current (A), I _DMBe differential mode noise electric current (A);

Step 5: pass through formula according to following common mode noise current:

$E_{\mathrm{\&theta;}}\&ap;j\frac{l{Z}_{0}I{\mathrm{\&beta;}}_0\sin \mathrm{\&theta;}}{4\mathrm{\&pi;r}}{e}^{-j{\mathrm{\&beta;}}_{0}r}$

, obtain to launch the radiated noise electric field to free space, in the formula, Z ₀Be the free space wave impedance, unit is Ω; L is a conductor length, and unit is m; I is an electric current, and unit is A; R is a measuring distance, and unit is m; β ₀Be 2 π/λ, wherein λ is the correlated frequency signal wavelength, and unit is m; E _θBe the radiated noise electric field, unit is dBuV/m; J is the imaginary part sign; θ is for measuring vector angle.

2. method according to claim 1 is characterized in that: also comprise: by noise separating network with live wire noise voltage (V _L), center line noise voltage (V _N) carry out vector and with the phasor difference computing, obtain differential mode conducted noise voltage (V _DM) and common mode conducted noise voltage (V _CM), with differential mode conducted noise in the discrete conductive noise and common mode conducted noise;

Concrete formula is as follows:

$\left|V_{\mathrm{DM}}\right|=\left|\frac{V_L-V_N}{2}\right|$

$\left|V_{\mathrm{CM}}\right|=\left|\frac{V_L+V_N}{2}\right|$

In the formula, V _LBe live wire noise voltage, V _NBe center line noise voltage, V _DMBe differential mode conducted noise voltage, V _CMBe common mode conducted noise voltage.

3. method according to claim 1 is characterized in that: it be the conducted noise separated network in frequency of operation is performance more than the 30MHz that described common mode is inserted loss and differential mode rejection ratio performance.

4. method according to claim 1 is characterized in that: described internal impedance is R ₂Be receiver internal impedance (Ω), L ₂Be sense (H) in the high-frequency current clamp, M is the volume mutual inductance (H) between circuit and the high-frequency current clamp, and ω is angular frequency (rad/s), and j is an imaginary part unit.

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