CN101010871B - Receiver and method for wireless communications terminal - Google Patents

Abstract

A wireless receiver (200) for receiving and demodulating a frequency modulated RF signal by a direct conversion procedure comprising an input signal path (101), a local oscillator (111), a first mixer(107) for mixing an output reference signal from the local oscillator with an input received signal to produce an in-phase component, a quadrature phase shifter (113) connected to the input signal path a second mixer (109) for mixing an output signal from the phase shifter with the input received signal to produce a quadrature component; and means (218) for producing an output demodulated information signal by producing a combining function of the in-phase and quadrature components; and characterised by means (214) for periodically detecting an error in the relative phase difference between the in-phase and quadrature components and for applying (217) a relative adjustment in phase difference to compensate for the detected error.

Description

The receiver and the method that are used for wireless communication terminal

Technical field

The terminal that the present invention relates to a kind of receiver that in radio communication, uses and method and use it.Particularly, the present invention relates to directly change receiver, it can come (radio frequency) signal of frequency, demodulation frequency modulation (FM) by homophase (I) and quadrature (Q) component that decomposes (resolution) and use modulation signal.

Background technology

The traditional F M wireless receiver that uses direct converting structure to detect the I of received signal and Q component and make up has the problem of the following stated.As described later, this receiver can produce error in relative phase between I and the Q component and amplitude.This error is sometimes referred to as " orthogonal unbalance ", can cause the distortion of gained output audio signal.This distortion under the condition when received signal declines or has low signal to noise ratio, is unacceptable particularly for the user.Prior art does not provide the solution for the satisfaction of this problem.

Summary of the invention

According to a first aspect of the invention, provide a kind of radio receiver.

According to a second aspect of the invention, provide a kind of wireless communications method.

According to a third aspect of the invention we, provide a kind of wireless communication terminal.

With reference to the accompanying drawings, embodiments of the invention are described in the mode of example, wherein:

Description of drawings

Fig. 1 is the theory diagram circuit diagram of known direct conversion RF receiver.

Fig. 2 is the inner product L that calculated and the figure of unbalance phase angle α, and useful in an embodiment of the present invention relation is described;

Fig. 3 is the theory diagram circuit diagram that embodies direct conversion RF receiver of the present invention.

Fig. 4 embodies of the present invention another directly to change the theory diagram circuit diagram of RF receiver.

Fig. 5 is detector output and the figure that local oscillator frequencies is offset, and relation useful in the circuit shown in Fig. 4 is described.

Fig. 6 be to use the simulation that embodies signal processing of the present invention actual unbalance with the figure of the unbalance phase imbalance angle of calculating and time-gap number (algorithm circulation).

Fig. 7 is to use the simulation that embodies signal processing of the present invention actual unbalance and the unbalance amplitude imbalance of calculating and the figure of time-gap number.

Embodiment

Fig. 1 illustrates known RF and directly changes FM receiver 100, and problem to be solved by this invention is described.The FM signal x (t) that imports into is delivered to two frequency mixers 107,109 respectively via the input path 101 with branch road connection 103,105.Local oscillator 111 generates the reference signal that has with input signal x (t) carrier frequency same frequency.First component of reference signal is applied directly to frequency mixer 107, and it and input signal x (t) multiply each other in this frequency mixer 107.The phase shift output that the second component of reference signal is applied to phase shifter 113 and phase shifter 113 is applied to frequency mixer 109, and it and input signal x (t) multiply each other in this frequency mixer 109.Although being intended to introduce, phase shifter 113 and frequency mixer 107 and 109 combinations have the phase shift of gain for 90 degree of " 1 " between the reference signal components that is applied to frequency mixer 107 and 109, but, in fact introduced phase shift slightly different and the gain slightly different with " 1 " with 90 degree.Output signal from frequency mixer 107 is passed through low pass filter (LPF) 115, exports in-phase component signal I (t) to produce, and passes through low pass filter (LPF) 117 from the output signal of frequency mixer 109, to produce output orthogonal component signal Q (t).Amplitude unbalance that is incorporated into frequency mixer 109 output is shown unbalance gain A in frame 119.

Mathematical analysis in this structure shown in Fig. 1 is as follows:

Input signal can be expressed as:

x(t)＝cos(ωt+φ(t)+γ)

Wherein be the RF carrier frequency of ω input rf signal x (t), γ is the oscillator arbitrary phase, and φ (t) is the frequency modulation(FM) of x (t) to be detected.

In addition, x (t)=I (t)+j*Q (t), wherein I (t) and Q (t) they are homophase and the quadrature component of x (t).

Wherein A represents amplitude imbalance, and a represents the phase imbalance angle between the phase angle of I (t) and Q (t).

With the embodiments of the invention of explanation, periodically handle I (t) and Q (t) in the mode of explanation after a while according to after a while, estimating and to eliminate that these are unbalance, and the adjusting component that makes up gained constructs modulation signal φ (t), so that audio signal output to be provided.

Following analysis illustrates how to determine phase imbalance a.

Consider the inner product (inner product) of I (t) and Q (t).Provide by following equation:

L＝Z+X

Based on the required value of the immunity to interference of noise being selected T.

Under normal conditions, X is not equal to zero.Yet, under given conditions X＜＜Z, that is:

$X=A\cos \left(\mathrm{\α}\right)\frac{1}{T}\underset{0}{\overset{T}{\∫}}\sin (\mathrm{\φ}\left(t\right)+\mathrm{\γ})\cos (\mathrm{\φ}\left(t\right)+\mathrm{\γ})\mathrm{dt}=0.$

Two examples in this case are as follows:

1. for tone FM modulation (audio tones of FM modulation); For example use the 150Hz PL tone of 500Hz offset modulation) and γ=0 we obtain X＜＜Z.I and Q quadrature (L ≈ Z) so.

2. for tone FM modulation and big modulation index, we obtain X＜＜Z (L ≈ Z). this is correct for any γ, is wrong for any local oscillator frequency error still.If local oscillator (LO) has 0, f _m/ 2, f _m, 3f _mThe frequency error of/2 grades, X will be not equal to zero so.Yet, if can detect this frequency error, can regulate the LO frequency, to overcome the problem of explanation after a while.

Therefore, if satisfy the condition that makes X=0 as these two examples, we obtain:

$L=A\cos \left(\mathrm{\α}\right)\frac{1}{T}\underset{0}{\overset{T}{\∫}}{\cos }^2(\mathrm{\φ}\left(t\right)+\mathrm{\γ})\mathrm{dt}$

For any value of φ (t), expression formula

Be restricted to 1.For α=0, L=0 also is like this.

Fig. 2 illustrates the relation between L and the α.Curve C 1 shown in Figure 2, it is the L (V of unit ²) be the figure of phase angle (α) PH2 of unit with the degree.As shown in Figure 2, when PH2 reached minimum value PH2_opt, curve C 1 reached trough.This is L (PH2_opt) corresponding to the minimum value of L.

Therefore, in two above-mentioned examples, we learn that so unbalance phase angle α is a minimum value if L is a minimum value.Therefore, use these examples, we are intended to minimize L by adaptive sampling and adjustment process.The embodiment adaptive configuration of the present invention of these examples is used in explanation in the example 1 and 2 below, is used for to find the minimum value of L.

Example 1

The circuit that in this example, uses 200 shown in Figure 3.In Fig. 3, have with the element of same reference numerals in Fig. 1 and have identical functions.By connecting 201, sampling is by the output signal I (t) of low pass filter (LPF) 115, and by connecting 203, sampling is by the output signal Q (t) of low pass filter (LPF) 117.Each sampled signal is provided as each input of processor 205,205 squares of each inputs of this processor, and estimate that (estimate) goes out the value of factor A, this factor A is the amplitude imbalance of estimating.It is determined by following formula:

$\tilde{A}=\frac{\mathrm{\&Sigma;}{\mathrm{<figure-callout\; id="2"\; label="Q"\; filenames="H2005800173955E00042.png,H2005800173955E00043.png"\; state="\{\{state\}\}">Q</figure-callout>}}^2}{\mathrm{\&Sigma;}{I}^2}$

The output signal of from processor 205 is amplitude imbalance correction signals, and it points out the value of 1/A.Via connecting 202 this correction signal is applied to amplitude modifier 207, this amplitude modifier 207 is with the factor of the amplitude correction 1/A of Q (t), to eliminate the amplitude imbalance A that is detected.

By connecting 210, the output signal of passing through low pass filter (LPF) 115 of sampled representation component I (t), this connection 210 has first branch road 223, and this first branch road 223 is connected to the DC estimator (estimator) 225 of estimating the DC value.By connecting 208, sampling is by the output signal Q (t) of low pass filter (LPF) 117, and this connection 208 has branch road 217, and this branch road 217 is connected to the DC estimator 219 of the DC value of estimated signal Q (t).By " DC value ", we refer to I (f) suitable when f=0 or the value of Q (f), and wherein I (f) is the Fourier transform of signal I (t), and Q (f) is the Fourier transform of signal Q (t).Output signal from DC estimator 219 and 225 is delivered to arbitrary phase estimator 221, and this arbitrary phase estimator 221 uses these two signals to estimate arbitrary phase angle γ in the following manner.By connecting 227, the output signal from arbitrary phase estimator 221 of representing arbitrary phase angle γ is offered processor 211 and 215, be further described below.Connect 210 and also be directly connected to processor 215.Connect 209 and be also connected to phase shifter PH2209, next phase shifter PH2209 is connected to processor 211.Processor 211,215 computing functions

Wherein i is the subscript of γ.Each value α for α _i, processor 211 calculates γ _i Processor 211 and 215 is for each value α _iWith Q (t) and I (t) phase shift γ _iFrom processor phase shifter 211 and 215 output signal are multiplied each other by frequency mixer 213, obtain offering the output signal of further processor 212, and this processor 212 is for each α _iCalculating parameter ' L '.The output signal of from processor 212 is represented aforementioned parameters L, and is applied in to storage and processing unit, and this storage and processing unit correspondingly write down the value of L.

Via connecting 216, the phase shift control signal is applied to phase shifter PH2 from storage and processing unit 214.The phase shift control signal is operating as at phase shifter PH2 and applies phase shift, and this phase shift has such phase shift angle, promptly, this angle is in single sweep (perhaps many scannings, wherein stride is becoming littler from the process that scans scanning) in, from-5 spend+strides of 5 degree, with 0.2 degree stride change.For each the phase shift angle value that is applied, 214 monitor the respective value of the L that generates at processor 212 in the unit, and write down the phase shift angle value that provides the L minimum value.This is corresponding to the minimum value of aforesaid α.Via connecting 229, will 214 be applied to phase shifter PH1231 from the unit corresponding to the phase shift control signal of the identical and inverse value of the phase angle that this calculated.Via connecting 208, will be applied to phase shifter PH1231 from low pass filter 117 corresponding to the signal of quadrature component Q (t).Phase shifter 231 applies the phase angle adjustment thus, the phase imbalance angle [alpha] that its compensation is detected.The output from phase shifter PH1231 corresponding to the phase place correction value of Q (t) is applied to processor 218.Via connecting 224, also will be applied to processor 218 as input corresponding to the signal of in-phase component I (t).Processor 218 calculates the value of discussing Q (t)/I (t) from its each input, and the signal of ecbatic is provided to processor 230.Processor 230 calculates arc tangent (arctg) value of the input signal of from processor 218.The output signal of from processor 230 is applied to another processor 232, the differential that this processor 232 calculates with respect to the time t of input signal.At last, the output signal of from processor 232 is applied to audio frequency output.Audio frequency output 233 comprises converter, audio tweeter for example, and this audio tweeter is converted to for example audio signal of voice messaging with the electronic signal output of from processor 232.

Be further to explain the also mathematical analysis of the operation of the circuit 200 shown in the key diagram 3 below.

(i) estimation of arbitrary phase angle γ:

We study the DC value of I and Q component:

$I_{\mathrm{DC}}=\frac{1}{T}\underset{0}{\overset{T}{\&Integral;}}\cos (\mathrm{\&phi;}\left(t\right)+\mathrm{\&gamma;})\mathrm{dt}=\cos \left(\mathrm{\&gamma;}\right)J_0\left(\mathrm{\&beta;}\right)$

$Q_{\mathrm{DC}}=\frac{1}{T}\underset{0}{\overset{T}{\&Integral;}}\sin (\mathrm{\&phi;}\left(t\right)+\mathrm{\&alpha;}+\mathrm{\&gamma;})\mathrm{dt}=\sin (\mathrm{\&alpha;}+\mathrm{\&gamma;})J_0\left(\mathrm{\&beta;}\right)$

J wherein ₀(.) is first kind of rank, zero Bessel equations.

Suppose we wish with+/-accuracy of α estimates arbitrarily angled γ.Therefore we can be similar to following expression:

$I_{\mathrm{DC}}=\frac{1}{T}\underset{0}{\overset{T}{\&Integral;}}\cos (\mathrm{\&phi;}\left(t\right)+\mathrm{\&gamma;})\mathrm{dt}=\cos \left(\mathrm{\&gamma;}\right)J_0\left(\mathrm{\&beta;}\right)$

$Q_{\mathrm{DC}}\&ap;\frac{1}{T}\underset{0}{\overset{T}{\&Integral;}}\sin (\mathrm{\&phi;}\left(t\right)+\mathrm{\&alpha;}+\mathrm{\&gamma;})\mathrm{dt}=\sin \left(\mathrm{\&gamma;}\right)J_0\left(\mathrm{\&beta;}\right)$

Use two last equatioies, can estimate γ by following calculating:

$\mathrm{\&gamma;}=\arctan \left(\frac{Q_{\mathrm{DC}}}{I_{\mathrm{DC}}}\right)$

This is carried out by the arbitrary phase estimator among Fig. 3 221.

The (ii) application of arbitrary phase angle correct:

For each α that applies by phase shifter PH2 _i, calculate corresponding γ _i

Therefore, for each α _i, the correction below carrying out:

$\left\{\right[I\cos \left({\mathrm{\&alpha;}}_i\right)-Q\sin \left({\mathrm{\&alpha;}}_i\right)]+\mathrm{jQ}\}{e}^{j{\mathrm{\&gamma;}}_i}$

Wherein

Calculating be the function of processor 211 and 215.

In fact the phase shift of being introduced by phase shifter PH2 (and by phase shifter PH1) according to following mathematical analysis realization:

I _corr＝I _in?cos(α _i)-Q _in?sin(α _i)

Q _corr＝Q _in

So I _Corr+ jQ _Corr={ [I _InCos (α _i)-Q _InSin (α _i)]+jQ}

I wherein _InAnd Q _InIt is input to PH2.

I _CorrAnd Q _CorrBe the output of PH2.In order to simplify, (Fig. 3) shows PH2 (and PH1) with the Q path in the accompanying drawings, but actual realization is to use top last group equation.

For the L=minimum value, we obtain:

$\widehat{\mathrm{\&alpha;}}={\mathrm{\&alpha;}}_i(L=\min )$

$\widehat{\mathrm{\&gamma;}}={\mathrm{\&gamma;}}_i(L=\min )$

For the value of above-mentioned α and γ, I and Q are quadratures.We suppose f _mThe signaling tone of frequency, it is modulated according to industrial standard TIA603FM.This is industrial standard and " Land Mobile FM or PM Communications Equipment andPerformance Standards " (" FM or PM communication equipment and performance standard are moved in land ") by name of being announced by TIA, it comprises following regulation: " CDCSS (" Continuous Digital Controlled SquelchSystem " (" squelch system of continuous number control ")) should define such system; promptly (one or more) radio receiver is equipped with tone or data response apparatus in this system; this tone or data response apparatus only when receiving the carrier wave that uses specific tone or data pattern modulation, allow audio signal to appear at the receiver audio output; to select for example speech processes of scramble; select between voice or data; perhaps control repeater function.For continuous audio frequency output, this tone or data pattern must be to occur continuously.As in CTCSS system (" continuous Tone controlled Squelch System " (" squelch system of continuous tone control ")), should use continuous tone to come the transmitter of modulate emission carrier wave, its frequency is with identical at the required audio frequency of receiver output operating audio response CTCSS equipment.In the CDCSS system, should be in a similar manner, use N continuous RZ fsk data stream to come the transmitter of modulate emission carrier wave, with detector in receiver output operating data sensitivity with correct pattern.The purpose of the system that limits is to minimize to listen to be directed to other people noise of communication of sharing same carrier frequencies or channel.By using specific tone or data system, can encode his carrier wave of each user prevents the carrier wave received audio signal of any not coding or different coding.

Therefore CTCSS/CDCSS is to use the sub-audio signaling of above-mentioned TIA agreement.

It is the waveform that needed to be used for forbidding receiver audio output before the RF carrier wave removes that CDCSS closes (turn off) sign indicating number.It is as noise elimination tail tag or noise eliminator.In order to realize this point, the CDCSS encoder should launch 134.4+/-the 0.5Hz tone continues 150 to 200 milliseconds.It can also be the PL/DPL tone.(PL=special circuit, DPL=digital private circuit).PL/DPL and sub-audio signaling are used to open the receiver noise elimination.PL/DPL is and the parallel emission of voice.

We can by when do not have speech activity during, this regulation is deferred in the application self-adapting correction, although even in fact our phase place adjustment algorithm of finding us is also worked under the situation of audio speech well existing.

For the I channel:

$I\left(t\right)\&ap;\cos \left(\mathrm{\&phi;}\left(t\right)\right)=\cos \left(\mathrm{\&beta;}\sin \left({\mathrm{\&omega;}}_{m}t\right)\right)=J_o\left(\mathrm{\&beta;}\right)+2\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)$

Close sign indicating number for the CDCSS audio frequency, modulation index is

${\mathrm{\&beta;}}_{\mathrm{CDCSS}\_\mathrm{Audio}\_\mathrm{turn}\_\mathrm{offf}}=\frac{500\mathrm{Hz}}{134.4\mathrm{Hz}}=3.72$

Bessel function J _o(β _{CDCSS_Audio_turn_offf})=J _o(3.72)=-0.4

Therefore close the situation of sign indicating number for the CDCSS audio frequency

$I\left(t\right)\&ap;2\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)$

For the Q channel:

$Q\left(t\right)\&ap;\sin (\mathrm{\&phi;}\left(t\right)-\mathrm{\&alpha;})=\cos ({\mathrm{\&beta;}}_d\sin \left({\mathrm{\&omega;}}_{m}t\right)-\mathrm{\&alpha;})=\sin \left(\mathrm{\&beta;}\sin \left({\mathrm{\&omega;}}_{m}t\right)\right)\cos \left(\mathrm{\&alpha;}\right)-$

$-\cos \left(\mathrm{\&beta;}\sin \left({\mathrm{\&omega;}}_{m}t\right)\right)\sin \left(\mathrm{\&alpha;}\right)=$

$=\cos \left(\mathrm{\&alpha;}\right)\left[2\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_{2k-1}\left(\mathrm{\&beta;}\right)\sin \left((2k-1){\mathrm{\&omega;}}_{m}t\right)\right]-\sin \left(\mathrm{\&alpha;}\right)[J_o\left(\mathrm{\&beta;}\right)+2\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)]$

Therefore close the situation of sign indicating number for the CDCSS audio frequency

$Q\left(t\right)\&ap;\cos \left(\mathrm{\&alpha;}\right)\left[2\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_{2k-1}\left(\mathrm{\&beta;}\right)\sin \left((2k-1){\mathrm{\&omega;}}_{m}t\right)\right]-\sin \left(\mathrm{\&alpha;}\right)\left[2\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)\right]$

As mentioned above, $L=\frac{1}{T}\underset{0}{\overset{T}{\&Integral;}}I\left(t\right)Q\left(t\right)\mathrm{dt}$

$L=\frac{1}{T}\underset{0}{\overset{T}{\&Integral;}}\begin{array}{c}\left\{2\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)\right\}\left\{\cos \left(\mathrm{\&alpha;}\right)\right[2\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_{2k-1}\left(\mathrm{\&beta;}\right)\sin \left((2k-1){\mathrm{\&omega;}}_{m}t\right)]\\ -\sin \left(\mathrm{\&alpha;}\right)[2\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)\end{array}\}\mathrm{dt}$

$L=\frac{1}{T}\underset{0}{\overset{T}{\&Integral;}}\begin{array}{c}\left\{2\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)\right\}\left\{\cos \left(\mathrm{\&alpha;}\right)\right[2\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_{2k-1}\left(\mathrm{\&beta;}\right)\sin \left((2k-1){\mathrm{\&omega;}}_{m}t\right)]\\ -\sin \left(\mathrm{\&alpha;}\right)\left[2\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)\right]\}\mathrm{dt}\end{array}$

$L=\frac{1}{T}\underset{\mathrm{<figure-callout\; id="4"\; label="o\; T"\; filenames="H2005800173955E00042.png,H2005800173955E00043.png"\; state="\{\{state\}\}">o</figure-callout>}}{\overset{T}{\&Integral;}}\begin{array}{c}4\cos \left(\mathrm{\&alpha;}\right)\left[\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_{2k-1}\left(\mathrm{\&beta;}\right)\sin \left((2k-1){\mathrm{\&omega;}}_{m}t\right)\right]\\ -4\sin \left(\mathrm{\&alpha;}\right)\left[\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)\right]\left[\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)\right]\mathrm{dt}\end{array}$

Because the integration of orthogonal function, the first of L is zero.

Therefore:

$L=4\sin \left(\mathrm{\&alpha;}\right)\frac{1}{T}\underset{o}{\overset{T}{\&Integral;}}\left[\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)\right]\left[\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J}_k\left(\mathrm{\&beta;}\right)\cos \left(2k{\mathrm{\&omega;}}_{m}t\right)\right]\mathrm{dt}$

Once more, because integration is applied to orthogonal function:

$L=4\sin \left(\mathrm{\&alpha;}\right)\frac{1}{T}\underset{o}{\overset{T}{\&Integral;}}\left[\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J^2}_k\left(\mathrm{\&beta;}\right){\cos }^2\left(2k{\mathrm{\&omega;}}_{m}t\right)\right]\mathrm{dt}$

$L=4\sin \left(\mathrm{\&alpha;}\right)\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J^2}_k\left(\mathrm{\&beta;}\right)\frac{1}{T}\underset{o}{\overset{T}{\&Integral;}}\left[{\cos }^2\left(2k{\mathrm{\&omega;}}_{m}t\right)\right]\mathrm{dt}$

But $\frac{1}{T}\underset{o}{\overset{T}{\&Integral;}}{\cos }^2\left(2k{\mathrm{\&omega;}}_{m}t\right)\mathrm{dt}=\frac{1}{2}$

Therefore we obtain:

$L=2\sin \left(\mathrm{\&alpha;}\right)\underset{k=1}{\overset{\&infin;}{\mathrm{\&Sigma;}}}{J^2}_k\left(\mathrm{\&beta;}\right)$

We can see from this last equation, for α=0, L=0.In this example, L=0 can be reduced to X=0.By the method for unit 214 with reference Fig. 3 description, scanning (sweep) by applying the phase place set-up procedure via phase shifter PH2 and in unit 214 output of recording processor 212 when provide minimum value, determine when X=0, perhaps more specifically, determine when that X is minimum.

(iii) demodulation is to provide audio output signal

Can use down relation of plane to make up audio output signal:

$\mathrm{audio}=\frac{d}{\mathrm{dt}}\left[\arctan \right\{\frac{Q\left(t\right)}{I\left(t\right)}\left\}\right]=\frac{d}{\mathrm{dt}}\left[\arctan \right\{\frac{A\sin (\mathrm{\&phi;}\left(t\right)+a)}{\cos \left(\mathrm{\&phi;}\left(t\right)\right)}\left\}\right]$

By aforesaid processor 218,230 and 233, in circuit 200, carry out the calculating of last expression formula.

Example 2

The circuit that in this example, uses shown in Figure 4.In this example, use the tone FM modulation of big modulation index, be used to provide above-mentioned L=X.Have among Fig. 4 with Fig. 1 or Fig. 3 in the element of element same reference numerals have and these element identical functions, will not describe once more.In Fig. 4, connect 305 and be directly connected to frequency mixer 213, and connect 327 and be directly connected to frequency mixer 213 from low pass filter 115 from phase shifter PH2209.Processor 212 carries out the calculating of L, is used for finding by the phase adjustment value that scanning applies at phase shifter PH2209 place the minimum value of L, thus, does not comprise as the correction for the arbitrary phase angle in the circuit 200 of Fig. 3.This is because for any γ in this example 2, and L=Z sets up, if still have any frequency shift (FS) (error) between the carrier frequency of the received signal X (t) that imports into and local oscillator (LO) frequency, then L=Z is false.If the LO frequency shift (FS) is with 0, f _m/ 2, f _m, 3f _mThe sequence of/2....Hz, X is with non-vanishing so.In other words, if there is problematic frequency shift (FS), then for the calculating of the L of phase place adjustment work improperly (providing error result).This problematic frequency shift (FS) is above-mentioned centrifugal pump.The bandwidth of problematic frequency shift (FS) (width of peak value in this sequence) exists

Near, and depend on the time of integration.For example for the time of integration of 150msec, problematic frequency range (bandwidth) is

For the time of integration of 1000msec, problematic frequency range (bandwidth) is (do not have the best total of points time, the time of integration is long more, and the problem bandwidth is narrow more.) therefore, circuit 300 detects and adjusts any problematic LO frequency shift (FS), and is as described below.

The connection 322 of being led to frequency error detector 320 by the quadrature component Q (t) that connects 208 samplings is further sampled.Similarly, by the connection 327 of leading to detector 320, sampling in-phase component I (t).Detector detects at sequence 0, f _m/ 2, f _m, 3f _mWhether there is frequency error in/2.

Fig. 5 illustrates that detector 320 is output as the function of frequency error or shifted by delta f (Hz).When exporting, detect error greater than threshold value THR.If detect this error, then generate correction signal, and this correction signal is applied to local oscillator 111 via connecting 321, to adjust the reference frequency that generates by local oscillator 111, compensate this error, make that local oscillator frequencies is not problematic frequency, for example, by local oscillator frequencies is moved 20Hz.

Local frequency detector is according to following analysis.

If receive deviation frequency

Wherein k is an integer, and then this algorithm will be forbidden detection.

Correction below detector 320 operations.

$\frac{{|\frac{1}{N}\underset{n=0}{\overset{N-1}{\mathrm{\&Sigma;}}}{I}_r\&CenterDot;I_{\mathrm{Ideal}}|}^2+{|\frac{1}{N}\underset{n=0}{\overset{N-1}{\mathrm{\&Sigma;}}}{Q}_r\&CenterDot;Q_{\mathrm{Ideal}}|}^2}{\frac{1}{N}(\underset{n=0}{\overset{N-1}{\mathrm{\&Sigma;}}}{I}_r^2+\underset{n=0}{\overset{N-1}{\mathrm{\&Sigma;}}}{Q}_r^2)}>\mathrm{th}$

Wherein,

$I_{\mathrm{Ideal}}=\mathrm{Cos}(2\&CenterDot;\mathrm{\&pi;}\&CenterDot;0.5*f_m\&CenterDot;t+2\&CenterDot;\mathrm{\&pi;}\&CenterDot;\mathrm{\&beta;}\&CenterDot;\underset{-\&infin;}{\overset{t}{\&Integral;}}m\left(\mathrm{\&tau;}\right)\&CenterDot;\mathrm{d\&tau;})$

$Q_{\mathrm{Ideal}}=\mathrm{Sin}(2\&CenterDot;\mathrm{\&pi;}\&CenterDot;0.5*f_m\&CenterDot;t+2\&CenterDot;\mathrm{\&pi;}\&CenterDot;\mathrm{\&beta;}\&CenterDot;\underset{-\&infin;}{\overset{t}{\&Integral;}}m\left(\mathrm{\&tau;}\right)\&CenterDot;\mathrm{d\&tau;})$

M (τ) is the PL of expectation or the sampling that finishes tone.I _IdealAnd Q _IdealBe stored in the memory that receives and be associated in.

The process of this example 2 has been described in following mathematical analysis:

We have from above stated specification:

I(t)≈cos(φ(t)-γ)

Q(t)≈sin(φ(t)-α-γ)

Wherein $\mathrm{\&phi;}\left(t\right)=2\mathrm{\&pi;}{f}_d\underset{0}{\overset{t}{\&Integral;}}{A}_m\cos \left({\mathrm{\&omega;}}_m\mathrm{\&tau;}\right)\mathrm{d\&tau;}=\mathrm{\&beta;}\sin \left({\mathrm{\&omega;}}_{m}t\right)$

Wherein

It is modulation index.

Equally, from above stated specification:

For as big modulation index I in this example 2, the path power that the I path power equals Q path power: I is

The path power of Q is

Therefore,

And

But under the situation of the input signal that is received owing to the decline that discrete local oscillator frequency error caused shown in Figure 5, using arcsin function calculation phase imbalance is problematic (providing inaccurate result).If, adjust local oscillator frequencies therefore as above-mentioned needs.

For the frequency Fs signal sampling speed of the tone time of the signal to noise ratio of 15dB, 150msec and 48KHz as the PL composite signal, the accuracy of determining in example 2, to use (algorithm keeps track performance) with reference to the process of Fig. 4 explanation.Fig. 6 and 7 illustrates the result who is obtained.In Fig. 6, curve C 3 has pointed out that simulation receiving phase angle is unbalance, and curve C 4 pointed out, use phase shifter PH2 209 and processor 212 and 214, use that the L minimum value estimates to calculate phase angle unbalance.As shown in Figure 6, unbalance by phase imbalance (degree) the tight tracking actual phase of adjustment algorithm calculating.In Fig. 7, curve C 5 has been pointed out the amplitude imbalance that receives of simulation, and curve C 6 has pointed out to use the amplitude imbalance of calculating of processor 205.As shown in Figure 7, the amplitude imbalance of being calculated by adjustment algorithm (%) tight tracking actual phase is unbalance.

In Fig. 6 and 7, " time-gap number " measured on trunnion axis is each integration period of algorithm.Therefore for example the time slot of 150msec is the integration period of 150msec.Therefore, in Fig. 6 and 7, algorithm operation 150msec calculates the required phase place (Fig. 6) or the amplitude (Fig. 7) of (for time slot 1) and adjusts in each case.Then, it moves another 150msec, and calculates the adjustment of (for time slot 2), and the rest may be inferred.

With reference in Fig. 3 and 4 described circuit 200 and 300, various processors have been described in the above.They can be discrete equipment, although can make up two or more these processors in single treatment facility.Aptly, each treatment facility can comprise digital signal processor, and it is programmed and operates in known in fact mode, to carry out signal processing or (one or more) computing function that needs.

In a word, the invention provides a kind of improved method and apparatus, be used at the adaptive quadrature imbalance compensation of directly changing receiver.

If the present invention is used for radio receiver, can programming wireless after manufacturing the memory of electricity equipment, with the table of storing initial imbalance values and RF frequency.In the operating period of wireless device, imbalance values will change along with the time.Therefore, can be as in the above-mentioned example in use collect the unbalance information of upgrading, and the compensation that above-mentioned unbalance information is used to provide suitable is kept the suitable quality of audio output signal.The unbalance information of this renewal can also be stored in the memory of wireless device, to replace the information of initial storage.

Unlike the prior art, the step of describing in the example 1 and 2 of reference Fig. 3 and 4 allows to operate under the condition that institute's received signal to noise ratio is low and/or generation Rayleigh (multipath) declines.

Receiver circuit 200 or 300 can be used for traditional mobile radio station, and this tradition mobile radio station uses the direct conversion of FM radio communication.

Claims (10)

1. a wireless receiver is used for receiving the warbled RF signal of reconciliation menstruation regulating by direct transfer process, comprising:

The input signal path is used to transmit RF input received signal;

Local oscillator, it is connected to described input signal path;

First frequency mixer is used to make output reference signal and the mixing of input received signal from described local oscillator, to produce the in-phase component of described input received signal;

Quadrature phase shifter is used for orthogonal phase shift is applied to output reference signal from described local oscillator, and described local oscillator is connected to described input signal path;

Second frequency mixer is used for output signal and described input from described phase shifter are received

Signal mixing is to produce the quadrature component of described input received signal;

Be used for by producing the device that described homophase and quadrature component composite function produce output demodulating information signal; And

Be used for detecting periodically the error of the relative phase difference between described homophase and the quadrature component

And the relative adjustment that is used to apply phase difference to be compensating the device of described detection error,

The wherein said device that is used for detecting periodically error is operating as, and determines the inner product L of described homophase and quadrature component, and wherein said inner product L is by expression formula

Definition, wherein I (t) is described in-phase component, and Q (t) is described quadrature component, and t is the time, and T is an integration period.

2. receiver as claimed in claim 1, the wherein said device that is used for detecting periodically error is operating as, and determines the approximation of L, when input signal be represented as x (t)=cos (wt+ φ (t)+γ),

And X＜＜during Z, L has Form, wherein A is an amplitude measurement, φ is pitch frequency modulation, a is the error of the phase angle between described input RF received signal in-phase component and the quadrature component, γ is the arbitrary phase angle, and t is the time.

3. receiver as claimed in claim 2, the wherein said device that is used for detecting periodically error is operating as, when following at least a situation, determine described inner product L, promptly working as a), arbitrary phase angle γ is zero, and b) described input RF received signal is to use the tone of big modulation index modulation, and c) when unbalance phase angle α is minimum value.

4. receiver as claimed in claim 3, the wherein said device that is used for detecting periodically error is operating as, sample described homophase and quadrature component, and apply the relative phase shift of the change between described sampling homophase and the quadrature component, and determine when that the phase shift of described relative variability provides the minimum value of described inner product L.

5. receiver as claimed in claim 4, the wherein said device that is used for detecting periodically error is operating as, and applies the relative phase shift between described sampling homophase and the quadrature component, and it changes with step-length.

6. receiver as claimed in claim 1, wherein said being used for is operating as by producing the device that composite function produces output demodulating information signal, calculates difference function

$\frac{d}{\mathrm{dt}}\left[\arctan \right\{\frac{Q\left(t\right)}{I\left(t\right)}\left\}\right].$

7. receiver as claimed in claim 1 comprises: the adjustment that is used for detecting the amplitude imbalance between described homophase and the quadrature component periodically and is used to apply relative amplitude is to compensate the unbalance device of described detection.

8. receiver as claimed in claim 1 comprises: it is unbalance and be used to apply the frequency adjustment of described output reference signal to compensate the unbalance device of described detection to be used for detecting periodically frequency between described local oscillations output reference signal and the described input received signal.

9. one kind receives the method for conciliating the warbled RF signal of menstruation regulating by direct transfer process, comprising: detect the homophase of input RF received signal and the error in the relative phase difference between the quadrature component periodically; And, apply the relative adjustment of phase difference, compensating the error of described detection,

The step that wherein detects error periodically comprises: determine the inner product L of described homophase and quadrature component, wherein said inner product L is by expression formula

Definition, wherein I (t) is described in-phase component, and Q (t) is described quadrature component, and t is the time, and T is an integration period.

10. method as claimed in claim 9 further comprises: by calculating difference function

Produce output demodulating information signal.

Images