CA1116241A - Radio broadcasting system with code signaling - Google Patents
Radio broadcasting system with code signalingInfo
- Publication number
- CA1116241A CA1116241A CA297,759A CA297759A CA1116241A CA 1116241 A CA1116241 A CA 1116241A CA 297759 A CA297759 A CA 297759A CA 1116241 A CA1116241 A CA 1116241A
- Authority
- CA
- Canada
- Prior art keywords
- signal
- frequency
- carrier
- sub
- phase
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired
Links
Classifications
-
- G—PHYSICS
- G08—SIGNALLING
- G08G—TRAFFIC CONTROL SYSTEMS
- G08G1/00—Traffic control systems for road vehicles
- G08G1/09—Arrangements for giving variable traffic instructions
- G08G1/091—Traffic information broadcasting
- G08G1/092—Coding or decoding of the information
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04H—BROADCAST COMMUNICATION
- H04H20/00—Arrangements for broadcast or for distribution combined with broadcast
- H04H20/28—Arrangements for simultaneous broadcast of plural pieces of information
- H04H20/33—Arrangements for simultaneous broadcast of plural pieces of information by plural channels
- H04H20/34—Arrangements for simultaneous broadcast of plural pieces of information by plural channels using an out-of-band subcarrier signal
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- Engineering & Computer Science (AREA)
- Multimedia (AREA)
- Physics & Mathematics (AREA)
- General Physics & Mathematics (AREA)
- Signal Processing (AREA)
- Stereo-Broadcasting Methods (AREA)
- Circuits Of Receivers In General (AREA)
Abstract
ABSTRACT
A radio broadcasting system with code signaling where-in a multiplex signal frequency-modulated on a main carrier is transmitted at the transmitter side. The multiplex signal con-tains an audio frequency information and, in the case of stereo transmission, a stereo information signal modulated on a sup-pressed sub-carrier and a stereo pilot, which serves for de-modulating the stereo information signal, whose frequency is loc-ated between the frequency spectra of the audio frequency in-formation signal and of the modulated stereo information signal.
The multiplex signal also contains a binary code signal which is modulated on a further sub-carrier located outside the frequency spectra of the audio frequency information signal and having an amplitude which causes the main carrier to deviate for not more than 1 KHz. The further sub-carrier is a harmonic of a sub-harmonic of the stereo pilot not coinciding with a harmonic of the stereo pilot and is derived at the transmitter side form the same frequency source as the stereo pilot. The code signal is binary phase modulated on this sub-carrier.
A radio broadcasting system with code signaling where-in a multiplex signal frequency-modulated on a main carrier is transmitted at the transmitter side. The multiplex signal con-tains an audio frequency information and, in the case of stereo transmission, a stereo information signal modulated on a sup-pressed sub-carrier and a stereo pilot, which serves for de-modulating the stereo information signal, whose frequency is loc-ated between the frequency spectra of the audio frequency in-formation signal and of the modulated stereo information signal.
The multiplex signal also contains a binary code signal which is modulated on a further sub-carrier located outside the frequency spectra of the audio frequency information signal and having an amplitude which causes the main carrier to deviate for not more than 1 KHz. The further sub-carrier is a harmonic of a sub-harmonic of the stereo pilot not coinciding with a harmonic of the stereo pilot and is derived at the transmitter side form the same frequency source as the stereo pilot. The code signal is binary phase modulated on this sub-carrier.
Description
. . . - PilN.8691C
`- . . LOOP/GILS/GK
10.2.1978 ` . .
- : ~
"Radio broadcasting system with code signaling"
The invention relates to a radio broadcasting system with code signaling wherein at the transmitter side a multiplex signal which is frequency modulated on ` a main carrier is transmitted, said multiplex signal com-5 prising: an audio frequency information signal, in the case of stereo transmission a stereo information signal modulated on a suppressed sub-carrier, a stereo pilot whose frequency is located between the frequency spectra of the audio frequency information signal and of the modulated stereo information signal and which serves for ~ . demodulating the stereo information signal, as well as .~ a binary code signal modulated on a further sub-carrier located outside said frequency spectra and having an am-c ~.~ ~<~ ~
plitude which causes the ub`carricr to deviate for not more than 1 KHz. In addition the invention relates to a ~ transmitter for transmitting signals in accordance with ;~' .
such a system as well as to a receiver for receiving such signals.
` When tuning the present FM-radio receivers the . :
~:. 20 user often experiences great difficulties because the ~,~
`~` tuning scale only mentions frequencies and/or channel ` numbers and not the names of the stati.ons. In addition, one given program is often transmi-tted by several trans-~' mitters so that the user does not well know if he has tuned to the strongest transmitter.
In order to provide the user with an easily - recognizable identlfication of the FM transmitters and/or . .
of the nature of the program transmitted by the trans- ;
.. .
.. . . .
:
~ - P~IN.8691C
10.2.1978
`- . . LOOP/GILS/GK
10.2.1978 ` . .
- : ~
"Radio broadcasting system with code signaling"
The invention relates to a radio broadcasting system with code signaling wherein at the transmitter side a multiplex signal which is frequency modulated on ` a main carrier is transmitted, said multiplex signal com-5 prising: an audio frequency information signal, in the case of stereo transmission a stereo information signal modulated on a suppressed sub-carrier, a stereo pilot whose frequency is located between the frequency spectra of the audio frequency information signal and of the modulated stereo information signal and which serves for ~ . demodulating the stereo information signal, as well as .~ a binary code signal modulated on a further sub-carrier located outside said frequency spectra and having an am-c ~.~ ~<~ ~
plitude which causes the ub`carricr to deviate for not more than 1 KHz. In addition the invention relates to a ~ transmitter for transmitting signals in accordance with ;~' .
such a system as well as to a receiver for receiving such signals.
` When tuning the present FM-radio receivers the . :
~:. 20 user often experiences great difficulties because the ~,~
`~` tuning scale only mentions frequencies and/or channel ` numbers and not the names of the stati.ons. In addition, one given program is often transmi-tted by several trans-~' mitters so that the user does not well know if he has tuned to the strongest transmitter.
In order to provide the user with an easily - recognizable identlfication of the FM transmitters and/or . .
of the nature of the program transmitted by the trans- ;
.. .
.. . . .
:
~ - P~IN.8691C
10.2.1978
2~
.
mitter, a radio broadcasting system with code signaling as defined in the preamble has already been suggested at the CCIR (Comité Consultatif International des Radio-- communications). In this system the code signal is trans-mitted by means of a suitable sub-carrier above the fre-quency spectrum of the stereo information signal. This sub-carrier is frequency-modulated with the binary code signal which, by means of a digital code, contains in-formation on, for example, the name of the program, the location of the transmitter, the nature of the program and the channel number, so that, for example, the fol-lowing lnformat~n, consisting of 16 characters, is re-ceived:
¦ Nedl Roerm KL 25 ¦
1S The receivers for such a system are provided ~ with a decoder which decodes the binary code signal from `~ the signal received and uses it, for example, for wholly ` or partly optically displaying the information thus ~`i`
transmitted, so that the user can immediately see to which transmitter his receiver is tuned. Alternatively, it is possible to arrange the receiver in such a way ' that at a preset code a portion of the receiver or of a tape recorder or reproducing apparatus is switched on or "~ off. In particular, if the code contains a special code ! 25 which is transmitted for traffic reports~ the code can be used to switch on the reproducing section of a car ~'``` :
~`. radio receiver or to stop a tape reproducing device which ~' is ln operation.
The above-mentioned prior art broadcasting `, .
system with code s:ignaling has been tested in practice
.
mitter, a radio broadcasting system with code signaling as defined in the preamble has already been suggested at the CCIR (Comité Consultatif International des Radio-- communications). In this system the code signal is trans-mitted by means of a suitable sub-carrier above the fre-quency spectrum of the stereo information signal. This sub-carrier is frequency-modulated with the binary code signal which, by means of a digital code, contains in-formation on, for example, the name of the program, the location of the transmitter, the nature of the program and the channel number, so that, for example, the fol-lowing lnformat~n, consisting of 16 characters, is re-ceived:
¦ Nedl Roerm KL 25 ¦
1S The receivers for such a system are provided ~ with a decoder which decodes the binary code signal from `~ the signal received and uses it, for example, for wholly ` or partly optically displaying the information thus ~`i`
transmitted, so that the user can immediately see to which transmitter his receiver is tuned. Alternatively, it is possible to arrange the receiver in such a way ' that at a preset code a portion of the receiver or of a tape recorder or reproducing apparatus is switched on or "~ off. In particular, if the code contains a special code ! 25 which is transmitted for traffic reports~ the code can be used to switch on the reproducing section of a car ~'``` :
~`. radio receiver or to stop a tape reproducing device which ~' is ln operation.
The above-mentioned prior art broadcasting `, .
system with code s:ignaling has been tested in practice
- 3 ~
PHN.8691~
` ~ 10.2.1978 .
:
with the following values: -` - The sub-carrier frequency was 66 KHz and the -- frequency sweep 1 KHz so that owing to the binary in-formation the frequency was switched between 65 KHz and - 5 67 KHz.
The code used was the 6-bit ASCII-code having 16 characters per information.
- The amplitude of the modulated code signal was chosen such that 1 KHz, i.e. 1,33 % of the total fre-quency sweep of 75 KHz available for the FM modulation of the main carrier, was occupied by the code signal.
The comparatively small amplitude (1 KHz) of this signal is opted for because experiments proved that a greater amplitude may cause interference noise in some FM re-~ 15 ceivers.
;- It appeared, however, that the necessarily ~ small amplitude of the modulated code signal and the .
~` ` comparatively high frequency thereof (66 KHz) result in a poo~ signal-to-noise ratio. In order to recover the ~` 20 code signal flawlessly, the receiver requires a high-. ~ .
grade filter having a good quality factor and a good "; temperature stability. In addition, it appeared that in ~", spite of the use of such a high-grade and expensive fil-~ - .
.; ter, decoding of the code signal no longer occurs flaw-lessly at aerial voltages below 10 /uV (at 60 Ohm) where-as the average FM receivers still furnish an acceptable . mono-reception at such aerial voltages.
It is an object of the invention to provide a : radio broadcasting system with code signaling which en-ables a substantially flawless decoding of the code signal at received aerial voltages where an acceptable .--, .
- .
, P~IN.8691C
10.~.19~
`
..
' mono-reception is not or hardly possible, whereas the signal reception in existing receivers is not or hardly disturbed and whereas, furthermore, high-grade and, con-sequently, expensive'filtering means in the receiver for the system according to the invention can be dispensed ; with and the radio broadcasting system according to the invention is therefore characterized in that said further - sub-carrier is a harmonic of a sub-harmonic of the stereo ~'` pilot not coinciding with a harmonic of this pilot which, ' 10 at the transmitter side, is derived from the same fre-: quency source as the stereo pilot and in that the code ~ signal is binary phase modulated on this su'b-carrier.
." The expression binary phase modulation must here, as customary, be understood to mean a phase mo-~ 15 dulation in which the phase of the sub-carrier is shifted "` 1800 by the binary code signal. This furnishes a modulated ` signal with a fully suppressed carrier.
Using phase modulation of the sub~carrier with the binary code signal (phase shift keying~ instead o~
- 20 frequency modulation (frequency shift keying~ results in an improvement of -the signal-to-noise ratio. In contrast ' to the demo~dulation of the frequency-modulated 'sub-carrler the demodulation of the phase-modulated sub-carrier requiresJ however, an unmodulatcd ("clean") sub-carrier. This sub-carrier is not present in the binary phase-modulated code signal because the sub-carrier it-` self is suppressed and sidebands only are transmitted.
Generating this sub-carrier at the receiving side can, however, be effected by squaring the incoming binary . 30 phase-modulated signal which results in a carrier having - double the frequency, by thereafter filtering this , i' I , ~ ' ' ' J ' ' . X (J '~ I C
~ ~r ~ ~ 10.~ 7~
.
carrier with double frequency, and subsequently having the carrier of the original frequency recovered from the carrier of double this frequency by means of a fre-quency divide-by-two divider.
With this method, in the case of poor signal-` to-noise ratios, the carrier of double the frequency must be obtained from a signal having a high degree of noise. If, for example, a so-called phase locked loop ` is used for this purpose, then this can indeed be ef-fected by using a low-pass filter having a low cut-off frequency in this loop in such a way that the phase of _ the voltage-controlled oscillator of the phase loop is ~ not modulated too much by noise, on the other hand such '~ ~ a low-pass filter having a low cut-off frequency reduces ~ 15 the pulling-in range of the phase locked loop to such t~; an extent that a voltage-controlled oscillator having a ~;~ very stable free-running frequency is now required. In ~m~ practice this can only be obtained with a crystal-con-trolled oscillator.
With the present invention wherein binary phase . modulation of the sub-carrier for the~code signal is `" used in combination with a ~requency relationship be-tween the stereo pilot and said sub-carrier, which ~-- relationship is fixed at the transmitter side, a system -~ 25 is obtained which can be decoded without high-grade means and which is comparatively insensitive to poor signal-to-noise conditions.
The sub-carrier for the transmitter ~dentifi-~; cation signal can now be recovered with much simpler means because the stereo pilot is modulated with a much ~ greater frequency sweep (10% of the total 75 KHz fre-! - 6 ~
` ~ l PHM.~691C
10.2.1978 2~.~
-quency sweep) on the main carrier than the transmitter identification signal itself ( 1 9 33~ of the total fre-quency sweep of 75 KHz). In a receiver for a system according to the invention phase errors may be produced owing to different delay times for the modulated code signal and for the stereo pilot in the tuner and in the intermediate frequency section of the receiver. Also phase multiplicities are produced because sub-carrier frequency (~k) of the modulated code signal is chosen , ........................................... .
; 10 equal to a "fractional" harmonic of the stereo pilot (c~);
this expression means that ~k = m W, where m and n are integers but n is not divisable by m. The frequency division required thereby in the transmitter ana in the receiver may produce these phase multiplicities.
In accordance with a further aspect of the invention an automatic phase corrector is used in a ~` receiver according to the invention which can adjust the phase of the unmodulated wave required for detecting the ,.~ .
modulated code signal relative to the modulated code ~; 20 signal i-tself. This phase corrector is controlled from a phase detector which compares the phase of the modulated code signal to the phase of the unmodulated wave obtained from the stereo pilot and depending on the result of this . :
comparison, the phase corrector corrects any phase errors.
As the carrler itself lS missing in the binary modulated code signal this, however, cannot be done without fur-ther measures.
A first method to solve this diffioulty is the use of frequency doubling of the modulated code signal which provides an unmodulated carrier of twice the sub-carrier frequency. This carrier of twice the sub-carrier . ~
- 7 - ~
~ ~ ' P~IN.8691C
: ~ ' ' ' 10.2.'1978 .
frequency is applied to one input of the phase detector, a wave of likewise twice the subcarrier frequency, which is cbtained by means of frequency multlplication and/or division of` the stereo pilot being applied to the other 5 input.
A second method consists in the use of a phase inverter in one of the input leads or in the output of the phase detector, which phase inverter is controlled ~r by the demodulated output signal of the synchronous detector. It appears that this results in both cases in à phase duplicity at the detection of-the code signal.
Thls phase duplicity is not disturbing if a code is used which is insensitive to such a duplicity, for example `. a so-called differential code; this is a code with which the two binary states are not transmitted by two phase conditions of the sub-carrier but by the occurrence or `
- non-occurrence, respectively, of a phase transition from ~` the one phase to the other or vice versa.
The frequency of the sub-carrier can, for example, ` 20 be chosen between the third and the fifth harmonic of the stereo pilot. Choosing lt below the third harmonic brings the sub-carrier too closely to the spectrum of the stereo information signal and choosing it above the fifth harmonic increases the chance for disturbances owing to adjacent transmitters.
Furthermore~ interference may occur in a number of stereo receivers between the sub-carrier for the transmitter identiflcation and the seoond harmonic of the 38 ~lz signal required for stereo de-teotion, which cor-responds to the fourth harmonic of the pilot. For this reason the sub carrier for the code signaling should not PHN.8691C
10.2.1978 ~,~
- be located too near this fourth harmonic.
Owing to the non-linear phase characteristic - of the intermedia-te freqllency section of the receiver an interference product having a frequency equal to the difference frequency between the sub-carrier and the stereo pilot occurs in the multiplex signal. This inter-ference product may, after detection with the 38 KHz . .
wave, cause audible noise if the sub-carrier is located . .~
too near the third harmonic of the stereo pilot.
The above-mentioned non-linear phase character-` istic furthermore causes noise in the region of the whole - harmonics of the stereo pilot. All these considerations lead to the choice that the sub-carrier for the code signal should notcoincide with a full harmonic of the stereo pilot. Consequently, it should be preferred to -' choose a "fractional" harmonic of the stereo pilot f`or the sub-carrier of the code signal and to remove the .` phase multiplicity then occurring in the receiver in the ~. .
; manner described above.
On the bases of the above consideration pre-ference should be given to a position of the sub-carrier half-way between two harmonics of the stereo pilot~ for example at 7/2 or 9/2 times the stereo pilot. The in-vention has been tested with a sub-carrier frequency of t 25 7/2 times the pilot frequency; for clearness' sake an ' ` embodiment for a sub-carrier frequency of 16~5 tirnes a pilot frequency is given.
A further improvement of a radio broadcastin~
; system with code signaling ln which, while maintaining a re]iable transmission of the code in~ormation, a reduced chance of disturbing existing receivers is possible, is ~- , `~ ~ 9 _ - ~ P~IN.8691C
_~ ' 10.~.1978 ;`' -: - characterized in that the additional sub-carrier with the ' modulated code signal is located in at least one of the . two halves of the frequency range, divided into two parts by the stereo pilot, between the upper limit of : -, 5 the frequency spectrum of the audio frequency information signal and the lower limit of the frequency spectrum of the modulated stereo information signal and in that the modulated code signal has an amplitude which causes the main carr.~er to deviate less than 1 KHz, preferably `, 10 0.25 KHz.
~ This measure has the following effects 1. Because the sub-.carrier for the code signaling ., .
is now remote from the higher harmonics of the 38 KHz stereo detection signal, these higher harmonics cannot : 1$ ~ produce audible noise in existing receivers.
2. Because the sub-carrier for the code signaling is now located much lower in the frequency spectrum of the multiplex signal, the signal-to-noise ratio is con-siderably more favourable. Consequently, the modulated code signaI can have a still smaller amplitude than was the case for a sub-carrier of, for example, 66.5 KHz.
. For comparison it should be noted that with the present - preferred embodlment, for obtaining a reliable code Slg-naling, the modulated code signal need only occupy ap-- 25 proximately 0.25 KHz of the maximum frequency sweep of 75 KHz. This requires approximately 1 KHz in case of a .66.5 KHz sub-carrier. Of course the much smaller sub-carrier amplitude considerably reduces the chance for interference noise.caused by other components of the multiplex signal.
A still further reduction of the chance for.
:,~ , . .
, .
:; . : . .
- PI~N.8691C
--- - 10.2.1978 ~; ~
. . - . .
- noise in existing receivers, particular in the case of monoreception, can be achieved in accordance with a fur-- ther characteristic of the invention when a sub-carrier whose phase is binary modulated by the code signal is located in each of the two halves of said frequency range, divided into two parts by the stereo pilot, and the two sub-carriers modulated by the code signal have equal amplitudes and such a phase relative to the stereo pilot that together with the stereo pilot they form a - 10 signal obtained by quadrature modulation of the stereo ;~ - pilot by a sub-carrier derived from the stereo pilot, this sub-carrier itself being binary phase modulated by the code signal.
.`. With such a signal each of the binary phase 1$ modulated sub-carrier signals may be considered to be a sideband of a double-sideband signal having the stereo pilot as the carrier. The stereo pilot is quadrature mo-dulated by a modulation signal which itself is binary --` phase modulated by the code signal. The modulation sig-nal has a frequency equal to the difference between the " frequency of the stereo pilot and of a sub-carrier. A
system according to this further feature of the invention, and tested in practice has, next to the stereo pilot with i ` a frequency fp of 19 KHz9 a first sub-carrier of 16.625 KHz (7/8 fp), whose phase is binary modulated by the code signal and a second sub~carri~r of 21.375 KHz (9/8 fp) whose phase is binary modula~ted by the code signal. In the case of equal amplitudes of the two sub~
carriers and a relative proper phase relation between the sub-carriers and the stereo pilot, the three signals together constitute a stereo pilot which is quadrature ~ ~ .
; P~N.8691C
, 10.2.1978 '' modulated by a sub-carrier signal of 1/8 f which itself ~, is binary phase modulated by the code signal. To this ~- end *he phase of one swb-carrier must lead the stereo ' pilot, shifted over 90, for the same amount as the other sub-carrier lags this 90-shifted stereo pilot, in other ' words the resultant of the two modulated sub-carriers has -, a 90 phase shift relative to the stereo pilot.
The sum of the stereo pilot and the two sub-carriers forms a pilot signal the amplitude of which is `- 10 substantially constant. As especially the amplitude va-" riations of the pilot give rise to distortion products owing to~the non-linear phase characteristic of the inter-~ mediate frequency section of the receivers the above-described measure offers an additional distortion reduction.
Within the framework of the invention it is ` alternatively possible to give the sub~carrier modulated by the code signals such a phase that the resultants al-ways coincide (0 or 1800) with the stereo pilot. The two sub-carriers functioning as sideband for the stereo pilot then cause an amplitude modulation of the stereo pllot by a carrier signal which itself is binary phase-modulated by the code signal.
When applying the stereo pilot, double~sideband quadrature or amplitude-modulated by the two sub-carriers or the stereo pllot single-sideband phase and amplitude-modulated by one sub-carrier, to the stereo decoder of radio receivers, the sub-carrier amplitudes which are al-ready small, are suppressed so much by the stereo pilo-t filter provided in such receivers, relative to the stereo pilot itself that disturbing the stereo detector does sub-- stantially not occur. Such a dis-turbance would be much : P~IN.c',~)~31C
~ 10.2.197~
. ~ ' ' . .
. . .
greater in the case of direct phase or amplitude modulation of the stereo pilot by the code signal.
This disturbance is, of course, also greater according as the sub-carriers are located nearer the ~tereo pilot (for example at 11/12 fp and/or 13~12 ~ ).
~" .
On the other hand~ in case of an excessive sub-carrier ~; stereo pilot distance the sub-carrier becomes located too near the frequency spectrum of the audio information signal or of the modulated stereo information signal. On the basis of these considerations a spacing of 1/8 fp between sub-carrier (sub-carriers respectively) and the stereo pilot should be preferred.
" As with the system according to the present ` embodiment the sub-carrier frequency is relatively near that of the stereo pilot it is preferred in the receiver arranged for receiving such signals to convert the mo-dulated sub-carrier (7/8 fp and/or 9/8 f ) first with the stereo pilot to an intermediate frequency (1/8 fp) which is harmonically related to the stereo pilot and which is equal to the difference between the sub-carrier frequency and stereo pilot rrequency. The synchronous detection of the code signal can then be effected at .~ this lower frequency in a corresponding manner as de-scribed above.
The invention will be further~exp]alned with ~ reference to the Figures in the accompanying drawings.
`` Herein: ~ ~
Figure 1 is a block diagram of an embodiment ` of a transmltter for a first implementation cf the system ;~ 30 according to the invention, Figure 2 shows the frequency spec-trum of the .
, .
P~IN.8691C
10.2.1978 ;~ multiplex signal generated in the first ernbodiment of the system according to the invention at the transmitter side and obtained at the receiver side after FM demodu-- lation, Figure 3 is a block diagram of a first embodi~
` ment of a receiver according to the invention, Figure 4 is a block diagram of a second embodi-ment of a receiver according to the invention, Figùre 5 and Figure 5a, respectively, are block diagrams of a transmitter for a second embodiment of the system according to the invention, Eigure 6 shows the frequency spectrum of the multiplex signal generated at the transmitter side and obtained at the receiver side after FM-demodulation according to the second embodiment of the system, Figure 7 is a block diagram of an embodiment .
of a receiver for receiving a signal as shown in Figure 6.
Figure 8 is a block diagram of a second embodi-ment of a receiver for receiving a signal as shown in Figure 6.
- The transmitter of Fig. 1 comprises a source . . .
` of left-hand audio signals 1 and a souroe of right-hand audio signals 2. Each of the left-hand and right-hand audio signals is applied via a pre-emphasis network 3 and -25 4, respectively, and via a low-pass filter 5 nnd 6, . respectively, having a cut-off frequency of 15 KHz,to an adder circuit 7 and to a subtractor circuit 8. There-. after the sum signal L-~R derived from the adder circuit is applied to an input 9 of a multiplexer 10. The dif-ference signal L-R of the subtrac-tor circuit 8 is modu-. latcd in a balance modulator 11 on a stereo sub-carrier ~ 1 1 1 ,s ,~
: . . . .
: - , . . .. , .. , . . ,. , ~ .. , P~IN.869lC
10.2.1978 ! ' .
of, for example, 38 KHz and the modulated stereo inform-ation signal thus obta:ined, which consists of two side-bands with suppressed stereo sub-carrier, is applied via a bandpass filter 12 to a second input 13 of the multiplexer 10.
In addition, the transmitter of Fig. 1 com-prises a stable oscillator 14, for example a crystal oscillator supplying a wave of, in general, 19 KHz which is used as stereo pilot. This stereo pilo-t is applied to .~ 10 a first input 15 of the multiplexer 10.
The stereo pilot of the oscillator 14 is also `~ applied to a so-called phase locked loop 16 which in-cludes a phase detector 16a, a low-pass filter 16b, a voltage-controlled oscillator 16c and a frequency-divide-1$ by-two divider 16d. The phase locked loop 16 is used for p.oducing a sub-carrier whose frequency (38 KHz) is equal to twice the frequency of the stereo pllot and which is locked to the stereo pilot. The operation of such a phase locked loop is known; the j8 KHz output signal of the oscillator 16c is converted in the divide-by-two divider ` 16d into a 19 KHz signal which is compared in the phase ~i~ detector 16a with the 19 KHz pilot of the oscillator 11~.
~ .
The output voltage of the phase detector 16a is filtered in low-pass filter 16b and applied as a control voltage : 25 to the oscillator 16c.
Via a phase shifter 17 the 38 I~Hz output signal of the phase locked loop 16 is applied as stereo sub-carrier to the modulator 11 for modu~ating the L-R signal.
The phase shifter 17 is used to give the sub-carrier the internationally prescribed phase relative to -the 19 KHz .
s-tereo pilot.
`; .
~, .
~ P~N.8691~
~ 2~ 10.2.l978 '' A second phase locked loop 18, connected to - the 19 KHz oscillator, comprises a phase de-tector 18a, a lowpass filter 18b, a voltage-controlled oscillator 18c and a 16-scaler 18d. The phase locked loop 18 operates in a similar manner as the phase locked loop 16 and supplies an output signal which is locked to the stereo pilot, of 304 KHz, i.e. 16 times the pilot frequency.
Thereafter the 304 KHz signal of the phase locked loop 18 is reduced in a divide-by-5 divider 19 to 60.8 KHz and the latter signal is applied as sub~carrier of the trans-mltter identification signal to the carrier input of a balanced modulator 20. The modulatlon input of this mo-dulator~is connected to a schematically shown arrangement 21 for generating a suitable binary code which includes the transmltter identification information, for example a code as defined in the preamble.
The modulator 20 may~ for example, be a ring ~` modulator or a dual long-tail circuit or any other known modulator which, under the influence of the bits derived `. 20 from the arrangement 21, shifts the phase of the 60.8 KHz signal from the divlde-by-5 divider 19 over 1800. The i 60.8 KHz signal, phase-modulated in this way, is applied via a bandpass filter 22 having a bandwidth of approxi-mately 4 KHz to a fourth input 23 of the multiplexer 10.
The multiplexer combines the signals at the inputs 9, 13, 15 and 23 and supplies them combined to a FM-trans-- mitt.er, not shown in the drawing.
- For further explanatlon, Figure 2 shows the frequency spectrum of the signal obtained at the output ~` 30 of the multiplexer. Between 0 and 15 KHz there is the sum signal I~R supplied v:La the input 9, at 19 K~Iz there I G
~ .
~ PflN.oG91C
~ r~ Z~ .~ . 0 . 2.1978 is the stereo pilot supplied via the ~nput 15~ between 23 and 53 KHz there lS the L-R signal which is modulated at 38 KHz and which is supplied via the lnput 13 and at 60.8 KHz there is the approximately 4 KHz wide trans-mitter identification signal which is supplied via the input 23. It should be noted that the relative amplitude ratios generally deviate more from one another than is indicated for clearness~ sake. In general the stereo pilot is approximately 9x smaller than the L~R and L-R
components and the amplitude of the transmi-tter identi-fication signal is preferably chosen to be approximately 10x smaller than that of the stereo pilot.
The receiver of Fig. 3 comprises a tuner 24, an lntermediate frequency amplifier 25 and an FM-detector 26. The multiplex signal which is composed of the com-' ponents shown in Fig. 2 is available at the output of ~` this FM detector. For a stereo receiver tbls multiplex -~ `signal is applied to a stereo decoder 27 which supplies the left-hand and right-hand audio signals which are ` ~ 20 supplied vla audio amplifiers 28 and 29 to a left-hand and a right-hand loudspeaker 30 and 31.
For demodulating the transmitter identification signal the multiplex signal is applied to a 19 KHz band-pass filter 32 for the stereo pilot and a 60.8 KHz band-pass filter 33 for the transmi-tter identifica-tion slgnal.
The stereo pilot ~iltered out by means of the filter 32 is additionally filtered and its frequency multiplied by a phase locked loop 34 which includes a phase detector ~- 34a, a low-pass filter 34b, a voltage-controlled oscil-lator 34c and a 1:32 frequency divider 34d. I-ts operation `~` is similar to that of the phase loclced loops 16 and 18 ~. , .
.
E)HN.8G9 1 c 10.2.1978 ,. ' -of Fig. 1.
The output wave of the phase locked loop 34, which has a frequency of 32x19=608 KHz, is thereafter reduced to 121.6 KHz in a divide-by-five divider 35, thereafter passed through a controllable phase shifter 36, ~hose function will be explained in the course of ` this description, thereafter divided in a divide-by-two divider 37 to 60.8 KHz and, finally, applied to a first input 38 of a synchronous demodula-tor 39.
The 60.8 KHz phase modulated transmitter iden-tification signal originating from the bandpass filter 33 is applied via a 45 phase shifter 40 to a second inpu-t 41 of the synchronous demodulator 39. The syn-chronous detection of the 60.8 KHz phase modulated trans-mitter identification signal at the input 41 by means of the unmodulated 60.8 KHz wave at the input 38 furnishes, at the output of the synchronous demodulator 39, the demodulated binarv transmitter identification signals.
This binary code signal is passed through a low-pass filter 42, thereafter it is formed into square pulses in a pulse shaper 43 and applied to a decoder 44. This decoder converts the binary transmitter identifica-tion ; signal iDto signals suitable for driving a "display" 45 For a proper synchronous detection in the - 25 demodulator 39, the unmodulated wave at the input 38 ` must have the proper phase relation relative to the . ~
modulated signal applied to the input 41. ID general this proper phase relation is not guaranteed owing to ~, . . .
the following causes: ~
~` 30 1. Owing to the insufficient linear phase charact-,~ `eristic of the intermediate frequency amplifier 25 -the ~, .
PIIN.8691C
- 10.2.1978 _ , . , 19 ~Hz stereo pilot and the 60.8 KHY. transrnitter iden-tification signal may be subjected to mutually different delay times.
2. The input filters 32 and 33 may effect unwanted phase shifts.
3. Owing to the frequency division by the divider 19 in the transmitter the phase of the transmitted 60.8 KHz transmitter identification signal is no longer unambiguously determined relative to the transmitted - 10 stereo pilot. A similar phase multiplicity is caused by ~` the frequency divider 35 in the receiver.
'~ In order to obvIate all these phase problems the circuit of Flg. 3 comprises the adjustable phase shifter 36 mentioned above. This shifter is controlled via ~ low-pass filter 46 from a phase detector 47. The phase detector 47 has two inputs 48 and 49, the input 48 being connected to the output wave o~ the phase shifter 36, the Input 49 being connected to the output of a device 50 which produces a modulatéd wave of double the frequency (namely 121.6 KH~) from the phase-modulated signal of the filter 33. To this end the device 50 has a non-linear characteristlc with even~power term, for example a squaring circuit or a full-wave rectifier.
Becauss the phase locked loop 34 multiplies the stereo pllot by a factor of 2 more than necessary for the synchronous detection, the frequency of the wave which IS applied to the input 48 of the phase detector is equal to double the carrier frequency. So measuring the phase by means of the phase detector 47 is ef~ected at double ths carrier frequency and the result of the measuremeIl1, is used to compensate ~or the above-mentioned ~, .
. :
. ~
- PHN.86~1C
10.2. 1978 .
unwanted phase shifts in the controllable phase shifter 36. It should be noted that the frequencies of the two signals which are connected to the phase detector 47 are always equal to one another so that no pulling-in pro-`blems can arise. The phase errors which are correctedtherewith vary only slowly and the low-pass filter 46 can, consequently, have a very low cut-off frequency (for example 10 Hz). Owing to this low cut-off frequency, rapid phase variations which might be produced owing to the noise in the transmission path 33-40-41 of the trans-mitter identification signal can be effectively suppressed.
By means of the specified measures it is possible to `~ obtain for the synchronous detector 39 an unmodulated wave of the proper frequency and the proper phase and - 15 whicll is noise-free to a sufficient degree. As, owing to the phase control by means of 36, 46, L~7 the phase is compensated at double the carrier frequency, the phase relation at the inputs of the synchronous detec-tor 39 is still not unambiguous (180 phase duplicity). When using a (differential) code which is insensitive thereto a proper transmission of the code signal can, however, ` yet be guaranteed.
.; In practice, the automatic phase control by means of the phase controller 36 always operates so . 25 that the two lnput signals of the phase detector L~7 are shifted 90 in phase relative to one another. Moreover it is desirable that the mutual phase relation between the input slgnals of the synchronous detector is 0 or 180. If the divide-by-two divider 37 is so constructed .
~` 30 that the zero-crossings of its output wave coincide with . ~ ~
`~ the zero-crossings of its input wave and :if the tre-., ~ .
. - .
, . PIIN.~91C
- - lO~J~ 1978 ` quency doubler 50 is implemented as a squar:ing clrcult ; ln which the tops of the input wave co:inc:ide with tops of the output wave, then this preferred phase relation is achieved automatlcally. In other cases an addltional phase correctlon may be necessary in one of the input leads of the detectors 47 and 39, for example a 90 correction for the double carrler frequency or a 45 correction for the carrier frequency itself. The 45 phase shifter 40 ls used for this purpose. It should be noted that several variants of the clrcult of Fig. 3 are possible. It is, for example, possible to replace the divide-by-two divider 37 by a frequency doubler in the input lead 48 of phase detector 47. The frequency mul-tiplication fac-tor of the phase locked loop 34 should then be a factor of 2 smaller. Alternatively it is, for example, possible to lnclude the phase controller 36 in the output lead of the filter 33.
If, instead of 16/5 times the stereo pilot, 7/2 times this pilot would, for example, be chosen for the carrier frequency of the transmitter identification i sO ~
signal, the ~i-~-idcnd of the divider 34d might be equal to 14 and that of the divider 35 might be equal to 2.
s O ~
It is then, of course, simpIer to choose the ~i~h~
of 34d to be equal to 7 so that the divider~35 can be dispensed with.
In the embodiment of Fig. 4 the~units corres-pondlng to the functional unlts of Fig. 3 have been given the same reference numerals.
Whereas in the embodiment of Fig. 3 the phase `
; 30 comparison for the con-trol of the phase correGtor 36 is effected at doubLe the carri.er frequerlcy, it is effected ~,:
., ' .
PlIN . 8 G 9 1 C
~~~~ l0.~.1978 2~.~
:.
for the embodiment of Fig. 4 a-t the carrier frequency it-self. To this end the frequency doubling circuit 50 and the frequency divide-by-two divider 37 are omitted and the di~idcnd of the frequency 34d is reduced to 16.
Via the phase shifter 36 the divider 35 now supplies an unmodulated carrier of the carrier frequency (60.8 KHz) to the input 48 of the phase detector 47.
A phase inverter 51 (balanced modulated) is included in the input lead to the input 49 of the phase 10 detector 47. The phase inverter 51 is controlled by the output signal of the synchronous detector 39 or by, alternatively, the output signal of the pulse shaper l13.
Each time the phase of the transmitter identification `signal is changed 1800 owing to the code signal, this 1S causes a transient in the output signal of the pulse shaper 43 which effects a phase reversal by the phase inverter 51 so that, at the input 49, the original phase reversal is cancelled. So the input 49 of the phase detector 47 is supplied with the 60.8 KHz transmitter 20 identification carrier from which the original phase modulation was removed. In phase detector 47 the phase of this unmodulated carrier is compared relative to the wave at input 48 and any phase errors are compensated again by the phase shifter 36 via low-pass filter 46.
Instead of having been included in the lead to the input 49 the phase inverter 51 may also be in-cluded in the supply lead to the input 48 of the phase detector 47. The 60.8 KlIz carrler supplied via the phase shifter 36 is then phase-modulated by the binary code sig~nal in the same manner as the transmitter identification .~ signal itself h,ls been modulated. 'l~hen the phase detector 4'7 .
PIIN.8G(J1C
10.2.1978 .
supplies again an output voltage which can be used for the phase correction.
; A third possibility is to include the phase inverter in the output lead of the phase detector 47, either before or after the filter 46. The phase detector 47 itself` then supplies the binary code signal but as the phase inverter changes state of each signal transient of this signal, the output signal of the phase inverter becomes a d.c. voltage which can be used for the phase correction.
A 90 phase shifter 52 in the input lead 38 of the synchronous detector has a similar function as the 45 phase shifter 40 of Fig. 2. Alternatively, the ~~ phase shifter 52 may be includecl in the input lead 41 of the synchronous detector 39 or In one of the input leads of the phase detector.
A further analysis of the circuit of Fig. 4 sIlows that the entire phase correction system has two stàble control conditions wherein the phase difference of the signal at the input 49 relative to the signal at the input 48 of the phase detector can be +90 or -90.
The detection of the binary code signal by means of the : synchronous detector 39 has, consequently~ the same ambiguity as in the receiver of Fig. 3.
The circuits shown in Figs. 3 and 4 do not require resonant circuits which satisfy high selectivity ~-- requirements, because a large part of the required ~`
; selectivIty can be reaIIzed at low frequency, that is to SAy by Iow-pass filters (34b, 46, 42).~ Consequently, the bandpass filters 32 and 33 need have moderate quality fac-tors (approximately 20) oniy. Recent tests have sho~n .' ' ' .
PTI,~'. 8G9 1 c 10.2.197~
that the stereo pilot filter 32 may even be dispensed with completely. In some cases it is also possible to obtain from the stereo decoder 27 a stereo pilot which has already been filtered. The input of the phase locked loop 34 is then connected to a suitable point in the stereo decoder 27.
The functional unit shown in the Figs. 1, 3 and
PHN.8691~
` ~ 10.2.1978 .
:
with the following values: -` - The sub-carrier frequency was 66 KHz and the -- frequency sweep 1 KHz so that owing to the binary in-formation the frequency was switched between 65 KHz and - 5 67 KHz.
The code used was the 6-bit ASCII-code having 16 characters per information.
- The amplitude of the modulated code signal was chosen such that 1 KHz, i.e. 1,33 % of the total fre-quency sweep of 75 KHz available for the FM modulation of the main carrier, was occupied by the code signal.
The comparatively small amplitude (1 KHz) of this signal is opted for because experiments proved that a greater amplitude may cause interference noise in some FM re-~ 15 ceivers.
;- It appeared, however, that the necessarily ~ small amplitude of the modulated code signal and the .
~` ` comparatively high frequency thereof (66 KHz) result in a poo~ signal-to-noise ratio. In order to recover the ~` 20 code signal flawlessly, the receiver requires a high-. ~ .
grade filter having a good quality factor and a good "; temperature stability. In addition, it appeared that in ~", spite of the use of such a high-grade and expensive fil-~ - .
.; ter, decoding of the code signal no longer occurs flaw-lessly at aerial voltages below 10 /uV (at 60 Ohm) where-as the average FM receivers still furnish an acceptable . mono-reception at such aerial voltages.
It is an object of the invention to provide a : radio broadcasting system with code signaling which en-ables a substantially flawless decoding of the code signal at received aerial voltages where an acceptable .--, .
- .
, P~IN.8691C
10.~.19~
`
..
' mono-reception is not or hardly possible, whereas the signal reception in existing receivers is not or hardly disturbed and whereas, furthermore, high-grade and, con-sequently, expensive'filtering means in the receiver for the system according to the invention can be dispensed ; with and the radio broadcasting system according to the invention is therefore characterized in that said further - sub-carrier is a harmonic of a sub-harmonic of the stereo ~'` pilot not coinciding with a harmonic of this pilot which, ' 10 at the transmitter side, is derived from the same fre-: quency source as the stereo pilot and in that the code ~ signal is binary phase modulated on this su'b-carrier.
." The expression binary phase modulation must here, as customary, be understood to mean a phase mo-~ 15 dulation in which the phase of the sub-carrier is shifted "` 1800 by the binary code signal. This furnishes a modulated ` signal with a fully suppressed carrier.
Using phase modulation of the sub~carrier with the binary code signal (phase shift keying~ instead o~
- 20 frequency modulation (frequency shift keying~ results in an improvement of -the signal-to-noise ratio. In contrast ' to the demo~dulation of the frequency-modulated 'sub-carrler the demodulation of the phase-modulated sub-carrier requiresJ however, an unmodulatcd ("clean") sub-carrier. This sub-carrier is not present in the binary phase-modulated code signal because the sub-carrier it-` self is suppressed and sidebands only are transmitted.
Generating this sub-carrier at the receiving side can, however, be effected by squaring the incoming binary . 30 phase-modulated signal which results in a carrier having - double the frequency, by thereafter filtering this , i' I , ~ ' ' ' J ' ' . X (J '~ I C
~ ~r ~ ~ 10.~ 7~
.
carrier with double frequency, and subsequently having the carrier of the original frequency recovered from the carrier of double this frequency by means of a fre-quency divide-by-two divider.
With this method, in the case of poor signal-` to-noise ratios, the carrier of double the frequency must be obtained from a signal having a high degree of noise. If, for example, a so-called phase locked loop ` is used for this purpose, then this can indeed be ef-fected by using a low-pass filter having a low cut-off frequency in this loop in such a way that the phase of _ the voltage-controlled oscillator of the phase loop is ~ not modulated too much by noise, on the other hand such '~ ~ a low-pass filter having a low cut-off frequency reduces ~ 15 the pulling-in range of the phase locked loop to such t~; an extent that a voltage-controlled oscillator having a ~;~ very stable free-running frequency is now required. In ~m~ practice this can only be obtained with a crystal-con-trolled oscillator.
With the present invention wherein binary phase . modulation of the sub-carrier for the~code signal is `" used in combination with a ~requency relationship be-tween the stereo pilot and said sub-carrier, which ~-- relationship is fixed at the transmitter side, a system -~ 25 is obtained which can be decoded without high-grade means and which is comparatively insensitive to poor signal-to-noise conditions.
The sub-carrier for the transmitter ~dentifi-~; cation signal can now be recovered with much simpler means because the stereo pilot is modulated with a much ~ greater frequency sweep (10% of the total 75 KHz fre-! - 6 ~
` ~ l PHM.~691C
10.2.1978 2~.~
-quency sweep) on the main carrier than the transmitter identification signal itself ( 1 9 33~ of the total fre-quency sweep of 75 KHz). In a receiver for a system according to the invention phase errors may be produced owing to different delay times for the modulated code signal and for the stereo pilot in the tuner and in the intermediate frequency section of the receiver. Also phase multiplicities are produced because sub-carrier frequency (~k) of the modulated code signal is chosen , ........................................... .
; 10 equal to a "fractional" harmonic of the stereo pilot (c~);
this expression means that ~k = m W, where m and n are integers but n is not divisable by m. The frequency division required thereby in the transmitter ana in the receiver may produce these phase multiplicities.
In accordance with a further aspect of the invention an automatic phase corrector is used in a ~` receiver according to the invention which can adjust the phase of the unmodulated wave required for detecting the ,.~ .
modulated code signal relative to the modulated code ~; 20 signal i-tself. This phase corrector is controlled from a phase detector which compares the phase of the modulated code signal to the phase of the unmodulated wave obtained from the stereo pilot and depending on the result of this . :
comparison, the phase corrector corrects any phase errors.
As the carrler itself lS missing in the binary modulated code signal this, however, cannot be done without fur-ther measures.
A first method to solve this diffioulty is the use of frequency doubling of the modulated code signal which provides an unmodulated carrier of twice the sub-carrier frequency. This carrier of twice the sub-carrier . ~
- 7 - ~
~ ~ ' P~IN.8691C
: ~ ' ' ' 10.2.'1978 .
frequency is applied to one input of the phase detector, a wave of likewise twice the subcarrier frequency, which is cbtained by means of frequency multlplication and/or division of` the stereo pilot being applied to the other 5 input.
A second method consists in the use of a phase inverter in one of the input leads or in the output of the phase detector, which phase inverter is controlled ~r by the demodulated output signal of the synchronous detector. It appears that this results in both cases in à phase duplicity at the detection of-the code signal.
Thls phase duplicity is not disturbing if a code is used which is insensitive to such a duplicity, for example `. a so-called differential code; this is a code with which the two binary states are not transmitted by two phase conditions of the sub-carrier but by the occurrence or `
- non-occurrence, respectively, of a phase transition from ~` the one phase to the other or vice versa.
The frequency of the sub-carrier can, for example, ` 20 be chosen between the third and the fifth harmonic of the stereo pilot. Choosing lt below the third harmonic brings the sub-carrier too closely to the spectrum of the stereo information signal and choosing it above the fifth harmonic increases the chance for disturbances owing to adjacent transmitters.
Furthermore~ interference may occur in a number of stereo receivers between the sub-carrier for the transmitter identiflcation and the seoond harmonic of the 38 ~lz signal required for stereo de-teotion, which cor-responds to the fourth harmonic of the pilot. For this reason the sub carrier for the code signaling should not PHN.8691C
10.2.1978 ~,~
- be located too near this fourth harmonic.
Owing to the non-linear phase characteristic - of the intermedia-te freqllency section of the receiver an interference product having a frequency equal to the difference frequency between the sub-carrier and the stereo pilot occurs in the multiplex signal. This inter-ference product may, after detection with the 38 KHz . .
wave, cause audible noise if the sub-carrier is located . .~
too near the third harmonic of the stereo pilot.
The above-mentioned non-linear phase character-` istic furthermore causes noise in the region of the whole - harmonics of the stereo pilot. All these considerations lead to the choice that the sub-carrier for the code signal should notcoincide with a full harmonic of the stereo pilot. Consequently, it should be preferred to -' choose a "fractional" harmonic of the stereo pilot f`or the sub-carrier of the code signal and to remove the .` phase multiplicity then occurring in the receiver in the ~. .
; manner described above.
On the bases of the above consideration pre-ference should be given to a position of the sub-carrier half-way between two harmonics of the stereo pilot~ for example at 7/2 or 9/2 times the stereo pilot. The in-vention has been tested with a sub-carrier frequency of t 25 7/2 times the pilot frequency; for clearness' sake an ' ` embodiment for a sub-carrier frequency of 16~5 tirnes a pilot frequency is given.
A further improvement of a radio broadcastin~
; system with code signaling ln which, while maintaining a re]iable transmission of the code in~ormation, a reduced chance of disturbing existing receivers is possible, is ~- , `~ ~ 9 _ - ~ P~IN.8691C
_~ ' 10.~.1978 ;`' -: - characterized in that the additional sub-carrier with the ' modulated code signal is located in at least one of the . two halves of the frequency range, divided into two parts by the stereo pilot, between the upper limit of : -, 5 the frequency spectrum of the audio frequency information signal and the lower limit of the frequency spectrum of the modulated stereo information signal and in that the modulated code signal has an amplitude which causes the main carr.~er to deviate less than 1 KHz, preferably `, 10 0.25 KHz.
~ This measure has the following effects 1. Because the sub-.carrier for the code signaling ., .
is now remote from the higher harmonics of the 38 KHz stereo detection signal, these higher harmonics cannot : 1$ ~ produce audible noise in existing receivers.
2. Because the sub-carrier for the code signaling is now located much lower in the frequency spectrum of the multiplex signal, the signal-to-noise ratio is con-siderably more favourable. Consequently, the modulated code signaI can have a still smaller amplitude than was the case for a sub-carrier of, for example, 66.5 KHz.
. For comparison it should be noted that with the present - preferred embodlment, for obtaining a reliable code Slg-naling, the modulated code signal need only occupy ap-- 25 proximately 0.25 KHz of the maximum frequency sweep of 75 KHz. This requires approximately 1 KHz in case of a .66.5 KHz sub-carrier. Of course the much smaller sub-carrier amplitude considerably reduces the chance for interference noise.caused by other components of the multiplex signal.
A still further reduction of the chance for.
:,~ , . .
, .
:; . : . .
- PI~N.8691C
--- - 10.2.1978 ~; ~
. . - . .
- noise in existing receivers, particular in the case of monoreception, can be achieved in accordance with a fur-- ther characteristic of the invention when a sub-carrier whose phase is binary modulated by the code signal is located in each of the two halves of said frequency range, divided into two parts by the stereo pilot, and the two sub-carriers modulated by the code signal have equal amplitudes and such a phase relative to the stereo pilot that together with the stereo pilot they form a - 10 signal obtained by quadrature modulation of the stereo ;~ - pilot by a sub-carrier derived from the stereo pilot, this sub-carrier itself being binary phase modulated by the code signal.
.`. With such a signal each of the binary phase 1$ modulated sub-carrier signals may be considered to be a sideband of a double-sideband signal having the stereo pilot as the carrier. The stereo pilot is quadrature mo-dulated by a modulation signal which itself is binary --` phase modulated by the code signal. The modulation sig-nal has a frequency equal to the difference between the " frequency of the stereo pilot and of a sub-carrier. A
system according to this further feature of the invention, and tested in practice has, next to the stereo pilot with i ` a frequency fp of 19 KHz9 a first sub-carrier of 16.625 KHz (7/8 fp), whose phase is binary modulated by the code signal and a second sub~carri~r of 21.375 KHz (9/8 fp) whose phase is binary modula~ted by the code signal. In the case of equal amplitudes of the two sub~
carriers and a relative proper phase relation between the sub-carriers and the stereo pilot, the three signals together constitute a stereo pilot which is quadrature ~ ~ .
; P~N.8691C
, 10.2.1978 '' modulated by a sub-carrier signal of 1/8 f which itself ~, is binary phase modulated by the code signal. To this ~- end *he phase of one swb-carrier must lead the stereo ' pilot, shifted over 90, for the same amount as the other sub-carrier lags this 90-shifted stereo pilot, in other ' words the resultant of the two modulated sub-carriers has -, a 90 phase shift relative to the stereo pilot.
The sum of the stereo pilot and the two sub-carriers forms a pilot signal the amplitude of which is `- 10 substantially constant. As especially the amplitude va-" riations of the pilot give rise to distortion products owing to~the non-linear phase characteristic of the inter-~ mediate frequency section of the receivers the above-described measure offers an additional distortion reduction.
Within the framework of the invention it is ` alternatively possible to give the sub~carrier modulated by the code signals such a phase that the resultants al-ways coincide (0 or 1800) with the stereo pilot. The two sub-carriers functioning as sideband for the stereo pilot then cause an amplitude modulation of the stereo pllot by a carrier signal which itself is binary phase-modulated by the code signal.
When applying the stereo pilot, double~sideband quadrature or amplitude-modulated by the two sub-carriers or the stereo pllot single-sideband phase and amplitude-modulated by one sub-carrier, to the stereo decoder of radio receivers, the sub-carrier amplitudes which are al-ready small, are suppressed so much by the stereo pilo-t filter provided in such receivers, relative to the stereo pilot itself that disturbing the stereo detector does sub-- stantially not occur. Such a dis-turbance would be much : P~IN.c',~)~31C
~ 10.2.197~
. ~ ' ' . .
. . .
greater in the case of direct phase or amplitude modulation of the stereo pilot by the code signal.
This disturbance is, of course, also greater according as the sub-carriers are located nearer the ~tereo pilot (for example at 11/12 fp and/or 13~12 ~ ).
~" .
On the other hand~ in case of an excessive sub-carrier ~; stereo pilot distance the sub-carrier becomes located too near the frequency spectrum of the audio information signal or of the modulated stereo information signal. On the basis of these considerations a spacing of 1/8 fp between sub-carrier (sub-carriers respectively) and the stereo pilot should be preferred.
" As with the system according to the present ` embodiment the sub-carrier frequency is relatively near that of the stereo pilot it is preferred in the receiver arranged for receiving such signals to convert the mo-dulated sub-carrier (7/8 fp and/or 9/8 f ) first with the stereo pilot to an intermediate frequency (1/8 fp) which is harmonically related to the stereo pilot and which is equal to the difference between the sub-carrier frequency and stereo pilot rrequency. The synchronous detection of the code signal can then be effected at .~ this lower frequency in a corresponding manner as de-scribed above.
The invention will be further~exp]alned with ~ reference to the Figures in the accompanying drawings.
`` Herein: ~ ~
Figure 1 is a block diagram of an embodiment ` of a transmltter for a first implementation cf the system ;~ 30 according to the invention, Figure 2 shows the frequency spec-trum of the .
, .
P~IN.8691C
10.2.1978 ;~ multiplex signal generated in the first ernbodiment of the system according to the invention at the transmitter side and obtained at the receiver side after FM demodu-- lation, Figure 3 is a block diagram of a first embodi~
` ment of a receiver according to the invention, Figure 4 is a block diagram of a second embodi-ment of a receiver according to the invention, Figùre 5 and Figure 5a, respectively, are block diagrams of a transmitter for a second embodiment of the system according to the invention, Eigure 6 shows the frequency spectrum of the multiplex signal generated at the transmitter side and obtained at the receiver side after FM-demodulation according to the second embodiment of the system, Figure 7 is a block diagram of an embodiment .
of a receiver for receiving a signal as shown in Figure 6.
Figure 8 is a block diagram of a second embodi-ment of a receiver for receiving a signal as shown in Figure 6.
- The transmitter of Fig. 1 comprises a source . . .
` of left-hand audio signals 1 and a souroe of right-hand audio signals 2. Each of the left-hand and right-hand audio signals is applied via a pre-emphasis network 3 and -25 4, respectively, and via a low-pass filter 5 nnd 6, . respectively, having a cut-off frequency of 15 KHz,to an adder circuit 7 and to a subtractor circuit 8. There-. after the sum signal L-~R derived from the adder circuit is applied to an input 9 of a multiplexer 10. The dif-ference signal L-R of the subtrac-tor circuit 8 is modu-. latcd in a balance modulator 11 on a stereo sub-carrier ~ 1 1 1 ,s ,~
: . . . .
: - , . . .. , .. , . . ,. , ~ .. , P~IN.869lC
10.2.1978 ! ' .
of, for example, 38 KHz and the modulated stereo inform-ation signal thus obta:ined, which consists of two side-bands with suppressed stereo sub-carrier, is applied via a bandpass filter 12 to a second input 13 of the multiplexer 10.
In addition, the transmitter of Fig. 1 com-prises a stable oscillator 14, for example a crystal oscillator supplying a wave of, in general, 19 KHz which is used as stereo pilot. This stereo pilo-t is applied to .~ 10 a first input 15 of the multiplexer 10.
The stereo pilot of the oscillator 14 is also `~ applied to a so-called phase locked loop 16 which in-cludes a phase detector 16a, a low-pass filter 16b, a voltage-controlled oscillator 16c and a frequency-divide-1$ by-two divider 16d. The phase locked loop 16 is used for p.oducing a sub-carrier whose frequency (38 KHz) is equal to twice the frequency of the stereo pllot and which is locked to the stereo pilot. The operation of such a phase locked loop is known; the j8 KHz output signal of the oscillator 16c is converted in the divide-by-two divider ` 16d into a 19 KHz signal which is compared in the phase ~i~ detector 16a with the 19 KHz pilot of the oscillator 11~.
~ .
The output voltage of the phase detector 16a is filtered in low-pass filter 16b and applied as a control voltage : 25 to the oscillator 16c.
Via a phase shifter 17 the 38 I~Hz output signal of the phase locked loop 16 is applied as stereo sub-carrier to the modulator 11 for modu~ating the L-R signal.
The phase shifter 17 is used to give the sub-carrier the internationally prescribed phase relative to -the 19 KHz .
s-tereo pilot.
`; .
~, .
~ P~N.8691~
~ 2~ 10.2.l978 '' A second phase locked loop 18, connected to - the 19 KHz oscillator, comprises a phase de-tector 18a, a lowpass filter 18b, a voltage-controlled oscillator 18c and a 16-scaler 18d. The phase locked loop 18 operates in a similar manner as the phase locked loop 16 and supplies an output signal which is locked to the stereo pilot, of 304 KHz, i.e. 16 times the pilot frequency.
Thereafter the 304 KHz signal of the phase locked loop 18 is reduced in a divide-by-5 divider 19 to 60.8 KHz and the latter signal is applied as sub~carrier of the trans-mltter identification signal to the carrier input of a balanced modulator 20. The modulatlon input of this mo-dulator~is connected to a schematically shown arrangement 21 for generating a suitable binary code which includes the transmltter identification information, for example a code as defined in the preamble.
The modulator 20 may~ for example, be a ring ~` modulator or a dual long-tail circuit or any other known modulator which, under the influence of the bits derived `. 20 from the arrangement 21, shifts the phase of the 60.8 KHz signal from the divlde-by-5 divider 19 over 1800. The i 60.8 KHz signal, phase-modulated in this way, is applied via a bandpass filter 22 having a bandwidth of approxi-mately 4 KHz to a fourth input 23 of the multiplexer 10.
The multiplexer combines the signals at the inputs 9, 13, 15 and 23 and supplies them combined to a FM-trans-- mitt.er, not shown in the drawing.
- For further explanatlon, Figure 2 shows the frequency spectrum of the signal obtained at the output ~` 30 of the multiplexer. Between 0 and 15 KHz there is the sum signal I~R supplied v:La the input 9, at 19 K~Iz there I G
~ .
~ PflN.oG91C
~ r~ Z~ .~ . 0 . 2.1978 is the stereo pilot supplied via the ~nput 15~ between 23 and 53 KHz there lS the L-R signal which is modulated at 38 KHz and which is supplied via the lnput 13 and at 60.8 KHz there is the approximately 4 KHz wide trans-mitter identification signal which is supplied via the input 23. It should be noted that the relative amplitude ratios generally deviate more from one another than is indicated for clearness~ sake. In general the stereo pilot is approximately 9x smaller than the L~R and L-R
components and the amplitude of the transmi-tter identi-fication signal is preferably chosen to be approximately 10x smaller than that of the stereo pilot.
The receiver of Fig. 3 comprises a tuner 24, an lntermediate frequency amplifier 25 and an FM-detector 26. The multiplex signal which is composed of the com-' ponents shown in Fig. 2 is available at the output of ~` this FM detector. For a stereo receiver tbls multiplex -~ `signal is applied to a stereo decoder 27 which supplies the left-hand and right-hand audio signals which are ` ~ 20 supplied vla audio amplifiers 28 and 29 to a left-hand and a right-hand loudspeaker 30 and 31.
For demodulating the transmitter identification signal the multiplex signal is applied to a 19 KHz band-pass filter 32 for the stereo pilot and a 60.8 KHz band-pass filter 33 for the transmi-tter identifica-tion slgnal.
The stereo pilot ~iltered out by means of the filter 32 is additionally filtered and its frequency multiplied by a phase locked loop 34 which includes a phase detector ~- 34a, a low-pass filter 34b, a voltage-controlled oscil-lator 34c and a 1:32 frequency divider 34d. I-ts operation `~` is similar to that of the phase loclced loops 16 and 18 ~. , .
.
E)HN.8G9 1 c 10.2.1978 ,. ' -of Fig. 1.
The output wave of the phase locked loop 34, which has a frequency of 32x19=608 KHz, is thereafter reduced to 121.6 KHz in a divide-by-five divider 35, thereafter passed through a controllable phase shifter 36, ~hose function will be explained in the course of ` this description, thereafter divided in a divide-by-two divider 37 to 60.8 KHz and, finally, applied to a first input 38 of a synchronous demodula-tor 39.
The 60.8 KHz phase modulated transmitter iden-tification signal originating from the bandpass filter 33 is applied via a 45 phase shifter 40 to a second inpu-t 41 of the synchronous demodulator 39. The syn-chronous detection of the 60.8 KHz phase modulated trans-mitter identification signal at the input 41 by means of the unmodulated 60.8 KHz wave at the input 38 furnishes, at the output of the synchronous demodulator 39, the demodulated binarv transmitter identification signals.
This binary code signal is passed through a low-pass filter 42, thereafter it is formed into square pulses in a pulse shaper 43 and applied to a decoder 44. This decoder converts the binary transmitter identifica-tion ; signal iDto signals suitable for driving a "display" 45 For a proper synchronous detection in the - 25 demodulator 39, the unmodulated wave at the input 38 ` must have the proper phase relation relative to the . ~
modulated signal applied to the input 41. ID general this proper phase relation is not guaranteed owing to ~, . . .
the following causes: ~
~` 30 1. Owing to the insufficient linear phase charact-,~ `eristic of the intermediate frequency amplifier 25 -the ~, .
PIIN.8691C
- 10.2.1978 _ , . , 19 ~Hz stereo pilot and the 60.8 KHY. transrnitter iden-tification signal may be subjected to mutually different delay times.
2. The input filters 32 and 33 may effect unwanted phase shifts.
3. Owing to the frequency division by the divider 19 in the transmitter the phase of the transmitted 60.8 KHz transmitter identification signal is no longer unambiguously determined relative to the transmitted - 10 stereo pilot. A similar phase multiplicity is caused by ~` the frequency divider 35 in the receiver.
'~ In order to obvIate all these phase problems the circuit of Flg. 3 comprises the adjustable phase shifter 36 mentioned above. This shifter is controlled via ~ low-pass filter 46 from a phase detector 47. The phase detector 47 has two inputs 48 and 49, the input 48 being connected to the output wave o~ the phase shifter 36, the Input 49 being connected to the output of a device 50 which produces a modulatéd wave of double the frequency (namely 121.6 KH~) from the phase-modulated signal of the filter 33. To this end the device 50 has a non-linear characteristlc with even~power term, for example a squaring circuit or a full-wave rectifier.
Becauss the phase locked loop 34 multiplies the stereo pllot by a factor of 2 more than necessary for the synchronous detection, the frequency of the wave which IS applied to the input 48 of the phase detector is equal to double the carrier frequency. So measuring the phase by means of the phase detector 47 is ef~ected at double ths carrier frequency and the result of the measuremeIl1, is used to compensate ~or the above-mentioned ~, .
. :
. ~
- PHN.86~1C
10.2. 1978 .
unwanted phase shifts in the controllable phase shifter 36. It should be noted that the frequencies of the two signals which are connected to the phase detector 47 are always equal to one another so that no pulling-in pro-`blems can arise. The phase errors which are correctedtherewith vary only slowly and the low-pass filter 46 can, consequently, have a very low cut-off frequency (for example 10 Hz). Owing to this low cut-off frequency, rapid phase variations which might be produced owing to the noise in the transmission path 33-40-41 of the trans-mitter identification signal can be effectively suppressed.
By means of the specified measures it is possible to `~ obtain for the synchronous detector 39 an unmodulated wave of the proper frequency and the proper phase and - 15 whicll is noise-free to a sufficient degree. As, owing to the phase control by means of 36, 46, L~7 the phase is compensated at double the carrier frequency, the phase relation at the inputs of the synchronous detec-tor 39 is still not unambiguous (180 phase duplicity). When using a (differential) code which is insensitive thereto a proper transmission of the code signal can, however, ` yet be guaranteed.
.; In practice, the automatic phase control by means of the phase controller 36 always operates so . 25 that the two lnput signals of the phase detector L~7 are shifted 90 in phase relative to one another. Moreover it is desirable that the mutual phase relation between the input slgnals of the synchronous detector is 0 or 180. If the divide-by-two divider 37 is so constructed .
~` 30 that the zero-crossings of its output wave coincide with . ~ ~
`~ the zero-crossings of its input wave and :if the tre-., ~ .
. - .
, . PIIN.~91C
- - lO~J~ 1978 ` quency doubler 50 is implemented as a squar:ing clrcult ; ln which the tops of the input wave co:inc:ide with tops of the output wave, then this preferred phase relation is achieved automatlcally. In other cases an addltional phase correctlon may be necessary in one of the input leads of the detectors 47 and 39, for example a 90 correction for the double carrler frequency or a 45 correction for the carrier frequency itself. The 45 phase shifter 40 ls used for this purpose. It should be noted that several variants of the clrcult of Fig. 3 are possible. It is, for example, possible to replace the divide-by-two divider 37 by a frequency doubler in the input lead 48 of phase detector 47. The frequency mul-tiplication fac-tor of the phase locked loop 34 should then be a factor of 2 smaller. Alternatively it is, for example, possible to lnclude the phase controller 36 in the output lead of the filter 33.
If, instead of 16/5 times the stereo pilot, 7/2 times this pilot would, for example, be chosen for the carrier frequency of the transmitter identification i sO ~
signal, the ~i-~-idcnd of the divider 34d might be equal to 14 and that of the divider 35 might be equal to 2.
s O ~
It is then, of course, simpIer to choose the ~i~h~
of 34d to be equal to 7 so that the divider~35 can be dispensed with.
In the embodiment of Fig. 4 the~units corres-pondlng to the functional unlts of Fig. 3 have been given the same reference numerals.
Whereas in the embodiment of Fig. 3 the phase `
; 30 comparison for the con-trol of the phase correGtor 36 is effected at doubLe the carri.er frequerlcy, it is effected ~,:
., ' .
PlIN . 8 G 9 1 C
~~~~ l0.~.1978 2~.~
:.
for the embodiment of Fig. 4 a-t the carrier frequency it-self. To this end the frequency doubling circuit 50 and the frequency divide-by-two divider 37 are omitted and the di~idcnd of the frequency 34d is reduced to 16.
Via the phase shifter 36 the divider 35 now supplies an unmodulated carrier of the carrier frequency (60.8 KHz) to the input 48 of the phase detector 47.
A phase inverter 51 (balanced modulated) is included in the input lead to the input 49 of the phase 10 detector 47. The phase inverter 51 is controlled by the output signal of the synchronous detector 39 or by, alternatively, the output signal of the pulse shaper l13.
Each time the phase of the transmitter identification `signal is changed 1800 owing to the code signal, this 1S causes a transient in the output signal of the pulse shaper 43 which effects a phase reversal by the phase inverter 51 so that, at the input 49, the original phase reversal is cancelled. So the input 49 of the phase detector 47 is supplied with the 60.8 KHz transmitter 20 identification carrier from which the original phase modulation was removed. In phase detector 47 the phase of this unmodulated carrier is compared relative to the wave at input 48 and any phase errors are compensated again by the phase shifter 36 via low-pass filter 46.
Instead of having been included in the lead to the input 49 the phase inverter 51 may also be in-cluded in the supply lead to the input 48 of the phase detector 47. The 60.8 KlIz carrler supplied via the phase shifter 36 is then phase-modulated by the binary code sig~nal in the same manner as the transmitter identification .~ signal itself h,ls been modulated. 'l~hen the phase detector 4'7 .
PIIN.8G(J1C
10.2.1978 .
supplies again an output voltage which can be used for the phase correction.
; A third possibility is to include the phase inverter in the output lead of the phase detector 47, either before or after the filter 46. The phase detector 47 itself` then supplies the binary code signal but as the phase inverter changes state of each signal transient of this signal, the output signal of the phase inverter becomes a d.c. voltage which can be used for the phase correction.
A 90 phase shifter 52 in the input lead 38 of the synchronous detector has a similar function as the 45 phase shifter 40 of Fig. 2. Alternatively, the ~~ phase shifter 52 may be includecl in the input lead 41 of the synchronous detector 39 or In one of the input leads of the phase detector.
A further analysis of the circuit of Fig. 4 sIlows that the entire phase correction system has two stàble control conditions wherein the phase difference of the signal at the input 49 relative to the signal at the input 48 of the phase detector can be +90 or -90.
The detection of the binary code signal by means of the : synchronous detector 39 has, consequently~ the same ambiguity as in the receiver of Fig. 3.
The circuits shown in Figs. 3 and 4 do not require resonant circuits which satisfy high selectivity ~-- requirements, because a large part of the required ~`
; selectivIty can be reaIIzed at low frequency, that is to SAy by Iow-pass filters (34b, 46, 42).~ Consequently, the bandpass filters 32 and 33 need have moderate quality fac-tors (approximately 20) oniy. Recent tests have sho~n .' ' ' .
PTI,~'. 8G9 1 c 10.2.197~
that the stereo pilot filter 32 may even be dispensed with completely. In some cases it is also possible to obtain from the stereo decoder 27 a stereo pilot which has already been filtered. The input of the phase locked loop 34 is then connected to a suitable point in the stereo decoder 27.
The functional unit shown in the Figs. 1, 3 and
4 are all known per se, and, consequently, require no further explanation.
The transmitter of Fig. 5 comprises a stereo multiplex encoder ~101 to which sources 102 and 103 of left-hand and right-hand audio signals, respectively, are connected and a 19 ~Hz oscillator 104 which gene-rates a stereo pilot fp. The encoder 101 composes in a similar manner as described with reference to Fig. 1 the standard multiplex signal from the applied signals, said standard multiplex signal comprising the audio frequency sum sLgnal L+R, the L-R stereo information signal modu-lated on a suppressed carrie~ of twice the pilot fre-quency, as well as the stereo pilot fp itself. It shouldbe assumed that the stereo pilot derived from the oscil lator 104 has the same phase as the pilot in the multi-plex signal.
The stereo pilot is applied directly to a first contact ~ and, in addition, via a 90 phase shifter ~ ;~5~
105 to a s~d contact ~ of a switch 106. The master contact c of the switch 106 is connected to a first iIl-put 107 oE a linear modulator 108. Therefore, switch 106 - being in the position shown in the drawing, a stereo pilot shifted 90 relative to the stereo pilot in the mtlltiplex signal :is appliecl -to this input. :~n the other zLI _ ... .
` P~IN.8691C
10.2.1978 position of the switch the input 107 of the modulator ~o8 receives the stereo pilot in phase with tl-lat in the multiplex signal.
Furthermore, the stereo pilot is applied via a pulse shaper 109 to a frequency divider 110 which fur-nishes a square wave of 1/8x the stereo pilot frequency (2.375 KHz)~ A band filter 111, tuned -to this frequency, filters the fundamental frequency so that a sinusoidal wave of 1/8x the pllot frequency is available at the second input 112 of the modulator 108.
The modulator 108 is a linear balanced modu-lator which produces the sum and the difference fre-quencies (fp-fp/8 and fp+fp/8) from the two applied sinusoidal signals, while the frequencies (fp and fp/8) originally applied, are rnissing from the output signal.
~o~
The output signal of the modulator ~e~ is ` thereafter directly applied to a first contact a of a three-position switch 113, and also, via a bandpass fil~
ter 114, tuned to 16.625 KHz (7/8 fp) to a second con-tact b and via a bandpass filter 115, tuned to 21. 375 KHz (9/8 fp) to a third contact c of the three-position switch 113. The master contact d of the three-position ~ switch 113 is connècted to a first input 116~ of a linear - balanced modulator 117. A device 120 wh-ch supplies the binary code signal comprising the transmitter identifi-cation information is connected via a trapezoidal wave-.
shaper 119 to the second input 118 of this modulator 117.
The trapezoldal wave-shaper 119 reduces the higher fre-quency component's contents, so that the code signal applied to the modulator comprises a limited frequency range (up to approxirnateJy 600 llz~.
.
P~IN.86'31C
10.2.1978 f3~
In the third position (c) of switch 113 the 9/8 fp sub-carrier passed by the filter 115 is binary phase-modulated in the modulator 117 with the code signal of the device 120. In the second position (b) of switch 113 the 7/8 fp sub-carrier passed by filter 11~ is binary phase-modulated by the code signal. In the first position (a) of switch 113 the two sub-carriers (7/8 f and 9/8 fp), originating from modulator 108, are both binary phase-modulated by the code signal. Finally, the output signal of the modulator 117 is added to the stereo multiplex signal of the encoder 101 in an adder stage 121, all this in such a way that the amplitude of the added sub-carrier or sub-carriers, respectively, is con-siderably (for example 30 times) smaller than the ampli-tude of the stereo pilot comprised in the multiple~ signal.
~inally, the output signal of the adder stage 121 is applied to a FM-transmitter, not shown.
In position (a~ of the two switches 106 and 113 the whole, transmitted, signal comprises next to the stereo pilot fp the sub-carriers f +1/8 f and f -1/8 f which are both binary phase-modulated by the code signal.
The resultant of the two sub-carriers is always shifted over 90 relative to the stereo pilot so that the stereo pilot with the two sub-carriers as sidebands form a sig-nal which is quadrature modulated, so that the stereo pilot is amplitude-modulated to a very limited extent ` only. The modulating signal itself is a sub-carrier of 1/8 fp which is binary phase-modulated by the code signal.
In the second position (b) of the switch 106 the resultant of the two sub-carriers JS in-phase or (1800 out of phase) with the stereo pi:lot so tha-t the stereo pilot with -the ~- - 26 -P~ G9l(, - 10.2.1978 .
two sub-carriers as sidebands forms a signal which is amplitude-modu:Lated,7but n~ot phase-modulated with the modulated 1/8 f signal.~
In the second or third positlon, respectively, of the switch 113 only the lower or upper sideband, respectively, is added to the stereo pilot of the multi-ple~ signal. S~itching over of switch 106 results indeed in a 90 phase shift of the single sideband relative to the stereo pilot but this of little practical im-portance.
It will ~be obvious that the diagram of Figure 5 relates to an eXperimental transmitter suitable for testing which system will be most satisfactory in practice.
In its definite version the transmitter needs only be suitable for one system and may, therefore, be of a simpler implementation. Thus, a transmitter wherein only one modulated sub-carrier of for example 7/8 fp or 9/8 fp is applied to the multiplex signaL (see ~ig. 5a) might comprise a phase locked loop 122 deriving a signal of 7f or 9fp from the stereo pllot, furtherrnore an 8-scaler 123 for providing a pulse-shaped signal of 7/8f or 9/8fp, furthermore a band pass filter 124 for converting tbe pulse-shaped signal into a sinusoidal signal of 7/8fp or 9/8fp, this sinusoidal signal being applied to the first input i16 of the modulator 117.
" .
In a definitive transmitter for a system with two modulated sub-carriers the elements 106, 113, 114, 115 of Fig. ~ can be dispensea wlth. The oscillator 104 can then be connested directly or via the phase shil-ter ~` 30 105 to the I`irs-t input 107 of the modulator 108 and the output of 1oS diL-ec-tly to -tlle flrs-t iIlpUt 106 o~ the .
~ 7 -. ' .8~lC
10.2.1978 f~Z~.~
modulator 117. Instead of first mixing the pilot fp with the f`p/8 signal and by rnodulating the result thereafter by the code signal it is also possible -to modulate the f /8 signal by the code signal and to mix it thereafter with the stereo pilot or to modulate the stereo pilot fp by the code signal and to mix it thereaf-ter with the fp/8 signal.
Fig. 6 shows the frequencv spectrum of the signal supplied by the adder stage ~t~. The Figure shows from 0-15KHz the audio frequency inf`ormation signal, at 19 KH~ the stereo pilot, at 23 KHz up to 53 KHz (not visible) the stereo information signal modulated at 38 KHz, and at 16.625 and 21.375 KHz the two binary phase modulated sub-carriers ? each having a bandwldth of ap-proximately 1200 Hz. It should be noted that the am-plitudes of the signal components differ co~d~rably more from one another than shown in the Figure for clearness sake. In practice the L-~R and L-R signal components may . ~
be approximately ~ times greater than the stereo pilot, while the two sub-carrier signals may, f`or example, be 30 times smaller than the stereo pilot.
The receiver of Fig. 7 ls especlally suitable `~ for a system in which only one binary phase modulated sub-carrier is transmitted at, for example 7/8fp . 25 (16.625 }~Hz). The customary rsceiver elements, such as `` high frequsncy, intermediate frequency and low frequency stages are not shown in Fig. ~. The multiplex signal ~--`. der~ed from the frequency discriminator of -the receiver ~i is applie-l to a bandpass filter 125 tuned to the sub-calrier frequency of 16.625 KHz and may have an effective ` quality f`ac1;or oI`, for exan1ple, 15. This filter passeC.
: - ~8 -.,. "
~;:
- PIIN.869lC
10.Z.197$
the moclulated sub-carrier f`requency as well as the stereo pilot itself which, although coinciding with an edge of the filter is still considerably greater than the sub-carrier signal. After having been amplified in an amplifier 126 the two signals are applied to a first in-put 127 of` a dual-function multiplier stage 128.
Firstly, stage 128 operates as a phase detector in a phase locked loop which comprises, in addition, a low-pass filter 129, a voltage-controlled 38 KHz oscil-lator 130 and a divide-by-two divider 131, the latter feeding a 19 KHz square wave back to a second input 132 of the multiplier stage 1280 This phase locked loop locks into the received stereo pilot and consequently supplies, at the output oP the divider 131 an 19 KHz square wave which is synchronized with the received stereo pilot. The low-pass filter 129 used for preventing the phase locked loop being influenced by other signal co~nponents than - the stereo pilot may have a cut-off frequency of, for example, 300 Hz and a slope above this cut-oPf frequency of 6 db/octave.
Secondly, the multiplier stage 128 operates as a mixer stage for the modulated 16.625 KHz (7/8fp) sub-carr1er. This sub-carrier is mixed with the 19 KHz (fp) square wave at the input 132 which results in a binary phase modulated intermediate frequency signal of ~` 2.37~ KHz (1/8fp), which is passed on via a low-pass filter 133 having a cut-oPf Prequency of, for example, 3 KHz and a hlgh f`requency slope oP 20 db/octave.
,~ Instead of a voltage-controlled 19 KHz oscil-~. :
lator a voltage-con-trolled 38 K~Iz oscillator 130 fol]owed by a div-ide-b~-two divider 131 :is used because, in . ~ . .
, - 29 -, .
P~l~.8691C
.2.1978 general, a divide-by-two divider supplies a more sym-metrical square wave than a voltage-controli.ed oscilla-tor. Consequently, stage 128 is controlle~d by a perfect-ly symmetrical square wave so that input signal compo-nents around the even harmonics of 19 KH~, in particular around 38 KHz, do not influence the output signal. of the stage 128. The detection of signal components around 57 I~Hz by stage 128 is prevented to a sufficient degree by filter 125 which has an adequate attenuation for these signal components. Consequently, by means of the elements 128, 129, i30 and 131 a flltered stereo pilot is avail-able at the output of the divide-by-two divider 131 and a con~erted binary modulated sub-carrier at the output - of stage 128. However, it will. be obvious that these fùnctions can be performed by any other suitable filter and converter arrangement.
The 19 KHz square wave of the divide-by-two divider 131 is divided in a divide-by-eight divider 134 : into a square wave having a frequency of 2.375 KHz (1/8fp). There:fore, a binary phase modulated 2.375 KHz~
; carrier signal is available at the output of filter 133 and an unmodulated 2.375~ KHz square wave, derived from .- the stereo pilot, at the output of the scaler 13~. The ... .
~, modulated carrier signal can now be demodulated syn-chronou.sly by means of the unmodulated wave and be pro-, .
cessed in.accordance with one of the methods descri.bed with reference to the Figures 3 and ~f. The actual detecti~n is effectecl in a synchronous.de-tector 135 to a flrst input :136 whcreo~ ~he modulated signa`l is applied ~; 30 vi.a an amplifier 137, while the unmodulated wave is app:ii.ed to a second inpu-t 138 v:La a controllable phass s ,. .
- 3(~ --, ,~ . , : . . . .
-. :- . . ~ "
P~-IN.~G~JlC
~ 10.2.1~)78 shifter l39. The detecte(l code signals are filterecl ln a low-pass filter 140 having a cut-off frequency of, for example, 350 Hz and a 20 db/octave slope, thereafter converted into square pulses by means of a pulse shaper 141, thereaf~r applied to a decoder 142 which converts the binary transmitter identification signal thus ob-tained into signals suitable for application to a load 143. The load 143 can be different, depending on the in-formation comprised in the code. In case the code com-prises information about the transmitter received and/or the program received, the load ~3 rnay comprise a display displaying this information, so that, for example, the customary tuning sca]e can be dispensed with. Alternatively ` the load 143 may comprise an automatic transmitter-search circuit so that the receiver only tunes to those stations broadcasting a given type of program, for example, classical music. If the code comprises time indication the device 143 can, for example, switch a tape recorder connected to it on or off at a preset time. If the code is a semaphoDe signal the device 143 is formed by a semaphone receiver.
- The phase shifter 139 serves for eliminating all phase errors wh:ich may be produced betwe~en the modu-~ lated 2.375 KHzsignal andthe unmodulated 2.375 KHz square '~ 25 wave. These phase errors can be produced at the divider : 110 in the transmitter and the divider 134 in the re-, ceiver and by delay time differences in the various fil-ters,~ for example in filter 1Z5. For the benefit~of this phase control the binary phase modulated 2.375 K~z "; 30 signal is converted by means oI a squarin~ de-vice 1LIL~
.~ ~ . , .
-;" and a pulse shaper ILl5 into a`4 75 KIIY. squaIe wave. The .', , lC
~ z ~ .2.197~
unlnodulated wave of the phase shifter 139 is converted into a L~.75 KHz square wave by means of a frequcncy mul-tiplier 146. The two ll,75 Kllz square waves are compared in a phase detector 147 which generates a control sig-nal from these square waves which is applied, after fil-tering in a low-pass filter ll~8 and amplification in an amplifier 1~l9, to the control input 150 of the controllable phase shifter 139. Thus, the phase shifter 139 ensures ~- that the 2.37~ KHz square wave and the 2.375 KHz signals e~
applied to the synchronous ~hr~t-~ have the same phase (or are 1800 out of phase respectively). The phase duplicity .
then still occurring can again be offset by the use of a code insensitive thereto. The phase shifter 139 as well as the phase shifter 36 of Figs. 3 and ll can, for example, consist of two cascade-arranged monostable circuits, the time constant of the first circuit being controlled by the control signal and that of the second circuit being equal to half a cycle of the signal to be delayed, the first circuit being started by the incoming signal and the second circuit by the trailing edge of the first circuit. Such a phase shifter is capable of shifting the phase of the signal over nearly 3600, which is more than . . , sufficient since the signal should be shifted over 180. -`` A preferred embodiment of a receiver for re-cèiving signals comprising two binary phase modulated ~,;
sub-carriers on either side of the stereo pilot, the stereo pilot being in quadrature to the resultant of the , ., two sub-carriers may be of the same shape as shown in '~ Fig. 7, it being understood that the filter 125 is not tuned to one sub-carrier but to the stereo pilot, whlle the passband ~idth of the f`:ilter must be sufficiently . , .~
. .
- '12 -.
' . 8~9 1 (~
10 . 2 1 ~78 large to pass the two sub-carriers. On the other hand, the attenuation outside the passband, in particular for signals around 57 KHz, must be sufficiently high to pre-vent disturbances~
In a receiver for receiving signals having a - binary phase modulated sub-carrier on either side of the stereo pilot, the resultant of the two sub-carriers being in phase with the stereo pilot, it is not possible to use the phase detector (128) of the phase locked loop also for down-converting the signal, because the phase detector and the mixer stage must then be controlled with stereo pilots which are shifted 90 relative to one another. Fig. 8 shows a possib]e embodiment of such a receiver in which corresponding elements have been given the same reference numerals as in Fig.- ~.
The output signal of the amplifier 126 is then applied to the phase detector 128 as well as to a second detector 1`51, which functions as a mixer stage. In this mixer stage the input signal is mixed with a 19 KHz square wave which is der1ved by means of frequency division by Q divide-by-two divider 15Z from the 38 KHz oscillator ;~ 130. The two dividers 131 and 152 are s~ energized that `. they supply 19 KHz waves whose phases are sh1fted 90 relative to one another.
.
.
`'; ' ' ' ' ; - 33 -
The transmitter of Fig. 5 comprises a stereo multiplex encoder ~101 to which sources 102 and 103 of left-hand and right-hand audio signals, respectively, are connected and a 19 ~Hz oscillator 104 which gene-rates a stereo pilot fp. The encoder 101 composes in a similar manner as described with reference to Fig. 1 the standard multiplex signal from the applied signals, said standard multiplex signal comprising the audio frequency sum sLgnal L+R, the L-R stereo information signal modu-lated on a suppressed carrie~ of twice the pilot fre-quency, as well as the stereo pilot fp itself. It shouldbe assumed that the stereo pilot derived from the oscil lator 104 has the same phase as the pilot in the multi-plex signal.
The stereo pilot is applied directly to a first contact ~ and, in addition, via a 90 phase shifter ~ ;~5~
105 to a s~d contact ~ of a switch 106. The master contact c of the switch 106 is connected to a first iIl-put 107 oE a linear modulator 108. Therefore, switch 106 - being in the position shown in the drawing, a stereo pilot shifted 90 relative to the stereo pilot in the mtlltiplex signal :is appliecl -to this input. :~n the other zLI _ ... .
` P~IN.8691C
10.2.1978 position of the switch the input 107 of the modulator ~o8 receives the stereo pilot in phase with tl-lat in the multiplex signal.
Furthermore, the stereo pilot is applied via a pulse shaper 109 to a frequency divider 110 which fur-nishes a square wave of 1/8x the stereo pilot frequency (2.375 KHz)~ A band filter 111, tuned -to this frequency, filters the fundamental frequency so that a sinusoidal wave of 1/8x the pllot frequency is available at the second input 112 of the modulator 108.
The modulator 108 is a linear balanced modu-lator which produces the sum and the difference fre-quencies (fp-fp/8 and fp+fp/8) from the two applied sinusoidal signals, while the frequencies (fp and fp/8) originally applied, are rnissing from the output signal.
~o~
The output signal of the modulator ~e~ is ` thereafter directly applied to a first contact a of a three-position switch 113, and also, via a bandpass fil~
ter 114, tuned to 16.625 KHz (7/8 fp) to a second con-tact b and via a bandpass filter 115, tuned to 21. 375 KHz (9/8 fp) to a third contact c of the three-position switch 113. The master contact d of the three-position ~ switch 113 is connècted to a first input 116~ of a linear - balanced modulator 117. A device 120 wh-ch supplies the binary code signal comprising the transmitter identifi-cation information is connected via a trapezoidal wave-.
shaper 119 to the second input 118 of this modulator 117.
The trapezoldal wave-shaper 119 reduces the higher fre-quency component's contents, so that the code signal applied to the modulator comprises a limited frequency range (up to approxirnateJy 600 llz~.
.
P~IN.86'31C
10.2.1978 f3~
In the third position (c) of switch 113 the 9/8 fp sub-carrier passed by the filter 115 is binary phase-modulated in the modulator 117 with the code signal of the device 120. In the second position (b) of switch 113 the 7/8 fp sub-carrier passed by filter 11~ is binary phase-modulated by the code signal. In the first position (a) of switch 113 the two sub-carriers (7/8 f and 9/8 fp), originating from modulator 108, are both binary phase-modulated by the code signal. Finally, the output signal of the modulator 117 is added to the stereo multiplex signal of the encoder 101 in an adder stage 121, all this in such a way that the amplitude of the added sub-carrier or sub-carriers, respectively, is con-siderably (for example 30 times) smaller than the ampli-tude of the stereo pilot comprised in the multiple~ signal.
~inally, the output signal of the adder stage 121 is applied to a FM-transmitter, not shown.
In position (a~ of the two switches 106 and 113 the whole, transmitted, signal comprises next to the stereo pilot fp the sub-carriers f +1/8 f and f -1/8 f which are both binary phase-modulated by the code signal.
The resultant of the two sub-carriers is always shifted over 90 relative to the stereo pilot so that the stereo pilot with the two sub-carriers as sidebands form a sig-nal which is quadrature modulated, so that the stereo pilot is amplitude-modulated to a very limited extent ` only. The modulating signal itself is a sub-carrier of 1/8 fp which is binary phase-modulated by the code signal.
In the second position (b) of the switch 106 the resultant of the two sub-carriers JS in-phase or (1800 out of phase) with the stereo pi:lot so tha-t the stereo pilot with -the ~- - 26 -P~ G9l(, - 10.2.1978 .
two sub-carriers as sidebands forms a signal which is amplitude-modu:Lated,7but n~ot phase-modulated with the modulated 1/8 f signal.~
In the second or third positlon, respectively, of the switch 113 only the lower or upper sideband, respectively, is added to the stereo pilot of the multi-ple~ signal. S~itching over of switch 106 results indeed in a 90 phase shift of the single sideband relative to the stereo pilot but this of little practical im-portance.
It will ~be obvious that the diagram of Figure 5 relates to an eXperimental transmitter suitable for testing which system will be most satisfactory in practice.
In its definite version the transmitter needs only be suitable for one system and may, therefore, be of a simpler implementation. Thus, a transmitter wherein only one modulated sub-carrier of for example 7/8 fp or 9/8 fp is applied to the multiplex signaL (see ~ig. 5a) might comprise a phase locked loop 122 deriving a signal of 7f or 9fp from the stereo pllot, furtherrnore an 8-scaler 123 for providing a pulse-shaped signal of 7/8f or 9/8fp, furthermore a band pass filter 124 for converting tbe pulse-shaped signal into a sinusoidal signal of 7/8fp or 9/8fp, this sinusoidal signal being applied to the first input i16 of the modulator 117.
" .
In a definitive transmitter for a system with two modulated sub-carriers the elements 106, 113, 114, 115 of Fig. ~ can be dispensea wlth. The oscillator 104 can then be connested directly or via the phase shil-ter ~` 30 105 to the I`irs-t input 107 of the modulator 108 and the output of 1oS diL-ec-tly to -tlle flrs-t iIlpUt 106 o~ the .
~ 7 -. ' .8~lC
10.2.1978 f~Z~.~
modulator 117. Instead of first mixing the pilot fp with the f`p/8 signal and by rnodulating the result thereafter by the code signal it is also possible -to modulate the f /8 signal by the code signal and to mix it thereafter with the stereo pilot or to modulate the stereo pilot fp by the code signal and to mix it thereaf-ter with the fp/8 signal.
Fig. 6 shows the frequencv spectrum of the signal supplied by the adder stage ~t~. The Figure shows from 0-15KHz the audio frequency inf`ormation signal, at 19 KH~ the stereo pilot, at 23 KHz up to 53 KHz (not visible) the stereo information signal modulated at 38 KHz, and at 16.625 and 21.375 KHz the two binary phase modulated sub-carriers ? each having a bandwldth of ap-proximately 1200 Hz. It should be noted that the am-plitudes of the signal components differ co~d~rably more from one another than shown in the Figure for clearness sake. In practice the L-~R and L-R signal components may . ~
be approximately ~ times greater than the stereo pilot, while the two sub-carrier signals may, f`or example, be 30 times smaller than the stereo pilot.
The receiver of Fig. 7 ls especlally suitable `~ for a system in which only one binary phase modulated sub-carrier is transmitted at, for example 7/8fp . 25 (16.625 }~Hz). The customary rsceiver elements, such as `` high frequsncy, intermediate frequency and low frequency stages are not shown in Fig. ~. The multiplex signal ~--`. der~ed from the frequency discriminator of -the receiver ~i is applie-l to a bandpass filter 125 tuned to the sub-calrier frequency of 16.625 KHz and may have an effective ` quality f`ac1;or oI`, for exan1ple, 15. This filter passeC.
: - ~8 -.,. "
~;:
- PIIN.869lC
10.Z.197$
the moclulated sub-carrier f`requency as well as the stereo pilot itself which, although coinciding with an edge of the filter is still considerably greater than the sub-carrier signal. After having been amplified in an amplifier 126 the two signals are applied to a first in-put 127 of` a dual-function multiplier stage 128.
Firstly, stage 128 operates as a phase detector in a phase locked loop which comprises, in addition, a low-pass filter 129, a voltage-controlled 38 KHz oscil-lator 130 and a divide-by-two divider 131, the latter feeding a 19 KHz square wave back to a second input 132 of the multiplier stage 1280 This phase locked loop locks into the received stereo pilot and consequently supplies, at the output oP the divider 131 an 19 KHz square wave which is synchronized with the received stereo pilot. The low-pass filter 129 used for preventing the phase locked loop being influenced by other signal co~nponents than - the stereo pilot may have a cut-off frequency of, for example, 300 Hz and a slope above this cut-oPf frequency of 6 db/octave.
Secondly, the multiplier stage 128 operates as a mixer stage for the modulated 16.625 KHz (7/8fp) sub-carr1er. This sub-carrier is mixed with the 19 KHz (fp) square wave at the input 132 which results in a binary phase modulated intermediate frequency signal of ~` 2.37~ KHz (1/8fp), which is passed on via a low-pass filter 133 having a cut-oPf Prequency of, for example, 3 KHz and a hlgh f`requency slope oP 20 db/octave.
,~ Instead of a voltage-controlled 19 KHz oscil-~. :
lator a voltage-con-trolled 38 K~Iz oscillator 130 fol]owed by a div-ide-b~-two divider 131 :is used because, in . ~ . .
, - 29 -, .
P~l~.8691C
.2.1978 general, a divide-by-two divider supplies a more sym-metrical square wave than a voltage-controli.ed oscilla-tor. Consequently, stage 128 is controlle~d by a perfect-ly symmetrical square wave so that input signal compo-nents around the even harmonics of 19 KH~, in particular around 38 KHz, do not influence the output signal. of the stage 128. The detection of signal components around 57 I~Hz by stage 128 is prevented to a sufficient degree by filter 125 which has an adequate attenuation for these signal components. Consequently, by means of the elements 128, 129, i30 and 131 a flltered stereo pilot is avail-able at the output of the divide-by-two divider 131 and a con~erted binary modulated sub-carrier at the output - of stage 128. However, it will. be obvious that these fùnctions can be performed by any other suitable filter and converter arrangement.
The 19 KHz square wave of the divide-by-two divider 131 is divided in a divide-by-eight divider 134 : into a square wave having a frequency of 2.375 KHz (1/8fp). There:fore, a binary phase modulated 2.375 KHz~
; carrier signal is available at the output of filter 133 and an unmodulated 2.375~ KHz square wave, derived from .- the stereo pilot, at the output of the scaler 13~. The ... .
~, modulated carrier signal can now be demodulated syn-chronou.sly by means of the unmodulated wave and be pro-, .
cessed in.accordance with one of the methods descri.bed with reference to the Figures 3 and ~f. The actual detecti~n is effectecl in a synchronous.de-tector 135 to a flrst input :136 whcreo~ ~he modulated signa`l is applied ~; 30 vi.a an amplifier 137, while the unmodulated wave is app:ii.ed to a second inpu-t 138 v:La a controllable phass s ,. .
- 3(~ --, ,~ . , : . . . .
-. :- . . ~ "
P~-IN.~G~JlC
~ 10.2.1~)78 shifter l39. The detecte(l code signals are filterecl ln a low-pass filter 140 having a cut-off frequency of, for example, 350 Hz and a 20 db/octave slope, thereafter converted into square pulses by means of a pulse shaper 141, thereaf~r applied to a decoder 142 which converts the binary transmitter identification signal thus ob-tained into signals suitable for application to a load 143. The load 143 can be different, depending on the in-formation comprised in the code. In case the code com-prises information about the transmitter received and/or the program received, the load ~3 rnay comprise a display displaying this information, so that, for example, the customary tuning sca]e can be dispensed with. Alternatively ` the load 143 may comprise an automatic transmitter-search circuit so that the receiver only tunes to those stations broadcasting a given type of program, for example, classical music. If the code comprises time indication the device 143 can, for example, switch a tape recorder connected to it on or off at a preset time. If the code is a semaphoDe signal the device 143 is formed by a semaphone receiver.
- The phase shifter 139 serves for eliminating all phase errors wh:ich may be produced betwe~en the modu-~ lated 2.375 KHzsignal andthe unmodulated 2.375 KHz square '~ 25 wave. These phase errors can be produced at the divider : 110 in the transmitter and the divider 134 in the re-, ceiver and by delay time differences in the various fil-ters,~ for example in filter 1Z5. For the benefit~of this phase control the binary phase modulated 2.375 K~z "; 30 signal is converted by means oI a squarin~ de-vice 1LIL~
.~ ~ . , .
-;" and a pulse shaper ILl5 into a`4 75 KIIY. squaIe wave. The .', , lC
~ z ~ .2.197~
unlnodulated wave of the phase shifter 139 is converted into a L~.75 KHz square wave by means of a frequcncy mul-tiplier 146. The two ll,75 Kllz square waves are compared in a phase detector 147 which generates a control sig-nal from these square waves which is applied, after fil-tering in a low-pass filter ll~8 and amplification in an amplifier 1~l9, to the control input 150 of the controllable phase shifter 139. Thus, the phase shifter 139 ensures ~- that the 2.37~ KHz square wave and the 2.375 KHz signals e~
applied to the synchronous ~hr~t-~ have the same phase (or are 1800 out of phase respectively). The phase duplicity .
then still occurring can again be offset by the use of a code insensitive thereto. The phase shifter 139 as well as the phase shifter 36 of Figs. 3 and ll can, for example, consist of two cascade-arranged monostable circuits, the time constant of the first circuit being controlled by the control signal and that of the second circuit being equal to half a cycle of the signal to be delayed, the first circuit being started by the incoming signal and the second circuit by the trailing edge of the first circuit. Such a phase shifter is capable of shifting the phase of the signal over nearly 3600, which is more than . . , sufficient since the signal should be shifted over 180. -`` A preferred embodiment of a receiver for re-cèiving signals comprising two binary phase modulated ~,;
sub-carriers on either side of the stereo pilot, the stereo pilot being in quadrature to the resultant of the , ., two sub-carriers may be of the same shape as shown in '~ Fig. 7, it being understood that the filter 125 is not tuned to one sub-carrier but to the stereo pilot, whlle the passband ~idth of the f`:ilter must be sufficiently . , .~
. .
- '12 -.
' . 8~9 1 (~
10 . 2 1 ~78 large to pass the two sub-carriers. On the other hand, the attenuation outside the passband, in particular for signals around 57 KHz, must be sufficiently high to pre-vent disturbances~
In a receiver for receiving signals having a - binary phase modulated sub-carrier on either side of the stereo pilot, the resultant of the two sub-carriers being in phase with the stereo pilot, it is not possible to use the phase detector (128) of the phase locked loop also for down-converting the signal, because the phase detector and the mixer stage must then be controlled with stereo pilots which are shifted 90 relative to one another. Fig. 8 shows a possib]e embodiment of such a receiver in which corresponding elements have been given the same reference numerals as in Fig.- ~.
The output signal of the amplifier 126 is then applied to the phase detector 128 as well as to a second detector 1`51, which functions as a mixer stage. In this mixer stage the input signal is mixed with a 19 KHz square wave which is der1ved by means of frequency division by Q divide-by-two divider 15Z from the 38 KHz oscillator ;~ 130. The two dividers 131 and 152 are s~ energized that `. they supply 19 KHz waves whose phases are sh1fted 90 relative to one another.
.
.
`'; ' ' ' ' ; - 33 -
Claims (18)
PROPERTY OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:
1. A method of radio broadcasting with code signal-ling wherein a multiplex signal frequency-modulated on a main carrier is transmitted at the transmitter side, said multiplex signal comprising: an audio frequency information signal, in the case of stereo transmission, a stereo information signal modulated on a suppressed sub-carrier, and a stereo pilot whose frequency is located between the frequency spectra of the audio frequency information signal and of the modulated stereo information signal and which serves for demodulating the stereo information signal, as well as a binary code signal which is modulated on a further sub-carrier located outside said frequency spectra and having an amplitude which causes the main carrier to deviate by not more than 1 KHz, characterized in that said further sub-carrier is a harmonic of a sub-harmonic of the stereo pilot not coinciding with a har-monic of the stereo pilot, and is derived at the trans-mitter side from the same frequency source as the stereo pilot and in that the code signal is binary phase modu-lated on this sub-carrier.
2. A method as claimed in Claim 1, characterized in that the frequency of said further sub-carrier is located in the middle between two harmonics of the stereo pilot.
3. A method as claimed in Claim 1, characterized in that the further sub-carrier with the modulated code signal is located in at least one of the two halves of the frequency range, divided in two by the stereo pilot, bet-ween the upper limit of the frequency spectrum of the audio frequency information signal and the lower limit of the frequency spectrum of the modulated stereo information signal and in that the modulated code signal has an ampli-tude which causes the main carrier to deviate by less than 1 KHz, preferably by 0.25 KHz.
4. A method as claimed in Claim 3, characterized in that a sub-carrier, binary phase modulated by the code signal is located in each of the two halves of the frequ-ency range divided into two by the stereo pilot.
5. A method as claimed in Claim 4, characterized in that the two sub-carriers, modulated by the code signal, have equal amplitudes and such a phase relative to the stereo pilot that together with the stereo pilot they form a signal produced by quadrature modulation of the stereo pilot by a sub-carrier derived from the stereo pilot, this sub-carrier being binary phase modulated by the code sig-nal.
6. A method as claimed in Claim 3, characterized in that the sub-carrier modulated by the code signal is located from the stereo pilot at a distance equal to 1/8x the frequency of the stereo pilot.
7. A receiver for receiving a radio broadcast according to the method claimed in claim 1 comprising a frequency discriminator for demodulating the received main carrier, characterized by a synchronous demodulator having first and second inputs PHN.8691C
10.2.1978 and an output, a first transmission path coupled to the frequency discriminator for applying the binary phase modulated code signal to the first input of the synchronous demodulator, a second transmission path coupled to the frequency discriminator for applying an unmodulated wave synchronized by the stereo pilot to the second input of the synchronous demodulator and an output circuit coupled to the output of the synchronous demodulator for the demodulated binary code signal.
10.2.1978 and an output, a first transmission path coupled to the frequency discriminator for applying the binary phase modulated code signal to the first input of the synchronous demodulator, a second transmission path coupled to the frequency discriminator for applying an unmodulated wave synchronized by the stereo pilot to the second input of the synchronous demodulator and an output circuit coupled to the output of the synchronous demodulator for the demodulated binary code signal.
8. A receiver as claimed in Claim 7, characterized by a device, included between the two transmission paths, for generating a control signal in dependency on the relative phase between the carrier of the binary phase modulated signal applied to the first input of the syn-chronous demodulator and of the unmodulated wave applied to the second input of the synchronous demodulator and by an electronically controllable phase shifter, included in one of the two transmission paths and controlled by said control signal, for controlling said relative phase.
9. A receiver as claimed in Claim 8, characterized in that the device for generating a control signal com-prises a phase detector with first and second inputs and an output as well as a frequency doubling circuit coupled between a connection point of the first transmission path and the first input of the phase detector, in that a con-necting point of the second transmission path is coupled to the second input of the phase detector and in that the output of the phase detector controls the electronically controllable phase shifter.
10. A receiver as claimed in Claim 8, characterized in that the device for generating a control signal com-PHN.8691C
10.2.1978 prises a phase detector with first and second inputs and an output, the first input being connected via a first connection to a connection point of the first transmission path and the second input via a second connection to a connection point of the second transmission path, the output of the phase detector controlling the electronic-ally controllable phase shifter via a third connection, as well as a phase converter, included in one of said connections and being controlled by the output signal of the synchronous demodulator.
10.2.1978 prises a phase detector with first and second inputs and an output, the first input being connected via a first connection to a connection point of the first transmission path and the second input via a second connection to a connection point of the second transmission path, the output of the phase detector controlling the electronic-ally controllable phase shifter via a third connection, as well as a phase converter, included in one of said connections and being controlled by the output signal of the synchronous demodulator.
11. A receiver as claimed in Claim 9 or 10, characterized in that the electronically controllable phase shifter is included in the second transmission path before said connection point of the second trans-mission path.
12. A receiver as claimed in Claim 7, characterized in that connected to the output of the frequency dis-criminator there is a filter and converter arrangement for outfiltering the stereo pilot and for converting by means of the outfiltered stereo pilot the subcarrier binary phase modulated by the code signal. into an intermediate frequency carrier, binary phase modulated by the code signal and having a frequency equal to the frequency spacing between the sub-carrier and the stereo pilot, in that said first transmission path is connected to an out-put of the filter and converter arrangement for applying the binary phase modulated intermediate frequency carrier to the first input of the synchronous demodulator, and in that an output of the filter and converter arrangement for the outfiltered stereo pilot is connected to the second transmission path, comprising one or more fre-quency dividers, for generating an unmodulated intermediate frequency wave, synchronized by the outfiltered stereo pilot, and for applying this unmodulated intermediate fre-quency wave to the second input of the synchronous demod-ulator.
13. A receiver as claimed in Claim 12, characterized in that said filter and converter arrangement comprises a phase locked loop implemented with a voltage-controlled oscillator, a filter and a phase detector, for filtering out the stereo pilot, the output signal of the frequency discriminator being applied to a first input of the phase detector and the filtered-out stereo pilot derived from the voltage-controlled oscillator to a second input of the phase detector and in that the phase detector of the phase locked loop also functions as a converter for the modulated sub-carrier because said first transmission path is con-nected to the output of the phase detector.
14. A receiver as claimed in Claim 12, characterized in that a bandpass filter tuned to the sub-carrier mod-ulated by the code signal is included between the output of the frequency discriminator and the input of the filter and converter arrangement, a slope of this bandpass filter passing the stereo pilot.
15. A transmitter for transmitting a radio broadcast in accordance with the method as claimed in claim 1, com-prising a device for generating an audio frequency informa-tion signal, in the case of a stereo transmitter a stereo information signal modulated on a suppressed stereo sub-carrier, as well as an oscillator for generating a stereo pilot whose frequency is located between the frequency spectra of the audio frequency in-PHN.8691C
10.2.1978 formation signal and of the modulated stereo information signal, characterized by a source of binary code signals and a modulating signal generator, connected to this source and to the said oscillator, for generating a sub-carrier which is binary phase modulated by the code signal this sub-carrier being a harmonic of a sub-harmonic of the stereo pilot not coinciding with the harmonic of this pilot and being located outside said frequency spectra.
10.2.1978 formation signal and of the modulated stereo information signal, characterized by a source of binary code signals and a modulating signal generator, connected to this source and to the said oscillator, for generating a sub-carrier which is binary phase modulated by the code signal this sub-carrier being a harmonic of a sub-harmonic of the stereo pilot not coinciding with the harmonic of this pilot and being located outside said frequency spectra.
16. A transmitter as claimed in Claim 15, character-ized in that the modulating signal generator generates a sub-carrier which is binary phase modulated by the code signal and is located in at least one of the two halves of the frequency range, divided into two by the stereo pilot, between the upper limit of the frequency spectrum of the audio frequency information signal and the lower limit of the frequency spectrum of the modulated stereo information signal.
17. A transmitter as claimed in Claim 15, character-ized in that the modulating signal generator comprises a sub-carrier generator for generating said sub-carrier as well as a modulator, connected to the sub-carrier generator and the source of binary code signals, for binary phase modulating the sub-carrier by the code signals.
18. A transmitter as claimed in Claim 15, character-ized in that the modulating signal generator comprises a frequency generator connected to the oscillator, for generating an intermediate frequency carrier having a frequency equal to the frequency spacing between the sub-carrier and the stereo pilot, as well as first and second modulators each comprising first and second inputs PHN.8691C
10.2.1978 and an output, the output of the first modulator being connected to the first input of the second modulator, of the two inputs of the first modulator and the second input of the second modulator one input being connected to the oscillator, a second input to the frequency gene-rator and a third input to the binary code signal source.
10.2.1978 and an output, the output of the first modulator being connected to the first input of the second modulator, of the two inputs of the first modulator and the second input of the second modulator one input being connected to the oscillator, a second input to the frequency gene-rator and a third input to the binary code signal source.
Applications Claiming Priority (4)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| NL7702019A NL7702019A (en) | 1977-02-25 | 1977-02-25 | RADIO BROADCASTING SYSTEM WITH TRANSMITTER CHARACTERIZATION. |
| NL7702019 | 1977-02-25 | ||
| NL7709619A NL7709619A (en) | 1977-09-01 | 1977-09-01 | RADIO BROADCASTING SYSTEM WITH CODE SIGNALING. |
| NL7709619 | 1977-09-01 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1116241A true CA1116241A (en) | 1982-01-12 |
Family
ID=26645294
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA297,759A Expired CA1116241A (en) | 1977-02-25 | 1978-02-27 | Radio broadcasting system with code signaling |
Country Status (19)
| Country | Link |
|---|---|
| JP (1) | JPS6033014B2 (en) |
| AR (1) | AR216939A1 (en) |
| AT (1) | AT375511B (en) |
| BE (1) | BE864272A (en) |
| BR (1) | BR7801160A (en) |
| CA (1) | CA1116241A (en) |
| CH (1) | CH627597A5 (en) |
| DE (1) | DE2807706C2 (en) |
| DK (1) | DK79978A (en) |
| ES (1) | ES467246A1 (en) |
| FI (1) | FI780597A (en) |
| FR (1) | FR2382135A1 (en) |
| GB (1) | GB1579985A (en) |
| HK (1) | HK22881A (en) |
| IT (1) | IT1156903B (en) |
| MX (1) | MX147105A (en) |
| NL (1) | NL7800581A (en) |
| NZ (1) | NZ186531A (en) |
| SE (1) | SE429704B (en) |
Families Citing this family (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| NL8000607A (en) * | 1980-01-31 | 1981-09-01 | Philips Nv | FM RECEIVER WITH TRANSMITTER CHARACTERIZATION. |
| US4388493A (en) * | 1980-11-28 | 1983-06-14 | Maisel Douglas A | In-band signaling system for FM transmission systems |
| NL8100419A (en) * | 1981-01-29 | 1982-08-16 | Philips Nv | FM BROADCASTING SYSTEM WITH TRANSMITTER CHARACTERIZATION. |
| NL8200560A (en) * | 1982-02-15 | 1983-09-01 | Philips Nv | SYSTEM FOR COMMUNICATION BY RE-MESSAGES TRANSMITTED MESSAGES AND STATIONS FOR USE IN SUCH A SYSTEM. |
| DE3536820A1 (en) * | 1985-10-16 | 1987-04-16 | Bosch Gmbh Robert | Traffic program decoder |
| JPH07114390B2 (en) * | 1986-10-29 | 1995-12-06 | 日本放送協会 | Subcarrier reproduction system |
Family Cites Families (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| FR1529069A (en) * | 1966-06-18 | 1968-06-14 | Philips Nv | A radio transmission system for stereophonic signals, as well as transmitters and receivers to be used in this system |
| DE2051034C3 (en) * | 1970-10-17 | 1978-11-02 | Hessischer Rundfunk, 6000 Frankfurt | VHF radio stereophonic transmission system |
-
1978
- 1978-01-18 NL NL7800581A patent/NL7800581A/en not_active Application Discontinuation
- 1978-02-22 GB GB7038/78A patent/GB1579985A/en not_active Expired
- 1978-02-22 NZ NZ186531A patent/NZ186531A/en unknown
- 1978-02-22 FI FI780597A patent/FI780597A/en not_active Application Discontinuation
- 1978-02-22 SE SE7802031A patent/SE429704B/en unknown
- 1978-02-22 DK DK79978A patent/DK79978A/en not_active Application Discontinuation
- 1978-02-22 CH CH194878A patent/CH627597A5/en not_active IP Right Cessation
- 1978-02-22 MX MX172521A patent/MX147105A/en unknown
- 1978-02-23 IT IT67376/78A patent/IT1156903B/en active
- 1978-02-23 ES ES467246A patent/ES467246A1/en not_active Expired
- 1978-02-23 DE DE2807706A patent/DE2807706C2/en not_active Expired
- 1978-02-23 BE BE185440A patent/BE864272A/en not_active IP Right Cessation
- 1978-02-23 AR AR271200A patent/AR216939A1/en active
- 1978-02-24 BR BR7801160A patent/BR7801160A/en unknown
- 1978-02-24 FR FR7805364A patent/FR2382135A1/en not_active Withdrawn
- 1978-02-24 AT AT0134578A patent/AT375511B/en not_active IP Right Cessation
- 1978-02-25 JP JP53020419A patent/JPS6033014B2/en not_active Expired
- 1978-02-27 CA CA297,759A patent/CA1116241A/en not_active Expired
-
1981
- 1981-05-28 HK HK228/81A patent/HK22881A/en unknown
Also Published As
| Publication number | Publication date |
|---|---|
| NL7800581A (en) | 1978-08-29 |
| IT7867376A0 (en) | 1978-02-23 |
| FI780597A (en) | 1978-08-26 |
| DK79978A (en) | 1978-08-26 |
| MX147105A (en) | 1982-10-06 |
| ES467246A1 (en) | 1978-11-16 |
| BR7801160A (en) | 1978-12-05 |
| CH627597A5 (en) | 1982-01-15 |
| GB1579985A (en) | 1980-11-26 |
| SE429704B (en) | 1983-09-19 |
| DE2807706A1 (en) | 1978-08-31 |
| ATA134578A (en) | 1983-12-15 |
| BE864272A (en) | 1978-08-23 |
| AR216939A1 (en) | 1980-02-15 |
| FR2382135A1 (en) | 1978-09-22 |
| DE2807706C2 (en) | 1985-02-14 |
| SE7802031L (en) | 1978-08-26 |
| NZ186531A (en) | 1981-12-15 |
| JPS53114301A (en) | 1978-10-05 |
| JPS6033014B2 (en) | 1985-07-31 |
| AT375511B (en) | 1984-08-10 |
| IT1156903B (en) | 1987-02-04 |
| HK22881A (en) | 1981-06-05 |
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