WO2024095330A1

WO2024095330A1 - Motor control device

Info

Publication number: WO2024095330A1
Application number: PCT/JP2022/040762
Authority: WO
Inventors: 健太郎松尾; 峻谷口; 俊幸安島
Original assignee: 日立Astemo株式会社
Priority date: 2022-10-31
Filing date: 2022-10-31
Publication date: 2024-05-10

Abstract

This motor control device comprises: a current detection unit that acquires a current detection value corresponding to current flowing through an AC motor; a current control unit that acquires a current command value for the AC motor, and that calculates a voltage command value for the AC motor, at each prescribed calculation period, on the basis of the current command value and the current detection value; and a voltage control unit that executes, at each calculation period, control calculation based on the voltage command value, and obtains a duty command for generating a gate signal. The current detection unit acquires the current detection value at least twice in each calculation period. The voltage control unit obtains at least two types of duty command in each calculation period and reflects the same on the gate signal.

Description

Motor Control Device

The present invention relates to a device for controlling a motor.

Traditionally, motors used for driving automobiles and other applications are required to be compact, have high output, and are highly efficient. To achieve these characteristics, in recent years there has been an increasing use of high-speed motors with faster rotation speeds than conventional motors.

When a three-phase AC synchronous motor driven by an inverter is rotated at high speed, the motor's electrical angular frequency becomes higher than when the motor is normally driven, and the time per electrical angular period becomes shorter. As a result, in a motor control device that detects the current flowing through the motor and controls the drive of the inverter's switching elements based on the current detection value to control the motor, the number of times that the current detection value can be reflected in motor control during one electrical angular period decreases. This reduction in the number of times that the current detection value is reflected in motor control at high speeds leads to a decrease in motor control performance and may cause errors, pulsation, overcurrent, etc. in the torque generated by the motor.

In order to increase the number of times that current detection values are reflected in motor control at high speeds, it is sufficient to speed up the motor control calculations performed by the motor control device and shorten the calculation cycle. However, the time required for motor control calculations is determined by the processing power of the microcomputer that executes the calculations, and improving the processing power of the microcomputer leads to increased manufacturing costs and power consumption. For this reason, there is a limit to how fast motor control calculations can be made.

For example, the technology of Patent Document 1 is known regarding motor control at high speeds. Patent Document 1 describes a control device for an electric motor drive device that, when synchronous control mode is selected, performs a current detection process IS for detecting the current flowing through the coil of an AC motor at every reference calculation period T0 set to 1/2 the carrier period, performs a voltage control process VC at a period N times (N is an integer equal to or greater than 2) the reference calculation period T0, and feeds back N current detection values detected by N current detection processes IS to perform one voltage control process VC. This keeps the overall calculation load within the processing capacity of the calculation processing unit, while appropriately reflecting current detection values detected at short periods to suppress the occurrence of aliasing in current sampling.

Japanese Patent Publication No. 2011-83068

The technology in Patent Document 1 makes it possible to speed up current detection processing, but the cycle of the voltage control processing is the same as the carrier cycle. Therefore, the number of times that the current detection value can be reflected in motor control during one electrical angle cycle remains unchanged, and it is not possible to increase the number of times that the current detection value is reflected in motor control during high speed rotation.

The main objective of the present invention is to solve the above problems by improving motor control performance while suppressing increases in processing load during motor control at high speeds.

The motor control device according to the present invention generates a gate signal for controlling the operation of a switching element of an inverter having a plurality of switching elements, and outputs the generated gate signal to the inverter to control the drive of an AC motor connected to the inverter. The motor control device includes a current detection unit that obtains a current detection value corresponding to the current flowing through the AC motor, a current control unit that obtains a current command value for the AC motor and calculates a voltage command value for the AC motor for each predetermined calculation period based on the current command value and the current detection value, and a voltage control unit that executes a control calculation based on the voltage command value for each calculation period and determines a duty command for generating the gate signal, the current detection unit obtains the current detection value at least twice for each calculation period, and the voltage control unit determines at least two or more types of duty commands for each calculation period and reflects them in the gate signal.

According to the present invention, it is possible to improve motor control performance while suppressing increases in processing load during motor control at high speeds.

1 is a diagram showing the overall configuration of a motor drive system including a motor control device according to an embodiment of the present invention; 1 is a block diagram showing a functional configuration of a motor control device according to an embodiment of the present invention; FIG. 2 is a block diagram of a current converter according to an embodiment of the present invention. FIG. 2 is a block diagram of a voltage control unit according to an embodiment of the present invention. 11A and 11B are diagrams comparing dead time compensation in a conventional example and the present invention; FIG. 11 is a comparison diagram of calculation processing between the conventional example and the present invention.

Below, a detailed description of an embodiment of the present invention will be given with reference to the drawings. In this embodiment, an example of application to a motor drive system mounted on and used in an electrically powered vehicle such as an electric vehicle or a hybrid vehicle will be described.

Figure 1 is an overall configuration diagram of a motor drive system equipped with a motor control device according to one embodiment of the present invention. In Figure 1, the motor drive system 100 includes a motor control device 1, an AC motor (hereinafter simply referred to as "motor") 2, an inverter 3, a rotational position detector 4, and a high-voltage battery 5.

The motor control device 1 controls the operation of the inverter 3 based on a torque command T* corresponding to a target torque required of the motor 2 by the vehicle, thereby generating a gate signal for controlling the drive of the motor 2. The generated gate signal is then output to the inverter 3. Details of the motor control device 1 will be explained later.

The inverter 3 has an inverter circuit 31, a gate drive circuit 32, and a smoothing capacitor 33. The gate drive circuit 32 generates a gate drive signal for controlling each switching element of the inverter circuit 31 based on a gate signal input from the motor control device 1, and outputs the gate drive signal to the inverter circuit 31. The inverter circuit 31 has switching elements corresponding to the upper and lower arms of the U, V, and W phases, respectively. By controlling each of these switching elements according to the gate drive signal input from the gate drive circuit 32, the DC power supplied from the high-voltage battery 5 is converted into AC power and output to the motor 2. The smoothing capacitor 33 smoothes the DC power supplied from the high-voltage battery 5 to the inverter circuit 31.

Motor 2 is a synchronous motor that is driven to rotate by AC power supplied from inverter 3, and has a stator and a rotor. When AC power input from inverter 3 is applied to armature coils Lu, Lv, and Lw provided in the stator, three-phase AC currents Iu, Iv, and Iw are conducted in motor 2, and armature magnetic flux is generated in each armature coil. Attractive and repulsive forces are generated between the armature magnetic flux of each armature coil and the magnetic flux of the permanent magnets arranged in the rotor, generating torque in the rotor and driving the rotor to rotate.

A rotational position sensor 8 is attached to the motor 2 to detect the rotational position θ of the rotor. The rotational position detector 4 calculates the rotational position θ from the input signal of the rotational position sensor 8. The calculation result of the rotational position θ by the rotational position detector 4 is input to the motor control device 1, and is used in the phase control of the AC power that the motor control device 1 performs in accordance with the phase of the induced voltage of the motor 2.

Here, a resolver consisting of an iron core and windings is more suitable for the rotational position sensor 8, but a sensor using a magnetic resistance element such as a GMR sensor or a Hall element is also acceptable. Also, the rotational position detector 4 may estimate the rotational position θ using the three-phase AC currents Iu, Iv, Iw flowing through the motor 2 and the three-phase AC voltages Vu, Vv, Vw applied to the motor 2 from the inverter 3, without using an input signal from the rotational position sensor 8.

A current sensor 7 is disposed between the inverter 3 and the motor 2. The current sensor 7 detects the three-phase AC currents Iu, Iv, Iw (U-phase AC current Iu, V-phase AC current Iv, and W-phase AC current Iw) passing through the motor 2. The current sensor 7 is configured using, for example, a Hall current sensor. The detection results of the three-phase AC currents Iu, Iv, Iw by the current sensor 7 are input to the motor control device 1 and are used for the phase control of AC power performed by the motor control device 1, similar to the calculation result of the rotational position θ by the rotational position detector 4. Note that while FIG. 2 shows an example in which the current sensor 7 is configured with three current detectors, it is also possible to use two current detectors and calculate the AC current of the remaining one phase because the sum of the three-phase AC currents Iu, Iv, Iw is zero. In addition, the pulsed DC current flowing from the high-voltage battery 5 to the inverter 3 may be detected by a shunt resistor or the like inserted between the smoothing capacitor 33 and the inverter 3, and the three-phase AC currents Iu, Iv, and Iw may be calculated based on this DC current and the three-phase AC voltages Vu, Vv, and Vw applied from the inverter 3 to the motor 2.

Next, the motor control device 1 will be described in detail. FIG. 2 is a block diagram showing the functional configuration of the motor control device 1 according to one embodiment of the present invention. In FIG. 2, the motor control device 1 has the following functional blocks: a current command generation unit 10, a speed calculation unit 11, a current detection unit 12, a current conversion unit 13, a current control unit 14, a carrier frequency control unit 15, a voltage control unit 16, and a gate signal generation unit 17. The motor control device 1 is configured, for example, by a microcomputer, and these functional blocks can be realized by executing a predetermined program in the microcomputer. Alternatively, some or all of these functional blocks may be realized using hardware circuits such as logic ICs or FPGAs.

The current command generating unit 10 calculates the d-axis current command Id* and the q-axis current command Iq* based on the input torque command T* and the voltage Vdc of the high-voltage battery 5. Here, the d-axis current command Id* and the q-axis current command Iq* corresponding to the torque command T* are calculated using, for example, a preset current command map or formula.

The speed calculation unit 11 calculates the motor rotation speed ω, which represents the rotation speed (revolutions) of the motor 2, from the change over time in the rotation position θ. Note that the motor rotation speed ω may be a value expressed as either an angular velocity (rad/s) or a rotation speed (rpm). These values may also be converted into each other for use.

The current detection unit 12 samples the detection signals of the three-phase AC currents Iu, Iv, Iw input from the current sensor 7 at predetermined sampling periods, and obtains the first three-phase AC current detection values Iua, Iva, Iwa (the first U-phase AC current detection value Iua, the first V-phase AC current detection value Iva, and the first W-phase AC current detection value Iwa) and the second three-phase AC current detection values Iub, Ivb, Iwb (the second U-phase AC current detection value Iub, the second V-phase AC current detection value Ivb, and the second W-phase AC current detection value Iwb) at different timings. Here, the current detection unit 12 samples the three-phase AC currents Iu, Iv, and Iw in a sampling period that is half the calculation period of the current control calculation performed by the current control unit 14, and stores the sampling results in the microcomputer register as the first three-phase AC current detection values Iua, Iva, and Iwa and the second three-phase AC current detection values Iub, Ivb, and Iwb. Here, the first three-phase AC current detection values Iua, Iva, and Iwa are currents detected half a cycle before the current control calculation task is started. The second three-phase AC current detection values Iub, Ivb, and Iwb are currents detected when the current control calculation task is started. In addition, in order to prevent detection errors due to delays in the current sensor and circuit, the timing of current detection is delayed by about several μs from the start of the current control calculation task and half a cycle before. That is, the current detection unit 12 obtains the current detection values twice for each calculation cycle of the current control calculation, and outputs the current detection values obtained the first time as the first three-phase AC current detection values Iua, Iva, Iwa, and the current detection values obtained the second time as the second three-phase AC current detection values Iub, Ivb, Iwb to the current conversion unit 13 for each calculation cycle.

The current conversion unit 13 performs dq conversion based on the rotational position θ and rotational speed ω on the first three-phase AC current detection values Iua, Iva, Iwa and the second three-phase AC current detection values Iub, Ivb, Iwb acquired by the current detection unit 12, and calculates the d-axis current value Id and the q-axis current value Iq based on the results of these dq conversions. At this time, the current conversion unit 13 calculates the average current values of the d-axis and q-axis based on the first d-axis current value Id1 and the first q-axis current value Iq1 obtained by dq conversion of the first three-phase AC current detection values Iua, Iva, Iwa, and the second d-axis current value Id2 and the second q-axis current value Iq2 obtained by dq conversion of the second three-phase AC current detection values Iub, Ivb, Iwb. Then, the current conversion unit 13 outputs the calculated average current values as the d-axis current value Id and the q-axis current value Iq. The method for calculating the d-axis current value Id and the q-axis current value Iq by the current converter 13 will be described in detail later.

The current control unit 14 calculates a d-axis voltage command Vd* and a q-axis voltage command Vq* according to the torque command T* based on the deviation between the d-axis current command Id* and the q-axis current command Iq* output from the current command generation unit 10 and the d-axis current value Id and the q-axis current value Iq output from the current conversion unit 13, so that these values match. Here, taking into account the calculation delay between the detection of the rotational position θ and the three-phase AC currents Iu, Iv, Iw and the output of a gate signal by the motor control device 1 to the inverter 3, the d-axis voltage command Vd* according to the deviation between the d-axis current command Id* and the d-axis current value Id, and the q-axis voltage command Vq* according to the deviation between the q-axis current command Iq* and the q-axis current value Iq are calculated at each predetermined calculation cycle by current control calculation using a control method such as PI control.

The carrier frequency control unit 15 calculates a carrier frequency fc representing the frequency of the carrier wave used to generate the PWM pulse signal based on the rotational position θ determined by the rotational position detector 4 and the rotational speed ω determined by the speed calculation unit 11. For example, the carrier frequency fc is calculated so that the number of carrier waves per rotation of the motor 2 matches a predetermined number Nc.

The voltage control unit 16 uses the rotational position θ, motor rotational speed ω, voltage Vdc of the high-voltage battery 5, and carrier frequency fc to determine the duty command values Dua, Dub, Dva, Dvb, Dwa, and Dwb corresponding to the first and second half of the carrier for each phase from the d-axis voltage command Vd* and q-axis voltage command Vq* calculated by the current control unit 14. At this time, the voltage control unit 16 performs a three-phase conversion on the d-axis voltage command Vd* and q-axis voltage command Vq* to determine the three-phase voltage commands Vu*, Vv*, and Vw* (U-phase voltage command value Vu*, V-phase voltage command value Vv*, and W-phase voltage command value Vw*). Then, PWM calculation is performed based on the obtained three-phase voltage commands Vu*, Vv*, Vw* and a triangular carrier wave that changes periodically with a carrier frequency fc, and the pulse width of the gate drive signal to be output to each switching element is determined to obtain the duty command values Dua, Dub, Dva, Dvb, Dwa, and Dwb. Details of the method of calculating the duty command values Dua, Dub, Dva, Dvb, Dwa, and Dwb by the voltage control unit 16 will be described later.

The gate signal generating unit 17 generates gate signals (PWM pulse signals) Gup, Gun, Gvp, Gvn, Gwp, and Gwn for controlling the operation of the inverter 3 based on the duty command values Dua, Dub, Dva, Dvb, Dwa, and Dwb obtained by the voltage control unit 16. For example, the gate signals Gup, Gun, Gvp, Gvn, Gwp, and Gwn are generated by converting the duty command values Dua, Dub, Dva, Dvb, Dwa, and Dwb into predetermined voltage signals. The gate signals generated by the gate signal generating unit 17 are output from the motor control device 1 to the gate drive circuit 32 of the inverter 3, and are converted into gate drive signals by the gate drive circuit 32. As a result, each switching element of the inverter circuit 31 is controlled to be turned on and off, and the output voltage of the inverter 3 is adjusted.

Next, the current conversion unit 13 in the motor control device 1 will be described in detail. FIG. 3 is a block diagram of the current conversion unit 13 according to one embodiment of the present invention. The current conversion unit 13 has the following functional blocks: a first current conversion unit 131, a second current conversion unit 132,

adders

133 and 134, and

multipliers

135 and 136.

Here, the current input to the current conversion unit 13 will be described. As described above, the current detection unit 12 detects the first three-phase AC current detection values Iua, Iva, Iwa and the second three-phase AC current detection values Iub, Ivb, Iwb in a cycle that is half the calculation cycle of the current control calculation performed by the current control unit 14, and stores them in a microcomputer register. The first three-phase AC current detection values Iua, Iva, Iwa are currents detected half a cycle before the current control calculation task is started. The second three-phase AC current detection values Iub, Ivb, Iwb are currents detected when the current control calculation task is started. The first three-phase AC current detection values Iua, Iva, Iwa and the second three-phase AC current detection values Iub, Ivb, Iwb stored in the microcomputer register for each control cycle are simultaneously input to the current conversion unit 13. Since the current control calculation is performed once per control calculation cycle, the current conversion unit 13 averages the current detected twice in the control calculation cycle and outputs it as two samples of data to the current control unit 14. The current is averaged by performing dq transformation on each of the first three-phase AC current detection values Iua, Iva, Iwa and the second three-phase AC current detection values Iub, Ivb, Iwb, and averaging them on the dq axis.

The first current converter 131 performs dq conversion on the first three-phase AC current detection values Iua, Iva, Iwa input from the current detector 12 based on the rotational position θ and the rotational speed ω. The first three-phase AC current detection values Iua, Iva, Iwa are currents detected half a cycle before the start of the current control calculation, and the rotational position θ and the rotational speed ω are the rotational position and the rotational speed detected at the start of the current control calculation. Therefore, the timing at which the current is detected is different from the timing at which the rotational position θ is detected. Therefore, it is necessary to calculate the rotational angle advanced from the current detection to the rotational position detection, estimate the rotational angle at the current detection timing, and perform the dq conversion. The phase _θa used in the dq conversion is calculated as shown in Equation (1).
θ _a = θ + (Ts [n-1] / 2) × ω ... (1)
In equation (1), Ts[n-1] is the control period at the time of the previous interrupt. If the change in the rotational angular velocity ω is slow and does not change during the control period, the rotation angle at the current detection timing can be calculated using equation (1). The obtained dq transformation result is then output as the first d-axis current value Id1 and the first q-axis current value Iq1.

The second current conversion unit 132 performs dq conversion on the second three-phase AC current detection values Iub, Ivb, Iwb input from the current detection unit 12 based on the rotational position θ and the rotational speed ω. The obtained dq conversion result is then output as the second d-axis current value Id2 and the second q-axis current value Iq2. Because the detection timing of the second three-phase AC current detection values Iub, Ivb, Iwb and the rotational position θ is the same, there is no need to correct the phase of the dq conversion as in the first current conversion unit.

The adder 133 sums the first d-axis current value Id1 output from the first current converter 131 and the second d-axis current value Id2 output from the second current converter 132. The multiplier 135 multiplies this sum by 0.5 and outputs the result as the d-axis current value Id. In this way, the d-axis current value Id is calculated from the average value of the first d-axis current value Id1 and the second d-axis current value Id2.

The adder 134 sums the first q-axis current value Iq1 output from the first current converter 131 and the second q-axis current value Iq2 output from the second current converter 132. The multiplier 136 multiplies this sum by 0.5 and outputs it as the q-axis current value Iq. In this way, the q-axis current value Iq is calculated from the average value of the first q-axis current value Iq1 and the second q-axis current value Iq2.

As described above, the current conversion unit 13 performs dq conversion on the first three-phase AC current detection values Iua, Iva, Iwa and the second three-phase AC current detection values Iub, Ivb, Iwb to obtain the d-axis current value Id and the q-axis current value Iq. This method makes it possible to double the current detection period without changing the current control calculation period.

Next, the voltage control unit 16 in the motor control device 1 will be described in detail. FIG. 4 is a block diagram of the voltage control unit 16 according to one embodiment of the present invention. The voltage control unit 16 has the following functional blocks: a first voltage conversion unit 161, a second voltage conversion unit 162, overmodulation

gain multiplication units

163a and 163b, zero-

phase addition units

164a and 164b,

duty conversion units

165a and 165b, dead-time

compensation addition units

166a and 166b, and a duty output unit 167.

The first voltage conversion unit 161 performs three-phase conversion on the d-axis voltage command Vd* and q-axis voltage command Vq* input from the current control unit 14 based on the rotational position θ and the rotational speed ω, and obtains three-phase voltage command values to be used in the PWM calculation. At this time, the first voltage conversion unit 161 obtains a three-phase voltage command value corresponding to the rotational position θ at a time point delayed by 1.25 control cycles from the time point at which the rotational position θ and the first three-phase AC current detection values Iua, Iva, Iwa were acquired, as the three-phase voltage command value for the first half of the next control cycle. The obtained three-phase conversion result is then output as the first three-phase voltage command values Vua*, Vva*, Vwa* (first U-phase voltage command value Vua*, first V-phase voltage command value Vva*, and first W-phase voltage command value Vwa*) corresponding to the first half of the carrier wave.

The second voltage conversion unit 162 performs three-phase conversion on the d-axis voltage command Vd* and q-axis voltage command Vq* input from the current control unit 14 based on the rotational position θ and the rotational speed ω, and obtains three-phase voltage command values to be used in the PWM calculation. At this time, the second voltage conversion unit 162 obtains, as the three-phase voltage command values for the latter half of the next control period, three-phase voltage command values corresponding to the rotational position θ at a time point delayed by 1.75 control periods from the time point at which the rotational position θ and the first three-phase AC current detection values Iua, Iva, Iwa were acquired. The obtained three-phase conversion results are then output as second three-phase voltage command values Vub*, Vvb*, Vwb* (second U-phase voltage command value Vub*, second V-phase voltage command value Vvb*, and second W-phase voltage command value Vwb*) corresponding to the latter half of the carrier wave.

The overmodulation

gain multiplication units

163a and 163b multiply the first three-phase voltage command values Vua*, Vva*, Vwa* output from the first voltage conversion unit 161 and the second three-phase voltage command values Vub*, Vvb*, Vwb* output from the second voltage conversion unit 162 by a predetermined overmodulation gain, based on the voltage Vdc of the high-voltage battery 5. The overmodulation gain is a gain used when performing overmodulation control to increase the output voltage of the inverter 3 above the amplitude of the fundamental wave component, and is set to a value of 1 or more, for example. If overmodulation control is not performed, the overmodulation gain can be set to 1.

The zero-

phase adders

164a and 164b add a predetermined zero-phase voltage value to the first three-phase voltage command values Vua*, Vva*, Vwa* and the second three-phase voltage command values Vub*, Vvb*, Vwb* multiplied by the overmodulation gain by the

overmodulation gain multipliers

163a and 163b. The zero-phase voltage value is a process for improving the AC voltage output relative to the voltage Vdc of the high-voltage battery 5 (improving the voltage utilization rate), and is determined, for example, according to a combination of the first three-phase voltage command values Vua*, Vva*, Vwa* or the second three-phase voltage command values Vub*, Vvb*, Vwb*, and the rotational position θ. Note that the addition of the zero-phase voltage value is not necessarily required, so the zero-

phase adders

164a and 164b may be omitted in the voltage control unit 16.

The

duty conversion units

165a and 165b convert the first three-phase voltage command values Vua*, Vva*, Vwa* and the second three-phase voltage command values Vub*, Vvb*, Vwb*, which are multiplied by the zero-phase voltage value by the zero-

phase addition units

164a and 164b, into duty values for the U, V, and W phases by performing PWM calculations using a carrier wave that changes periodically at the carrier frequency fc. As described above, the first three-phase voltage command values Vua*, Vva*, Vwa* are generated corresponding to the first half of the carrier wave, and the second three-phase voltage command values Vub*, Vvb*, Vwb* are generated corresponding to the second half of the carrier wave. Therefore, the duty conversion unit 165a calculates the duty value of each phase corresponding to the first half of the carrier wave based on the first three-phase voltage command values Vua*, Vva*, Vwa*. Meanwhile, the duty conversion unit 165b calculates the duty values of each phase corresponding to the latter half of the carrier wave based on the second three-phase voltage command values Vub*, Vvb*, and Vwb*.

The dead time

compensation addition units

166a and 166b perform dead time compensation by adding a predetermined dead time compensation value Udt, Vdt, or Wdt for each phase to the duty value of each phase calculated by the

duty conversion units

165a and 165b, respectively. The dead time compensation value is a value that is added to the duty value of each phase as necessary to prevent the switching elements of the upper arm and lower arm of each phase from being turned on simultaneously. Details of the dead time compensation by the dead time

compensation addition units

166a and 166b will be described later.

The duty output unit 167 outputs duty command values Dua, Dub, Dva, Dvb, Dwa, and Dwb based on the duty values of each phase for which dead time compensation has been performed by the dead

time compensation adders

166a and 166b. At this time, the duty output unit 167 outputs the dead time compensated duty values Dua, Dva, and Dwa corresponding to the first half of the carrier wave, which are output from the dead time compensation adder 166a based on the first three-phase voltage command values Vua*, Vva*, and Vwa*, and the dead time compensated duty values Dub, Dvb, and Dwb corresponding to the second half of the carrier wave, which are output from the dead time compensation adder 166b based on the second three-phase voltage command values Vub*, Vvb*, and Vwb*.

As described above, the voltage control unit 16 can obtain the duty command values Dua, Dub, Dva, Dvb, Dwa, and Dwb corresponding to the first and second half of the carrier for each phase from the d-axis voltage command Vd* and the q-axis voltage command Vq*.

Next, the details of the dead time compensation by the dead time

compensation adding units

166a and 166b will be described with reference to FIG. 5. FIG. 5 is a comparison diagram of the dead time compensation in the conventional example and the present invention.

FIG. 5(a) shows the state of dead time compensation in a conventional example to which the present invention is not applied. When the current flowing through the motor 2 is positive, for example, a target pulse shown on the left side of FIG. 5(a) is input as the duty value before dead time compensation for one of the phases. In this case, by adding a predetermined compensation duty dt equivalent to the dead time compensation value before and after the target pulse, the positions of the falling edge and rising edge of the target pulse move, respectively, and a pulse waveform after dead time compensation as shown in FIG. 5(a) is generated. Based on the pulse waveform after dead time compensation generated in this way, pulse waveforms for the upper arm and lower arm are generated with a dead time difference between them, and an output pulse corresponding to the original target pulse is generated.

Also, when the current flowing through the motor 2 is negative, for example, a target pulse as shown on the right side of Figure 5(a) is input as the duty value before dead time compensation for one of the phases. In this case, by subtracting a predetermined compensation duty dt equivalent to the dead time compensation value from the beginning and end of the target pulse, the positions of the falling edge and rising edge of the target pulse are moved, respectively, and a pulse waveform after dead time compensation as shown in Figure 5(a) is generated. Based on the pulse waveform after dead time compensation generated in this way, pulse waveforms for the upper arm and lower arm are generated with a dead time difference between them, and an output pulse corresponding to the original target pulse is generated.

It can be seen that, regardless of whether the current is positive or negative, the output pulse of the conventional example generated as described above has a delay of the compensation duty dt compared to the original target pulse before dead time compensation.

FIG. 5(b) shows the state of dead-time compensation in an embodiment to which the present invention is applied. When the current flowing through the motor 2 is positive, for example, the target pulse shown on the left side of FIG. 5(b) is input to the dead-

time compensation adders

166a and 166b as the duty value before dead-time compensation for either phase. In this case, the dead-time compensation adder 166a does not perform dead-time compensation, and the dead-time compensation adder 166b adds a value obtained by multiplying the compensation duty dt by two to the target pulse, thereby moving the position of the rising edge of the target pulse and generating a pulse waveform after dead-time compensation as shown in FIG. 5(b). Based on the pulse waveform after dead-time compensation thus generated, the pulse waveforms of the upper arm and the lower arm are generated with a dead-time difference between them, and an output pulse corresponding to the original target pulse is generated.

Also, when the current flowing through the motor 2 is negative, for example, the target pulse shown on the right side of FIG. 5(b) is input to the dead

time compensation adders

166a and 166b as the duty value before dead time compensation for either phase. In this case, the dead time compensation adder 166b does not perform dead time compensation, and the dead time compensation adder 166a subtracts a value obtained by multiplying the compensation duty dt described above by two from the target pulse, thereby moving the position of the falling edge of the target pulse and generating a pulse waveform after dead time compensation as shown in FIG. 5(b). Based on the pulse waveform after dead time compensation generated in this way, pulse waveforms for the upper arm and lower arm are generated with a dead time difference between them, and an output pulse corresponding to the original target pulse is generated.

It can be seen that, regardless of whether the current is positive or negative, the output pulse of the present invention generated as described above does not have a delay as in the conventional example, compared to the original target pulse before dead time compensation.

Figure 6 is a comparison diagram of the calculation processing in the conventional example and the present invention.

FIG. 6(a) shows the state of the calculation processing in a conventional example to which the present invention is not applied. In the conventional example, a series of calculation processing from current detection to PWM duty command is repeatedly performed within each period of a triangular carrier wave that changes periodically with carrier frequency fc, and a gate signal of an output pulse corresponding to the torque command T* is output from the motor control device 1 to the inverter 3.

Here, in the conventional calculation process shown in FIG. 6(a), current detection is performed only once in each control cycle. Furthermore, the result of the calculation process performed in each control cycle is reflected only once in the valley of the carrier wave for the duty of the next control cycle. For example, during a control cycle Ts[n], current detection is first performed in the valley of the carrier wave, and calculation processing is executed based on the current detection result. The duty command value obtained by the calculation process in the control cycle Ts[n] is output in the valley of the carrier wave as a gate signal for adjusting the output voltage of the inverter 3 in the next control cycle Ts[n+1].

FIG. 6(b) shows the state of the calculation processing in an embodiment to which the present invention is applied. In this embodiment, current detection is performed twice within each period of a triangular carrier wave that changes periodically with carrier frequency fc. Then, a series of calculation processing similar to that in the conventional example is repeated using the results of the two immediately preceding current detections, and a gate signal of an output pulse corresponding to the torque command T* is output from the motor control device 1 to the inverter 3.

Here, in the calculation process of this embodiment shown in FIG. 6(b), current detection is performed twice in each control cycle. The results of the calculation process performed in each control cycle are reflected in the valley and peak of the carrier wave for the duty of the next control cycle. For example, in each of the control cycles Ts[n-1] and Ts[n], current detection is performed in the valley and peak of the carrier wave. The calculation process of the control cycle Ts[n] is performed based on the current value detected in the peak of the carrier wave in the immediately preceding control cycle Ts[n-1] and the current value detected in the valley of the carrier wave at the beginning of the control cycle Ts[n]. The duty command value obtained by the calculation process of the control cycle Ts[n] is output in the valley and peak of the carrier wave as a gate signal for adjusting the output voltage of the inverter 3 in the next control cycle Ts[n+1].

In the motor control device 1 of this embodiment, the motor 2 is controlled via the inverter 3 by the calculation process described above. As a result, the current detection value is obtained twice per calculation cycle, and two types of duty commands are determined and reflected in the control of the motor 2. Therefore, without increasing the number of current control calculations per carrier wave, it is possible to realize control of the motor 2 that more finely reflects the detection results of the current flowing through the motor 2 than in the conventional example, thereby improving motor control performance.

In this embodiment, an example has been described in which a current detection value is obtained twice per calculation cycle, and two types of duty commands are calculated per calculation cycle based on these current detection values, but the present invention is not limited to this, and the number of times that a current detection value is obtained per calculation cycle and the number of types of duty commands calculated per calculation cycle may each be three or more. The present invention is applicable as long as a current detection value is obtained at least twice per calculation cycle and at least two or more types of duty commands are calculated per calculation cycle.

The embodiment of the present invention described above provides the following effects.

(1) The motor control device 1 generates a gate signal for controlling the operation of a switching element for an inverter 3 having multiple switching elements, and outputs the generated gate signal to the inverter 3 to control the drive of an AC motor 2 connected to the inverter 3. The motor control device 1 includes a current detection unit 12 that acquires a current detection value corresponding to the current flowing through the AC motor 2, a current control unit 14 that acquires a current command value for the AC motor 2 and calculates a voltage command value for the AC motor 2 for each predetermined calculation period based on the current command value and the current detection value, and a voltage control unit 16 that executes a control calculation based on the voltage command value for each calculation period and obtains a duty command for generating a gate signal. The current detection unit 12 acquires the current detection value at least twice for each calculation period, and the voltage control unit 16 obtains at least two types of duty commands for each calculation period and reflects them in the gate signal. As a result, it is possible to improve motor control performance while suppressing an increase in processing load during motor control at high speeds.

(2) The motor control device 1 includes a current converter 13 that converts the detection values of the three-phase AC current acquired by the current detector 12 into a d-axis current value Id and a q-axis current value Iq. The current detector 12 acquires first three-phase AC current detection values Iua, Iva, Iwa and second three-phase AC current detection values Iub, Ivb, Iwb at different timings within a calculation period. The current converter 13 calculates, by the first current converter 131 and the second current converter 132, a first d-axis current value Id1 and a first q-axis current value Iq1 based on the first three-phase AC current detection values Iua, Iva, Iwa, and a second d-axis current value Id2 and a second q-axis current value Iq2 based on the second three-phase AC current detection values Iub, Ivb, Iwb. Then, the

adders

133, 134 and the

multipliers

135, 136 calculate the d-axis current value Id from the average value of the first d-axis current value Id1 and the second d-axis current value Id2, and calculate the q-axis current value Iq from the average value of the first q-axis current value Iq1 and the second q-axis current value Iq2. The current control unit 14 calculates the d-axis voltage command Vd* and the q-axis voltage command Vq*, which are voltage command values for the AC motor 2, based on the d-axis current command Id* and the q-axis current command Iq* for the AC motor 2 and the d-axis current value Id and the q-axis current value Iq calculated by the current conversion unit 13. In this way, the current control unit 14 can appropriately calculate the voltage command value for the AC motor 2 using the current detection values acquired twice per calculation period by the current detection unit 12.

(3) The current control unit 14 calculates the d-axis voltage command Vd* and the q-axis voltage command Vq* for each predetermined calculation period. The voltage control unit 16 calculates, using the first voltage conversion unit 161 and the second voltage conversion unit 162, first three-phase voltage command values Vua*, Vva*, and Vwa* at a first timing within the calculation period and second three-phase voltage command values Vub*, Vvb*, and Vwb* at a second timing different from the first timing within the calculation period, based on the d-axis voltage command Vd* and the q-axis voltage command Vq*. In addition, the overmodulation

gain multiplication units

163a and 163b, the zero-

phase addition units

164a and 164b, the

duty conversion units

165a and 165b, and the dead time

compensation addition units

166a and 166b calculate a first duty value based on the first three-phase voltage command values Vua*, Vva*, and Vwa*, and a second duty value based on the second three-phase voltage command values Vub*, Vvb*, and Vwb*. Then, the duty output unit 167 calculates the duty command values Dua, Dub, Dva, Dvb, Dwa, and Dwb based on these duty values. In this way, the voltage control unit 16 can reliably calculate two types of duty commands for each calculation period and reflect them in the gate signal.

(4) The voltage control unit 16 performs dead time compensation by adding a predetermined dead time compensation value to the first three-phase voltage command values Vua*, Vva*, Vwa* and the second three-phase voltage command values Vub*, Vvb*, Vwb*, respectively, using the dead time

compensation adder units

166a, 166b, and calculates the first duty value and the second duty value based on the first three-phase voltage command values Vua*, Vva*, Vwa* and the second three-phase voltage command values Vub*, Vvb*, Vwb* after the dead time compensation. In this way, dead time compensation is reliably performed for the duty command values Dua, Dub, Dva, Dvb, Dwa, Dwb, and it is possible to prevent the upper arm and the lower arm of each phase from being turned on simultaneously in the inverter 3.

The present invention is not limited to the above-described embodiment, and various modifications are possible without departing from the spirit of the present invention.

1...motor control device, 2...motor, 3...inverter, 4...rotational position detector, 5...high voltage battery, 7...current sensor, 8...rotational position sensor, 10...current command generator, 11...speed calculator, 12...current detector, 13...current converter, 14...current controller, 15...carrier frequency controller, 16...voltage controller, 17...gate signal generator, 31...inverter circuit, 32...gate driver circuit, 33...smoothing capacitor, 100...motor driver system

Claims

A motor control device that generates a gate signal for controlling an operation of an inverter having a plurality of switching elements, and outputs the generated gate signal to the inverter, thereby controlling driving of an AC motor connected to the inverter,
a current detection unit for obtaining a current detection value corresponding to a current flowing through the AC motor;
a current control unit that acquires a current command value for the AC motor, and calculates a voltage command value for the AC motor for each predetermined calculation period based on the current command value and the current detection value;
a voltage control unit that executes a control calculation based on the voltage command value for each calculation period to obtain a duty command for generating the gate signal,
The current detection unit obtains the current detection value at least twice in each calculation period,
the voltage control unit obtains at least two types of the duty commands for each calculation period and reflects them in the gate signal;
Motor control device.
2. The motor control device according to claim 1,
a current converter for converting the current detection value into a d-axis current value and a q-axis current value,
the current detection unit obtains a first current detection value and a second current detection value at different timings within the calculation period;
The current conversion unit is
calculating a first d-axis current value and a first q-axis current value based on the first current detection value, and a second d-axis current value and a second q-axis current value based on the second current detection value;
determining the d-axis current value from an average value of the first d-axis current value and the second d-axis current value;
determining the q-axis current value from an average value of the first q-axis current value and the second q-axis current value;
the current control unit calculates the voltage command value based on a d-axis current command value and a q-axis current command value for the AC motor and the d-axis current value and the q-axis current value calculated by the current conversion unit.
Motor control device.
2. The motor control device according to claim 1,
the current control unit calculates a d-axis voltage command value and a q-axis voltage command value for each calculation cycle;
The voltage control unit is
calculating a first three-phase voltage command value at a first timing within the calculation cycle and a second three-phase voltage command value at a second timing within the calculation cycle that is different from the first timing, based on the d-axis voltage command value and the q-axis voltage command value;
calculating a first duty value based on the first three-phase voltage command value and a second duty value based on the second three-phase voltage command value;
determining the duty command based on the first duty value and the second duty value;
Motor control device.
4. The motor control device according to claim 3,
the voltage control unit performs dead time compensation by adding a predetermined dead time compensation value to the first three-phase voltage command value and the second three-phase voltage command value, respectively, and determines the first duty value and the second duty value based on the first three-phase voltage command value and the second three-phase voltage command value after the dead time compensation.
Motor control device.