WO2024017769A2

WO2024017769A2 - Multi-phase power converter with fixed resonant capacitor and switch-controlled capacitor

Info

Publication number: WO2024017769A2
Application number: PCT/EP2023/069575
Authority: WO
Inventors: Mojtaba Forouzesh; Xiang Yu; Bo SHENG; Yang Chen; Samuel Dylan Webb; Yan-Fei Liu; Philip KORTA; Oliver OBERLEITNER; Michael NEUDORFHOFER; Lakshmi Varaha IYER; Martin Winter
Original assignee: Magna powertrain gmbh & co kg
Priority date: 2022-07-18
Filing date: 2023-07-13
Publication date: 2024-01-25
Also published as: WO2024017769A3

Abstract

A multi-phase power converter includes a plurality of converter phases. Each of the converter phase includes a resonant tank and an inverter stage configured to switch an input power at an operating frequency to apply a switched power to the resonant tank. The switched power approximates an alternating current (AC) waveform. A converter phase of the plurality of converter phases includes a switch-controlled capacitor (SCC) configured to vary a capacitance of a corresponding resonant capacitor. Another converter phase of the plurality of converter phases is configured as a non-SCC phase, without an SCC, and including a non-SCC resonant capacitor having a fixed capacitance value.

Description

MULTI-PHASE POWER CONVERTER WITH FIXED RESONANT CAPACITOR AND SWITCH-CONTROLLED CAPACITOR

FIELD

The present disclosure relates generally to multi-phase power converters, such as induc- tor-inductor-capacitor (LLC) type power converters for use as a low-voltage DC-DC converter (LDC) in an electric vehicle (EV).

BACKGROUND

With advancements of power conversion technology and power electronics devices, high efficiency and high power density become a major challenge for front-end DC-DC converters. LLC resonant converter has been widely accepted in recent years by providing both high efficiency and high power density for numerous applications, such as servers, flat panel TVs, and LED lighting.

Increasing load demands call for high-power converters. For example, high power electronic converters may be used in Electric Vehicles (EVs) for charging a high-capacity high voltage battery (such as a battery with a nominal voltage of 250 V - 450 V) as well as a low voltage battery (such as a battery with a nominal voltage of 10 V - 16 V) that are installed on a typical EV. The high voltage battery may provide power to drive electric traction motors that move the vehicle. The high voltage battery may also be used to charge the low voltage battery that supplies numerous auxiliary loads of the EV. Hence, a high- power DC-DC converter with a wide input and output voltage range is needed to maintain the state of charge of the low voltage battery. The charging requirement of the low voltage battery charger can easily exceed 200 A, which can be challenging for a single-phase converter design. Multi-phase converters may be employed in this application for distributing current stress.

Inductor-inductor-capacitor (LLC) type resonant power converters are used in various industrial applications which is due to their soft-switching performance and ability to achieve

2022P00096WQ high conversion efficiencies. Due to the sensitivity of the resonant tank impedance to component tolerances, it can be challenging to connect such LLC resonant converters in a parallel arrangement. Small differences in component values may result in a large current imbalance between the paralleled phases, which can lead to failure in high current applications. Several different solutions have been proposed to achieve current sharing in multi-phase resonant converters, including current-controlled inductors, star connection of transformer primary windings, and adjusting turn-on timing control for secondary side switches. However, all these methods have limitations. Converters with current-controlled inductors may require extra inductors, which can be large and costly, Star-connected transformer primary windings may require a three-phase LLC interleaving topology. Turnon timing control for secondary side may add complexity to the operation of secondary side switches.

Switch-Controlled Capacitors (SCC) may be used to control a voltage gain in resonant converters. A full-wave SCC has been successfully implemented in a three-phase induc- tor-inductor-capacitor (LLC) type power converter for wide input/output voltage applications. However, multiphase power converters that employ SCCs generally use SCCs in each of the phases.

Multi-phase resonant converters can achieve current sharing between the paralleled phases using SCC circuits. However, with the conventional resonant tank design method, one SCC circuit included in each paralleled phase. That increases the cost and control complexity.

SUMMARY

The present disclosure provides a multi-phase power converter. The multi-phase power converter includes a plurality of converter phases. Each of the converter phase includes a resonant tank and an inverter stage configured to switch an input power at an operating frequency to apply a switched power to the resonant tank. The switched power approximates an alternating current (AC) waveform. A converter phase of the plurality of converter phases includes a switch-controlled capacitor (SCC) configured to vary a capacitance of a corresponding resonant capacitor of the resonant capacitors. Another converter phase of the plurality of converter phases is configured as a non-SCC phase, without an SCC and including a non-SCC resonant capacitor of the resonant capacitors having a fixed capacitance value.

The present disclosure also provides a method for operating a multi-phase power converter. The method includes: providing a plurality of converter phases, each including an inverter stage and a resonant tank having a resonant capacitor; switching, by the inverter stage of each of the plurality of converter phases, an input power to apply a switched power approximating an alternating current (AC) waveform to a corresponding one of the resonant tanks; and varying, by a switch-controlled capacitor (SCC), a capacitance of a corresponding resonant capacitor in the resonant tank of at least one converter phase of the plurality of converter phases. The plurality of converter phases of the multi-phase power converter further includes another converter phase that is configured as a non-SCC phase, without an SCC and including a non-SCC resonant capacitor of the resonant capacitors having a fixed capacitance value.

BRIEF DESCRIPTION OF THE DRAWINGS

Further details, features and advantages of designs of the invention result from the following description of embodiment examples in reference to the associated drawings.

FIG. 1 shows a schematic diagram of a multi-phase LLC power converter with switch- controlled capacitors (SCC) in each of two phases;

FIG. 2 shows a schematic diagram of a multi-phase LLC power converter with an SCC in only one of the phases, in accordance with some embodiments of the present disclosure;

FIGS. 3A-3C show graphs illustrating voltage gain vs. switching frequency for an LLC power converter at different operating conditions;

FIG. 4 shows an enlarged portion of FIG. 2; FIG. 5 shows a graph with plots of voltages and currents in an LLC power converter over a common time scale in accordance with some embodiments of the present disclosure.

FIGS. 6A-6D show graphs illustrating voltage gain comparisons of a two-phase LLC power converter of the present disclosure for various operating conditions;

FIGS. 7A-7D show graphs illustrating voltage gain comparisons of a two-phase LLC power converter of the present disclosure for various operating conditions; and

FIG. 8 shows a flow chart illustrating steps in a method for operating a multi-phase power converter.

DETAILED DESCRIPTION

Referring to the drawings, the present invention will be described in detail in view of following embodiments. The present disclosure provides a new resonant tank design approach for multi-phase resonant converters that only requires one switch-controlled capacitor (SCC) circuit, which significantly reduces complexity and implementation cost when compared with conventional designs.

Example embodiments of a multi-phase power converter are provided. The multi-phase power converters include a plurality of converter phases each including a resonant tank and an inverter stage configured to switch an input power at an operating frequency to apply a switched power to the resonant tank. The multi-phase power converters of the present disclosure include inductor-inductor-capacitor (LLC) type resonant power converters, where the resonant tank includes two inductors and a capacitor. However, aspects of the present disclosure are applicable to multi-phase power converters having other types of resonant tanks. For example, aspects of the present disclosure may be used with resonant tanks having an inductor-capacitor resonance, such as an inductor-capacitor-ca- pacitor (LCC) configuration or a capacitor-inductor-inductor (CLL) configuration, etc. The multi-phase power converters of the present disclosure include one non-SCC phase with a resonant capacitor having a fixed capacitance value, and one or more SCC phases each including an SCC configured to vary a capacitance of the corresponding resonant capacitor. FIG. 1 shows a schematic diagram of a multi-phase LLC converter 10 with full-wave switch-controlled capacitor (SCC) and two transformers on each phase for high current applications. The main idea is to be able to tune the impedance of each LLC tank by modulating the resonant capacitance of each phase to compensate for the impedance mismatch caused by component tolerances. While the mentioned method is quite effective in balancing the three-phase current sharing, the design and control implementation are complex. Half-wave SCC have been implemented in an interleaved two-phase LCLC resonant converter for data center applications with a wide input voltage range. The SCC circuit has been used in both phases as because of component tolerances the voltage gain of one phase is larger than the other one at minimum input voltage condition and it is vice versa at the maximum input voltage condition.

As shown in FIG. 1 , a first multi-phase LLC converter 10 may receive DC power from a DC source 20, such as a battery, and having an input voltage \Z_/n via input conductors 22p, 22n. The first multi-phase LLC converter 10 may supply output power to a load 24 via output conductors 26p, 26n. The output power may be supplied at an output voltage Vo, which may be higher than or lower than the input voltage Vm. The first multi-phase LLC converter 10 includes a plurality of LLC phases 30a, 30n connected in parallel. The first multi-phase LLC converter 10 of FIG. 1 includes two of the LLC phases 30a, 30n. However, a multi-phase LLC converter may include any number n of the LLC phases 30a, 30n, where n is a number greater than 1 . Each of the LLC phases 30a, 30n may have a similar or identical construction. For the simplicity of the description, only one of the LLC phases 30a, 30n is described herein. The LLC phases 30a, 30n include a first LLC phase 30a, which may also be called Phase 1 .

The first LLC phase 30a includes an input capacitor CM connected across the input conductors 22p, 22n. The first LLC phase 30a also includes an inverter stage 32 having a plurality of switching transistors Qu, Q21, Q31, Q41 that function to switch an input power at an operating frequency to generate a switched power that approximates an alternating current (AC) waveform. The first LLC phase 30a also includes a transformer 34. The transformer 34 may include two or more transformers which may include series-connected primary windings for enhanced current carrying capability. The transformer 34 may have a 1 :1 winding ratio and may provide electrical isolation between the input conductors 22p, 22n and the output conductors 26p, 26n. However, other configurations may be used, and the transformer 34 may have a different winding ratio to provide a step-up or a step-down conversion.

The first LLC phase 30a also includes a first resonant tank 35, which includes a resonant inductor L_ri, a SCC 40, a series capacitor Cri, and a parallel inductance L_mi. In some embodiments, the parallel inductance L_mi is a physical device. Alternatively or additionally, the parallel inductance L_mi may represent inductance characteristics of the transformer 34. The SCC 40 and the series capacitor Cri are connected in series, with the series combination of the SCC 40 and the series capacitor Cri together providing a resonant capacitance C_ri.

The SCC 40 includes a switched capacitance Cai connected in parallel with one or more switches Sn, S21. The two or more switches S11, S21 may include field-effect transistors (FETs), as shown in FIG. 1. However, other types of switching devices may be used, such as junction transistors. The two or more switches may be arranged to form a full-wave switch configured to selectively conduct current in either of two opposite polarities to bypass the switched capacitor C_ai. For example, the FETs may be arranged as shown in FIG. 1 , in a back-to-back configuration, with each of their source terminals connected to a common node.

The first LLC phase 30a also includes a rectifier stage 36 connected to a secondary side of the transformer 34, opposite from the first resonant tank 35 and configured to rectify AC power therefrom and to provide the DC output power on the output conductors 26p, 26n. The rectifier stage 36 may be configured as a synchronous rectifier including a plurality of transistors SR11, SR21 , SR31 , SR41. However, the rectifier stage 36 may have a different configuration and/or arrangement. The first LLC phase 30a also includes an output capacitor C01 connected across the output conductors 26p, 26n to smooth ripples in the DC output power generated by the rectifier stage 36.

FIG. 2 presents a second resonant converter 110, which includes a single SCC 40. The second resonant converter 110 of the present disclosure may achieve current sharing over a wide input/output voltage range. The proposed design approach only requires one SCC circuit to achieve current balancing and hence it reduces the control complexity and implementation costs of the conventional approach. In this design, the impedance of one phase is kept always larger than the other phase over the switching frequency and load range. Hence, it is possible to reduce the impedance of one phase by the SCC circuit to increase the load share on that phase and achieve balanced current sharing between the phases.

As shown in FIG. 2, the second resonant converter 110 includes two LLC phases 130a, 130b including a static LLC phase 130a that is similar to the first LLC phase 30a of FIG. 1 , except with a second resonant tank 135 that is different from the first resonant tank 35 of the first LLC phase 30a of FIG. 1. In particular, the second resonant tank 135 does not include a SCC 40. The two LLC phases 130a, 130b of the second resonant converter 110 also include a second LLC phase 30b that is similar or identical to the to the first LLC phase 30a of FIG. 1 , including the first resonant tank 35 having the SCC 40.

The impedances of the resonant tanks in paralleled resonant converters determines the current distribution between them. The phase with a resonant tank with a larger impedance will carry less output current. On the other hand, the SCC 40 can be operated to reduce an equivalent resonant capacitance compared with a resonant tank that is not so equipped. Hence, the SCC 40 may function to increase the output current in a phase. Figure 2 shows the proposed second resonant converter 110 in a two-phase configuration, with one SCC 40 located in the second LLC phase 30b.

It is crucial in the design procedure of the proposed converter to make sure that when the SCC 40 is removed, the second LLC phase 30b always carries less output current, and when the SCC 40 is operating, the output current of the second LLC phase 30b can be increased to achieve a balanced current sharing. For the sake of manufacturing convenience, the magnetics are built with the same values (i.e. , L_ri = Lr2 and L_pi = L_P2) and only the resonant capacitance of the second LLC phase 30b is chosen to be larger than the resonant capacitance of the first LLC phase 30a (i.e., Cr2 > Cri). The first step in the design is the selection of the transformer turn ratio and then the resonant tank parameters of the static LLC phase 130a for half-load operation. The resonant tank of the second LLC phase 30b can be designed such that the impedance of the second LLC phase 30b (i.e. with the SCC 40) is always smaller than the impedance of the static LLC phase 130a over an entire range of operation conditions. Then, the SCC 40 should be designed such that the impedance matching can be achieved based on the predefined component tolerances.

The turn ratio of the transformer is determined the same as that of the conventional design approaches. As the operation range is desired to be kept below the series resonant frequency (i.e., f_r), the resonant frequency where the voltage gain of the LLC tank is unity should be set for the maximum input voltage and minimum output voltage.

where N_p and N_s are the transformer's primary and secondary number of turns.

For a low voltage onboard charger of an EV, the high voltage battery side may have a DC voltage of 250 V-450 V, and the low voltage battery side may have a DC voltage of 10 V to 16 V. With 450V maximum input and 10V minimum output, the equivalent transformer turn ratio should be 450V/10V = 45. The transformer's final turn ratio is selected as 44, with the turn ratio being 22: 1 : 1 (n: 1 : 1 ) for each transformer.

Resonant components design in the phase without an SCC circuit

In the static LLC phase 130a, the design parameters include L_ri, C_r-i, and L_pi. The parameters may be designed the same as conventional LLC converters. The design objective is to meet the desired frequency range and all the input/output voltage and load regulations and at the same time reduce the circuiting current at the primary side of the transformers.

The worst-case scenario for the voltage gain requirement of the LLC tank is for the highest load condition at the highest gain. The following corner conditions are considered for half of the rated power of the EV low voltage onboard battery charger:

The maximum voltage Gain=2.82 for 250 V input, 16 V output, and 110 A load.

The worst-case scenario with Gain=1 .93 for 320V input, 14 V output, and 140 A load. The minimum voltage Gain=0.98 for 450 V input, 10 V output, and 140 A load.

The design is based on the well-known First Harmonic Approximation (FHA) method, and then the parameters need to be fine-tuned based on PSM simulation results. It should be mentioned that the voltage gain found from FHA is lower than the actual gain for the switching frequencies far away from the resonant frequency. In this design, the desired switching frequency is from 250 kHz to 450 kHz. A relatively small quality factor (i.e., Q=0.3) for the worst-case scenario is considered as a wide input/output voltage and load range need to be covered. Moreover, the circulating currents can be reduced by using a large inductance ratio. Hence, a relatively large inductance ratio is considered in the design (i.e., LmlL^Q). The final resonant tank parameters of the static LLC phase 130a are listed in Table 1 . FIGS. 3A-3C show voltage curves for different operating conditions.

Table 1. Parameters of Single-Phase SCC-LLC Resonant Converter

(i.e. the static LLC phase 130a)

Resonant Components Design in the Phase with an SCC Circuit

The magnetics in the first resonant tank 35 of the second LLC phase 30b may be identical to the magnetics in the second resonant tank 135 of the static LLC phase 130a (i.e., L_r2 = L_ri and L_m2 = L_mi), and hence only the resonant capacitance Cr2 and the SCC capacitance Ca2 may need to be designed. The design criteria are such that the voltage gain of the static LLC phase 130a 2 is kept below the voltage gain of the static LLC phase 130a over the switching frequency range in all operating conditions. Both capacitances are designed based on the operation principles of the SCC circuit. The SCC circuit is shown in FIG. 4 and its operation is illustrated in FIG. 5. FIG. 4 shows the structure of the SCC 40, which is configured as a full wave device, operable to switch current in either of two opposite directions, and which includes the switched capacitor Ca in parallel with the two MOSFETs SC1 , SC2.

FIG. 5 is a graph 200 showing plots of voltages and currents in an LLC power converter over a common time scale. Graph 200 includes plot 202 having line 204 showing current IAB, plot 206 having line 208 showing gate voltage V_gsi of SCC switch SC1 , and plot 210 having line 212 showing gate voltage V_gS2 of SCC switch SC2. The gate voltages V_gsi, V_gS2 are driven between a high, or asserted state to cause the corresponding SCC switch SC1 , SC2 to be in a conductive state (which may also be called “on”), and a low, or deasserted state to cause the corresponding SCC switch SC1 , SC2 to be in a non-conduc- tive state (which may also be called “off’). Graph 200 also includes plot 214 having line 216 showing current /ca into the switched capacitor Ca, and plot 218 having line 219 showing voltage Vca across the switched capacitor Ca.

Assuming a sinusoidal current IAB is flowing through SCC, as shown in FIG. 5, the current zero-crossing points are at angles 0, IT, 2TT, .... etc. For a positive half cycle, Si is turned OFF at an angle of 2nir+a. After Si is turned OFF, the current flows from A to B via C_a and charges the capacitor until the next current zero-crossing point at (2n+1)n. Then, the current reverse direction, and begins to discharge C_a. After C_a is fully discharged, the negative current is about to flow from B to A via the body diode of Si. To prevent its body diode from conducting, Si is turned ON. It remains ON for the rest of the cycle and turns OFF again at angle (2n+2)ir+a. Following the same procedure, S2 controls the negative half cycle.

The equivalent capacitance of SCC, C_a is modulated by angle a and can be expressed using equation (2), below:

where C_a is the SCC capacitor. The equivalent resonant capacitance can be derived using equation (3):

where C_r is the series resonant capacitor. Based on the above two equations (2) and (3).

The equivalent resonant capacitance can be rewritten as equation (4), below:

With the angle a being from TT/2 to IT, the equivalent capacitance changes from a minimum value to a maximum value. In extreme cases, when a = TT/2, current IAB will flow through C_a and bypass SCC MOSFETs. Thus, the equivalent resonant capacitance is at the minimum value which is equal to C_r and C_a connected in series. When a = TT, current IAB will flow through SCC MOSFETs and bypass capacitor C_a, which makes the equivalent resonant capacitance toward its maximum value C_r. To have a continuous SCC operation and for safety reasons, a minimum and maximum operating angle is considered in practice

By modulating the phase angle, the equivalent resonant capacitance in the second LLC phase 30b is modified, which modifies the impedance of the first resonant tank 35 in the second LLC phase 30b. When the phase angle is a certain value between the minimum and the maximum, the impedance of the static LLC phase 130a matches that of the second LLC phase 30b, and current sharing is achieved. If the phase angle increases, the impedances of the second LLC phase 30b increase, and the second LLC phase 30b carry smaller currents. On the contrary, if the phase angle decreases, the impedances of the second LLC phase 30b decrease, and the second LLC phase 30b carries larger currents. Thus, the design criteria can be summarized as follows:

When the phase angle is maximum, in any components’ tolerances and operation points, the second LLC phase 30b carries less current than the static LLC phase 130a. When the phase angle is minimum, in any components’ tolerances and operation points, the second LLC phase 30b carries more current than the static LLC phase 130a.

The tolerances of components (i.e. differences in actual values of design characteristics, such as resistance, capacitance, and inductance) in the LLC converter are the main reason causing impedances unmatched and current unbalancing, which is compensated by the SCC circuit. In this design, +/-5% tolerances for all the resonant elements and the SCC capacitor are considered. The two design criteria can be expressed by equations (5) - (6), below:

where l_{o i} and l₀2 are the output currents of the static LLC phase 130a and the second LLC phase 30b, respectively. M is the LLC tank voltage gain, f_sw is the switching frequency, Cr_ eq(a = max) and Cr_ _eq(a = min) are the equivalent capacitances when the phase angles are maximum and minimum, respectively. Note that /Wand f_sw indicate different operating conditions. The equivalent resonant capacitance C_r_eq satisfies equation (7):

When components’ tolerances are considered, the following equations can be derived:

where the subscripts _min and _max indicate the minimum and maximum values of the associated variables due to components’ tolerances. C_r_ eq(a = max, Ca2 = min, Cr2 = min) is the equivalent resonant capacitance when cr is maximum, Ca2 is minimum and Cr2 is minimum, Cr_eq(a = min, ca2 = max, cr2 = max) is the equivalent resonant capacitance when a is minimum, Ca2 is maximum and Cr2 is maximum.

Increasing any component’s value in a passive impedance network will increase the total impedance, and a larger impedance will contribute to a smaller output current. Then, the two criteria can be expressed by equations (12) and (13):

where X_ max — 1.05xXand X_ min — 0.95*X considering +/-5% tolerances, X represents L_mi, Lm2, L_ri, L_r2, C_ri, C_r2 and C_a2. The two equivalent capacitances C_r_ eq(a = max, Ca2 = min, Cr2 = min) and C_r_eq(a = min, ca2 = max, cr2 = max) which satisfy the design criteria are determined first.

Current sharing performance becomes worse at heavy loads, and it is less influenced by the voltage gain. Thus, initial numbers are found from FHA and then tuned with the help of PSIM simulations. The two equivalent capacitances are determined first in the heaviest load operation at the nominal condition voltage gain, then are verified and adjusted in other operation points. After some iterations, the two equivalent capacitances are determined as equations (14) - (15):

With the help of the expression of the equivalent resonant capacitance and PSIM simulations, and considering the minimum and maximum a angle, Cr2 and Ca2 are designed as Cr2 = 11 nF, Ca2 = 9.5 nF. Hence, the final designed parameters are shown in Table 2.

Table 2. Designed parameters of the second resonant converter 110.

In order to check that the designed parameters can meet the desired tolerances, some analysis has been done. FIGS. 6A-6D show the voltage gain comparison of the two-phase converter considering a 5% increase in the resonant components of tank 1 and considering a 5% decrease in the resonant components of tank 2, including the SCC capacitor. In other words, FIGS. 6A-6D show the two-phase converter with L_ri+5%, L_mi+5%, C_ri+5%, Lr2-5%, Lm2-5%, C_r2~5°/o, C_a2~5°/o, and cr=160°. In this case, the SCC a is kept at its maximum to see if the voltage gains of the second LLC phase 30b are kept smaller than the static LLC phase 130a in all the operating conditions. From FIGS. 6A-6D it can be seen that the condition of equation (12) is satisfied over the input/output voltage range and operating at rated power.

FIG. 7 shows the voltage gain comparison of the two-phase converter considering a 5% decrease in the resonant components of tank 1 and considering a 5% increase in the resonant components of tank 2 including the SCC capacitor. In other words, FIGS. 7A-7D show the two-phase converter with L_ri-5%,

C_ri-5%, L_r2+5%, L_m2+5%, Cr2+5%,

Ca2+5%, and cr=100°. In this case, the SCC a is kept at its minimum to see if the voltage gains of the second LLC phase 30b are kept larger than the static LLC phase 130a in all the operating conditions. From Figure 7 it can be seen that the condition described in equation (13) is satisfied over the input/output voltage range and operating at rated power.

A method 300 for operating a multi-phase power converter is shown in the flow chart of FIG. 8. As can be appreciated in light of the disclosure, the order of operation within the method is not limited to the sequential execution as illustrated in FIG. 8, but may be performed in one or more varying orders as applicable and in accordance with the present disclosure. The method 300 includes providing, at step 302, a plurality of converter phases, with each of the converter phases including an inverter stage and a resonant tank having a resonant capacitor.

The method 300 also includes switching, at step 304, and by the inverter stage of each of the plurality of converter phases, an input power to apply a switched power approximating an alternating current (AC) waveform to a corresponding one of the resonant tanks.

The method 300 also includes varying, at step 306, and by a switch-controlled capacitor (SCC), a capacitance of a corresponding resonant capacitor in the resonant tank of at least one converter phase of the plurality of converter phases.

The plurality of converter phases of the multi-phase power converter further includes another converter phase that is configured as a non-SCC phase, without an SCC and including a non-SCC resonant capacitor of the resonant capacitors having a fixed capacitance value.

In some embodiments, at least one converter phase of the plurality of converter phases is configured as an LLC phase with the resonant tank including a resonant inductor, the resonant capacitor, and a parallel inductance.

In some embodiments, at least one converter phase of the plurality of converter phases further includes a transformer having a primary winding and a secondary winding, with the primary winding connected in series with the resonant capacitor.

In some embodiments, the plurality of converter phases includes exactly two converter phases including one converter phase having the SCC, and one converter phase configured as the non-SCC phase.

In some embodiments, the SCC is configured as a full-wave SCC circuit including a switched capacitor and a full-wave switch configured to selectively conduct current in either of two opposite polarities to bypass the switched capacitor. In some embodiments, the full-wave switch includes two field-effect transistors (FETs) in a series configuration, and with the series configuration of the FETs connected in parallel with the switched capacitor.

In some embodiments, the resonant capacitor of the converter phase including the SCC further includes a series capacitor connected in series with the SCC.

The foregoing description is not intended to be exhaustive or to limit the disclosure. Individual elements or features of a particular embodiment are generally not limited to that particular embodiment, but, where applicable, are interchangeable and can be used in a selected embodiment, even if not specifically shown or described. The same may also be varied in many ways. Such variations are not to be regarded as a departure from the disclosure, and all such modifications are intended to be included within the scope of the disclosure.

Claims

1 . A multi-phase power converter comprising: a plurality of converter phases, each including a resonant tank having a resonant capacitor and an inverter stage configured to switch an input power at an operating frequency to apply a switched power approximating an alternating current (AC) waveform to the resonant tank; a converter phase of the plurality of converter phases including a switch-controlled capacitor (SCC) configured to vary a capacitance of a corresponding resonant capacitor of the resonant capacitors; and another converter phase of the plurality of converter phases configured as a non- SCC phase, without an SCC and including a non-SCC resonant capacitor of the resonant capacitors having a fixed capacitance value.

2. The multi-phase power converter of Claim 1 , wherein at least one converter phase of the plurality of converter phases is configured as an LLC phase with the resonant tank including a resonant inductor, the resonant capacitor, and a parallel inductance.

3. The multi-phase power converter of Claim 1 , wherein at least one converter phase of the plurality of converter phases further includes a transformer having a primary winding and a secondary winding, with the primary winding connected in series with the resonant capacitor.

4. The multi-phase power converter of Claim 1 , wherein the plurality of converter phases includes exactly two converter phases including one converter phase having the SCC, and one converter phase configured as the non-SCC phase.

5. The multi-phase power converter of Claim 1 , wherein the SCC is configured as a full-wave SCC circuit including a switched capacitor and a full-wave switch configured to selectively conduct current in either of two opposite polarities to bypass the switched capacitor.

6. The multi-phase power converter of Claim 5, wherein the full-wave switch includes two field-effect transistors (FETs) in a series configuration, and with the series configuration of the FETs connected in parallel with the switched capacitor.

7. The multi-phase power converter of Claim 1 , wherein the resonant capacitor of the converter phase including the SCC further includes a series capacitor connected in series with the SCC.

8. The multi-phase power converter of Claim 1 , wherein each of the plurality of converter phases each further includes a synchronous rectifier having a plurality of switching transistors configured to rectify an AC power to generate a direct current (DC) output power.

9. A method for operating multi-phase power converter, said method comprising: providing a plurality of converter phases, each including an inverter stage and a resonant tank having a resonant capacitor; switching, by the inverter stage of each of the plurality of converter phases, an input power to apply a switched power approximating an alternating current (AC) waveform to a corresponding one of the resonant tanks; and varying, by a switch-controlled capacitor (SCC), a capacitance of a corresponding resonant capacitor in the resonant tank of at least one converter phase of the plurality of converter phases, wherein the plurality of converter phases of the multi-phase power converter further includes another converter phase that is configured as a non-SCC phase, without an SCC and including a non-SCC resonant capacitor of the resonant capacitors having a fixed capacitance value.

10. The method of Claim 9, wherein at least one converter phase of the plurality of converter phases is configured as an LLC phase with the resonant tank including a resonant inductor, the resonant capacitor, and a parallel inductance.

11 . The method of Claim 9, wherein at least one converter phase of the plurality of converter phases further includes a transformer having a primary winding and a secondary winding, with the primary winding connected in series with the resonant capacitor.

12. The method of Claim 9, wherein the plurality of converter phases includes exactly two converter phases including one converter phase having the SCC, and one converter phase configured as the non-SCC phase.

13. The method of Claim 9, wherein the SCC is configured as a full-wave SCC circuit including a switched capacitor and a full-wave switch configured to selectively conduct current in either of two opposite polarities to bypass the switched capacitor.

14. The method of Claim 13, wherein the full-wave switch includes two fieldeffect transistors (FETs) in a series configuration, and with the series configuration of the FETs connected in parallel with the switched capacitor.

15. The method of Claim 9, wherein the resonant capacitor of the converter phase including the SCC further includes a series capacitor connected in series with the SCC.