WO2023010829A1 - 用于天线的双极化辐射单元、天线以及天线系统 - Google Patents

用于天线的双极化辐射单元、天线以及天线系统 Download PDF

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Publication number
WO2023010829A1
WO2023010829A1 PCT/CN2022/077196 CN2022077196W WO2023010829A1 WO 2023010829 A1 WO2023010829 A1 WO 2023010829A1 CN 2022077196 W CN2022077196 W CN 2022077196W WO 2023010829 A1 WO2023010829 A1 WO 2023010829A1
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Prior art keywords
dual
antenna
radiation unit
polarized
polarized radiation
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PCT/CN2022/077196
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English (en)
French (fr)
Inventor
H·普里亚南达
胡中皓
J·德西亚
D·达马维里亚
Original Assignee
普罗斯通信技术（苏州）有限公司
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Priority to EP22851556.5A priority Critical patent/EP4383455A1/en
Publication of WO2023010829A1 publication Critical patent/WO2023010829A1/zh
Priority to US18/431,071 priority patent/US20240178564A1/en

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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/36—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/52—Means for reducing coupling between antennas; Means for reducing coupling between an antenna and another structure
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q13/00—Waveguide horns or mouths; Slot antennas; Leaky-waveguide antennas; Equivalent structures causing radiation along the transmission path of a guided wave
    • H01Q13/10—Resonant slot antennas
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q21/00—Antenna arrays or systems
    • H01Q21/06—Arrays of individually energised antenna units similarly polarised and spaced apart
    • H01Q21/061—Two dimensional planar arrays
    • H01Q21/062—Two dimensional planar arrays using dipole aerials
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q21/00—Antenna arrays or systems
    • H01Q21/24—Combinations of antenna units polarised in different directions for transmitting or receiving circularly and elliptically polarised waves or waves linearly polarised in any direction
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q21/00—Antenna arrays or systems
    • H01Q21/24—Combinations of antenna units polarised in different directions for transmitting or receiving circularly and elliptically polarised waves or waves linearly polarised in any direction
    • H01Q21/26—Turnstile or like antennas comprising arrangements of three or more elongated elements disposed radially and symmetrically in a horizontal plane about a common centre
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/40—Imbricated or interleaved structures; Combined or electromagnetically coupled arrangements, e.g. comprising two or more non-connected fed radiating elements
    • H01Q5/48—Combinations of two or more dipole type antennas

Definitions

  • the present disclosure relates to the technical field of radio frequency communication, and in particular, the present disclosure relates to a dual-polarization radiation unit for an antenna, an antenna including the dual-polarization radiation unit for an antenna, and an antenna system including the antenna.
  • the so-called sub-6 GHz band In the broad spectrum range between 400 MHz and 6 GHz, the so-called sub-6 GHz band, these spectrum ranges are allotted to telecommunications for communication.
  • analog components such as filters, phase shifters, radiating elements, and amplifiers for such a wide bandwidth. Therefore, the sub-6GHz frequency band is further divided into multiple sub-bands and operated separately to enable the design of analog components.
  • the frequency band below 6 GHz into the following four separate working sub-bands: 600 MHz to 1 GHz, 1.4 GHz to 3 GHz, 3 GHz to 4.2 GHz, and 5 GHz-6 GHz.
  • Base station antennas for LTE, 5G or 3G communications consist of arrays of multiple radiating elements operating in different frequency bands.
  • the inventors of the present disclosure propose the following dual-polarized radiation unit for antennas based on the above considerations, the dual-polarized radiation unit includes: four dipoles, wherein the four The radiating arms of a dipole are planar structures and symmetrical about two mutually perpendicular lines, wherein the two mutually perpendicular lines divide the radiating unit into four areas equally, and the centers of the four areas Partial DC conduction, and wherein each of the four regions has a hollow region.
  • the dual-polarized radiating unit proposed according to the present disclosure has a hollow area, so that its interference with electromagnetic wave signals generated by the radiating unit used in other frequency bands is reduced, so that the dual-polarized radiation proposed according to the present disclosure
  • the unit is more friendly to system integration, and the radiation performance of the integrated multi-band radiation unit will be better than that of the radiation unit in the prior art.
  • the central part of the dual-polarized radiating unit proposed in the present disclosure is conduction with direct current, its manufacturing process is simpler and more convenient, and the dual-polarized radiating unit according to the present disclosure can be constructed integrally , so its product consistency is better, and its radiation performance can be better ensured.
  • the first side of the dual-polarized radiation unit is perpendicular to the corresponding line of the two mutually perpendicular lines, wherein the first side is perpendicular to the corresponding The lines intersect the edges.
  • the two perpendicular lines are usually diagonal lines, that is to say, the edge of the dual-polarized radiation unit to which the diagonal extends is usually one of the dual-polarized radiation units. Corners, not sides.
  • the edge of the dual-polarized radiation unit to which two mutually perpendicular lines of the dual-polarized radiation unit proposed according to the present disclosure extend is a side of the dual-polarized radiation unit, and the side is connected to two mutually perpendicular lines The corresponding line in the line is vertical.
  • the dual-polarized radiation unit according to the present disclosure adopts a corner cutting method, so that the dual-polarized radiation unit proposed according to the present disclosure is relatively Compared with the traditional dual-polarization radiation unit, the area is smaller, thereby saving more materials, and further reducing the manufacturing cost.
  • the dual-polarized radiation unit has an octagonal shape.
  • the dual-polarized radiating unit according to the present disclosure adopts a corner cutting method, so that the dual-polarized radiating unit proposed according to the present disclosure For example, it has an octagonal shape.
  • the area is smaller, thereby saving more materials, thereby reducing manufacturing costs.
  • the dual-polarization radiation unit further includes: four slots, and the four slots are respectively located at adjacent positions of two adjacent areas in the four areas. Two of the four slots point to one polarization direction, for example, a +45° polarization direction, and the other two of the four slots point to another polarization direction, for example, a ⁇ 45° polarization direction.
  • the size of the slot such as the length of the slot, is reduced by such as the aforementioned corner cutting, the resonance frequency of the current loop can be moved higher until the resonance exceeds the working range of the antenna, thereby reducing the impact of other frequency bands on the basis of the present disclosure. Effect of the proposed dual-polarization radiating element.
  • the dual-polarized radiation unit further includes: four feed lines, the four feed lines are associated with the four slots, and each of the four feed lines The wire extends from the middle area of the dual-polarized radiating element to the feeding point of the slot associated therewith, wherein the length of the feeding wire is associated with the matching impedance.
  • the dual-polarization radiating unit further includes: a common mode choke circuit, the common mode choke circuit is configured to be arranged near the feeding point and Associated with one of the four slots.
  • the common mode choke circuit further includes a first track and a second track, wherein the first track and the second track are connected by an inductance coil The form is constructed on both sides of the radial arm parallel to each other, and wherein the electrical lengths of the first track and the second track are equal, and the coils of the first track and the second track are wound around The row direction is the same.
  • the dual-polarization radiating unit further includes: a parallel filter, the parallel filter is configured to be inward from the edge of the hollow region Extended wire.
  • the parallel filter is configured as an open line.
  • the dual-polarization radiating unit further includes: an inductance element, the inductance element is directed from an edge of the hollow area far away from the central area of the dual-polarization radiating unit to the The above-mentioned central area extends.
  • the second aspect of the present disclosure provides an antenna, the antenna comprising: at least one dual-polarized radiation unit according to the first aspect of the present disclosure; and a radiation unit matching circuit.
  • a third aspect of the present disclosure provides an antenna system, the antenna system comprising: the antenna provided according to the second aspect of the present disclosure; and at least one second antenna, wherein the at least one first The working frequency band of the second antenna is higher than the working frequency band of the antenna.
  • the antenna and the second antenna are arranged alternately.
  • the number of columns of the antenna is equal to the number of columns of the second antenna, or the number of columns of the second antenna is equal to the number of columns of the antenna double.
  • the dual-polarized radiation unit proposed according to the present disclosure has a hollow area, so that its interference with electromagnetic wave signals generated by radiation units used in other frequency bands is reduced, so that the proposed dual-polarization radiation unit according to the present disclosure
  • the dual-polarization radiating unit is more friendly to system integration, and the radiation performance of the integrated multi-band radiating unit will be better than that of the radiating unit in the prior art.
  • the central part of the dual-polarized radiating unit proposed in the present disclosure is conduction with direct current, its manufacturing process is simpler and more convenient, and the dual-polarized radiating unit according to the present disclosure can be constructed integrally , so its product consistency is better, and its radiation performance can be better ensured.
  • FIG. 1A shows a perspective view of an antenna system 100 including a dual-polarized radiation unit 110 and a high-frequency radiation unit 120 according to an embodiment of the prior art
  • FIG. 1B shows a side view of an antenna system 100 including a dual-polarized radiating unit 110 and a high-frequency radiating unit 120 according to an embodiment of the prior art
  • Figure 2A shows a schematic diagram of a dual-polarized radiating element 210 for an antenna according to one embodiment of the present disclosure
  • FIG. 2B shows a partially enlarged schematic diagram of a common-mode choke circuit for a dual-polarized radiating element 210 of an antenna according to an embodiment of the present disclosure
  • FIG. 2C shows a side view of a dual-polarized radiating element 210 for an antenna according to one embodiment of the present disclosure
  • Figure 2D shows a side perspective view of a dual polarized radiating element 210 for an antenna according to one embodiment of the present disclosure
  • FIG. 2E shows a bottom perspective view of a dual-polarized radiating element 210 for an antenna according to an embodiment of the present disclosure
  • FIG. 2F shows an exploded view of a dual polarized radiating element 210 for an antenna according to one embodiment of the present disclosure
  • FIG. 3 shows a schematic diagram of an antenna system 300 according to an embodiment of the present disclosure
  • FIG. 4A shows a schematic diagram of an antenna system 400A according to another embodiment of the present disclosure.
  • FIG. 4B shows a schematic diagram of an antenna system 400B according to another embodiment of the present disclosure.
  • FIG. 1A shows a perspective view of an antenna system 100 including a dual-polarized radiating unit 110 and a high-frequency radiating unit 120 according to an embodiment of the prior art
  • FIG. 1B shows an antenna system including a dual-polarization radiating unit 120 according to an embodiment of the prior art.
  • a side view of the antenna system 100 of the dual-polarization radiating unit 110 and the high-frequency radiating unit 120 It can be seen from the two drawings in FIG. 1 that the antenna system 100 disclosed in WO2015/124573A outlines a low-frequency band radiating unit 110 constructed using a four-slot feeding method.
  • the vertical element pitch of the radiating elements 120 of the high-band array is limited to half of the pitch of the radiating elements 110 of the low-band array.
  • the spacing of the radiating elements 110 of the low-frequency band array is usually about 0.7 ⁇ , and a reasonable level of grating lobe can be achieved at a large downtilt angle, wherein the grating lobe is a side lobe that increases as the angle of the scanning beam increases, and
  • the spacing between the radiating elements is directly proportional to the ratio of the wavelengths of the radiating elements of the array. Larger spacing between radiating elements results in higher grating lobes.
  • Grating lobes reduce the gain of the array's radiating elements and cannot be reduced by adjusting the amplitude or phase taper of the array's radiating elements.
  • the frequency spectrum at high frequencies is wider, it is more sensitive to the effects of grating lobes at higher frequencies within the band.
  • Practical spacing between radiating elements used in industry is 0.78 ⁇ to 0.85 ⁇ . If the spacing between the radiating elements of the high frequency band is half of that of the radiating elements of the low frequency band, it is greater than 0.9 ⁇ of the high frequency, resulting in significant grating lobes.
  • the grating lobe of the radiating element of the 0.78 ⁇ array inclined downward at 10 degrees is about 13.5 dB, and the grating lobe of the radiating element of the 0.91 ⁇ array inclined downward at 10 degrees is about 7.5 dB.
  • the inventors of the present disclosure innovatively thought of a method to improve the grating lobes of the high-frequency radiation elements, that is, to allow the spacing between the radiation elements of the high-frequency array to be free (usually less than 0.7 wavelengths). This can only be achieved by placing the radiating elements of the high-band array side by side with the radiating elements of the low-band array.
  • the radiating elements of conventional low-band elements are not hidden at high-band frequencies, if the radiating elements of the high-band array are very close to the radiating elements of the low-band array, the modes in the high-band will be severely distorted.
  • the radiating elements of the high-band array must be placed side by side with the radiating elements of the low-band array and kept close to the radiating elements of the low-band array. greater distance. This limits the possibility of having multiple radiating elements of the high-band array and radiating elements of the low-band array in a narrow-width antenna.
  • the radiating elements of the low-band array are invisible to the high-band frequencies, which provides flexibility for the radiating elements of the high-band array and the radiating elements of the low-band array to be interleaved. Furthermore, there is no restriction on the spacing between the radiating elements of the high-band array, which provides degrees of freedom for optimal grating lobe and gain performance.
  • FIG. 2A shows a dual-polarized radiating unit 210 for an antenna according to an embodiment of the present disclosure.
  • FIG. 2B shows a partially enlarged schematic diagram of a common mode choke circuit for a dual-polarized radiating element 210 of an antenna according to an embodiment of the present disclosure
  • FIG. 2C shows an implementation of the present disclosure
  • the dual-polarized radiating unit 210 for an antenna proposed according to the present disclosure includes four dipoles, that is, a pair of dipoles is formed by a diagonal line in FIG. 2A , for example along
  • the two poles at one end of the diagonal 211 form a dipole
  • the two dipoles at both ends of the diagonal 211 form a pair of dipoles
  • the two poles at one end along the diagonal 213 One dipole is formed, and the two dipoles at opposite ends of the diagonal 213 form a pair of dipoles.
  • the four dipoles jointly form two pairs of dipoles along the two diagonals 211 and 213, wherein the radiation arms of the four dipoles (that is, the most extended gray region) is a planar structure and is axisymmetric about two mutually perpendicular lines 211 and 213, wherein the two mutually perpendicular lines 211 and 213 divide the dual-polarized radiation unit 210 into four regions equally, and the The central parts of the four regions are DC-conductive, that is, the four regions are DC-conductive at the crossing regions of the two diagonal lines 211 and 213, rather than being hollow and disconnected, and wherein, in the four regions Each region of has a hollow area.
  • each region divided by the two diagonal lines 211 and 213 into the dual-polarized radiation unit 210 has a blank hollow region in the illustration. Since the dual-polarized radiation unit 210 is provided with the hollow area, high-frequency signals are allowed to pass through, so that the dual-polarized radiation unit 210 shields less high-frequency oscillators and minimizes diffraction.
  • a dual-polarized (eg +45°, -45°) radiating element 210 operating at a low frequency band (sub-band 1) is designed to be sensitive to high Frequency radiating elements are not visible.
  • the dual polarization radiating element 210 has inherently good radiation performance at the return loss bandwidth over sub-band 1 .
  • the dual-polarized radiating unit 210 can be realized by PCB, for example, but it can also be formed by using a metal plate or a die-casting plate, because the main body of the low-frequency dual-polarized radiating unit 210 is DC conducting, which means that the low-frequency
  • the dual-polarized radiation unit 210 can be integrally formed by metal casting or sheet metal processing.
  • the dielectric plate is omitted in FIG. 2 , that is, the microstrip transmission line 214 and the microstrip transmission line 216 on one plane are shown, and the plane 223 where the dipoles represented by the gray area are located is on the other plane.
  • the two planes can be parallel to each other, and the dielectric plate can, for example, be located between the two parallel planes.
  • the dual-polarized radiating unit 210 proposed according to the present disclosure has a simpler and more convenient manufacturing process because its central part is conducted in direct current, and the dual-polarized radiating unit 210 according to the present disclosure can be integrally constructed , so its product consistency is better, and its radiation performance can be better ensured.
  • the dual-polarized radiating unit 210 proposed according to the present disclosure has a hollow area, so that its interference with electromagnetic wave signals generated by radiating units used in other frequency bands is reduced, so that the dual-polarized radiation unit 210 proposed according to the present disclosure
  • the polarized radiation unit 210 is more friendly to system integration, and the radiation performance of the integrated multi-band radiation unit will be better than that of the radiation unit in the prior art.
  • the size of the slot 212 is reduced by about 1/4 of the high-frequency wavelength, for example, by edge chamfering.
  • the high-band radiating element is shielded by the low-band radiating element (such as the dual-polarized radiating element 210 here)
  • a current loop at the high-band frequency will be established at the slot 212 and the edge of the slot 212 .
  • the resonant frequency of the current loop can be shifted higher until the resonance exceeds the operating range of the antenna.
  • other features of the geometry were tuned to maintain impedance bandwidth (return loss better than 12dB) at low-band frequencies.
  • the first side (such as 211a or 211b) of the dual-polarized radiation unit 210 is perpendicular to the corresponding line (such as the diagonal line 211) in the two mutually perpendicular lines; and the The other side (such as 213a or 213b) of the dual-polarized radiation unit 210 is perpendicular to the other corresponding line (such as the diagonal line 213) of the two mutually perpendicular lines, wherein the first side (such as 211a Or 211b) is the edge that intersects said corresponding line (eg diagonal 211).
  • the two mutually perpendicular lines are usually diagonal lines, that is to say, the edge of the dual-polarized radiation unit 110 to which the diagonal line extends is usually the dual-polarized radiation unit A corner of 110, not a side.
  • the edge of the dual-polarization radiation unit 210 to which the two mutually perpendicular lines 211 and 213 of the dual-polarization radiation unit 210 proposed according to the present disclosure extend is a side of the dual-polarization radiation unit 210 , and the side It is perpendicular to the corresponding one of the two mutually perpendicular lines 211 and 213 .
  • the dual-polarized radiation unit 210 adopts a corner cutting method, so that the dual-polarized radiation proposed according to the present disclosure Compared with the traditional dual-polarized radiation unit 110 , the area of the unit 210 is smaller, thereby saving materials and further reducing manufacturing costs.
  • the dual-polarized radiation unit 210 has an octagonal shape.
  • the structure of the radiation unit 210 is composed of a conductive plate with four slots 212 on the diagonals 211 and 213 .
  • Each slot 212 takes the form of a slot, where the two slots on the diagonal 213 point to the +45° vector and the two slots on the diagonal 211 point to the -45° vector, the two polarizations together constitute The intended polarization direction of the dual-polarized radiating element 210 .
  • each slot uses a microstrip transmission line 214 and a corresponding microstrip transmission line 216 to feed in another slot to form the polarization of the antenna.
  • FIG. 1 the embodiment shown in FIG.
  • the feed point connected to the microstrip transmission line 214 produces +45° polarized radiation when viewed from the top, while the feed point connected to the microstrip transmission line 216 produces a -45° polarized radiation. ° of polarized radiation.
  • the position of the feed point along the slot 212 determines the impedance of the feed point, and the impedance of the circuit can be matched to 50 ohms by using an impedance transformation line and a power divider.
  • the two slots for each polarization are fed with equal amplitude and phase.
  • the support structure is shown in Figure 2C.
  • the current design is made of 2 interdigitated PCBs 215 and 222 each feeding polarization through a power splitter.
  • the output of the power divider is connected to the microstrip line on the radiating element 210 .
  • the feed structure includes an equal power divider and a section of impedance transformation line, the impedance transformation line can be matched to 50 ohms, or 75 ohms in other embodiments, or other values according to specific application requirements. These are realized by means of microstrip circuits, where the back of the feed structure also acts as a balun for the radiating element. It is not necessary to use a PCB for the feed structure and radiating elements, alternative versions can be realized using sheet metal or die-cast metal.
  • the dual-polarized radiation unit 210 includes four slots 212, and the four slots 212 are respectively located at the adjoining positions of two adjacent areas in the four areas.
  • Two of the four slots 212 (two slots on the same diagonal line) point to a polarization direction, for example, a +45° polarization direction, while the other two slots (on the other Two slots on a diagonal) point to another polarization direction, for example -45° polarization direction.
  • the size of the slot such as the length of the slot, is reduced by such as the aforementioned corner cutting, the resonance frequency of the current loop can be moved higher until the resonance exceeds the working range of the antenna, thereby reducing the impact of other frequency bands on the basis of the present disclosure.
  • the dual-polarized radiating unit 210 includes four feed lines 214 and 216, the four feed lines 214 and 216 are associated with the four slots 212, and each feed line in the four feed lines 214 and 216
  • the wire extends from the middle area of the dual-polarized radiation unit 210 (for example, the area where two diagonal lines 211 and 213 intersect) to the feeding point of the slot 212 associated therewith, wherein the feeding line 214 and the length of 216 are associated with matching impedance.
  • the dual-polarization radiating unit 210 also includes a common mode choke circuit 218, and the common mode choke circuit 218 is configured to be arranged near the feeding point and communicate with the One of the four gaps is associated.
  • the dual-polarized radiation unit 210 further includes a parallel filter 219 configured as a metal wire extending inward from the edge of the hollow region, preferably, these parallel Filter 219 is designed as a symmetrical structure.
  • a parallel filter 219 is added in the hollow area hollowed out inside.
  • the shunt filter 219 is connected to the body of the dual-polarized radiating unit 210, generating an impedance disturbance at the connection location.
  • the frequency at which impedance disturbance occurs is controlled by the electrical length of the parallel filter 219.
  • the electrical length is greater than ⁇ /6 and less than or equal to ⁇ /4, and the parallel filter 219 acts as a band-stop filter.
  • the location of the shunt filter 219 is optimized to suppress localized high-band currents flowing around the hollow hollow region of the hollow hollow. Since the lines of the parallel filter 219 are added to the continuous conductive track, the parallel filter 219 acts as a parallel circuit. Further preferably, the parallel filter 219 is configured as an open line.
  • the dual-polarization radiating unit 210 further includes an inductance element 217, and the inductance element 217 is connected from an edge (such as an outer edge) of the hollow area away from the central area of the dual-polarization radiating unit 210 extending toward the central region. Create a high impedance section at the outer edge. Due to the reduced size of the slots 212 on the diagonal, the resonant frequency of the low frequency radiating element (such as the dual polarized radiating element 210 ) will oscillate to a higher frequency than the designed low frequency band working center. In order to rebalance the resonant frequency in the low frequency band, a reactive part 217 is added. This feature increases the electrical length of the radiating element, although the diagonal dimension is smaller than ideal.
  • each hollow area may have an open-circuit parallel filter 219 and one inductance element 217 .
  • FIG. 2B shows a partially enlarged schematic diagram of the common mode choke circuit 218 used in the dual-polarized radiating unit 210 of the antenna according to an embodiment of the present disclosure.
  • the common mode choke circuit 218 further includes a first track 2181 and a second track 2182, wherein the first track 2181 and the second track 2182 are connected by an inductance coil
  • the forms are connected on the radiation arms parallel to each other and the coil winding directions of the first track 2181 and the second track 2182 are consistent. Since they are constructed on both sides of the radiation arms, there are multiple
  • the via hole is the via hole 2183 shown in FIG. 2B , and the electrical lengths of the first track 2181 and the second track 2182 are equal.
  • a common mode choke circuit 218, such as a common mode choke, is implemented on a balun, as shown in FIG. 2B.
  • High-band frequencies in the 1.4GHz to 2.7GHz range set a common mode at the low-band balun.
  • the low band currents in this region are differential, so the common mode choke circuit 218 is used to suppress the high band currents without adversely affecting the low band impedance.
  • Figure 2B shows a close-up of the structure.
  • Two conductive tracks 2181 and 2182 circulate parallel to each other on the two layers of the PCB. These two rails act as coupled inductors. When common-mode currents pass through the structure, the magnetic flux generated by each branch adds up, creating a large inductance.
  • the differential currents produce magnetic fluxes that cancel each other out, so the differential mode sees very little inductance.
  • the high frequency band current is effectively choked while the passing low frequency band current is relatively unaffected.
  • the upper and lower choke coils of the common-mode filter circuit 218 should be wound as tightly as possible to increase the common-mode choke effect and differential conduction mechanism.
  • the common mode choke circuit 218 has a length of at least 1/8 wavelength of the high frequency. The choke effect is more pronounced at longer lengths.
  • FIG. 2D shows a side perspective view of a dual-polarized radiating unit 210 for an antenna according to an embodiment of the present disclosure
  • FIG. 2E shows a dual-polarized radiation unit 210 for an antenna according to an embodiment of the present disclosure.
  • FIG. 2F shows an exploded view of a dual polarized radiating element 210 for an antenna according to one embodiment of the present disclosure.
  • the microstrip transmission line 214, the microstrip transmission line 216 and the common mode choke circuit 218 can be located on one side of the dielectric board 221, for example, and the dielectric board 221 can be connected by two PCB circuit boards 215 and 222. support.
  • the plane 223 where the dipole is located can be located on the other side of the dielectric plate 221 , for example.
  • the common mode choke circuit 218 includes a first track 2181 and a second track 2182 located on two different planes, wherein, for example, the first track 2181 can be located where the dipole is located.
  • the second track 2182 can be located in the plane where the microstrip transmission line 214 and the microstrip transmission line 216 lie, for example, and the two tracks 2181 and 2182 are connected through a via hole passing through the dielectric board 221 .
  • the two tracks 2181 and 2182 can be connected to the radiation arm in the form of an inductance coil, and the coil winding directions of the first track 2181 and the second track 2182 are consistent, and the first track 2181 and the second track 2181 The electrical lengths of the tracks 2182 are equal.
  • the second aspect of the present disclosure proposes an antenna, which includes at least one dual-polarized radiation unit 210 and a radiation unit matching circuit shown in FIG. 2A , FIG. 2B or FIG. 2C above. Since the dual-polarized radiating unit 210 proposed according to the present disclosure does not have to overlap with radiating units of other frequency bands, the height d of the multi-band antenna system formed by the dual-polarized radiating unit according to the present disclosure can be significantly smaller than that according to the The height of the multi-band antenna system constructed in the prior art.
  • a third aspect of the present disclosure provides an antenna system, the antenna system comprising: the antenna provided according to the second aspect of the present disclosure; and at least one second antenna, wherein the at least one first The working frequency band of the second antenna is higher than the working frequency band of the antenna.
  • the antennas and the second antennas are arranged alternately.
  • FIG. 3 shows a schematic diagram of an antenna system 300 according to one embodiment of the present disclosure. It can be seen from FIG.
  • the antenna system 300 proposed according to the present disclosure includes at least antennas for radiating radio frequency signals of two different frequency bands, wherein, for example, the first antenna 310 is configured for radiating signals of frequency band 1, namely The first antenna 310 is configured as a low-frequency antenna; correspondingly, for example, the second antenna 320 is configured to radiate signals in the frequency band 2, the frequency band 3 or the frequency band 4, that is, the second antenna 320 is configured as a high-frequency antenna.
  • the first antenna 310 and the second antenna 320 are arranged alternately, that is, for example, first arrange a row of low-frequency antennas 310, then arrange two rows of high-frequency antennas 320, and then arrange a row of low-frequency antennas 310, and so on.
  • the first antenna 310 at least partially covers the second antenna 320 to reduce the distance between two adjacent columns of high-frequency antennas 320 .
  • the number of columns of the antenna is equal to the number of columns of the second antenna, or the number of columns of the second antenna is equal to the number of columns of the antenna double.
  • FIGS. 4A and 4B show a schematic diagram of an antenna system 400A according to another embodiment of the present disclosure
  • FIG. 4B shows a schematic diagram of an antenna system 400B according to another embodiment of the present disclosure.
  • the antenna system 400a includes two columns of low-frequency antennas 410a and two columns of high-frequency antennas 420a, that is, the number of columns of the low-frequency antennas 410a and the number of columns of the high-frequency antennas 420a
  • the numbers are equal; differently, it can be seen from FIG.
  • the antenna system 400b includes two columns of low-frequency antennas 410b and four columns of high-frequency antennas 420b, that is to say, the number of columns of the low-frequency antennas 410a and The number of columns of the high-frequency antenna 420a is not equal, and the number of columns of the high-frequency antenna 420b is twice that of the low-frequency antenna 420a.
  • the low-frequency antenna 410b at least partially covers two rows of high-frequency antennas 420b, so as to reduce the distance between two adjacent rows of high-frequency antennas 420b.
  • the radiating elements of the low-band array and the radiating elements of the high-band array are placed very close to each other, causing the radiating elements of the low-band array to partially obstruct the radiating elements of the high-band array. Therefore, if the radiating elements of the low-band array are not designed to be transparent to the high-band, then the radiation pattern of the high-band will be distorted. Distortion is the result of two mechanisms: diffraction and resonance. Diffraction occurs when an electromagnetic wave interacts with an obstacle causing the wave to twist around the object. In this case, if the radiating element of the low-band array is a continuous piece of metal, it will cause a higher degree of diffraction. Diffraction that occurs in the near field of the radiating element can cause significant distortion of the far field pattern.
  • the radiating units of the low frequency band array When the radiation in the high frequency band excites the radiating units of the low frequency band array, the radiating units of the low frequency band array will locally resonate in the high frequency band.
  • Current is sensitive to the geometry of the local structure in terms of electrical length and impedance. These currents themselves cause unwanted radiation that distorts the high-band far-field pattern.
  • the radiating unit of the low-band array according to the present invention adopts various means to solve the above distortion mechanism.
  • the radiating unit of the low-band array has a hollow part in the structure to allow the high-band energy to pass through (the maximum dimension of the hollow part should be larger than 1/2 wavelength).
  • the radiation unit of the low frequency band according to the present disclosure for example, the dipole radiation unit according to the present disclosure is processed at the four corners, so that the radiation unit of the low frequency band is more sensitive to the high frequency Less shadowing with arrays.
  • High-impedance points are introduced at certain locations in the low-band radiating element geometry, thereby suppressing high-band resonances.
  • the high impedance is frequency selective and tuned to reject high frequency band currents without adversely affecting low frequency band currents.
  • the common mode choke coil is used to suppress the common mode resonance of the high frequency band at certain positions of the low frequency band radiating unit.
  • the first is a common-mode filter used on low-frequency oscillators.
  • the filter is composed of two tightly wound inductances in the same direction, which creates high reactance at high frequencies, and the resistance of the high reactance is used to reduce the high-frequency resonance generated on the low-frequency oscillator. It is also capable of conducting differential low frequency currents.
  • the filter circuit should be 1/8 wavelength longer than the choke frequency. The longer the effect, the more obvious.
  • the second type is a filter formed by a high-frequency 1/4 wavelength impedance line connected in parallel on a low-frequency vibrator. The low-frequency vibrator with the characteristics of the above two filter circuits and the hollow feature can reduce the diffraction of the high-frequency vibrator.
  • the dual-polarized radiation unit 210 proposed according to the present disclosure has a hollow area, so that it interferes with the electromagnetic wave signal generated by the radiation unit used for other frequency bands (such as the radiation unit 220 of the high frequency band) can be reduced, so that the dual-polarization radiating unit 210 proposed according to the present disclosure is more friendly to system integration, and the radiation performance of the integrated multi-band radiating unit will also be better than that of the radiating unit in the prior art better.
  • the central part of the dual-polarized radiation unit 210 proposed according to the present disclosure is conducted with direct current, its manufacturing process is simpler and more convenient, and the dual-polarized radiation unit 210 according to the present disclosure can be integrated structure, so its product consistency is better and its radiative performance can be better ensured.

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Abstract

本公开内容涉及用于天线的双极化辐射单元、天线以及天线系统。其中，依据本公开内容的用于天线的双极化辐射单元包括：四个偶极子，其中，所述四个偶极子的辐射臂为平面结构并且关于两条相互垂直的线轴对称，其中，所述两条相互垂直的线将所述辐射单元均分为四个区域，并且所述四个区域的中心部分直流导通，并且其中，所述四个区域中的每个区域均具有中空区域。

Description

用于天线的双极化辐射单元、天线以及天线系统 技术领域
本公开内容涉及射频通信技术领域，具体地，本公开内容涉及一种用于天线的双极化辐射单元、包括该用于天线的双极化辐射单元的天线以及包括该天线的天线系统。
背景技术
在400MHz到6GHz之间的广谱范围内，即所谓的6GHz以下频段的范围内，这些频谱范围均被分配给电信用于通信。但是从技术上来说，目前尚无法为如此宽的带宽设计模拟组件，例如滤波器、移相器、辐射单元以及放大器等。因此，6GHz以下频段会进一步划分为多个子频段并单独运行，以此实现模拟组件的设计。例如，通常业界将6GHz以下的频段划分为以下四个单独的工作子频段：600MHz至1GHz、1.4GHz至3GHz、3GHz至4.2GHz以及5GHz-6GHz。用于LTE、5G或者3G通信的基站天线由工作在不同频段的多个辐射单元阵列组成。
这些分离的频带需要单独的组件，例如滤波器、移相器、放大器、辐射单元。所有这些组件都不能相互干扰，隔离度最小需要在20dB左右，并且每个信号通道之间的隔离度最好为30dB。对于诸如滤波器、移相器之类的屏蔽通道来说相对容易达到这一目标，在这些通道中所有信号都被微带线或带状线屏蔽。但是，由于辐射单元辐射到空气中并且非常容易耦合，所以隔离辐射单元是相对困难的。如果隔离无法达到该级别，则将存在严重的模式失真和端口间隔离问题。这些问题将降低网络性能。如果辐射单元之间可以相隔多于两个最低频率处的波长，则隔离性可以得到增强，但是这样的相隔距离将会降低天线系统的集成度，增加天线系统的体积。
另一方面，由于需要覆盖多个频带、运营商和扇区，因此在基站塔上安装天线的空间非常有限。在过去的五至七年中，该行业正在朝着将多个子带系统组合到同一天线罩或产品中的方向发展，导致隔离成为挑战。
发明内容
如前所述，现有技术中存在以下技术问题，即现有的不同频段的辐射单元之间会形成干扰，而且由于低频辐射单元和高频辐射单元之间的干扰问题的存在，使得低频辐射单元和高频辐射单元之间存在固定的位置关系，不能相互解耦，从而限制了多频段辐射单元的设计。
为了解决以上的技术问题，本公开内容的发明人基于以上思考提出了如下的用于天线的双极化辐射单元，所述双极化辐射单元包括：四个偶极子，其中，所述四个偶极子的辐射臂为平面结构并且关于两条相互垂直的线轴对称，其中，所述两条相互垂直的线将所述辐射单元均分为四个区域，并且所述四个区域的中心部分直流导通，并且其中，所述四个区域中的每个区域均具有中空区域。
依据本公开内容所提出的双极化辐射单元具有中空区域，从而使得其对于用于其他频段的辐射单元所产生的电磁波信号的干扰得以降低，从而使得依据本公开内容所提出的双极化辐射单元对于系统集成更为友好，而且所集成产生的多频段辐射单元的辐射性能也将比现有技术中的辐射单元的辐射性能更优。此外，依据本公开内容所提出的双极化辐射单元由于其中心部分是直流导通的，所以其制造过程更为简单方便，而且依据本公开内容的双极化辐射单元由于能够一体化进行构造，所以其产品一致性更好，而且其辐射性能也能够更好地得以确保。
在依据本公开内容的一个实施例之中，所述双极化辐射单元的第一边与所述两条相互垂直的线中相应的线垂直，其中，所述第一边是与所述相应的线相交叉的边。在传统的双极化辐射单元之中，其两条相互垂直的线通常为对角线，也就是说对角线延伸至的双极化辐射单元的边缘通常为该双极化辐射单元的一个角，而非边。在依据本公开内容所提出的双极化辐射单元的两条相互垂直的线延伸至的双极化辐射单元的边缘为该双极化辐射单元的一个边，而且该边与两条相互垂直的线中相应的线垂直。形象地说，相较于传统的双极化辐射单元来说，依据本公开内容的双极化辐射单元采取了切角的处理方式，从而使得依据本公开内容所提出的双极化辐射单元相较于传统的双极化辐射单元的面积更小，从而更为节省材料，进而 降低制造成本。
优选地，在依据本公开内容的一个实施例之中，所述双极化辐射单元具有八边形的形状。相较于传统的双极化辐射单元具有例如四边形的形状而言，依据本公开内容的双极化辐射单元采取了切角的处理方式，从而使得依据本公开内容所提出的双极化辐射单元例如具有八边形的形状。相较于传统的双极化辐射单元的面积更小，从而更为节省材料，进而降低制造成本。
在依据本公开内容的一个实施例之中，所述双极化辐射单元还包括：四条缝隙，所述四条缝隙分别位于所述四个区域中的两个相邻的区域的邻接处。这四条缝隙中的两条缝隙指向一个极化方向，例如+45°的极化方向，而这四条缝隙中的另外两条缝隙指向另一个极化方向，例如-45°的极化方向。通过诸如前述的切角处理虽然缩小了缝隙的尺寸，例如缝隙的长度，电流环路的谐振频率可以移得更高，直到谐振超出天线的工作范围，由此能够降低其他频段对于依据本公开内容所提出的双极化辐射单元的影响。
在依据本公开内容的一个实施例之中，所述双极化辐射单元还包括：四条馈电线，所述四条馈电线与所述四条缝隙相关联，并且所述四条馈电线中的每条馈电线从所述双极化辐射单元的中间区域延伸至与之相关联的缝隙的馈电点处，其中，所述馈电线的长度与匹配阻抗相关联。
优选地，在依据本公开内容的一个实施例之中，所述双极化辐射单元还包括：共模扼流电路，所述共模扼流电路被构造为设置在所述馈电点附近并且与所述四条缝隙中的一条缝隙相关联。进一步优选地，在依据本公开内容的一个实施例之中，所述共模扼流电路进一步包括第一轨道和第二轨道，其中，所述第一轨道和所述第二轨道被以电感线圈的形式相互平行地构造在所述辐射臂的两侧上，并且其中，所述第一轨道和所述第二轨道的电长度相等，并且所述第一轨道和所述第二轨道的线圈绕行方向一致。
可选地或者附加地，在依据本公开内容的一个实施例之中，所述双极化辐射单元还包括：并联滤波器，所述并联滤波器被构造为从所述中空区域的边缘向内延伸的金属丝。优选地，在依据本公开内容的一个实施例之中，所述并联滤波器被构造为开路线。
在依据本公开内容的一个实施例之中，所述双极化辐射单元还包括： 电感元件，所述电感元件从所述中空区域的远离所述双极化辐射单元的中心区域的边缘朝向所述中心区域延伸。
此外，本公开内容的第二方面提出了一种天线，所述天线包括：至少一个根据本公开内容的第一方面所述的双极化辐射单元；以及辐射单元匹配电路。
再者，本公开内容的第三方面提出了一种天线系统，所述天线系统包括：根据本公开内容的第二方面所提出的天线；以及至少一个第二天线，其中，所述至少一个第二天线的工作频段比所述天线的工作频段更高。
在依据本公开内容的一个实施例之中，所述天线与所述第二天线交错设置。可选地，在依据本公开内容的一个实施例之中，所述天线的列数和所述第二天线的列数相等，或者所述第二天线的列数是所述天线的列数的两倍。
综上所述，依据本公开内容所提出的双极化辐射单元具有中空区域，从而使得其对于用于其他频段的辐射单元所产生的电磁波信号的干扰得以降低，从而使得依据本公开内容所提出的双极化辐射单元对于系统集成更为友好，而且所集成产生的多频段辐射单元的辐射性能也将比现有技术中的辐射单元的辐射性能更优。此外，依据本公开内容所提出的双极化辐射单元由于其中心部分是直流导通的，所以其制造过程更为简单方便，而且依据本公开内容的双极化辐射单元由于能够一体化进行构造，所以其产品一致性更好，而且其辐射性能也能够更好地得以确保。
附图说明
结合附图并参考以下详细说明，本公开内容的各实施例的特征、优点及其他方面将变得更加明显，在此以示例性而非限制性的方式示出了本公开内容的若干实施例，在附图中：
图1A示出了根据现有技术的一个实施例的包括双极化辐射单元110与高频辐射单元120的天线系统100的立体图；
图1B示出了根据现有技术的一个实施例的包括双极化辐射单元110与高频辐射单元120的天线系统100的侧视图；
图2A示出了根据本公开内容的一个实施例的用于天线的双极化辐射 单元210的示意图；
图2B示出了根据本公开内容的一个实施例的用于天线的双极化辐射单元210的共模扼流电路的局部放大示意图；
图2C示出了根据本公开内容的一个实施例的用于天线的双极化辐射单元210的侧视图；
图2D示出了根据本公开内容的一个实施例的用于天线的双极化辐射单元210的侧视立体图；
图2E示出了根据本公开内容的一个实施例的用于天线的双极化辐射单元210的仰视立体图；
图2F示出了根据本公开内容的一个实施例的用于天线的双极化辐射单元210的分解视图；
图3示出了根据本公开内容的一个实施例的天线系统300的示意图；
图4A示出了根据本公开内容的另一个实施例的天线系统400A的示意图；以及
图4B示出了根据本公开内容的另一个实施例的天线系统400B的示意图。
具体实施方式
以下参考附图详细描述本公开内容的各个示例性实施例。虽然以下所描述的示例性方法、装置包括在其它组件当中的硬件上执行的软件和/或固件，但是应当注意，这些示例仅仅是说明性的，而不应看作是限制性的。例如，考虑在硬件中独占地、在软件中独占地、或在硬件和软件的任何组合中可以实施任何或所有硬件、软件和固件组件。因此，虽然以下已经描述了示例性的方法和装置，但是本领域的技术人员应容易理解，所提供的示例并不用于限制用于实现这些方法和装置的方式。
此外，附图中的流程图和框图示出了根据本公开内容的各种实施例的方法和系统的可能实现的体系架构、功能和操作。应当注意，方框中所标注的功能也可以按照不同于附图中所标注的顺序发生。例如，两个接连地表示的方框实际上可以基本并行地执行，或者它们有时也可以按照相反的顺序执行，这取决于所涉及的功能。同样应当注意的是，流程图和/或框图 中的每个方框、以及流程图和/或框图中的方框的组合，可以使用执行规定的功能或操作的专用的基于硬件的系统来实现，或者可以使用专用硬件与计算机指令的组合来实现。
如前所述，现有技术中存在以下技术问题，即现有的不同频段的辐射单元之间会形成干扰。本发明的目的是提供一种工作在1GHz以下的双极化低频带辐射单元或元件，即使两个或三个不同的元件在很大程度上重叠(高度在不同维度上)，也能很好地隔离其他高频带的干扰(如上所述的子频带2、3、4)，使得高频方向图保持不变，就像顶部没有低频单元一样。此外，对低频带单元自身的性能(例如辐射方向图和回波损耗)影响被最小化。
WO2015/124573A公开了一种类似的辐射单元的形式。图1A示出了根据现有技术的一个实施例的包括双极化辐射单元110与高频辐射单元120的天线系统100的立体图，而图1B示出了根据现有技术的一个实施例的包括双极化辐射单元110与高频辐射单元120的天线系统100的侧视图。从图1的两张附图之中可以看出，WO2015/124573A公开的天线系统100概述了采用四槽馈电方法构造的低频带辐射单元110。它进一步概述了一种通过将高频带辐射单元120放置在低频带元件110的顶部，并用一个附加金属片130作为高频带元件120的反射器的方法来获得多频带的天线系统100的方法。这样，高频带阵列120将与低频带阵列110成一直线，其中一半的高频单元120在低频单元110的顶部，而另一半的高频单元120在低频单元110之间。在这种天线系统100的辐射单元110和辐射单元120的设计方式中，其设计人没有试图使高频带辐射单元120看不见低频带辐射单元110，而是以使低频带辐射单元110不会使高频带辐射单元120模糊的方式来布置高频带辐射单元120。但是，此方法有一些缺点：
首先，高频段阵列的辐射单元120的垂直单元间距被限制为低频段阵列的辐射单元110间距的一半。低频带阵列的的辐射单元110的间距通常约为0.7λ，可在较大下倾斜角度达到合理的栅瓣水平，其中，栅瓣是一个随扫描波束的角度加大而增加的旁瓣，与辐射元件之间的间距与阵列的辐射单元的波长之比成正比例。较大的辐射元件之间的间距会导致较高的栅瓣。栅瓣会减小阵列的辐射单元的增益，并且无法通过调整阵列的辐射单 元的幅度或相位锥度来减小。
其次，由于高频的频谱更宽，因此对频带内较高频率处的栅瓣影响更敏感。工业上使用的实际的辐射单元之间的间距为0.78λ至0.85λ。如果高频带的辐射单元之间的间距是低频带的辐射单元之间的间距的一半，则大于高频的0.9λ，从而导致显著的栅瓣。向下倾斜10度的0.78λ的阵列的辐射单元的栅瓣约为13.5dB，而向下倾斜10度的0.91λ的阵列的辐射单元的栅瓣约为7.5dB。
针对以上问题，本公开内容的发明人创新地想到改善高频段的辐射单元的栅瓣的方法，即允许高频段阵列的辐射单元之间的间距自由化(通常小于0.7波长)。这只能通过将高频段阵列的辐射单元与低频段阵列的辐射单元并排放置来实现。然而，由于传统的低频带元件的辐射单元在高频带频率处没有隐藏，如果高频带阵列的辐射单元非常接近低频带阵列的辐射单元，则高频带的模式会严重失真。
要在常规低频带阵列的辐射单元下获得良好的高频垂直面和水平面方向图性能，高频带阵列的辐射单元必须与低频带阵列的辐射单元并排放置，并与低频带阵列的辐射单元保持较大距离。这限制了在窄宽度天线中具有多个高频带阵列的辐射单元和低频带阵列的辐射单元的可能性。
在依据本公开内容的设计中，低频带阵列的辐射单元对高频段频率不可见，这为高频段阵列的辐射单元与低频带阵列的辐射单元交错放置提供了灵活性。此外，对高频带阵列的辐射单元之间的间距没有限制，这为获得最佳栅瓣和增益性能提供了自由度。
以下结合图2A和图2B来介绍依据本公开内容所提出的双极化辐射单元210，其中，图2A示出了根据本公开内容的一个实施例的用于天线的双极化辐射单元210的示意图，图2B示出了根据本公开内容的一个实施例的用于天线的双极化辐射单元210的共模扼流电路的局部放大示意图，而图2C示出了根据本公开内容的一个实施例的用于天线的双极化辐射单元210的侧视图。从图2A之中可以看出：依据本公开内容所提出的用于天线的双极化辐射单元210包括四个偶极子，即图2A中一个对角线构成一对偶极子，例如沿着对角线211的一端的两个极子形成一个偶极子，而对角线211的两端的两个偶极子形成一对偶极子，而沿着对角线213的一端的两个极子 形成一个偶极子，对角线213的两端的两个偶极子形成一对偶极子。换句话说，四个偶极子沿着两条对角线211和对角线213共同形成了两对偶极子，其中，所述四个偶极子的辐射臂(即图中最外延的灰色区域)为平面结构并且关于两条相互垂直的线211和213轴对称，其中，所述两条相互垂直的线211和213将所述双极化辐射单元210均分为四个区域，并且所述四个区域的中心部分直流导通，即这四个区域在两条对角线211和213的交叉区域是直流导通的，而非空心不连接的，并且其中，所述四个区域中的每个区域均具有中空区域。即两条对角线211和213将所述双极化辐射单元210所分成的每个区域的中间均具有图示中为空白的中空区域。由于所述双极化辐射单元210设置有所述中空区域，从而允许高频信号通过，进而使得所述双极化辐射单元210对高频振子的遮蔽较少，最大限度地减少了衍射。换句话说，在低频段(子频段1)下工作的双极化(例如+45°，-45°)辐射单元210被设计为对工作在高频段(子频段2、3、4)的高频辐射单元不可见。双极化辐射单元210在子频段1上的回波损耗带宽下具有固有的良好辐射性能。此外，双极化辐射单元210例如能够通过PCB实现，但也可采用金属板或压铸板来形成，因为低频段的双极化辐射单元210的主体是直流导通的，这意味着低频段的双极化辐射单元210可以通过金属铸造或者钣金加工一体化成型。
在此，在图2中缺省了介质板，即示出了位于一个平面上的微带传输线214和微带传输线216，而灰色区域所表示的偶极子所在的平面223在另一个平面之上，这两个平面可以相互平行，而且介质板例如可以位于这两个平行的平面之间。
依据本公开内容所提出的双极化辐射单元210由于其中心部分是直流导通的，所以其制造过程更为简单方便，而且依据本公开内容的双极化辐射单元210由于能够一体化进行构造，所以其产品一致性更好，而且其辐射性能也能够更好地得以确保。此外，依据本公开内容所提出的双极化辐射单元210具有中空区域，从而使得其对于用于其他频段的辐射单元所产生的电磁波信号的干扰得以降低，从而使得依据本公开内容所提出的双极化辐射单元210对于系统集成更为友好，而且所集成产生的多频段辐射单元的辐射性能也将比现有技术中的辐射单元的辐射性能更优。与传统设计 相比，槽212的尺寸例如通过边沿切角的方式减少了约1/4高频波长。当高频带辐射单元被低频带辐射单元(例如此处的双极化辐射单元210)遮挡时，会在槽212和槽212的边缘处建立高频带频率的电流环路。通过缩小诸如缝隙的槽212的尺寸，电流环路的谐振频率可以移得更高，直到谐振超出天线的工作范围。尽管减小了缝隙尺寸，但仍对几何形状的其他特征进行了调整，以在低频带频率下保持阻抗带宽(回波损耗优于12dB)。
从图2A可以看出：所述双极化辐射单元210的第一边(例如211a或者211b)与所述两条相互垂直的线中相应的线(例如对角线211)垂直；而所述双极化辐射单元210的另一条边(例如213a或者213b)与所述两条相互垂直的线中另外一条相应的线(例如对角线213)垂直，其中，所述第一边(例如211a或者211b)是与所述相应的线(例如对角线211)相交叉的边。在传统的双极化辐射单元110之中，其两条相互垂直的线通常为对角线，也就是说对角线延伸至的双极化辐射单元110的边缘通常为该双极化辐射单元110的一个角，而非边。在依据本公开内容所提出的双极化辐射单元210的两条相互垂直的线211和213延伸至的双极化辐射单元210的边缘为该双极化辐射单元210的一个边，而且该边与两条相互垂直的线211和213中相应的线垂直。形象地说，相较于传统的双极化辐射单元110来说，依据本公开内容的双极化辐射单元210采取了切角的处理方式，从而使得依据本公开内容所提出的双极化辐射单元210相较于传统的双极化辐射单元110的面积更小，从而更为节省材料，进而降低制造成本。优选地，如图2A所示，所述双极化辐射单元210具有八边形的形状。
具体而言，该辐射单元210的结构由在对角线211和213上具有四个槽212的导电板组成。每个槽212均表现为缝隙的形式，其中，对角线213上的两个缝隙指向+45°向量，而对角线211上的两个缝隙指向-45°向量，两个极化一起构成该双极化辐射单元210的预期极化方向。如图2A所示，每个缝隙使用一个微带传输线214与对应的微带传输线216在另一个缝隙馈电组合成天线的极化。在图2A所示出的实施例之中，从顶部看微带传输线214所连接的馈电点会产生+45°的极化辐射，而微带传输线216所连接的馈电点会产生-45°的极化辐射。沿着槽212的馈电点位置决定了馈电点的阻抗，利用阻抗变换线和功率分配器可以使得电路的阻抗匹配到50欧姆。每 个极化的两个缝隙以相等的幅度和相位馈电。支撑结构如图2C所示。当前设计由2个相互交叉的PCB 215和222制成，每个PCB 215和222通过功率分配器馈入极化。功率分配器的输出连接到辐射单元210上的微带线路。馈电结构包括一个等功率分配器和一截阻抗变换线，该阻抗变换线能够例如匹配至50欧姆，或者在其他实施例中匹配至75欧姆或者根据具体应用需要匹配至其他值。这些通过微带电路实现，其中馈电结构背面还充当辐射单元的平衡-不平衡转换器。馈电结构和辐射单元不是必须使用PCB，也可以使用金属板或压铸金属来实现替代版本。
概括地讲，所述双极化辐射单元210包括四条缝隙212，所述四条缝隙212分别位于所述四个区域中的两个相邻的区域的邻接处。这四条缝隙212中的两条缝隙(在同一条对角线上的两条缝隙)指向一个极化方向，例如+45°的极化方向，而这四条缝隙中的另外两条缝隙(在另一条对角线上的两条缝隙)指向另一个极化方向，例如-45°的极化方向。通过诸如前述的切角处理虽然缩小了缝隙的尺寸，例如缝隙的长度，电流环路的谐振频率可以移得更高，直到谐振超出天线的工作范围，由此能够降低其他频段对于依据本公开内容所提出的双极化辐射单元的影响。优选地，所述双极化辐射单元210包括四条馈电线214和216，所述四条馈电线214和216与所述四条缝隙212相关联，并且所述四条馈电线214和216中的每条馈电线从所述双极化辐射单元210的中间区域(例如两条对角线211和213相交叉的区域)延伸至与之相关联的缝隙212的馈电点处，其中，所述馈电线214和216的长度与匹配阻抗相关联。
此外，从图2A还可以看出，所述双极化辐射单元210还包括共模扼流电路218，所述共模扼流电路218被构造为设置在所述馈电点附近并且与所述四条缝隙中的一条缝隙相关联。可选地或者附加地，所述双极化辐射单元210还包括并联滤波器219，所述并联滤波器219被构造为从所述中空区域的边缘向内延伸的金属丝，优选地，这些并联滤波器219被构造为对称的结构。内部镂空的中空区域中添加了并联滤波器219。该并联滤波器219连接到双极化辐射单元210的主体，在连接位置处产生阻抗扰动。出现阻抗扰动的频率由并联滤波器219的电长度控制通常电长度大于λ/6小于等于λ/4，并联滤波器219充当带阻滤波器。并联滤波器219的位置经过优化以 抑制围绕空心镂空的中空区域流动的局部高频带电流。由于并联滤波器219的线路添加到连续的导电轨道，因此并联滤波器219充当并联电路。进一步优选地，所述并联滤波器219被构造为开路线。
此外，可选地，所述双极化辐射单元210还包括电感元件217，所述电感元件217从所述中空区域的远离所述双极化辐射单元210的中心区域的边缘(例如外边缘)朝向所述中心区域延伸。在外边缘处创建一个高阻抗部分。由于对角线上的槽212的尺寸缩小，因此低频辐射单元(例如双极化辐射单元210)的谐振频率会振荡到比所设计的低频带工作中心的频率更高的频率。为了重新平衡低频段的谐振频率，增加了电抗部分217。尽管对角线尺寸小于理想尺寸，但此功能增加了辐射单元的电长度。
在此，虽然只标注了一个开路的并联滤波器219和一个电感元件217，本领域的技术人员应当了解，每个中空区域均可以具有一个开路的并联滤波器219和一个电感元件217。
再者，图2B示出了根据本公开内容的一个实施例的用于天线的双极化辐射单元210的共模扼流电路218的局部放大示意图。由图2B可以看出：优选地，所述共模扼流电路218进一步包括第一轨道2181和第二轨道2182，其中，所述第一轨道2181和所述第二轨道2182被以电感线圈的形式相互平行地连接在所述辐射臂上并且所述第一轨道2181和所述第二轨道2182的线圈绕行方向一致，由于被构造在辐射臂的两侧，所以在辐射臂上存在多个过孔，即图2B中所示出的过孔2183，并且其中，所述第一轨道2181和所述第二轨道2182的电长度相等。诸如共模扼流圈的共模扼流电路218在巴伦上实现，如图2B所示。1.4GHz至2.7GHz范围内的高频段频率在低频段的巴伦处设置了共模。该区域的低频带电流是差分的，因此使用共模扼流电路218来抑制高频带电流而不会对低频带阻抗产生不利影响。图2B示出了该结构的特写。两个导电轨道2181和2182在PCB的两层上相互平行地循环。这两条轨道充当耦合电感器。当共模电流通过该结构时，每个分支产生的磁通量相加，从而产生大的电感。差动电流产生相互抵消的磁通量，因此差模看到的电感非常小。因此，高频带电流被有效扼流，而通过的低频带电流相对不受影响。共模滤波电路218上下两层扼流圈应尽量紧密缠绕，以增加共模扼流效果和差分导通机制。共模扼流电路218长 度至少为高频的1/8波长。长度更长时，扼流效果更加明显。
为了更为形象地示出依据本公开内容的双极化辐射单元210。其中，图2D示出了根据本公开内容的一个实施例的用于天线的双极化辐射单元210的侧视立体图，图2E示出了根据本公开内容的一个实施例的用于天线的双极化辐射单元210的仰视立体图，而图2F示出了根据本公开内容的一个实施例的用于天线的双极化辐射单元210的分解视图。
从图2D之中可以看出，微带传输线214和微带传输线216以及共模扼流电路218能够例如位于介质板221的一侧，该介质板221能够通过两个PCB电路板215和222进行支撑。而从图2E之中可以看出，偶极子所在的平面223能够例如位于介质板221的另一侧。从图2F之中可以看出，共模扼流电路218包括位于两个不同的平面上的第一轨道2181和第二轨道2182，其中，所述第一轨道2181例如能够位于偶极子所在的平面223之中，而第二轨道2182例如能够位于微带传输线214和微带传输线216所处于的平面之中，这两个轨道2181和2182通过穿过介质板221的过孔进行连接。这两个轨道2181和2182例如能够以电感线圈的形式连接在所述辐射臂上并且所述第一轨道2181和所述第二轨道2182的线圈绕行方向一致，并且第一轨道2181和第二轨道2182的电长度相等。
此外，本公开内容的第二方面提出了一种天线，所述天线包括至少一个根据以上图2A、图2B或者图2C所示出的双极化辐射单元210以及辐射单元匹配电路。由于依据本公开内容所提出的双极化辐射单元210并非必须与其他频段的辐射单元叠置，所以依据本公开内容的双极化辐射单元所构成的多频段天线系统的高度d可以明显小于依据现有技术所构建的多频段天线系统的高度。
再者，本公开内容的第三方面提出了一种天线系统，所述天线系统包括：根据本公开内容的第二方面所提出的天线；以及至少一个第二天线，其中，所述至少一个第二天线的工作频段比所述天线的工作频段更高。优选地，所述天线与所述第二天线交错设置。以下借助于图3来描述依据本公开内容所提出的天线系统。图3示出了根据本公开内容的一个实施例的天线系统300的示意图。从图3可以看出，依据本公开内容所提出的天线系统300至少包括用于辐射两个不同频段的射频信号的天线，其中，例如 第一天线310被构造用于辐射频段1的信号，即第一天线310被构造为低频天线；与之相对应地，例如第二天线320被构造用于辐射频段2、频段3或者频段4的信号，即第二天线320被构造为高频天线。在此，第一天线310和第二天线320是交错设置的，即先例如布置一列低频天线310，然后布置两列高频天线320，然后再布置一列低频天线310，以此类推地进行布置。优选地，所述第一天线310至少部分地覆盖所述第二天线320，以降低相邻的两列高频天线320之间的距离。
可选地，在依据本公开内容的一个实施例之中，所述天线的列数和所述第二天线的列数相等，或者所述第二天线的列数是所述天线的列数的两倍。以下借助于图4A和图4B来描述借助于依据本公开内容的双极化辐射单元210来构建多频段的天线系统。图4A示出了根据本公开内容的另一个实施例的天线系统400A的示意图，而图4B示出了根据本公开内容的另一个实施例的天线系统400B的示意图。
从图4A可以看出，依据本公开内容的天线系统400a包括两列低频天线410a和两列高频天线420a，也就是说，所述低频天线410a的列数和所述高频天线420a的列数相等；与之不同地，从图4B可以看出，依据本公开内容的天线系统400b包括两列低频天线410b和四列高频天线420b，也就是说，所述低频天线410a的列数和所述高频天线420a的列数不相等，并且高频天线420b的列数是低频天线420a的列数的两倍。优选地，所述低频天线410b至少部分覆盖两列高频天线420b，以缩小相邻的两列高频天线420b之间的间距。
在紧凑型的多频段天线之中，低频段阵列的辐射单元和高频段阵列的辐射单元放置得很近，从而导致低频段阵列的辐射单元部分阻碍高频段阵列的辐射单元。因此，如果低频段阵列的辐射单元没有设计为对于高频段是透明的，那么会导致高频段的辐射模式失真。失真是两种机制的结果：衍射和谐振。当电磁波与障碍物相互作用导致电磁波在物体周围扭曲时，就会发生衍射。在这种情况下，如果低频段阵列的辐射单元是一块连续的金属，它会引起较高程度的衍射。发生在辐射单元近场的衍射会导致远场方向图的显着失真。
当高频带的辐射激发低频带阵列的辐射单元时，低频段阵列的辐射单 元在局部会发生高频段的谐振。在电长度和阻抗方面，电流对局部结构的几何形状很敏感。这些电流本身会导致不需要的辐射，从而导致高频带远场方向图失真。
依据本发明的低频段阵列的辐射单元采用多种手段来解决以上失真机制。为了最大限度地减少衍射，低频带阵列的辐射单元在结构中具有中空部分，允许高频带能量通过(中空部分最大尺寸应大于1/2波长)。与传统的辐射单元相比，依据本公开内容的低频带的辐射单元，例如依据本公开内容的双极子辐射单元在四个角处进行了切角处理，从而使低频带辐射单元对高频带阵列的遮蔽更少。
为了解决谐振问题，使用了以下几种方法。在低频段辐射单元几何结构的某些位置引入了高阻抗点，从而抑制了高频段共振。高阻抗是频率选择性的，经过调整以抑制高频带电流，同时不会对低频带电流产生不利影响。共模扼流圈用于抑制低频段辐射单元某些位置的高频段共模谐振。第一种是用在低频振子上的共模滤波器。该滤波器通过两条紧密缠绕的同方向电感构成，在高频率处制造高电抗，高电抗的阻值用来减少低频振子上产生的高频谐振。而且还能够导通差分低频电流。该滤波电路要长于扼流频率1/8波长。越长效果越明显。第二种是由并联在低频振子上高频1/4波长阻抗线形成的滤波器。具有以上两种滤波电路的特征具有镂空特征的低频振子能够减少对高频振子的衍射。
综上所述，依据本公开内容所提出的双极化辐射单元210具有中空区域，从而使得其对于用于其他频段的辐射单元(例如高频带的辐射单元220)所产生的电磁波信号的干扰得以降低，从而使得依据本公开内容所提出的双极化辐射单元210对于系统集成更为友好，而且所集成产生的多频段辐射单元的辐射性能也将比现有技术中的辐射单元的辐射性能更优。此外，依据本公开内容所提出的双极化辐射单元210由于其中心部分是直流导通的，所以其制造过程更为简单方便，而且依据本公开内容的双极化辐射单元210由于能够一体化进行构造，所以其产品一致性更好，而且其辐射性能也能够更好地得以确保。
以上所述仅为本公开内容的实施例可选实施例，并不用于限制本公开内容的实施例，对于本领域的技术人员来说，本公开内容的实施例可以有 各种更改和变化。凡在本公开内容的实施例的精神和原则之内，所作的任何修改、等效替换、改进等，均应包含在本公开内容的实施例的保护范围之内。
虽然已经参考若干具体实施例描述了本公开内容的实施例，但是应当理解，本公开内容的实施例并不限于所公开的具体实施例。本公开内容的实施例旨在涵盖在所附权利要求的精神和范围内所包括的各种修改和等同布置。权利要求的范围符合最宽泛的解释，从而包含所有这样的修改及等同结构和功能。

Claims (14)

  1. 一种用于天线的双极化辐射单元，其特征在于，所述双极化辐射单元包括：
    四个偶极子，其中，所述四个偶极子的辐射臂为平面结构并且关于两条相互垂直的线轴对称，其中，所述两条相互垂直的线将所述辐射单元均分为四个区域，并且所述四个区域的中心部分直流导通，
    并且其中，所述四个区域中的每个区域均具有中空区域。
  2. 根据权利要求1所述的双极化辐射单元，其特征在于，所述双极化辐射单元的第一边与所述两条相互垂直的线中相应的线垂直，其中，所述第一边是与所述相应的线相交叉的边。
  3. 根据权利要求2所述的双极化辐射单元，其特征在于，所述双极化辐射单元具有八边形的形状。
  4. 根据权利要求1所述的双极化辐射单元，其特征在于，所述双极化辐射单元还包括：
    四条缝隙，所述四条缝隙分别位于所述四个区域中的两个相邻的区域的邻接处。
  5. 根据权利要求4所述的双极化辐射单元，其特征在于，所述双极化辐射单元还包括：
    四条馈电线，所述四条馈电线与所述四条缝隙相关联，并且所述四条馈电线中的每条馈电线从所述双极化辐射单元的中间区域延伸至与之相关联的缝隙的馈电点处，其中，所述馈电线的长度与匹配阻抗相关联。
  6. 根据权利要求5所述的双极化辐射单元，其特征在于，所述双极化辐射单元还包括：
    共模扼流电路，所述共模扼流电路被构造为设置在所述馈电点附近并 且与所述四条缝隙中的一条缝隙相关联。
  7. 根据权利要求6所述的双极化辐射单元，其特征在于，所述共模扼流电路进一步包括第一轨道和第二轨道，其中，所述第一轨道和所述第二轨道被以电感线圈的形式相互平行地构造在所述辐射臂的两侧上，并且其中，所述第一轨道和所述第二轨道的电长度相等，并且所述第一轨道和所述第二轨道的线圈绕行方向一致。
  8. 根据权利要求1所述的双极化辐射单元，其特征在于，所述双极化辐射单元还包括：
    并联滤波器，所述并联滤波器被构造为从所述中空区域的边缘向内延伸的金属丝。
  9. 根据权利要求8所述的双极化辐射单元，其特征在于，所述并联滤波器被构造为开路线。
  10. 根据权利要求1所述的双极化辐射单元，其特征在于，所述双极化辐射单元还包括：
    电感元件，所述电感元件从所述中空区域的远离所述双极化辐射单元的中心区域的边缘朝向所述中心区域延伸。
  11. 一种天线，其特征在于，所述天线包括：
    至少一个根据权利要求1至10中任一项所述的双极化辐射单元；以及
    辐射单元匹配电路。
  12. 一种天线系统，其特征在于，所述天线系统包括：
    根据权利要求11所述的天线；以及
    至少一个第二天线，其中，所述至少一个第二天线的工作频段比所述天线的工作频段更高。
  13. 根据权利要求12所述的天线系统，其特征在于，所述天线与所述第二天线交错设置。
  14. 根据权利要求13所述的天线系统，其特征在于，所述天线的列数和所述第二天线的列数相等，或者所述第二天线的列数是所述天线的列数的两倍。
PCT/CN2022/077196 2021-08-02 2022-02-22 用于天线的双极化辐射单元、天线以及天线系统 WO2023010829A1 (zh)

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CN1591976A (zh) * 2003-08-27 2005-03-09 广州埃信科技有限公司 双极化天线
WO2015124573A1 (en) 2014-02-18 2015-08-27 Filtronic Wireless Ab Broadband antenna, multiband antenna unit and antenna array
US20180097293A1 (en) * 2016-10-05 2018-04-05 Kathrein-Werke Kg Antenna for mobile communication
CN113140893A (zh) * 2020-01-20 2021-07-20 康普技术有限责任公司 用于基站天线应用的紧凑型宽带双极化辐射元件

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Publication number Priority date Publication date Assignee Title
CN1591976A (zh) * 2003-08-27 2005-03-09 广州埃信科技有限公司 双极化天线
WO2015124573A1 (en) 2014-02-18 2015-08-27 Filtronic Wireless Ab Broadband antenna, multiband antenna unit and antenna array
US20180097293A1 (en) * 2016-10-05 2018-04-05 Kathrein-Werke Kg Antenna for mobile communication
CN113140893A (zh) * 2020-01-20 2021-07-20 康普技术有限责任公司 用于基站天线应用的紧凑型宽带双极化辐射元件

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