WO2021243989A1 - 一种基于多传感器融合的快响应直流比较仪 - Google Patents

一种基于多传感器融合的快响应直流比较仪 Download PDF

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Publication number
WO2021243989A1
WO2021243989A1 PCT/CN2020/133573 CN2020133573W WO2021243989A1 WO 2021243989 A1 WO2021243989 A1 WO 2021243989A1 CN 2020133573 W CN2020133573 W CN 2020133573W WO 2021243989 A1 WO2021243989 A1 WO 2021243989A1
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magnetic
signal
current
output
core
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English (en)
French (fr)
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韩小涛
张绍哲
王奕霖
谢剑峰
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Huazhong University of Science and Technology
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Huazhong University of Science and Technology
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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R15/00—Details of measuring arrangements of the types provided for in groups G01R17/00 - G01R29/00, G01R33/00 - G01R33/26 or G01R35/00
    • G01R15/14—Adaptations providing voltage or current isolation, e.g. for high-voltage or high-current networks
    • G01R15/18—Adaptations providing voltage or current isolation, e.g. for high-voltage or high-current networks using inductive devices, e.g. transformers
    • G01R15/183—Adaptations providing voltage or current isolation, e.g. for high-voltage or high-current networks using inductive devices, e.g. transformers using transformers with a magnetic core
    • G01R15/185—Adaptations providing voltage or current isolation, e.g. for high-voltage or high-current networks using inductive devices, e.g. transformers using transformers with a magnetic core with compensation or feedback windings or interacting coils, e.g. 0-flux sensors
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R15/00—Details of measuring arrangements of the types provided for in groups G01R17/00 - G01R29/00, G01R33/00 - G01R33/26 or G01R35/00
    • G01R15/14—Adaptations providing voltage or current isolation, e.g. for high-voltage or high-current networks
    • G01R15/20—Adaptations providing voltage or current isolation, e.g. for high-voltage or high-current networks using galvano-magnetic devices, e.g. Hall-effect devices, i.e. measuring a magnetic field via the interaction between a current and a magnetic field, e.g. magneto resistive or Hall effect devices
    • G01R15/202—Adaptations providing voltage or current isolation, e.g. for high-voltage or high-current networks using galvano-magnetic devices, e.g. Hall-effect devices, i.e. measuring a magnetic field via the interaction between a current and a magnetic field, e.g. magneto resistive or Hall effect devices using Hall-effect devices
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R19/00—Arrangements for measuring currents or voltages or for indicating presence or sign thereof
    • G01R19/0038—Circuits for comparing several input signals and for indicating the result of this comparison, e.g. equal, different, greater, smaller (comparing pulses or pulse trains according to amplitude)
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R19/00—Arrangements for measuring currents or voltages or for indicating presence or sign thereof
    • G01R19/18—Arrangements for measuring currents or voltages or for indicating presence or sign thereof using conversion of DC into AC, e.g. with choppers
    • G01R19/20—Arrangements for measuring currents or voltages or for indicating presence or sign thereof using conversion of DC into AC, e.g. with choppers using transductors, i.e. a magnetic core transducer the saturation of which is cyclically reversed by an AC source on the secondary side
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only

Definitions

  • the invention belongs to the technical field of current measurement, and more specifically, relates to a fast-response DC comparator based on multi-sensor fusion.
  • the magnetic modulation DC comparator is the most accurate current sensor in existence, with a measurement accuracy of up to 10 ppm, and has extremely low temperature drift, high stability and high sensitivity. Therefore, current sensor giants such as LEM, Hitech, and Bergoz have the precision Current sensors all adopt this type.
  • LEM, Hitech, and Bergoz have the precision Current sensors all adopt this type.
  • False balance phenomenon refers to the fact that the magnetic core material of the DC comparator is completely saturated, the output of the magnetic modulation current sensor is zero and it does not respond to any input, and the system enters a false balance state.
  • the reasons for this phenomenon are: (1) Before the power is turned on, the measured current passing through the magnetic core of the magnetic modulation current sensor already exists; (2) The measured current exceeds the compensation range of the DC current comparator; 3) The amplitude of the measured current changes rapidly in a wide range. At this time, the sensor cannot work at all, and cannot automatically exit from the abnormal state. False balance greatly limits the application range of the magnetic modulation DC current comparator.
  • the commonly used false balance solutions can be divided into two types: the system restart method and the secondary side current scanning method.
  • the system restart method means: turn off the entire system, then turn on the DC current comparator first, and then load the measured current after the PI feedback loop starts to work.
  • This method is extremely inefficient and cannot be widely used. Therefore, the secondary-side current scanning method is widely used to solve the false balance problem, that is, a scanning module is added to the circuit, which is usually composed of a low-frequency triangular wave generator and a voltage comparator. The module monitors the working status in real time. When the false balance is detected, the secondary winding disconnects the PI feedback loop to connect to the scanning module, and the circuit enters the scanning state, trying to find the zero magnetic flux state. Once the scan reaches the zero magnetic flux state, the scan module exits its work, returns the control to the PI feedback loop, and the circuit returns to its normal working state.
  • the current scanning process is very slow. When false balance occurs, the magnitude of the measured current, that is, the balance point is unknown. In order to avoid the system being too late to react and miss the balance point, scanning must be performed slowly; and the inductance of the secondary coil of the large-range DC comparator is very large, and the fast scanning speed means the supply voltage When the equilibrium state is found, the current needs to stabilize quickly to avoid instability again. At this time, a large induced voltage may be generated and the equipment may be damaged. (2) The dynamic response performance of the system has not been improved. Even if it leaves the false balance state and the measured current experiences the same large-scale fluctuation again, the system will still enter the false balance and must start scanning again, which is inefficient.
  • the purpose of the present invention is to provide a fast-response DC comparator based on multi-sensor fusion, which aims to not affect the measurement accuracy, stability, time drift, temperature drift and other steady-state performance of the DC comparator.
  • Improve the dynamic response capability of the DC comparator avoid false balance during the dynamic process of the DC comparator, and realize the automatic recovery of the false balance state caused by live switching, overload, etc., so as to realize the precision of large-range dynamic DC high current Measurement.
  • the invention also provides a fast-response DC comparator based on multi-sensor fusion, which includes: a magnetic modulation current sensor, a current correction module, an excitation transformer, an AC detection filter circuit, a phase-sensitive demodulation filter system, a PI controller, and a power Amplifier; the excitation voltage input terminal of the magnetic modulation current sensor is connected to the output terminal of the excitation transformer, and the feedback input terminal is connected to the output terminal of the power amplifier, which is used to excite the magnetic core of the magnetic modulation current sensor through the excitation voltage.
  • One signal output terminal outputs the balance judgment signal to the phase-sensitive demodulation filter system, and the second signal output terminal outputs the AC induced voltage signal to the AC detection filter circuit;
  • the current correction module is used to measure the measured current on the primary side and obtain the feedforward signal, The feedforward signal and the PI control signal are amplified and then output is used to control the magnetic modulation core to quickly exit or avoid entering the false equilibrium state of magnetic saturation, and to keep the output of the magnetic modulation current sensor stable;
  • the input terminal of the excitation transformer is used for connection Power frequency alternating current, the first output terminal is connected to the excitation voltage input terminal of the magnetic modulation current sensor, and the second output terminal is connected to the second input terminal of the phase-sensitive demodulation filtering system, which is used to provide periodicity for the magnetic modulation current sensor And provide the required square wave signal to the phase-sensitive demodulation filter system;
  • the input end of the AC detection filter circuit is connected to the second signal output end of the magnetic modulation current sensor for The AC induced voltage signal output by the current
  • the present invention due to the fusion of the magnetic modulation current sensor and the Hall current sensor, it has a stronger dynamic response capability, and can respond quickly when faced with a wide range of current amplitude changes, which greatly reduces The possibility of malfunction of the DC comparator due to the false balance problem caused by the saturation of the magnetic core.
  • the present invention can control the magnetic core to quickly exit the magnetic saturation state through the feedforward output current of the Hall current sensor without restarting or scanning the system for a long time, which greatly reduces the return of the DC comparator to normal operation Response time.
  • the magnetic modulation current sensor includes: a first magnetic modulation magnetic core C 1 , a second magnetic modulation magnetic core C 2 , an AC feedback magnetic core C 3 , a first winding W 1 , a second winding W 2 , an AC induction The winding W 3 and the secondary winding W S ; the first magnetic modulation magnetic core C 1 and the second magnetic modulation magnetic core C 2 are arranged in parallel to each other, and the first magnetic modulation magnetic core C 1 and the second magnetic modulation magnetic core C 1 The magnetic cores C 2 are respectively wrapped by insulating materials and arranged in the AC feedback magnetic core C 3 ; the first winding W 1 is wound on the first magnetic modulation magnetic core C 1 , and the second winding W 2 wound around the second magnetic core C 2 modulation, and said first winding and said second winding W 1 W 2 in series, for detecting a modulation of the first magnetic core and said C 1 The magnitude of the magnetic flux in the second magnetically modulated magnetic core C 2 and outputs a balance judgment signal; the induction wind
  • first winding W 1 and the second winding W 2 are connected in reverse series, and the moment of the alternating magnetic flux caused by the excitation current in the first magnetic modulation core C 1 and the second magnetic modulation core C 2 Keep the same size and the opposite direction.
  • the materials of the first magnetic modulation core C 1 and the second magnetic modulation core C 2 are ferromagnetic materials with a relatively high rectangular coefficient and a relatively low coercivity.
  • the materials of the first magnetic modulation core C 1 and the second magnetic modulation core C 2 are both cold-rolled silicon steel sheets with grain orientation, permalloy or amorphous alloy.
  • the shapes of the first magnetic modulation core C 1 and the second magnetic modulation core C 2 are toroidal shapes, and their diameters are determined by the number of turns of the measured bus bar and the secondary winding, and a large DC current must be ensured During the measurement, the bus bar can pass through the inside of the modulating iron core ring.
  • the current correction module plays a feed-forward function, and adjusts the secondary side feedback current instantly according to the magnitude of the measured current on the primary side.
  • the Hall current sensor branch and the magnetic core
  • the mutual inductance works together, and the instant output of the power amplifier avoids the saturation of the magnetic core and strengthens the mutual inductance of the magnetic core.
  • the mutual inductance can make the system enter the zero flux state faster.
  • the feedback loop of the magnetic modulation current sensor only needs to provide a small voltage to ensure zero magnetic flux. Since the Hall current sensor is outside the feedback loop, it will not affect the zero magnetic flux detection, so the current measurement accuracy and stability will not be affected.
  • the current correction module includes: a Hall current sensor chip and a signal conditioning circuit; the Hall current sensor chip is used to detect the current to be measured, and according to the current to be measured and the feedforward signal provided by the power amplifier to the magnetic modulation current sensor To control the magnetic modulation core to quickly exit or avoid entering the false equilibrium state of magnetic saturation, so that the output of the magnetic modulation current sensor is stable; the signal conditioning circuit is used to filter and size the output signal of the Hall current sensor chip and the PI controller adjust.
  • the number of Hall current sensor chips is eight, and the eight Hall current sensor chips are divided into four groups that are perpendicular to each other and symmetrically arranged on the inner side of the ring-shaped magnetic modulation current sensor.
  • the phase-sensitive demodulation filter system includes: a frequency selection circuit, a synchronous square wave circuit, a second-order band pass filter, an analog multiplier, and a second-order low-pass filter; the input of the frequency selection circuit is used as a phase-sensitive demodulator
  • the first input terminal of the filter system is used to extract the second harmonic signal in the input signal of the winding
  • the input terminal of the synchronous square wave circuit is used as the second input terminal of the phase-sensitive demodulation filter system, which is used to process the excitation voltage and Ensure that the output phase reference square wave signal is in phase synchronization with the second harmonic signal output by the frequency selection circuit
  • the input end of the second-order band pass filter is connected to the output end of the frequency selection circuit for the second harmonic signal Perform filtering
  • the first input of the analog multiplier is connected to the output of the second-order bandpass filter, and the second input is connected to the output of the synchronous square wave circuit for filtering the second harmonic signal and phase
  • the reference square wave signal is phase-sensitive de
  • the multi-sensor fusion DC comparator provided by the present invention adopts the design of fusion with the Hall current sensor, and has stronger dynamic response ability, and faces a wide range of current amplitude changes. It can respond quickly, which greatly reduces the possibility of failure of the DC comparator due to the false balance problem caused by the saturation of the magnetic core; in the case that the false balance has occurred, the sensor fusion design of the present invention can pass the Hall current sensor
  • the feedforward output current makes the magnetic core quickly exit the magnetic saturation state without restarting or scanning the system for a long time, which greatly reduces the time for the DC comparator to resume normal operation and response.
  • Fig. 1 is a structural principle diagram of a fast-response DC comparator based on multi-sensor fusion provided by an embodiment of the present invention
  • FIG. 2 is a schematic diagram of the positional relationship of the Hall current sensor in the current correction module of the fast response DC comparator provided by the embodiment of the present invention
  • FIG. 3 is a geometric structure diagram of a magnetically modulated current sensor coil in a fast-response DC comparator provided by an embodiment of the present invention
  • FIG. 4 is a schematic diagram of the connection relationship between the phase-sensitive demodulation filter system and the excitation transformer in the fast-response DC comparator provided by an embodiment of the present invention
  • FIG. 5 is a schematic diagram of the module structure of a phase-sensitive demodulation filtering system provided by an embodiment of the present invention.
  • 1 is a magnetic modulation current sensor
  • 21 is a dot matrix Hall current sensor
  • 22 is a signal conditioning circuit
  • 3 is an excitation transformer
  • 4 is an AC detection filter circuit
  • 5 is a phase-sensitive demodulation filter system
  • 6 is the PI controller
  • 7 is the power amplifier.
  • the invention provides a fast-response DC comparator based on multi-sensor fusion, which can improve the dynamic response of the DC comparator without affecting the steady-state performance such as measurement accuracy, stability, time drift and temperature drift of the DC comparator It avoids false balance in the dynamic process of the DC comparator, and realizes the automatic recovery of false balance due to live switching, overload, etc., so as to realize the precise measurement of dynamic DC large current.
  • Fig. 1 shows the structure of a fast-response DC comparator based on multi-sensor fusion provided by an embodiment of the present invention.
  • Fig. 1 shows the structure of a fast-response DC comparator based on multi-sensor fusion provided by an embodiment of the present invention.
  • the parts related to the embodiment of the present invention are shown, which are detailed as follows:
  • the fast response DC comparator based on multi-sensor fusion includes: magnetic modulation current sensor 1, current correction module, excitation transformer 3, AC detection filter circuit 4, phase-sensitive demodulation filter system 5, PI controller 6 and power amplifier 7; Wherein, the excitation voltage input terminal of the magnetic modulation current sensor 1 is connected to the output terminal of the excitation transformer 3, and the feedback input terminal is connected to the output terminal of the power amplifier 7 for controlling the magnetic modulation current sensor 1 through the excitation voltage.
  • the magnetic core is excited, the first signal output terminal outputs the balance judgment signal to the phase-sensitive demodulation filter system 5, and the second signal output terminal outputs the AC induced voltage signal to the AC detection filter circuit 4;
  • the current correction module is used for one measurement Side measured current and obtain the feedforward signal, the feedforward signal and PI control signal are amplified and output is used to control the magnetic modulation core to quickly exit or avoid entering the false equilibrium state of magnetic saturation, and to keep the output of the magnetic modulation current sensor Stable;
  • the input end of the excitation transformer 3 is used to connect the power frequency alternating current, the first output end is connected to the excitation voltage input end of the magnetic modulation current sensor 1, and the second output end is connected to the second input of the phase-sensitive demodulation filter system 5
  • the terminal is used to provide periodic excitation voltage for the magnetic modulation current sensor 1 and to provide the required square wave signal to the phase-sensitive demodulation filter system 5;
  • the input terminal of the AC detection filter circuit 4 is connected to the magnetic modulation current sensor 1
  • the Hall sensor By adopting the fusion method of the Hall current sensor and the magnetic modulation current sensor, the Hall sensor provides a reverse magnetic flux to the magnetic core in real time according to the magnitude of the measured current on the primary side, which can prevent the magnetic core from being saturated or saturated. When the magnetic core is "drag" out of the saturation state. In this way, the current change range that the instrument can measure and withstand is much larger than that of the original magnetic modulation current sensor alone, that is, "improved dynamic response performance".
  • the output of the Hall sensor is not like a magnetic modulation current sensor with a very complicated closed loop, but as soon as there is a measured current on the primary side
  • the reverse magnetic potential is provided to the magnetic core, so it can make the balance faster, that is, "fast response”. And because there is no need to restart or scan, the overall work performance is greatly improved.
  • the current correction module is used to measure the measured current on the primary side through the current sensor, and the resulting sensor output is proportional to the feedforward signal of the measured current on the primary side and the PI control signal output by the PI controller 6 is connected to the signal conditioning
  • the circuit input terminal, after filtering and amplitude modulation processing, the feedforward signal and PI control signal are output to the input terminal of the power amplifier 6.
  • the feedforward signal and PI control signal are amplified to provide feedforward input and closed loop feedback to the magnetic modulation current sensor 1 Input, control the magnetic modulation core to quickly exit or avoid entering the false equilibrium state of magnetic saturation, so that the output of the magnetic modulation current sensor remains stable.
  • the current correction module can use the sensor to perform feedforward correction on the magnetic modulation current sensor.
  • the current correction module includes: a Hall current sensor chip 21 and a signal conditioning circuit 22; the Hall current sensor chip 21 is used to detect the current to be measured, and according to the current to be measured and the power amplifier provided to the magnetic modulation current sensor. The signal is used to control the magnetic modulation core to quickly exit or avoid entering the false equilibrium state of magnetic saturation, so that the output of the magnetic modulation current sensor is stable; the signal conditioning circuit 22 is used to filter and adjust the output signal of the Hall current sensor chip .
  • the number of Hall current sensor chips 21 can be eight, and the eight Hall current sensor chips 21 are divided into four groups that are perpendicular to each other and symmetrically arranged on the inner side of the ring-shaped magnetic modulation current sensor.
  • the Hall current sensor chip 21 may also be arranged outside the magnetic modulation current sensor.
  • the excitation transformer, the magnetic modulation core and its windings are used as the DC feedback path of the DC comparator to detect DC and low-frequency magnetic field components;
  • the magnetic modulation current sensor adopts a differential structure, and the first winding and the second winding The windings are wound in reverse series on two magnetic modulation cores to detect and output the magnetic flux in the magnetic modulation cores;
  • the secondary side of the excitation transformer is connected to the first winding and the second winding respectively;
  • the two magnetic The alternating magnetic flux caused by the excitation current in the modulating magnetic core is reversed in series because the first and second windings are always equal in size and opposite in direction. Their external performance and the combined magnetic flux are zero, so the excitation current will not cause excitation.
  • the output is generated in the transformer.
  • the secondary side of the excitation transformer 3 is respectively connected to the first winding W 1 and the second winding W 2 of the magnetic modulation sensor 1, and the output ends of the first winding W 1 and the second winding W 2 are connected to the phase-sensitive demodulation filter system.
  • induction winding W 3 modulated output of the detection sensor is connected to the AC filter circuit
  • a phase sensitive detector demodulates the filter system AC filter circuit connected to the output terminal of the PI controller
  • a PI controller the output of the signal conditioning circuit connected to the input terminal of the current correction module, Its output end is connected to the power amplifier, and the output end of the power amplifier is connected to the secondary winding W S of the magnetic modulation sensor.
  • the modulating core material is usually a ferromagnetic material with a higher rectangular coefficient and a lower coercivity, such as grain-oriented cold-rolled silicon steel sheet, permalloy, and amorphous.
  • the shape of the modulating magnetic core is usually a circular ring, and its diameter is determined by factors such as the measured bus bar and the number of turns of the secondary winding. It is necessary to ensure that the bus bar can pass through the modulating iron core ring when measuring a large DC current.
  • the selection of the cross-sectional area of the modulated iron core should ensure the slenderness ratio of the iron core. For example, the ratio of the average circumference of the ring to the cross-sectional area can be selected to be greater than 50, and the optimal effect is determined by experiment.
  • the modulating magnetic core is wound by a high-permeability amorphous alloy, and the average diameter of the sensor may be 1124 mm.
  • FIG. 3 shows a geometric structure diagram of a magnetic modulation current sensor coil provided by an embodiment of the present invention.
  • the first winding W 1 and the second winding W 2 are wound in reverse series on the first magnetic modulation core C 1 and the second magnetic
  • the modulation magnetic core C 2 is used to detect and output the magnetic flux in the magnetic modulation magnetic core
  • the induction winding W 3 is wound on the AC feedback magnetic core C 3 to detect the magnetic flux in the AC feedback magnetic core.
  • the structural parameters of the first winding W 1 and the second winding W 2 are kept the same, and they are wound in reverse series.
  • the first magnetic modulation core C 1 and the second magnetic modulation core C 2 The alternating magnetic flux caused by the excitation current keeps equal in magnitude and opposite direction at all times, and their external manifestation is a composite magnetic flux of zero, so the excitation current does not cause output in the excitation transformer.
  • the magnetic characteristics of the modulating core are determined by the material.
  • the number of turns of the first winding W 1 and the second winding W 2 can generally be 1000 turns.
  • the wire diameter is determined according to the size of the excitation current, and the current density of the close-wound coil according to the national standard is not greater than 2.5A/ mm 2 , and then determine the output resistance value and the excitation voltage value according to the size of the output signal.
  • the cross-sectional size of the first magnetic modulation core C 1 and the second magnetic modulation core C 2 is 10mm ⁇ 8mm, and the corresponding excitation first winding W 1 and second winding W 2 are wound with 0.2mm enameled wire, and the turns
  • the number is 1000 turns. There is no special requirement for the number of turns of the excitation winding, as long as it can carry the current ampere-turns required at the peak magnetic potential.
  • the thickness of the iron core shielding layer is usually 5mm, and it is wound with high-permeability material strips, such as grain-oriented cold-rolled silicon steel sheets, permalloy, and amorphous.
  • the gap between the iron core shielding layer and the modulated iron core is filled with electrical potting glue or other insulating fillers.
  • the secondary winding W S is wound on the first magnetic modulation magnetic core C 1 , the second magnetic modulation magnetic core C 2 and the AC feedback magnetic core C 3 . Its function is to balance the magnetic potential of the primary-side measured current I 1 with the DC magnetic potential generated by the secondary-side current I 2, and the primary-side measured current value is calculated by the magnitude of the secondary-side current.
  • the number of turns of the secondary winding depends on the magnitude of the measured current on the primary side and the current rating of the secondary side.
  • N 1 I 1 N 2 I 2
  • the wire diameter is selected according to the national standard that the current density of the close-wound coil is not more than 2.5A/mm 2 and I 2 .
  • the measured maximum current is 30 kA
  • the number of winding turns is 10000
  • the enameled wire with a wire diameter of 1.2 mm is used for winding.
  • the AC feedback magnetic core C 3 and its induction winding are used as the AC feedback path of the DC comparator to detect the intermediate frequency AC magnetic field component and suppress the induced ripple of the system.
  • the induction winding is wound on the AC feedback magnetic core to detect the magnetic flux in the AC feedback magnetic core; the output voltage of the induction winding after the AC detection filter circuit and the output voltage of the magnetic modulation current sensor are used as the PI controller
  • the control quantity through the power amplifier, drives the secondary winding to form a zero-flux closed-loop feedback control.
  • the AC feedback magnetic core C 3 serves as the magnetic shield of the AC detection magnetic core, the transformer magnetic core and the magnetic modulation magnetic core at the same time. Its section wall thickness is 10mm, the section height is 45mm, the section width is 28mm, and the upper left corner of the magnetic core has 1mm insulation. It is filled to prevent the formation of a loop in the winding direction to generate an eddy current path.
  • AC induction winding W 3 is detecting an open state in which no current flows, so the diameter is not required, the present embodiment employed for enameled wire wound 0.2mm, the number of turns associated with sensitivity detection branch exchange, but also its output signal It needs to go through a signal conditioning circuit, so the selection of the number of turns is not very strict. In this embodiment, it is selected to be 1000 turns.
  • the design of the feedback winding needs to comprehensively consider the range and the measurement of the secondary side current to determine the number of winding turns, that is, the ratio of the primary current to the secondary side current.
  • a large number of turns means that the secondary side current is small, which is conducive to the measurement of the secondary side current. It also means that the winding resistance and inductance are large, which will increase the design difficulty of the power amplifier, and the large number of turns will also increase the division of the winding. parameter.
  • the excitation amplifier of the fast-response DC comparator based on multi-sensor fusion selects power frequency 50Hz as the magnetic modulation excitation frequency, and the excitation source is directly obtained from the commercial power through a power frequency transformer with a center tap.
  • Fig. 4 shows the connection relationship between the phase-sensitive demodulation filter system 5 and the excitation transformer 3 in the fast-response DC comparator provided by the embodiment of the present invention
  • the square wave required for phase-sensitive demodulation is generated by a comparator after the excitation voltage is rectified, and then undergoes a phase adjustment to compensate the phase error generated by the filter to ensure the phase of the second harmonic voltage v(2f) and the square wave voltage v(s) Synchronize.
  • v(2f) and v(s) undergo phase-sensitive demodulation through the analog multiplier AD633, and then output through a second-order low-pass filter.
  • FIG. 5 shows the module structure of the phase-sensitive demodulation filter system provided by an embodiment of the present invention.
  • the phase-sensitive demodulation filter system 5 includes: a frequency selection circuit 51, a synchronous square wave circuit 52, a second-order band pass filter 53, an analog The multiplier 54 and the second-order low-pass filter 55; the input end of the frequency selection circuit 51 is used as the first input end of the phase-sensitive demodulation filter system 5 for extracting the second harmonic signal in the winding input signal; synchronous square wave
  • the input terminal of the circuit 52 is used as the second input terminal of the phase-sensitive demodulation filter system 5, which is used to process the excitation voltage and ensure that the output phase reference square wave signal is in phase synchronization with the second harmonic signal output by the frequency selection circuit;
  • the input end of the first-order band pass filter 53 is connected to the output end of the frequency selection circuit 51 for filtering the second harmonic signal; the first input end of the analog multiplier 54 is connected to the output of the second-order band pass filter 53 The second input terminal
  • the functions of the PI controller 6 and the power amplifier 7 are now explained in detail.
  • the voltage signal adjusted by the power amplifier is output to the secondary winding Ws of the magnetic modulation current sensor.
  • the coupling effect of each branch in the transition phase of the frequency band is not considered for the time being.
  • the entire measurement system can be regarded as In the case of a linear system, the working process of the DC comparator.
  • the primary current I 1 When the primary current I 1 is in the lower frequency band (including DC), it is generally below f ex /2. At this time, the output voltage of the AC detection induction winding W 3 is zero and there is no connection between the measured current loop and the secondary winding on the primary side. Mutual inductance; among them, f ex is the excitation voltage frequency of the excitation transformer, generally 50 Hz.
  • the dot matrix Hall current sensor detects the primary side current, outputs a constant voltage, and enters the power amplifier after passing through the signal conditioning circuit; the output of the magnetic modulation amplifier passes through the PI controller and the signal conditioning circuit and then enters the power amplifier.
  • the dot matrix Hall current sensor detects the AC current, outputs a constant voltage, and enters the power amplifier after passing through the signal conditioning circuit; the output voltage of the magnetic modulation amplifier branch is zero, so no one is considered the measured mutual inductance between the current loop and the secondary-side winding, the primary current to be measured AC induced voltage is generated in the detected inductive winding W 3, after the PI controller input of the power amplifier and signal conditioning circuit.
  • the magnetic modulation current sensor acts as an AC current comparator to maintain a zero magnetic flux state.
  • the adjustment process is similar to the low frequency, except that the magnetic flux error control signal comes from the electromagnetic induction voltage of the AC detection induction winding W 3 and the output voltage of the dot matrix Hall current sensor.
  • the above three working states are coupled with each other.
  • the mutual inductance between the output of the Hall current sensor and the magnetic modulation current sensor first acts.
  • the mutual inductance core is not saturated, zero magnetic flux can be maintained in the secondary winding under the action of the Hall sensor and the mutual inductance. state.
  • the current-time product ampere-second product
  • the mutual inductance magnetic core gradually saturates.
  • the Hall sensor, magnetic modulation current sensor and AC detection core work together to control the output of the power amplifier, drive the secondary winding, and maintain the zero magnetic field state.
  • the measured current on the primary side has a large variation range.
  • the saturation of the mutual inductance core during the dynamic process causes the mutual inductance to drop rapidly. If the power amplifier cannot produce output in time, it can maintain the secondary side current. The current will drop rapidly, which will cause the unbalanced current to exceed the linear range of the magnetic modulation amplifier, and the closed-loop system of the DC comparator will become unstable and enter a false equilibrium state.
  • the magnetic modulation amplifier in the multi-sensor fusion system cannot work due to the saturation of the magnetic core, and the dot matrix Hall current sensor outputs a voltage that changes in the same proportion according to the measured primary current. , Directly control the power amplifier to drive the secondary winding, so that the magnetic core of the magnetic modulation amplifier exits the magnetic saturation and returns to the normal working state.
  • the Hall current sensor branch plays a feed-forward role.
  • the real-time output of the power amplifier avoids the saturation of the mutual inductance magnetic core, strengthens the mutual inductance of the magnetic modulation amplifier, and the mutual inductance can make the system enter the zero flux state faster, making the DC comparator more dynamic Responsiveness, it can respond quickly when faced with a wide range of current amplitude changes.
  • the output of the Hall sensor gives the power amplifier a static operating point, and the feedback loop of the magnetic modulation current sensor only needs to provide a small voltage to ensure zero magnetic flux.

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Abstract

一种基于多传感器融合的快响应直流比较仪，包括：磁调制式电流传感器（1）、电流矫正模块、激励变压器（3）、交流检测滤波电路（4）、相敏解调滤波系统（5）、PI控制器（6）和功率放大器（7）；电流矫正模块测量一次侧被测电流并获得前馈信号，将前馈信号与PI控制信号进行放大后输出用于控制磁调制磁芯快速退出或避免进入磁饱和的虚假平衡状态，且使磁调制式电流传感器（1）输出保持稳定。快响应直流比较仪将磁调制式电流传感器（1）与霍尔电流传感器进行融合，降低了因磁芯饱和导致虚假平衡问题而发生故障的可能性；在虚假平衡已经产生后可通过前馈输出电流控制磁芯快速退出磁饱和状态，无需重启或长时间扫描系统，减少了恢复正常工作响应的时间。

Description

一种基于多传感器融合的快响应直流比较仪 【技术领域】
本发明属于电流测量技术领域，更具体地，涉及一种基于多传感器融合的快响应直流比较仪。
【背景技术】
大电流(具体是指大于等于1kA的高强度电流)在工业生产、直流输电、高速铁路、科学实验等各行业都有广泛的应用，因此大电流测量技术一直以来都是重要研究课题。目前常见的直流大电流的测量方法有分流器、直流互感器、罗氏线圈传感器、光纤传感器、霍尔效应直流传感器等，但是绝大多数的大电流传感器存在测量精度较低、温度漂移较大等问题，无法满足直流大电流的10ppm级高精度快速稳定测量要求。
近几十年来，基于磁通门法的直流大电流测量技术受到了广泛研究与快速发展。其中，磁调制式直流比较仪作为现存最精确的电流传感器，测量精度可达10ppm，并且具有极低的温度漂移和高稳定性与高灵敏度，因此目前LEM、Hitech、Bergoz等电流传感器巨头的精密电流传感器都是采用该类型。虽然目前磁调制式直流电流比较仪产业已经成熟，但是在虚假平衡问题的解决方案、动态响应性能等方面需要进行研究和改进。
“虚假平衡现象”是指直流比较仪磁芯材料完全饱和、磁调制式电流传感器输出为零且对任何输入均不响应，系统进入一种虚假平衡状态。产生这一现象的原因有：(1)电源上电前，穿过磁调制式电流传感器磁芯的被测电流已经存在；(2)被测电流大小超过直流电流比较仪可补偿的范围；(3)被测电流幅值大范围快速变化。此时，传感器完全无法工作，且无法从异常状态自动退出。虚假平衡大大限制了磁调制式直流电流比较仪的应用范围。
目前常用的虚假平衡解决方案可分为系统重启法与二次侧电流扫描法两种。系统重启法是指：关闭整个系统，然后先开启直流电流比较仪，待PI反馈回路开始工作后，再加载被测电流。该方法效率极低，基本无法普及使用。故目前广泛采用二次侧电流扫描法解决虚假平衡问题，即在电路中增加扫描模块，通常由低频三角波发生器与电压比较器组成。模块实时监测工作状态，当监测到虚假平衡发生时，二次绕组断开PI反馈回路连接扫描模块，电路进入扫描状态，试图寻找零磁通状态。一旦扫描到零磁通状态，扫描模块退出工作，将控制权交回PI反馈回路，电路回归工作正常状态。
虽然二次侧电流扫描法应用已经成熟，但是该方法依然存在两个问题：(1)电流扫描过程十分缓慢。当发生虚假平衡时，被测电流大小即平衡点未知，为了避免系统来不及反应而错过平衡点，扫描必须缓慢进行；且大量程直流比较仪二次线圈电感非常大，扫描速度快意味着供电电压大，而且当寻得平衡状态时，电流需要快速稳定以免再次失稳，此时可能产生较大感应电压而造成设备损坏。(2)该系统的动态响应性能未得到改善。即使脱离虚假平衡状态，被测电流再次经历同样的大范围波动，系统依旧会进入虚假平衡，必须再次开始扫描，效率较低。
【发明内容】
针对现有技术缺陷，本发明的目的在于提供一种基于多传感器融合的快响应直流比较仪，旨在不影响直流比较仪的测量精度、稳定度、时漂和温漂等稳态性能的前提下改善直流比较仪的动态响应能力，避免直流比较仪动态过程中发生虚假平衡，并实现由于带电合闸、过载等原因造成的虚假平衡状态的自动恢复，从而实现大范围动态直流大电流的精密测量。
本发明还提供了一种基于多传感器融合的快响应直流比较仪，包括：磁调制式电流传感器、电流矫正模块、激励变压器、交流检测滤波电路、相敏解调滤波系统、PI控制器和功率放大器；磁调制式电流传感器的励磁 电压输入端连接至激励变压器的输出端，反馈输入端连接至功率放大器的输出端，用于通过励磁电压对磁调制式电流传感器的磁芯进行励磁，由第一信号输出端输出平衡判断信号给相敏解调滤波系统，由第二信号输出端输出交流感应电压信号给交流检测滤波电路；电流矫正模块用于测量一次侧被测电流并获得前馈信号，将前馈信号与PI控制信号进行放大后输出用于控制磁调制磁芯快速退出或避免进入磁饱和的虚假平衡状态，且使磁调制式电流传感器输出保持稳定；激励变压器的输入端用于连接工频交流电，第一输出端连接至磁调制式电流传感器的励磁电压输入端，第二输出端连接至相敏解调滤波系统的第二输入端，用于为磁调制式电流传感器提供周期性的励磁电压，并向相敏解调滤波系统提供所需方波信号；所述交流检测滤波电路的输入端连接至所述磁调制式电流传感器的第二信号输出端，用于对磁调制式电流传感器输出的交流感应电压信号进行检测与滤波；相敏解调滤波系统的第一输入端连接至磁调制式电流传感器的第一输出端，第二输入端连接至激励变压器的第三输出端，用于对平衡判断信号进行相敏解调处理并输出平衡调节信号；PI控制器第一输入端连接至交流检测滤波电路的输出端，第二输入端连接至相敏解调滤波系统的输出端，用于对滤波后的交流感应电压信号与平衡调节信号进行比例积分控制并输出稳定的PI控制信号；功率放大器的输入端连接至电流矫正模块的输出端，用于对前馈信号与PI控制信号进行功率放大与滤波处理后输出反馈信号。
本发明中，由于采用了将磁调制式电流传感器与霍尔电流传感器进行融合，具有更强的动态响应能力，面对电流幅值大范围的变化时都能快速响应，极大程度地降低了直流比较仪因磁芯饱和导致虚假平衡问题而产生故障的可能性。另外，在虚假平衡已经产生的情况下，本发明可通过霍尔电流传感器的前馈输出电流控制磁芯快速退出磁饱和状态，无需重启或长时间扫描系统，大大减少了直流比较仪恢复正常工作响应的时间。
更进一步地，磁调制式电流传感器包括：第一磁调制磁芯C 1、第二磁 调制磁芯C 2、交流反馈磁芯C 3、第一绕组W 1、第二绕组W 2、交流感应绕组W 3和二次侧绕组W S；第一磁调制磁芯C 1和第二磁调制磁芯C 2相互平行设置，且所述第一磁调制磁芯C 1和所述第二磁调制磁芯C 2分别由绝缘材料包裹后设置于所述交流反馈磁芯C 3中；所述第一绕组W 1绕制在所述第一磁调制磁芯C 1上，所述第二绕组W 2绕制在所述第二磁调制磁芯C 2上，且所述第一绕组W 1与所述第二绕组W 2串联，用于检测所述第一磁调制磁芯C 1和所述第二磁调制磁芯C 2中的磁通大小，并输出平衡判断信号；所述感应绕组W 3绕制于所述交流反馈磁芯C 3上，所述感应绕组W 3用于检测交流反馈磁芯中的磁通大小，并输出交流感应电压信号；所述二次侧绕组W S绕制在包裹了所述交流反馈磁芯C 3的绝缘材料上，用于保证通过二次侧电流I 2产生的直流磁势与一次侧被测电流I 1的磁势平衡，且根据二次侧电流的大小计算待测电流值。
更进一步地，第一绕组W 1和第二绕组W 2反向串联，且第一磁调制磁芯C 1和所述第二磁调制磁芯C 2中的激励电流引起的交变磁通时刻保持大小相等方向相反。
更进一步地，第一磁调制磁芯C 1和所述第二磁调制磁芯C 2的材料均为矩形系数较高、矫顽力较小的铁磁材料。
更进一步地，第一磁调制磁芯C 1和所述第二磁调制磁芯C 2的材料均为晶粒取向的冷轧硅钢片、坡莫合金或非晶合金。
更进一步地，第一磁调制磁芯C 1和所述第二磁调制磁芯C 2的形状为圆环形，其直径由被测母线和二次侧绕组匝数决定，需保证直流大电流测量时母线可以从调制铁芯圆环内部穿过。
在本发明实施例中，电流矫正模块起到了前馈作用，根据一次侧被测电流的大小即时调整二次侧反馈电流，在一次侧被测电流范围变化时霍尔电流传感器支路和磁芯互感共同作用，功率放大器的即时输出避免了磁芯饱和，加强了磁芯互感能力，互感作用又可以使系统更快地进入零磁通状 态。一次侧被测电流进入稳态后，由于霍尔电流传感器的输出已经给予了功率放大器一个静态工作点，所以磁调制式电流传感器反馈回路只需提供一个较小电压保证零磁通即可。由于霍尔电流传感器在反馈环之外，不会影响零磁通的检测，所以电流的测量精度和稳定性不会受到影响。
更进一步地，电流矫正模块包括：霍尔电流传感器芯片和信号调理电路；霍尔电流传感器芯片用于检测待测电流，并根据待测电流以及功率放大器向磁调制式电流传感器提供的前馈信号来控制磁调制磁芯快速退出或避免进入磁饱和的虚假平衡状态，使磁调制式电流传感器输出保持稳定；信号调理电路用于对霍尔电流传感器芯片及PI控制器的输出信号进行滤波与大小调整。
更进一步地，霍尔电流传感器芯片的数量为八个，且八个霍尔电流传感器芯片分为四组互相垂直且中心对称的设置在环形的磁调制式电流传感器内侧。
更进一步地，相敏解调滤波系统包括：选频电路、同步方波电路、二阶带通滤波器、模拟乘法器和二阶低通滤波器；选频电路的输入端作为相敏解调滤波系统的第一输入端，用于提取绕组输入信号中的二次谐波信号；同步方波电路的输入端作为相敏解调滤波系统的第二输入端，用于对激励电压进行处理并保证输出的相位参考方波信号与所述选频电路输出的二次谐波信号相位同步；二阶带通滤波器的输入端连接至选频电路的输出端，用于对二次谐波信号进行滤波；模拟乘法器的第一输入端连接至二阶带通滤波器的输出端，第二输入端连接至同步方波电路的输出端，用于对滤波后的二次谐波信号以及相位参考方波信号进行相敏解调后输出平衡调节信号；二阶低通滤波器的输入端连接至模拟乘法器的输出端，用于对平衡调节信号进行滤波后输出无高频噪声的平衡调节信号。
本发明所提供的多传感器融合的直流比较仪，与现有技术相比，由于采用了与霍尔电流传感器融合的设计，具有更强的动态响应能力，面对电 流幅值大范围的变化时都能快速响应，极大程度上降低了直流比较仪因磁芯饱和导致虚假平衡问题而故障的可能性；在虚假平衡已经产生的情况下，本发明中传感器融合设计可通过霍尔电流传感器的前馈输出电流令磁芯快速退出磁饱和状态，无需重启或长时间扫描系统，大大减少了直流比较仪恢复正常工作响应的时间。
【附图说明】
图1为本发明实施实例提供的基于多传感器融合的快响应直流比较仪的结构原理图；
图2为本发明实施例提供的快响应直流比较仪中电流矫正模块中霍尔电流传感器的位置关系示意图；
图3为本发明实施例提供的快响应直流比较仪中磁调制式电流传感器线圈的几何结构图；
图4为本发明实施例提供的快响应直流比较仪中相敏解调滤波系统与激励变压器的连接关系示意图；
图5为本发明实施例提供的相敏解调滤波系统的模块结构示意图。
在所附图中，1为磁调制式电流传感器，21为点阵式霍尔电流传感器，22为信号调理电路，3为激励变压器，4为交流检测滤波电路，5为相敏解调滤波系统，6为PI控制器，7为功率放大器。
【具体实施方式】
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本发明进行进一步详细说明。应当理解，此处所描述的具体实施实例仅仅用以解释本发明，并不用于限定本发明。
本发明提供了一种基于多传感器融合的快响应直流比较仪，可以在不影响直流比较仪的测量精度、稳定度、时漂和温漂等稳态性能的前提下改善直流比较仪的动态响应，避免了直流比较仪动态过程中发生虚假平衡，并实现由于带电合闸、过载等原因造成的虚假平衡状态的自动恢复，从而 实现动态直流大电流的精密测量。
图1示出了本发明实施实例提供的基于多传感器融合的快响应直流比较仪的结构，为了便于说明，仅示出了与本发明实施例相关的部分，详述如下：
基于多传感器融合的快响应直流比较仪包括：磁调制式电流传感器1，电流矫正模块，激励变压器3，交流检测滤波电路4、相敏解调滤波系统5，PI控制器6和功率放大器7；其中，磁调制式电流传感器1的励磁电压输入端连接至所述激励变压器3的输出端，反馈输入端连接至功率放大器7的输出端，用于通过励磁电压对所述磁调制式电流传感器1的磁芯进行励磁，由第一信号输出端输出平衡判断信号给相敏解调滤波系统5，由第二信号输出端输出交流感应电压信号给交流检测滤波电路4；电流矫正模块用于测量一次侧被测电流并获得前馈信号，将前馈信号与PI控制信号进行放大后输出用于控制磁调制磁芯快速退出或避免进入磁饱和的虚假平衡状态，且使磁调制式电流传感器输出保持稳定；激励变压器3的输入端用于连接工频交流电，第一输出端连接至磁调制式电流传感器1的励磁电压输入端，第二输出端连接至相敏解调滤波系统5的第二输入端，用于为磁调制式电流传感器1提供周期性的励磁电压，并向相敏解调滤波系统5提供所需方波信号；交流检测滤波电路4的输入端连接至磁调制式电流传感器1的第二信号输出端，用于对磁调制式电流传感器1输出的交流感应电压信号进行检测与滤波；相敏解调滤波系统5的第一输入端连接至磁调制式电流传感器1的第一输出端，第二输入端连接至激励变压器3的第三输出端，用于对平衡判断信号进行相敏解调处理并输出平衡调节信号；PI控制器6第一输入端连接至交流检测滤波电路4的输出端，第二输入端连接至相敏解调滤波系统5的输出端，用于对滤波后的交流感应电压信号与平衡调节信号进行比例积分控制并输出稳定的PI控制信号；功率放大器7的输入端连接至电流矫正模块的输出端，用于对前馈信号与PI控制信号进行功率放大与 滤波处理后输出反馈信号。
本发明通过采用霍尔电流传感器和磁调制式电流传感器融合的方式，霍尔传感器根据一次侧被测电流的大小实时给磁芯一个反向的磁通，可以防止磁芯饱和，或者在已经饱和时，把磁芯从饱和状态“拖”出来。这样，仪器可以测量与承受的电流变化范围比原来单独磁调制式电流传感器时大了很多，即“动态响应性能提高”。由于直流比较仪的“响应时间”，就是磁芯中磁势达到平衡的时间，霍尔传感器的输出不像磁调制式电流传感器里面有很复杂的闭环，而是一有一次侧被测电流立即向磁芯提供反向磁势，所以可以令平衡更快，也即“快响应”。并且由于不需要重启或扫描，整体工作性能大大提高。
在本发明实施例中，电流矫正模块用于通过电流传感器测量一次侧被测电流，所得传感器输出正比于一次侧被测电流的前馈信号以及PI控制器6输出的PI控制信号连接至信号调理电路输入端，经滤波、调幅处理后，前馈信号与PI控制信号输出至功率放大器6输入端，前馈信号与PI控制信号经放大后向磁调制式电流传感器1提供前馈输入与闭环反馈输入，控制磁调制磁芯快速退出或避免进入磁饱和的虚假平衡状态，使磁调制式电流传感器输出保持稳定。
如图2所示，作为本发明的一个实施例，电流矫正模块可以利用传感器对磁调制式电流传感器进行前馈矫正。具体地，电流矫正模块包括：霍尔电流传感器芯片21和信号调理电路22；霍尔电流传感器芯片21用于检测待测电流，并根据待测电流以及功率放大器向磁调制式电流传感器提供的前馈信号来控制磁调制磁芯快速退出或避免进入磁饱和的虚假平衡状态，使磁调制式电流传感器输出保持稳定；信号调理电路22用于对霍尔电流传感器芯片的输出信号进行滤波与大小调整。
其中，霍尔电流传感器芯片21的数量可以为八个，且八个霍尔电流传感器芯片21分为四组互相垂直且中心对称的设置在环形的磁调制式电流传 感器内侧。
在本发明实施例中，霍尔电流传感器芯片21还可以设置在磁调制式电流传感器外侧。
在本发明实施例中，激励变压器、磁调制磁芯及其绕组作为直流比较仪的直流反馈通路，用于检测直流和低频磁场分量；磁调制式电流传感器采用差分结构，第一绕组与第二绕组反向串联绕制在两个磁调制磁芯上，用于检测与输出磁调制磁芯中的磁通大小；激励变压器的副边分别与第一绕组和第二绕组相连接；两个磁调制磁芯中的激励电流引起的交变磁通由于第一、第二绕组反向串联绕制，时刻保持大小相等方向相反，它们的对外表现合成磁通为零，所以激励电流不会使激励变压器中产生输出。
激励变压器3的副边分别与磁调制式传感器1的第一绕组W 1和第二绕组W 2相连接，第一绕组W 1和第二绕组W 2输出端连接相敏解调滤波系统，磁调制式传感器的感应绕组W 3输出端连接交流检测滤波电路，相敏解调滤波系统与交流检测滤波电路输出端连接PI控制器，PI控制器输出端连接电流矫正模块中信号调理电路输入端，其输出端连接功率放大器，功率放大器输出端连接磁调制式传感器的二次侧绕组W S。
调制磁芯材料通常为矩形系数较高、矫顽力较小的铁磁材料，如晶粒取向的冷轧硅钢片、坡莫合金、非晶等。调制磁芯形状通常为圆环形，其直径由被测母线、二次侧绕组匝数等因素综合决定，需保证直流大电流测量时母线可以从调制铁芯圆环内部穿过。调制铁芯截面积的选择应保证铁芯的细长比，例如，可以选择圆环平均周长和截面积之比大于50，最优效果根据实验确定。本实施例中，调制磁芯采用高磁导率非晶合金绕制，传感器平均直径可以为1124mm。
图3示出了本发明实施例提供的磁调制式电流传感器线圈几何结构图，第一绕组W 1和第二绕组W 2反向串联绕制在第一磁调制磁芯C 1和第二磁调制磁芯C 2上，用于检测与输出磁调制磁芯中的磁通大小；感应绕组W 3绕 制在交流反馈磁芯C 3上，用于检测交流反馈磁芯中的磁通大小。
作为本发明的一个实施例，第一绕组W 1和第二绕组W 2结构参数保持一致，且反向串联绕制，第一磁调制磁芯C 1和第二磁调制磁芯C 2中的激励电流引起的交变磁通时刻保持大小相等方向相反，它们的对外表现合成磁通为零，所以激励电流不会使所述激励变压器中产生输出。
调制磁芯磁特性由材料决定，第一绕组W 1和第二绕组W 2的匝数一般可以取1000匝，根据励磁电流大小确定线径，根据国家标准密绕线圈电流密度不大于2.5A/mm 2，再根据输出信号大小确定输出电阻阻值和激励电压值。本实例中第一磁调制磁芯C 1和第二磁调制磁芯C 2截面尺寸为10mm×8mm，其对应的激励第一绕组W 1和第二绕组W 2采用0.2mm漆包线绕制，匝数为1000匝，激励绕组匝数没有特殊要求，只要能承载峰值磁势时所需电流安匝数即可。
铁芯屏蔽层厚度通常为5mm，采用高导磁材料带材绕制，如晶粒取向的冷轧硅钢片、坡莫合金、非晶等。铁芯屏蔽层与调制铁芯之间的空隙用电气灌封胶或其他绝缘填充物填充。
二次侧绕组W S绕制在第一磁调制磁芯C 1和第二磁调制磁芯C 2与交流反馈磁芯C 3上。其作用为通过流过二次侧电流I 2产生直流磁势与一次侧被测电流I 1磁势平衡，通过二次侧电流的大小计算出一次侧被测电流值。二次绕组匝数取决于一次侧被测电流大小和二次侧电流额定值。
根据零磁通原理，当磁势平衡时有N 1I 1＝N 2I 2，进行电流测量时一次侧被测电流母线穿过磁芯，故N 1＝1，即I 2＝(N 1I 1)/N 2。线径的选取，根据国家标准密绕线圈电流密度不大于2.5A/mm 2和I 2确定。本实施例中测量最大电流为30kA，取绕组匝数为10000，采用线径为1.2mm的漆包线进行绕组。
交流反馈磁芯C 3及其感应绕组作为直流比较仪的交流反馈通路，用于检测中频交流磁场分量及抑制系统感应纹波。感应绕组绕制在交流反馈磁芯上，用于检测交流反馈磁芯中的磁通大小；感应绕组经过交流检测滤波 电路后的输出电压和磁调制式电流传感器的输出电压和作为PI控制器的控制量，经过功率放大器，驱动二次侧绕组形成零磁通闭环反馈控制。
交流反馈磁芯C 3同时作为交流检测磁芯、互感器磁芯和磁调制磁芯的磁屏蔽，其截面壁厚为10mm，截面高度为45mm，截面宽度为28mm，磁芯左上角有1mm绝缘填充以防止形成绕线方向上的回路产生涡流通路。交流检测感应绕组W 3是开路状态没有电流流过，所以对线径没有要求，本实施例中采用0.2mm漆包线进行绕制，绕组匝数与交流检测支路的灵敏度相关，但是其输出信号还需要经过信号调理电路，所以匝数的选取不十分严格，本实施例中选取为1000匝。反馈绕组的设计需要综合考虑量程和二次侧电流的测量以确定绕组匝数，即一次电流和二次侧电流的变比。匝数多意味着二次侧电流小，有利于二次侧电流的测量，同时也意味着绕组电阻和电感大，这会增加功率放大器的设计难度，而且匝数多也会增加绕组的分部参数。
本发明基于多传感器融合的快响应直流比较仪的激励放大器选取工频50Hz作为磁调制激励频率，激励源直接采用市电经过带有中心抽头的工频变压器获得。
图4示出了本发明实施例提供的快响应直流比较仪中相敏解调滤波系统5与激励变压器3的连接关系；T 1、T 2、T 3、T 4分别是激励第一绕组W 1和第二绕组W 2的接入端，两个绕组的差分电压经过双T选频网络和二阶带通滤波器输出二次谐波电压v(2f)，其中心频率f 1＝100Hz，品质因数Q 1＝5。相敏解调所需方波由激励电压整流后通过比较器产生，然后经过相位调整补偿滤波器产生的相位误差，保证二次谐波电压v(2f)和方波电压v(s)的相位同步。v(2f)和v(s)经过模拟乘法器AD633进行相敏解调，然后经过二阶低通滤波器输出，低通滤波器截止频率为20Hz，品质因数Q 2＝0.5。
图5示出了本发明实施例提供的相敏解调滤波系统的模块结构，相敏 解调滤波系统5包括：选频电路51、同步方波电路52、二阶带通滤波器53、模拟乘法器54和二阶低通滤波器55；选频电路51的输入端作为相敏解调滤波系统5的第一输入端，用于提取绕组输入信号中的二次谐波信号；同步方波电路52的输入端作为相敏解调滤波系统5的第二输入端，用于对激励电压进行处理并保证输出的相位参考方波信号与选频电路输出的二次谐波信号相位同步；二阶带通滤波器53的输入端连接至选频电路51的输出端，用于对二次谐波信号进行滤波；模拟乘法器54的第一输入端连接至二阶带通滤波器53的输出端，第二输入端连接至同步方波电路52的输出端，用于对滤波后的二次谐波信号以及相位参考方波信号进行相敏解调后输出平衡调节信号；二阶低通滤波器55的输入端连接至模拟乘法器的输出端，用于对平衡调节信号进行滤波后输出无高频噪声的平衡调节信号。
现详细说明PI控制器6和功率放大器7的功能作用，交流检测绕组输出信号v ac经过20kHz二阶低滤波器以消除噪声干扰；v ac和v o共同作为PI控制器输入，PI控制器参数为K p＝0.6，τ＝0.22s；PI控制器和霍尔电流传感器共同输出至功率放大器。
功率放大器7可以采用AB类线性放大器，其由8路并联的MJL4302和MJL4281互补对管组成，供电电压为±140V；为了增长系统稳定性，引入了由RC局部负反馈网络；设计完成后功率放大器增益KPA＝14。经功率放大器调整后的电压信号输出至磁调制式电流传感器二次绕组Ws。
为了更进一步的说明本发明实施例提供的基于多传感器融合的快响应直流比较仪，现结合附图详述其工作过程如下：
为了便于叙述，暂不考虑频段过渡阶段各支路的耦合作用。首先介绍正常工作状态时，即当被测一次电流值不变或在小范围内变化时，动态过程中磁芯不发生饱和，互感作用不受磁芯饱和的影响，整个测量系统可以看成是线性系统时，直流比较仪的工作过程。
当一次侧电流I 1在较低频段(包括直流)时，一般为f ex/2以下，此时 交流检测感应绕组W 3输出电压为零且一次侧被测电流回路和二次绕组之间没有互感作用；其中，f ex为激励变压器的激励电压频率，一般为50Hz。此时，点阵式霍尔电流传感器检测一次侧电流，输出恒定电压，通过信号调理电路后输入功率放大器；磁调制式放大器输出经过PI控制器与信号调理电路后输入功率放大器。两个传感器融合的总输出经功率放大器后，经由二次绕组W S和采样电阻R M形成闭环回路。当测量系统处于稳定状态时，即有N 1I 1＝N 2I 2，磁芯中被测磁势为零，此时磁调制式放大器输出信号为零，PI控制器的积分作用使其保持稳定输出；而霍尔电流传感器输出仅与被测一次电流有关，故稳定输出不变。此时被测一次电流大小I 1＝(N 2I 2)/N 1。
当一次侧电流I 1在中高频段时，点阵式霍尔电流传感器检测交流电流，输出恒定电压，通过信号调理电路后输入功率放大器；磁调制式放大器支路输出电压为零，不考虑一次侧被测电流回路和二次绕组之间的互感作用，一次侧被测电流在交流检测感应绕组W 3中产生感应电压，经过PI控制器与信号调理电路后输入功率放大器。此时，磁调制式电流传感器作为交流电流比较仪维持零磁通状态。调节过程与低频时类似，只是磁通误差控制信号来自交流检测感应绕组W 3的电磁感应电压及点阵式霍尔电流传感器输出电压。
实际正常工作时，上述三种工作状态相互耦合。当被测信号发生变化时，霍尔电流传感器的输出与磁调制式电流传感器的互感首先起作用，互感磁芯不饱和时，在霍尔传感器与互感作用下二次绕组内可以保持零磁通状态。随着电流时间乘积(安秒积)的增大互感磁芯逐渐饱和，霍尔传感器、磁调制式电流传感器和交流检测铁芯共同作用控制功率放大器输出，驱动二次绕组，维持零磁场状态。
非正常工作状态时，即一次侧被测电流的变化范围较大，动态过程中互感磁芯的饱和导致互感能力迅速下降，如果功率放大器不能及时产生输出，来维持二次侧电流，二次侧电流就会快速下降，进而导致不平衡电流 超出磁调制式放大器的线性范围，直流比较仪闭环系统失稳而进入虚假平衡状态。
当被测一次电流大范围变化，虚假平衡已经发生时，多传感器融合系统中磁调制式放大器因磁芯饱和无法工作，点阵式霍尔电流传感器输出根据被测一次电流发生同比例变化的电压，直接控制功率放大器驱动二次绕组，使磁调制式放大器的磁芯退出磁饱，回归正常工作状态。
当被测一次电流大范围变化，虚假平衡还未发生时，霍尔电流传感器支路起到了前馈作用，当被测一次电流发生变化时，点阵式霍尔传感器和磁调制式放大器的互感作用共同工作，功率放大器的即时输出避免了互感磁芯发生饱和，加强了磁调制式放大器互感能力，互感作用又可以使系统更快地进入零磁通状态，使直流比较仪具有更强的动态响应能力，面对电流幅值大范围的变化时都能快速响应。一次电流进入稳态后，霍尔传感器的输出给予功放一个静态工作点，磁调制式电流传感器反馈回路只需提供一个较小电压保证零磁通即可。
本领域的技术人员容易理解，以上所述仅为本发明的较佳实施例而已，并不用以限制本发明，凡在本发明的精神和原则之内所作的任何修改、等同替换和改进等，均应包含在本发明的保护范围之内。

Claims (9)

  1. 一种基于多传感器融合的快响应直流比较仪，其特征在于，包括：磁调制式电流传感器(1)、电流矫正模块、激励变压器(3)、交流检测滤波电路(4)、相敏解调滤波系统(5)、PI控制器(6)和功率放大器(7)；
    所述磁调制式电流传感器(1)的励磁电压输入端连接至所述激励变压器(3)的输出端，反馈输入端连接至所述功率放大器(7)的输出端，用于通过励磁电压对所述磁调制式电流传感器(1)的磁芯进行励磁，由第一信号输出端输出平衡判断信号给相敏解调滤波系统(5)，由第二信号输出端输出交流感应电压信号给交流检测滤波电路(4)；
    所述电流矫正模块用于测量一次侧被测电流并获得前馈信号，将所述前馈信号与PI控制信号进行放大后输出用于控制磁调制磁芯快速退出或避免进入磁饱和的虚假平衡状态，且使所述磁调制式电流传感器输出保持稳定；
    所述激励变压器(3)的输入端用于连接工频交流电，第一输出端连接至所述磁调制式电流传感器(1)的励磁电压输入端，第二输出端连接至所述相敏解调滤波系统(5)的第二输入端，用于为所述磁调制式电流传感器(1)提供周期性的励磁电压，并向所述相敏解调滤波系统(5)提供所需方波信号；
    所述交流检测滤波电路(4)的输入端连接至所述磁调制式电流传感器(1)的第二信号输出端，用于对所述磁调制式电流传感器(1)输出的交流感应电压信号进行检测与滤波；
    所述相敏解调滤波系统(5)的第一输入端连接至所述磁调制式电流传感器(1)的第一输出端，第二输入端连接至所述激励变压器(3)的第三输出端，用于对平衡判断信号进行相敏解调处理并输出平衡调节信号；
    所述PI控制器(6)第一输入端连接至所述交流检测滤波电路(4)的 输出端，第二输入端连接至所述相敏解调滤波系统(5)的输出端，用于对滤波后的交流感应电压信号与所述平衡调节信号进行比例积分控制并输出稳定的PI控制信号；
    所述功率放大器(7)的输入端连接至所述电流矫正模块的输出端，用于对所述前馈信号与PI控制信号进行功率放大与滤波处理后输出所述反馈信号。
  2. 如权利要求1所述的快响应直流比较仪，其特征在于，所述磁调制式电流传感器(1)包括：第一磁调制磁芯C 1、第二磁调制磁芯C 2、交流反馈磁芯C 3、第一绕组W 1、第二绕组W 2、交流感应绕组W 3和二次侧绕组W S；
    所述第一磁调制磁芯C 1和所述第二磁调制磁芯C 2相互平行设置，且所述第一磁调制磁芯C 1和所述第二磁调制磁芯C 2分别由绝缘材料包裹后设置于所述交流反馈磁芯C 3中；
    所述第一绕组W 1绕制在所述第一磁调制磁芯C 1上，所述第二绕组W 2绕制在所述第二磁调制磁芯C 2上，且所述第一绕组W 1与所述第二绕组W 2串联，用于检测所述第一磁调制磁芯C 1和所述第二磁调制磁芯C 2中的磁通大小，并输出平衡判断信号；
    所述感应绕组W 3绕制于所述交流反馈磁芯C 3上，所述感应绕组W 3用于检测交流反馈磁芯中的磁通大小，并输出交流感应电压信号；
    所述二次侧绕组W S绕制在包裹了所述交流反馈磁芯C 3的绝缘材料上，用于保证通过二次侧电流I 2产生的直流磁势与一次侧被测电流I 1的磁势平衡，且根据二次侧电流的大小计算待测电流值。
  3. 如权利要求2所述的快响应直流比较仪，其特征在于，所述第一绕组W 1和所述第二绕组W 2反向串联，且所述第一磁调制磁芯C 1和所述第二磁调制磁芯C 2中的激励电流引起的交变磁通时刻保持大小相等方向相反。
  4. 如权利要求2或3所述的快响应直流比较仪，其特征在于，所述第 一磁调制磁芯C 1和所述第二磁调制磁芯C 2的材料均为矩形系数较高、矫顽力较小的铁磁材料。
  5. 如权利要求4所述的快响应直流比较仪，其特征在于，所述第一磁调制磁芯C 1和所述第二磁调制磁芯C 2的材料均为晶粒取向的冷轧硅钢片、坡莫合金或非晶合金。
  6. 如权利要求2-5任一项所述的快响应直流比较仪，其特征在于，所述第一磁调制磁芯C 1和所述第二磁调制磁芯C 2的形状为圆环形，其直径由被测母线和二次侧绕组匝数决定，需保证直流大电流测量时母线可以从调制铁芯圆环内部穿过。
  7. 如权利要求1或2所述的快响应直流比较仪，其特征在于，所述电流矫正模块包括：霍尔电流传感器芯片(21)和信号调理电路(22)；
    所述霍尔电流传感器芯片(21)用于检测待测电流，并根据所述待测电流以及所述功率放大器(7)向所述磁调制式电流传感器(1)提供的前馈信号来控制磁调制磁芯快速退出或避免进入磁饱和的虚假平衡状态，使所述磁调制式电流传感器输出保持稳定；
    所述信号调理电路(22)用于对所述霍尔电流传感器芯片(21)及所述PI控制器(6)的输出信号进行滤波与大小调整。
  8. 如权利要求7所述的快响应直流比较仪，其特征在于，所述霍尔电流传感器芯片(21)的数量为八个，且八个霍尔电流传感器芯片(21)分为四组互相垂直且中心对称的设置在环形的磁调制式电流传感器内侧。
  9. 如权利要求1-8任一项所述的快响应直流比较仪，其特征在于，所述相敏解调滤波系统(5)包括：选频电路(51)、同步方波电路(52)、二阶带通滤波器(53)、模拟乘法器(54)和二阶低通滤波器(55)；
    所述选频电路(51)的输入端作为所述相敏解调滤波系统(5)的第一输入端，用于提取绕组输入信号中的二次谐波信号；
    所述同步方波电路(52)的输入端作为所述相敏解调滤波系统(5)的 第二输入端，用于对激励电压进行处理并保证输出的相位参考方波信号与所述选频电路输出的二次谐波信号相位同步；
    所述二阶带通滤波器(53)的输入端连接至所述选频电路(51)的输出端，用于对所述二次谐波信号进行滤波；
    所述模拟乘法器(54)的第一输入端连接至所述二阶带通滤波器(53)的输出端，第二输入端连接至所述同步方波电路(52)的输出端，用于对滤波后的二次谐波信号以及相位参考方波信号进行相敏解调后输出平衡调节信号；所述二阶低通滤波器(55)的输入端连接至所述模拟乘法器的输出端，用于对平衡调节信号进行滤波后输出无高频噪声的平衡调节信号。
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