WO2021199304A1 - Wireless power transmission device - Google Patents

Wireless power transmission device Download PDF

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Publication number
WO2021199304A1
WO2021199304A1 PCT/JP2020/014870 JP2020014870W WO2021199304A1 WO 2021199304 A1 WO2021199304 A1 WO 2021199304A1 JP 2020014870 W JP2020014870 W JP 2020014870W WO 2021199304 A1 WO2021199304 A1 WO 2021199304A1
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Prior art keywords
resonance
current
power transmission
switching
circuit
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PCT/JP2020/014870
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French (fr)
Japanese (ja)
Inventor
淳史 田中
井戸　寛
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マクセルホールディングス株式会社
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Application filed by マクセルホールディングス株式会社 filed Critical マクセルホールディングス株式会社
Priority to PCT/JP2020/014870 priority Critical patent/WO2021199304A1/en
Priority to JP2021508010A priority patent/JP7066046B2/en
Publication of WO2021199304A1 publication Critical patent/WO2021199304A1/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/10—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling
    • H02J50/12—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling of the resonant type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode

Definitions

  • the present invention relates to a wireless power transmission device that transmits power to a power receiving device in a non-contact manner, that is, wirelessly.
  • an electromagnetic induction type wireless power transmission device using electromagnetic induction between a power transmission coil provided in the power transmission device and a power reception coil provided in the power reception device is known.
  • This electromagnetic induction type wireless power transmission device in order to improve the power transmission efficiency, it is necessary to arrange the coils on the power transmission side and the power reception side close to each other, and there is a problem that the distance at which power can be wirelessly transmitted is short. Have.
  • Magnetic resonance coupling is a non-radiative and coupling type power transmission principle, in which the power transmission coil of the power transmission resonance circuit and the power reception coil of the power reception resonance circuit are coupled (resonant) by a magnetic field in a state where the power transmission resonance circuit and the power reception resonance circuit resonate with each other. ), And power transmission is performed.
  • the magnetic field resonance coupling is more efficient than the electromagnetic induction method, and can transmit power with high efficiency even when a large air gap or misalignment occurs.
  • the magnetic resonance coupling method by matching the resonance frequencies of the power transmission resonance circuit and the power reception resonance circuit and operating the power transmission resonance circuit in a state where the impedance is optimized, high-efficiency power transmission is possible even if the coupling coefficient is very small.
  • the magnetic resonance coupling type wireless power transmission device includes a power transmission device having a power transmission resonance circuit composed of a power transmission coil and a capacitor, and a power reception device having a power reception resonance circuit composed of a power reception coil and a capacitor. Power is transmitted from the power transmitting device to the power receiving device in a non-contact manner by utilizing the fact that the coil and the power receiving coil resonate in a magnetic field. That is, when the power transmission resonance circuit and the power reception resonance circuit resonate at the resonance frequency in the magnetic field resonance coupling state, high power transmission efficiency can be obtained.
  • Patent Document 1 describes that in a magnetic field resonance coupling type non-contact power transmission device, the resonance frequencies of the power transmission resonance circuit and the power reception resonance circuit are matched, and the frequency of the drive voltage of the power transmission resonance circuit is also matched with the resonance frequency. , A technique for magnetically resonating a power transmitting coil and a power receiving coil is disclosed.
  • the resonance frequency of the power transmission resonance circuit, the resonance frequency of the power reception resonance circuit, and the drive frequency of the power transmission resonance circuit are all the aged deterioration of the circuit element, the operating environment temperature, the distance between the power transmission coil and the power reception coil, and the like. It fluctuates due to various factors such as the surrounding environment. For example, as the distance between the power transmission coil and the power reception coil deviates from the optimum distance, the mutual inductance changes, the coupling coefficient changes, and the resonance frequency also deviates slightly. Further, when a foreign substance such as a metal piece is inserted between the power transmission coil and the power reception coil, the coupling coefficient changes and the resonance frequency changes. Further, the Q value of the resonance circuit also changes due to the fluctuation of the power load of the power receiving device, which contributes to the fluctuation of the resonance frequency.
  • Patent Document 1 discloses that the output frequency of the drive voltage of the transmission resonance circuit can be freely adjusted, but the output frequency of the drive voltage is adjusted according to the fluctuation of the resonance frequency of the transmission resonance circuit and the power reception resonance circuit. Is not disclosed to automatically follow. Therefore, it is considered that the power transmission resonance circuit operates in a state where the drive voltage frequency is set to a predetermined value, but there is a problem that the power transmission efficiency is lowered if the drive frequency of the power transmission resonance circuit remains a constant value.
  • Patent Document 2 describes a switching element that outputs a drive voltage of a predetermined frequency to a transmission resonance circuit, a current detection unit that detects a current flowing through the transmission resonance circuit, and a time from when the switching element is turned on to a current zero crossing point.
  • a magnetic resonance coupling type non-contact power transmission device including a timer for measuring the voltage and a frequency setting unit for setting the frequency of the drive voltage of the transmission resonance circuit based on the time measured by the timer is disclosed.
  • Patent Documents 3 and 4 disclose a technique of detecting the zero crossing point of the resonance current of the power transmission resonance circuit, switching the switching element at the timing of the zero crossing of the resonance current, and continuing the resonance state.
  • Japanese Unexamined Patent Publication No. 2010-130878 Japanese Unexamined Patent Publication No. 2012-120253 Japanese Unexamined Patent Publication No. 2019-54629 Japanese Unexamined Patent Publication No. 2018-183056
  • Patent Documents 2 to 4 are based on the premise that the zero crossing point of the resonance current of the power transmission resonance circuit can be accurately detected, but in reality, the current detection is delayed. Further, after detecting the zero crossing point, there is a delay in the switching operation of the switching element by the command from the control unit.
  • the switching element is driven at a frequency lower than the optimum driving frequency because a delay occurs in principle.
  • the delay is not a problem, but when the drive frequency is relatively high, the delay of the above-mentioned current detection circuit and drive circuit cannot be ignored, and the frequency deviates from the optimum resonance frequency.
  • the transmission power is lowered and the transmission efficiency is lowered.
  • the power transmission power will be increased and the power transmission efficiency will be improved by creating a high resonance state by using a power transmission resonance circuit consisting of a power transmission coil and a capacitor.
  • a power transmission resonance circuit consisting of a power transmission coil and a capacitor.
  • the drive frequency deviates from the resonance frequency of the transmission resonance circuit, the transmission power will decrease and the transmission efficiency will decrease.
  • the switching circuit is accurately switched and operated at the zero crossing point of the resonance current of the power transmission resonance circuit, so that a high frequency of a more appropriate frequency is used. It is an object of the present invention to provide a wireless transmission device capable of supplying a voltage to a transmission resonance circuit.
  • the wireless power transmission device is a wireless power transmission device that transmits power to a power receiving device in a non-contact manner, and uses a resonance circuit having a transmission coil and a capacitor and power supplied from a power source to transmit high voltage to the resonance circuit.
  • a switching circuit configured to be able to alternately switch between a first switching mode and a second switching mode in order to apply a voltage, a control unit that controls switching of the switching mode of the switching circuit, and the resonance circuit.
  • a current detection unit for detecting the generated resonance current is provided.
  • the control unit Based on the value detected by the current detection unit of the resonance current, the control unit performs the next zero crossing point of the actual current waveform of the resonance current before the actual current waveform of the resonance current reaches the next zero crossing point. It may be provided with a current waveform prediction unit for predicting.
  • the zero crossing point of the high-frequency current waveform means a time point at which the high-frequency current becomes zero when the high-frequency current changes from positive to negative or negative to positive, that is, a predetermined time on the time axis.
  • control unit may be configured to switch the switching mode of the switching circuit at the predicted next zero crossing point. If it is predicted that the next zero crossing point will be the time T, the control unit may be configured to switch the switching mode of the switching circuit at the time T, but the switching mode may be switched only slightly. In consideration of this delay, the time T is the time when the switching mode of the switching circuit is switched by configuring the control unit to output the switching mode switching control command of the switching circuit immediately before the time T. It is preferable to match with.
  • the current waveform prediction unit can be configured to sample the current detected by the current detection unit at predetermined time intervals that are sufficiently small with respect to the resonance current. According to this, the current waveform of the resonance current can be grasped more accurately.
  • the sampling interval can be preferably 1/16 or less, more preferably 1/32 or less, and further preferably 1/64 or less of one cycle of the resonance current.
  • the current waveform predictor predicts the next zero cross point based on the measurement time from a predetermined time point to the detection of the peak of the detected current waveform based on the detected value of the resonance current. It may be configured as follows. For example, the measurement time is t, the time corresponding to the delay of the peak detection operation of the detected current waveform by the current detection circuit is ⁇ t, and the time when 2 ⁇ (t ⁇ t) elapses from a predetermined time point is set as the next zero crossing point. Can be predicted.
  • the predetermined time point may be the time point when the control unit performs the switching immediately before the switching mode, or the time when the current waveform prediction unit outputs the switching mode switching command most recently. It may be the previous zero cross point predicted for the previous switching mode switching. In any case, appropriate correction processing can be performed in predicting the next zero crossing point.
  • the current waveform predictor may be configured to detect the peak of the detected current waveform of the resonance current based on the differential value of the detected current waveform, or the absolute value of the detected current may increase. It can also be detected by detecting that the current state has changed to a decrease. In consideration of the influence of noise and the like, it is also possible to set the peak detection condition that the absolute value of the detected current starts to decrease and then continuously decreases a predetermined number of times. In this case, since the current detection time of a predetermined number of times has elapsed, it is preferable to calculate the measurement time assuming that the time point before the lapse of the current detection time of the predetermined number of times is the peak of the detected current waveform.
  • the current waveform prediction unit is next to the actual current waveform of the resonance current based on the time when the absolute value of the detection value of the resonance current by the current detection unit transitions from the reference value or more to less than the reference value. It can also be configured to predict the zero cross point. According to this, it is determined whether or not the transition from the reference value or more to less than the reference value is performed in a region where the amount of current fluctuation per unit time is larger than that near the peak of the current waveform, preferably immediately before the next zero crossing point of current. be able to.
  • the reference value may be a constant value regardless of the amplitude of the resonance current. However, even if the amplitude of the resonance current fluctuates due to various factors, the time from the time when the absolute value of the detected value of the resonance current transitions from the reference value or more to less than the reference value to the current zero crossing point is made uniform. Therefore, preferably, the reference value for the previous period can be obtained by calculation based on the peak value immediately before the detection current waveform based on the detection value of the resonance current. For example, the reference value can be obtained by multiplying the absolute value of the peak value immediately before the detected current waveform by a predetermined coefficient greater than 0 and less than 1. The predetermined coefficient can be determined in consideration of the delay in the detection operation of the current detection circuit, the delay in the operation of the control unit and the switching circuit, and the like.
  • the zero cross point of the actual resonance current waveform of the resonance circuit is predicted in advance, and the switching mode of the switching circuit is switched at the predicted zero cross point to transmit a high frequency voltage having a more appropriate frequency. It can be applied to a resonance circuit, and the transmitted power can be improved or the transmission efficiency can be improved.
  • FIG. 1 shows the configuration of a non-contact power transmission device according to an embodiment of the present invention.
  • This device is composed of a power transmitting device 10 and a power receiving device 20, and transmits electric power from the power transmitting device 10 to the power receiving device 20 by a magnetic resonance coupling method in a non-contact manner.
  • one power transmitting device 10 and one power receiving device 20 operate as a pair, but one power transmitting device 10 can be configured to transmit power to a plurality of power receiving devices 20, and a plurality of power receiving devices 20 can be transmitted.
  • the power transmission device 10 may be configured to transmit electric power to one power receiving device 20, or the plurality of power transmitting devices 10 and the plurality of power receiving devices 20 may be configured to be usable at the same time.
  • the power receiving device 20 uses a power receiving resonance circuit 21 that receives power transmitted from the power transmitting resonance circuit 13 of the power transmitting device 10, a rectifier circuit 22 that rectifies the received power and converts it into DC power, and a rectified DC power. It is provided with an output unit 23 that outputs to a built-in or external load.
  • the power receiving device 20 may have the same device configuration as the power transmitting device 10, and the switching circuit 12 is made into a rectifier circuit 22 composed of a diode bridge by turning off all the switching elements Q1 to Q4 of the switching circuit 12 described later. Can be made to work.
  • the power receiving resonance circuit 21 is configured as an LC resonance circuit in which a power receiving coil and a capacitor are connected in series or in parallel.
  • the configuration is such that the resonance frequency is substantially the same as that of the power transmission resonance circuit 13, and the magnetic field generated from the power transmission resonance circuit 13 is received and converted into AC power.
  • the rectifier circuit 22 rectifies the AC power received by the power receiving resonance circuit 21 and converts it into DC power.
  • the rectifier circuit 22 is composed of a diode bridge or the like.
  • the output unit 23 can be configured to perform smoothing, voltage conversion, or the like as necessary when outputting electric power to the load.
  • the power transmission device 10 includes a power source 11, a switching circuit 12 that converts the power supplied from the power source 11 into a high-frequency drive power for non-contact transmission, and a transmission coil L that converts the high-frequency drive power output by the switching circuit 12 into a magnetic field. It is provided with a power transmission resonance circuit 13 having a power transmission resonance circuit 13, a current detection circuit 14 for detecting a resonance current flowing through the power transmission resonance circuit 13, and a control unit 15 for generating and outputting a drive command signal voltage of the switching circuit 12. Further, the control unit 15 performs switching drive control of the switching circuit 12 at the current waveform prediction unit 16 that predicts the zero cross point of the resonance current based on the detection value of the resonance current by the current detection circuit 14 and the predicted current zero cross point. It includes a drive control unit 17.
  • the power supply 11 is typically a power supply that supplies DC power, and can be composed of a converter that converts a commercial AC power supply into a DC power supply, a storage battery, or the like.
  • the switching circuit 12 is configured to be able to alternately switch between a first switching mode and a second switching mode in order to apply a high frequency voltage to the power transmission resonance circuit 13 using the electric power supplied from the power supply 11.
  • the switching circuit 12 is composed of an inverter having a half-bridge or full-bridge configuration, and has a voltage polarity applied to the transmission coil L in the first switching mode and a voltage applied to the transmission coil L in the second switching mode.
  • the polarity can be configured to be positive or negative inverted.
  • FIG. 2 shows a simplified circuit configuration example of a power transmission device 10 having a switching circuit 12 composed of a full-bridge inverter, and the switching circuit 12 includes four FETs as switching elements Q1 to Q4. More specifically, the first leg on the left side of the drawing and the second leg on the right side of the drawing are connected in parallel between the power supply 11 and the ground, and the first switching element is attached to the upper arm (power supply side arm) of the first leg.
  • Q1 is provided
  • a second switching element Q2 is provided on the lower arm (ground side arm) of the first leg
  • a third switching element Q3 is provided on the upper arm of the second leg
  • the lower arm of the second leg is provided. Is provided with a fourth switching element Q4.
  • Each switching element Q1 to Q4 operates on / off according to the drive command signal voltage output from the drive control unit 17.
  • the first and fourth switching elements Q1 and Q4 are turned on, and the second and third switching elements Q2 and Q3 are turned off, whereby the power supply voltage is operated via the switching element Q1.
  • the second and third switching elements Q2 and Q3 are turned on, and the first and fourth switching elements Q1 and Q4 are turned off, whereby the power supply voltage is operated via the switching element Q3. Is supplied to the other terminal of the power transmission resonant circuit.
  • a dead time is provided to turn off all the switching elements Q1 to Q4 in order to prevent the power supply from being short-circuited to the ground.
  • a high-frequency drive voltage composed of a square wave is typically output to the resonance circuit 13 by continuing the on or off operation of each switching element, but the switching element to be turned on is controlled by PWM.
  • PWM pulse-driving, a high-frequency drive voltage having a waveform close to a sine wave may be output to the resonance circuit 13.
  • the switching elements Q1 to Q4 are composed of NMOSFETs
  • the body diodes included in the NMOSFETs are all arranged in a direction in which a current flows only from the ground side to the power supply side.
  • the switching circuit 12 functions as a full-wave rectifier composed of four diodes and rectifies the high-frequency power on the resonance circuit 13 side. It can be supplied to the power supply 11 side. Therefore, it is also possible to operate the power transmission device 10 as a power receiving device.
  • the transmission resonance circuit 13 is configured as an LC resonance circuit in which a transmission coil L and a capacitor C are connected in series, and when a high-frequency drive voltage generated by the switching circuit 12 is supplied, a resonance voltage and a resonance current having the same period are transmitted. A high-frequency magnetic field is generated in the coil L.
  • the resonance current generated in the transmission coil L has substantially the same phase as the high frequency drive voltage, but the resonance voltage has a lead phase of 90 degrees.
  • the current detection circuit 14 is a circuit that detects the resonance current flowing through the resonance circuit 13, and may be an appropriate circuit such as a method using a shunt resistor, a method using a current sensor, or a method using a Hall element.
  • the potential difference between both ends of the shunt resistor R connected in series with the power transmission coil L is amplified by the amplifier 14a, and the output voltage is input to the analog signal input terminal of the control unit 15 as the detection value of the resonance current. ..
  • the control unit 15 predicts the next zero cross point of the actual current waveform of the resonance current before the actual current waveform of the resonance current reaches the next zero cross point based on the detected value of the resonance current by the current detection circuit 14.
  • a waveform prediction unit 16 and a drive control unit 17 that outputs a drive command signal voltage to each of the switching elements Q1 to Q4 so as to switch the switching mode of the switching circuit at the next zero crossing point predicted are provided.
  • the drive control unit 17 drives the switching control units 17a that generate and output the control signals of the switching elements Q1 to Q4, and the switching control units Q1 to Q4 based on the control signals output by the switching control units 17a. It is equipped with a gate driver 17b that outputs a command signal voltage.
  • the current waveform prediction unit 16 and the switching control unit 17a are shown as functional block diagrams, but these are configured by a single microprocessor, FPGA, or a circuit integrally mounted on a substrate. It may be composed of physically different circuits. Further, the switching control unit 17a and the gate driver 17b may also be configured by a driver LSI having these functions integrally, or may be configured as physically different circuits. Further, the current waveform prediction unit 16, the switching control unit 17a, and the gate driver 17b may be configured by a single driver LSI or the like.
  • the current waveform prediction unit 16 is a circuit unit that predicts in advance the next zero crossing point of the resonance current waveform as the optimum timing for switching based on the detection value of the resonance current by the current detection circuit 14, and is the core of the present invention. It is the part that makes up.
  • the current waveform prediction unit 16 is preferably configured by a microprocessor having an analog input and a digital output, but can also be configured by an FPGA or an individual electronic circuit.
  • the current waveform prediction unit 16 may be configured to output a switching mode switching command to the drive control unit 17 at the timing when the predicted current zero cross point is reached, or data on the predicted current zero cross point may be output. It can also be configured to be passed to the drive control unit 17 in advance so that the drive control unit 17 voluntarily switches the switching mode at the current zero crossing point.
  • the drive control unit 17 generates signals necessary for driving the switching elements Q1 to Q4 of the switching circuit 12. That is, a voltage for driving the gates of the switching elements Q1 to Q4 of the switching circuit 12 is generated, and a dead time is generated so that a through current does not flow in the switching circuit 12.
  • the drive control unit 17 may be configured such that the dead time processing and other correction processing of the switching circuit 12 are performed by the switching control unit 17a or the gate driver 17b.
  • the power transmission device 10 detects the resonance current of the power transmission resonance circuit 13 and performs a self-excited oscillation operation in which the next switching is performed using the detected resonance current waveform. However, since switching is not performed at the start of the operation, self-excitation is performed. Oscillation operation does not start. Therefore, after the start, the switching circuit 12 is driven at a predetermined initial frequency to start power transmission (S101). As the initial frequency, the resonance frequency f obtained from the rated inductance of the power transmission coil L and the rated capacity of the capacitor C can be used. It is also possible to store the frequency at the time of the previous power transmission operation and start power transmission using that frequency as the initial frequency. In one example, the resonance frequency of the power transmission resonance circuit 13 may be 70 to 100 kHz.
  • the power transmission operation is continued until a predetermined power transmission end condition (for example, when a power transmission end operation is performed or an abnormality is detected) is satisfied (S104).
  • a predetermined power transmission end condition for example, when a power transmission end operation is performed or an abnormality is detected
  • the switching waveform shows the gate signals (drive command signal voltage) of the first and fourth switching elements Q1 and Q4, and the gate signals of the second and third switching elements Q2 and Q3 are those of Q1 and Q4. It is not shown because it has the opposite phase of the gate signal.
  • the switching waveform when switching is performed without delay at the current zero crossing point, the switching waveform has no delay, and the resonance current waveform becomes the target current waveform.
  • the target current waveform is in an ideal resonance state in which switching is performed at a frequency that matches the resonance frequency when the frequency characteristics of the transmission resonance circuit 13 fluctuate due to various external factors and magnetic resonance coupling with the power reception resonance circuit 21. It is a resonance current waveform at the time.
  • the next half cycle is also operated in the same manner, and the gate signals of Q2 and Q3 are switched from on to off and the gates of Q1 and Q4 are switched from off to on at the zero crossing point where the resonance current changes from negative to positive.
  • the resonance state continues at an ideal frequency that matches the resonance frequency of the resonator.
  • the conventional power transmission device performs the next switching immediately after detecting the zero crossing point of the resonance current waveform, as shown by the solid line in FIG. 4, before switching is performed from the current zero crossing point of the actual resonance current waveform.
  • a delay ⁇ t occurs. Due to this delay ⁇ t, switching is not performed even if the current zero cross point is exceeded, and as a result, the actual resonance current waveform is distorted and its period becomes longer by the amount corresponding to the delay ⁇ t.
  • the current detection circuit 14 takes a certain amount of time to detect a change in current, and the delay ⁇ t becomes large especially when a current transformer or a Hall element is used as a current sensor. In the method using a shunt resistor, the delay ⁇ t is relatively small, but when an element such as an operational amplifier 14a is used, a delay always occurs.
  • the delay ⁇ t since the delay ⁇ t also occurs at the next current zero crossing point, the delay ⁇ t accumulates every half cycle, and the actual resonance current waveform cycle is larger than the ideal target current waveform cycle. Is also 2 ⁇ t longer. When there is a delay ⁇ t in the current detection in this way, switching continues at a frequency lower than the ideal resonance frequency.
  • FIG. 6 is a graph in which both the power transmission resonance circuit and the power reception resonance circuit are graphs in which the power transmission power and the power transmission efficiency are measured while changing the switching frequency using a non-contact wireless power transmission device in which the resonance frequency is adjusted to 85 kHz. At 85 kHz, both transmission power and transmission efficiency are maximum. When the drive control method of the conventional zero-cross point detection type power transmission device is used as it is, it operates at 84.3 kHz, which is lower than the ideal resonance frequency, which is the maximum power transmission power and power transmission efficiency. It fell below the maximum value.
  • the delay time changes depending on the current sensor and drive control circuit used, the frequency of self-excited oscillation changes, but since there is always a delay, it always operates at a frequency lower than the ideal resonance frequency, and the transmitted power Both transmission efficiency will decrease.
  • the delay of the drive control unit 17 is relatively not a problem, but when the frequency is high, the delay of the operation of the drive control circuit 17 is also a problem.
  • the resonance current waveform when the control unit 15 including the current waveform prediction unit 16 according to the present embodiment is used is shown in FIG. 5 in comparison with the resonance current waveform by the conventional current zero cross point detection method.
  • the control unit 15 according to the present embodiment predicts the resonance current waveform based on the detection value of the resonance current by the current detection circuit 14, and considers the delay of the current sensor 14a and the drive control circuit 17, and actually considers the resonance current waveform.
  • the drive command signal voltage is output to the switching circuit 12 so that the switching can be performed accurately at the zero crossing point.
  • FIG. 7 shows an example of the operation flowchart of the current waveform prediction unit 16, and the operation of the current waveform prediction unit 16 is performed after the start of the self-excited oscillation operation (S103) shown in FIG.
  • the current waveform prediction unit 16 starts counting for measuring the time from a predetermined time point, for example, the latest last switching command output time point with respect to the drive control unit 17 (S201). Since the last switching command output does not exist at the time of the first operation, it is possible to measure the time from an appropriate time point such as measuring the time from the start time of the self-excited oscillation operation.
  • the peak detection method of the resonance current waveform may be appropriate, but for example, the time when it is detected that the absolute value of the detected current has changed from an increasing state to a decreasing state is determined as the peak of the resonance current waveform. can. At this time, when it is assumed that noise is mixed in the current waveform, it can be determined that the peak is detected when the absolute value of the detected current decreases continuously a plurality of times.
  • the time when the differential value of the resonance current waveform satisfies a predetermined condition can be determined as the peak of the resonance current waveform. Since the resonance current waveform is a roughly sine wave, the differentiated waveform is a roughly cosine wave. Therefore, as an example of the predetermined condition, it is the time when the differential value becomes close to 0. Since the fact that the differential value becomes almost 0 coincides with the peak time of the current waveform, waiting until the differential value becomes almost 0 means waiting until the peak of the current waveform.
  • the counter for time measurement is read out, and as shown in step (1) in FIG. 5, the time from the latest last switching command output time to the peak of the resonance current waveform is calculated. (S203).
  • the zero cross point of the next resonance current waveform is predicted as shown in step (2) in FIG. 5 (S204).
  • the time from the previous switching command output time to the peak of the resonance current waveform is t, and the current sensor delay and the drive control circuit delay are ⁇ t
  • the ideal switching waveform has no delay from the previous switching command output time.
  • the period until the next zero crossing point of the resonance current waveform can be calculated by (t ⁇ t) ⁇ 2. Further, the period may be set in consideration of the switching loss of the semiconductor.
  • the drive control unit 17 waits until the predicted current zero cross time (S205), and at the predicted current zero cross time as shown in step (3) in FIG. A switching command signal is output to (S206).
  • the current waveform prediction unit 16 repeatedly continues the above operations of steps S201 to S206 during the power transmission operation.
  • high-power and highly efficient wireless power transmission can be performed by continuing switching with a waveform equivalent to the switching waveform without delay and controlling wireless power transmission by the resonance frequency of the resonant circuit.
  • the current waveform prediction unit 16 can also be configured to predict the next zero cross point by a method other than peak detection of the resonance current waveform. For example, if it is detected that the absolute value of the resonance current detected by the current detection circuit 14 has transitioned from a predetermined threshold value A or more to less than the threshold value A, a switching command is immediately output to the drive control unit 17, and the following It can be configured to predict that switching will occur exactly at the zero-current crossing point of.
  • the threshold A may be a fixed value set in advance, but resonance is performed so that switching can be performed accurately at the next zero crossing point even when the amplitude of the resonance current waveform fluctuates due to operating conditions or the like. It can be assumed that the amplitude of the peak immediately before the current waveform is multiplied by a predetermined coefficient greater than 0 and less than 1. The coefficient is obtained when the degree of delay in the operation of the current detection circuit 14 and the drive control unit 17 is measured in advance and it is detected that the absolute value of the detected value of the resonance current has transitioned from the threshold value A or more to the threshold value A or less. By outputting the switching command to the drive control unit 17 at the immediate timing, it is possible to set so that the switching is accurately performed at the next current zero crossing point by just canceling the above delay.
  • the present invention is not limited to the above embodiment, and the design can be changed as appropriate.
  • the present invention is applied to a magnetic field resonance coupling type wireless power transmission device, but the present invention can also be applied to an electromagnetic induction type wireless power transmission device having a resonance circuit on the power transmission side.
  • the present invention can be particularly suitably used for a wireless power transmission device such as an underwater vehicle, for example, a submarine or an underwater drone, in which the optimum power transmission frequency tends to change due to a position change in seawater.
  • Wireless power transmission device 11 Power supply 12 Switching circuit 13 Power transmission resonance circuit 14 Current detection circuit 15 Control unit 16 Current waveform prediction unit 17 Drive control unit

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Abstract

The present disclosure provides a wireless power transmission device for transmitting power to a power reception device in a non-contact manner.　A wireless power transmission device 10 includes: a resonance circuit 13 having a power transmission coil L and a capacitor C; a switching circuit 12 which can switch a switching mode in order to apply a high-frequency voltage to the resonance circuit 13 by using power supplied from a power supply 11; a control unit 15 which performs switching control of the switching circuit 12; and a current detection unit 14 which detects resonance current generated in the resonance circuit 13. The control unit 15 comprises a current waveform prediction unit 16 which, on the basis of the resonance current detection value of the current detection unit 14, predicts the next zero cross point of the actual current waveform of the resonance current before the actual current waveform of the resonance current reaches the next zero cross point. The control unit operates so as to switch the switching mode of the switching circuit 12 at the predicted next zero cross point.

Description

ワイヤレス送電装置Wireless power transmission device
　本発明は、非接触、すなわちワイヤレスで受電装置に対して電力を伝送するワイヤレス送電装置に関する。 The present invention relates to a wireless power transmission device that transmits power to a power receiving device in a non-contact manner, that is, wirelessly.
　近年、電気自動車や産業用機器、携帯用電子機器等への非接触電力伝送技術が注目を浴びている。特に、電動歯ブラシや電気シェーバー等の水まわりで使う電化製品やコードレス電話機、携帯電話機等の分野においてこの技術が重宝され、一部の製品において実用化されている。 In recent years, non-contact power transmission technology for electric vehicles, industrial equipment, portable electronic equipment, etc. has been attracting attention. In particular, this technology is useful in the fields of electric appliances such as electric toothbrushes and electric shavers used around water, cordless telephones, mobile phones, etc., and has been put into practical use in some products.
　現在実用化されている非接触電力伝送装置として、送電装置に設けられた送電コイルと、受電装置に設けられた受電コイルとの間での電磁誘導を利用した電磁誘導型ワイヤレス電力伝送装置が知られている。この電磁誘導型ワイヤレス電力伝送装置においては、電力の伝送効率を高めるために、送電側と受電側の各々のコイルを近接させて配置させる必要があり、電力を無線伝送できる距離が短いという課題を有する。 As a non-contact power transmission device currently in practical use, an electromagnetic induction type wireless power transmission device using electromagnetic induction between a power transmission coil provided in the power transmission device and a power reception coil provided in the power reception device is known. Has been done. In this electromagnetic induction type wireless power transmission device, in order to improve the power transmission efficiency, it is necessary to arrange the coils on the power transmission side and the power reception side close to each other, and there is a problem that the distance at which power can be wirelessly transmitted is short. Have.
　そこで、数ｍ離れた機器にワイヤレスで電力を供給する技術も開発されている。それは送電コイルと受電コイルとの磁界共振結合（磁界共鳴）を利用して電力伝送を行う磁界共振結合式ワイヤレス電力伝送技術である。 Therefore, a technology to wirelessly supply power to devices several meters away has also been developed. It is a magnetic field resonance coupling type wireless power transmission technology that transmits power by utilizing the magnetic field resonance coupling (magnetic field resonance) between the power transmission coil and the power reception coil.
　磁界共振結合は非放射型かつ結合型の電力伝送原理であり、送電共振回路及び受電共振回路がそれぞれ共振した状態において送電共振回路の送電コイルと受電共振回路の受電コイルとが磁界によって結合（共鳴）して、電力伝送を行なう。磁界共振結合は、電磁誘導方式に比して高効率であり、大きなエアギャップや位置ずれが生じた場合でも高効率の電力伝送が可能である。磁界共振結合方式は、送電共振回路及び受電共振回路の共振周波数を一致させ、インピーダンスを最適化した状態で動作させることにより、結合係数が非常に小さくても高効率の電力伝送が可能となる。 Magnetic resonance coupling is a non-radiative and coupling type power transmission principle, in which the power transmission coil of the power transmission resonance circuit and the power reception coil of the power reception resonance circuit are coupled (resonant) by a magnetic field in a state where the power transmission resonance circuit and the power reception resonance circuit resonate with each other. ), And power transmission is performed. The magnetic field resonance coupling is more efficient than the electromagnetic induction method, and can transmit power with high efficiency even when a large air gap or misalignment occurs. In the magnetic resonance coupling method, by matching the resonance frequencies of the power transmission resonance circuit and the power reception resonance circuit and operating the power transmission resonance circuit in a state where the impedance is optimized, high-efficiency power transmission is possible even if the coupling coefficient is very small.
　磁界共振結合式のワイヤレス電力伝送装置は、送電コイル及びコンデンサにより構成された送電共振回路を有する送電装置と、受電コイル及びコンデンサにより構成された受電共振回路を有する受電装置とを備えており、送電コイルと受電コイルとが磁界的に共振することを利用して送電装置から受電装置に非接触で電力を伝送する。すなわち、送電共振回路と受電共振回路とが、磁界共振結合状態における共振周波数で共振するとき、高い電力伝送効率が得られる。 The magnetic resonance coupling type wireless power transmission device includes a power transmission device having a power transmission resonance circuit composed of a power transmission coil and a capacitor, and a power reception device having a power reception resonance circuit composed of a power reception coil and a capacitor. Power is transmitted from the power transmitting device to the power receiving device in a non-contact manner by utilizing the fact that the coil and the power receiving coil resonate in a magnetic field. That is, when the power transmission resonance circuit and the power reception resonance circuit resonate at the resonance frequency in the magnetic field resonance coupling state, high power transmission efficiency can be obtained.
　特許文献１には、磁界共振結合型の非接触電力伝送装置において、送電共振回路及び受電共振回路の共振周波数を一致させるとともに、送電共振回路の駆動電圧の周波数をも共振周波数に一致させることにより、送電コイルと受電コイルとを磁界的に共振させる技術が開示されている。 Patent Document 1 describes that in a magnetic field resonance coupling type non-contact power transmission device, the resonance frequencies of the power transmission resonance circuit and the power reception resonance circuit are matched, and the frequency of the drive voltage of the power transmission resonance circuit is also matched with the resonance frequency. , A technique for magnetically resonating a power transmitting coil and a power receiving coil is disclosed.
　しかしながら、送電共振回路の共振周波数、受電共振回路の共振周波数、及び、送電共振回路の駆動周波数はいずれも、回路素子の経年劣化や動作環境温度、送電コイルと受電コイルとの間の距離やその他の周囲環境等の様々な要因によって変動する。例えば、送電コイルと受電コイルとの距離が最適距離からずれるにつれて相互インダクタンスが変化して結合係数が変化し、共振周波数も微妙にずれていく。また、送電コイルと受電コイルとの間に金属片などの異物が挿入された場合にも結合係数が変化し、共振周波数が変化してしまう。また、受電装置の電力負荷の変動によっても共振回路のＱ値が変化し、共振周波数の変動の一因となる。 However, the resonance frequency of the power transmission resonance circuit, the resonance frequency of the power reception resonance circuit, and the drive frequency of the power transmission resonance circuit are all the aged deterioration of the circuit element, the operating environment temperature, the distance between the power transmission coil and the power reception coil, and the like. It fluctuates due to various factors such as the surrounding environment. For example, as the distance between the power transmission coil and the power reception coil deviates from the optimum distance, the mutual inductance changes, the coupling coefficient changes, and the resonance frequency also deviates slightly. Further, when a foreign substance such as a metal piece is inserted between the power transmission coil and the power reception coil, the coupling coefficient changes and the resonance frequency changes. Further, the Q value of the resonance circuit also changes due to the fluctuation of the power load of the power receiving device, which contributes to the fluctuation of the resonance frequency.
　特許文献１には、送電共振回路の駆動電圧の出力周波数を自由に調整可能であることが開示されているが、送電共振回路及び受電共振回路の共振周波数の変動に合わせて駆動電圧の出力周波数も自動追従することは開示されていない。したがって、送電共振回路の駆動電圧周波数が所定値に設定された状態で動作するものと考えられるが、送電共振回路の駆動周波数が一定値のままでは送電効率の低下が発生する問題がある。 Patent Document 1 discloses that the output frequency of the drive voltage of the transmission resonance circuit can be freely adjusted, but the output frequency of the drive voltage is adjusted according to the fluctuation of the resonance frequency of the transmission resonance circuit and the power reception resonance circuit. Is not disclosed to automatically follow. Therefore, it is considered that the power transmission resonance circuit operates in a state where the drive voltage frequency is set to a predetermined value, but there is a problem that the power transmission efficiency is lowered if the drive frequency of the power transmission resonance circuit remains a constant value.
　特許文献２には、所定周波数の駆動電圧を送電共振回路に出力するスイッチング素子と、送電共振回路を流れる電流を検出する電流検出部と、スイッチング素子をオン動作させてから電流ゼロクロス点までの時間を計測するタイマーと、タイマーにより計測される時間に基づいて送電共振回路の駆動電圧の周波数を設定する周波数設定部とを備える磁界共振結合型の非接触電力伝送装置が開示されている。 Patent Document 2 describes a switching element that outputs a drive voltage of a predetermined frequency to a transmission resonance circuit, a current detection unit that detects a current flowing through the transmission resonance circuit, and a time from when the switching element is turned on to a current zero crossing point. A magnetic resonance coupling type non-contact power transmission device including a timer for measuring the voltage and a frequency setting unit for setting the frequency of the drive voltage of the transmission resonance circuit based on the time measured by the timer is disclosed.
　また、特許文献３及び４には、送電共振回路の共振電流のゼロクロス点を検出して、共振電流のゼロクロスのタイミングでスイッチング素子の切り替えを行い、共振状態を継続させる技術が開示されている。 Further, Patent Documents 3 and 4 disclose a technique of detecting the zero crossing point of the resonance current of the power transmission resonance circuit, switching the switching element at the timing of the zero crossing of the resonance current, and continuing the resonance state.
特開２０１０－１３０８７８号公報Japanese Unexamined Patent Publication No. 2010-130878 特開２０１２－１２０２５３号公報Japanese Unexamined Patent Publication No. 2012-120253 特開２０１９－５４６２９号公報Japanese Unexamined Patent Publication No. 2019-54629 特開２０１８－１８３０５６号公報Japanese Unexamined Patent Publication No. 2018-183056
　特許文献２～４に開示された技術は、送電共振回路の共振電流のゼロクロス点を正確に検出可能であることを前提とするものであるが、実際には電流検出には遅れが生じる。さらに、ゼロクロス点を検出した後、制御部からの指令によるスイッチング素子の切り換え動作にも遅れが生じる。 The techniques disclosed in Patent Documents 2 to 4 are based on the premise that the zero crossing point of the resonance current of the power transmission resonance circuit can be accurately detected, but in reality, the current detection is delayed. Further, after detecting the zero crossing point, there is a delay in the switching operation of the switching element by the command from the control unit.
　したがって、共振電流のゼロクロス時点を検出してからスイッチング素子を駆動する方法では、原理的に遅れが生じるため、最適な駆動周波数よりも低い周波数で駆動してしまうという問題がある。駆動周波数が比較的低い場合には相対的に遅れが問題とならないが、駆動周波数が比較的高くなると、上記した電流検出回路や駆動回路の遅れが無視できなくなり、最適な共振周波数からずれた周波数で駆動が行なわれる結果、送電電力の低下や送電効率の低下が発生する。 Therefore, in the method of driving the switching element after detecting the zero crossing point of the resonance current, there is a problem that the switching element is driven at a frequency lower than the optimum driving frequency because a delay occurs in principle. When the drive frequency is relatively low, the delay is not a problem, but when the drive frequency is relatively high, the delay of the above-mentioned current detection circuit and drive circuit cannot be ignored, and the frequency deviates from the optimum resonance frequency. As a result of being driven by the frequency, the transmission power is lowered and the transmission efficiency is lowered.
　また、電磁誘導型のワイヤレス送電装置においても、送電コイルとコンデンサとによる送電共振回路を利用して高い共振状態を作り出すことで、送電電力の増加や電力伝送効率の向上を期待できるが、この場合も送電共振回路の共振周波数に対し駆動周波数がずれると送電電力の低下や送電効率の低下が発生する。 Also, in an electromagnetic induction type wireless power transmission device, it is expected that the power transmission power will be increased and the power transmission efficiency will be improved by creating a high resonance state by using a power transmission resonance circuit consisting of a power transmission coil and a capacitor. However, if the drive frequency deviates from the resonance frequency of the transmission resonance circuit, the transmission power will decrease and the transmission efficiency will decrease.
　本発明は、送電装置を構成する各回路の動作に遅れが生じる場合であっても、送電共振回路の共振電流のゼロクロス点で正確にスイッチング回路を切り換え動作させることにより、より適切な周波数の高周波電圧を送電共振回路に供給し得るワイヤレス送電装置を提供することを目的とする。 According to the present invention, even when the operation of each circuit constituting the power transmission device is delayed, the switching circuit is accurately switched and operated at the zero crossing point of the resonance current of the power transmission resonance circuit, so that a high frequency of a more appropriate frequency is used. It is an object of the present invention to provide a wireless transmission device capable of supplying a voltage to a transmission resonance circuit.
　本発明によるワイヤレス送電装置は、非接触により受電装置へ電力を伝送するワイヤレス送電装置であって、送電コイルとコンデンサとを有する共振回路と、電源から供給される電力を用いて前記共振回路に高周波電圧を印加させるべく第１のスイッチングモードと第２のスイッチングモードとに交互に切り換え可能に構成されたスイッチング回路と、前記スイッチング回路の前記スイッチングモードの切換制御を行う制御部と、前記共振回路に生じる共振電流を検出する電流検出部と、を備える。 The wireless power transmission device according to the present invention is a wireless power transmission device that transmits power to a power receiving device in a non-contact manner, and uses a resonance circuit having a transmission coil and a capacitor and power supplied from a power source to transmit high voltage to the resonance circuit. A switching circuit configured to be able to alternately switch between a first switching mode and a second switching mode in order to apply a voltage, a control unit that controls switching of the switching mode of the switching circuit, and the resonance circuit. A current detection unit for detecting the generated resonance current is provided.
　前記制御部は、前記電流検出部による前記共振電流の検出値に基づいて、前記共振電流の実際の電流波形が次のゼロクロス点に至る前に前記共振電流の実際の電流波形の次のゼロクロス点を予測する電流波形予測部を備えていてよい。なお、本明細書において、高周波電流波形のゼロクロス点とは、高周波電流が正から負、或いは負から正に変化する際にゼロとなる時点、すなわち、時間軸上の所定の時点を意味する。 Based on the value detected by the current detection unit of the resonance current, the control unit performs the next zero crossing point of the actual current waveform of the resonance current before the actual current waveform of the resonance current reaches the next zero crossing point. It may be provided with a current waveform prediction unit for predicting. In the present specification, the zero crossing point of the high-frequency current waveform means a time point at which the high-frequency current becomes zero when the high-frequency current changes from positive to negative or negative to positive, that is, a predetermined time on the time axis.
　さらに、前記制御部は、予測した次のゼロクロス点で前記スイッチング回路のスイッチングモードを切り換えるよう構成されていてよい。なお、次のゼロクロス点が時刻Ｔであると予測した場合、時刻Ｔとなった時点でスイッチング回路のスイッチングモードの切り換えを行うよう制御部を構成してもよいが、スイッチングモードの切り換えにも僅かに遅れが生じるため、この遅れを考慮して時刻Ｔとなる直前にスイッチング回路のスイッチングモードの切換制御指令を出力するよう制御部を構成することにより、スイッチング回路のスイッチングモードが切り替わる時点を時刻Ｔに合致させることが好ましい。 Further, the control unit may be configured to switch the switching mode of the switching circuit at the predicted next zero crossing point. If it is predicted that the next zero crossing point will be the time T, the control unit may be configured to switch the switching mode of the switching circuit at the time T, but the switching mode may be switched only slightly. In consideration of this delay, the time T is the time when the switching mode of the switching circuit is switched by configuring the control unit to output the switching mode switching control command of the switching circuit immediately before the time T. It is preferable to match with.
　好ましくは、電流波形予測部は、共振電流に対して十分に小さい所定時間毎に電流検出部による検出電流をサンプリングするよう構成することができる。これによれば、共振電流の電流波形をより正確に把握できる。サンプリング間隔は、好ましくは共振電流の１周期の１／１６以下、より好ましくは１／３２以下、さらに好ましくは１／６４以下とすることができる。 Preferably, the current waveform prediction unit can be configured to sample the current detected by the current detection unit at predetermined time intervals that are sufficiently small with respect to the resonance current. According to this, the current waveform of the resonance current can be grasped more accurately. The sampling interval can be preferably 1/16 or less, more preferably 1/32 or less, and further preferably 1/64 or less of one cycle of the resonance current.
　本発明によるワイヤレス送電装置において、前記電流波形予測部は、所定の時点から前記共振電流の検出値に基づく検出電流波形のピークが検出されるまでの計測時間に基づいて次のゼロクロス点を予測するよう構成されていてよい。例えば、前記計測時間をｔ、電流検出回路による検出電流波形のピーク検出動作の遅れに相当する時間をΔｔとして、所定の時点から２×（ｔ－Δｔ）を経過した時点を次のゼロクロス点として予測することができる。 In the wireless power transmission device according to the present invention, the current waveform predictor predicts the next zero cross point based on the measurement time from a predetermined time point to the detection of the peak of the detected current waveform based on the detected value of the resonance current. It may be configured as follows. For example, the measurement time is t, the time corresponding to the delay of the peak detection operation of the detected current waveform by the current detection circuit is Δt, and the time when 2 × (t−Δt) elapses from a predetermined time point is set as the next zero crossing point. Can be predicted.
　前記所定の時点は、前記制御部が前記スイッチングモードの直前の切換えを行った時点であってもよいし、電流波形予測部がスイッチングモードの切り換え指令を直近で最後に出力した時点であってもよいし、また、直前のスイッチングモードの切り換えのために予測された直前のゼロクロス点であってもよい。いずれの場合でも、次のゼロクロス点を予測するにあたり、適切な補正処理を行うことができる。 The predetermined time point may be the time point when the control unit performs the switching immediately before the switching mode, or the time when the current waveform prediction unit outputs the switching mode switching command most recently. It may be the previous zero cross point predicted for the previous switching mode switching. In any case, appropriate correction processing can be performed in predicting the next zero crossing point.
　前記電流波形予測部は、前記共振電流の前記検出電流波形のピークを、当該検出電流波形の微分値に基づいて検出するよう構成されていてもよいし、また、検出電流の絶対値が増加していく状態から減少に転じたことを検出することにより検出することもできる。ノイズ等の影響を考慮して、検出電流の絶対値が減少に転じた後、連続して所定回数減少したことを検出したことをピーク検出条件とすることもできる。この場合、所定回数の電流検出時間が経過しているため、この所定回数の電流検出時間経過前の時点が検出電流波形のピークであるとして上記計測時間を算定することが好ましい。 The current waveform predictor may be configured to detect the peak of the detected current waveform of the resonance current based on the differential value of the detected current waveform, or the absolute value of the detected current may increase. It can also be detected by detecting that the current state has changed to a decrease. In consideration of the influence of noise and the like, it is also possible to set the peak detection condition that the absolute value of the detected current starts to decrease and then continuously decreases a predetermined number of times. In this case, since the current detection time of a predetermined number of times has elapsed, it is preferable to calculate the measurement time assuming that the time point before the lapse of the current detection time of the predetermined number of times is the peak of the detected current waveform.
　また、前記電流波形予測部は、前記電流検出部による前記共振電流の検出値の絶対値が基準値以上から前記基準値未満へ遷移した時点に基づいて前記共振電流の実際の電流波形の次のゼロクロス点を予測するよう構成することもできる。これによれば、電流波形のピーク近傍よりも単位時間あたりの電流変動量が大きな領域、好ましくは次の電流ゼロクロス点の直前で基準値以上から基準値未満へ遷移したか否かの判定を行うことができる。 Further, the current waveform prediction unit is next to the actual current waveform of the resonance current based on the time when the absolute value of the detection value of the resonance current by the current detection unit transitions from the reference value or more to less than the reference value. It can also be configured to predict the zero cross point. According to this, it is determined whether or not the transition from the reference value or more to less than the reference value is performed in a region where the amount of current fluctuation per unit time is larger than that near the peak of the current waveform, preferably immediately before the next zero crossing point of current. be able to.
　前記基準値は、共振電流の振幅にかかわらず一定の値であってもよい。しかし、種々の要因によって共振電流の振幅が変動しても共振電流の検出値の絶対値が基準値以上から前記基準値未満へ遷移した時点から電流ゼロクロス点までの時間が均一となるようにするために、好ましくは、前記共振電流の検出値に基づく検出電流波形の直前のピーク値に基づいて演算により前期基準値を求めることができる。例えば、検出電流波形の直前のピーク値の絶対値に、０より大きく１未満の所定の係数を乗じることによって、基準値を求めることができる。所定の係数は、電流検出回路の検出動作の遅れや、制御部およびスイッチング回路の動作の遅れ等を考慮して定めることができる。 The reference value may be a constant value regardless of the amplitude of the resonance current. However, even if the amplitude of the resonance current fluctuates due to various factors, the time from the time when the absolute value of the detected value of the resonance current transitions from the reference value or more to less than the reference value to the current zero crossing point is made uniform. Therefore, preferably, the reference value for the previous period can be obtained by calculation based on the peak value immediately before the detection current waveform based on the detection value of the resonance current. For example, the reference value can be obtained by multiplying the absolute value of the peak value immediately before the detected current waveform by a predetermined coefficient greater than 0 and less than 1. The predetermined coefficient can be determined in consideration of the delay in the detection operation of the current detection circuit, the delay in the operation of the control unit and the switching circuit, and the like.
　本発明によれば、共振回路の実際の共振電流波形のゼロクロス点を事前に予測して、予測したゼロクロス点においてスイッチング回路のスイッチングモードの切り換えを行うことによって、より適切な周波数の高周波電圧を送電共振回路に印加することができ、送電電力の向上、若しくは、送電効率の向上を図ることができる。 According to the present invention, the zero cross point of the actual resonance current waveform of the resonance circuit is predicted in advance, and the switching mode of the switching circuit is switched at the predicted zero cross point to transmit a high frequency voltage having a more appropriate frequency. It can be applied to a resonance circuit, and the transmitted power can be improved or the transmission efficiency can be improved.
本発明の一実施形態に係るワイヤレス送電装置の概略ブロック構成図である。It is a schematic block block diagram of the wireless power transmission device which concerns on one Embodiment of this invention. 同ワイヤレス送電装置の概略回路図である。It is a schematic circuit diagram of the wireless power transmission device. 同ワイヤレス送電装置の全体動作フローチャートである。It is an overall operation flowchart of the wireless power transmission device. 従来の送電装置における共振電流波形及びスイッチング波形を示す波形図である。It is a waveform diagram which shows the resonance current waveform and the switching waveform in the conventional power transmission apparatus. 本発明による送電装置における共振電流波形及びスイッチング波形を示す波形図である。It is a waveform diagram which shows the resonance current waveform and the switching waveform in the power transmission apparatus by this invention. スイッチング周波数を変更しながら送電電力と送電効率とを測定したグラフである。It is a graph which measured transmission power and transmission efficiency while changing a switching frequency. 実施形態に係る電流波形予測部のフローチャートである。It is a flowchart of the current waveform prediction part which concerns on embodiment.
　図１は本発明の一実施形態に係る非接触電力伝送装置の構成を示す。この装置は、送電装置１０と受電装置２０とにより構成され、磁界共振結合方式で非接触により送電装置１０から受電装置２０へ電力を伝送する。なお、典型的には一つの送電装置１０と一つの受電装置２０が対となって動作するが、一つの送電装置１０から複数の受電装置２０に電力伝送可能に構成することもできるし、複数の送電装置１０から一つの受電装置２０に電力伝送するよう構成してもよいし、また、複数の送電装置１０と複数の受電装置２０とを同時利用可能に構成することもできる。 FIG. 1 shows the configuration of a non-contact power transmission device according to an embodiment of the present invention. This device is composed of a power transmitting device 10 and a power receiving device 20, and transmits electric power from the power transmitting device 10 to the power receiving device 20 by a magnetic resonance coupling method in a non-contact manner. Typically, one power transmitting device 10 and one power receiving device 20 operate as a pair, but one power transmitting device 10 can be configured to transmit power to a plurality of power receiving devices 20, and a plurality of power receiving devices 20 can be transmitted. The power transmission device 10 may be configured to transmit electric power to one power receiving device 20, or the plurality of power transmitting devices 10 and the plurality of power receiving devices 20 may be configured to be usable at the same time.
　受電装置２０は、送電装置１０の送電共振回路１３から伝送される電力を受電する受電共振回路２１と、受電した電力を整流して直流電力に変換する整流回路２２と、整流された直流電力を内蔵もしくは外部の負荷へ出力する出力部２３とを備えている。なお、受電装置２０は、送電装置１０と同様の装置構成であってよく、後述するスイッチング回路１２のスイッチング素子Ｑ１～Ｑ４をすべてオフ動作させることによってスイッチング回路１２をダイオードブリッジからなる整流回路２２として機能させることができる。 The power receiving device 20 uses a power receiving resonance circuit 21 that receives power transmitted from the power transmitting resonance circuit 13 of the power transmitting device 10, a rectifier circuit 22 that rectifies the received power and converts it into DC power, and a rectified DC power. It is provided with an output unit 23 that outputs to a built-in or external load. The power receiving device 20 may have the same device configuration as the power transmitting device 10, and the switching circuit 12 is made into a rectifier circuit 22 composed of a diode bridge by turning off all the switching elements Q1 to Q4 of the switching circuit 12 described later. Can be made to work.
　受電共振回路２１は、受電コイルとコンデンサとを直列または並列に接続したＬＣ共振回路として構成される。送電共振回路１３と共振周波数が略同一となるような構成とし、送電共振回路１３から発生する磁界を受け取り、交流電力に変換する。整流回路２２は受電共振回路２１が受け取った交流電力を整流し直流電力に変換する。整流回路２２はダイオードブリッジなどにより構成される。出力部２３は、負荷へ電力を出力する際に、必要に応じて平滑や電圧変換等を行うよう構成できる。 The power receiving resonance circuit 21 is configured as an LC resonance circuit in which a power receiving coil and a capacitor are connected in series or in parallel. The configuration is such that the resonance frequency is substantially the same as that of the power transmission resonance circuit 13, and the magnetic field generated from the power transmission resonance circuit 13 is received and converted into AC power. The rectifier circuit 22 rectifies the AC power received by the power receiving resonance circuit 21 and converts it into DC power. The rectifier circuit 22 is composed of a diode bridge or the like. The output unit 23 can be configured to perform smoothing, voltage conversion, or the like as necessary when outputting electric power to the load.
　送電装置１０は、電源１１と、電源１１から供給される電力を非接触送電用の高周波駆動電力に変換するスイッチング回路１２と、スイッチング回路１２が出力する高周波駆動電力を磁界に変換する送電コイルＬを有する送電共振回路１３と、送電共振回路１３に流れる共振電流を検出する電流検出回路１４と、スイッチング回路１２の駆動指令信号電圧を生成出力する制御部１５とを備えている。さらに、制御部１５は、電流検出回路１４による共振電流の検出値に基づいて共振電流のゼロクロス点を予測する電流波形予測部１６と、予測した電流ゼロクロス点でスイッチング回路１２のスイッチング駆動制御を行う駆動制御部１７とを備えている。 The power transmission device 10 includes a power source 11, a switching circuit 12 that converts the power supplied from the power source 11 into a high-frequency drive power for non-contact transmission, and a transmission coil L that converts the high-frequency drive power output by the switching circuit 12 into a magnetic field. It is provided with a power transmission resonance circuit 13 having a power transmission resonance circuit 13, a current detection circuit 14 for detecting a resonance current flowing through the power transmission resonance circuit 13, and a control unit 15 for generating and outputting a drive command signal voltage of the switching circuit 12. Further, the control unit 15 performs switching drive control of the switching circuit 12 at the current waveform prediction unit 16 that predicts the zero cross point of the resonance current based on the detection value of the resonance current by the current detection circuit 14 and the predicted current zero cross point. It includes a drive control unit 17.
　電源１１は典型的には直流電力を供給する電源であり、商用交流電源を直流電源に変換するコンバータや、蓄電池などから構成できる。 The power supply 11 is typically a power supply that supplies DC power, and can be composed of a converter that converts a commercial AC power supply into a DC power supply, a storage battery, or the like.
　スイッチング回路１２は、電源１１から供給される電力を用いて送電共振回路１３に高周波電圧を印加させるべく第１のスイッチングモードと第２のスイッチングモードとに交互に切り換え可能に構成されている。典型的には、スイッチング回路１２はハーフブリッジもしくはフルブリッジ構成のインバータにより構成され、第１のスイッチングモードにおける送電コイルＬへの印加電圧極性と、第２のスイッチングモードにおける送電コイルＬへの印加電圧極性とが、正負反転するよう構成できる。 The switching circuit 12 is configured to be able to alternately switch between a first switching mode and a second switching mode in order to apply a high frequency voltage to the power transmission resonance circuit 13 using the electric power supplied from the power supply 11. Typically, the switching circuit 12 is composed of an inverter having a half-bridge or full-bridge configuration, and has a voltage polarity applied to the transmission coil L in the first switching mode and a voltage applied to the transmission coil L in the second switching mode. The polarity can be configured to be positive or negative inverted.
　図２はフルブリッジインバータからなるスイッチング回路１２を有する送電装置１０の簡略回路構成例を示しており、このスイッチング回路１２は４個のＦＥＴをスイッチング素子Ｑ１～Ｑ４として備えている。より詳細には、図示左側の第１レグと図示右側の第２レグとが電源１１とグラウンドとの間に並列に接続され、第１レグの上アーム（電源側アーム）に第１のスイッチング素子Ｑ１が設けられ、第１レグの下アーム（グラウンド側アーム）に第２のスイッチング素子Ｑ２が設けられ、第２レグの上アームに第３のスイッチング素子Ｑ３が設けられ、第２レグの下アームに第４のスイッチング素子Ｑ４が設けられている。 FIG. 2 shows a simplified circuit configuration example of a power transmission device 10 having a switching circuit 12 composed of a full-bridge inverter, and the switching circuit 12 includes four FETs as switching elements Q1 to Q4. More specifically, the first leg on the left side of the drawing and the second leg on the right side of the drawing are connected in parallel between the power supply 11 and the ground, and the first switching element is attached to the upper arm (power supply side arm) of the first leg. Q1 is provided, a second switching element Q2 is provided on the lower arm (ground side arm) of the first leg, a third switching element Q3 is provided on the upper arm of the second leg, and the lower arm of the second leg is provided. Is provided with a fourth switching element Q4.
　各スイッチング素子Ｑ１～Ｑ４は、駆動制御部１７から出力される駆動指令信号電圧によってそれぞれオン／オフ動作する。第１のスイッチングモードでは、第１および第４のスイッチング素子Ｑ１，Ｑ４がオン動作するとともに、第２および第３のスイッチング素子Ｑ２，Ｑ３がオフ動作し、これによりスイッチング素子Ｑ１を介して電源電圧が送電共振回路１３の一方の端子に供給される。第２のスイッチングモードでは、第２および第３のスイッチング素子Ｑ２，Ｑ３がオン動作するとともに、第１および第４のスイッチング素子Ｑ１，Ｑ４がオフ動作し、これによりスイッチング素子Ｑ３を介して電源電圧が送電共振回路の他方の端子に供給される。 Each switching element Q1 to Q4 operates on / off according to the drive command signal voltage output from the drive control unit 17. In the first switching mode, the first and fourth switching elements Q1 and Q4 are turned on, and the second and third switching elements Q2 and Q3 are turned off, whereby the power supply voltage is operated via the switching element Q1. Is supplied to one terminal of the transmission resonance circuit 13. In the second switching mode, the second and third switching elements Q2 and Q3 are turned on, and the first and fourth switching elements Q1 and Q4 are turned off, whereby the power supply voltage is operated via the switching element Q3. Is supplied to the other terminal of the power transmission resonant circuit.
　スイッチングモードの切り替え時には、電源がグラウンドに短絡することを回避するため、すべてのスイッチング素子Ｑ１～Ｑ４をオフ動作させるデッドタイムが設けられる。なお、各スイッチングモード中、典型的には各スイッチング素子のオン動作またはオフ動作を継続させることにより矩形波からなる高周波駆動電圧を共振回路１３に出力するが、オン動作させるスイッチング素子をＰＷＭ制御によりパルス駆動することにより正弦波に近い波形の高周波駆動電圧を共振回路１３に出力してもよい。 When switching the switching mode, a dead time is provided to turn off all the switching elements Q1 to Q4 in order to prevent the power supply from being short-circuited to the ground. In each switching mode, a high-frequency drive voltage composed of a square wave is typically output to the resonance circuit 13 by continuing the on or off operation of each switching element, but the switching element to be turned on is controlled by PWM. By pulse-driving, a high-frequency drive voltage having a waveform close to a sine wave may be output to the resonance circuit 13.
　スイッチング素子Ｑ１～Ｑ４をＮＭＯＳＦＥＴにより構成した場合、ＮＭＯＳＦＥＴが備えるボディダイオードが、いずれもグラウンド側から電源側へのみ電流を流す方向に配置される。この結果、すべてのスイッチング素子Ｑ１～Ｑ４がいずれもオフ動作状態の場合には、スイッチング回路１２は４つのダイオードによって構成された全波整流器として機能し、共振回路１３側の高周波電力を整流して電源１１側に供給可能である。したがって、送電装置１０を受電装置として動作させることも可能である。 When the switching elements Q1 to Q4 are composed of NMOSFETs, the body diodes included in the NMOSFETs are all arranged in a direction in which a current flows only from the ground side to the power supply side. As a result, when all the switching elements Q1 to Q4 are in the off operating state, the switching circuit 12 functions as a full-wave rectifier composed of four diodes and rectifies the high-frequency power on the resonance circuit 13 side. It can be supplied to the power supply 11 side. Therefore, it is also possible to operate the power transmission device 10 as a power receiving device.
　送電共振回路１３は、送電コイルＬとコンデンサＣとを直列に接続したＬＣ共振回路として構成され、スイッチング回路１２が生成した高周波駆動電圧が供給されると、同じ周期の共振電圧および共振電流が送電コイルＬに生じて高周波の磁界が発生する。なお、送電コイルＬに生じる共振電流は高周波駆動電圧とほぼ同位相となるが、共振電圧は９０度の進み位相となる。共振回路１３の共振周波数（ｆ＝１／２π√ＬＣ）と高周波駆動電圧の周波数（すなわちスイッチング回路１２のスイッチング周波数）とが同じであるとき、共振電圧および共振電流の振幅が最大となる。 The transmission resonance circuit 13 is configured as an LC resonance circuit in which a transmission coil L and a capacitor C are connected in series, and when a high-frequency drive voltage generated by the switching circuit 12 is supplied, a resonance voltage and a resonance current having the same period are transmitted. A high-frequency magnetic field is generated in the coil L. The resonance current generated in the transmission coil L has substantially the same phase as the high frequency drive voltage, but the resonance voltage has a lead phase of 90 degrees. When the resonance frequency (f = 1 / 2π√LC) of the resonance circuit 13 and the frequency of the high frequency drive voltage (that is, the switching frequency of the switching circuit 12) are the same, the amplitudes of the resonance voltage and the resonance current are maximized.
　本実施形態の送電装置１０では、共振電流のゼロクロス点でスイッチング回路１２のスイッチングモードを切り替えることにより、正帰還をかけながら共振するようになり、高周波駆動電圧の周波数が共振回路１３の共振周波数に収束していく。この動作原理は上記特許文献４に開示されているものと同様であるので、詳細説明を省略する。 In the power transmission device 10 of the present embodiment, by switching the switching mode of the switching circuit 12 at the zero crossing point of the resonance current, resonance is performed while applying positive feedback, and the frequency of the high frequency drive voltage becomes the resonance frequency of the resonance circuit 13. It will converge. Since this operating principle is the same as that disclosed in Patent Document 4, detailed description thereof will be omitted.
　電流検出回路１４は、共振回路１３を流れる共振電流を検出する回路であって、シャント抵抗を使った方式、カレントセンサを使った方式、ホール素子を使った方式など適宜の回路であってよい。図示例では、送電コイルＬに直列に接続されたシャント抵抗Ｒの両端の電位差をアンプ１４ａで増幅し、その出力電圧を共振電流の検出値として制御部１５のアナログ信号入力端子に入力している。 The current detection circuit 14 is a circuit that detects the resonance current flowing through the resonance circuit 13, and may be an appropriate circuit such as a method using a shunt resistor, a method using a current sensor, or a method using a Hall element. In the illustrated example, the potential difference between both ends of the shunt resistor R connected in series with the power transmission coil L is amplified by the amplifier 14a, and the output voltage is input to the analog signal input terminal of the control unit 15 as the detection value of the resonance current. ..
　制御部１５は、電流検出回路１４による共振電流の検出値に基づいて共振電流の実際の電流波形が次のゼロクロス点に至る前に共振電流の実際の電流波形の次のゼロクロス点を予測する電流波形予測部１６と、予測した次のゼロクロス点で前記スイッチング回路のスイッチングモードを切り換えるよう各スイッチング素子Ｑ１～Ｑ４に駆動指令信号電圧を出力する駆動制御部１７とを備えている。図示実施例では、駆動制御部１７は、各スイッチング素子Ｑ１～Ｑ４の制御信号を生成出力するスイッチング制御部１７ａと、スイッチング制御部１７ａが出力する制御信号に基づいて各スイッチング素子Ｑ１～Ｑ４に駆動指令信号電圧を出力するゲートドライバ１７ｂとを備えている。なお、図２においては電流波形予測部１６とスイッチング制御部１７ａとを機能ブロック図として図示したが、これらは単一のマイクロプロセッサやＦＰＧＡ若しくは基板上に一体的に実装された回路によって構成してもよいし、物理的に異なる回路によって構成されていてもよい。また、スイッチング制御部１７ａおよびゲートドライバ１７ｂについても、これらの機能を一体的に有するドライバＬＳＩによって構成してもよいし、物理的に異なる回路として構成されていてもよい。また、電流波形予測部１６とスイッチング制御部１７ａとゲートドライバ１７ｂとを単一のドライバＬＳＩ等によって構成してもよい。 The control unit 15 predicts the next zero cross point of the actual current waveform of the resonance current before the actual current waveform of the resonance current reaches the next zero cross point based on the detected value of the resonance current by the current detection circuit 14. A waveform prediction unit 16 and a drive control unit 17 that outputs a drive command signal voltage to each of the switching elements Q1 to Q4 so as to switch the switching mode of the switching circuit at the next zero crossing point predicted are provided. In the illustrated embodiment, the drive control unit 17 drives the switching control units 17a that generate and output the control signals of the switching elements Q1 to Q4, and the switching control units Q1 to Q4 based on the control signals output by the switching control units 17a. It is equipped with a gate driver 17b that outputs a command signal voltage. In FIG. 2, the current waveform prediction unit 16 and the switching control unit 17a are shown as functional block diagrams, but these are configured by a single microprocessor, FPGA, or a circuit integrally mounted on a substrate. It may be composed of physically different circuits. Further, the switching control unit 17a and the gate driver 17b may also be configured by a driver LSI having these functions integrally, or may be configured as physically different circuits. Further, the current waveform prediction unit 16, the switching control unit 17a, and the gate driver 17b may be configured by a single driver LSI or the like.
　電流波形予測部１６は、電流検出回路１４による共振電流の検出値に基づいて、スイッチングに最適なタイミングとしての共振電流波形の次のゼロクロス点を事前に予測する回路ユニットであり、本発明の中核をなす部分である。電流波形予測部１６は、アナログ入力とディジタル出力を持つマイクロプロセッサにより好適に構成されるが、ＦＰＧＡや個別の電子回路によっても構成することができる。 The current waveform prediction unit 16 is a circuit unit that predicts in advance the next zero crossing point of the resonance current waveform as the optimum timing for switching based on the detection value of the resonance current by the current detection circuit 14, and is the core of the present invention. It is the part that makes up. The current waveform prediction unit 16 is preferably configured by a microprocessor having an analog input and a digital output, but can also be configured by an FPGA or an individual electronic circuit.
　また、電流波形予測部１６は、予測した電流ゼロクロス点となるタイミングで駆動制御部１７に対してスイッチングモードの切り替え指令を出力するように構成してもよいし、予測した電流ゼロクロス点に関するデータを駆動制御部１７に予め渡しておき、駆動制御部１７が電流ゼロクロス点でスイッチングモードの切り替えを自発的に行うように構成することもできる。 Further, the current waveform prediction unit 16 may be configured to output a switching mode switching command to the drive control unit 17 at the timing when the predicted current zero cross point is reached, or data on the predicted current zero cross point may be output. It can also be configured to be passed to the drive control unit 17 in advance so that the drive control unit 17 voluntarily switches the switching mode at the current zero crossing point.
　駆動制御部１７はスイッチング回路１２のスイッチング素子Ｑ１～Ｑ４を駆動するために必要な信号を生成する。すなわち、スイッチング回路１２のスイッチング素子Ｑ１～Ｑ４のゲートを駆動する電圧を生成したり、スイッチング回路１２で貫通電流が流れないようにデッドタイムを生成したりする。なお、駆動制御部１７は、スイッチング回路１２のデッドタイム処理その他の補正処理等を、スイッチング制御部１７ａで行うように構成してもよいし、ゲートドライバ１７ｂで行うように構成してもよい。 The drive control unit 17 generates signals necessary for driving the switching elements Q1 to Q4 of the switching circuit 12. That is, a voltage for driving the gates of the switching elements Q1 to Q4 of the switching circuit 12 is generated, and a dead time is generated so that a through current does not flow in the switching circuit 12. The drive control unit 17 may be configured such that the dead time processing and other correction processing of the switching circuit 12 are performed by the switching control unit 17a or the gate driver 17b.
　次に、実施形態に係る送電装置１０の動作を図３の全体動作フローチャートを用いて説明する。 Next, the operation of the power transmission device 10 according to the embodiment will be described with reference to the overall operation flowchart of FIG.
　送電装置１０は、送電共振回路１３の共振電流を検出して、検出した共振電流波形を用いて次のスイッチングを行う自励発振動作を行うが、動作開始時にはスイッチングを行っていないため、自励発振動作が開始しない。そのため、開始後はあらかじめ決められた初期周波数でスイッチング回路１２を駆動して送電を開始する（Ｓ１０１）。初期周波数としては、送電コイルＬの定格インダクタンスとコンデンサＣの定格容量とから求められる共振周波数ｆを用いることができる。また、前回の送電動作時の周波数を記憶しておき、その周波数を初期周波数として送電開始することもできる。なお、一例において、送電共振回路１３の共振周波数は７０～１００ｋＨｚであってよい。 The power transmission device 10 detects the resonance current of the power transmission resonance circuit 13 and performs a self-excited oscillation operation in which the next switching is performed using the detected resonance current waveform. However, since switching is not performed at the start of the operation, self-excitation is performed. Oscillation operation does not start. Therefore, after the start, the switching circuit 12 is driven at a predetermined initial frequency to start power transmission (S101). As the initial frequency, the resonance frequency f obtained from the rated inductance of the power transmission coil L and the rated capacity of the capacitor C can be used. It is also possible to store the frequency at the time of the previous power transmission operation and start power transmission using that frequency as the initial frequency. In one example, the resonance frequency of the power transmission resonance circuit 13 may be 70 to 100 kHz.
　次に、送電が安定するまで待機する（Ｓ１０２）。所定の時間を経過するまで待機してもよいし、送電共振回路の共振電圧及び／又は共振電流が安定するまで待機してもよい。 Next, wait until the power transmission stabilizes (S102). It may wait until a predetermined time elapses, or it may wait until the resonance voltage and / or the resonance current of the transmission resonance circuit becomes stable.
　次に、一定の初期周波数でスイッチング回路１２を駆動することによる初期送電動作から、本発明による電流波形予測部１６を用いた自励発振動作に切り替える（Ｓ１０３）。 Next, the initial power transmission operation by driving the switching circuit 12 at a constant initial frequency is switched to the self-excited oscillation operation using the current waveform prediction unit 16 according to the present invention (S103).
　その後、所定の送電終了条件（例えば送電終了操作が行われたり、異常を検出した場合など）を満たすまで、送電動作を継続する（Ｓ１０４）。 After that, the power transmission operation is continued until a predetermined power transmission end condition (for example, when a power transmission end operation is performed or an abnormality is detected) is satisfied (S104).
　ここで、本発明による送電動作と、電流波形予測部を備えていない従来の送電装置による送電動作とを、図４及び図５を用いて対比説明する。なお、スイッチング波形は第１及び第４のスイッチング素子Ｑ１，Ｑ４のゲート信号（駆動指令信号電圧）を図示しており、第２及び第３のスイッチング素子Ｑ２，Ｑ３のゲート信号はＱ１及びＱ４のゲート信号の逆相となるため、図示省略している。 Here, the power transmission operation according to the present invention and the power transmission operation by the conventional power transmission device not provided with the current waveform prediction unit will be compared and described with reference to FIGS. 4 and 5. The switching waveform shows the gate signals (drive command signal voltage) of the first and fourth switching elements Q1 and Q4, and the gate signals of the second and third switching elements Q2 and Q3 are those of Q1 and Q4. It is not shown because it has the opposite phase of the gate signal.
　図４に点線で示すように、電流ゼロクロス点で遅延なくスイッチングが行われた場合には、遅延の無いスイッチング波形となり、共振電流波形は目標電流波形となる。目標電流波形は、様々な外因や受電共振回路２１との磁界共振結合による送電共振回路１３の周波数特性の変動時における共振周波数と一致する周波数でスイッチングが行われている理想的な共振状態にあるときの共振電流波形である。共振電流が正から負へ変化するゼロクロス点でＱ１とＱ４のゲート信号をオンからオフへ、Ｑ２とＱ３のゲート信号をオフからオンへスイッチングを行うことにより、次の半周期の動作となる。次の半周期も同様に動作させ、共振電流が負から正へ変化するゼロクロス点でＱ２とＱ３のゲート信号をオンからオフへ、Ｑ１とＱ４のゲートをオフからオンへスイッチングを行う。この動作を繰り返すことで、共振器の共振周波数と一致する理想的な周波数で共振状態が継続する。 As shown by the dotted line in FIG. 4, when switching is performed without delay at the current zero crossing point, the switching waveform has no delay, and the resonance current waveform becomes the target current waveform. The target current waveform is in an ideal resonance state in which switching is performed at a frequency that matches the resonance frequency when the frequency characteristics of the transmission resonance circuit 13 fluctuate due to various external factors and magnetic resonance coupling with the power reception resonance circuit 21. It is a resonance current waveform at the time. By switching the gate signals of Q1 and Q4 from on to off and the gate signals of Q2 and Q3 from off to on at the zero crossing point where the resonance current changes from positive to negative, the operation is performed in the next half cycle. The next half cycle is also operated in the same manner, and the gate signals of Q2 and Q3 are switched from on to off and the gates of Q1 and Q4 are switched from off to on at the zero crossing point where the resonance current changes from negative to positive. By repeating this operation, the resonance state continues at an ideal frequency that matches the resonance frequency of the resonator.
　しかし、従来の送電装置は、共振電流波形のゼロクロス点を検出した直後に次のスイッチングを行うため、図４に実線で示すように実際の共振電流波形の電流ゼロクロス点からスイッチングが行われるまでに遅れΔｔが生じる。この遅れΔｔに起因して電流ゼロクロス点を超えてもスイッチングが行われない結果、実際の共振電流波形が歪んでその周期が遅れΔｔに相当する分だけ長くなる。 However, since the conventional power transmission device performs the next switching immediately after detecting the zero crossing point of the resonance current waveform, as shown by the solid line in FIG. 4, before switching is performed from the current zero crossing point of the actual resonance current waveform. A delay Δt occurs. Due to this delay Δt, switching is not performed even if the current zero cross point is exceeded, and as a result, the actual resonance current waveform is distorted and its period becomes longer by the amount corresponding to the delay Δt.
　電流検出回路１４は、電流の変化を検出するのにある程度の時間を要し、特にカレントトランスやホール素子を電流センサとして用いた場合に遅れΔｔが大きくなる。シャント抵抗を使った方式では遅れΔｔが比較的小さいが、オペアンプ１４ａなどの素子を使うと必ず遅れが発生する。 The current detection circuit 14 takes a certain amount of time to detect a change in current, and the delay Δt becomes large especially when a current transformer or a Hall element is used as a current sensor. In the method using a shunt resistor, the delay Δt is relatively small, but when an element such as an operational amplifier 14a is used, a delay always occurs.
　さらに、その次の電流ゼロクロス点においても同様に遅れΔｔが生じるため、半周期毎に遅れΔｔが蓄積していくこととなり、実際の共振電流波形の周期が、理想的な目標電流波形の周期よりも２Δｔ長くなってしまう。このように電流検出に遅れΔｔがあるときには、理想的な共振周波数よりも低い周波数でスイッチングが継続することとなる。 Further, since the delay Δt also occurs at the next current zero crossing point, the delay Δt accumulates every half cycle, and the actual resonance current waveform cycle is larger than the ideal target current waveform cycle. Is also 2Δt longer. When there is a delay Δt in the current detection in this way, switching continues at a frequency lower than the ideal resonance frequency.
　なお、電流検出だけが遅れるものとして説明したが、実際には駆動制御部１７の動作にも遅れもあるため、さらに遅れ大きくなる。 Although it was explained that only the current detection is delayed, the delay is further increased because the operation of the drive control unit 17 is also delayed.
　図６は、送電共振回路および受電共振回路ともに共振周波数を８５ｋＨｚに調整した非接触無線電力伝送装置を用いて、スイッチング周波数を変更しながら送電電力と送電効率とを測定したグラフであり、周波数が８５ｋＨｚのときに送電電力・送電効率ともに最大となっている。従来の電流ゼロクロス点検出式の送電装置の駆動制御方法をそのまま用いた場合、最大の送電電力・送電効率となる理想的な共振周波数よりも低い８４．３ｋＨｚで動作し、送電電力・送電効率ともに最大値よりも低下した。利用する電流センサや駆動制御回路により遅れの時間は変わるので、自励発振の周波数は変わってくるが、必ず遅れがあるため、必ず理想的な共振周波数よりも低い周波数で動作し、送電電力・送電効率ともに低下する。非接触電力伝送に使う周波数が比較的低いときには駆動制御部１７の遅れは相対的に問題とならないが、周波数が高くなると駆動制御回路１７の動作の遅れも問題となってくる。 FIG. 6 is a graph in which both the power transmission resonance circuit and the power reception resonance circuit are graphs in which the power transmission power and the power transmission efficiency are measured while changing the switching frequency using a non-contact wireless power transmission device in which the resonance frequency is adjusted to 85 kHz. At 85 kHz, both transmission power and transmission efficiency are maximum. When the drive control method of the conventional zero-cross point detection type power transmission device is used as it is, it operates at 84.3 kHz, which is lower than the ideal resonance frequency, which is the maximum power transmission power and power transmission efficiency. It fell below the maximum value. Since the delay time changes depending on the current sensor and drive control circuit used, the frequency of self-excited oscillation changes, but since there is always a delay, it always operates at a frequency lower than the ideal resonance frequency, and the transmitted power Both transmission efficiency will decrease. When the frequency used for non-contact power transmission is relatively low, the delay of the drive control unit 17 is relatively not a problem, but when the frequency is high, the delay of the operation of the drive control circuit 17 is also a problem.
　一方、本実施形態による電流波形予測部１６を備える制御部１５を利用した場合の共振電流波形を、従来の電流ゼロクロス点検出方式による共振電流波形と対比して図５に図示する。本実施形態による制御部１５は、電流検出回路１４による共振電流の検出値に基づいて共振電流波形の予測を行い、電流センサ１４ａや駆動制御回路１７の遅れを考慮して、実際の共振電流波形のゼロクロス点で正確にスイッチングが行われるよう、スイッチング回路１２に駆動指令信号電圧を出力する動作を行う。 On the other hand, the resonance current waveform when the control unit 15 including the current waveform prediction unit 16 according to the present embodiment is used is shown in FIG. 5 in comparison with the resonance current waveform by the conventional current zero cross point detection method. The control unit 15 according to the present embodiment predicts the resonance current waveform based on the detection value of the resonance current by the current detection circuit 14, and considers the delay of the current sensor 14a and the drive control circuit 17, and actually considers the resonance current waveform. The drive command signal voltage is output to the switching circuit 12 so that the switching can be performed accurately at the zero crossing point.
　図７は、電流波形予測部１６の動作フローチャートの一例を示しており、この電流波形予測部１６の動作は、図３に示す自励発振動作（Ｓ１０３）の開始後に行われる。 FIG. 7 shows an example of the operation flowchart of the current waveform prediction unit 16, and the operation of the current waveform prediction unit 16 is performed after the start of the self-excited oscillation operation (S103) shown in FIG.
　電流波形予測部１６は、所定の時点、例えば駆動制御部１７に対する直近の最後のスイッチング指令出力時点、からの時間を計測するためのカウントを開始する（Ｓ２０１）。なお、初回動作時は、直近の最後のスイッチング指令出力が存在しないため、自励発振動作の開始時点からの時間を計測するなど、適宜の時点からの時間を計測することができる。 The current waveform prediction unit 16 starts counting for measuring the time from a predetermined time point, for example, the latest last switching command output time point with respect to the drive control unit 17 (S201). Since the last switching command output does not exist at the time of the first operation, it is possible to measure the time from an appropriate time point such as measuring the time from the start time of the self-excited oscillation operation.
　次に、カウントを継続しつつ、共振電流波形のピークを検出するまで待機する（Ｓ２０２）。共振電流波形のピーク検出方法は適宜のものであってよいが、例えば、検出電流の絶対値が増加している状態から減少する状態となったことを検出した時点を共振電流波形のピークとして判定できる。このとき、電流波形にノイズが混入していることが想定されるときには検出電流の絶対値が複数回連続して減少したときにピークを検出したと判定することもできる。 Next, while continuing the count, it waits until the peak of the resonance current waveform is detected (S202). The peak detection method of the resonance current waveform may be appropriate, but for example, the time when it is detected that the absolute value of the detected current has changed from an increasing state to a decreasing state is determined as the peak of the resonance current waveform. can. At this time, when it is assumed that noise is mixed in the current waveform, it can be determined that the peak is detected when the absolute value of the detected current decreases continuously a plurality of times.
　また、共振電流波形の微分値が所定条件を満たした時点を共振電流波形のピークとして判定することもできる。共振電流波形は概略正弦波であるため、微分した波形は概略余弦波となる。したがって、所定条件の一例としては、微分値が０に近い値になった時点である。微分値がほぼ０になるということは、電流波形のピークの時刻と同時であるため、微分値がほぼ０になるまで待機することによって電流波形のピークまで待機することとなる。 Further, the time when the differential value of the resonance current waveform satisfies a predetermined condition can be determined as the peak of the resonance current waveform. Since the resonance current waveform is a roughly sine wave, the differentiated waveform is a roughly cosine wave. Therefore, as an example of the predetermined condition, it is the time when the differential value becomes close to 0. Since the fact that the differential value becomes almost 0 coincides with the peak time of the current waveform, waiting until the differential value becomes almost 0 means waiting until the peak of the current waveform.
　共振電流波形のピークを検出すると、時間計測のためのカウンタを読み出し、図５にもステップ（１）で示すように、直近の最後のスイッチング指令出力時点から共振電流波形のピークまでの時間を算出する（Ｓ２０３）。 When the peak of the resonance current waveform is detected, the counter for time measurement is read out, and as shown in step (1) in FIG. 5, the time from the latest last switching command output time to the peak of the resonance current waveform is calculated. (S203).
　次に、前回のスイッチング指令出力時点から共振電流波形のピークまでの時間を用いて、図５にもステップ（２）で示すように、次の共振電流波形のゼロクロス点を予測する（Ｓ２０４）。例えば、前回のスイッチング指令出力時点から共振電流波形のピークまでの時間をｔ、電流センサの遅れと駆動制御回路の遅れがΔｔのとき、前回のスイッチング指令出力時点から遅れの無いスイッチング波形による理想的な共振電流波形の次のゼロクロス点までの期間は、（ｔ－Δｔ）×２により算出できる。さらに半導体のスイッチング損失などを考慮した期間とすることもできる。 Next, using the time from the time of the previous switching command output to the peak of the resonance current waveform, the zero cross point of the next resonance current waveform is predicted as shown in step (2) in FIG. 5 (S204). For example, when the time from the previous switching command output time to the peak of the resonance current waveform is t, and the current sensor delay and the drive control circuit delay are Δt, the ideal switching waveform has no delay from the previous switching command output time. The period until the next zero crossing point of the resonance current waveform can be calculated by (t−Δt) × 2. Further, the period may be set in consideration of the switching loss of the semiconductor.
　共振電流波形の次のゼロクロス点の予測が完了すると、予測された電流ゼロクロス時刻まで待機し（Ｓ２０５）、図５にもステップ（３）で示すように予測された電流ゼロクロス時刻で駆動制御部１７にスイッチング指令信号を出力する（Ｓ２０６）。 When the prediction of the next zero cross point of the resonance current waveform is completed, the drive control unit 17 waits until the predicted current zero cross time (S205), and at the predicted current zero cross time as shown in step (3) in FIG. A switching command signal is output to (S206).
　電流波形予測部１６は、以上のステップＳ２０１～Ｓ２０６の動作を送電動作中、繰り返し継続する。これにより、遅れの無いスイッチング波形と同等の波形でスイッチングを継続し、共振回路の共振周波数によるワイヤレス電力伝送の制御を行うことにより、大電力かつ効率の高い無線電力伝送を行うことができる。 The current waveform prediction unit 16 repeatedly continues the above operations of steps S201 to S206 during the power transmission operation. As a result, high-power and highly efficient wireless power transmission can be performed by continuing switching with a waveform equivalent to the switching waveform without delay and controlling wireless power transmission by the resonance frequency of the resonant circuit.
　電流波形予測部１６は、共振電流波形のピーク検出以外の方法で次の電流ゼロクロス点の予測を行うよう構成することもできる。例えば、電流検出回路１４による共振電流の検出値の絶対値が所定の閾値Ａ以上から閾値Ａ未満に遷移したことを検出した時点で、いますぐにスイッチング指令を駆動制御部１７に出力すれば次の電流ゼロクロス点でスイッチングがちょうど行われるものと予測するよう構成できる。 The current waveform prediction unit 16 can also be configured to predict the next zero cross point by a method other than peak detection of the resonance current waveform. For example, if it is detected that the absolute value of the resonance current detected by the current detection circuit 14 has transitioned from a predetermined threshold value A or more to less than the threshold value A, a switching command is immediately output to the drive control unit 17, and the following It can be configured to predict that switching will occur exactly at the zero-current crossing point of.
　閾値Ａは、予め設定した固定値であっても良いが、共振電流波形の振幅が動作条件等によって変動した場合にも次の電流ゼロクロス点で正確にスイッチングが行われるようにするために、共振電流波形の直前のピークの振幅に、０より大きく１未満の所定の係数を乗算したものとすることができる。係数は、予め電流検出回路１４や駆動制御部１７の動作の遅れの程度を計測しておき、共振電流の検出値の絶対値が閾値Ａ以上から閾値Ａ未満に遷移したことを検出した場合に即時のタイミングでスイッチング指令を駆動制御部１７に出力することで、上記の遅れをちょうど相殺して次の電流ゼロクロス点で正確にスイッチングが行われるよう設定できる。 The threshold A may be a fixed value set in advance, but resonance is performed so that switching can be performed accurately at the next zero crossing point even when the amplitude of the resonance current waveform fluctuates due to operating conditions or the like. It can be assumed that the amplitude of the peak immediately before the current waveform is multiplied by a predetermined coefficient greater than 0 and less than 1. The coefficient is obtained when the degree of delay in the operation of the current detection circuit 14 and the drive control unit 17 is measured in advance and it is detected that the absolute value of the detected value of the resonance current has transitioned from the threshold value A or more to the threshold value A or less. By outputting the switching command to the drive control unit 17 at the immediate timing, it is possible to set so that the switching is accurately performed at the next current zero crossing point by just canceling the above delay.
　本発明は上記実施形態に限定されるものではなく、適宜設計変更できる。例えば、上記実施形態では磁界共鳴結合式のワイヤレス送電装置に本発明を適用した例を示したが、送電側に共振回路を有する電磁誘導式のワイヤレス送電装置に本発明を適用することもできる。また、本発明は、海水中での位置変動による最適送電周波数の変化が起こりがちな、水中航走体、例えば潜水艦や水中ドローンなど、のワイヤレス送電装置に特に好適に利用することができる。 The present invention is not limited to the above embodiment, and the design can be changed as appropriate. For example, in the above embodiment, the present invention is applied to a magnetic field resonance coupling type wireless power transmission device, but the present invention can also be applied to an electromagnetic induction type wireless power transmission device having a resonance circuit on the power transmission side. Further, the present invention can be particularly suitably used for a wireless power transmission device such as an underwater vehicle, for example, a submarine or an underwater drone, in which the optimum power transmission frequency tends to change due to a position change in seawater.
　１０　ワイヤレス送電装置
　１１　電源
　１２　スイッチング回路
　１３　送電共振回路
　１４　電流検出回路
　１５　制御部
　１６　電流波形予測部
　１７　駆動制御部 10 Wireless power transmission device 11 Power supply 12 Switching circuit 13 Power transmission resonance circuit 14 Current detection circuit 15 Control unit 16 Current waveform prediction unit 17 Drive control unit

Claims (6)

  1. 　送電コイルとコンデンサとを有する共振回路と、
    　電源から供給される電力を用いて前記共振回路に高周波電圧を印加させるべく第１のスイッチングモードと第２のスイッチングモードとに交互に切り換え可能に構成されたスイッチング回路と、
    　前記スイッチング回路の前記スイッチングモードの切換制御を行う制御部と、
    　前記共振回路に生じる共振電流を検出する電流検出部と、
    　を備え、非接触により受電装置へ電力を伝送するワイヤレス送電装置において、
    　前記制御部は、前記電流検出部による前記共振電流の検出値に基づいて前記共振電流の実際の電流波形が次のゼロクロス点に至る前に前記共振電流の実際の電流波形の次のゼロクロス点を予測する電流波形予測部を備え、予測した次のゼロクロス点で前記スイッチング回路のスイッチングモードを切り換えるよう構成されている、ワイヤレス送電装置。     A resonant circuit with a power transmission coil and a capacitor,
    A switching circuit configured to be able to alternately switch between a first switching mode and a second switching mode in order to apply a high frequency voltage to the resonance circuit using electric power supplied from a power source.
    A control unit that controls switching of the switching mode of the switching circuit,
    A current detection unit that detects the resonance current generated in the resonance circuit, and
    In a wireless power transmission device that transmits power to a power receiving device in a non-contact manner
    Based on the value detected by the current detection unit of the resonance current, the control unit sets the next zero cross point of the actual current waveform of the resonance current before the actual current waveform of the resonance current reaches the next zero cross point. A wireless power transmission device including a current waveform prediction unit for prediction, which is configured to switch the switching mode of the switching circuit at the next zero crossing point predicted.
  2. 　請求項１に記載のワイヤレス送電装置において、
    　前記電流波形予測部は、所定の時点から前記共振電流の検出値に基づく検出電流波形のピークが検出されるまでの計測時間に基づいて次のゼロクロス点を予測するよう構成されている、ワイヤレス送電装置。     In the wireless power transmission device according to claim 1,
    The current waveform predictor is configured to predict the next zero crossing point based on the measurement time from a predetermined time point to the detection of the peak of the detected current waveform based on the detected value of the resonant current. Device.
  3. 　請求項２に記載のワイヤレス送電装置において、
    　前記時点は、前記制御部が前記スイッチングモードの直前の切換えを行った時点である、ワイヤレス送電装置。     In the wireless power transmission device according to claim 2.
    The time point is a time point when the control unit performs switching immediately before the switching mode, that is, a wireless power transmission device.
  4. 　請求項２又は３に記載のワイヤレス送電装置において、
    　前記電流波形予測部は、前記共振電流の前記検出電流波形のピークを、当該検出電流波形の微分値に基づいて検出するよう構成されている、ワイヤレス送電装置。     In the wireless power transmission device according to claim 2 or 3.
    The current waveform prediction unit is a wireless power transmission device configured to detect a peak of the detected current waveform of the resonance current based on a differential value of the detected current waveform.
  5. 　請求項１に記載のワイヤレス送電装置において、
    　前記電流波形予測部は、前記電流検出部による前記共振電流の検出値の絶対値が基準値以上から前記基準値未満へ遷移した時点に基づいて前記共振電流の実際の電流波形の次のゼロクロス点を予測するよう構成されている、ワイヤレス送電装置。     In the wireless power transmission device according to claim 1,
    The current waveform prediction unit is the next zero crossing point of the actual current waveform of the resonance current based on the time when the absolute value of the detection value of the resonance current by the current detection unit transitions from the reference value or more to less than the reference value. A wireless transmitter that is configured to predict.
  6. 　請求項５に記載のワイヤレス送電装置において、
    　前記基準値は、前記共振電流の検出値に基づく検出電流波形の直前のピーク値に基づいて演算により求められる値である、ワイヤレス送電装置。     In the wireless power transmission device according to claim 5.
    The reference value is a value obtained by calculation based on the peak value immediately before the detection current waveform based on the detection value of the resonance current, and is a wireless power transmission device.
PCT/JP2020/014870 2020-03-31 2020-03-31 Wireless power transmission device WO2021199304A1 (en)

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Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2016032345A (en) * 2014-07-29 2016-03-07 日立マクセル株式会社 Contactless power transmission device
JP2019503160A (en) * 2015-12-17 2019-01-31 コーニンクレッカ フィリップス エヌ ヴェＫｏｎｉｎｋｌｉｊｋｅ Ｐｈｉｌｉｐｓ Ｎ．Ｖ． Control circuit and method for controlling resonant converter, and power inverter including resonant converter and control circuit
JP2019512162A (en) * 2016-02-08 2019-05-09 ワイトリシティ コーポレーションＷｉｔｒｉｃｉｔｙ Ｃｏｒｐｏｒａｔｉｏｎ PWM capacitor control

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2016032345A (en) * 2014-07-29 2016-03-07 日立マクセル株式会社 Contactless power transmission device
JP2019503160A (en) * 2015-12-17 2019-01-31 コーニンクレッカ フィリップス エヌ ヴェＫｏｎｉｎｋｌｉｊｋｅ Ｐｈｉｌｉｐｓ Ｎ．Ｖ． Control circuit and method for controlling resonant converter, and power inverter including resonant converter and control circuit
JP2019512162A (en) * 2016-02-08 2019-05-09 ワイトリシティ コーポレーションＷｉｔｒｉｃｉｔｙ Ｃｏｒｐｏｒａｔｉｏｎ PWM capacitor control

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