WO2021125911A1

WO2021125911A1 - Method and apparatus for reducing spike voltage of switching power supply

Info

Publication number: WO2021125911A1
Application number: PCT/KR2020/018764
Authority: WO
Inventors: 박찬웅
Original assignee: 박찬웅
Priority date: 2019-12-20
Filing date: 2020-12-21
Publication date: 2021-06-24
Also published as: KR20210080265A; WO2021125911A4

Abstract

A switching power supply including an input filter capacitor and a switching element according to the present invention comprises: a transformer including a transformer core, a first input winding which is wound on the transformer core and in which the flow of current is controlled by a switching operation of the switching element, a second input winding which is wound on the transformer core and in which the flow of current is controlled by a switching operation of the switching element, and an output winding which is wound on the transformer core and coupled to the first input winding and the second input winding to emit energy; an inductor; and a first capacitor, wherein the first input winding of the transformer is closely coupled to the second input winding, the inductor and the first input winding are connected in series, the first capacitor is connected between one end and the other end of the inductor and the first input winding connected in series, and the inductor and the first input winding, and the second input winding and the switching element are connected in series between two terminals of the input filter capacitor.

Description

Method and device for reducing the spike voltage of a switching power supply

The present invention relates to a method and apparatus for lowering a spike voltage generated by a leakage inductance of an input winding of a transformer of a switching power supply, and more particularly, to generate a spike voltage with very low energy stored in the leakage inductance of an input winding of a transformer By removing the clamp circuit that limits the spike voltage, the noise generated in the clamp circuit is completely removed, and a very low spike voltage is induced in the auxiliary winding, thereby reducing the high-frequency noise generated by the diode rectifying the voltage of the auxiliary winding. It relates to a method and apparatus for significantly reducing the influence of noise from a power supply to the outside by significantly lowering it.

In the prior art switching power supply, after the switching element is turned off, a high spike voltage is generated by energy accumulated in the leakage inductance of the input winding of the transformer, and a clamp circuit composed of a diode, a capacitor, and a resistor is used. In the process of limiting the spike voltage, large high-frequency noise is generated by the clamp diode, and large high-frequency noise is generated by the diode that supplies the power voltage to the control unit of the power supply by rectifying the high spike voltage induced in the auxiliary winding of the transformer. , the power supply greatly affects the noise to the outside.

A brief description of the prior art is as follows.

1 shows a power supply device of the prior art, and FIGS. 2 and 3 show voltage and current waveforms of each part of FIG. 1 .

In the power supply device of FIG. 1 , the AC input voltage is rectified and smoothed by the input filter capacitor 11 . One terminal of the input winding 231 of the transformer 23 is connected to the “+” terminal of the input filter capacitor 11 , and the other terminal of the input winding 231 is connected to one terminal of the switching element 12 . . The other terminal of the switching element 12 is connected to the "-" terminal of the input filter capacitor 11 . The control unit 13 controls the switching operation of the switching element 12 to adjust the amount of energy transferred from the input winding 231 to the output winding 233 of the transformer 23 . The energy transferred to the output winding 233 is rectified by the diode 16 and smoothed by the capacitor 17, and is supplied to the output. The voltage induced in the auxiliary winding 232 is rectified by the diode 14 and smoothed by the capacitor 15 to supply a power supply voltage to the control unit 13 . When the switching element 12 is turned off, the energy accumulated in the leakage inductance of the input winding 231 that is not magnetically coupled to the output winding 233 is determined by the distribution capacity of the input winding 231 and the switching element ( 12) is discharged while charging the capacity between the terminals, and a large spike voltage is generated in this process. The spike voltage is suppressed as it is rectified by diode 18 and smoothed by capacitor 19 and consumed by resistor 20 . The combination of resistor 50, diode 18, capacitor 19 and resistor 20 is called a clamp circuit. here. Reference numeral 21 in FIG. 1 denotes an input line, and 22 denotes an output line.

In the power supply device of the prior art of FIG. 1, the process of generating a spike voltage with high energy accumulated in the leakage inductance of the input winding 231 and the process of generating noise when the spike voltage is suppressed by the clamp circuit It will be described in detail with reference to the waveform diagrams of FIGS. 2 and 3 .

FIG. 2 is a waveform diagram of each part of the power supply device of FIG. 1 when a clamp circuit composed of a resistor 50, a diode 18, a capacitor 19, and a resistor 20 is not included, and FIG. 3 is FIG. It is a waveform diagram of each part when the clamp circuit is included in the power supply of

In the power supply device of the prior art of FIG. 1 and the waveform diagram of FIG. 2 , when the switching element 12 is in a conductive state, a current flows through the input winding 231 and magnetic energy is accumulated in the transformer 23 . At time t1 just before the switching element 12 is turned off, the current flowing through the switching element 12 becomes the maximum value. Immediately after the switching element 12 is turned off, the magnetic energy accumulated in the input winding 231 is the capacitance of the snubber circuit connected to the power supply, the distributed capacitance of each winding, and the capacitance between the terminals of the switching element 12 . to charge At time t3, when the voltage of the output winding 233 becomes higher than the output voltage, the diode 16 is turned on and the magnetic energy accumulated in the transformer 23 is withdrawn to the output winding 233 to supply power to the load. . Usually, the time from t1 to t3 is within several tens of nS. The energy accumulated in the leakage inductance of the input winding 231 is discharged while charging the distributed capacitance of the input winding 231 and the capacitance between the terminals of the switching element 12 after time t3. When there is no clamp circuit composed of a diode 18, a capacitor 19, and a resistor 20 in the power device of FIG. 1, the voltage between both terminals of the switching element 12 is the waveform diagram B. Vds of FIG. 2 and Similarly, it rises very high in the period from time t3 to time t4, making it impossible to secure the reliability of the switching element 12. In general, in order to secure reliability, when the breakdown voltage of the switching element 12 is 700V, the maximum voltage between both terminals of the switching element 12 must be limited to 600V or less by the clamp circuit.

In the prior art power supply device of FIG. 1 and the waveform diagram of FIG. 3, at time t3, the voltage Vds of the switching element 12 is applied to the voltage of the "+" terminal of the input filter capacitor 11 and the capacitor of the clamp circuit. When it becomes higher than the voltage plus the charging voltage of (19), a forward voltage is applied to the diode (18). After time t3, diode 18 still does not conduct during Forward Recovery Time, despite forward voltage being applied. When the current starts to flow in the diode 18 at time t4, the forward voltage drop of the diode 18 gradually decreases, and the voltage charged in the distributed capacitance of the input winding 231 is transferred to the capacitor 19 through the diode 18. ) decreases while charging. At time t5, the diode 18 conducts stably. The energy transferred to capacitor 19 is dissipated as heat in resistor 20 . The faster the rising rate of the forward voltage applied to the diode 18 in the period from time t3 to t4, or the faster the rising rate of the current flowing through the diode 18 in the period of time t4 to t5, the diode 18 is , as shown in the period t3 to t5 of the waveform of C. ID18 in Fig. 3, generates a larger high-frequency noise. The resistor 50 limits the peak value of the current flowing through the diode 18 after the diode 18 conducts, thereby lowering the noise generated when the diode 18 conducts. On the other hand, the auxiliary winding 232 is wound in close contact with the input winding 231 , and also induces a spike voltage component generated by the leakage inductance of the input winding 231 . The high spike voltage induced in the auxiliary winding 232 causes the diode 14 to generate a large noise voltage while turning the diode 14 on. The resistor 51 limits the peak value of the current flowing through the diode 14 to lower the high-frequency noise generated by the diode 14 . Since the high-frequency noise generated by the diode 18 and the diode 14 affects the outside of the power supply device, the cost of the noise filter increases in order to lower the EMI (Electro-Magnetic Interference) of the power supply device to a specified value or less.

On the other hand, in the power supply device of the prior art of FIG. 1 and the waveform diagram of FIG. 3 , the discharge of energy accumulated in the input winding 231 of the transformer 23 is completed at time t6. After time t6 , the voltage charged in the distributed capacitance of the input winding 231 starts to be discharged through the input winding 231 , and a reverse voltage is applied to the diode 18 . In the prior art of FIG. 1 , a forward voltage with a fast rising rate is applied to the diode 18 during the period from time t3 to time t5 so that a large charge is accumulated at the junction of the diode 18, and after time t6, the diode 18 ), the diode 18 recovers to “0” after allowing a large reverse current to flow while discharging the charge accumulated at the junction. In general, a diode with a faster switching speed tends to generate a large noise current, so engineers try to use a slow speed diode to reduce noise. However, in the prior art of FIG. 1, when a Slow Speed Diode is used for the diode 18, the period during which the reverse current flows is too long, so that the capacitor 19 of the clamp circuit is discharged through the diode 18, resulting in unnecessary loss. This may occur or a problem in which the diode 18 is overheated may occur.

In the prior art power supply device of FIG. 1 , the capacitor 52 and the resistor 53 indicated by dotted lines reduce the energy that turns on the diode 18 when the diode 18 is turned on, and the diode 18 ) to reduce the magnitude of high-frequency noise. However, the capacitor 52 has a disadvantage that, when the switching device 12 is turned on, a charged voltage is discharged through the switching device 12 and a large power loss occurs.

4 shows another power supply device of the prior art.

In the prior art power supply device of FIG. 4 , the AC input voltage is rectified and smoothed by the input filter capacitor 11 . One terminal of the second input winding 242 of the transformer 24 is connected to the “+” terminal of the input filter capacitor 11 , and the other terminal of the second input winding 242 is one terminal of the switching element 12 . connected to the terminal. The other terminal of the switching element 12 is connected to one terminal of the first input winding 241 . The other terminal of the first input winding 241 is connected to the "-" terminal of the input filter capacitor 11 . The control unit 13 controls the switching operation of the switching element 12 , and the amount of energy transferred from the first input winding 241 and the second input winding 242 to the output winding 243 of the transformer 24 . adjust the The energy transferred to the output winding 243 is rectified by the diode 16a and smoothed by the capacitor 17, and is supplied to the output. The voltage induced in the first input winding 241 is rectified by the diode 14b and smoothed by the capacitor 15b, and is supplied to the control unit 13 as a power supply voltage. When the switching element 12 is turned off, the energy accumulated in the leakage inductance of the first input winding 241 and the second input winding 242 that are not magnetically coupled to the output winding 243 is, It is discharged while charging and discharging the distributed capacitance of the switching element 12 and the capacitance between the terminals of the switching element 12, and a large spike voltage is generated in this process. The spike voltage is suppressed as it is rectified by diode 18 and smoothed by capacitor 19 and consumed by resistor 20 . here. Reference numeral 21 in FIG. 1 denotes an input line, and 22 denotes an output line.

In the prior art power supply device of FIG. 4, when the diode 18 of the clamp circuit clamps the spike voltage of the second input winding 242, the waveform shown in the period t3 to t5 of C. ID18 of FIG. As can be seen, the diode 18 generates large high-frequency noise. Further, the spike voltage of the second input winding 242 is induced in the first input winding 241 and applied to the diode 14b. Therefore, the diode 14b generates a large high-frequency noise when conducting.

In the prior art power supply device of FIG. 4 , the high-frequency noise generated by the diode 18 and the diode 14b affects the outside of the power supply device, so the cost of the noise filter to lower the EMI of the power supply device to a specified value or less it rises In addition, since it is necessary to use a clamp circuit composed of a resistor 50, a diode 18, a capacitor 19, and a resistor 20, it becomes complicated and the cost increases.

The resistor 51 limits the peak value of the current flowing through the diode 14b, thereby slightly lowering the high-frequency noise generated by the diode 14b.

In FIG. 4 as well, an attempt is made to connect a resistor and a capacitor to both ends of the first input winding 241 in order to lower the noise voltage generated in the first input winding 241, but the power loss due to the capacitor is within the allowable value. It is very difficult to significantly lower the high-frequency noise generated by the diode 14b while maintaining it.

5 shows a prior art power supply for reducing noise generated by a diode of a clamp circuit.

5, the input winding 251 of the transformer 25 is divided into a first input winding 251a and a second input winding 251b. A first capacitor 27 is connected between both terminals of the first input winding 251a.

FIG. 6 shows an embodiment of a transformer 25 used in the power supply of FIG. 5 .

In FIG. 6 , the second input winding 251b of the transformer 25 is wound around the entire winding surface of the core 256 so as to be magnetically coupled to the output winding 253 . The first input winding 251a is wound so that the leakage inductance not coupled to the second input winding 251b has a large value.

In the prior art power supply device of FIG. 5, the leakage inductance of the first input winding 251a not coupled to the second input winding 251b and the first capacitor 27, the switching element 12 is turned off Immediately after the resonance occurs, when the diode 18 is turned on by the spike voltage generated by the input winding 251, the energy to turn on the diode 18 is lowered, and the high-frequency noise generated by the diode 18 is reduced. The size is much lower compared to the prior art technique of FIG. 1 .

In the power supply device of the prior art of FIG. 5, most of the magnetic energy accumulated in the leakage inductance of the second input winding 251b is generated by the first input winding 251a immediately after the switching element 12 is turned off. It cannot be withdrawn through , and the distribution capacity of the second input winding 251b is charged and a high spike voltage is generated. The spike voltage is limited by a clamp circuit consisting of a resistor (50), a diode (18), a capacitor (19) and a resistor (20). After the switching element 12 is turned off, a portion of the magnetic flux generated by the leakage inductance of the second input winding 251b and the magnetic flux of the core 256 of the transformer 25 cause the first input winding 251a The voltage induced to and the voltage of the first capacitor 27 are applied to the leakage inductance of the first input winding 251a, and the leakage inductance of the first input winding 251a and the first capacitor 27 generate resonance. causes Compared to the voltage applied to the first input winding 251a, the leakage inductance of the first input winding 251a and the resonance current delayed by 1/4 of the resonance period of the first capacitor 27 are generated in the first input winding ( 251a). Therefore, when the spike voltage reaches the voltage that conducts the diode 18 of the clamp circuit, the current flowing out through the output winding 253 is reduced by the current flowing out through the first input winding 251a, The voltage at both ends of the leakage inductance of the output winding 253 is also lowered, the charging voltage of the distributed capacity of the output winding 253 is also lowered, and the charging voltage of the distributed capacity of the second input winding 251b is also lowered. Further, when the spike voltage reaches the voltage that makes the diode 18 of the clamp circuit become conductive, compared to when no current flows from the first input winding 251a to the first capacitor 27, the second input winding The magnetic energy of the leakage inductance of (251b) is much more escaped to the distribution capacitance of the input winding (251). As a result, since a small amount of magnetic energy remaining in the leakage inductance of the second input winding 251b conducts the diode 18 of the clamp circuit, high-frequency noise generated by the diode 18 is significantly reduced. However, since the high spike voltage of the second input winding 251b is induced very high in the auxiliary winding 252, the diode 14 rectifying the voltage of the auxiliary winding 252 still generates a large high-frequency noise. The leakage inductance of the first input winding 251a that is not coupled to the second input winding 251b transfers the electrostatic energy of the first capacitor 27 as magnetic energy when the first capacitor 27 is discharged. Therefore, power loss due to discharging of the first capacitor 27 does not occur.

However, in the power supply device of the prior art of FIG. 5, the leakage inductance of the second input winding 251b is still reduced by the clamp circuit consisting of the resistor 50, the diode 18, the capacitor 19, and the resistor 20. There is a disadvantage that the generated spike voltage must be limited, and the noise generated by the diode 18 cannot be completely removed. In addition, since the spike voltage generated by the energy accumulated in the leakage inductance of the second input winding 251b is induced in the auxiliary winding 252 and applied to the diode 14, when the diode 14 is turned on, The diode 14 has a disadvantage that large high-frequency noise is generated.

Energy accumulated in the leakage inductance of the input winding of the prior art transformer of FIG. 1 causes a high spike voltage to be applied between both terminals of the switching element. In order to protect the switching element from such spike voltage, a clamp circuit must be used. The clamp circuit increases the number of parts of the power supply device, increases the cost, deteriorates the EMI of the power supply device due to high-frequency noise generated by the diode of the clamp circuit, and increases the noise filter cost. In addition, the spike voltage generated in the input winding is induced in the auxiliary winding, and when the diode supplying the power supply voltage to the control unit by rectifying the voltage of the auxiliary winding is turned on, a large high frequency noise is generated in the diode. In addition, when a Slow Speed Diode is used in the clamp circuit, it has a disadvantage in that unnecessary loss or overheating of the diode occurs.

Although the prior art power supply of Fig. 5 significantly lowers noise generated by the diode of the clamp circuit compared to the prior art power supply of Fig. 1, it is still necessary to limit the spike voltage using the clamp circuit. The use of the clamp circuit has disadvantages in that the number of parts of the power supply device is increased, the cost is increased, and the high-frequency noise generated by the diode of the clamp circuit cannot be completely removed. In addition, the spike voltage generated in the input winding is induced in the auxiliary winding, and the diode that supplies the power voltage to the control unit by rectifying the voltage of the auxiliary winding is turned on so that the diode generates high frequency noise. Accordingly, the power supply device of the prior art of FIG. 5 has a disadvantage in that noise is influenced to the outside.

The present invention is to solve all of these problems of the prior art.

an input filter capacitor for achieving the above object; A switching type power supply including a switching element,

the core of the transformer; a first input winding wound around the core of the transformer and controlling the flow of current by a switching operation of the switching element; a second input winding wound around the core of the transformer and controlling the flow of current by a switching operation of the switching element; and a transformer wound around the core of the transformer and including an output winding coupled to the first input winding and the second input winding to extract energy;

an inductor; And

including a first capacitor,

The first input winding of the transformer is tightly coupled with the second input winding,

The inductor and the first input winding are connected in series, and a first capacitor is connected between the serially connected inductor and one end of the first input winding and the other end of the first input winding,

The inductor, the first input winding, the second input winding, and the switching element are connected in series between the two terminals of the input filter capacitor.

Also provided is a product comprising the above-described power supply device according to the present invention.

The present invention eliminates the clamp circuit by generating a spike voltage with low energy accumulated in the leakage inductance of the input winding of the transformer, thereby completely removing the high-frequency noise generated in the clamp circuit, and the spike voltage in the auxiliary winding is very By lowering the induction, the high frequency noise generated by the rectifier diode for auxiliary power is significantly reduced, and thus the influence of noise from the power device to the outside is greatly reduced, thereby reducing the number of parts of the power device and reducing the noise filter cost.

1 is a block diagram of a power supply device according to the prior art.

Fig. 2 is a waveform diagram of voltage and current of each part when the clamp circuit is removed in Fig. 1;

Fig. 3 is a waveform diagram of voltage and current of each part in the case where there is a clamp circuit in Fig. 1;

4 is another configuration diagram of a power supply device according to the prior art.

5 is another configuration diagram of a power supply device according to the prior art.

Fig. 6 is an embodiment of a transformer used in the power supply device of Fig. 5;

7 is an embodiment of a power supply device according to the present invention;

Fig. 8 is an embodiment of a transformer used in the power supply device of Fig. 7;

Fig. 9 is a waveform diagram of voltage and current of each part of the power supply device of Fig. 7;

Fig. 10 is a waveform diagram of voltage and current of each part of the power supply device of Fig. 7;

Fig. 11 is another embodiment of a transformer used in the power supply device of Fig. 7;

12 and 13 are explanatory views of the transformer of FIG. 11;

14 is another embodiment of a power supply device according to the present invention.

15 is another embodiment of a power supply device according to the present invention;

Fig. 16 is an embodiment of a transformer used in the power supply device of Fig. 15;

Fig. 17 is a waveform diagram of voltage and current of each part of the power supply device of Fig. 15;

18 is another embodiment of a power supply device according to the present invention;

Fig. 19 is an embodiment of a transformer used in the power supply device of Fig. 18;

Although this invention is applied to all switched power supply devices, the description according to the embodiment will be described with an example of a flyback converter.

Hereinafter, with reference to the accompanying drawings, the energy accumulated in the leakage inductance of the input winding of the transformer to generate a low spike voltage will be described in more detail and the apparatus.

In the accompanying drawings, black dots or circles filled with black indicated on each winding indicate the starting point of the winding.

7 is an embodiment of a power supply according to the present invention in which energy stored in the leakage inductance of the input winding 261 of the transformer 26 generates a very low spike voltage.

In FIG. 7 which is an embodiment of the power supply device according to the present invention, the AC input voltage is rectified and smoothed by the input filter capacitor 11 . The input winding 261 of the transformer 26 is divided into a first input winding 261a and a second input winding 261b. The inductor 30 and the first input winding 261a are connected in series. A first capacitor 27 is connected between one end and the other end of the inductor 30 connected in series and the first input winding 261a. One end of the inductor 30 connected in series and the first input winding 261a is connected to one terminal of the second input winding 261b. The inductor 30 connected in series and the other end of the first input winding 261a and the other terminal of the second input winding 261b are connected to the “+” terminal of the input filter capacitor 11 and the switching element 12 connected between one terminal. The other terminal of the switching element 12 is connected to the “-” terminal of the input filter capacitor 11 . 7, reference numeral 14 denotes a diode, 15 denotes a capacitor, 21 denotes an input line, 22 denotes an output line, 28 denotes a resistor, 262 denotes an auxiliary winding, 263 denotes an output winding, and 266 denotes the core of the transformer.

In FIG. 7 which is an embodiment of the power supply device according to the present invention, the control unit 13 controls the switching operation of the switching element 12, and the first input winding 261a and the second input winding of the transformer 26 are Controls the amount of energy transferred from 261b to the output winding 263 . The energy transferred to the output winding 263 is rectified by the diode 16 and smoothed by the capacitor 17, and is supplied to the output. The voltage induced in the auxiliary winding 262 is rectified by the diode 14 and smoothed by the capacitor 15 , and is supplied to the control unit 13 as a power supply voltage. The inductor 30 may include a leakage inductance component of the first input winding 261a that is not magnetically coupled to the second input winding 261b and the output winding 263 . The inductance value of the inductor 30 is much larger than the leakage inductance value of the first input winding 261a that is not magnetically coupled to the second input winding 261b and the output winding 263 .

FIG. 8 is an embodiment of a transformer 26 used in the power supply of FIG. 7 .

In the embodiment of Figure 8, the first input winding 261a and the second input winding 261b are formed over the entire main winding section above the third angle 266-3 of the core 266 of the transformer 26. It is wound so as to be magnetically coupled to the output winding 263 . In particular, in order to increase the degree of coupling between the first input winding 261a and the second input winding 261b, the second winding layer of the second input winding 261b after the first winding layer of the second input winding 261b is wound. When this winding starts, it is wound side by side with the rest of the second input winding 261b. Accordingly, most of the magnetic energy accumulated in the leakage inductance of the second input winding 261b, which is not magnetically coupled to the output winding 263, is also extracted from the first input winding 261a.

In Fig. 7, which is an embodiment of the power supply device according to the present invention, the energy accumulated in the leakage inductance of the input winding 261 of the transformer 26 generates the spike voltage very low together with the waveform diagram of Fig. 9 Describe in detail.

In the embodiment of FIG. 7 and the waveform diagram of FIG. 9 , when the switching element 12 is in a conduction state, a current flows through the input winding 261 of the transformer 26 and magnetic energy is accumulated in the transformer 26 . . The current flowing in the first input winding 261a of the transformer 26 flows through the inductor 30 . When the switching element 12 is in the conduction state, the inductance of the input winding 261 is much larger than the inductance of the inductor 30, so the voltage across the input filter capacitor 11 is mostly the first input winding 261a and The second input winding 261b is divided according to the ratio of the number of turns and applied with the polarity indicated in the circle, and the voltage of the first input winding 261a is charged to the polarity indicated in the circle in the first capacitor 27 . When the switching element 12 is turned off at time t1, the magnetic energy accumulated in the first input winding 261a and the second input winding 261b charges the distribution capacity of each winding, and the distribution capacity of each winding is charged. The voltage moves from the voltage of polarity in the circle to the voltage of polarity in the square. At time t3a, when the charging voltage of the distributed capacitance of the output winding 263 becomes higher than the output voltage, the diode 16 is turned on and the magnetic energy accumulated in the transformer 26 is withdrawn through the output winding 263 . The period from time t1 to time t3a is usually within several tens of nS. When the switching element 12 is in a conductive state, the voltage of the first capacitor 27 and the voltage of the first input winding 261a are similar in magnitude and opposite in polarity to each other, and the voltage applied to the inductor 30 is low. After the time t1 when the switching element 12 is turned off, the voltage of the polarity in the circle of the first input winding 261a is quickly lowered and the voltage of the polarity in the circle of the first capacitor 27 is lowered much more slowly, so that the inductor At 30, a voltage of the difference between the voltage of the first capacitor 27 and the voltage of the first input winding 261a is applied, so that the rising speed of the current of the inductor 30 is increased. The rate at which the voltage of the polarity in the circle of the first input winding 261a decreases is much faster than the rate at which the voltage of the polarity in the circle of the first capacitor 27 decreases.

In the embodiment of Fig. 7 and the waveform diagram of Fig. 9, the magnetic energy accumulated in the input winding 261 of the transformer 26 until time t1 is transferred through the inductor through the first input winding 261a after time t1. Current is supplied to (30) and the first capacitor (27). The first input winding 261a is closely coupled to the second input winding 261b, and most of the energy accumulated in the leakage inductance of the second input winding 261b that is not coupled to the output winding 263 is induced. A current is supplied to the inductor 30 and the first capacitor 27 . Accordingly, a significant portion of the energy accumulated in the leakage inductance of the second input winding 261b that is not coupled to the output winding 263 is released while supplying current to the inductor 30 and the first capacitor 27 . The value of the leakage inductance of the second input winding 261b not coupled to the first input winding 261a is determined by the input winding 231 not coupled to the output winding 233 of the transformer 23 of the prior art of FIG. It is much lower than the value of the leakage inductance of . Accordingly, the energy accumulated in the leakage inductance of the second input winding 261b that is not coupled to the first input winding 261a is very small, thus generating a very low spike voltage. Energy accumulated in the leakage inductance of the first input winding 261a that is not coupled to the output winding 263 is released while supplying current to the inductor 30 and the first capacitor 27, so that no spike voltage is generated. don't let

In the embodiment of FIG. 7 and the waveform diagram of FIG. 9 , the first capacitor 27 is charged by the current of the inductor 30 after time t1 to change from the voltage of the polarity in the circle to the voltage of the polarity in the square. Move. At a time t5a when the voltage of the polarity in the square of the first capacitor 27 becomes the same as the voltage of the polarity in the square of the first input winding 261a, the current of the inductor 30 becomes the maximum value.

In the embodiment of FIG. 7 and the waveform diagram of FIG. 9 , during a period of time t5a to t5b, the inductor 30 releases the accumulated magnetic energy, and through the first input winding 261a of the transformer 26 A current flows through the first capacitor 27 , and a voltage higher than the voltage of the polarity in the square of the first input winding 261a is charged to the first capacitor 27 . At time t5b, when the inductor 30 releases all of the accumulated magnetic energy, the voltage of the polarity in the square of the first capacitor 27 becomes the maximum value. After time t5b, the inductor 30 and the first capacitor 27 continue to resonate until energy is dissipated.

In the embodiment of Fig. 7 and the waveform diagram of Fig. 9, the magnetic energy accumulated in the input winding 261 of the transformer 26 before time t1 is, during the period of time t1 to t5b, the first input winding 261a ) through the inductor 30 and the first capacitor 27, the current continuously flows. During this period, the current drawn through the output winding 263 decreases due to the current flowing into the inductor 30 and the first capacitor 27 through the first input winding 261a. Meanwhile, during the period from time t5a to t5b, the leakage inductance component of the first input winding 261a and the inductor 30 charge and discharge the accumulated magnetic energy to the first capacitor 27, so that it has a sinusoidal voltage. . During the period of t5b to t5c, the difference between the voltage of the polarity in the square of the first capacitor 27 and the voltage of the polarity in the square of the first input winding 261a is the leakage inductance component of the first input winding 261a and the inductor (30) is applied, causing a “negative” current to flow in the inductor 30 . During this period, the negative current flowing through the first input winding 261a increases the current drawn through the output winding 263 . Therefore, a part of the electrostatic energy accumulated in the first capacitor 27 at time t5b is withdrawn to the load. At time t5c, the voltage of the first capacitor 27 reaches the voltage of the polarity in the square of the first input winding 261a to end the discharge, and the leakage inductance component of the first input winding 261a and the inductor 30 ), the “negative” current flowing through it becomes the maximum. During the period of time t5c to t5d, the leakage inductance component of the first input winding 261a and the magnetic energy accumulated in the inductor 30 are released while reducing the voltage of the polarity in the square of the first capacitor 27 . During the period of time t5b to t5d, the leakage inductance component of the first input winding 261a and the “negative” current flowing by the inductor 30 increase the current drawn through the output winding 263 . Accordingly, the leakage inductance component of the first input winding 261a and a part of the magnetic energy accumulated in the inductor 30 are withdrawn to the load.

In the embodiment of FIG. 7 and the waveform diagram of FIG. 9, at time t5a, the leakage inductance component of the first input winding 261a of the transformer 26 and the magnetic energy accumulated in the inductor 30 are Accordingly, during the process of repeating the resonance with the first capacitor 27 described above, energy is supplied to the load and disappears.

In the embodiment of FIG. 7 and the waveform diagram of FIG. 9 , the first input winding 261a and the second input winding 261b of the transformer 26 with respect to how the voltages at both ends of the switching element 12 are generated. ) will be described with an example of a turn ratio of 2:8. The voltage at both ends of the switching element 12 is highest at a time t5b when the voltage of the polarity in the square of the first capacitor 27 becomes the maximum value. When AC 270V, which is the maximum input voltage, is applied to the power supply, the voltage of the "+" terminal of the input filter capacitor 11 becomes 380V. During the period in which the switching element 12 is conducting, 76V is applied to both ends of the first input winding 261a in a circle polarity, and the first capacitor 27 has 76V in a circle polarity. Assuming that the voltage across the input winding 261 becomes 100V as the polarity in the box at t3a, which is the time when the diode 16 is turned on, 80V appears at both ends of the second input winding 261b as the polarity in the box, At both ends of the first input winding 261a, 20V appears as a polarity in a square. During the period from time t1 to t5a, the voltage of the first capacitor 27 changes by 96V from 76V with a polarity in a circle to 20V in a polarity in a square. During the period from time t5a to t5b, the voltage across the first capacitor 27 changes by 96V again due to the resonance action with the inductor 30 . Therefore, at time t5b, the voltage across the first capacitor 27 becomes 116V. The voltage Vp2 across the switching element 12 at time t5b is 380V, which is the voltage at the “+” terminal of the input filter capacitor 11, 116V, which is the voltage across the first capacitor 27, and the second input winding 261b. ), the voltage across both ends of 80V is added to make 576V. When a MOS-FET having a breakdown voltage of 700V is used as the switching device 12 , the switching device 12 has a sufficient voltage margin. During the period from time t3a to t4a, the magnetic energy accumulated in the leakage inductance of the second input winding 261b that is not magnetically coupled to the first input winding 261a and the output winding 263 is applied to the second input winding 261a and the output winding 263. Most of the magnetic energy accumulated in the leakage inductance of the second input winding 261b that charges the distribution capacity of 261b and generates a spike voltage, but is not magnetically coupled to the output winding 263, is generated in the first input winding Since it is induced and drawn out at 261a, the spike voltage generated by the leakage inductance of the second input winding 261b is very low. The voltage Vp1 across the switching element 12 at time t4a is the sum of 380V, which is the voltage at the “+” terminal of the input filter capacitor 11, 80V, which is the voltage across the second input winding 261b, and the spike voltage. This voltage is lower than 576V which is the voltage Vp2 of both ends of the switching element 12 at time t5b.

Therefore, the power supply of FIG. 7 according to the present invention is switched even after eliminating the clamp circuit consisting of the resistor 50, the diode 18, the capacitor 19, and the resistor 20 of the prior art power supply of FIG. It has the advantage that the margin of the voltage at both ends of the element 12 can be sufficiently secured and that the high-frequency noise generated in the clamp circuit of the power supply device of the prior art is completely removed. In addition, after the switching element 12 is turned off, instead of the spike voltage at both ends of the input winding 261 of the transformer 26 having a fast rising speed, the resonance of the inductor 30 and the first capacitor 27 causes the Since the generated sine wave voltage appears, the component of high frequency noise included in the voltage of the input winding 261 of the transformer 26 is very low. In addition, since the input winding 261 of the transformer 26 generates a very low spike voltage, a very low spike voltage is also induced in the auxiliary winding 262, and the diode 14 rectifying the voltage of the auxiliary winding 262 also It generates much less high-frequency noise. Accordingly, the power supply device of FIG. 7 has an advantage that the influence of high-frequency noise to the outside through the line is significantly lower than that of the power supply device of the related art of FIG. 1 .

Hereinafter, differences between an embodiment of the power supply device according to the present invention of FIG. 7 and the power supply device of the related art of FIG. 5 will be described.

7, the first input winding 261a has a very high coupling degree with the second input winding 261b. For this reason, the first input winding 261a induces most of the magnetic energy accumulated in the leakage inductance of the second input winding 261b immediately after the switching element 12 is turned off, thereby inducing the inductor 30 and the second input winding 261a. Since the resonant current of the capacitor 27 is supplied, a spike voltage between both terminals of the second input winding 261b is generated very low enough to remove the clamp circuit, and since the clamp circuit is removed, the prior art of FIG. The noise generated in the clamp circuit of the power supply unit is completely removed. In addition, since a very low spike voltage is also induced in the auxiliary winding 262 , the high frequency noise generated by the diode 14 rectifying the voltage of the auxiliary winding 262 is also significantly reduced.

In contrast, in the prior art of FIG. 5, most of the magnetic energy accumulated in the leakage inductance of the second input winding 251b of the transformer 25 cannot be withdrawn through the first input winding 251a, The distribution capacity of the two input windings 251b is charged and a high spike voltage is generated. The spike voltage is limited by a clamp circuit consisting of a resistor (50), a diode (18), a capacitor (19) and a resistor (20). After the switching element 12 is turned off, the current flowing out through the first input winding 251a by the leakage inductance of the first input winding 251a and the resonance of the first capacitor 27 is, Compared to when the current does not flow through the input winding 251a, the magnetic energy of the leakage inductance of the second input winding 251b escapes to the distribution capacity of the input winding 251 much more at a point in time, a spike Let the voltage reach the voltage that conducts the diode 18 of the clamp circuit. As a result, since a small amount of magnetic energy remaining in the leakage inductance of the second input winding 251b conducts the diode 18 of the clamp circuit, high-frequency noise generated by the diode 18 is significantly reduced. However, since the high spike voltage of the second input winding 251b is induced very high in the auxiliary winding 252, the diode 14 rectifying the voltage of the auxiliary winding 252 still generates a large high-frequency noise.

Differences between an embodiment of the power supply device according to the present invention of FIG. 7 and the power supply device of the prior art of FIG. 5 are summarized.

First, while the first input winding 261a of the transformer 26 of the power supply device according to the present invention of FIG. 7 has a very high coupling degree with the second input winding 261b, the power supply device of the prior art of FIG. There is a big difference in that the first input winding 251a of the transformer 25 has a very low coupling degree with the second input winding 251b.

Second, in the power supply device according to the present invention of FIG. 7 , most of the magnetic energy accumulated in the leakage inductance of the second input winding 261b of the transformer 26 is withdrawn by the first input winding 261a and the inductor 30 ) and the resonance energy of the first capacitor 27, the leakage inductance of the second input winding 261b generates a very low spike voltage, whereas the transformer 25 of the prior art power supply of FIG. Since most of the magnetic energy accumulated in the leakage inductance of the second input winding 251b charges the distribution capacity of the second input winding 251b, the leakage inductance of the second input winding 251b generates a very high spike voltage. There is a big difference in creating.

Third, the power supply device according to the present invention of FIG. 7 can eliminate the clamp circuit, whereas the power supply device of the prior art of FIG. 5 lowers only the high-frequency noise generated by the diode 18 of the clamp circuit.

Fourth, in one embodiment of the power supply device according to the present invention of Fig. 7, since the clamp circuit is removed, high-frequency noise generated in the clamp circuit of the power supply device of the prior art is completely removed, whereas the power supply device of the prior art of Fig. 5 There is a big difference in that it reduces the noise generated by the clamp circuit a lot, but cannot completely eliminate it.

Fifth, in one embodiment of the power supply device according to the present invention of FIG. 7, since the input winding 261 of the transformer 26 generates a very low spike voltage, a very low spike voltage is also induced in the auxiliary winding 262, While the diode 14 rectifying the voltage of the auxiliary winding 262 generates much less high-frequency noise, in the power supply device of the prior art of FIG. 5 , the second input winding 251b of the transformer 25 generates Since a very high spike voltage is induced in the auxiliary winding 252, there is a big difference that the diode 14 rectifying the voltage of the auxiliary winding 252 generates high frequency noise.

Such big differences well explain that the embodiment of the power supply device according to the present invention of FIG. 7 has completely different components and operational effects compared to the power supply device of the prior art of FIG. 5 and is much more advanced.

Hereinafter, an operation when the switching device 12 of the embodiment of FIG. 7 is turned on again will be described in detail with the waveform diagram of FIG. 10 .

In the embodiment of Fig. 7 and the waveform diagram of Fig. 10, the discharge of the magnetic energy accumulated in the input winding 261 is completed at time t10. During the period from time t10 to time t11, the charging voltage of the distributed capacitances of the first capacitor 27 and each winding and the capacitance between the terminals of the switching element 12 discharges through each winding, thereby accumulating magnetic energy in each winding. During the period from time t11 to time t12, the magnetic energy of each winding is released again, further discharging the distributed capacitances of each winding and the voltage of the first capacitor 27 . A case in which the turn ratio of the first input winding 261a and the second input winding 261b is 2:8 is taken as an example. At time t10, when the voltage at both ends of the input winding 261 is 100V with the polarity shown in the square shown in FIG. 7 and the voltage at the “+” terminal of the input filter capacitor 11 is 380V, the switching element 12 is turned on At the immediately preceding time t12, the voltage across the input winding 261 becomes 100V with the polarity in the circle shown in FIG. 7, and the voltage across the first input winding 261a and the voltage at the first capacitor 27 are It becomes 20V with the polarity inside. Accordingly, the voltage at the point Q, which is the connection point between the first input winding 261a and the second input winding 261b, becomes 360V. When the switching element 12 is turned on at time t12, a voltage of 360V is applied to the second input winding 261b, and thus a voltage of 90V is induced at both ends of the first input winding 261a. At this time, the voltage across the input winding 261 reaches 450V. Immediately after the switching element 12 is turned on, a voltage of 70V, which is a difference between a voltage of 20V of the first capacitor 27 and a voltage of 90V across both ends of the first input winding 261a, is applied to the inductor 30 . Accordingly, the inductor 30 and the first capacitor 27 generate resonance. At time t14, the voltage across the input winding 261 becomes 380V, which is the voltage at the “+” terminal of the input filter capacitor 11, and the voltage across the second input winding 261b becomes 304V according to the ratio of the number of turns. and the voltage across the first input winding 261a becomes 76V, and the first capacitor 27 is charged up to a voltage of 76V.

In the embodiment of FIG. 7 and the waveform diagram of FIG. 10 according to the present invention, during a period from time t12 to t14, the voltage across the input winding 261 is applied across the input winding 231 of the prior art of FIG. It is higher than the voltage, and since a high reverse voltage may be applied to the diode 16 , consideration such as adjusting the number of turns of the output winding 263 is required.

In FIG. 7 , the resistor 28 connected in series to the path through which the resonance current of the inductor 30 and the first capacitor 27 flows causes the resonance of the inductor 30 and the first capacitor 27 to disappear quickly. . Resistor 28 is optionally used as needed.

When the transformer 26 of FIG. 8 is used in the power supply device of FIG. 7, the inductor 30 is configured as an independent element.

11 is another embodiment of a transformer 26 used in an embodiment of the power supply device according to the present invention of FIG. 7 of FIG. 7 , wherein the inductor 30 is wound around the core of the transformer 26. .

In FIG. 11 , the illustrated reference numerals 1 to 9 are pin numbers of the transformer 26 .

In Fig. 11, the first input winding 261a and the second input winding 261b are connected to the output winding 263 over the entire main winding section above the third angle 266-3 of the core 266 of the transformer 26. ) and coiled to magnetically combine well. In particular, the first input winding 261a is closely coupled to the second input winding 261b, and magnetic energy is accumulated in the leakage inductance of the second input winding 261b that is not magnetically coupled to the output winding 263 . withdraw most of The inductor 30 is divided into a first winding 30a and a second winding 30b connected in series. The first winding 30a of the inductor 30 is wound on the first angle 266-1 of the core 266 of the transformer 26, and the second winding 30b of the inductor 30 is of the transformer 26. It is wound on the second leg 266 - 2 of the core 266 .

12 is an explanatory diagram for explaining a magnetic path generated by the first winding 30a and the second winding 30b of the inductor 30 .

In FIG. 12 , the magnetic flux generated by the first winding 30a of the inductor 30 at the first angle 266-1 of the core 266 is the second angle 266-2 and the second angle 266-2 of the core 266. It flows through the triangle 266-3. The magnetic flux generated by the second winding 30b of the inductor 30 in the second angle 266 - 2 of the core 266 is the first angle 266 - 1 and the third angle 266 - 1 of the core 266 . 3) flows through The magnetic flux generated by the first winding 30a of the inductor 30 and flowing through the third angle 266 - 3 of the core 266 and the second winding 30b of the inductor 30 are generated by the core 266 . The magnetic fluxes flowing through the third angle 266-3 of the are equal in magnitude and in opposite directions, so they cancel each other out, and the first winding 30a and the second winding 30b of the inductor 30 form the third angle of the core 266. The sum of the magnetic fluxes generated at (266-3) approaches “zero”. Accordingly, the magnetic flux generated by the inductor 30 has little effect on the voltages of the input winding 261 and the output winding 263 of the transformer 26 wound around the third angle 266 - 3 of the core 266 . does not

13 is an explanatory diagram for explaining a magnetic path generated by the input winding 261 of the transformer 26 .

In FIG. 13 , the magnetic flux generated in the third angle 266 - 3 of the core 266 by the first input winding 261a and the second input winding 261b of the transformer 26 is the inductor 30 . Since the first winding 30a and the second winding 30b of the inductor 30 generate voltages having opposite polarities and the same magnitude, the input winding 261 of the transformer 26 is connected to both terminals of the inductor 30. Almost no voltage is induced.

14 is another embodiment of the power supply device according to the present invention, and shows the configuration of High Side Switching in which one terminal of the switching element 12 is connected to a “+” input voltage.

14 , the AC input voltage is rectified and smoothed by the input filter capacitor 11 . The input winding 311 of the transformer 31 is divided into a first input winding 311a and a second input winding 311b.

The inductor 30 and the first input winding 311a are connected in series. A first capacitor 27 is connected between one end of the inductor 30 connected in series and one end of the first input winding 311a and the other end of the first input winding 311a. One end of the inductor 30 connected in series and the first input winding 311a is connected to one terminal of the second input winding 311b. The inductor 30 connected in series and the other end of the first input winding 311a and the other terminal of the second input winding 311b are connected to the “-” terminal of the input filter capacitor 11 and the switching element 12 . connected between one terminal. The other terminal of the switching element 12 is connected to the “+” terminal of the input filter capacitor 11 .

14, the voltage on the input winding 311 of the transformer 31 is rectified and smoothed by the diode 18 and the capacitor 19, through the resistor 20a and the capacitor 54, It is supplied as a power supply voltage to the control unit 13 . The resistor 50 limits the peak value of the current flowing through the diode 18 , thereby lowering the high frequency noise generated by the diode 18 . The roles of the first input winding 311a, the second input winding 311b, and the first capacitor 27 of FIG. 14 are the first input winding 261a, the second input winding 261b, and the first capacitor of FIG. It corresponds to the roles in (27). 14, reference numeral 21 denotes an input line, 22 denotes an output line, 28 denotes a resistor, 313 denotes an output winding, 316 denotes a core, and 54 denotes a capacitor. The inductance value of the inductor 30 is much larger than the leakage inductance value of the first input winding 261a that is not magnetically coupled to the second input winding 261b and the output winding 263 .

In the embodiment of FIG. 14 , like the embodiment of FIG. 7 , the energy accumulated in the leakage inductance of the first input winding 311a and the second input winding 311b of the transformer 31 generates a very low spike voltage. The voltage across both ends of the switching element 12 is kept low, and when the diode 18 is turned on, a spike voltage having a fast rising speed is not applied, so noise is reduced when the diode 18 is turned on or off. It has the advantage of generating very low.

In FIG. 14, a resistor 28 connected in series to the path through which the resonance current of the inductor 30 and the first capacitor 27 flows causes the resonance of the inductor 30 and the first capacitor 27 to disappear quickly. . Resistor 28 is optionally used as needed.

15 is another embodiment of a power supply device according to the present invention. 16 is an embodiment of a structural diagram of a transformer 32 used in the power supply device of FIG. 15 , and FIG. 17 is a waveform diagram of each part of the power supply device of FIG. 15 .

In one embodiment of the power supply device according to the present invention of FIG. 15 , the AC input voltage is rectified and smoothed by the input filter capacitor 11 . The input winding of the transformer 32 is divided into a first input winding 321 and a second input winding 322 . The second input winding 322 is connected between the “+” side terminal of the input filter capacitor 11 and one terminal of the switching element 12 . The inductor 30 and the first input winding 321 are connected in series. A first capacitor 27 is connected between one end of the inductor 30 connected in series and the other end of the first input winding 321 . One end and the other end of the inductor 30 and the first input winding 321 connected in series are connected between the other terminal of the switching element 12 and the “-” terminal of the input filter capacitor 11 .

In FIG. 16 which is an embodiment of the transformer 32 used in the power supply device of FIG. 15 , the first input winding 321 and the second input winding 322 are the third angles of the core 326 of the transformer 32 . (326-3) is wound to magnetically couple well with the output winding 323 over the entire main winding section above. In particular, the first input winding 321 is well coupled to the second input winding 322, and the magnetic energy accumulated in the leakage inductance of the second input winding 322 that is not magnetically coupled to the output winding 323 is reduced. withdraw most of The inductor 30 is divided into a first winding 30a and a second winding 30b, and is connected to the first and second angles 326-1 and 326-2 of the core 326 of the transformer 32. cold As described in FIG. 12 , the magnetic flux generated by the inductor 30 wound around the first angle 326-1 and the second angle 326-2 of the core 326 is the third angle ( 326-3) has little influence on the voltages of the first input winding 321, the second input winding 322, and the output winding 323 of the transformer 32. Also, as described with reference to FIG. 13 , the magnetic flux generated by the first input winding 321 and the second input winding 322 of the transformer 32 in the third angle 326 - 3 of the core 326 is the core 326 . ) of the first and

second windings

30a and 30b of the inductor 30 wound around the first and second angles 326-1 and 326-2 of the Therefore, the voltages of the first input winding 321 and the second input winding 322 of the transformer 32 are hardly induced to both terminals of the inductor 30 .

In the embodiment of the power supply device according to the present invention of FIG. 15 , the inductor 30 may be configured as an independent element separated from the transformer 32 .

15, the voltage of the first input winding 321 of the transformer 32 is rectified and smoothed by the diode 14b and the capacitor 15b, and the control unit 13 ) is supplied as the power supply voltage. The roles of the first input winding 321, the second input winding 322, and the first capacitor 27 of the transformer 32 of FIG. 15 are the first input winding 261a and the second input winding 261a of the transformer 26 of FIG. Corresponds to the roles of the two input windings 261b and the first capacitor 27 . The resistor 51 limits the peak value of the current flowing through the diode 14b to lower the high-frequency noise generated by the diode 14b. 16, reference numeral 21 denotes an input line, 22 denotes an output line, 323 denotes an output winding, 326 denotes a core, and 51 denotes a resistor. The inductance value of the inductor 30 is much larger than the leakage inductance value of the first input winding 321 that is not magnetically coupled to the second input winding 322 and the output winding 323 .

In one embodiment of the power supply according to the invention of Figure 15, energy accumulated in the leakage inductances of the first input winding 321 and the second input winding 322 of the transformer 32 substantially generates a spike voltage. The process of not doing it will be described in detail together with the waveform diagram of FIG. 17 .

In the embodiment of FIG. 15 and the waveform diagram of FIG. 17 , when the switching element 12 is in a conductive state, the first input winding 321 , the second input winding 322 , and the inductor 30 of the transformer 32 . ) flows and magnetic energy is accumulated in the transformer 32 and the inductor 30 . When the switching element 12 is in the conduction state, the inductance of the first input winding 321 and the second input winding 322 of the transformer 32 is much larger than the inductance of the inductor 30, so the input filter capacitor ( 11) is mostly applied to the first input winding 321 and the second input winding 322 of the transformer 32 in the ratio of the number of turns and applied with the polarity indicated in the circle, and the first capacitor 27 1 The voltage of the input winding 321 is charged with the polarity indicated in the circle. When the switching element 12 is turned off at time t1, the magnetic energy accumulated in the first input winding 321 and the second input winding 322 of the transformer 32 charges the distribution capacity of each winding, and each winding The charging voltage of the distributed capacitance of is shifted from the voltage of polarity in the circle to the voltage of polarity in the square. At time t3a, when the charging voltage of the distributed capacitance of the output winding 323 becomes higher than the output voltage, the diode 16a is turned on and the magnetic energy accumulated in the transformer 32 is supplied to the output winding 323. The time from time t1 to time t3a is usually within several tens of nS. When the switching element 12 is in a conductive state, the voltage of the first capacitor 27 and the voltage of the first input winding 321 of the transformer 32 are similar in size and opposite in polarity to each other, and are applied to the inductor 30 voltage is low. After the time t1 when the switching element 12 is turned off, the voltage of the polarity in the circle of the first input winding 321 of the transformer 32 is quickly lowered and the voltage of the polarity in the circle of the first capacitor 27 is much higher Since it is slowly lowered, the voltage of the difference between the voltage of the first capacitor 27 and the voltage of the first input winding 321 of the transformer 32 is applied to the inductor 30, so that the rising speed of the current of the inductor 30 is speed up

In the embodiment of FIG. 15 and the waveform diagram of FIG. 17, after time t1, the magnetic energy accumulated in the first input winding 321 and the second input winding 322 of the transformer 32 is transferred to the first input winding ( A current is supplied to the inductor 30 and the first capacitor 27 through 321 . Energy accumulated in the leakage inductance of the first input winding 321 of the transformer 32 is tightly coupled to the second input winding 322 and not coupled to the output winding 323 . Inducing most of the current is supplied to the inductor 30 and the first capacitor 27 . Accordingly, most of the energy accumulated in the leakage inductance of the second input winding 322 that is not coupled to the output winding 323 is emitted while supplying current to the inductor 30 and the first capacitor 27 . The value of the leakage inductance of the second input winding 322 not coupled to the first input winding 321 is determined by the input winding 231 not coupled to the output winding 233 of the conventional transformer 23 of FIG. 1 . It is much lower than the value of the leakage inductance of . Accordingly, energy accumulated in the leakage inductance of the second input winding 322 that is not coupled to the first input winding 321 of the transformer 32 generates a very low spike voltage. Energy accumulated in the leakage inductance of the first input winding 321 that is not coupled to the output winding 323 of the transformer 32 is released while supplying current to the inductor 30 and the first capacitor 27, It does not generate any spike voltage.

In the embodiment of Fig. 15 and the waveform diagram of Fig. 17, the first capacitor 27 is charged by the current of the inductor 30 after time t1 to move from the voltage of the polarity in the circle to the voltage of the polarity in the square. do. The current in the inductor 30 becomes the maximum value at time t5a when the voltage of the polarity in the square of the first capacitor 27 becomes the same as the voltage in the polarity in the square of the first input winding 321 of the transformer 32 .

In the embodiment of FIG. 15 and the waveform diagram of FIG. 17 , during the period of time t5a to t5b, the inductor 30 emits the accumulated magnetic energy, and is discharged through the first input winding 321 of the transformer 32. The first capacitor 27 allows a current to flow, and the first capacitor 27 is charged with a voltage higher than the voltage of the polarity in the square of the first input winding 321 of the transformer 32 . At time t5b, when the inductor 30 releases all of the accumulated magnetic energy, the voltage of the polarity in the square of the first capacitor 27 becomes the maximum value. After time t5b, the inductor 30 and the first capacitor 27 continue to resonate until energy is dissipated.

In the embodiment of Fig. 15 and the waveform diagram of Fig. 17, the magnetic energy accumulated in the second input winding 322 of the transformer 32 before time t1, during the period of time t1 to t5b, is A current continuously flows into the inductor 30 and the first capacitor 27 through the first input winding 321 . During this period, the current drawn through the output winding 323 decreases due to the current flowing through the first input winding 321 of the transformer 32 into the inductor 30 and the first capacitor 27 . Meanwhile, during the period from time t5a to t5b, the leakage inductance component of the first input winding 321 of the transformer 32 and the inductor 30 charge and discharge the accumulated magnetic energy to the first capacitor 27 . Therefore, it has a sinusoidal voltage. During the period of t5b to t5c, the difference between the voltage of the polarity in the square of the first capacitor 27 and the voltage of the polarity in the square of the first input winding 321 of the transformer 32 is the first input winding of the transformer 32 The leakage inductance component of 321 and applied to the inductor 30 causes a “negative” current to flow in the inductor 30 . During this period, the negative current flowing through the first input winding 321 of the transformer 32 increases the current drawn through the output winding 323 . Accordingly, a part of the electrostatic energy accumulated in the first capacitor 27 is withdrawn to the load. At time t5c, the voltage of the first capacitor 27 reaches the voltage of the polarity in the square of the first input winding 321 of the transformer 32 to end the discharge, and the first input winding 321 of the transformer 32 ) and the “negative” current flowing through the inductor 30 become the maximum value. During the period of time t5c to t5d, the leakage inductance component of the first input winding 321 of the transformer 32 and the magnetic energy accumulated in the inductor 30 further increase the voltage of the polarity in the square of the first capacitor 27. lowered and released During the period of time t5b to t5d, the leakage inductance component of the first input winding 321 of the transformer 32 and the “negative” current flowing by the inductor 30 increase the current drawn through the output winding 323. increase Accordingly, the leakage inductance component of the first input winding 321 of the transformer 32 and a part of the magnetic energy accumulated in the inductor 30 are withdrawn to the load.

In the embodiment of Fig. 15 and the waveform diagram of Fig. 17, at time t5a, the leakage inductance component of the first input winding 321 of the transformer 32 and the magnetic energy accumulated in the inductor 30 change over time. Accordingly, during the process of repeating the resonance with the first capacitor 27 described above, energy is supplied to the load and disappears.

In the embodiment of FIG. 15 and the waveform diagram of FIG. 17 , the first input winding 321 and the second input winding 322 of the transformer 32 with respect to how the voltage at both ends of the switching element 12 is generated. A case where the turn ratio of ' is 2:8 will be described as an example. The voltage at both ends of the switching element 12 is highest at a time t5b when the voltage of the polarity in the square of the first capacitor 27 becomes the maximum value. When the input voltage of the power supply is AC 270V which is the maximum value, the voltage of the "+" terminal of the input filter capacitor 11 becomes about 380V. 76V is applied to both ends of the first input winding 321 of the transformer 32 during the period in which the switching element 12 is conducting, and 304V is applied to both ends of the second input winding 322 with the polarity in a circle. is applied, and 76V is charged to both ends of the first capacitor 27 with the polarity in a circle. Assuming that the sum of the voltages of the first input winding 321 and the second input winding 322 of the transformer 32 at t3a, which is the time when the diode 16a is turned on, is 100V with the polarity in the square, the second input At both ends of the winding 322 , 80V appears as a polarity in a square, and 20V appears as a polarity in a square at both ends of the first input winding 321 . During the period from time t1 to t5a, the voltage of the first capacitor 27 changes by 96V from 76V with a polarity in a circle to a voltage of 20V in a polarity in a square. During the period from time t5a to t5b, the voltage of the first capacitor 27 is changed by 96V again due to the resonance action with the inductor 30 . Therefore, at time t5b, the voltage across the first capacitor 27 becomes 116V. The voltage Vp2 across the switching element 12 at time t5b is 380V, the maximum voltage of the "+" terminal of the input filter capacitor 11, 116V, the voltage across the first capacitor 27, and the second input winding ( 322) is added to 80V, which is the voltage across both ends, and it becomes 576V. When a MOS-FET having a breakdown voltage of 700V is used as the switching device 12 , the switching device 12 has a sufficient voltage margin. During the period from time t3a to t4a, the magnetic energy accumulated in the leakage inductance of the second input winding 322 that is not magnetically coupled to the first input winding 321 and the output winding 323 is applied to the second input winding 321 and the output winding 323. Most of the magnetic energy accumulated in the leakage inductance of the second input winding 322 that is not magnetically coupled to the output winding 323 while charging the distribution capacity of the 322 is generated by the first input winding. Since it is induced and pulled out at 321, the generated spike voltage is very low. The voltage Vp1 across both ends of the switching element 12 at time t4a is the sum of 380V, the maximum voltage of the "+" terminal of the input filter capacitor 11, 80V, the voltage across the second input winding 322, and the spike voltage. value, this voltage is lower than 576V which is the voltage Vp2 across both ends of the switching element 12 at time t5b.

Accordingly, an embodiment of the power supply device according to the present invention of FIG. 15 is composed of a resistor 50, a diode 18, a capacitor 19, and a resistor 20 used in the power supply device of the prior art of FIG. It has the advantage of being able to secure a margin of voltage at both ends of the switching element 12 even after removing the clamp circuit and that high-frequency noise generated in the clamp circuit of the power supply device of the prior art is completely removed. In addition, after the switching element 12 is turned off, the inductor 30 and the first input winding 321 and the second input winding 322 of the transformer 32 instead of the spike voltage with a fast rising speed. Since the voltage of a sine wave generated by the resonance of the capacitor 27 appears, the component of high-frequency noise included in the voltages of the first input winding 321 and the second input winding 322 of the transformer 32 is very low. have an advantage In addition, since the second input winding 322 of the transformer 32 generates a very low spike voltage, a very low spike voltage is also induced in the first input winding 321 , and the voltage of the first input winding 321 is rectified. The diode 14b also generates much smaller high-frequency noise. Accordingly, the power supply device of FIG. 15 has an advantage that the influence of high-frequency noise to the outside through the line is significantly lower than that of the power supply device of the related art of FIG. 1 .

In FIG. 15, a resistor 28 connected in series to the path through which the resonance current of the inductor 30 and the first capacitor 27 flows causes the resonance of the inductor 30 and the first capacitor 27 to disappear quickly. . Resistor 28 is optionally used as needed.

In one embodiment of the power supply device according to the present invention of FIG. 15, as described with reference to FIG. 7, when the switching element 12 is turned on again from the OFF state, the second input winding 322 of the transformer 32 Since the voltage applied to the voltage is high and a high reverse voltage can be applied to the diode 16a, it is necessary to consider adjusting the number of turns of the output winding 323.

18 shows an embodiment modified from the embodiment of FIG. 15 .

15, the ratio of the number of turns of the first input winding 321 to the sum of the number of turns of the first input winding 321 of the transformer 32 and the number of turns of the second input winding 322 is In a large case, the reverse voltage applied to the diode 16a connected to the output winding 323 of the transformer 32 may become excessively high when the switching element 12 is turned on again from the OFF state. 18, in preparation for such a case, may be used to prevent the reverse voltage applied to the diode connected to the output winding from becoming excessively high.

In one embodiment of the power supply device according to the present invention of FIG. 18 , the AC input voltage is rectified and smoothed by the input filter capacitor 11 . The input winding of the transformer 33 is divided into a first input winding 331a, a second input winding 332, and a third input winding 331b. The second input winding 332 is connected between the “+” side terminal of the input filter capacitor 11 and one terminal of the switching element 12 . The inductor 30 and the first input winding 331a are connected in series. A first capacitor 27 is connected between one end of the inductor 30 connected in series and one end of the first input winding 331a and the other end of the first input winding 331a. One end of the inductor 30 connected in series and the first input winding 331a is connected to one terminal of the third input winding 331b. The other end of the inductor 30 and the first input winding 331a connected in series and the other terminal of the third input winding 331b are connected to the other terminal of the switching element 12 and the “ -" is connected between terminals.

In one embodiment of the power supply device according to the present invention of FIG. 18 , the sum of the second input winding 332 and the third input winding 331b of the transformer 33 is the second input of the transformer 32 of FIG. 15 . Corresponding to the winding 322 , the first input winding 331a corresponds to the first input winding 321 of the transformer 32 of FIG. 15 . The inductor 30 and the first capacitor 27 correspond to the inductor 30 and the first capacitor 27 of FIG. 15 . The inductance value of the inductor 30 is much larger than the leakage inductance value of the first input winding 331a that is not magnetically coupled to the second input winding 331b and the output winding 333 of the transformer 33 . have 18 shows the sum of the number of turns of the second input winding 332 of the transformer 33 and the number of turns of the third input winding 331b of the transformer 33 and the first input winding 331a ) to properly maintain the ratio of the number of turns, and has the operational characteristics and advantages of the embodiment of FIG. 15 .

19 is an embodiment of a transformer 33 used in the power supply device of FIG. 18 .

Reference numerals 1 to 7 shown in FIG. 19 are pin numbers of the transformer 33 . In Fig. 19, the first input winding 331a, the second input winding 332, and the third input winding 331b are the main ones on the third angle 336-3 of the core 336 of the transformer 33. It is wound so as to be magnetically coupled to the output winding 333 throughout the winding section. In particular, the first input winding 331a is closely coupled to the second input winding 332 , and the magnetic energy accumulated in the leakage inductance of the second input winding 332 that is not magnetically coupled to the output winding 333 is reduced. Coiled to withdraw most.

When the transformer 33 of FIG. 19 is used in the power supply device of FIG. 18, the inductor 30 is configured as an independent element. Although not shown, the inductor 30 is divided into a first winding 30a and a second winding 30b, as shown in FIG. 16 , and the first angle 336 of the core 336 of the transformer 33 of FIG. 19 . -1) and the second angle 336-2 may be wound separately.

As described above, in the power supply device of the present invention, the energy accumulated in the leakage inductance of the input winding of the transformer generates a very low spike voltage, thereby removing the clamp circuit limiting the spike voltage and completely removing the noise generated in the clamp circuit. In addition, a very low spike voltage is induced to the auxiliary winding, and the high-frequency noise generated by the diode rectifying the voltage of the auxiliary winding is significantly lowered, thereby providing a solution that significantly reduces the influence of noise from the power supply to the outside. Nevertheless, those of ordinary skill in the art can apply the solution provided by the present invention to appropriately lower the magnitude of the spike voltage generated by the energy accumulated in the leakage inductance of the input winding of the transformer and use a clamp circuit. , it can also be used as a solution to effectively lower the high-frequency noise generated by the clamp circuit. That is, in this case, the embodiments of FIG. 7 or 14 or 15 or 18 according to the present invention include the inductor 30 and the first input winding 261a and the second input winding 261b of FIG. 7 or The inductor 30 and the first input winding 311a and the second input winding 311b of FIG. 14 or the inductor 30, the first input winding 321 and the second input winding 322 of FIG. 15 or the second input winding 322 of FIG. 18 The voltage between the inductor 30, the first input winding 331a, the second input winding 332, and the third input winding 331b is rectified with a diode and a capacitor to resist the smoothed voltage. It may further include a clamp circuit for discharging by. Or in this case, one embodiment of FIG. 7 or 14 or 15 or 18 according to the present invention limits the voltage between both terminals of the switching element 12 of FIG. 7 or 14 or 15 or 18 To this end, it may further include a clamp circuit including a diode, a capacitor, and a resistor. However, the application and use of the present invention in this way will also fall within the scope of the technical idea of the present invention.

Although the technical idea of this invention has been described with the accompanying drawings in the above, this is an exemplary description of the best embodiment of the present invention and does not limit the present invention. It is a clear fact that anyone with ordinary skill in the art can make various combinations, modifications and imitations without departing from the scope of the technical spirit of the present invention.

11 is an input filter capacitor, 12 is a switching element, 13 is a control unit, 14 and 14b are a diode, 15 and 15b are a capacitor, 16 and 16a are a rectifier, 17 is a capacitor, 18 is a diode, 19 is a capacitor, 20 and 20a are a resistor , 21 is an input line, 22 is an output line, 23 to 26 a transformer, 27 a capacitor, 28 a resistor, 30 an inductor, 31 to 34 a transformer, 50 and 51 a resistor, 52 a capacitor, 53 a resistor, 54 is the capacitor.

Claims

an input filter capacitor; A switched power supply including a switching element, comprising:

the core of the transformer; a first input winding wound around the core of the transformer and controlling the flow of current by a switching operation of the switching element; a second input winding wound around the core of the transformer and controlling the flow of current by a switching operation of the switching element; and a transformer wound around the core of the transformer and including an output winding coupled to the first input winding and the second input winding to extract energy;

an inductor; And

including a first capacitor,

The first input winding of the transformer is tightly coupled with the second input winding,

The inductor and the first input winding are connected in series, and a first capacitor is connected between the serially connected inductor and one end of the first input winding and the other end of the first input winding,

and the inductor, the first input winding, the second input winding, and the switching element are connected in series between the two terminals of the input filter capacitor.
The power supply device of claim 1, wherein one end of the inductor and the first input winding connected in series is connected to one terminal of the input filter capacitor, and the other end of the serially connected inductor and the first input winding is connected to one terminal of the second input winding, the other terminal of the second input winding is connected to one terminal of the switching element, and the other terminal of the switching element is connected to the other terminal of the input filter capacitor. Switched power supply, characterized in that.
The power supply device of claim 1, wherein one terminal of the second input winding is connected to one terminal of the input filter capacitor, and the other terminal of the second input winding is connected in series between the inductor and the first input winding. The inductor connected to one end and the other end of the first input winding connected in series is connected to one terminal of the switching element, and the other terminal of the switching element is connected to the other terminal of the input filter capacitor. A switching type power supply, characterized in that.
The power supply device of claim 1, wherein the inductor connected in series and one end and the other end of the first input winding are connected between one terminal of the input filter capacitor and one terminal of the switching element, and the second input A winding is connected between the other terminal of the switching element and the other terminal of the input filter capacitor.
The power supply device according to claim 1, wherein the core of the transformer has one leg on each side and one leg in the middle, and the first input winding, the second input winding, and the output winding of the transformer are wound around a middle leg of a core of a transformer, the inductor is divided into a first winding and a second winding, the first winding of the inductor is wound on a leg next to the core of the transformer, and the second winding of the inductor is A switched power supply device, characterized in that the first winding of the inductor and the second winding of the inductor are connected in series by being wound around a leg on the other side of the core of the transformer.
The power supply device of claim 1, wherein the transformer further comprises a third input winding wound around a core of the transformer and controlling the flow of current by a switching operation of the switching element, wherein the first input winding of the transformer is tightly coupled to the second input winding and the third input winding, and the inductor, the first input winding, the second input winding, the third input winding, and the switching element are two terminals of the input filter capacitor. A switched power supply, characterized in that it is connected in series between.
The power supply device according to claim 6, wherein the core of the transformer has one leg on each side and one leg in the middle, and the first input winding, the second input winding, and the third input winding of the transformer and the output winding is wound on a middle leg of the core of the transformer, the inductor is divided into a first winding and a second winding, the first winding of the inductor is wound on a leg next to the core of the transformer, and the inductor A second winding of the transformer is wound around a leg on the other side of the core of the transformer, so that the first winding of the inductor and the second winding of the inductor are connected in series.
The power supply device of claim 6, wherein the second input winding is connected between one terminal of the input filter capacitor and one terminal of the switching element, and the other terminal of the switching element is a terminal of the third input winding. connected, the other terminal of the third input winding is connected to one end of the inductor and the first input winding connected in series, and the other end of the inductor and the first input winding connected in series is connected to the input filter capacitor Switching power supply, characterized in that connected to the other terminal of the.
The power supply device of claim 6, wherein the second input winding is connected between one terminal of the input filter capacitor and one terminal of the switching element, and the other terminal of the switching element is connected to the inductor and the first terminal in series. The inductor connected to one end of the input winding and the other end of the first input winding connected in series is connected to one terminal of the third input winding, and the other terminal of the third input winding is the input filter capacitor. Switching power supply, characterized in that connected to the other terminal of the.
The power supply device of claim 1,

The inductor connected in series and the first capacitor 27 connected between one end and the other end of the first input winding are connected, and the inductor and the first input winding are connected in series, connected in series and a first resistor connected in series with a first capacitor connected between the inductor and one end of the first input winding and the other end of the first input winding, wherein the first resistance is a resonance between the inductor and the first input winding. A switched power supply, characterized in that it is rapidly dissipated.
The power supply device of claim 1, wherein a voltage between both terminals of at least one of the first input winding, the second input winding and the inductor is rectified and smoothed by a second diode and a second capacitor, and both terminals of the second capacitor are rectified and smoothed. A switched power supply further comprising a clamp circuit for discharging a voltage smoothed by a second resistor connected therebetween. Between one end of the inductor and the first input winding connected in series and the other end of the other end The first capacitor 27 connected to is connected, the inductor and the first input winding are connected in series, and the inductor connected in series and one end of the first input winding and the other end are connected and a first resistor connected in series with a first capacitor, wherein the first resistor causes resonance between the inductor and the first input winding to be quickly extinguished.
The power supply device of claim 6, wherein a voltage between both terminals of the first input winding, the second input winding, the third input winding, and at least one of the inductor is rectified and smoothed by a second diode and a second capacitor; A switching type power supply further comprising a clamp circuit for discharging a voltage smoothed by a second resistor connected between both terminals of a second capacitor. The inductor connected in series and one side of the first input winding The first capacitor 27 connected between one end and the other end is connected, the inductor and the first input winding are connected in series, and the inductor connected in series and one end of the first input winding are different from each other. A switched power supply device, characterized in that it further comprises a first resistor connected in series with a first capacitor connected between one end, wherein the first resistance causes the resonance of the inductor and the first input winding to quickly disappear. .
A manufactured article comprising the switched power supply of any one of claims 1 to 12.