WO2020152792A1 - Drive apparatus, compressor, refrigeration air conditioning apparatus, and electric motor driving method - Google Patents

Drive apparatus, compressor, refrigeration air conditioning apparatus, and electric motor driving method Download PDF

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Publication number
WO2020152792A1
WO2020152792A1 PCT/JP2019/001935 JP2019001935W WO2020152792A1 WO 2020152792 A1 WO2020152792 A1 WO 2020152792A1 JP 2019001935 W JP2019001935 W JP 2019001935W WO 2020152792 A1 WO2020152792 A1 WO 2020152792A1
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WO
WIPO (PCT)
Prior art keywords
electric motor
inverter
carrier frequency
frequency
voltage
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Application number
PCT/JP2019/001935
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French (fr)
Japanese (ja)
Inventor
勇二 廣澤
昌弘 仁吾
Original Assignee
三菱電機株式会社
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Application filed by 三菱電機株式会社 filed Critical 三菱電機株式会社
Priority to PCT/JP2019/001935 priority Critical patent/WO2020152792A1/en
Priority to JP2020567286A priority patent/JPWO2020152792A1/en
Publication of WO2020152792A1 publication Critical patent/WO2020152792A1/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters

Definitions

  • the present invention relates to a drive device that drives an electric motor.
  • a pulse width modulation control method is used to control an inverter that applies a voltage to an electric motor.
  • the carrier frequency of the inverter affects the switching loss in the inverter and the harmonic iron loss in the electric motor. Therefore, a driving method of an electric motor that switches the carrier frequency according to the state of the current waveform output from the inverter has been proposed (for example, refer to Patent Document 1).
  • the conventional technology has a problem that the total efficiency of the inverter and the electric motor is poor because the total loss of the inverter and the electric motor, such as switching loss and harmonic iron loss, is large.
  • the purpose of the present invention is to improve the overall efficiency of the inverter and the electric motor.
  • a drive device is A drive device for driving an electric motor having a coil, An inverter for applying a voltage to the coil, A controller for controlling the carrier frequency of the inverter for adjusting the control frequency of the voltage applied to the coil, and switching the carrier frequency according to the rotation speed of the electric motor.
  • a compressor includes the drive device.
  • a refrigerating and air-conditioning apparatus includes the compressor.
  • a method of driving an electric motor according to another aspect of the present invention is A method for driving an electric motor having a coil, comprising: Setting the carrier frequency of the inverter for adjusting the control frequency of the voltage applied to the coil, The carrier frequency is switched according to the rotation speed of the electric motor.
  • the overall efficiency of the inverter and the electric motor can be improved.
  • FIG. 1 It is sectional drawing which shows the structure of an electric motor schematically. It is a block diagram which shows the structure of a drive device. It is a figure which shows an example of the carrier wave and inverter output voltage command value for generating a PWM control signal. It is a figure which shows an example of the PWM control signal produced
  • FIG. 1 is a sectional view schematically showing the structure of the electric motor 1.
  • FIG. 1 shows the structure of the electric motor 1 in a plane orthogonal to the rotation axis of the rotor 20 of the electric motor 1.
  • the electric motor 1 is, for example, a permanent magnet embedded electric motor.
  • the electric motor 1 is used, for example, in a rotary compressor.
  • the electric motor 1 has a stator 10 and a rotor 20 rotatably provided inside the stator 10.
  • An air gap exists between the inner peripheral surface of the stator 10 and the outer peripheral surface of the rotor 20.
  • the width of the air gap is, for example, 0.3 mm to 1 mm.
  • the axial direction of the rotor 20 (that is, the direction of the rotation axis C1) is simply referred to as the “axial direction”.
  • the direction along the outer circumference of the stator 10 or the rotor 20 is simply referred to as “circumferential direction”.
  • the radial direction of the stator 10 or the rotor 20 is simply referred to as “radial direction”.
  • the stator 10 includes a stator core 11, at least one coil (three coils 3U, three coils 3V, and three coils 3W in FIG. 1) and at least one insulator 14.
  • the stator core 11 is made of, for example, a plurality of electromagnetic steel plates.
  • the plurality of magnetic steel sheets are laminated in the axial direction.
  • the plurality of magnetic steel sheets are fixed by caulking.
  • the thickness of each electromagnetic steel sheet is, for example, 0.1 mm to 0.7 mm. In the present embodiment, the thickness of each electromagnetic steel plate is 0.35 mm.
  • Each electromagnetic steel sheet is formed into a predetermined shape by press working such as punching.
  • the stator core 11 has an annular yoke portion 13 and a plurality of teeth portions 12.
  • the yoke portion 13 extends in the circumferential direction.
  • Each tooth part 12 extends in the radial direction.
  • each tooth portion 12 projects from the yoke portion 13 toward the rotation axis C1 of the rotor 20.
  • the plurality of tooth portions 12 are arranged at equal intervals in the circumferential direction and extend radially.
  • the stator core 11 has nine teeth portions 12.
  • a slot exists between two adjacent tooth portions 12.
  • Each tooth portion 12 has a main body portion that extends in the radial direction and a tooth tip portion that extends in the circumferential direction.
  • Each tooth tip is located at the tip of each tooth 12.
  • Each of the coils 3U, 3V, and 3W is wound around the insulator 14 fixed to the stator core 11.
  • Coils 3U, 3V and 3W are made of at least one stator winding. That is, at least one stator winding is wound around the stator core 11.
  • the at least one stator winding is, for example, a magnet wire.
  • each of coils 3U, 3V, and 3W is wound 110 turns around insulator 14 by concentrated winding.
  • the number of turns and the wire diameter of each of the coils 3U, 3V, and 3W are determined according to the characteristics (for example, rotation speed, torque) required for the electric motor 1, the supply voltage, or the cross-sectional area of the slot.
  • the stator core 11 may be formed of, for example, a plurality of blocks (also referred to as split cores).
  • the magnet wire can be wound around each tooth portion 12 in a state where these plural blocks are linearly arranged. After winding the magnet wire around the tooth portion 12, the plurality of blocks are folded in an annular shape, and both ends of the plurality of blocks are welded.
  • the coil is composed of U-phase, V-phase, and W-phase three-phase coils (that is, coils 3U, 3V, and 3W).
  • the coil 3U has U-phase coil portions Ua, Ub, and Uc.
  • the coil 3V has V-phase coil portions Va, Vb, and Vc.
  • Coil 3W has W-phase coil portions Wa, Wb, and Wc. Both terminals of the coil of each phase are open. That is, the coil has a total of six terminals.
  • the insulator 14 is made of, for example, a film made of polyethylene terephthalate (PET).
  • the insulator 14 made of a thin film is effective in increasing the number of turns of the coil in the slot. Furthermore, the stator core 11 made of a plurality of blocks is effective in increasing the number of turns of the coil in the slot.
  • the stator core 11 is not limited to the structure made of a plurality of blocks.
  • the rotor 20 has a rotor core 21 and a plurality of permanent magnets 25 attached to the rotor core 21.
  • the rotor core 21 is made of, for example, a plurality of electromagnetic steel plates.
  • the plurality of magnetic steel sheets are laminated in the axial direction.
  • the plurality of magnetic steel sheets are fixed by caulking.
  • the thickness of each electromagnetic steel sheet is, for example, 0.1 mm to 0.7 mm. In the present embodiment, the thickness of each electromagnetic steel plate is 0.35 mm.
  • Each electromagnetic steel sheet is formed into a predetermined shape by press working such as punching.
  • the rotor core 21 has a plurality of magnet insertion holes 22 and a shaft hole 27. A shaft is inserted into the shaft hole 27. The shaft is fixed to the rotor core 21 by a method such as shrink fitting and press fitting.
  • the rotor core 21 has six magnet insertion holes 22. On the surface orthogonal to the axial direction, the plurality of magnet insertion holes 22 are arranged in the circumferential direction. Each magnet insertion hole 22 corresponds to each magnetic pole of the rotor 20. The number of magnetic poles of the rotor 20 is 2 or more. In the present embodiment, the rotor 20 has six magnetic poles. One or more permanent magnets 25 are arranged in each magnet insertion hole.
  • each magnet insertion hole 22 On the plane orthogonal to the axial direction, the central portion of the magnet insertion hole 22 projects toward the rotation axis C1. That is, each magnet insertion hole 22 has a V shape in the plane orthogonal to the axial direction.
  • the shape of each magnet insertion hole 22 is not limited to the V shape, and may be, for example, a straight shape.
  • two permanent magnets 25 are arranged in one magnet insertion hole 22. That is, two permanent magnets 25 are arranged for one magnetic pole. Therefore, in this embodiment, the rotor 20 has twelve permanent magnets 25.
  • Each permanent magnet 25 is a flat magnet that is long in the axial direction.
  • Each permanent magnet 25 is, for example, a rare earth magnet containing neodymium (Nd), iron (Fe), and boron (B).
  • Two permanent magnets 25 arranged in one magnet insertion hole 22 serve as one magnetic pole of the rotor 20.
  • the rotor core 21 further has a plurality of flux barriers 26.
  • Flux barriers 26 are located on both sides of each magnet insertion hole 22 in the circumferential direction.
  • Each flux barrier 26 is a void communicating with the magnet insertion hole 22.
  • Each flux barrier 26 reduces the leakage magnetic flux between adjacent magnetic poles (that is, the magnetic flux passing between the magnetic poles).
  • the width of the thin portion between each flux barrier 26 and the outer peripheral surface of the rotor core 21 on the surface orthogonal to the axial direction is, for example, the same as the thickness of the electromagnetic steel sheet of the rotor core 21.
  • each magnet insertion hole 22 has a first magnet holding portion 23, which is a protrusion, formed in the central portion in the circumferential direction. Further, in the rotor core 21, second magnet holding portions 24, which are protrusions, are formed at both ends in the circumferential direction of each magnet insertion hole 22. The first magnet holding portion 23 and the second magnet holding portion 24 hold the permanent magnet 25 in each magnet insertion hole 22.
  • FIG. 2 is a block diagram showing the configuration of the driving device 101.
  • the drive device 101 includes a converter 102 that rectifies an output of a power supply, an inverter 103 that applies a voltage (specifically, an AC voltage) to the coil 3 of the electric motor 1, and a control device 50.
  • the coils 3U, 3V, and 3W are collectively referred to as a coil 3.
  • Electric power is supplied to the converter 102 from a power supply which is an alternating current (AC) power supply.
  • the converter 102 applies a voltage to the inverter 103.
  • the voltage applied from the converter 102 to the inverter 103 is also referred to as “converter voltage”.
  • the bus voltage of converter 102 is supplied to control device 50.
  • the inverter voltage that drives the electric motor 1, that is, the voltage applied to the coil 3 of the electric motor 1 is generated by the PWM control method.
  • the coil 3 of the electric motor 1 is, for example, a three-phase coil.
  • the inverter 103 has at least one inverter switch corresponding to each phase, and each inverter switch has one set of switching elements (two switching elements in the present embodiment).
  • the waveform of the inverter voltage is generated by controlling the on/off time ratio of the inverter switch corresponding to each phase.
  • a desired output waveform from the inverter 103 can be obtained.
  • the inverter switch in the inverter 103 is turned on, a voltage is supplied from the inverter 103 to the coil 3 and the inverter voltage increases.
  • the inverter switch is off, the voltage supply from the inverter 103 to the coil 3 is cut off and the inverter voltage drops.
  • the difference between the inverter voltage and the induced voltage is supplied to the coil 3, the electric motor current is generated, and the rotational force of the electric motor 1 is generated.
  • a desired output waveform from the inverter 103 can be obtained by controlling the on/off time ratio of the inverter switch so as to match the target electric motor current value.
  • FIG. 3 is a diagram showing an example of a carrier wave and an inverter output voltage command value for generating a PWM control signal.
  • FIG. 4 is a diagram showing an example of the PWM control signal generated in the control device 50.
  • FIG. 5 is a diagram showing an example of the electric motor current generated based on the PWM control signal.
  • the on/off timing of each inverter switch is determined based on the carrier wave.
  • the carrier wave is composed of a triangular wave having a constant amplitude.
  • the pulse width modulation period in the PWM control method is determined by the carrier frequency which is the frequency of the carrier wave.
  • a predetermined carrier wave pattern or a predetermined carrier frequency is stored in control device 50.
  • the control device 50 controls the carrier frequency and controls on/off of each inverter switch. As a result, the control device 50 controls the output from the inverter 103 supplied to the coil 3.
  • the carrier frequency which is the frequency of the carrier wave, is also referred to as the “carrier frequency of the inverter 103”. That is, the carrier frequency of the inverter 103 is the control frequency of the voltage applied to the coil 3, and the control device 50 controls the carrier frequency of the inverter 103.
  • the inverter 103 has three inverter switches (that is, six switching elements), but one of the three inverter switches, that is, the U phase, the V phase, Alternatively, control for one inverter switch corresponding to the W phase will be described. However, the control for the one inverter switch can be applied to the control for the other two inverter switches.
  • the control device 50 compares the voltage value of the carrier wave with the inverter output voltage command value.
  • the inverter output voltage command value is calculated, for example, in the control device 50 based on the target electric motor current value.
  • the inverter output voltage command value is set, for example, based on a signal input to the control device 50 from a remote controller of a refrigerating and air conditioning device such as an air conditioner.
  • the inverter output voltage command value corresponds to, for example, the air volume from the refrigeration air conditioning device set by the user of the refrigeration air conditioning device.
  • control device 50 When the voltage value of the carrier wave is smaller than the inverter output voltage command value, the control device 50 turns on the PWM control signal so that the inverter switch turns on. When the voltage value of the carrier wave is equal to or higher than the inverter output voltage command value, control device 50 turns off the PWM control signal so that the inverter switch turns off. As a result, the inverter voltage approaches the target value.
  • control device 50 generates the PWM control signal based on the difference between the inverter output voltage command value and the carrier wave voltage value.
  • the control device 50 outputs a control signal such as an inverter drive signal based on the PWM control signal to the inverter 103 to control ON/OFF of the inverter switch.
  • the inverter drive signal may be the same signal as the PWM control signal or a signal different from the PWM control signal.
  • the inverter voltage is output from the inverter 103 when the inverter switch is on.
  • the inverter voltage is supplied to the coil 3, and a motor current (specifically, U-phase current, V-phase current, and W-phase current) is generated in the motor 1.
  • a motor current specifically, U-phase current, V-phase current, and W-phase current
  • the inverter voltage is converted into the rotational force of the electric motor 1 (specifically, the rotor 20).
  • the electric motor current is measured by a measuring device such as a current sensor, and the measurement result (for example, a signal indicating a current value) is transmitted to the control device 50.
  • the control device 50 is composed of, for example, a processor and a memory.
  • the control device 50 is a microcomputer.
  • the control device 50 may be configured with a processing circuit as dedicated hardware such as a single circuit or a composite circuit.
  • FIG. 6 is a diagram showing the relationship between the actual inverter voltage and the line voltage.
  • FIG. 7 is a diagram showing an example of the waveform of the electric motor current during low speed rotation.
  • FIG. 8 is a diagram showing an example of the waveform of the electric motor current during rotation at a medium speed.
  • FIG. 9 is a diagram showing an example of the waveform of the electric motor current during high speed rotation.
  • an induced voltage is generated in the coil by electromagnetic induction between the permanent magnet and the coil (for example, three-phase coil).
  • the higher the rotational speed of the electric motor that is, the rotational speed of the electric motor, the larger the induced voltage and the line voltage.
  • the difference between the inverter voltage and the line voltage is converted into the rotational force of the electric motor.
  • the instantaneous inverter output voltage ie, the actual inverter voltage
  • the switching loss of the inverter increases.
  • the carrier frequency that optimizes the efficiency is determined from the balance between the harmonic iron loss and the switching loss, and the smaller the proportion of the harmonic iron loss, the smaller the carrier frequency that optimizes the efficiency.
  • the control device 50 switches the carrier frequency according to the rotation speed of the electric motor 1.
  • the rotation speed of the electric motor 1 means the rotation speed of the electric motor 1. That is, the control device 50 switches the carrier frequency according to the rotation speed of the electric motor 1.
  • FIG. 10 is a flowchart showing an example of the driving method of the electric motor 1.
  • FIG. 11 is a diagram showing an example of the relationship between the rotation speed of the electric motor 1 and the carrier frequency.
  • control device 50 sets the carrier frequency to the frequency f1 as the first frequency (step ST1).
  • the control device 50 switches the carrier frequency according to the rotation speed of the electric motor 1. Specifically, control device 50 determines whether the rotation speed of electric motor 1 has reached threshold value Nd (step ST2). When the rotation speed of the electric motor 1 is equal to or higher than the threshold value Nd (Yes in step ST2), the control device 50 sets the carrier frequency to the frequency f2 as the second frequency smaller than the frequency f1 (step ST3). When the rotation speed of the electric motor 1 has not reached the threshold value Nd (No in step ST2), the control device 50 does not switch the carrier frequency.
  • the frequency f2 is equal to or higher than the minimum carrier frequency fm.
  • the minimum carrier frequency fm is the minimum value that can generate the waveform of the electric motor current required to control the electric motor 1.
  • FIG. 12 is a diagram showing the relationship between the rotation speed of the electric motor 1 and the line voltage.
  • the line voltage increases as the rotation speed of the electric motor 1 increases and saturates at the maximum output of the inverter 103. Therefore, field weakening control is started in order to further increase the rotation speed of the electric motor 1.
  • field weakening control tends to reduce the efficiency of the electric motor.
  • the line voltage is saturated with respect to the converter voltage while the electric motor 1 is driven by the field weakening control.
  • the field weakening control since the difference between the inverter voltage and the line voltage is small, the high frequency component of the motor current is unlikely to occur. Therefore, in the field weakening control, the ratio of the harmonic iron loss to the iron loss generated in the electric motor 1 is particularly small.
  • the control device 50 starts the field weakening control for the electric motor 1, the control device 50 lowers the carrier frequency more than before the field weakening control is started.
  • the control device 50 starts field weakening control and lowers the carrier frequency.
  • FIG. 13 is a diagram showing the relationship between the ratio of the on time of the inverter switch during the pulse width modulation control of the inverter switch of the inverter 103 and the carrier frequency.
  • the pulse width modulation control for the inverter switch of the inverter 103 is also referred to as “PWM control”.
  • the control device 50 may set the carrier frequency according to the proportion of the ON time of the inverter switch during PWM control. For example, when the ratio of the on time of the inverter switch during PWM control is smaller than the threshold value Nt, the control device 50 sets the carrier frequency to the first frequency and the ratio of the on time of the inverter switch is equal to or greater than the threshold value Nt. At this time, the control device 50 sets the carrier frequency to the second frequency which is lower than the first frequency.
  • the ratio of the on time of the inverter switch reaches the threshold value Nt.
  • the carrier frequency is the frequency f1 as the first frequency.
  • the ON time ratio of the inverter switch of the inverter 103 is equal to or higher than the threshold value Nt, the carrier frequency is the frequency f2 as the second frequency smaller than the frequency f1.
  • the frequency f1 is, for example, 9000 Hz
  • the frequency f2 is, for example, 4500 Hz.
  • the threshold value Nt is 1/ ⁇ 2.
  • the threshold value Nt may be 1/ ⁇ 2 or more.
  • the electric motor 1 when the ratio of the on time of the inverter switch of the inverter 103 is 1/ ⁇ 2 or more, the electric motor 1 is driven by the field weakening control.
  • the line voltage is saturated with respect to the converter voltage while the electric motor 1 is driven by the field weakening control.
  • the field weakening control since the difference between the inverter voltage and the line voltage is small, the high frequency component of the motor current is unlikely to occur. Therefore, in the field weakening control, the ratio of the harmonic iron loss to the iron loss generated in the electric motor 1 is particularly small.
  • the control device 50 lowers the carrier frequency as compared with before the field weakening control. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved.
  • the converter 102 may have a booster circuit that boosts the voltage applied to the inverter 103, that is, the converter voltage. As the converter voltage becomes insufficient with respect to the line voltage, the field weakening current during field weakening control increases and copper loss increases. Therefore, in the field weakening control, it is desirable to boost the converter voltage until it balances with the line voltage.
  • FIG. 14 is a diagram showing the relationship between the converter voltage and the line voltage. That is, in the field weakening control, as shown in FIG. 14, it is desirable to boost the converter voltage to a voltage at which the field weakening control is started. As a result, the line voltage is saturated with respect to the converter voltage, and the difference between the inverter voltage and the line voltage is reduced. As a result, it is possible to suppress an increase in harmonic iron loss.
  • converter 102 (specifically, the booster circuit) starts boosting the converter voltage
  • control device 50 starts the field weakening control.
  • FIG. 15 is a diagram showing the relationship between the rotation speed of the electric motor 1 and the ratio of the ON time of the inverter switch during PWM control.
  • the booster circuit of the converter 102 may boost the voltage applied to the inverter 103 so that the ON time ratio of the inverter switch is 1/ ⁇ 2.
  • the rotation speed of the electric motor 1 reaches Nd [rpm]
  • the on-time ratio of the inverter switch reaches 1/ ⁇ 2.
  • the converter 102 may boost the converter voltage.
  • the booster circuit of the converter 102 boosts the voltage applied to the inverter 103 so that the ON time ratio of the inverter switch of the inverter 103 becomes 1/ ⁇ 2.
  • the line voltage is saturated with respect to the converter voltage, and the difference between the inverter voltage and the line voltage is reduced. As a result, it is possible to suppress an increase in harmonic iron loss.
  • FIG. 16 is a diagram showing carrier frequencies before and after boosting the converter voltage.
  • the carrier frequency after the voltage applied to the inverter 103 is boosted is lower than the carrier frequency before the voltage applied to the inverter 103 is boosted. That is, after boosting the converter voltage, control device 50 lowers the carrier frequency as compared with before boosting the converter voltage.
  • controller 50 sets the carrier frequency to f1 before the converter voltage is boosted, and controller 50 sets the carrier frequency to f2 after the converter voltage is boosted.
  • switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved.
  • a silicon carbide (SiC) element for example, is used for at least one switching element of the inverter 103. Thereby, switching loss can be reduced.
  • SiC silicon carbide
  • a gallium nitride (GaN) element may be used for at least one switching element of the inverter 103, for example. Thereby, switching loss can be reduced. In particular, when the electric motor 1 rotates at high speed, the ratio of switching loss to harmonic iron loss tends to increase, so that the efficiency of the electric motor 1 during high speed rotation of the electric motor 1 can be improved.
  • GaN gallium nitride
  • the threshold value of the rotation speed of the electric motor 1 is not limited to one. Therefore, two or more threshold values may be set. In this case, the control device 50 lowers the carrier frequency stepwise. That is, the carrier frequency may be set in two or more patterns.
  • FIG. 17 is a diagram showing the relationship between the rotation speed of the electric motor 1 and the carrier frequency.
  • the frequencies f1 [Hz] to f3 [Hz] satisfy f1>f2>f3 ⁇ fm.
  • the rotation speeds N1 [rpm] and N2 [rpm] satisfy N1 ⁇ N2.
  • the control device 50 switches the carrier frequency stepwise according to the rotation speed of the electric motor 1. Specifically, the control device 50 gradually lowers the carrier frequency according to the rotation speed of the electric motor 1. That is, in the example shown in FIG. 17, when the rotation speed of electric motor 1 reaches N1, control device 50 lowers the carrier frequency from f1 to f2. When the rotation speed of the electric motor 1 reaches from N1 to N2, the control device 50 lowers the carrier frequency from f2 to f3.
  • FIG. 18 is a diagram showing the relationship between the rotation speed of the electric motor 1 and the carrier frequency.
  • controller 50 may decrease the carrier frequency as the rotation speed of electric motor 1 increases.
  • the control device 50 continuously lowers the carrier frequency as the rotation speed of the electric motor 1 increases. Further, when the rotation speed of the electric motor 1 reaches Nd [rpm], the control device 50 starts the field weakening control.
  • the carrier frequency is constant during the field weakening control. That is, during the field weakening control, the control device 50 keeps the carrier frequency constant. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved.
  • FIG. 19 is a diagram showing the relationship between the on-time ratio of the inverter switch and the carrier frequency during pulse width modulation control of the inverter switch of the inverter 103. As shown in FIG. 19, the controller 50 may decrease the carrier frequency as the ratio of the on time of the inverter switch increases.
  • the control device 50 continuously lowers the carrier frequency as the ratio of the on time of the inverter switch increases. Further, when the ratio of the on time of the inverter switch reaches the threshold value Nt, the control device 50 starts the field weakening control.
  • the carrier frequency is constant during the field weakening control. That is, during the field weakening control, the control device 50 keeps the carrier frequency constant. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved.
  • control device 50 switches the carrier frequency according to the rotation speed of electric motor 1. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved. As a result, the efficiency of the driving device 101 can be improved.
  • the control device 50 preferably lowers the carrier frequency according to the rotation speed of the electric motor 1. As a result, switching loss is effectively reduced during high-speed rotation of the electric motor 1, and the overall efficiency of the inverter 103 and the electric motor 1 can be effectively improved. As a result, the efficiency of the driving device 101 can be effectively improved.
  • the control device 50 when the control device 50 starts the field weakening control for the electric motor 1, the control device 50 lowers the carrier frequency compared to before the field weakening control is started. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved. As a result, the efficiency of the driving device 101 can be improved.
  • the electric motor 1 is driven by the field weakening control.
  • the control device 50 lowers the carrier frequency compared to before the field weakening control. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved. As a result, the efficiency of the driving device 101 can be improved.
  • ⁇ Field weakening control increases field weakening current and copper loss. Therefore, in the field weakening control, it is desirable to boost the converter voltage until it balances with the line voltage. This alleviates the field weakening current and reduces copper loss.
  • boosting the converter voltage it is desirable to weaken the converter voltage to a voltage at which field control is started. As a result, the line voltage is saturated with respect to the converter voltage, and the difference between the inverter voltage and the line voltage is reduced. As a result, it is possible to suppress an increase in harmonic iron loss.
  • the control device 50 when boosting the converter voltage, the control device 50 lowers the carrier frequency more than before it boosts it.
  • the lowering of the carrier frequency contributes to the reduction of switching loss.
  • the total efficiency of the inverter 103 and the electric motor 1 can be improved.
  • the efficiency of the driving device 101 can be improved.
  • a silicon carbide (SiC) element or a gallium nitride (GaN) element is used for at least one switching element of the inverter 103.
  • switching loss can be reduced.
  • the ratio of harmonic iron loss is small and the ratio of switching loss easily increases, so that the efficiency of the electric motor 1 at high speed rotation of the electric motor 1 can be improved.
  • the efficiency of the driving device 101 can be improved.
  • FIG. 20 is a sectional view schematically showing the structure of the compressor 6 according to the second embodiment.
  • the compressor 6 includes the drive device 101 according to the first embodiment, an electric motor 60 as an electric element driven by the drive device 101, a closed container 61 as a housing, and a compression mechanism 62 as a compression element.
  • the compressor 6 is a rotary compressor.
  • the compressor 6 is not limited to the rotary compressor.
  • the electric motor 1 described in the first embodiment is applied to the electric motor 60.
  • the electric motor 60 drives the compression mechanism 62.
  • the closed container 61 covers the electric motor 60 and the compression mechanism 62.
  • the closed container 61 is a cylindrical container. Refrigerating machine oil that lubricates the sliding portion of the compression mechanism 62 is stored at the bottom of the closed container 61.
  • the compressor 6 further includes a glass terminal 63 fixed to the closed container 61, an accumulator 64, a suction pipe 65, and a discharge pipe 66.
  • the compression mechanism 62 is attached to a cylinder 62a, a piston 62b, an upper frame 62c (also referred to as a first frame), a lower frame 62d (also referred to as a second frame), an upper frame 62c and a lower frame 62d. And a plurality of mufflers 62e.
  • the compression mechanism 62 further has a vane that divides the inside of the cylinder 62a into a suction side and a compression side.
  • the compression mechanism 62 is driven by the electric motor 60.
  • the electric motor 60 is fixed in the closed container 61 by press fitting or shrink fitting. Instead of press fitting and shrink fitting, the electric motor 60 may be directly attached to the closed container 61 by welding.
  • Electric power is supplied to the coil of the electric motor 60 (for example, the coil 3 described in the first embodiment) from the drive device 101 through the glass terminal 63.
  • the rotor 20 of the electric motor 60 (specifically, one side of the shaft 67) is rotatably supported by bearings provided in each of the upper frame 62c and the lower frame 62d.
  • a shaft 67 is inserted through the piston 62b.
  • a shaft 67 is rotatably inserted through the upper frame 62c and the lower frame 62d.
  • the upper frame 62c and the lower frame 62d close the end surface of the cylinder 62a.
  • the accumulator 64 supplies a refrigerant (for example, refrigerant gas) to the cylinder 62a through the suction pipe 65.
  • the refrigerant supplied from the accumulator 64 is sucked into the cylinder 62a from the suction pipe 65 fixed to the closed container 61.
  • the piston 62b fitted on the shaft 67 rotates in the cylinder 62a.
  • the refrigerant is compressed in the cylinder 62a.
  • the refrigerant passes through the muffler 62e and rises in the closed container 61. In this way, the compressed refrigerant is supplied to the high pressure side of the refrigeration cycle through the discharge pipe 66.
  • refrigerant of the compressor 6 R410A, R407C, R22 or the like can be used.
  • the refrigerant of the compressor 6 is not limited to these types.
  • a refrigerant having a small GWP (global warming potential) for example, the following refrigerant can be used.
  • the GWP of HFO-1234yf is 4.
  • a hydrocarbon having a carbon double bond in the composition for example, R1270 (propylene) may be used.
  • R1270 has a GWP of 3, which is lower than that of HFO-1234yf, but higher than that of HFO-1234yf.
  • HFO-1234yf a mixture containing at least either a halogenated hydrocarbon having a carbon double bond in the composition or a hydrocarbon having a carbon double bond in the composition, for example, a mixture of HFO-1234yf and R32.
  • HFO-1234yf is a low-pressure refrigerant, it tends to have a large pressure loss, which may lead to a reduction in the performance of the refrigeration cycle (especially the evaporator). Therefore, it is practically desirable to use a mixture with R32 or R41, which is a higher pressure refrigerant than HFO-1234yf.
  • the compressor 6 according to the second embodiment has the advantages described in the first embodiment.
  • the compressor 6 according to the second embodiment has the drive device 101, the efficiency of the compressor 6 can be improved.
  • FIG. 21 is a diagram schematically showing the configuration of the refrigerating and air-conditioning apparatus 7 according to the third embodiment.
  • the refrigeration/air-conditioning apparatus 7 includes a compressor 6, a four-way valve 71, a condenser 72, a decompression device 73 (also called an expander), an evaporator 74, a refrigerant pipe 75, and a control unit according to the second embodiment. 76 and.
  • the compressor 6, the condenser 72, the decompression device 73, and the evaporator 74 are connected by a refrigerant pipe 75 to form a refrigeration cycle.
  • the compressor 6 compresses the sucked refrigerant and sends out a high-temperature and high-pressure gas refrigerant.
  • the four-way valve 71 switches the flow direction of the refrigerant. In the example shown in FIG. 21, the four-way valve 71 sends the refrigerant sent from the compressor 6 to the condenser 72.
  • the condenser 72 condenses the refrigerant by exchanging heat between the refrigerant sent from the compressor 6 and air (for example, outdoor air), and sends out the liquefied refrigerant.
  • the decompression device 73 expands the refrigerant (that is, the liquefied refrigerant) sent from the condenser 72, and sends the low-temperature and low-pressure liquefied refrigerant.
  • the evaporator 74 evaporates the refrigerant by exchanging heat between the low-temperature low-pressure liquefied refrigerant sent from the decompression device 73 and air (for example, indoor air), and the evaporated refrigerant (that is, gas). (Refrigerant) is sent out.
  • the air from which heat has been removed by the evaporator 74 is supplied to the target space (for example, the room) by, for example, a blower.
  • the operations of the four-way valve 71 and the compressor 6 are controlled by the controller 76.
  • the constituent elements other than the compressor 6 are not limited to the constituent elements described in the third embodiment.
  • the refrigerating and air-conditioning system 7 according to the third embodiment has the advantages described in the second embodiment.
  • the refrigeration air conditioning system 7 has the compressor 6, the efficiency of the refrigeration air conditioning system 7 can be improved.
  • the drive device 101 described in the first embodiment can be applied to a drive device in a device such as a blower, a ventilation fan, a home electric appliance, or a machine tool, in addition to the compressor 6 and the refrigerating and air conditioning device 7.
  • a device such as a blower, a ventilation fan, a home electric appliance, or a machine tool, in addition to the compressor 6 and the refrigerating and air conditioning device 7.
  • 1,60 electric motors 3,3U, 3V, 3W coils, 6 compressors, 7 refrigeration and air conditioning units, 50 control units, 101 drive units, 102 converters, 103 inverters.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

A drive apparatus (101) drives an electric motor (1) having a coil (3). The drive apparatus (101) has: an inverter (103) that applies voltage to the coil (3); and a control device (50). The control device (50) controls a carrier frequency, and switches the carrier frequency in accordance with the rotation speed of the electric motor (1).

Description

駆動装置、圧縮機、冷凍空調装置、および電動機の駆動方法Drive device, compressor, refrigerating and air-conditioning device, and method of driving electric motor
　本発明は、電動機を駆動する駆動装置に関する。 The present invention relates to a drive device that drives an electric motor.
　一般的に、電動機に電圧を印加するインバータを制御するため、パルス幅変調制御方式が用いられている。パルス幅変調制御方式では、インバータのキャリア周波数が、インバータにおけるスイッチング損失および電動機における高調波鉄損に影響を与えることが知られている。そのため、インバータから出力される電流波形の状態に応じてキャリア周波数を切り替える電動機の駆動方法が提案されている（例えば、特許文献１参照）。 Generally, a pulse width modulation control method is used to control an inverter that applies a voltage to an electric motor. In the pulse width modulation control method, it is known that the carrier frequency of the inverter affects the switching loss in the inverter and the harmonic iron loss in the electric motor. Therefore, a driving method of an electric motor that switches the carrier frequency according to the state of the current waveform output from the inverter has been proposed (for example, refer to Patent Document 1).
特開２００５－１７６４３７号公報JP, 2005-176437, A
　しかしながら、従来の技術では、スイッチング損失および高調波鉄損などの、インバータおよび電動機における総合損失が大きいため、インバータおよび電動機の総合効率が悪いという課題がある。 However, the conventional technology has a problem that the total efficiency of the inverter and the electric motor is poor because the total loss of the inverter and the electric motor, such as switching loss and harmonic iron loss, is large.
　本発明の目的は、インバータおよび電動機の総合効率を改善することである。 The purpose of the present invention is to improve the overall efficiency of the inverter and the electric motor.
　本発明の一態様に係る駆動装置は、
　コイルを有する電動機を駆動する駆動装置であって、
　前記コイルに電圧を印加するインバータと、
　前記コイルに印加される前記電圧の制御周波数を調整するための前記インバータのキャリア周波数を制御し、前記電動機の回転数に応じて前記キャリア周波数を切り替える制御装置と
　を備える。
　本発明の他の態様に係る圧縮機は、前記駆動装置を備える。
　本発明の他の態様に係る冷凍空調装置は、前記圧縮機を備える。
　本発明の他の態様に係る電動機の駆動方法は、
　コイルを有する電動機の駆動方法であって、
　前記コイルに印加される電圧の制御周波数を調整するためのインバータのキャリア周波数を設定し、
　前記電動機の回転数に応じて前記キャリア周波数を切り替える。 A drive device according to one aspect of the present invention is
A drive device for driving an electric motor having a coil,
An inverter for applying a voltage to the coil,
A controller for controlling the carrier frequency of the inverter for adjusting the control frequency of the voltage applied to the coil, and switching the carrier frequency according to the rotation speed of the electric motor.
A compressor according to another aspect of the present invention includes the drive device.
A refrigerating and air-conditioning apparatus according to another aspect of the present invention includes the compressor.
A method of driving an electric motor according to another aspect of the present invention is
A method for driving an electric motor having a coil, comprising:
Setting the carrier frequency of the inverter for adjusting the control frequency of the voltage applied to the coil,
The carrier frequency is switched according to the rotation speed of the electric motor.
　本発明によれば、インバータおよび電動機の総合効率を改善することができる。 According to the present invention, the overall efficiency of the inverter and the electric motor can be improved.
電動機の構造を概略的に示す断面図である。It is sectional drawing which shows the structure of an electric motor schematically. 駆動装置の構成を示すブロック図である。It is a block diagram which shows the structure of a drive device. ＰＷＭ制御信号を生成するためのキャリア波およびインバータ出力電圧指令値の一例を示す図である。It is a figure which shows an example of the carrier wave and inverter output voltage command value for generating a PWM control signal. 制御装置において生成されたＰＷＭ制御信号の一例を示す図である。It is a figure which shows an example of the PWM control signal produced|generated in the control apparatus. ＰＷＭ制御信号に基づいて生成される電動機電流の一例を示す図である。It is a figure which shows an example of the electric motor electric current produced|generated based on a PWM control signal. 実際のインバータ電圧と線間電圧との関係を示す図である。It is a figure which shows the relationship between an actual inverter voltage and line voltage. 低速回転時における電動機電流の波形の例を示す図である。It is a figure which shows the example of the waveform of the electric motor current at the time of low speed rotation. 中速回転時における電動機電流の波形の例を示す図である。It is a figure which shows the example of the waveform of the electric motor electric current at the time of medium speed rotation. 高速回転時における電動機電流の波形の例を示す図である。It is a figure which shows the example of the waveform of the electric motor current at the time of high speed rotation. 電動機の駆動方法の一例を示すフローチャートである。It is a flow chart which shows an example of a drive method of a motor. 電動機の回転数とキャリア周波数との関係の一例を示す図である。It is a figure which shows an example of the relationship between the rotation speed of an electric motor, and carrier frequency. 電動機の回転数と線間電圧との関係を示す図である。It is a figure which shows the relationship between the rotation speed of an electric motor, and the line voltage. インバータのインバータスイッチに対するパルス幅変調制御中におけるインバータスイッチのオン時間の割合とキャリア周波数との関係を示す図である。It is a figure which shows the relationship between the ratio of the ON time of an inverter switch and carrier frequency during pulse width modulation control with respect to the inverter switch of an inverter. コンバータ電圧と線間電圧との関係を示す図である。It is a figure which shows the relationship between a converter voltage and a line voltage. 電動機の回転数とＰＷＭ制御中におけるインバータスイッチのオン時間の割合との関係を示す図である。It is a figure which shows the relationship between the rotation speed of an electric motor, and the ratio of the ON time of the inverter switch during PWM control. コンバータ電圧の昇圧前後におけるキャリア周波数を示す図である。It is a figure which shows the carrier frequency before and behind the boost of a converter voltage. 電動機の回転数とキャリア周波数との関係を示す図である。It is a figure which shows the relationship between the rotation speed of an electric motor, and a carrier frequency. 電動機の回転数とキャリア周波数との関係を示す図である。It is a figure which shows the relationship between the rotation speed of an electric motor, and a carrier frequency. インバータのインバータスイッチに対するパルス幅変調制御中におけるインバータスイッチのオン時間の割合とキャリア周波数との関係を示す図である。It is a figure which shows the relationship between the ratio of the ON time of an inverter switch and carrier frequency during pulse width modulation control with respect to the inverter switch of an inverter. 本発明の実施の形態２に係る圧縮機の構造を概略的に示す断面図である。It is sectional drawing which shows roughly the structure of the compressor which concerns on Embodiment 2 of this invention. 本発明の実施の形態３に係る冷凍空調装置の構成を概略的に示す図である。It is a figure which shows schematically the structure of the refrigerating air-conditioning apparatus which concerns on Embodiment 3 of this invention.
　本発明の実施の形態について説明する。
実施の形態１．
＜電動機１の構造＞
　図１は、電動機１の構造を概略的に示す断面図である。図１では、電動機１のロータ２０の回転軸に直交する面における電動機１の構造が示されている。
　電動機１は、例えば、永久磁石埋込型電動機である。電動機１は、例えばロータリー圧縮機に用いられる。 An embodiment of the present invention will be described.
Embodiment 1.
<Structure of electric motor 1>
FIG. 1 is a sectional view schematically showing the structure of the electric motor 1. FIG. 1 shows the structure of the electric motor 1 in a plane orthogonal to the rotation axis of the rotor 20 of the electric motor 1.
The electric motor 1 is, for example, a permanent magnet embedded electric motor. The electric motor 1 is used, for example, in a rotary compressor.
　電動機１は、ステータ１０と、ステータ１０の内側に回転可能に設けられたロータ２０とを有する。ステータ１０の内周面とロータ２０の外周面との間には、エアギャップが存在する。そのエアギャップの幅は、例えば、０．３ｍｍから１ｍｍである。 The electric motor 1 has a stator 10 and a rotor 20 rotatably provided inside the stator 10. An air gap exists between the inner peripheral surface of the stator 10 and the outer peripheral surface of the rotor 20. The width of the air gap is, for example, 0.3 mm to 1 mm.
　ロータ２０の軸方向（すなわち、回転軸Ｃ１の方向）を、単に「軸方向」と称する。ステータ１０またはロータ２０の外周に沿った方向を、単に「周方向」と称する。ステータ１０またはロータ２０の半径方向を、単に「径方向」と称する。 The axial direction of the rotor 20 (that is, the direction of the rotation axis C1) is simply referred to as the “axial direction”. The direction along the outer circumference of the stator 10 or the rotor 20 is simply referred to as “circumferential direction”. The radial direction of the stator 10 or the rotor 20 is simply referred to as “radial direction”.
　ステータ１０は、ステータコア１１と、少なくとも１つのコイル（図１では、３つのコイル３Ｕ，３つのコイル３Ｖ，および３つのコイル３Ｗ）と、少なくとも１つの絶縁体１４とを有する。 The stator 10 includes a stator core 11, at least one coil (three coils 3U, three coils 3V, and three coils 3W in FIG. 1) and at least one insulator 14.
　ステータコア１１は、例えば、複数の電磁鋼板で作られている。この場合、複数の電磁鋼板は、軸方向に積層されている。複数の電磁鋼板は、カシメで固定されている。各電磁鋼板の厚さは、例えば、０．１ｍｍから０．７ｍｍである。本実施の形態では、各電磁鋼板の厚さは、０．３５ｍｍである。各電磁鋼板は、打ち抜き処理などのプレス加工で予め定められた形状に形成されている。 The stator core 11 is made of, for example, a plurality of electromagnetic steel plates. In this case, the plurality of magnetic steel sheets are laminated in the axial direction. The plurality of magnetic steel sheets are fixed by caulking. The thickness of each electromagnetic steel sheet is, for example, 0.1 mm to 0.7 mm. In the present embodiment, the thickness of each electromagnetic steel plate is 0.35 mm. Each electromagnetic steel sheet is formed into a predetermined shape by press working such as punching.
　ステータコア１１は、環状のヨーク部１３と、複数のティース部１２とを有している。ヨーク部１３は、周方向に延在する。各ティース部１２は、径方向に延在する。言い換えると、各ティース部１２は、ヨーク部１３からロータ２０の回転軸Ｃ１に向けて突出している。複数のティース部１２は、周方向に等間隔で配置されており、放射状に延在する。 The stator core 11 has an annular yoke portion 13 and a plurality of teeth portions 12. The yoke portion 13 extends in the circumferential direction. Each tooth part 12 extends in the radial direction. In other words, each tooth portion 12 projects from the yoke portion 13 toward the rotation axis C1 of the rotor 20. The plurality of tooth portions 12 are arranged at equal intervals in the circumferential direction and extend radially.
　図１に示される例では、ステータコア１１は、９個のティース部１２を有する。隣り合う２つのティース部１２の間には、スロットが存在する。各ティース部１２は、径方向に延在する本体部と、周方向に延在する歯先部とを有する。各歯先部は、各ティース部１２の先端部に位置する。 In the example shown in FIG. 1, the stator core 11 has nine teeth portions 12. A slot exists between two adjacent tooth portions 12. Each tooth portion 12 has a main body portion that extends in the radial direction and a tooth tip portion that extends in the circumferential direction. Each tooth tip is located at the tip of each tooth 12.
　コイル３Ｕ，３Ｖ，および３Ｗの各々は、ステータコア１１に固定された絶縁体１４の周りに巻き付けられている。コイル３Ｕ，３Ｖ，および３Ｗは、少なくとも１つのステータ巻線で作られている。すなわち、ステータコア１１に、少なくとも１つのステータ巻線が巻き付けられている。少なくとも１つのステータ巻線は、例えば、マグネットワイヤである。本実施の形態では、コイル３Ｕ，３Ｖ，および３Ｗの各々が、集中巻で絶縁体１４の周りに１１０ターン巻き付けられている。 Each of the coils 3U, 3V, and 3W is wound around the insulator 14 fixed to the stator core 11. Coils 3U, 3V and 3W are made of at least one stator winding. That is, at least one stator winding is wound around the stator core 11. The at least one stator winding is, for example, a magnet wire. In the present embodiment, each of coils 3U, 3V, and 3W is wound 110 turns around insulator 14 by concentrated winding.
　コイル３Ｕ，３Ｖ，および３Ｗの各々の巻き数および線径は、電動機１に要求される特性（例えば、回転数、トルク）、供給電圧、またはスロットの断面積に応じて決定される。 The number of turns and the wire diameter of each of the coils 3U, 3V, and 3W are determined according to the characteristics (for example, rotation speed, torque) required for the electric motor 1, the supply voltage, or the cross-sectional area of the slot.
　ステータコア１１は、例えば、複数のブロック（分割コアとも称する）で形成されていてもよい。この場合、これらの複数のブロックを直線的に配列した状態で、各ティース部１２にマグネットワイヤを巻き付けることができる。ティース部１２にマグネットワイヤを巻き付けた後、複数のブロックを環状に折り畳み、複数のブロックの両端を溶接する。 The stator core 11 may be formed of, for example, a plurality of blocks (also referred to as split cores). In this case, the magnet wire can be wound around each tooth portion 12 in a state where these plural blocks are linearly arranged. After winding the magnet wire around the tooth portion 12, the plurality of blocks are folded in an annular shape, and both ends of the plurality of blocks are welded.
　コイルは、Ｕ相、Ｖ相およびＷ相の３相コイル（すなわち、コイル３Ｕ，３Ｖ，および３Ｗ）で構成されている。コイル３Ｕは、Ｕ相のコイル部分Ｕａ，Ｕｂ，およびＵｃを有する。コイル３Ｖは、Ｖ相のコイル部分Ｖａ，Ｖｂ，およびＶｃを有する。コイル３Ｗは、Ｗ相のコイル部分Ｗａ，Ｗｂ，およびＷｃを有する。各相のコイルの両端子は開放されている。すなわち、コイルは、合計６つの端子を有している。絶縁体１４は、例えば、ポリエチレンテレフタレート（ＰＥＴ）で作られたフィルムで構成されている。 The coil is composed of U-phase, V-phase, and W-phase three-phase coils (that is, coils 3U, 3V, and 3W). The coil 3U has U-phase coil portions Ua, Ub, and Uc. The coil 3V has V-phase coil portions Va, Vb, and Vc. Coil 3W has W-phase coil portions Wa, Wb, and Wc. Both terminals of the coil of each phase are open. That is, the coil has a total of six terminals. The insulator 14 is made of, for example, a film made of polyethylene terephthalate (PET).
　薄いフィルムで作られた絶縁体１４は、スロット内のコイルの巻き数の増加に有効である。さらに、複数のブロックで作られたステータコア１１は、スロット内のコイルの巻き数の増加に有効である。ただし、ステータコア１１は、複数のブロックで作られた構造には限定されない。 The insulator 14 made of a thin film is effective in increasing the number of turns of the coil in the slot. Furthermore, the stator core 11 made of a plurality of blocks is effective in increasing the number of turns of the coil in the slot. However, the stator core 11 is not limited to the structure made of a plurality of blocks.
　ロータ２０は、ロータコア２１と、ロータコア２１に取り付けられた複数の永久磁石２５とを有する。 The rotor 20 has a rotor core 21 and a plurality of permanent magnets 25 attached to the rotor core 21.
　ロータコア２１は、例えば、複数の電磁鋼板で作られている。この場合、複数の電磁鋼板は、軸方向に積層されている。複数の電磁鋼板は、カシメで固定されている。各電磁鋼板の厚さは、例えば、０．１ｍｍから０．７ｍｍである。本実施の形態では、各電磁鋼板の厚さは、０．３５ｍｍである。各電磁鋼板は、打ち抜き処理などのプレス加工で予め定められた形状に形成されている。 The rotor core 21 is made of, for example, a plurality of electromagnetic steel plates. In this case, the plurality of magnetic steel sheets are laminated in the axial direction. The plurality of magnetic steel sheets are fixed by caulking. The thickness of each electromagnetic steel sheet is, for example, 0.1 mm to 0.7 mm. In the present embodiment, the thickness of each electromagnetic steel plate is 0.35 mm. Each electromagnetic steel sheet is formed into a predetermined shape by press working such as punching.
　ロータコア２１は、複数の磁石挿入孔２２と、シャフト孔２７とを有する。シャフト孔２７には、シャフトが挿入される。シャフトは、焼き嵌め、圧入などの方法で、ロータコア２１に固定される。 The rotor core 21 has a plurality of magnet insertion holes 22 and a shaft hole 27. A shaft is inserted into the shaft hole 27. The shaft is fixed to the rotor core 21 by a method such as shrink fitting and press fitting.
　本実施の形態では、ロータコア２１は、６個の磁石挿入孔２２を有する。軸方向に直交する面において、複数の磁石挿入孔２２は、周方向に配列されている。各磁石挿入孔２２は、ロータ２０の各磁極に対応する。ロータ２０の磁極数は、２以上である。本実施の形態では、ロータ２０の磁極数は、６極である。各磁石挿入孔には、１以上の永久磁石２５が配置されている。 In this embodiment, the rotor core 21 has six magnet insertion holes 22. On the surface orthogonal to the axial direction, the plurality of magnet insertion holes 22 are arranged in the circumferential direction. Each magnet insertion hole 22 corresponds to each magnetic pole of the rotor 20. The number of magnetic poles of the rotor 20 is 2 or more. In the present embodiment, the rotor 20 has six magnetic poles. One or more permanent magnets 25 are arranged in each magnet insertion hole.
　軸方向に直交する面において、磁石挿入孔２２の中央部は、回転軸Ｃ１に向けて突出している。すなわち、軸方向に直交する面において、各磁石挿入孔２２はＶ字形状を有している。各磁石挿入孔２２の形状は、Ｖ字形状に限定されるものではなく、例えばストレート形状であってもよい。 On the plane orthogonal to the axial direction, the central portion of the magnet insertion hole 22 projects toward the rotation axis C1. That is, each magnet insertion hole 22 has a V shape in the plane orthogonal to the axial direction. The shape of each magnet insertion hole 22 is not limited to the V shape, and may be, for example, a straight shape.
　本実施の形態では、１つの磁石挿入孔２２内には、２つの永久磁石２５が配置される。すなわち、１磁極について２つの永久磁石２５が配置される。したがって、本実施の形態では、ロータ２０は、１２個の永久磁石２５を有する。 In the present embodiment, two permanent magnets 25 are arranged in one magnet insertion hole 22. That is, two permanent magnets 25 are arranged for one magnetic pole. Therefore, in this embodiment, the rotor 20 has twelve permanent magnets 25.
　各永久磁石２５は、軸方向に長い平板状の磁石である。各永久磁石２５は、例えば、ネオジウム（Ｎｄ）、鉄（Ｆｅ）、およびボロン（Ｂ）を含む希土類磁石である。１つの磁石挿入孔２２内に配置された、２つの永久磁石２５は、ロータ２０の１つの磁極の役目をする。 Each permanent magnet 25 is a flat magnet that is long in the axial direction. Each permanent magnet 25 is, for example, a rare earth magnet containing neodymium (Nd), iron (Fe), and boron (B). Two permanent magnets 25 arranged in one magnet insertion hole 22 serve as one magnetic pole of the rotor 20.
　ロータコア２１は、複数のフラックスバリア２６をさらに有する。各磁石挿入孔２２の周方向両側に、フラックスバリア２６が位置する。各フラックスバリア２６は、磁石挿入孔２２に連通している空隙である。各フラックスバリア２６は、隣り合う磁極間の漏れ磁束（すなわち、極間を通る磁束）を低減する。軸方向に直交する面において、各フラックスバリア２６とロータコア２１の外周面との間の薄肉部の幅は、例えば、ロータコア２１の電磁鋼板の厚さと同じである。 The rotor core 21 further has a plurality of flux barriers 26. Flux barriers 26 are located on both sides of each magnet insertion hole 22 in the circumferential direction. Each flux barrier 26 is a void communicating with the magnet insertion hole 22. Each flux barrier 26 reduces the leakage magnetic flux between adjacent magnetic poles (that is, the magnetic flux passing between the magnetic poles). The width of the thin portion between each flux barrier 26 and the outer peripheral surface of the rotor core 21 on the surface orthogonal to the axial direction is, for example, the same as the thickness of the electromagnetic steel sheet of the rotor core 21.
　ロータコア２１において、各磁石挿入孔２２の周方向の中央部には、突起である第１の磁石保持部２３が形成されている。さらに、ロータコア２１において、各磁石挿入孔２２の周方向の両端部には、突起である第２の磁石保持部２４が形成されている。第１の磁石保持部２３および第２の磁石保持部２４は、各磁石挿入孔２２内において永久磁石２５を保持する。 In the rotor core 21, each magnet insertion hole 22 has a first magnet holding portion 23, which is a protrusion, formed in the central portion in the circumferential direction. Further, in the rotor core 21, second magnet holding portions 24, which are protrusions, are formed at both ends in the circumferential direction of each magnet insertion hole 22. The first magnet holding portion 23 and the second magnet holding portion 24 hold the permanent magnet 25 in each magnet insertion hole 22.
＜駆動装置１０１の構成＞
　次に、コイル３を有する電動機１を駆動する駆動装置１０１について説明する。
　図２は、駆動装置１０１の構成を示すブロック図である。
　駆動装置１０１は、電源の出力を整流するコンバータ１０２と、電動機１のコイル３に電圧（具体的には、交流電圧）を印加するインバータ１０３と、制御装置５０とを有する。コイル３Ｕ，３Ｖ，および３Ｗの全体を、コイル３と称する。 <Structure of drive device 101>
Next, the drive device 101 for driving the electric motor 1 having the coil 3 will be described.
FIG. 2 is a block diagram showing the configuration of the driving device 101.
The drive device 101 includes a converter 102 that rectifies an output of a power supply, an inverter 103 that applies a voltage (specifically, an AC voltage) to the coil 3 of the electric motor 1, and a control device 50. The coils 3U, 3V, and 3W are collectively referred to as a coil 3.
　コンバータ１０２には、交流（ＡＣ）電源である電源から電力が供給される。コンバータ１０２は、インバータ１０３に電圧を印加する。コンバータ１０２からインバータ１０３に印加される電圧を「コンバータ電圧」とも称する。コンバータ１０２の母線電圧は制御装置５０に供給される。 Electric power is supplied to the converter 102 from a power supply which is an alternating current (AC) power supply. The converter 102 applies a voltage to the inverter 103. The voltage applied from the converter 102 to the inverter 103 is also referred to as “converter voltage”. The bus voltage of converter 102 is supplied to control device 50.
　電動機１を駆動するインバータ電圧、すなわち、電動機１のコイル３に印加される電圧は、ＰＷＭ制御方式で生成される。上述のように、電動機１のコイル３は、例えば、３相コイルである。この場合、インバータ１０３は、各相に対応する少なくとも１つのインバータスイッチを有し、各インバータスイッチは、１組のスイッチング素子（本実施の形態では、２個のスイッチング素子）を有する。 The inverter voltage that drives the electric motor 1, that is, the voltage applied to the coil 3 of the electric motor 1 is generated by the PWM control method. As described above, the coil 3 of the electric motor 1 is, for example, a three-phase coil. In this case, the inverter 103 has at least one inverter switch corresponding to each phase, and each inverter switch has one set of switching elements (two switching elements in the present embodiment).
　ＰＷＭ制御方式では、各相に対応するインバータスイッチのオンオフの時間割合を制御することでインバータ電圧の波形を生成する。これにより、インバータ１０３からの所望の出力波形を得ることができる。具体的には、インバータ１０３においてインバータスイッチがオンのとき、インバータ１０３からコイル３へ電圧が供給され、インバータ電圧が増大する。インバータスイッチがオフのとき、インバータ１０３からコイル３への電圧供給は遮断され、インバータ電圧が降下する。インバータ電圧と誘起電圧との差分がコイル３に供給され、電動機電流が発生し、電動機１の回転力が生じる。目標とする電動機電流値に一致するようインバータスイッチのオンオフの時間割合を制御することで、インバータ１０３からの所望の出力波形を得ることができる。 In the PWM control method, the waveform of the inverter voltage is generated by controlling the on/off time ratio of the inverter switch corresponding to each phase. Thereby, a desired output waveform from the inverter 103 can be obtained. Specifically, when the inverter switch in the inverter 103 is turned on, a voltage is supplied from the inverter 103 to the coil 3 and the inverter voltage increases. When the inverter switch is off, the voltage supply from the inverter 103 to the coil 3 is cut off and the inverter voltage drops. The difference between the inverter voltage and the induced voltage is supplied to the coil 3, the electric motor current is generated, and the rotational force of the electric motor 1 is generated. A desired output waveform from the inverter 103 can be obtained by controlling the on/off time ratio of the inverter switch so as to match the target electric motor current value.
　図３は、ＰＷＭ制御信号を生成するためのキャリア波およびインバータ出力電圧指令値の一例を示す図である。
　図４は、制御装置５０において生成されたＰＷＭ制御信号の一例を示す図である。
　図５は、ＰＷＭ制御信号に基づいて生成される電動機電流の一例を示す図である。 FIG. 3 is a diagram showing an example of a carrier wave and an inverter output voltage command value for generating a PWM control signal.
FIG. 4 is a diagram showing an example of the PWM control signal generated in the control device 50.
FIG. 5 is a diagram showing an example of the electric motor current generated based on the PWM control signal.
　各インバータスイッチのオンオフのタイミングは、キャリア波に基づいて決定される。キャリア波は、一定の振幅を持つ三角波で構成される。ＰＷＭ制御方式におけるパルス幅変調周期は、キャリア波の周波数であるキャリア周波数によって決まる。本実施の形態では、予め定められたキャリア波のパターンまたは予め定められたキャリア周波数が制御装置５０に格納されている。制御装置５０は、キャリア周波数を制御し、各インバータスイッチのオンオフを制御する。これにより、制御装置５０は、コイル３に供給されるインバータ１０３からの出力を制御する。 -The on/off timing of each inverter switch is determined based on the carrier wave. The carrier wave is composed of a triangular wave having a constant amplitude. The pulse width modulation period in the PWM control method is determined by the carrier frequency which is the frequency of the carrier wave. In the present embodiment, a predetermined carrier wave pattern or a predetermined carrier frequency is stored in control device 50. The control device 50 controls the carrier frequency and controls on/off of each inverter switch. As a result, the control device 50 controls the output from the inverter 103 supplied to the coil 3.
　キャリア波の周波数であるキャリア周波数を「インバータ１０３のキャリア周波数」とも称する。すなわち、インバータ１０３のキャリア周波数は、コイル３に印加される電圧の制御周波数であり、制御装置５０は、インバータ１０３のキャリア周波数を制御する。 The carrier frequency, which is the frequency of the carrier wave, is also referred to as the “carrier frequency of the inverter 103”. That is, the carrier frequency of the inverter 103 is the control frequency of the voltage applied to the coil 3, and the control device 50 controls the carrier frequency of the inverter 103.
　本実施の形態では、インバータ１０３は、３個のインバータスイッチ（すなわち、６個のスイッチング素子）を有するが、３個のインバータスイッチの内の１個のインバータスイッチ、すなわち、Ｕ相、Ｖ相、またはＷ相に対応する１個のインバータスイッチに対する制御について説明する。ただし、その１個のインバータスイッチに対する制御は、他の２個のインバータスイッチに対する制御にも適用可能である。 In the present embodiment, the inverter 103 has three inverter switches (that is, six switching elements), but one of the three inverter switches, that is, the U phase, the V phase, Alternatively, control for one inverter switch corresponding to the W phase will be described. However, the control for the one inverter switch can be applied to the control for the other two inverter switches.
　制御装置５０は、キャリア波の電圧値と、インバータ出力電圧指令値とを比較する。インバータ出力電圧指令値は、例えば、制御装置５０において、目標電動機電流値に基づいて計算される。インバータ出力電圧指令値は、例えば、空気調和機などの冷凍空調装置のリモコンから制御装置５０に入力された信号に基づいて設定される。この場合、インバータ出力電圧指令値は、例えば、冷凍空調装置のユーザによって設定された冷凍空調装置からの風量に対応する。 The control device 50 compares the voltage value of the carrier wave with the inverter output voltage command value. The inverter output voltage command value is calculated, for example, in the control device 50 based on the target electric motor current value. The inverter output voltage command value is set, for example, based on a signal input to the control device 50 from a remote controller of a refrigerating and air conditioning device such as an air conditioner. In this case, the inverter output voltage command value corresponds to, for example, the air volume from the refrigeration air conditioning device set by the user of the refrigeration air conditioning device.
　キャリア波の電圧値がインバータ出力電圧指令値よりも小さいとき、制御装置５０は、インバータスイッチがオンになるようにＰＷＭ制御信号をオンにする。キャリア波の電圧値がインバータ出力電圧指令値以上であるとき、制御装置５０は、インバータスイッチがオフになるようにＰＷＭ制御信号をオフにする。これにより、インバータ電圧が目標値に近づく。 When the voltage value of the carrier wave is smaller than the inverter output voltage command value, the control device 50 turns on the PWM control signal so that the inverter switch turns on. When the voltage value of the carrier wave is equal to or higher than the inverter output voltage command value, control device 50 turns off the PWM control signal so that the inverter switch turns off. As a result, the inverter voltage approaches the target value.
　上述のように、制御装置５０は、インバータ出力電圧指令値とキャリア波の電圧値との差に基づいてＰＷＭ制御信号を生成する。 As described above, the control device 50 generates the PWM control signal based on the difference between the inverter output voltage command value and the carrier wave voltage value.
　制御装置５０は、ＰＷＭ制御信号に基づくインバータ駆動信号などの制御信号をインバータ１０３に出力し、インバータスイッチのオンオフ制御を行う。インバータ駆動信号は、ＰＷＭ制御信号と同じ信号でもよく、ＰＷＭ制御信号と異なる信号でもよい。 The control device 50 outputs a control signal such as an inverter drive signal based on the PWM control signal to the inverter 103 to control ON/OFF of the inverter switch. The inverter drive signal may be the same signal as the PWM control signal or a signal different from the PWM control signal.
　インバータスイッチがオンのときにインバータ電圧がインバータ１０３から出力される。インバータ電圧はコイル３に供給され、電動機１において電動機電流（具体的には、Ｕ相電流、Ｖ相電流、およびＷ相電流）が発生する。これにより、インバータ電圧は電動機１（具体的には、ロータ２０）の回転力に変換される。電動機電流は、電流センサなどの計測器で計測され、制御装置５０に計測結果（例えば、電流値を示す信号）が送信される。 The inverter voltage is output from the inverter 103 when the inverter switch is on. The inverter voltage is supplied to the coil 3, and a motor current (specifically, U-phase current, V-phase current, and W-phase current) is generated in the motor 1. As a result, the inverter voltage is converted into the rotational force of the electric motor 1 (specifically, the rotor 20). The electric motor current is measured by a measuring device such as a current sensor, and the measurement result (for example, a signal indicating a current value) is transmitted to the control device 50.
　制御装置５０は、例えば、プロセッサおよびメモリで構成される。例えば、制御装置５０は、マイクロコンピュータである。制御装置５０は、単一回路または複合回路などの専用のハードウェアとしての処理回路で構成されてもよい。 The control device 50 is composed of, for example, a processor and a memory. For example, the control device 50 is a microcomputer. The control device 50 may be configured with a processing circuit as dedicated hardware such as a single circuit or a composite circuit.
　図６は、実際のインバータ電圧と線間電圧との関係を示す図である。
　図７は、低速回転時における電動機電流の波形の例を示す図である。
　図８は、中速回転時における電動機電流の波形の例を示す図である。
　図９は、高速回転時における電動機電流の波形の例を示す図である。 FIG. 6 is a diagram showing the relationship between the actual inverter voltage and the line voltage.
FIG. 7 is a diagram showing an example of the waveform of the electric motor current during low speed rotation.
FIG. 8 is a diagram showing an example of the waveform of the electric motor current during rotation at a medium speed.
FIG. 9 is a diagram showing an example of the waveform of the electric motor current during high speed rotation.
　一般に、永久磁石埋込型電動機などの電動機では、永久磁石とコイル（例えば、３相コイル）との間の電磁誘導によって、コイルに誘起電圧が発生する。電動機の回転数（すなわち、電動機の回転速度）が高いほど誘起電圧が増大し、線間電圧も増大する。インバータ電圧と線間電圧との差分が電動機の回転力に変換される。瞬間的なインバータ出力電圧（すなわち、実際のインバータ電圧）は、インバータスイッチのオンオフで制御されているため、ゼロまたは母線電圧に等しい。電動機の回転数が低いほど、すなわち線間電圧が小さいほど、インバータスイッチがオンである時の実際のインバータ電圧と線間電圧との乖離が大きくなり、図７に示されるように、電動機電流の高調波成分が発生する。これにより、高調波鉄損が発生する。したがって、電動機が低速回転であるとき、電動機における全損失に占める高調波鉄損の割合が大きく、電動機が高速回転であるとき、高調波鉄損の割合が小さい。 Generally, in an electric motor such as a permanent magnet embedded type electric motor, an induced voltage is generated in the coil by electromagnetic induction between the permanent magnet and the coil (for example, three-phase coil). The higher the rotational speed of the electric motor (that is, the rotational speed of the electric motor), the larger the induced voltage and the line voltage. The difference between the inverter voltage and the line voltage is converted into the rotational force of the electric motor. The instantaneous inverter output voltage (ie, the actual inverter voltage) is equal to zero or the bus voltage because it is controlled by turning on and off the inverter switch. The lower the rotational speed of the electric motor, that is, the smaller the line voltage, the larger the deviation between the actual inverter voltage and the line voltage when the inverter switch is on, and as shown in FIG. Harmonic components are generated. This causes harmonic iron loss. Therefore, when the electric motor is rotating at low speed, the proportion of the harmonic iron loss in the total loss in the electric motor is large, and when the electric motor is rotating at high speed, the proportion of the harmonic iron loss is small.
　キャリア周波数が高いほどＰＷＭ制御信号の周期は短いため、電動機電流は所望の値から乖離せず、波形の形状が望ましい形状に近づく。この場合、すなわち、電動機電流の高周波成分が抑制される。その結果、電動機電流の高周波成分に起因する高調波鉄損が低減する。一方、インバータのスイッチング回数が多くなるにつれて、インバータのスイッチング損失が増大する。高調波鉄損とスイッチング損失とのバランスから効率最適となるキャリア周波数が決まり、高調波鉄損の割合が小さいほど効率最適となるキャリア周波数は小さくなる。 The higher the carrier frequency, the shorter the cycle of the PWM control signal, so the motor current does not deviate from the desired value, and the waveform shape approaches the desired shape. In this case, that is, the high frequency component of the electric motor current is suppressed. As a result, the harmonic iron loss due to the high frequency component of the electric motor current is reduced. On the other hand, as the number of switching times of the inverter increases, the switching loss of the inverter increases. The carrier frequency that optimizes the efficiency is determined from the balance between the harmonic iron loss and the switching loss, and the smaller the proportion of the harmonic iron loss, the smaller the carrier frequency that optimizes the efficiency.
　本実施の形態では、制御装置５０は、電動機１の回転数に応じてキャリア周波数を切り替える。電動機１の回転数とは、電動機１の回転速度を意味する。すなわち、制御装置５０は、電動機１の回転速度に応じてキャリア周波数を切り替える。 In the present embodiment, the control device 50 switches the carrier frequency according to the rotation speed of the electric motor 1. The rotation speed of the electric motor 1 means the rotation speed of the electric motor 1. That is, the control device 50 switches the carrier frequency according to the rotation speed of the electric motor 1.
　図１０は、電動機１の駆動方法の一例を示すフローチャートである。
　図１１は、電動機１の回転数とキャリア周波数との関係の一例を示す図である。 FIG. 10 is a flowchart showing an example of the driving method of the electric motor 1.
FIG. 11 is a diagram showing an example of the relationship between the rotation speed of the electric motor 1 and the carrier frequency.
　最初に、制御装置５０は、キャリア周波数を第１の周波数としての周波数ｆ１に設定する（ステップＳＴ１）。 First, the control device 50 sets the carrier frequency to the frequency f1 as the first frequency (step ST1).
　制御装置５０は、電動機１の回転数に応じてキャリア周波数を切り替える。具体的には、制御装置５０は、電動機１の回転数が閾値Ｎｄに到達したか判定する（ステップＳＴ２）。電動機１の回転数が閾値Ｎｄ以上であるとき（ステップＳＴ２においてＹｅｓ）、制御装置５０は、キャリア周波数を周波数ｆ１よりも小さい第２の周波数としての周波数ｆ２に設定する（ステップＳＴ３）。電動機１の回転数が閾値Ｎｄに到達していないとき（ステップＳＴ２においてＮｏ）、制御装置５０は、キャリア周波数を切り替えない。 The control device 50 switches the carrier frequency according to the rotation speed of the electric motor 1. Specifically, control device 50 determines whether the rotation speed of electric motor 1 has reached threshold value Nd (step ST2). When the rotation speed of the electric motor 1 is equal to or higher than the threshold value Nd (Yes in step ST2), the control device 50 sets the carrier frequency to the frequency f2 as the second frequency smaller than the frequency f1 (step ST3). When the rotation speed of the electric motor 1 has not reached the threshold value Nd (No in step ST2), the control device 50 does not switch the carrier frequency.
　すなわち、電動機１の回転数が閾値Ｎｄよりも小さいとき、キャリア周波数は第１の周波数としての周波数ｆ１であり、電動機１の回転数が閾値Ｎｄ以上であるとき、キャリア周波数は第１の周波数よりも小さい第２の周波数としての周波数ｆ２である。この場合、例えばｆ１＝９０００Ｈｚであり、例えばｆ２＝４５００Ｈｚである。すなわち、電動機１の回転数が閾値Ｎｄよりも小さいとき、制御装置５０は、キャリア周波数をｆ１に設定し、電動機１の回転数が閾値Ｎｄ以上であるとき、制御装置５０は、キャリア周波数をｆ２に設定する。 That is, when the rotation speed of the electric motor 1 is lower than the threshold value Nd, the carrier frequency is the frequency f1 as the first frequency, and when the rotation speed of the electric motor 1 is equal to or higher than the threshold value Nd, the carrier frequency is higher than the first frequency. Is the frequency f2 as the second frequency which is also smaller. In this case, for example, f1=9000 Hz and, for example, f2=4500 Hz. That is, when the rotation speed of the electric motor 1 is lower than the threshold value Nd, the control device 50 sets the carrier frequency to f1, and when the rotation speed of the electric motor 1 is equal to or higher than the threshold value Nd, the control device 50 sets the carrier frequency to f2. Set to.
　ただし、周波数ｆ２は、最小キャリア周波数ｆｍ以上である。最小キャリア周波数ｆｍは、電動機１を制御するために必要な電動機電流の波形を生成可能な最小値である。 However, the frequency f2 is equal to or higher than the minimum carrier frequency fm. The minimum carrier frequency fm is the minimum value that can generate the waveform of the electric motor current required to control the electric motor 1.
　図１２は、電動機１の回転数と線間電圧との関係を示す図である。
　図１２に示されるように、線間電圧は、電動機１の回転数の上昇に従って高くなり、インバータ１０３の最大出力で飽和する。そのため、電動機１の回転数をさらに高めるために、弱め界磁制御が開始される。通常、弱め界磁制御では、コイルの抵抗に起因する銅損が増加する。その結果、弱め界磁制御では、電動機の効率が低下する傾向がある。 FIG. 12 is a diagram showing the relationship between the rotation speed of the electric motor 1 and the line voltage.
As shown in FIG. 12, the line voltage increases as the rotation speed of the electric motor 1 increases and saturates at the maximum output of the inverter 103. Therefore, field weakening control is started in order to further increase the rotation speed of the electric motor 1. Usually, in field weakening control, copper loss due to coil resistance increases. As a result, the field weakening control tends to reduce the efficiency of the electric motor.
　図１２に示されるように、電動機１が弱め界磁制御で駆動されている間、線間電圧は、コンバータ電圧に対して飽和している。弱め界磁制御では、インバータ電圧と線間電圧との差分が小さいため、電動機電流の高周波成分が発生しにくい。そのため、弱め界磁制御では、電動機１に生じる鉄損に対する高調波鉄損の割合が特に小さい。 As shown in FIG. 12, the line voltage is saturated with respect to the converter voltage while the electric motor 1 is driven by the field weakening control. In the field weakening control, since the difference between the inverter voltage and the line voltage is small, the high frequency component of the motor current is unlikely to occur. Therefore, in the field weakening control, the ratio of the harmonic iron loss to the iron loss generated in the electric motor 1 is particularly small.
　したがって、制御装置５０が電動機１に対する弱め界磁制御を開始したとき、制御装置５０は、弱め界磁制御を開始する前よりもキャリア周波数を下げる。図１２に示される例では、電動機１の回転数が閾値Ｎｄに到達したとき、制御装置５０は、弱め界磁制御を開始し、キャリア周波数を下げる。これにより、弱め界磁制御において、スイッチング損失が低減され、インバータ１０３および電動機１の総合効率を改善することができる。 Therefore, when the control device 50 starts the field weakening control for the electric motor 1, the control device 50 lowers the carrier frequency more than before the field weakening control is started. In the example shown in FIG. 12, when the rotation speed of the electric motor 1 reaches the threshold value Nd, the control device 50 starts field weakening control and lowers the carrier frequency. Thereby, in the field weakening control, the switching loss is reduced, and the overall efficiency of the inverter 103 and the electric motor 1 can be improved.
　図１３は、インバータ１０３のインバータスイッチに対するパルス幅変調制御中におけるインバータスイッチのオン時間の割合とキャリア周波数との関係を示す図である。インバータ１０３のインバータスイッチに対するパルス幅変調制御を「ＰＷＭ制御」とも称する。 FIG. 13 is a diagram showing the relationship between the ratio of the on time of the inverter switch during the pulse width modulation control of the inverter switch of the inverter 103 and the carrier frequency. The pulse width modulation control for the inverter switch of the inverter 103 is also referred to as “PWM control”.
　電動機１の回転が高速になるほどＰＷＭ制御中におけるインバータスイッチのオン時間の割合が増えるので、制御装置５０は、ＰＷＭ制御中におけるインバータスイッチのオン時間の割合に従ってキャリア周波数を設定してもよい。例えば、ＰＷＭ制御中におけるインバータスイッチのオン時間の割合が閾値Ｎｔよりも小さいとき、制御装置５０は、キャリア周波数を第１の周波数に設定し、インバータスイッチのオン時間の割合が閾値Ｎｔ以上であるとき、制御装置５０は、キャリア周波数を第１の周波数よりも小さい第２の周波数に設定する。 The higher the rotation speed of the electric motor 1, the greater the proportion of the ON time of the inverter switch during PWM control. Therefore, the control device 50 may set the carrier frequency according to the proportion of the ON time of the inverter switch during PWM control. For example, when the ratio of the on time of the inverter switch during PWM control is smaller than the threshold value Nt, the control device 50 sets the carrier frequency to the first frequency and the ratio of the on time of the inverter switch is equal to or greater than the threshold value Nt. At this time, the control device 50 sets the carrier frequency to the second frequency which is lower than the first frequency.
　図１３に示される例では、電動機１の回転数が閾値Ｎｄに到達したとき、インバータスイッチのオン時間の割合が閾値Ｎｔに到達する。この場合、ＰＷＭ制御中におけるインバータスイッチのオン時間の割合が閾値Ｎｔよりも小さいとき、キャリア周波数は第１の周波数としての周波数ｆ１である。一方、インバータ１０３のインバータスイッチのオン時間の割合が閾値Ｎｔ以上であるとき、キャリア周波数は周波数ｆ１よりも小さい第２の周波数としての周波数ｆ２である。この場合、周波数ｆ１は、例えば、９０００Ｈｚであり、周波数ｆ２は、例えば、４５００Ｈｚである。 In the example shown in FIG. 13, when the rotation speed of the electric motor 1 reaches the threshold value Nd, the ratio of the on time of the inverter switch reaches the threshold value Nt. In this case, when the ratio of the on time of the inverter switch during PWM control is smaller than the threshold value Nt, the carrier frequency is the frequency f1 as the first frequency. On the other hand, when the ON time ratio of the inverter switch of the inverter 103 is equal to or higher than the threshold value Nt, the carrier frequency is the frequency f2 as the second frequency smaller than the frequency f1. In this case, the frequency f1 is, for example, 9000 Hz, and the frequency f2 is, for example, 4500 Hz.
　図１３に示される例では、閾値Ｎｔは、１／√２である。ただし、閾値Ｎｔは、１／√２以上でもよい。 In the example shown in FIG. 13, the threshold value Nt is 1/√2. However, the threshold value Nt may be 1/√2 or more.
　本実施の形態では、インバータ１０３のインバータスイッチのオン時間の割合が１／√２以上のとき、電動機１は、弱め界磁制御で駆動される。上述のように、電動機１が弱め界磁制御で駆動されている間、線間電圧は、コンバータ電圧に対して飽和している。弱め界磁制御では、インバータ電圧と線間電圧との差分が小さいため、電動機電流の高周波成分が発生しにくい。そのため、弱め界磁制御では、電動機１に生じる鉄損に対する高調波鉄損の割合が特に小さい。したがって、インバータ１０３のインバータスイッチのオン時間の割合が１／√２に到達したとき、制御装置５０は、弱め界磁制御前に比べてキャリア周波数を下げる。これにより、スイッチング損失が低減され、インバータ１０３および電動機１の総合効率を改善することができる。 In the present embodiment, when the ratio of the on time of the inverter switch of the inverter 103 is 1/√2 or more, the electric motor 1 is driven by the field weakening control. As described above, the line voltage is saturated with respect to the converter voltage while the electric motor 1 is driven by the field weakening control. In the field weakening control, since the difference between the inverter voltage and the line voltage is small, the high frequency component of the motor current is unlikely to occur. Therefore, in the field weakening control, the ratio of the harmonic iron loss to the iron loss generated in the electric motor 1 is particularly small. Therefore, when the ratio of the on time of the inverter switch of the inverter 103 reaches 1/√2, the control device 50 lowers the carrier frequency as compared with before the field weakening control. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved.
　コンバータ１０２は、インバータ１０３に印加される電圧、すなわち、コンバータ電圧を昇圧する昇圧回路を有してもよい。線間電圧に対してコンバータ電圧が不足しているほど、弱め界磁制御中の弱め界磁電流が増大し、銅損が増大する。そのため、弱め界磁制御では、コンバータ電圧を線間電圧と釣り合うまで昇圧することが望ましい。 The converter 102 may have a booster circuit that boosts the voltage applied to the inverter 103, that is, the converter voltage. As the converter voltage becomes insufficient with respect to the line voltage, the field weakening current during field weakening control increases and copper loss increases. Therefore, in the field weakening control, it is desirable to boost the converter voltage until it balances with the line voltage.
　図１４は、コンバータ電圧と線間電圧との関係を示す図である。
　すなわち、弱め界磁制御では、図１４に示されるように、コンバータ電圧を弱め界磁制御が開始される電圧まで昇圧することが望ましい。これにより、コンバータ電圧に対して線間電圧が飽和し、インバータ電圧と線間電圧との差分が減少する。その結果、高調波鉄損の増加を抑えることができる。 FIG. 14 is a diagram showing the relationship between the converter voltage and the line voltage.
That is, in the field weakening control, as shown in FIG. 14, it is desirable to boost the converter voltage to a voltage at which the field weakening control is started. As a result, the line voltage is saturated with respect to the converter voltage, and the difference between the inverter voltage and the line voltage is reduced. As a result, it is possible to suppress an increase in harmonic iron loss.
　図１４に示される例では、電動機１の回転数がＮｄに到達したとき、インバータ１０３のインバータスイッチのオン時間の割合が１／√２に到達する。この場合、コンバータ１０２（具体的には、昇圧回路）は、コンバータ電圧の昇圧を開始し、制御装置５０は、弱め界磁制御を開始する。 In the example shown in FIG. 14, when the rotation speed of the electric motor 1 reaches Nd, the on-time ratio of the inverter switch of the inverter 103 reaches 1/√2. In this case, converter 102 (specifically, the booster circuit) starts boosting the converter voltage, and control device 50 starts the field weakening control.
　図１５は、電動機１の回転数とＰＷＭ制御中におけるインバータスイッチのオン時間の割合との関係を示す図である。
　本実施の形態では、コンバータ電圧が線間電圧と釣り合うとき、インバータスイッチのオン時間の割合が１／√２である。したがって、インバータスイッチのオン時間の割合が１／√２になるように、コンバータ１０２の昇圧回路は、インバータ１０３に印加される電圧を昇圧してもよい。図１５に示される例では、電動機１の回転数がＮｄ［ｒｐｍ］に到達したとき、インバータスイッチのオン時間の割合が１／√２に到達する。 FIG. 15 is a diagram showing the relationship between the rotation speed of the electric motor 1 and the ratio of the ON time of the inverter switch during PWM control.
In the present embodiment, when the converter voltage balances with the line voltage, the ratio of the on time of the inverter switch is 1/√2. Therefore, the booster circuit of the converter 102 may boost the voltage applied to the inverter 103 so that the ON time ratio of the inverter switch is 1/√2. In the example shown in FIG. 15, when the rotation speed of the electric motor 1 reaches Nd [rpm], the on-time ratio of the inverter switch reaches 1/√2.
　制御装置５０が弱め界磁制御を開始した後、コンバータ１０２（具体的には、昇圧回路）は、コンバータ電圧を昇圧してもよい。この場合、電動機１に対する弱め界磁制御中に、コンバータ１０２の昇圧回路は、インバータ１０３のインバータスイッチのオン時間の割合が１／√２になるようにインバータ１０３に印加される電圧を昇圧する。これにより、コンバータ電圧に対して線間電圧が飽和し、インバータ電圧と線間電圧との差分が減少する。その結果、高調波鉄損の増加を抑えることができる。 After the control device 50 starts the field weakening control, the converter 102 (specifically, the booster circuit) may boost the converter voltage. In this case, during the field weakening control for the electric motor 1, the booster circuit of the converter 102 boosts the voltage applied to the inverter 103 so that the ON time ratio of the inverter switch of the inverter 103 becomes 1/√2. As a result, the line voltage is saturated with respect to the converter voltage, and the difference between the inverter voltage and the line voltage is reduced. As a result, it is possible to suppress an increase in harmonic iron loss.
　図１６は、コンバータ電圧の昇圧前後におけるキャリア周波数を示す図である。
　インバータ１０３に印加される電圧が昇圧された後におけるキャリア周波数は、インバータ１０３に印加される電圧が昇圧される前におけるキャリア周波数よりも小さい。すなわち、コンバータ電圧の昇圧後、制御装置５０は、コンバータ電圧の昇圧前に比べてキャリア周波数を下げる。図１６に示される例では、コンバータ電圧の昇圧前において、制御装置５０は、キャリア周波数をｆ１に設定し、コンバータ電圧の昇圧後において、制御装置５０は、キャリア周波数をｆ２に設定する。この場合、例えばｆ１＝９０００Ｈｚであり、例えばｆ２＝４５００Ｈｚである。これにより、スイッチング損失が低減され、インバータ１０３および電動機１の総合効率を改善することができる。 FIG. 16 is a diagram showing carrier frequencies before and after boosting the converter voltage.
The carrier frequency after the voltage applied to the inverter 103 is boosted is lower than the carrier frequency before the voltage applied to the inverter 103 is boosted. That is, after boosting the converter voltage, control device 50 lowers the carrier frequency as compared with before boosting the converter voltage. In the example shown in FIG. 16, controller 50 sets the carrier frequency to f1 before the converter voltage is boosted, and controller 50 sets the carrier frequency to f2 after the converter voltage is boosted. In this case, for example, f1=9000 Hz and, for example, f2=4500 Hz. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved.
　インバータ１０３の少なくとも１つのスイッチング素子には、例えば、シリコンカーバイド（ＳｉＣ）素子が用いられる。これにより、スイッチング損失を低減することができる。特に、電動機１の高速回転時では、高調波鉄損に対するスイッチング損失の比率が増加しやすいため、電動機１の高速回転時における電動機１の効率を改善することができる。 A silicon carbide (SiC) element, for example, is used for at least one switching element of the inverter 103. Thereby, switching loss can be reduced. In particular, when the electric motor 1 rotates at high speed, the ratio of switching loss to harmonic iron loss tends to increase, so that the efficiency of the electric motor 1 during high speed rotation of the electric motor 1 can be improved.
　インバータ１０３の少なくとも１つのスイッチング素子には、例えば、窒化ガリウム（ＧａＮ）素子が用いられてもよい。これにより、スイッチング損失を低減することができる。特に、電動機１の高速回転時では、高調波鉄損に対するスイッチング損失の比率が増加しやすいため、電動機１の高速回転時における電動機１の効率を改善することができる。 A gallium nitride (GaN) element may be used for at least one switching element of the inverter 103, for example. Thereby, switching loss can be reduced. In particular, when the electric motor 1 rotates at high speed, the ratio of switching loss to harmonic iron loss tends to increase, so that the efficiency of the electric motor 1 during high speed rotation of the electric motor 1 can be improved.
　電動機１の回転数の閾値は１つに限られない。したがって、２つ以上の閾値が設定されていてもよい。この場合、制御装置５０は、キャリア周波数を段階的に下げる。すなわち、キャリア周波数は、２パターン以上の段階に設定されてもよい。 -The threshold value of the rotation speed of the electric motor 1 is not limited to one. Therefore, two or more threshold values may be set. In this case, the control device 50 lowers the carrier frequency stepwise. That is, the carrier frequency may be set in two or more patterns.
　図１７は、電動機１の回転数とキャリア周波数との関係を示す図である。周波数ｆ１［Ｈｚ］からｆ３［Ｈｚ］は、ｆ１＞ｆ２＞ｆ３≧ｆｍを満たす。回転数Ｎ１［ｒｐｍ］およびＮ２［ｒｐｍ］は、Ｎ１＜Ｎ２を満たす。
　図１７に示される例では、制御装置５０は、電動機１の回転数に応じてキャリア周波数を段階的に切り替える。具体的には、制御装置５０は、電動機１の回転数に応じてキャリア周波数を段階的に下げる。すなわち、図１７に示される例では、電動機１の回転数がＮ１に到達したとき、制御装置５０は、キャリア周波数をｆ１からｆ２に下げる。電動機１の回転数がＮ１からＮ２に到達したとき、制御装置５０は、キャリア周波数をｆ２からｆ３に下げる。 FIG. 17 is a diagram showing the relationship between the rotation speed of the electric motor 1 and the carrier frequency. The frequencies f1 [Hz] to f3 [Hz] satisfy f1>f2>f3≧fm. The rotation speeds N1 [rpm] and N2 [rpm] satisfy N1<N2.
In the example shown in FIG. 17, the control device 50 switches the carrier frequency stepwise according to the rotation speed of the electric motor 1. Specifically, the control device 50 gradually lowers the carrier frequency according to the rotation speed of the electric motor 1. That is, in the example shown in FIG. 17, when the rotation speed of electric motor 1 reaches N1, control device 50 lowers the carrier frequency from f1 to f2. When the rotation speed of the electric motor 1 reaches from N1 to N2, the control device 50 lowers the carrier frequency from f2 to f3.
　上述の例では、キャリア周波数は、段階的に変更される。しかしながら、制御装置５０は、キャリア周波数を連続的に下げてもよい。
　図１８は、電動機１の回転数とキャリア周波数との関係を示す図である。
　例えば、図１８に示されるように、電動機１の回転数が増加するにつれて、制御装置５０は、キャリア周波数を下げてもよい。 In the above example, the carrier frequency is changed stepwise. However, the control device 50 may continuously reduce the carrier frequency.
FIG. 18 is a diagram showing the relationship between the rotation speed of the electric motor 1 and the carrier frequency.
For example, as shown in FIG. 18, controller 50 may decrease the carrier frequency as the rotation speed of electric motor 1 increases.
　図１８に示される例では、電動機１の回転数が増加するにつれて、制御装置５０は、キャリア周波数を連続的に下げる。さらに、電動機１の回転数がＮｄ［ｒｐｍ］に到達したとき、制御装置５０は、弱め界磁制御を開始する。弱め界磁制御中におけるキャリア周波数は一定である。すなわち、弱め界磁制御中では、制御装置５０は、キャリア周波数を一定にする。これにより、スイッチング損失が低減され、インバータ１０３および電動機１の総合効率を改善することができる。 In the example shown in FIG. 18, the control device 50 continuously lowers the carrier frequency as the rotation speed of the electric motor 1 increases. Further, when the rotation speed of the electric motor 1 reaches Nd [rpm], the control device 50 starts the field weakening control. The carrier frequency is constant during the field weakening control. That is, during the field weakening control, the control device 50 keeps the carrier frequency constant. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved.
　図１９は、インバータ１０３のインバータスイッチに対するパルス幅変調制御中におけるインバータスイッチのオン時間の割合とキャリア周波数との関係を示す図である。
　図１９に示されるように、インバータスイッチのオン時間の割合が増加するにつれて、制御装置５０は、キャリア周波数を下げてもよい。 FIG. 19 is a diagram showing the relationship between the on-time ratio of the inverter switch and the carrier frequency during pulse width modulation control of the inverter switch of the inverter 103.
As shown in FIG. 19, the controller 50 may decrease the carrier frequency as the ratio of the on time of the inverter switch increases.
　図１９に示される例では、インバータスイッチのオン時間の割合が増加するにつれて、制御装置５０は、キャリア周波数を連続的に下げる。さらに、インバータスイッチのオン時間の割合が閾値Ｎｔに到達したとき、制御装置５０は、弱め界磁制御を開始する。弱め界磁制御中におけるキャリア周波数は一定である。すなわち、弱め界磁制御中では、制御装置５０は、キャリア周波数を一定にする。これにより、スイッチング損失が低減され、インバータ１０３および電動機１の総合効率を改善することができる。 In the example shown in FIG. 19, the control device 50 continuously lowers the carrier frequency as the ratio of the on time of the inverter switch increases. Further, when the ratio of the on time of the inverter switch reaches the threshold value Nt, the control device 50 starts the field weakening control. The carrier frequency is constant during the field weakening control. That is, during the field weakening control, the control device 50 keeps the carrier frequency constant. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved.
　以上に説明したように、実施の形態１に係る駆動装置１０１では、制御装置５０は、電動機１の回転数に応じてキャリア周波数を切り替える。これにより、スイッチング損失が低減され、インバータ１０３および電動機１の総合効率を改善することができる。その結果、駆動装置１０１の効率を改善することができる。 As described above, in drive device 101 according to the first embodiment, control device 50 switches the carrier frequency according to the rotation speed of electric motor 1. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved. As a result, the efficiency of the driving device 101 can be improved.
　電動機１の回転数が増加するほど、電動機１における全損失に占める高調波鉄損の割合が小さい。したがって、電動機１の回転数が増加したとき、制御装置５０は、電動機１の回転数に応じてキャリア周波数を下げることが望ましい。これにより、電動機１の高速回転時においてスイッチング損失が効果的に低減され、インバータ１０３および電動機１の総合効率を効果的に改善することができる。その結果、駆動装置１０１の効率を効果的に改善することができる。 The higher the number of revolutions of the electric motor 1, the smaller the proportion of the harmonic iron loss in the total loss in the electric motor 1. Therefore, when the rotation speed of the electric motor 1 increases, the control device 50 preferably lowers the carrier frequency according to the rotation speed of the electric motor 1. As a result, switching loss is effectively reduced during high-speed rotation of the electric motor 1, and the overall efficiency of the inverter 103 and the electric motor 1 can be effectively improved. As a result, the efficiency of the driving device 101 can be effectively improved.
　本実施の形態では、制御装置５０が電動機１に対する弱め界磁制御を開始したとき、制御装置５０は、弱め界磁制御を開始する前よりもキャリア周波数を下げる。これにより、スイッチング損失が低減され、インバータ１０３および電動機１の総合効率を改善することができる。その結果、駆動装置１０１の効率を改善することができる。 In the present embodiment, when the control device 50 starts the field weakening control for the electric motor 1, the control device 50 lowers the carrier frequency compared to before the field weakening control is started. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved. As a result, the efficiency of the driving device 101 can be improved.
　本実施の形態では、インバータ１０３のインバータスイッチのオン時間の割合が１／√２以上のとき、電動機１は、弱め界磁制御で駆動される。この場合、制御装置５０は、弱め界磁制御前に比べてキャリア周波数を下げる。これにより、スイッチング損失が低減され、インバータ１０３および電動機１の総合効率を改善することができる。その結果、駆動装置１０１の効率を改善することができる。 In the present embodiment, when the ratio of the on time of the inverter switch of the inverter 103 is 1/√2 or more, the electric motor 1 is driven by the field weakening control. In this case, the control device 50 lowers the carrier frequency compared to before the field weakening control. As a result, switching loss is reduced and the overall efficiency of the inverter 103 and the electric motor 1 can be improved. As a result, the efficiency of the driving device 101 can be improved.
　弱め界磁制御では、弱め界磁電流が増大し、銅損が増大する。そのため、弱め界磁制御では、コンバータ電圧を線間電圧と釣り合うまで昇圧することが望ましい。これにより、弱め界磁電流が緩和され、銅損が低減される。コンバータ電圧を昇圧する場合、コンバータ電圧を弱め界磁制御が開始される電圧まで昇圧することが望ましい。これにより、コンバータ電圧に対して線間電圧が飽和し、インバータ電圧と線間電圧との差分が減少する。その結果、高調波鉄損の増加を抑えることができる。 ㆍField weakening control increases field weakening current and copper loss. Therefore, in the field weakening control, it is desirable to boost the converter voltage until it balances with the line voltage. This alleviates the field weakening current and reduces copper loss. When boosting the converter voltage, it is desirable to weaken the converter voltage to a voltage at which field control is started. As a result, the line voltage is saturated with respect to the converter voltage, and the difference between the inverter voltage and the line voltage is reduced. As a result, it is possible to suppress an increase in harmonic iron loss.
　例えば、図１６に示されるように、コンバータ電圧を昇圧するとき、制御装置５０は、昇圧する前よりもキャリア周波数を下げる。キャリア周波数の低下は、スイッチング損失の低減に寄与する。これにより、インバータ１０３および電動機１の総合効率を改善することができる。その結果、駆動装置１０１の効率を改善することができる。 For example, as shown in FIG. 16, when boosting the converter voltage, the control device 50 lowers the carrier frequency more than before it boosts it. The lowering of the carrier frequency contributes to the reduction of switching loss. Thereby, the total efficiency of the inverter 103 and the electric motor 1 can be improved. As a result, the efficiency of the driving device 101 can be improved.
　インバータ１０３の少なくとも１つのスイッチング素子には、例えば、シリコンカーバイド（ＳｉＣ）素子または窒化ガリウム（ＧａＮ）素子が用いられる。これにより、スイッチング損失を低減することができる。特に、電動機１の高速回転時では、高調波鉄損の割合が小さく、スイッチング損失の割合が増加しやすいため、電動機１の高速回転時における電動機１の効率を改善することができる。その結果、駆動装置１０１の効率を改善することができる。 A silicon carbide (SiC) element or a gallium nitride (GaN) element is used for at least one switching element of the inverter 103. Thereby, switching loss can be reduced. In particular, when the electric motor 1 rotates at high speed, the ratio of harmonic iron loss is small and the ratio of switching loss easily increases, so that the efficiency of the electric motor 1 at high speed rotation of the electric motor 1 can be improved. As a result, the efficiency of the driving device 101 can be improved.
実施の形態２．
　本発明の実施の形態２に係る圧縮機６について説明する。
　図２０は、実施の形態２に係る圧縮機６の構造を概略的に示す断面図である。 Embodiment 2.
A compressor 6 according to the second embodiment of the present invention will be described.
FIG. 20 is a sectional view schematically showing the structure of the compressor 6 according to the second embodiment.
　圧縮機６は、実施の形態１に係る駆動装置１０１と、駆動装置１０１によって駆動される電動要素としての電動機６０と、ハウジングとしての密閉容器６１と、圧縮要素としての圧縮機構６２とを有する。本実施の形態では、圧縮機６は、ロータリー圧縮機である。ただし、圧縮機６は、ロータリー圧縮機に限定されない。 The compressor 6 includes the drive device 101 according to the first embodiment, an electric motor 60 as an electric element driven by the drive device 101, a closed container 61 as a housing, and a compression mechanism 62 as a compression element. In the present embodiment, the compressor 6 is a rotary compressor. However, the compressor 6 is not limited to the rotary compressor.
　電動機６０には、実施の形態１で説明した電動機１が適用される。電動機６０は、圧縮機構６２を駆動する。 The electric motor 1 described in the first embodiment is applied to the electric motor 60. The electric motor 60 drives the compression mechanism 62.
　密閉容器６１は、電動機６０および圧縮機構６２を覆う。密閉容器６１は、円筒状の容器である。密閉容器６１の底部には、圧縮機構６２の摺動部分を潤滑する冷凍機油が貯留されている。 The closed container 61 covers the electric motor 60 and the compression mechanism 62. The closed container 61 is a cylindrical container. Refrigerating machine oil that lubricates the sliding portion of the compression mechanism 62 is stored at the bottom of the closed container 61.
　圧縮機６は、さらに、密閉容器６１に固定されたガラス端子６３と、アキュムレータ６４と、吸入パイプ６５と、吐出パイプ６６とを有する。 The compressor 6 further includes a glass terminal 63 fixed to the closed container 61, an accumulator 64, a suction pipe 65, and a discharge pipe 66.
　圧縮機構６２は、シリンダ６２ａと、ピストン６２ｂと、上部フレーム６２ｃ（第１のフレームとも称する）と、下部フレーム６２ｄ（第２のフレームとも称する）と、上部フレーム６２ｃおよび下部フレーム６２ｄに取り付けられた複数のマフラ６２ｅとを有する。圧縮機構６２は、さらに、シリンダ６２ａ内を吸入側と圧縮側とに分けるベーンを有する。圧縮機構６２は、電動機６０によって駆動される。 The compression mechanism 62 is attached to a cylinder 62a, a piston 62b, an upper frame 62c (also referred to as a first frame), a lower frame 62d (also referred to as a second frame), an upper frame 62c and a lower frame 62d. And a plurality of mufflers 62e. The compression mechanism 62 further has a vane that divides the inside of the cylinder 62a into a suction side and a compression side. The compression mechanism 62 is driven by the electric motor 60.
　電動機６０は、圧入または焼き嵌めで密閉容器６１内に固定されている。圧入および焼き嵌めの代わりに溶接で電動機６０を密閉容器６１に直接取り付けてもよい。 The electric motor 60 is fixed in the closed container 61 by press fitting or shrink fitting. Instead of press fitting and shrink fitting, the electric motor 60 may be directly attached to the closed container 61 by welding.
　電動機６０のコイル（例えば、実施の形態１で説明したコイル３）には、ガラス端子６３を通して駆動装置１０１から電力が供給される。 Electric power is supplied to the coil of the electric motor 60 (for example, the coil 3 described in the first embodiment) from the drive device 101 through the glass terminal 63.
　電動機６０のロータ２０（具体的には、シャフト６７の片側）は、上部フレーム６２ｃおよび下部フレーム６２ｄの各々に備えられた軸受けによって回転自在に支持されている。 The rotor 20 of the electric motor 60 (specifically, one side of the shaft 67) is rotatably supported by bearings provided in each of the upper frame 62c and the lower frame 62d.
　ピストン６２ｂには、シャフト６７が挿通されている。上部フレーム６２ｃおよび下部フレーム６２ｄには、シャフト６７が回転自在に挿通されている。上部フレーム６２ｃおよび下部フレーム６２ｄは、シリンダ６２ａの端面を閉塞する。アキュムレータ６４は、吸入パイプ６５を通して冷媒（例えば、冷媒ガス）をシリンダ６２ａに供給する。 A shaft 67 is inserted through the piston 62b. A shaft 67 is rotatably inserted through the upper frame 62c and the lower frame 62d. The upper frame 62c and the lower frame 62d close the end surface of the cylinder 62a. The accumulator 64 supplies a refrigerant (for example, refrigerant gas) to the cylinder 62a through the suction pipe 65.
　次に、圧縮機６の動作について説明する。アキュムレータ６４から供給された冷媒は、密閉容器６１に固定された吸入パイプ６５からシリンダ６２ａ内へ吸入される。電動機６０が回転することにより、シャフト６７に嵌合されたピストン６２ｂがシリンダ６２ａ内で回転する。これにより、シリンダ６２ａ内で冷媒が圧縮される。 Next, the operation of the compressor 6 will be described. The refrigerant supplied from the accumulator 64 is sucked into the cylinder 62a from the suction pipe 65 fixed to the closed container 61. When the electric motor 60 rotates, the piston 62b fitted on the shaft 67 rotates in the cylinder 62a. As a result, the refrigerant is compressed in the cylinder 62a.
　冷媒は、マフラ６２ｅを通り、密閉容器６１内を上昇する。このようにして、圧縮された冷媒が、吐出パイプ６６を通って冷凍サイクルの高圧側へ供給される。 The refrigerant passes through the muffler 62e and rises in the closed container 61. In this way, the compressed refrigerant is supplied to the high pressure side of the refrigeration cycle through the discharge pipe 66.
　圧縮機６の冷媒として、Ｒ４１０Ａ、Ｒ４０７Ｃ、またはＲ２２等を用いることができる。ただし、圧縮機６の冷媒は、これらの種類に限られない。圧縮機６の冷媒として、ＧＷＰ（地球温暖化係数）が小さい冷媒、例えば、下記の冷媒を用いることができる。 As the refrigerant of the compressor 6, R410A, R407C, R22 or the like can be used. However, the refrigerant of the compressor 6 is not limited to these types. As the refrigerant of the compressor 6, a refrigerant having a small GWP (global warming potential), for example, the following refrigerant can be used.
（１）組成中に炭素の二重結合を有するハロゲン化炭化水素、例えばＨＦＯ（Ｈｙｄｒｏ－Ｆｌｕｏｒｏ－Ｏｒｅｆｉｎ）－１２３４ｙｆ（ＣＦ３ＣＦ＝ＣＨ２）を用いることができる。ＨＦＯ－１２３４ｙｆのＧＷＰは４である。
（２）組成中に炭素の二重結合を有する炭化水素、例えばＲ１２７０（プロピレン）を用いてもよい。Ｒ１２７０のＧＷＰは３であり、ＨＦＯ－１２３４ｙｆより低いが、可燃性はＨＦＯ－１２３４ｙｆより高い。
（３）組成中に炭素の二重結合を有するハロゲン化炭化水素または組成中に炭素の二重結合を有する炭化水素の少なくとも何れかを含む混合物、例えばＨＦＯ－１２３４ｙｆとＲ３２との混合物を用いてもよい。上述したＨＦＯ－１２３４ｙｆは低圧冷媒のため圧損が大きくなる傾向があり、冷凍サイクル（特に蒸発器）の性能低下を招く可能性がある。そのため、ＨＦＯ－１２３４ｙｆよりも高圧冷媒であるＲ３２またはＲ４１との混合物を用いることが実用上は望ましい。 (1) A halogenated hydrocarbon having a carbon double bond in the composition, for example, HFO (Hydro-Fluoro-Orefin)-1234yf (CF3CF=CH2) can be used. The GWP of HFO-1234yf is 4.
(2) A hydrocarbon having a carbon double bond in the composition, for example, R1270 (propylene) may be used. R1270 has a GWP of 3, which is lower than that of HFO-1234yf, but higher than that of HFO-1234yf.
(3) Using a mixture containing at least either a halogenated hydrocarbon having a carbon double bond in the composition or a hydrocarbon having a carbon double bond in the composition, for example, a mixture of HFO-1234yf and R32. Good. Since the above-mentioned HFO-1234yf is a low-pressure refrigerant, it tends to have a large pressure loss, which may lead to a reduction in the performance of the refrigeration cycle (especially the evaporator). Therefore, it is practically desirable to use a mixture with R32 or R41, which is a higher pressure refrigerant than HFO-1234yf.
　実施の形態２に係る圧縮機６は、実施の形態１で説明した利点を有する。 The compressor 6 according to the second embodiment has the advantages described in the first embodiment.
　さらに、実施の形態２に係る圧縮機６が駆動装置１０１を有するので、圧縮機６の効率を改善することができる。 Further, since the compressor 6 according to the second embodiment has the drive device 101, the efficiency of the compressor 6 can be improved.
実施の形態３．
　本発明の実施の形態３に係る冷凍空調装置７について説明する。
　図２１は、実施の形態３に係る冷凍空調装置７の構成を概略的に示す図である。 Embodiment 3.
The refrigerating and air-conditioning apparatus 7 according to Embodiment 3 of the present invention will be described.
FIG. 21 is a diagram schematically showing the configuration of the refrigerating and air-conditioning apparatus 7 according to the third embodiment.
　冷凍空調装置７は、実施の形態２に係る圧縮機６と、四方弁７１と、凝縮器７２と、減圧装置７３（膨張器ともいう）と、蒸発器７４と、冷媒配管７５と、制御部７６とを有する。図２１に示される例では、圧縮機６、凝縮器７２、減圧装置７３、および蒸発器７４は、冷媒配管７５によって連結され、冷凍サイクルを構成している。 The refrigeration/air-conditioning apparatus 7 includes a compressor 6, a four-way valve 71, a condenser 72, a decompression device 73 (also called an expander), an evaporator 74, a refrigerant pipe 75, and a control unit according to the second embodiment. 76 and. In the example shown in FIG. 21, the compressor 6, the condenser 72, the decompression device 73, and the evaporator 74 are connected by a refrigerant pipe 75 to form a refrigeration cycle.
　冷凍空調装置７の動作の一例について説明する。圧縮機６は、吸入した冷媒を圧縮し、高温高圧のガス冷媒を送り出す。四方弁７１は、冷媒の流れ方向を切り換える。図２１に示される例では、四方弁７１は、圧縮機６から送り出された冷媒を凝縮器７２に送る。凝縮器７２は、圧縮機６から送り出された冷媒と空気（例えば、室外の空気）との熱交換を行うことにより、冷媒を凝縮し、液化された冷媒を送り出す。減圧装置７３は、凝縮器７２から送り出された冷媒（すなわち、液化された冷媒）を膨張させて、低温低圧の液化された冷媒を送り出す。 An example of the operation of the refrigerating and air-conditioning apparatus 7 will be described. The compressor 6 compresses the sucked refrigerant and sends out a high-temperature and high-pressure gas refrigerant. The four-way valve 71 switches the flow direction of the refrigerant. In the example shown in FIG. 21, the four-way valve 71 sends the refrigerant sent from the compressor 6 to the condenser 72. The condenser 72 condenses the refrigerant by exchanging heat between the refrigerant sent from the compressor 6 and air (for example, outdoor air), and sends out the liquefied refrigerant. The decompression device 73 expands the refrigerant (that is, the liquefied refrigerant) sent from the condenser 72, and sends the low-temperature and low-pressure liquefied refrigerant.
　蒸発器７４は、減圧装置７３から送り出された低温低圧の液化された冷媒と空気（例えば、室内の空気）との熱交換を行うことにより、冷媒を気化させ、気化された冷媒（すなわち、ガス冷媒）を送り出す。蒸発器７４で熱が奪われた空気は、例えば、送風機により、対象空間（例えば室内）に供給される。四方弁７１および圧縮機６の動作は、制御部７６によって制御される。 The evaporator 74 evaporates the refrigerant by exchanging heat between the low-temperature low-pressure liquefied refrigerant sent from the decompression device 73 and air (for example, indoor air), and the evaporated refrigerant (that is, gas). (Refrigerant) is sent out. The air from which heat has been removed by the evaporator 74 is supplied to the target space (for example, the room) by, for example, a blower. The operations of the four-way valve 71 and the compressor 6 are controlled by the controller 76.
　冷凍空調装置７において、圧縮機６以外の構成要素は、実施の形態３で説明された構成要素に限定されない。 In the refrigerating and air-conditioning apparatus 7, the constituent elements other than the compressor 6 are not limited to the constituent elements described in the third embodiment.
　実施の形態３に係る冷凍空調装置７は、実施の形態２で説明した利点を有する。 The refrigerating and air-conditioning system 7 according to the third embodiment has the advantages described in the second embodiment.
　さらに、冷凍空調装置７が圧縮機６を有するので、冷凍空調装置７の効率を改善することができる。 Further, since the refrigeration air conditioning system 7 has the compressor 6, the efficiency of the refrigeration air conditioning system 7 can be improved.
　実施の形態１で説明した駆動装置１０１は、圧縮機６および冷凍空調装置７以外に、送風機、換気扇、家電機器、または工作機などの機器における駆動装置に適用できる。 The drive device 101 described in the first embodiment can be applied to a drive device in a device such as a blower, a ventilation fan, a home electric appliance, or a machine tool, in addition to the compressor 6 and the refrigerating and air conditioning device 7.
　本発明の望ましい実施の形態について具体的に説明したが、本発明は上記の実施の形態に限定されるものではなく、本発明の要旨を逸脱しない範囲において、種々の改良または変形を行なうことができる。 Although the preferred embodiments of the present invention have been specifically described, the present invention is not limited to the above embodiments, and various improvements and modifications can be made without departing from the scope of the present invention. it can.
　１，６０　電動機、　３，３Ｕ，３Ｖ，３Ｗ　コイル、　６　圧縮機、　７　冷凍空調装置　、　５０　制御装置、　１０１　駆動装置、　１０２　コンバータ、　１０３　インバータ。 1,60 electric motors, 3,3U, 3V, 3W coils, 6 compressors, 7 refrigeration and air conditioning units, 50 control units, 101 drive units, 102 converters, 103 inverters.

Claims (16)

  1. 　コイルを有する電動機を駆動する駆動装置であって、
    　前記コイルに電圧を印加するインバータと、
    　前記コイルに印加される前記電圧の制御周波数を調整するための前記インバータのキャリア周波数を制御し、前記電動機の回転数に応じて前記キャリア周波数を切り替える制御装置と
    　を備えた駆動装置。     A drive device for driving an electric motor having a coil,
    An inverter for applying a voltage to the coil,
    A drive device, comprising: a control device that controls the carrier frequency of the inverter for adjusting the control frequency of the voltage applied to the coil, and switches the carrier frequency according to the rotation speed of the electric motor.
  2. 　前記制御装置が前記電動機に対する弱め界磁制御を開始したとき、前記制御装置は、前記弱め界磁制御を開始する前よりも前記キャリア周波数を下げる請求項１に記載の駆動装置。 The drive device according to claim 1, wherein when the control device starts field-weakening control for the electric motor, the control device lowers the carrier frequency more than before the field-weakening control is started.
  3. 　前記電動機の回転数が閾値よりも小さいとき、前記キャリア周波数は第１の周波数であり、
    　前記電動機の回転数が前記閾値以上であるとき、前記キャリア周波数は前記第１の周波数よりも小さい第２の周波数である
    　請求項１または２に記載の駆動装置。     When the rotation speed of the electric motor is smaller than a threshold value, the carrier frequency is a first frequency,
    The drive device according to claim 1, wherein when the rotation speed of the electric motor is equal to or higher than the threshold value, the carrier frequency is a second frequency smaller than the first frequency.
  4. 　前記インバータは、少なくとも１つのインバータスイッチを有し、
    　前記少なくとも１つのインバータスイッチに対するパルス幅変調制御中における前記少なくとも１つのインバータスイッチのオン時間の割合が閾値よりも小さいとき、前記キャリア周波数は第１の周波数であり、
    　前記少なくとも１つのインバータスイッチの前記オン時間の割合が前記閾値以上であるとき、前記キャリア周波数は前記第１の周波数よりも小さい第２の周波数である
    　請求項１または２に記載の駆動装置。     The inverter has at least one inverter switch,
    When the ratio of the on-time of the at least one inverter switch during pulse width modulation control for the at least one inverter switch is less than a threshold value, the carrier frequency is a first frequency,
    The drive device according to claim 1, wherein the carrier frequency is a second frequency smaller than the first frequency when the ratio of the on-time of the at least one inverter switch is equal to or higher than the threshold value.
  5. 　前記閾値は、１／√２以上である請求項４に記載の駆動装置。 The drive device according to claim 4, wherein the threshold value is 1/√2 or more.
  6. 　前記インバータにコンバータ電圧を印加するコンバータをさらに備え、
    　前記コンバータは、前記コンバータ電圧を昇圧する昇圧回路を有し、
    　前記電動機に対する弱め界磁制御中に、前記昇圧回路は、前記オン時間の割合が１／√２になるように前記コンバータ電圧を昇圧する
    　請求項４または５に記載の駆動装置。     Further comprising a converter for applying a converter voltage to the inverter,
    The converter has a booster circuit that boosts the converter voltage,
    The drive device according to claim 4, wherein during the field-weakening control for the electric motor, the booster circuit boosts the converter voltage so that the ratio of the on-time becomes 1/√2.
  7. 　前記インバータにコンバータ電圧を印加するコンバータをさらに備え、
    　前記コンバータは、前記コンバータ電圧を昇圧する昇圧回路を有する請求項１から５のいずれか１項に記載の駆動装置。     Further comprising a converter for applying a converter voltage to the inverter,
    The drive device according to claim 1, wherein the converter includes a booster circuit that boosts the converter voltage.
  8. 　前記コンバータ電圧が昇圧された後における前記キャリア周波数は、前記コンバータ電圧が昇圧される前における前記キャリア周波数よりも小さい請求項６または７に記載の駆動装置。 The drive device according to claim 6 or 7, wherein the carrier frequency after the converter voltage is boosted is smaller than the carrier frequency before the converter voltage is boosted.
  9. 　前記少なくとも１つのインバータスイッチにはシリコンカーバイド素子が用いられている請求項４に記載の駆動装置。 The drive device according to claim 4, wherein a silicon carbide element is used for the at least one inverter switch.
  10. 　前記少なくとも１つのインバータスイッチには窒化ガリウム素子が用いられている請求項４に記載の駆動装置。 The drive device according to claim 4, wherein a gallium nitride element is used for the at least one inverter switch.
  11. 　前記電動機の回転数が増加するにつれて、前記制御装置は、前記キャリア周波数を下げる請求項１に記載の駆動装置。 The drive unit according to claim 1, wherein the control unit lowers the carrier frequency as the rotation speed of the electric motor increases.
  12. 　電動機と、請求項１から１１のいずれか１項に記載の駆動装置を備えた圧縮機。 A compressor equipped with an electric motor and the drive device according to any one of claims 1 to 11.
  13. 　請求項１２に記載の圧縮機を備えた冷凍空調装置。 A refrigerating and air-conditioning apparatus equipped with the compressor according to claim 12.
  14. 　コイルを有する電動機の駆動方法であって、
    　前記コイルに印加される電圧の制御周波数を調整するためのインバータのキャリア周波数を設定し、
    　前記電動機の回転数に応じて前記キャリア周波数を切り替える
    　電動機の駆動方法。     A method for driving an electric motor having a coil, comprising:
    Setting the carrier frequency of the inverter for adjusting the control frequency of the voltage applied to the coil,
    A method for driving an electric motor, wherein the carrier frequency is switched according to the rotation speed of the electric motor.
  15. 　前記電動機に対する弱め界磁制御を開始したとき、前記弱め界磁制御を開始する前よりも前記キャリア周波数を下げる請求項１４に記載の電動機の駆動方法。 The method of driving an electric motor according to claim 14, wherein when the field weakening control for the electric motor is started, the carrier frequency is lowered more than before the field weakening control is started.
  16. 　前記電動機の回転数が閾値よりも小さいとき、前記キャリア周波数は第１の周波数であり、
    　前記電動機の回転数が前記閾値以上であるとき、前記キャリア周波数は前記第１の周波数よりも小さい第２の周波数である
    　請求項１４または１５に記載の電動機の駆動方法。     When the rotation speed of the electric motor is smaller than a threshold value, the carrier frequency is a first frequency,
    The method for driving an electric motor according to claim 14 or 15, wherein when the rotation speed of the electric motor is equal to or higher than the threshold value, the carrier frequency is a second frequency smaller than the first frequency.
PCT/JP2019/001935 2019-01-23 2019-01-23 Drive apparatus, compressor, refrigeration air conditioning apparatus, and electric motor driving method WO2020152792A1 (en)

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JP2005176437A (en) * 2003-12-09 2005-06-30 Matsushita Electric Ind Co Ltd Method and device for driving brushless dc motor
JP2010221856A (en) * 2009-03-24 2010-10-07 Hitachi Automotive Systems Ltd Steering control device
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Publication number Priority date Publication date Assignee Title
JP2001186787A (en) * 1999-12-24 2001-07-06 Mitsubishi Electric Corp Permanent-magnet motor, method and apparatus for controlling permanent-magnet motor, compressor, and refrigerator-air conditioner
JP2005176437A (en) * 2003-12-09 2005-06-30 Matsushita Electric Ind Co Ltd Method and device for driving brushless dc motor
JP2010221856A (en) * 2009-03-24 2010-10-07 Hitachi Automotive Systems Ltd Steering control device
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