WO2020104764A1

WO2020104764A1 - A method of controlling a brushless permanent magnet motor

Info

Publication number: WO2020104764A1
Application number: PCT/GB2019/052950
Authority: WO
Inventors: Nathan CROFT; David Evans
Original assignee: Dyson Technology Limited
Priority date: 2018-11-22
Filing date: 2019-10-16
Publication date: 2020-05-28
Also published as: GB2579185A; CN113169686A; GB2579185B; GB201819005D0

Abstract

A method of controlling a brushless permanent magnet motor (14) includes measuring (102) an input power of the motor (14). The method includes comparing (104) the measured input power to a target input power. The method includes applying (106) a first correction factor to an excitation timing parameter of the motor (14) to match the measured input power to the target input power. The method includes applying (116,120) a second correction factor to the excitation timing parameter to alter the speed of the motor (14). The method includes exciting the motor (14) in accordance with the excitation timing parameter.

Description

A Method of Controlling a Brushless Permanent Magnet Motor

FIELD OF THE INVENTION

The present invention relates to a method of controlling a brushless permanent magnet motor.

BACKGROUND OF THE INVENTION

Brushless permanent magnet motors comprise phase windings which are excited via a power source in such a manner that the magnetic fields induced in the windings drive rotation of a rotor of the motor. Commutation of the phase windings occurs relative to zero crossings of back EMF induced in the windings, and it is known to advance commutation of a winding by a so-called advance angle, such that commutation of the winding occurs prior to a zero-crossing of back EMF induced in the winding. Phase current is driven into the phase winding for a period of time which is often referred to as a conduction period, or phase current is driven into the phase winding for a pre-determ ined ratio of a cycle which is known as a duty cycle.

It has previously been proposed to utilise look-up tables to determine desired advance angles, conduction periods, and duty cycles.

SUMMARY OF THE INVENTION

According to a first aspect of the present invention there is provided a method of controlling a brushless permanent magnet motor, the method comprising measuring an input power of the motor, comparing the measured input power to a target input power, applying a first correction factor to an excitation timing parameter of the motor to substantially match the measured input power to the target input power, applying a second correction factor to the excitation timing parameter to alter the speed of the motor, and exciting the motor in accordance with the excitation timing parameter.

The method according to the present invention may be advantageous principally as the method comprises applying a first correction factor to an excitation timing parameter of the motor to substantially match the measured input power to the target input power, applying a second correction factor to the excitation timing parameter to alter the speed of the motor, and exciting the motor in accordance with the excitation timing parameter.

In particular, for a brushless permanent magnet motor, for example operating as a fan or compressor, peak operating efficiency may occur when the motor operates at its peak speed for a given input power. Application of the first correction factor to the excitation timing parameter may ensure that the motor is operating at the target input power, whilst application of the second correction factor may enable the speed to be altered such that the motor is operating at or near its peak operating speed for the given target input power, thereby ensuring that the motor is operating at or near its most efficient operating point for a given target input power.

Use of the first and second correction factors in response to a measured input power may remove or reduce the need for the use of look-up tables or equations to determine desired excitation timing parameters such that the motor operates at a desired input power at its greatest efficiency for said input power. The lack of, or reduced, need for look-up tables or equations may simplify the processing requirements and/or memory requirements of a processor for the control of the motor, which may result in the use of a physically smaller and/or a less expensive processor, ie a less expensive processing method, than arrangements known in the art. The method according to the first aspect of the present invention may also enable a motor to optimise its own control parameters in use, and hence may minimise or remove the need for end of line calibration of the motor during manufacture.

The method may comprise measuring an input power of a phase winding of the motor, and exciting the phase winding of the motor in accordance with the excitation timing parameter.

The method may comprise applying the first correction factor until the measured input power substantially matches the target input power. The method may comprise applying the second correction factor to the excitation timing parameter once the measured input power substantially matches the target input power.

The method may comprise applying the first correction factor in response to the measured input power. The method may comprise applying the first correction factor prior to applying the second correction factor.

The method may comprise applying the second correction factor to the excitation timing parameter such that the motor is operating substantially at peak speed for the measured input power.

By operating substantially at peak speed is meant that the motor is operating at, or near to, its peak operating speed for the target input power. The method may control the motor to operate at a speed which is within 10%, within 5%, within 4%, within 3%, within 2%, or within 1 %, or within 0.5%, or within 0.1 %, of the peak operating speed for the measured input power.

The method may comprise measuring the speed of the motor pre- and/or post- applying the second correction factor. Measuring the input power of the motor, for example the input power of a phase winding of the motor, may comprise measuring the supply voltage and/or the phase current through the motor. Measuring the input power of the motor may comprise measuring the supply voltage and/or the supply current. Measuring the supply current as opposed to measuring the phase current, for example the current through the phase winding of the motor, may be beneficial as it may provide a more accurate measurement of the current driven into the motor, as switching losses may not need to be taken into account.

The excitation timing parameter may comprise a parameter which defines when a phase winding of the motor is commutated and/or when excitation of the motor, for example a phase winding of the motor, begins and/or how long the motor, for example a phase winding of the motor, is excited for. Excitation of the motor may comprise driving a phase current through a phase winding of the motor.

The excitation timing parameter may comprise an advance angle, for example a phase angle at which commutation of a phase winding of the motor occurs and hence a phase angle at which excitation of the phase winding begins. The advance angle may be measured relative to a zero-crossing of back EMF induced in the phase winding. Whilst referred to as an advance angle, which may imply that commutation of the phase winding occurs in advance of a zero-crossing of back EMF induced in the phase winding, the advance angle may comprise a positive, negative, or zero value. Thus commutation of the phase winding may occur in advance of, after, or synchronously with, a zero-crossing of back EMF induced in the phase winding. Where the advance angle takes on a negative value, the advance angle may instead be referred to as a retard angle, as commutation of the phase winding may be retarded relative to a zero-crossing of back EMF induced in the phase winding. The excitation timing parameter may comprise a conduction period, for example a period of time over which excitation of the motor occurs. The excitation timing parameter may comprise a duty cycle, for example a ratio defining how often excitation occurs over a cycle. The excitation timing parameter may comprise an advance angle and a conduction period, or may comprise an advance angle and a duty cycle. The excitation timing parameter may comprise a sine wave amplitude, or other waveform amplitude.

The first correction factor may comprise a first advance angle correction factor and/or a first conduction period correction factor and/or a first duty cycle correction factor. The second correction factor may comprise a second advance angle correction factor and/or a second conduction period correction factor and/or a second duty cycle correction factor. The method may comprise applying a first advance angle correction factor to an advance angle and/or applying a first conduction period correction factor to a conduction period and/or applying a first duty cycle correction factor to a duty cycle. The method may comprise applying a second advance angle correction factor to an advance angle and/or applying a second conduction period correction factor to a conduction period and/or applying a second duty cycle correction factor to a duty cycle. The first advance angle correction factor may be proportional to the first conduction period correction factor and/or to the first duty cycle correction factor. The second advance angle correction factor may be proportional to the second conduction period correction factor and/or the second duty cycle correction factor.

The first correction factor may increase an advance angle and/or increase a conduction angle and/or increase a duty cycle. By increasing an advance angle the measured power may increase, as current may be driven into the phase winding at an earlier point in time. By increasing a conduction angle the measured power may increase, as current may be driven into the phase winding over a longer period of time. By increasing a duty cycle the measured power may increase, as current may be driven into the phase winding for a greater percentage of a cycle. The first correction factor may decrease an advance angle and/or decrease a conduction angle and/or decrease a duty cycle. By decreasing an advance angle the measured power may decrease, as current may be driven into the phase winding at a later point in time. By decreasing a conduction angle the measured power may decrease, as current may be driven into the phase winding over a shorter period in time. By decreasing a duty cycle the measured power may decrease, as current may be driven into the phase winding for a smaller percentage of a cycle.

The second correction factor may increase and/or decrease an advance and/or conduction angle and/or a duty cycle.

The target input power may comprise a power set by a user. For example the target input power may comprise a power set by toggling of a switch by a user.

The method may comprise operating the brushless permanent magnet motor using an initial excitation timing parameter relationship in response to the comparison of the measured input power to the target input power, for example prior to applying the first correction factor. This may be beneficial as it may move the measured input power closer to the target input power before application of the first correction factor, and hence may avoid the need for large correction factors which may cause unstable and/or inefficient operation. Applying the first correction factor to an excitation timing parameter of the motor may comprise applying the first correction factor to an excitation timing parameter of the initial excitation timing parameter relationship. The initial excitation timing parameter relationship may comprise a relationship between at least one excitation timing parameter and an input power, for example which excitation timing parameter, or combination of excitation timing parameters, achieve which input power. The method may comprise a first closed control loop for applying the first correction factor, and a second closed control loop for applying the second correction factor. The method may comprise a first closed control loop for ensuring the measured input power substantially matches the target input power. The method may comprise a second closed control loop for altering the speed of the motor. The method may comprise a first closed control loop for altering the measured input power through the motor, and a second closed control loop for altering the speed of the motor. The first closed control loop may comprise an outer control loop, and the second closed control loop may comprise an inner control loop. The first closed control loop may operate on a quicker timescale than the second closed control loop. This may be beneficial as the first closed control loop may act quickly to ensure that the measured input power substantially matches the target input power, whilst the second closed control loop may then correct to achieve the most efficient operating point for the given measured input power.

The method may comprise controlling a brushless permanent magnet motor operating in a steady state. The method may comprise controlling a brushless permanent magnet motor operating at a speed in excess of 20krpm. The method may comprise controlling a single phase brushless permanent magnet motor.

Applying the second correction factor may comprise making a first measurement of the speed of the motor, applying a first polarity correction factor to the excitation timing parameter of the motor, making a second measurement of the speed of the motor and: i) if the second measurement is greater than the first measurement, continuing to apply the first polarity correction factor and subsequently measuring the speed of the motor until the speed of the motor decreases; ii) if the second measurement is less than the first measurement, applying a second polarity correction factor having a polarity opposite to the first polarity correction factor, continuing to apply the second polarity correction factor and subsequently measuring the speed of the motor until the speed of the motor decreases.

This may be advantageous as application of the first correction factor to the excitation timing parameter may ensure that the motor is operating at the target input power, whilst application of the first or second polarity correction factors may ensure that the motor is operating at or near its peak operating speed for the given measured input power, thereby ensuring that the motor is operating at or near its most efficient operating point for a given measured input power.

Where the second measurement is greater than the first measurement, this may indicate that the motor has moved closer to its most efficient operating point for the given power. Continuing to apply the first polarity correction factor and subsequently measuring the speed of the motor until the speed of the motor decreases may indicate that the motor is operating at or near its peak speed for a given power. Where the second measurement is less than the first measurement, applying a second polarity correction factor having a polarity opposite to the first polarity correction factor may ensure that the speed of the operating motor increases, and that the motor has moved closer to its most efficient operating point for the given power. Continuing to apply the second polarity correction factor and subsequently measuring the speed of the motor until the speed of the motor decreases may indicate that the motor is operating at or near its peak speed for a given power.

Applying the second correction factor may comprise applying a positive polarity correction factor to the excitation timing parameter, making a first measurement of the speed of the motor, applying a negative polarity correction factor to the excitation timing parameter, making a second measurement of the speed of the motor; and i) if the second measurement is greater than the first measurement, continuing to apply the negative polarity correction factor and subsequently measuring the speed of the motor until the speed of the motor decreases; ii) if the second measurement is less than the first measurement, reapplying the positive polarity correction factor, continuing to apply the positive polarity correction factor and subsequently measuring the speed of the motor until the speed of the motor decreases.

This may be advantageous as application of the first correction factor to the excitation timing parameter may ensure that the motor is operating at the target input power, whilst application of the positive polarity correction factor or the negative polarity correction factor may ensure that the motor is operating at or near its peak operating speed for the given measured input power, thereby ensuring that the motor is operating at or near its most efficient operating point for a given measured input power.

Measuring the speed of the motor may comprise measuring the speed of a rotor of the motor, for example by utilising a position sensor of the motor or an estimated position of the rotor.

The value of the second correction factor may be chosen to be small enough to prevent significant overshoot past the peak operating speed of the motor for the given measured input power. This may ensure that the motor is operating at or near its most efficient operating point for a given measured input power at the operating point. In a closed loop system, the method may comprise substantially continuously altering the speed of the motor, for example such that the speed of the motor oscillates about the peak operating speed. In such a manner a controller of the motor may substantially continuously hunt the peak operating speed of the motor for the given input power.

According to a further aspect of the present invention there is provided a data carrier comprising machine readable instructions for the operation of one or more processors of a controller of a brushless permanent magnet motor to measure an input power of a the motor, compare the measured input power to a target input power, apply a first correction factor to an excitation timing parameter of the motor to substantially match the measured input power to the target input power, apply a second correction factor to the excitation timing parameter to alter the speed of the motor, and excite motor in accordance with the excitation timing parameter.

According to a further aspect of the present invention there is provided a brushless permanent magnet motor comprising a controller configured to measure an input power of the motor, compare the measured input power to a target input power, apply a first correction factor to an excitation timing parameter of the motor to substantially match the measured input power to the target input power, apply a second correction factor to the excitation timing parameter to alter the speed of the motor, and excite the motor in accordance with the excitation timing parameter.

The motor may comprise a current sensor for measuring a power supply current. The current sensor may be located between a power supply and an inverter of the motor when the motor is connected to the power supply. This may be beneficial as it may provide for a more accurate measurement of the input current, for example compared to a current sensor located between the inverter and a zero-volt rail. The current sensor may be located on a high-side rail.

Preferential features of aspects of the present invention may be equally applied to other aspects of the present invention, where appropriate.

BRIEF DESCRIPTION OF THE DRAWINGS In order to better understand the present invention, and to show more clearly how the invention may be put into effect, the invention will now be described, by way of example, with reference to the following drawings:

Figure 1 is a block diagram of a motor system in accordance with the present invention;

Figure 2 is a schematic diagram of the motor system of Figure 1 ;

Figure 3 details allowed states of an inverter of the motor system of Figure 1 in response to control signals issued by a controller of the motor system;

Figure 4 is a first schematic block diagram illustrating a method according to the present invention;

Figure 5 is a second schematic block diagram illustrating an inner control loop of the method of Figure 4;

Figure 6 is a schematic chart illustrating the variation of speed in response to an applied correction factor in accordance with the present invention;

Figure 7 is a schematic diagram of an alternative embodiment of a motor system according to the present invention;

Figure 8 is a schematic block diagram illustrating a further method according to the present invention.

DETAILED DESCRIPTION The motor system 10 of Figures 1 and 2 is powered by a DC power supply 12 and comprises a brushless motor 14 and a control circuit 16.

The motor 14 comprises a four-pole permanent-magnet rotor 18 that rotates relative to a four-pole stator 20. Although depicted here as a four-pole embodiment, it will be recognised by a person skilled in the art that the teaching of the present application is applicable to other embodiments, for example an eight-pole embodiment, of brushless permanent magnet motors. Conductive wires wound about the stator 20 are coupled together to form a single phase winding 22. Whilst described here as a single phase motor, it will be recognised by a person skilled in the art that the teachings of the present application may also be applicable to multi-phase, for example three phase, motors.

The control circuit 16 comprises a filter 24, an inverter 26, a gate driver module 28, a current sensor 30, a voltage sensor 32, a position sensor 34, and a controller 36.

The filter 24 comprises a link capacitor C1 that smoothes the relatively high- frequency ripple that arises from switching of the inverter 26.

The inverter 26 comprises a full bridge of four power switches Q1 -Q4 that couple the phase winding 22 to the voltage rails. Each of the switches Q1 -Q4 includes a freewheel diode.

The gate driver module 28 drives the opening and closing of the switches Q1 -Q4 in response to control signals received from the controller 36.

The current sensor 30 comprises a shunt resistor R1 located between the inverter and the zero-volt rail. The voltage across the current sensor 30 provides a measure of the current in the phase winding 22 when connected to the power supply 12. The voltage across the current sensor 30 is output to the controller 36 as signal, l_SENSE.

The voltage sensor 32 comprises a potential divider R2,R3 located between the DC voltage rail and the zero volt rail. The voltage sensor outputs a signal, V_DC, to the controller 36 that represents a scaled-down measure of the supply voltage provided by the power supply 12.

The position sensor 34 comprises a Hall-effect sensor located in a slot opening of the stator 20. The sensor 34 outputs a digital signal, HALL, that is logically high or low depending on the direction of magnetic flux through the sensor 34. The HALL signal therefore provides a measure of the angular position of the rotor 18. Embodiments are also envisaged in which the position sensor 34 is omitted, and sensorless control schemes are implemented. Such sensorless control schemes are known, and will not be described here for the sake of brevity. In such sensorless schemes, the HALL signal may be replaced by a BACK_EMF signal, which is representative of the period of the back EMF.

The controller 36 comprises a microcontroller having a processor, a memory device, and a plurality of peripherals (e.g. ADC, comparators, timers etc.). In an alternative embodiment, the controller 36 may comprise a state machine. The memory device stores instructions for execution by the processor, as well as control parameters that are employed by the processor during operation. The controller 36 is responsible for controlling the operation of the motor 14 and generates four control signals S1 -S4 for controlling each of the four power switches Q1 -Q4. The control signals are output to the gate driver module 28, which in response drives the opening and closing of the switches Q1 -Q4.

Figure 3 summarises the allowed states of the switches Q1 -Q4 in response to the control signals S1 -S4 output by the controller 36. Hereafter, the terms‘set’ and ‘clear’ will be used to indicate that a signal has been pulled logically high and low respectively. As can be seen from Figure 3, the controller 36 sets S1 and S4, and clears S2 and S3 in order to excite the phase winding 22 from left to right. Conversely, the controller 36 sets S2 and S3, and clears S1 and S4 in order to excite the phase winding 22 from right to left. The controller 36 clears S1 and S3, and sets S2 and S4 in order to freewheel the phase winding 22. Freewheeling enables current in the phase winding 22 to re-circulate around the low-side loop of the inverter 26. In the present embodiment, the power switches Q1 -Q4 are capable of conducting in both directions. Accordingly, the controller 36 closes both low-side switches Q2,Q4 during freewheeling such that current flows through the switches Q2,Q4 rather than the less efficient diodes. Conceivably, the inverter 26 may comprise power switches that conduct in a single direction only. In this instance, the controller 36 would clear S1 , S2 and S3, and set S4 so as to freewheel the phase winding 22 from left to right. The controller 36 would then clear S1 , S3 and S4, and set S2 in order to freewheel the phase winding 22 from right to left. Current in the low-side loop of the inverter 26 then flows down through the closed low-side switch (e.g. Q4) and up through the diode of the open low-side switch (e.g. Q2).

Where the speed of the rotor 18 is above a pre-determ ined threshold, for example above 60krpm, the controller 36 operates in a steady-state mode. The speed of the rotor 18 is determined from the interval between successive edges of the FIALL signal, which will hereafter be referred to as the FIALL period.

The controller 36 commutates the phase winding 22 in response to edges of the FIALL signal. Each HALL edge corresponds to a change in the polarity of the rotor 18, and thus a change in the polarity of the back EMF induced in the phase winding 22. More particularly, each HALL edge corresponds to a zero-crossing in the back EMF. Commutation involves reversing the direction of current through the phase winding 22. Consequently, if current is flowing through the phase winding 22 in a direction from left to right, commutation involves exiting the winding from right to left.

The controller 36 may advance, synchronise or retard commutation relative to the HALL edges, and hereafter the phase angle at which the controller 36 commutates the phase winding 22 will be referred to as the advance angle, irrespective of whether commutation is advanced, synchronised or retarded. The period over which current is driven into the phase winding 22 is hereafter referred to as the conduction period, and the controller 36 may vary the conduction period to obtain desired operating characteristics.

A method according to the present invention will now be described with reference to Figures 4 to 6.

During operation of the motor system 10, it may be desired to drive a target input power into the phase winding 22. To ensure that the target input power is driven into the phase winding 22, the controller 36 uses the signal l_SENSE received from the current sensor 30, and the signal V_DC received from the voltage sensor 32, to calculate 102 a measured power P_SENSE in accordance with the equation P_SENSE=V_DC. I_SENSE. The measured power P_SENSE is compared 104 to the target input power, for example by a comparator or the like. If the measured power P_SENSE does not equal the target input power, the controller 36 operates the motor 14 using an initial excitation timing parameter relationship, ie an initial combination of advance and conduction angles, to move the measured power P_SENSE closer to the target input power.

The controller 36 then alters 106 the value of the advance angle used to commutate the phase winding 22, and the conduction period over which phase current is driven into the phase winding 22, to attempt to correct any remaining error between the measured input power P_SENSE and the target input power, and communicates the new advance angle and conduction period to the gate drive module 28 by way of signals S1 -S4. The gate drive module 28 then operates 108 switches Q1 -Q4 accordingly to commutate the phase winding 22 at the corrected advance angle, and to provide phase current to the phase winding 22 for the corrected conduction period. The signals l_SENSE and V_DC are again used to calculate an updated measured power P_SENSE, with correction of the advance angle and conduction period occurring if needed.

Thus the correction of the measured power P_SENSE to correspond to the target input power to be driven into the phase winding 22 forms a closed outer feedback loop 1 10 which acts to ensure that the motor 14 is operating at a desired input power. The use of the closed outer feedback loop 1 10 may ensure that the motor 14 operates at a substantially constant input power. The closed outer feedback loop 1 10 acts at a relatively fast rate to ensure a desired input power is achieved.

For a motor operating at a certain input power, the most efficient operating point for that motor is when the motor is operating at its highest speed for that power. To ensure that this occurs, a closed inner feedback loop 1 12 is provided. The closed inner feedback loop 1 12 is illustrated schematically in Figure 5. The closed inner feedback loop 1 12 alters the advance angle and conduction period such that a maximum operating speed is obtained for the motor 14 at the desired constant input power, therefore ensuring peak operating efficiency for the given input power.

In a presently preferred embodiment, the closed inner feedback loop 1 12 initially measures 1 14 the speed of the rotor 18 once the motor 14 is determined to be operating at a constant input power by the closed outer feedback loop 1 10, and the measured speed is stored by the controller 36. A first polarity correction factor, for example plus or minus a specified number of degrees, is applied 1 16 to the advance and conduction angles, before the speed of the motor 14 is measured 1 14 again.

If the speed of the motor 14 has increased between the initial measurement and the measurement after the first polarity correction factor is applied 1 16, then the controller 36 continues to apply 1 16 the first polarity correction factor and subsequently measure 1 14 the speed of the motor 14 until the speed of the motor 14 decreases. Once the speed of the motor 14 decreases the controller 36 determines that the peak speed, and hence peak operating efficiency, has been passed. The controller 36 then applies 1 18 an opposite polarity correction factor such that the speed of the motor 14 increases, and the opposite polarity correction factor is applied until the speed of the motor 14 decreases again, whereby the first polarity correction factor is re-applied 1 16. This process is repeated such that the controller 36 continuously hunts for the peak speed, and hence the peak operating efficiency, and the motor 14 effectively oscillates about its peak operating efficiency.

If the speed of the motor 14 has decreased between the initial measurement and the measurement after the first polarity correction factor is applied 1 16, a second polarity correction factor having a polarity opposite to that of the first polarity correction factor is applied 120 to the advance and conduction angles, such that the speed of the motor 14 increases. The controller 36 continues to apply 120 the second polarity correction factor and subsequently measure 1 14 the speed of the motor 14 until the speed of the motor 14 decreases. Once the speed of the motor 14 decreases, the controller 36 then re-applies 122 the first polarity correction factor such that the speed of the motor 14 increases, and the first polarity correction factor is applied 122 until the speed of the motor 14 decreases again. This process is repeated such that the controller 36 continuously hunts for the peak operating efficiency, and the motor 14 effectively oscillates about its peak operating efficiency. This ensures that the motor 14 is operating at or near its peak speed for the measured input power, and hence that the motor 14 is operating at or near its most efficient operating point.

In particular, as can be seen from Figure 6, the speed of the motor follows a curve which varies in accordance with the correction factor applied, and the curve has a peak. If the first polarity correction factor causes an increase in speed relative to the initial speed measurement, then it can be inferred that the speed is moving along the curve in Figure 6 toward the peak speed. By continuing to apply the first polarity correction factor, the speed of the motor 14 will increase along the curve. Once a decrease in speed is measured, it can be inferred that the speed of the motor 14 has passed its peak, and a second, opposite, polarity correction factor is applied such that the speed of the motor 14 increases along the curve. Once a decrease in speed is measured, it can be inferred that the speed of the motor 14 has again passed its peak, and so the first polarity correction factor is reapplied. In such a manner the motor 14 operates to hunt the peak speed.

If the first polarity correction factor causes a decrease in speed relative to the initial speed measurement, then it can be inferred that the speed is moving along the curve in Figure 6 away from the peak speed. By applying a second polarity correction factor having a polarity opposite to that of the first polarity correction factor, the speed of the motor 14 may be moved along the curve of Figure 6 toward the peak. By continuing to apply the second polarity correction factor, the speed of the motor 14 will increase along the curve. Once a decrease in speed is measured, it can be inferred that the speed of the motor 14 has passed its peak, and so application 120 of the second polarity correction factor is ceased, and the first polarity correction factor is re-applied to hunt the peak, as discussed above. Thus the closed inner feedback loop 1 12 controls the operating speed of the motor 14 such that the motor 14 operates substantially at or near to its peak operating speed, and hence its peak efficiency, for a given measured input power. The value of the correction factors utilised in the closed inner feedback loop 1 12 are sufficiently small that when application of the correction factors are ceased (ie the speed of the motor decreases after steadily increasing), the speed of the motor is still sufficiently close to the peak speed that the speed can be said to be at or near the peak speed.

The use of the closed outer control loop 1 10 and the closed inner control loop 1 12 may remove or reduce the need for the controller 36 to utilise lookup tables for the advance and conduction angles necessary to obtain a desired operating power at peak efficiency, and may allow for real time correction of the operating characteristics of the motor 14. This may remove or reduce the need for end-of- line calibration of the motor 14, thereby saving time and cost.

It will be recognised that alternative peak hunting algorithms and methods may be employed for the closed inner control loop 1 12 to find the peak operating speed, and hence the peak operating efficiency. For example, a positive polarity correction factor may be applied to the advance and conduction angles, before making a first measurement of the speed of the motor. A negative polarity correction factor may then be applied to the advance and conduction angles, before making a second measurement of the speed of the motor. If the second measurement is greater than the first measurement, the negative polarity correction factor continues to be applied with subsequent measurements of the speed of the motor until the speed of the motor decreases. If the second measurement is less than the first measurement, the positive polarity correction factor is reapplied with subsequent measurements of the speed of the motor until the speed of the motor decreases. In such a manner the closed inner feedback loop 1 12 may control the operating speed of the motor 14 such that the motor 14 operates substantially at or near to its peak operating speed, and hence its peak efficiency.

Whilst the method described above applies correction factors to the advance and conduction angles, it will be recognised that this is dependent on the type of motor being controlled, and that alternatively the method may comprise applying correction factors to an advance angle and a duty cycle, for example,

In an alternative embodiment, shown schematically in Figure 7, the current sensor 30 comprises a current sense resistor located on the high-side rail between the power supply 12 and the filter 24. The voltage across the current sensor 30 provides a measure of the input current when connected to the power supply 12. The voltage across the current sensor 30 is output to the controller 36 as signal, l_SENSE, as per the previously described embodiment.

Locating the current sensor 30 on the high-side rail between the power supply 12 and the filter 24, prior to the inverter 26, may provide a more accurate measure of the input current, as switching losses do not need to be accounted for, and hence allow for more accurate calculation of the measured input power.

Whilst the current sensor 30 is shown in Figure 7 as being on the high-side rail between the power supply 12 and the filter 24, it is also envisaged that the current sensor 30 may be located on the low-side rail between the power supply 12 and the filter 24, whilst still allowing for a more accurate power measurement than if located in the position shown in Figure 2.

As mentioned above, locating the current sensor 30 between the power supply 12 and the filter 24 may provide for a more accurate power measurement. However, removing the current sensor 30 from the position shown in Figure 2 means that the current sensor 30 can no longer be used to provide a measure of instantaneous current in the phase winding 22. A measure of instantaneous current in the phase winding 22 may be essential to operation of the motor 10, and may be used to prevent overcurrent events and hence failure of the motor 10.

Accordingly, the inventors of the present application have devised a method of measuring the instantaneous current in the phase winding 22, namely by providing a measure of the instantaneous current through the low side power switch Q4. Although described here with reference to the low side power switch Q4, it will be recognised that the method that follows may be used to provide a measure of the instantaneous current through any of the power switches Q1 -Q4.

The low side power switch Q4 is an n-channel MOSFET, and when current flows through the power switch Q4, the power switch has a drain-source on resistance RDS(ON). The inventors of the present application have determined that RDS(ON) can be estimated using the following equation:

_D Mean Phase V

RDS(ON) _|\/_|ean□ø Q_urrent where Mean Phase V is the mean phase current supplied to the winding 22, and Mean DC current is the mean DC current measured at the power supply, for example using current sensor 30 located on the high-side rail between the power supply 12 and the filter 24. The above equation assumes that the Mean DC Current is approximately equal to the mean phase current in the winding 22. By using values Mean Phase V and Mean DC Current, the value of R_DS(ON) varies in real time, and so variation of R_DS(ON) with temperature is accounted for. The mean phase voltage across the power switch Q4 can be estimated using the following equation:

V Limit

Mean Phase V « —

Duty where V Limit is the VDS (drain to source voltage) of the power switch Q4 which is used to set a current limit, and Duty is the proportion of time over which current is supplied to the phase winding 22.

V Limit will be known by the controller 36, as will the duty, and hence Mean Phase

V will be known by the controller 36. The current sensor 30 can be used to measure the Mean DC Current, and hence RDS(ON) can be estimated.

The voltage drop across the power switch Q4, VDS can be measured, for example using existing VDS sensing circuitry. VDS and RDS(ON) can then be utilised to calculated the instantaneous current through the power switch Q4 in accordance with the following equation:

IQ4, the instantaneous current through the power switch Q4, can then be used by the controller 36 during operation of the brushless motor 14 to determine whether or not the current flowing through the power switch Q4 exceeds a pre-determ ined threshold, set by V Limit, and enables the controller 36 to act accordingly, for example by opening switch Q1 such that current freewheels through the low side of the circuit.

The method 200 for calculating IQ4 is shown in the flow chart of Figure 8. The method 200 includes setting 202 V Limit, the maximum voltage allowed across the power switch Q4 (which corresponds to the maximum current allowed through the power switch Q4), and this is typically set by the control algorithm depending upon the mode of operation of the brushless motor 14. The duty of the current supplied to the motor is measured 204 by the controller 36, and the duty and V Limit are used to calculate 206 Mean Phase V. The Mean DC Current is measured 208 by the current sensor 30, before being fed to the controller 36 as signal l_SENSE.

The controller 36 uses the Mean Phase V and the Mean DC Current to estimate 21 0 RDS(ON) of the power switch Q4. VDS, the voltage drop across the power switch Q4, VDS is measured 212, for example using existing VDS sensing circuitry. RDS(ON) and VDS are then used by the control algorithm to calculate 214 IQ4.

Thus the method 200 may be used to calculate the instantaneous current through the power switch Q4, and the instantaneous current may then be used, for example, to prevent an overcurrent event. Additionally or alternatively, the instantaneous current may be used to determine a position of the rotor 18 in the absence of a position sensor 34, for example as disclosed in WO2013/132249. The method 200 may further be used to estimate the instantaneous current flowing through any of the power switches Q1 -Q4 during freewheeling, whilst including only a single shunt resistor in the circuit, thereby obtaining increased functionality for little to no additional cost.

Claims

1 . A method of controlling a brushless permanent magnet motor, the method comprising measuring an input power of the motor, comparing the measured input power to a target input power, applying a first correction factor to an excitation timing parameter of the motor to match the measured input power to the target input power, applying a second correction factor to the excitation timing parameter to alter the speed of the motor, and exciting the motor in accordance with the excitation timing parameter.

2. A method as claimed in Claim 1 , wherein the excitation timing parameter comprises an advance angle and/or a conduction period and/or a duty cycle.

3. A method as claimed in Claim 1 or Claim 2, wherein the brushless permanent magnet motor comprises a single phase motor.

4. A method as claimed in any preceding claim, wherein applying the second correction factor comprises making a first measurement of the speed of the motor, applying a correction factor having a first polarity to the excitation timing parameter of the motor, making a second measurement of the speed of the motor and: i) if the second measurement is greater than the first measurement, continuing to apply the correction factor having a first polarity and subsequently measuring the speed of the motor until the speed of the motor decreases; ii) if the second measurement is less than the first measurement, applying second polarity correction factor having a polarity opposite to the first polarity correction factor, continuing to apply the second polarity correction factor and subsequently measuring the speed of the motor until the speed of the motor decreases.

5. A method as claimed in any of Claims 1 to 3, wherein applying the second correction factor comprises applying a positive polarity correction factor to the excitation timing parameter, making a first measurement of the speed of the motor, applying a negative polarity correction factor to the excitation timing parameter, making a second measurement of the speed of the motor; and i) if the second measurement is greater than the first measurement, continuing to apply the negative polarity correction factor and subsequently measuring the speed of the motor until the speed of the motor decreases; ii) if the second measurement is less than the first measurement, reapplying the positive polarity correction factor, continuing to apply the positive polarity correction factor and subsequently measuring the speed of the motor until the speed of the motor decreases.

6. A method as claimed in any preceding claim, wherein the method comprises a first closed control loop for applying the first correction factor, and a second closed control loop for applying the second correction factor, the first closed control loop comprising an outer control loop and the second closed control loop comprising an inner control loop.

7. A data carrier comprising machine readable instructions for the operation of one or more processors of a controller of a brushless permanent magnet motor to measure an input power of the motor, compare the measured input power to a target input power, apply a first correction factor to an excitation timing parameter of the motor to match the measured input power to the target input power, apply a second correction factor to the excitation timing parameter to alter the speed of the motor, and excite the motor in accordance with the excitation timing parameter.

8. A brushless permanent magnet motor comprising a controller configured to measure an input power of the motor, compare the measured input power to a target input power, apply a first correction factor to an excitation timing parameter of the motor to match the measured input power to the target input power, apply a second correction factor to the excitation timing parameter to alter the speed of the motor, and excite the motor in accordance with the excitation timing parameter.

9. A brushless permanent magnet motor as claimed in Claim 8, wherein the motor comprises a voltage sensor for measuring a power supply voltage and a current sensor for measuring a power supply current, being located between a power supply and an inverter of the motor when the motor is connected to the power supply.