WO2020070167A1

WO2020070167A1 - Multi-phase, electrically isolated dc-to-dc converter

Info

Publication number: WO2020070167A1
Application number: PCT/EP2019/076657
Authority: WO
Inventors: Florian KRISMER; Johann Walter Kolar; Julian Marius BÖHLER
Original assignee: Egston Power Electronics Gmbh
Priority date: 2018-10-04
Filing date: 2019-10-01
Publication date: 2020-04-09

Abstract

A multi-phase, electrically isolated DC-to-DC converter (10) comprises an HF inverter (1), an HF transformer assembly (3), and m HF rectifiers (5). A controller uses at least the following control parameters, which are adaptable during operation, wherein 1 ≤ kb≤ m: d_k _,p duty factor of the control signal of the kth half-bridge of the HF inverter; d_a _k _,s duty factor of the control signal of the first half-bridge of the kth HF rectifier; d_b _k _,s duty factor of the control signal of the second half-bridge of the kth HF rectifier; α _k phase angle between the rising flank of the control signal of the first half-bridge of the HF inverter and the kth half-bridge of the HF inverter; β _a _k phase angle between the rising flank of the control signal of the first half-bridge of the HF inverter (s _1,p) and the rising flank of the control signal of the first half-bridge of the kth HF rectifier (s _a _k, _s); β _b _k phase angle between the rising flank of the control signal s _1,p and the rising flank of the control signal of the second half-bridge of the kth HF rectifier (s _b _k _,s).

Description

Multi-phase, isolated DC converter

The invention relates to the field of electronic converters, in particular a floating DC voltage converter for generating a plurality of floating DC voltages.

background

For an internal power supply of a power hardware ln-the-loop (P-HIL) test system, a certain number of internal DC voltage connections have to be supplied with energy, whereby each internal DC voltage connection has a plus and a minus connection, both of which have a reference potential (= Earth potential) and must be galvanically isolated from the plus and minus connections of all other internal DC voltage connections. In addition, not only energy consumption, but also energy feed-in can occur at every internal DC connection. The number of internal DC voltage connections to be supplied is referred to below as m, m> 1.

In the simplest case, m DC converters, each with a potential-isolated DC voltage connection, supply the m system-internal DC voltage connections, i.e. for 1 <k <m: the plus connection of the c-th electrically isolated DC voltage connection is connected to the plus connection of the / c-th system-internal DC voltage connection and the minus connection of the / c-th electrically isolated DC voltage connection is connected to the minus connection of the kth internal DC voltage connection. The DC voltage converters, each with a potential-isolated DC voltage connection, perform three functions: o the stabilization of the voltages present between the plus and minus connections of the isolated DC voltage connections,

o the electrical isolation between a power supply network supplying the test system and the electrically isolated DC voltage connections and the electrical isolation between the m electrically isolated DC voltage connections and

o bidirectional power operation, i.e. Feeding and regenerating electrical energy, at all m isolated DC connections.

For reasons of economy, efficiency and / or the required construction volume, it may be sensible to replace the m DC voltage converters, each with a potential-isolated DC voltage output, by a multi-phase DC voltage converter, as shown in FIG. 1. Such DC converters are known for three phases. The multi-phase DC voltage converter (10) has a non-electrically isolated DC voltage connection on the primary side and m electrically isolated DC voltage connections on the secondary side. Between the non-isolated DC voltage connection and the isolated DC voltage connections, the multi-phase DC voltage converter has a bidirectional high-frequency inverter circuit (RF inverter) 1, m high-frequency transformers (RF transformers) 3, m primary-side high-frequency networks (primary-side RF networks), secondary RF (secondary networks) Networks)

4 and m bidirectional high-frequency rectifier circuits (HF rectifiers) 5 on: o The HF inverter consists of m half bridges 11, each with two circuit breakers, each with the parallel connection of an actual switching element and an anti-parallel diode (if, for example, a MOSFET as a circuit breaker is used, the anti-parallel diode is already present due to the internal semiconductor structure of the MOSFET). The two circuit breakers of each half-bridge are connected in series so that the anode of the anti-parallel diode in the first circuit breaker is connected to the cathode of the anti-parallel diode in the second circuit breaker. This series connection thus provides three connections: a switching voltage connection of the half bridge, a plus connection of the half bridge and a minus connection of the half bridge. The switching voltage connection of the half bridge is between the series circuit breakers, the plus connection of the half bridge is at the cathode of the anti-parallel diode of the first circuit breaker and the minus connection of the half bridge is at the anode of the anti-parallel diode of the second circuit breaker. The plus connections of all m half bridges of the HF inverter are connected to the plus connection of the non-isolated DC voltage connection. The minus connections of all m half bridges of the HF inverter are connected to the minus connection of the non-isolated DC voltage connection. The HF inverter has at least one DC link capacitor on the input side, having a plus and a minus connection. The positive connection of the DC link capacitor on the input side is connected to the positive connection of the non-isolated DC voltage connection and the negative connection of the DC link capacitor on the input side is connected to the negative connection of the non-isolated DC voltage connection. The voltage Up is present between the plus and minus connections of the non-electrically isolated direct voltage connection.

Each of the m HF rectifiers consists of two half bridges 51, 52, which are interconnected as full bridges as follows: the plus connections of the two half bridges of the / c-th HF rectifier, 1 <k <m, are connected to the plus connection of the k th isolated DC connection. The minus connections of the two half bridges 51, 52 of the k-th HF rectifier, 1 <k <m, are connected to the minus connection of the k-th isolated DC voltage connection. Each HF rectifier has at least one intermediate circuit capacitor on the output side, having a plus and a minus connection. The plus connection of the output-side DC link capacitor of the k-th HF rectifier is connected to the plus connection of the k-th electrically isolated DC voltage connection, and the minus connection of the output-side DC link capacitor of the k-th HF rectifier is connected to the minus connection of the k-th isolated DC voltage connection. For 1 <k <m the following applies: An output current of the k-th secondary-side RF rectifier is defined as the mean value of the current rectified by the k-th secondary-side RF rectifier, which is used at its electrically isolated DC voltage connection, e.g. to supply a system-internal DC voltage connection or to charge the output side connected there Gen DC link capacitor is available. The voltage Uk, s is present between the plus and minus connections of the th-isolated DC voltage connection.

Each of the m HF transformers has a primary-side winding, having two primary-side transformer connections, and a secondary-side winding, having two secondary-side transformer connections. The winding senses of the primary and secondary windings are defined as follows: If a positive voltage is applied from the first to the second primary transformer connection, then a positive voltage also results from the first to the second secondary transformer connection. For 1 <k <m the following applies: each HF transformer realizes a transformation ratio rik, which is defined as the ratio of the number of turns of the primary-side winding to the number of turns of the secondary-side winding: m _f = Nk, p / Mc, s. To increase the magnetic coupling between the primary and secondary windings, each HF transformer can have a magnetic core

The primary and secondary RF networks are networks with passive components, e.g. Inductors and / or capacitors. Each of the primary and secondary RF networks has a first and a second input terminal and a first and a second output terminal. Examples of practically frequently used RF networks are:

Each primary RF network: a capacitor connects the first input terminal to the first output terminal; the second input terminal is connected directly to the second output terminal. In a practical implementation, the capacitor serves to keep DC voltage components possibly generated by the circuit breakers away from the connected HF transformers. This RF network is referred to as an RF capacitor network.

^■ Each secondary-side RF network: a series resonant circuit consisting of coil and capacitor connects the first input terminal to the first output terminal; the second input terminal is connected directly to the second output terminal. The series resonant circuit is used to adapt the portable power. In practical implementation, the capacitor also serves to keep any DC voltage components generated by the circuit breakers away from the connected HF transformer. This RF network is referred to as an RF series resonant circuit network. In the simplest case, the capacitance of the capacitor of the series resonant circuit is chosen so large that this capacitor only keeps the DC voltage components away from the connected HF transformer, but is not effective for adapting the transmissible power. The voltage and current profiles shown in the exemplary embodiments are based on this case.

* The HF networks can be fully or partially integrated into the HF transformers (for example, the leakage inductance of the k-th HF transformer can be part or all of the inductance of the coil of the series resonant circuit of the k-th secondary-side HF network realize).

RF inverters, primary-side RF networks, RF transformers, secondary-side RF networks and RF rectifiers are connected as follows. The first primary-side transformer connection of the first H FT transformer is connected to the first output terminal of the first primary-side HF network, the first input terminal of the first primary-side HF network is connected to the switching voltage connection of the first half bridge of the HF inverter, the second primary-side Transformer connection of the first RF transformer is connected to the second output terminal of the first primary-side RF network and the second input terminal of the first primary-side RF network is connected to the first input terminal of the second primary-side RF network. This type of connection is continued for the remaining RF transformers and RF networks in this sense, ie for 1 <k <m the following applies: the first transformer connection of the kth RF transformer on the primary side is the one with the first output terminal of the k-th primary-side RF network, the first input terminal of the k-th primary-side RF network is connected to the switching voltage connection of the k-th half-bridge of the RF inverter and the second primary-side transformer connection of the k-th HF-T transformer is connected to the second output terminal of the k-th primary-side RF network. For 1 <k <m, the following applies: the second input terminal of the k-th primary-side RF network is connected to the first input terminal of the (k + 1) -th primary-side RF network. The second input terminal of the m th primary-side RF network is connected to the first input terminal of the first primary-side RF network. The following applies for 1 <k <m: the first secondary-side transformer connection of the kth HF transformer is connected to the first input terminal of the k-th secondary-side RF network connected, the second secondary-side transformer connection of the Zi-th RF transformer is connected to the second input terminal of the Zc-th secondary-side RF network, the first output terminal of the Zc-th secondary-side RF network is connected to the switching voltage connection of the first half-bridge of the Zc-th HF rectifier and the second output terminal of the Zc-th secondary-side HF network is connected to the switching voltage connection of the second half-bridge of the Zc-th HF rectifier.

The regulation of the electrical power delivered or consumed at the isolated DC voltage connections takes place via the suitable control of all half bridges. Each half-bridge is controlled on the one hand via a control signal of the half-bridge and on the other hand via an activation signal of the half-bridge. The control signal of the half bridge denotes a signal for switching the voltage at the switching voltage connection of the half bridge. The control signal of the half-bridge is rectangular and can have the values 0 or 1: 0 means that the switches of the half-bridge are switched so that the switching voltage connection of the half-bridge is connected to the minus connection of the half-bridge and 1 means that the switches of the half-bridge are switched in this way are that the switching voltage connection of the half bridge is connected to the positive connection of the half bridge. The activation signal of the half-bridge designates a signal for activating the circuit-breakers of the half-bridge and can take the values 0 or 1: 0 means that, regardless of the value of the control signal of the half-bridge, all circuit breakers of the assigned half-bridge are switched off and only the anti-parallel diodes of the circuit breakers are effective and 1 means that the control signal of the half-bridge determines the voltage at the switching voltage connection of the half-bridge.

The dead times to be taken into account in a practical implementation when the control signal changes from 0 to 1 and from 1 to 0, in order to avoid short-circuits within the half-bridge, are not dealt with in detail for reasons of comprehensibility of the explanations, since this is to be assumed that the dead times are very short in relation to the period of the control signal and their effects on the resulting voltage and current profiles of the multiphase DC converter under consideration are negligible. In a current publication [4], the concept of the multi-phase DC voltage converter is taken up again in the context of the analysis of resonance converters with HF integrated transformers. In this publication, three different embodiments of a three-phase resonance converter are presented, one of these representations coinciding with the multiphase DC converter with star connection on the primary side and m = 3. However, since the focus of this publication is on FIF-transformers integrated in the printed circuit board, their description is limited to m = 3. The control method used to control the half bridges is not discussed.

State of the art literature

[1] H. Wrede, V. Staudt, A. Steimel, “Design of an Electronic Power Transformer”, 28th Annual Conference of the IEEE Industrial Electronics Society (IECON 2002), Sevilla, Spain, p. 1380-1385.

[2] H. Wrede, V. Staudt, A. Steimel, “A Soft-Switched Dual Active Bridge 3DC-to- 1 DC Converter employed in a High-Voltage Electronic Power Transformer”, 10th European Conference on Power Electronics and Applications ( EPE 2003), Toulouse,

France.

[3] H. Wrede, “Contributions to increasing security of supply and power quality in the transmission and distribution of electrical energy through power electronic equipment”, dissertation, Ruhr University Bochum, 2004.

[4] B. Li, Q. Li, F. C. Lee, “A WBG Based Three Phase 12.5 kW 500 kHz CLLC Resonant Converter with Integrated PCB Winding Transformer”, Proc. IEEE Applied Power Electronics Conference and Exposition (AP EC), San Antonio, TX, USA, March 4-8, 2018, p. 469-475.

Schäfer Jannik et al: “Multi-port multi-cell DC / DC Converter topology for electric vehicles power distribution networks”, 2017 IEEE 18TH WORKSHOP ON CONTROL AND MODELING FOR POWER ELECTRONICS (COMPEL), IEEE, July 9, 2017 a converter with a large number of identical converter modules, on which an overall power flow is divided. Presentation of the invention

A possible object of the present invention is to create a multiphase DC voltage converter which allows a comprehensive use of the possibilities inherent in it.

Another possible object is to create a multi-phase DC converter that improves the operation of the considered multi-phase DC converter.

Another possible task is to create a multi-phase DC voltage converter which, with a constant number of DC voltage connections to be supplied, m achieves a reduction in the required circuit breakers and thus a reduction in circuit complexity.

At least one of these tasks is at least partially solved by the subject matter according to the corresponding patent claims.

The considered multi-phase DC voltage converter has m half bridges of the HF inverter and 2 m half bridges of the HF rectifiers and therefore 3 m control signals, whereby one switching period denotes the period of the control signal of the half bridge and has the same value for the control signals of all half bridges. Overall, there are 9 m degrees of freedom of control for the regulation of the electrical powers present at the electrically isolated DC voltage connections, see FIGS. 2 and 3:

o the switching period Ts,

o 3 m activation signals of the half bridges.

o 3 m duty cycles of the control signals of the half bridges, a duty cycle of the control signal denoting the relative duration of the value 1 of the control signal in relation to the duration of the switching period, and the duty cycle can therefore assume values between 0 and 1. The control signal of the k-th half-bridge of the HF inverter 1 is designated S _k .p, the control signal of the first half-bridge 51 of the k-th HF rectifier 5 is Sak, s and the control signal of the second half-bridge 52 of the k-th HF Rectifier 5 with Sbk.s. The duty cycle of the control signal of the k-th half-bridge of the HF inverter 1 is denoted by d _k .p, the duty cycle of the control signal of the first half-bridge 51 of the k-th HF Rectifier 5 with dak.s and the duty cycle of the control signal of the second half bridge 52 of the / c-th HF rectifier 5 with d «c, s

om - 1 phase relationships between the rising edges of the control signals of the half-bridges of the HF inverter. For 1 <k <m the following applies: each phase relationship is characterized by the phase angle Ok, which is present between the rising edge of the control signal of the first half bridge of the HF inverter 1 and the / c-th half bridge of the HF inverter. The / c-th half-bridge of the HF inverter 1 converts the control signal s _/ c, _{P assigned to it} into one between the switching voltage connection of the / c-th half-bridge of the HF inverter 1 and the minus connection of the / c-th half-bridge of the HF inverter present switching voltage of the / c-th half bridge of the HF inverter 1 according to Uk, p = Up Sk, p um.

o 2 m phase relationships ßak and ßbk for 1 <k <m: ßak denotes the phase angle between the rising edge of the control signal of the first half-bridge of the HF inverter (si, _P ) and the rising edge of the control signal of the first half-bridge 51 of / c-th HF rectifier (Sa _k .s); βb _k denotes the phase angle between the rising edge of the control signal si, _P and the rising edge of the control signal of the second half bridge 52 of the / c-th HF rectifier (Sbk, s). The following applies for 1 <k <m: the first half bridge 51 of the / c-th HF rectifier 5 converts the control signal Sak, s assigned to it into one between the switching voltage and negative connections of the first half bridge 51 of the / c-th HF Rectifier 5 present switching voltage of the first half bridge 51 of the / c-th HF rectifier according to Uak, s = iA, s Sak.s and the second half bridge 52 of the / c-th HF rectifier 5 converts the control signal Sak.s assigned to it into a switching voltage of the second half bridge 52 of the / c-th HF rectifier according to Ub _k .s = U _k .s Sb _k .s between the switching voltage and minus connections of the second half bridge 52 of the kth HF rectifier 5.

The following applies for 1 <k <m: phase angles ak> 0, βak> 0, βbk> 0 characterize a lag in the control signal of the relevant half-bridge in relation to the control signal of the first half-bridge of the HF inverter. Phase angles are given in radians. Due to the structure of the multiphase DC converter under consideration (FIG. 1), for 1 <k <m, the voltage t // _{c, / c} lies between the first and second input terminals of the k-th primary-side RF network _{+ i, P} = U _{k, P} - u _{/ + i, P} an. The voltage u _m, 1, _P = u _{m, p} - ui, _{P is present} between the first and the second input terminal of the m-th HF network. The voltage Uk, s = Uak.s - Ubk, s is present between the first and second secondary-side output terminals of the k-th RF network. With the choice of the 9 m degrees of freedom, the time profiles of the voltages uy _{f + i, Pl} Um, I .P and Uk.s are stamped. The time courses of the voltages Uk, k + i. , u _{m, i, P} and Uk, s, the implementations of the primary and secondary RF networks and the selected transmission ratios of the RF transformers determine the temporal courses of the primary and secondary windings of the RF transformers Currents and, subsequently, the electrical power output or consumed at the isolated DC voltage connections.

So far, the multiphase DC voltage converter on which this invention is based has been described in restricted form in [1, 2, 3]:

o Only a three-phase DC voltage converter was considered, as shown in FIG. 5. This corresponds to the embodiment of the multiphase DC converter for m = 3.

o Descriptions for the operation of the three-phase DC converter

restrict themselves to symmetrical operation, i.e. It is assumed that the same electrical power is present on all isolated DC voltage connections.

o In an in-depth study of the three-phase DC converter in

[2,3] the losses to be expected from various tax procedures are compared with each other, although the tax procedures used are severely restricted with regard to the use of the existing degrees of freedom. With regard to the achievable efficiency, a simplified control procedure is identified as best suited in [2,3], which is characterized as follows:

The activation signals of all half bridges are 1,

• dk, p ⁼ dak, s ⁼ dbk, s ⁼ 50%,

0f _{/ <} = k 2TT / 3,

Ssak = CT _f c + <p / (2), Ssb _k = a _k + (2p / 3) + [2 - L / 2] f.

This control method applies to all 1 <k <m = 3. This leaves a single degree of freedom, f, which is used to regulate the electrical powers that are present at all three electrically isolated DC voltage connections.

In a first step, an improvement goal can be defined, e.g. a mathematical optimization criterion (e.g. being the sum of all master losses or the sum of all losses occurring in the multi-phase DC voltage converter) or a practical goal (starting up the converter with limited output current). In a second step, the values for the degrees of freedom of the control required to meet the improvement goal are to be determined. This is typically done by means of a suitable analysis, for example, in the case of a mathematical optimization criterion, an optimization algorithm suitable for the search for a global optimum, which varies the degrees of freedom of the control in such a way that, assuming stationary operation at a given operating point, as a result optimization results in the values of the degrees of freedom of the control required to meet the optimization criterion. The operating point is defined by:

- the voltage applied to the non-isolated DC voltage connection,

- the voltages present at all m isolated DC connections,

- The electrical power available at all m isolated DC connections.

The stationary operation of the multiphase DC converter under consideration is characterized by constant values of all voltages and powers of the operating point.

The multi-phase, isolated DC voltage converter is used to exchange electrical energy between a primary-side DC voltage connection and m secondary-side isolated DC voltage connections. It has the following subunits:

• an RF inverter; connected to it m primary RF networks;

• connected to it m windings on the primary side of an HF transformer arrangement;

• secondary windings of the HF transformer arrangement;

• connected to it m secondary RF networks;

• connected to it m HF rectifier; and a control of the subunits, which is designed to determine control parameters, corresponding to degrees of freedom of the control, and to adapt them to an operating state of the DC voltage converter during operation of the DC voltage converter, wherein the control parameters that can be adjusted during operation include at least one for 1 <k <m: d _k .p Duty cycle of the control signal of the kth half-bridge of the HF inverter; since _k .s duty cycle of the control signal of the first half bridge of the k-th HF rectifier; dbk.s duty cycle of the control signal of the second half bridge of the k-th HF rectifier; ctk phase angle between the rising edge of the control signal of the first half-bridge of the HF inverter and the k-th half-bridge of the HF inverter; ßak phase angle between the rising edge of the control signal of the first half bridge of the HF inverter (si, _P ) and the rising edge of the control signal of the first half bridge of the kth HF rectifier (Sa _f es); ßbk phase angle between the rising edge of the control signal si, _P and the rising edge of the control signal of the second half-bridge of the k-th HF rectifier (Sbk, s). Two subunits connected to each other allow power to flow in one or both directions between the subunits, so “connected” also means “designed to exchange energy”.

In embodiments, the subunits are each designed for a bidirectional power flow.

In embodiments, the control parameters that can be adjusted during operation additionally include at least one of the following two control parameters: Ts (switching period) and activation signals of the half-bridges.

In embodiments, the controller is designed to minimize conduction losses

• to operate all half bridges with a duty cycle of 50% in a stationary operation;

• activate all half bridges;

• to determine the phase angles a _k, ßa _k , ßb _k to at least approximately minimize the optimization criterion f = min (5 _{= 1} /? _Fc / f _P ), where, p is the effective value of the current in the primary winding of the / c-th HF transformer and R _{k represents} a corresponding equivalent resistor.

In embodiments, the DC voltage converter has m = 3 electrically isolated DC voltage connections on the secondary side, the controller being designed, with symmetrical loading of the DC voltage connections on the secondary side, to design the phase angles cn = 0, ai = 2p / 3, az = 4p / 3 set, and to set first phase angle differences 8 _ai - ai = ßai - ai = ßaz - az, and to set second phase angle differences ßm - j8ai = ßbi - ß 2 = ßbz - ßaz, and in particular, the phase angle differences depending on the ratio of the power P. to a maximum power P _m a _x by • for P between zero and P _max / 2 the first phase angle difference rises from zero and falls back to zero, and the second phase angle difference rises monotonically from 2p / 3 to p; and

• for P between Pmax / 2 and Pmax, the first phase angle difference increases monotonically from zero to p / 3, and the second phase angle difference remains constant at p ver.

In embodiments, the phase angles and phase angle differences run according to Table 1.

In embodiments, the DC voltage converter has m = 3 secondary-side isolated DC voltage connections, the controller being designed to provide power Pi, _s , P2, s and Pz.s when the secondary-side DC voltage connections are asymmetrically loaded, and for different voltages i / i, _s , Uz , _s and l / s, s adjust the phase angles and the phase angle differences according to Table 2 on the secondary-side DC voltage connections.

In embodiments, the controller is designed to set the activation signals in an operating mode for starting up the DC voltage converter as follows, where 1 <k <m:

o activation of all half bridges (11) of the HF inverter (1);

o Deactivation of all half bridges (51, 52) of the HF rectifiers (5).

So it applies

o the activation signals of all half bridges of the HF inverter are 1 and o the activation signals of all half bridges of the HF rectifiers are 0, i.e. all HF rectifiers are operated as diode rectifiers and accordingly, ßak and ßbk are naturally set depending on the operating point.

The control method for starting up the multi-phase DC voltage converter with limited output current is based on the fact that the simplified control method and the control method described above for minimal conduction losses during the startup of the multi-phase DC voltage converter due to the charging of the connected to the electrically isolated DC voltage connections. DC link capacitors, very high currents in all power semiconductors and cause in all HF transformers of the multiphase DC converter under consideration.

In embodiments, a simultaneous reduction of the rms values of the currents in all power semiconductors and in all HF transformers can be achieved if, through suitable measures, a substantial reduction in the rms values of the voltages present between the first and second input terminals of all primary-side HF networks down to values is reached close to zero, which can be achieved due to the concatenation of the voltages present between the first and second input terminals of the primary-side RF networks, if the duty cycle of the control signals of the half-bridges of the RF inverter deviate from 50%, ie d _k .p F 50% for 1 <k <m.

The following technically meaningful restriction is also implemented in embodiments:

o ST = _i 4, p <1 or S = _i 4, _P > m - 1,

since a partial decoupling of the individual phases of the multiphase DC converter can also be achieved with this.

In embodiments, a simplification in the control, with simultaneous partial decoupling of the individual phases of the multiphase DC converter, is also achieved with the following condition:

o either 0 <<4 _P <1 I m or (m - 1) / m <<4, _P <1.

In embodiments, the following is also selected: d _{k, P} = di _P , ie the duty cycles of the separate DC voltage connections are the same. This is useful for symmetrical loads.

In embodiments, the pulse duty factor d _{k, p is} adapted to control an output current of a k-th secondary-side HF rectifier.

In embodiments, the controller is designed to set the control parameters as follows in the case of operation with low powers, where 1 <k <m:

o activation of all half-bridges of the HF inverter (1) and the HF rectifier (5), ie all activation signals are 1; o dk, p F 50%.

o Eif _{= l, P} <1 or E? = i _P > m - 1,

since a partial decoupling of the individual phases of the multiphase DC converter can also be achieved with this. A simplification in the control, with simultaneous partial decoupling of the individual phases of the multiphase DC converter, is achieved with the following condition:

o either 0 <d _{k, p} <1 I m or (m - 1) / m <d _k .p <1.

In addition, in combination with one or more of the relationships already mentioned, one or more of the following relationships can be implemented by the control:

o 0 <dk.p + dak, s <1 I m, i.e. in each module the sum of the duty cycles on the primary and secondary side is between zero and 1 I m. This condition represents a simplification for the control, since it prevents a positive voltage between the first and second input terminals of the k-th primary-side RF network and a negative voltage between the first and second output terminals of the k-th secondary-side RF network is present.

o dk, p = di, _P , ie the duty cycles of the separate DC voltage connections are the same. This is useful for symmetrical loads.

In embodiments, the controller is designed to set one or more of the further control parameters when operating at low powers, as follows, where at 1 <k <m:

o dk, p ⁼ i, _P ;

O dak, s ⁼ dbk, s

oc _k = (k - 1) 2tt / irr,

O ßtok - ßak = 2 p / m.

In embodiments, the controller has chosen to set the control parameters according to one of the following three variants in the case of operation with low powers, where 1 <k <m: first variant:

o 0 <i, p + da _k .s <1 I m,

O dak, s ⁼ dbk, s,

o ßbk - ßak = 2 p / m, second variant:

o 0 <c? i, _P + (ß _bk - ßak) / (2 tt) <1 / m,

O dak.s ⁼ dbk, s ⁼ 1 / PΊ, third variant:

o 0 <di, _P + (ßak - jSb _k ) / (2 p) <1 / m,

o dak.s = db / _{<, s} = (m - 1) / m.

In embodiments, the controller is designed to determine the control parameters that can be adjusted during operation in a mode of operation of the multiphase DC voltage converter, referred to as half-cycle dis-continuous conduction mode, hereinafter referred to as HC-DCM, so that at least one control parameter can be determined Element of the primary-side RF networks and / or the secondary-side RF networks forms a current characteristic characteristic of HC-DCM. In particular, this can be the current profile of an input or output current of the relevant RF network.

An operating mode of the multi-phase DC voltage converter, known as half-cycle discontinuous conduction mode (HC-DCM), is used, for example, to ensure that, during operation with a given voltage at the DC voltage connection of the HF converter, largely load-independent voltages at the electrically isolated DC voltage connections of the HF rectifiers to adjust.

When operating the multi-phase DC voltage converter with HC-DCM and when transferring energy from the HF inverter to the HF rectifier, such characteristic curves for operation with HC-DCM can be set at the outputs of one or more of the secondary-side HF networks.

If the direction of energy transmission is reversed in the case of an HF rectifier, that is to say from the HF rectifier to the HF inverter, such characteristic curves for operation with HC-DCM can occur at the input of the HF Rectifier-associated primary RF network can be set. The reversal of the energy transmission direction can affect a single HF rectifier, several HF rectifiers or all HF rectifiers, and thus also assigned individual, several or all primary-side HF networks.

Current curves characteristic for operation with HC-DCM are characterized by alternating positive and negative, essentially sinusoidal, half-oscillations, which begin with vanishing current and end again after the end of each (sine) half-oscillation with vanishing current. Between these (sine) semi-oscillations, the currents (at the outputs of the secondary RF networks or at the inputs of the primary RF networks) remain negligibly small for a certain time, i.e. Between the alternating positive and negative (sine) semi-oscillations used for energy transmission there are time intervals with vanishing current and therefore also vanishing energy transmission.

A vanishing current means a current value that is essentially zero. In particular, it is significantly smaller than an extremum of the same current profile over a half period. For example, the amount is less than a twentieth or less than a fiftieth or less than a hundredth of the maximum current value (a positive half-wave) or the amount of the minimum current value (a negative half-wave) of the half-period.

The time intervals with vanishing current have a duration of, for example, at least one twentieth, in particular at least one tenth, of the duration of a half period of the corresponding current profile.

In embodiments, the controller is designed to set the control parameters that can be adjusted during operation in the HC-DCM, where 1 <k <m:

Activation of all half-bridges of the HF inverter; • Activation of both half bridges in at least one of the HF rectifiers during time intervals in which the HF rectifier has a semi-oscillatory course of its input current, and deactivation of one or in the half bridges of the HF rectifier during time intervals in which the HF rectifier has a vanishing input current The input current of an HF rectifier is defined as a current flowing through the switching voltage connections of the HF rectifier on the AC voltage side;

• Operation, in a stationary operation, of all half bridges with a duty cycle of 50%;

• Setting the phase angle ct _k = (k- 1) 2 tt / m;

• Setting the first phase angle differences ßak - Ok = 0;

• Setting second phase angle differences ßbk - ct _k = TT.

The half-bridges of the HF rectifiers are thus with a duty cycle of 50%; operated, switching operations taking place only when half-bridges are activated.

It is thus possible, without active regulation of the HF rectifiers, to have an essentially load-independent voltage at the isolated DC voltage connections of the HF rectifiers (for energy transmission from the HF inverter to the HF rectifiers) or at the DC voltage connections of the HF inverter (at Energy transfer from the HF rectifiers to the HF inverter).

The input current of an RF rectifier is equal to the output current of the connected secondary RF network. It flows into the switching voltage connection or midpoint connection of one of the half bridges and out of the switching voltage connection of the other half bridge of the HF rectifier.

In embodiments, in the case of energy transmission from the HF inverter to the HF rectifiers, in order to form a current profile characteristic of HC-DCM at the output of at least one of the secondary-side HF networks, some or all of them Switch of at least one HF rectifier connected to it deactivated. In particular, diodes present in parallel with the deactivated switches allow a natural commutation process.

In embodiments, when transmitting energy from the HF rectifiers to the HF inverter, to form a current characteristic characteristic of HC-DCM at the input of at least one of the primary-side FIF networks, individual or all switches of the HF inverter connected to it are deactivated. In particular, diodes present in parallel with the deactivated switches allow a natural commutation process.

A loss reduction by synchronous rectification can thus be realized during this deactivation time.

In embodiments, there are m = 3 electrically isolated DC voltage connections on the secondary side, and the control is designed so that, in HC-DCM operation of the resonance DC voltage converter with a given voltage at the DC voltage connection of the HF inverter, essentially load-independent voltages are established at the electrically isolated DC voltage connections of the HF rectifiers.

In embodiments, the HF rectifiers each have a first half bridge with circuit breakers and a second half bridge with capacitors. In embodiments, the HF transformer arrangement has m individual HF transformers and, in particular, windings of these HF transformers on the primary side form a delta connection or a star connection or can be switched between a star and delta connection. This means that a phase-modular converter can be implemented.

In embodiments, the RF transformers each have two secondary windings, and the secondary windings of all RF transformers form a zigzag circuit. In embodiments, the RF transformer arrangement has a multi-phase RF transformer with m phases. This means that a phase-integrated converter can be implemented.

Brief description of the drawings

The subject matter of the invention is explained in more detail below on the basis of preferred exemplary embodiments which are illustrated in the accompanying drawings. Show it:

Fig. 1: Circuit diagram of a DC voltage supply with several electrically isolated outputs.

2: (a) control signals of the flaling bridges of the HF inverter, (b) with FIG. 2 (a) corresponding voltages between the switching voltage connections of the k-th and (c + 1) -th fialb-bridge of the Fl F change richters (for 1 <k <m) or between the

Switching voltage connections of the m-th and first half-bridge of the HF inverter.

3: (a) control signals of the half-bridges of the k-th HF rectifier, (b) with FIG. 3 (a) corresponding voltage between the switching voltage connections of the first and the second half-bridge of the k-th HF rectifier (for 1 < k <m).

Fig. 4: Example of the formation of the current in the secondary winding of the first RF transformer, k = 1: (a) between the first and second input terminals of the first primary RF network voltage and between the first and second output terminals of the first secondary RF network voltage present; (b) with Fig. 4 (a) corresponding voltage and current profiles in the coil of the first secondary RF network under the assumptions that the primary RF networks are realized by RF capacitor networks and the secondary RF networks by RF series resonant network networks and the capacitance of the capacitors of all RF networks are chosen so large that these capacitors only keep the DC components away from the H FT transformers.

Fig. 5: embodiment of a three-phase DC voltage supply.

Fig. 6: Example of a control characteristic for control according to Wrede [2] for m = 3, depending on the powers Fi, _s = P2, s = Pz, s available at all electrically isolated DC connections and for symmetrical operation using the following Characteristics: T _s = 20 ps, m = m = m = 1, Li = L2 = U = 222 mH, U _P = 700 V, (7i, _s = Lh, s = ü _3, s = 700 V.

Fig. 7: Control characteristics for m = 3 and U _p = ri _k U _k determined with the control method for minimum conduction losses of the multiphase DC voltage converter, depending on the power available at all electrically isolated DC voltage connections and related to the maximum convertible power, P / Pmax , for Pmax =

0.112 Ts U _P ² / L and m ² U = m ² Li = m ² Lz = L

Fig. 8: Comparison of the guide losses resulting with the simplified control method according to [2] (shown in dashed lines) to the guide losses which result with the control method for minimum guide losses of the multiphase DC voltage converter (drawn in solid lines) for the control characteristics shown in Fig. 7: m = 3 and U _P = ri _k LJ _k , depending on the power available at all electrically isolated DC voltage connections and related to the maximum convertible power, P / Pmax, for Pmax = 0.112 Ts U _P ² / L and r? I ² Li = m ² Lz = ² Lz = L

Fig. 9: With the control method for minimal conduction losses of the multi-phase DC converter, control curves for m = 3, depending on the power Pi, _s , 0 <present at the first electrically isolated DC connection

P1, S <0.8 Pmax, for three different powers Pz, s, P2, s / Pmax = (0.2, 0.4, 0.8}, available at the second electrically isolated DC voltage connections, and using the following characteristic data: mi / i, _s / U _P = n 2 Ife.s / U _P = m Uz, _s IU _P = 1.0, Pj.s / Pmax = 0.8, Pmax = 0.112 Ts U _P ² IL and m ² Li = ² L2 = m ² Lz = L.Fig. 10: With the control method for minimum conduction losses of the multiphase DC voltage converter, control characteristic curves for m = 3, depending on the power Pi, _s , 0 <Pi, _s <0.8 Pmax present at the first electrically isolated DC voltage connection, for three different DC voltages present at the first electrically isolated DC voltage connection L / is, ni L / i, _s / L / _p = {0.75, 1.0, 1.5}, and using the following characteristics:

Ü1, S / (7p = P2 (72, s / (7p = m (7s, s / (7 _P = 1.0, P _3, s / Pmax = 0.8, Pmax = 0.112 Ts (7 _P ² / L and m ² Li = ri2 ² L.2 = H3 ² L3 = L.

Fig. 11: With the control method for minimum conduction losses of the multi-phase direct voltage converter, control characteristic curves for m = 3, depending on the power Pi, _S 0 <present at the first electrically isolated direct voltage connection P1, _S <0.8 Pmax, for three different DC voltages U _{2, s>} m U _{2 s} l Up = {0.75, 1.0, 1.5} applied to the second electrically isolated DC voltage connection, and using the following characteristic data: m L / i, _s I Up = n ₂ U _{2, s} / Up = Uz, _s / Up = 1.0,

Ps.s / Pmax = 0.8, Pmax = 0.112 Ts L / _P ² / L and m ² Li = ² L ₂ = m ² = L

Fig. 12: Comparison of the guide losses resulting with the simplified control method according to [2] (shown in dashed lines) to the guide losses which result with the control process for minimal guide losses of the multi-phase DC voltage converter (shown with solid lines), ie for those in FIGS. 7, 8 and 9 shown control characteristic curves: m = 3, depending on the power Pi, _s present at the first electrically isolated DC voltage connection and, unless otherwise indicated in the figure, using the following characteristic data: Ts = 20 ps, m = m = = 1, Li =

L2 = LZ = 222 pH, Up = 700 V, Lh.s = Ui, s = Uz, _s = 700 V, P ₂ , _s = P _3, s = 4 kW (note: for Uk, s = 525 V leaves the simplified control procedure does not exceed Pk, s = 3.7 kW, k = {1, 2,3}, zu). For the examples considered, the improvement achieved is typically in the range between 5% and 20%.

Fig. 13: (a) Simulated voltage and current profiles with the simplified control method and (b) the control method for minimum conduction losses of the multi-phase DC converter.

Fig. 14: With the control method for starting up the multiphase DC converter with limited output current, control characteristics determined for m = 3, Up = 700 V and various output voltages (üi, _s = U _2, s = Uz, _s = {0, 200 V, 400 V, 600 V}). The curves of the duty cycles di, _P = cfe.p = cfe, _P are shown as a function of the output current of the first, the second and the third HF rectifier. Selected characteristics: = m = m = 1, Li = L ₂ = = 222 pH, Ts = 20 ps.

Fig. 15: Simulated voltage and current profiles using the control method for starting up the multi-phase DC-DC converter with limited output current during the start-up process.

Fig. 16: Control characteristic curves for operation with low powers on all isolated DC voltage connections of the multi-phase DC voltage converter for m = 3, Up = 700 V and different voltages on the isolated DC voltage connections: i / i, _s = U _{2, s} = U _s = { 525 V (dash-dotted), 700 V (solid), 1050 V (dashed lines)}. Shown in (a) are the courses of the duty cycles di, p = 02, p = cfe, p and in (b) are the courses of the duty cycles dai.s = dbi .s = da2, s = db2, s = d _a 3, s = db3, s depending on the output current of the first, second and third RF rectifiers .; in addition, cr _#c = (k- 1) 2 p / 3, ßa _k = a _k + 2 pd _k . and ßb _k - ßa _k = 2 p / 3 for all 1 <k <m. Selected characteristics: m = m = m = 1, Li = L2 =

Lz = 222 pH, Ts = 20 ps.

Fig. 17: Simulated voltage and current curves for operation with low powers at all electrically isolated DC voltage connections of the multi-phase DC voltage converter.

Fig. 18: Multi-phase DC half-bridge converter.

Fig. 19: Version as a multi-phase DC voltage half-bridge converter with primary star connection.

Fig. 20: Execution as a multi-phase DC voltage half-bridge converter with primary-side star connection and secondary-side zigzag connection.

Fig. 21: Execution as a multi-phase switchable DC voltage half-bridge converter.

Fig. 22: Circuit variant with a multi-phase RF transformer.

Ways of Carrying Out the Invention

In the following, three embodiments of the control method explained as examples are presented, corresponding to different operating modes of the DC voltage converter. The first embodiment of the control method realizes minimal Leitver losses of the multi-phase DC converter, the second embodiment of the control method enables starting up the multi-phase DC converter with limited output current and the third embodiment, operation with low power at the isolated DC voltage connections of the multi-phase DC converter. Basically, the principle of operation of the converter is that it can be controlled and regulated by connecting voltages from external or internal sources, typically capacitors, to the inductors. There are voltage differences that drive the currents in the inductors. The currents in turn can change the voltages of capacitances or flow into an input or an output.

Correspondingly, a control system can determine currents for correcting these voltages on the basis of desired output voltages and actually existing voltages on output capacitances, and in turn voltages that must be connected to the inductors in order to generate these currents.

The power transmitted there results from voltages and currents at an input or output. A voltage or power occurring at an output or input during operation of the converter can be related to a corresponding maximum voltage or power.

The control method is typically implemented using a control unit, the control method being permanently implemented or being carried out by a program-controlled microprocessor. Input variables for the control process originate in particular from voltage measurements and current measurements, output variables are switching signals for the switches.

The control unit can be set up to implement two or more of the subsequently presented operating modes of the direct voltage converter and to switch between them.

The embodiment of the control method for minimal conduction losses of the multiphase DC converter uses, for technical reasons, duty cycles of 50%, constant switching periods and permanently activated half bridges:

1. In stationary operation, all half bridges are operated with a duty cycle of 50%, ie c, _P = dak.s = dbk, s = 50% for 1 <k <m. This ensures half-wave symmetrical and therefore mean-free curves of the voltages and applied to the primary and secondary transformer connections serves to avoid a possible possible saturation of the magnetic cores of the HF transformers.

2. A constant switching period is assumed to avoid high switching periods

Switching losses, in the event that the embodiment of the control method explained here would determine low values of the switching period, and to avoid saturation of the magnetic cores of the RF transformers, in the event that the embodiment of the control method explained here high values of the switching period would determine, since the optimization criterion, which is used in the embodiment of the control method explained here, does not include switching losses or saturation of the magnetic cores of the FIF-T transformers.

3. The activation signals of all half bridges are 1, to ensure bidirectional power operation, i.e. Feeding and regenerating electrical energy, at all m isolated DC connections.

The optimization criterion of the control procedure for minimal routing losses is:

where k, the effective value of the current in the primary-side winding of the k-th HF transformer and R _{k represents} an equivalent resistance, for calculating the conductance losses in the part of the multiphase DC converter that is to be assigned to the k-th HF transformer.

For optimization, a simulation of the behavior of the converter can be carried out in a manner known per se. Certain control parameters can be specified and others can be varied as part of the optimization. For example, voltages, currents and powers are calculated in the simulation, and from this a target function that serves as an optimization criterion.

The resulting control characteristics of the control method for minimal Leitver losses of the multi-phase DC converter include the values of the degrees of freedom available for its optimization, ie the values of ak, ßak and ß k for 1 <k <m, depending on the operating point. To determine the optimal values for%, ßak and ßbk, 1 <k <m, no closed solutions or equations can be found, therefore numerical solution methods are used. For technical reasons, lookup tables are preferably used for technical reasons, i.e. for various operating points that are technically meaningfully distributed within the operating range of the DC-DC converter, the optimized values for Of _f c, ßak and ßbk, 1 <k <m, are determined in advance calculated, stored in the lookup tables and approved and applied by the controlling or regulating system of the multi-phase DC voltage converter by means of interpolation for the current operating point.

The number of required lookup tables is 3m (for Qk, ßak and ßbk, 1 <k <m) and the dimension of each lookup table is 2 +1 (due to the definition of the operating point by Up, Uk.s, Pk.s ), ie already for the embodiment of the three-phase DC converter, m = 3, and 10 support points per dimension of each lookup table, a total of 90 million values are generated (9-10 ⁷ ). Due to the memory modules available today, this does not represent a technical obstacle for a practical implementation, but it is considered useful for the typical characterization of the results for ak, ßak and ßbk, 1 <k <m, a typical design and a typical operating case choose. These are the design and the operating case, which have been considered in previous publications as part of investigations of the multiphase DC converter with simplified control procedures:

• m = 3,

• The primary-side RF networks are realized by HF capacitor networks and the secondary-side RF networks by RF series resonant circuit networks, the inductance of the kth RF series resonant circuit network being designated Lk (1 <k <m) and the capacitances of the capacitors of all HF networks are chosen so large that these capacitors only keep the DC components away from the HF transformers, but have no notable effects on the current profiles in the windings of the HF transformers,

Ri _k ² L _k = L for 1 <k <m,

• Up = ri _k Uk, s and Pk, s = P, for 1 <k <m and 0 <P <Pmax with Pmax = 0.112 T _s U _P ² / L. FIG. 7 shows the results obtained by means of optimization calculations for the above-mentioned embodiment and the above-mentioned operating case (si = 0 and 1 <k <m), which are listed in the following table:

Table 1

FIG. 8 compares the leading losses resulting for the previously known simplified control method with the leading losses, and those with the optimized control method for minimal leading losses for a multiphase direct voltage converter with m = 3 and for Up = m _< U _{k, s} and P ^ s = P, for 1 <k <m and 0 <Pf Pmax <1, using the following characteristics, result: T _s = 20 gs, m = n ₂ = m = 1 , Li = L ₂ = L3 = 222 mH, Pi = P ₂ =

P3 = 1.5 W, L / _P = 700 V, I / 1, _s = L / _{2, S} = 5/3, s = 700 V. There is always a reduction in the conduction losses, for P> Pmax / 2 this is achieved improvement in the range between 3% and 5%.

FIGS. 9, 10 and 11 exemplify further results of the control characteristic curves of the control method for minimum conduction losses of the multi-phase DC voltage converter with = 3 for the operating points listed below, based on the assumptions that the primary-side RF networks by RF capacitor networks and the secondary-side RF networks are implemented by RF series resonant circuit networks, that the inductance of the / c-th RF series resonant circuit network is designated L _k (1 <k <m) and that the capacitors of all RF networks are capacitors should be chosen so large that these capacitors only keep the DC components away from the HF transformers, but have no significant effects on the current profiles in the windings of the HF transformers:

- Fig. 9:

m L / i, _s I Up = P2 1/2, s / Up = m / 3, s / Up = 1.0,

0 <Pl, s <0.8 Pmax, P2, s / Pmax = {0.2, 0.4, 0.8} And P3, s / Pmax = 0.8.

- Fig. 10:

m / i, _s / Ü _P = {0.75, 1.0, 1.5}, m U _2, s / Up = m iis.s / t7 _P = 1.0,

0 <Pl, s <0.8 Pmax, P2, s / Pmax ⁼ P3, s / Pmax ⁼ 0.8.

- Fig. 11:

n _{2 (i2 S} / Up = {0.75, 1.0, 1.5}, m (7i, _s / t / _P = Lfe.s / (i _P = 1.0,

0 <Pi, s <0.8 Pmax, P _{2, s} / Pmax ⁼ P3, s / Pmax ⁼ 0.8.

Table 2:

P3 / Pmax - 0.8 and nzU3 _t s / L / _P - 1.0

12 compares the guide losses resulting for the (previously known) simplified control method with the guide losses which are achieved with the optimized control method for minimum guide losses for a multiphase DC voltage converter with m = 3, depending on the power Pi, _s and... Present at the first potential-separated DC voltage connection , unless otherwise stated in the figure, using the following characteristics, result: T _s = 20 ps, = nz = nz = 1, Li = L ₂ = Ls =

222 pH, Pi = Rz = Rz = 1.5 W, Up = 700 V, üi, _s = ü _{2, s} = Uz, s = 700 V, P _{2, s} = Pz, s = 4 kW For the examples considered there is always there is a reduction in the lead losses and the improvement achieved is typically in the range between 5% and 20%.

13 shows voltage and current curves simulated according to (a) using the simplified control method and (b) the control method for minimum conduction losses of the multiphase DC voltage converter; curves shown in (a) and (b): top: t / i, _{2 P} , t / i, s and the current / i through Li; Middle: U2, z, p, uz, s and the current / ₂ through Lz \ below: 1/3, i _P , uz, s and the current / 3 through Lz. The curves shown were determined for: m = 3,

U _P = 700 V, Ui, _s = 700 V, Uz, _s = 525 V, Uz.s = 700 V, Pi, _s = 1 kW, P _2.s = 3.7 kW, Pz, s = 4 kW; In contrast to the simplified control method, the control method also uses os - <12> (2p / 3) and 2 p -> (2p / 3) for minimal conduction losses, achieving a reduction in the effective values of the currents in the windings of the second and third HF transformers and overall a reduction in the total of all line losses. Selected characteristics: m = m = m = 1, Li = L2 = L3 = 222 pH, Ts = 20 ps.

The embodiment of the control method for starting up the multiphase DC voltage converter with limited output current is based on the fact that the simplified control method and the above-described control method for minimal conduction losses during startup of the multiphase DC voltage converter, due to the charging of the intermediate circuit capacitors connected to the isolated DC voltage connections , cause very high currents in all power semiconductors and in all HF transformers of the multiphase DC converter under consideration. The simultaneous reduction of the effective values of the currents in all power semiconductors and in all HF transformers can only be achieved if suitable measures significantly reduce the effective values of the voltages present between the first and second input terminals of all primary-side HF networks Values close to zero are reached, which can only be achieved if the duty cycles of the control signals of the half-bridges of the HF inverter deviate from 50%, ie, because of the concatenation of the voltages present between the first and second input terminals of the primary-side HF networks d _{k, p} F 50% for 1 <k <m.

The control method for starting up the multiphase DC converter with limited output current is characterized by the following definitions, where 1 <k <m:

o the activation signals of all half-bridges of the HF inverter are 1 and o the activation signals of all half-bridges of the HF rectifiers are 0, i.e. all HF rectifiers are operated as diode rectifiers and accordingly ßa _k and ßb _{k are} naturally set depending on the operating point.

o either 0 <dp <1 I m or (m - 1) / m <d _k . <1. Optionally, for parallel startup of all k modules, the primary-side duty cycles can be set the same:

od _{k, p} = c / i, p

Typically, one or both of the following relationships also apply:

O da _{k, s} ⁼ d _{k, s} ⁼ 50%,

oa _k = {k- 1) lirlm.

In order to control the output current of the kth secondary RF rectifier, the duty cycle d _{k, P} and the switching period Ts remain as degrees of freedom. Given the value for Ts, d _k, p is selected such that this secondary RF rectifier makes sense or is based on provide the currents required to charge the output capacitors. The switching period can be set so that a desired goal, for example minimal losses, is achieved in the present operating point.

14 shows examples of courses of di, _P for a multi-phase DC voltage converter with m = 3 for different operating points, on the assumption that the primary-side RF networks are realized by RF capacitor networks and the secondary-side RF networks by RF series resonant circuit networks be that the inductance of the kth HF series resonant circuit network is denoted by L _k (1 <k <m) and that the capacitance of the capacitors of all HF networks is chosen to be so large that these capacitors only have the DC components from the HF-T Keep transformers away, but have no significant influence on the current profiles in the windings of the HF transformers. The first range of validity mentioned above, 0 <di, _P <1 I m, was chosen as the range of validity of di, _P (the same result would result for 1-di, _P ). The following applies for 1 <k <m: the curves of di, _P are shown as a function of the output current of the / c-th secondary rectifier and for different values of 14, _s = {0, 200 V, 400 V, 600 V} . The operating points selected for the determination of the curves shown in FIG. 14 are also characterized by m = m = m = 1, Li = L = Lz = 222 pH, U _P = 700 V and Ts = 20 ps, ie to simplify the illustration a constant switching period was selected. The result is a monotonically increasing duty cycle di, _P for a monotonically increasing output current of the / c-th DC converter and a constant value for 14, _s . For constant duty cycle and monotonically increasing 14, s of the k-th DC converter results in a monotonically falling output current of the k-th DC converter, which disappears for U _{k, s} = U _P f hk. Because of this connection, the control method for starting up the multiphase DC converter with limited output current inherently ensures that this limit is not exceeded, so in a practical application 14, s <Up I ri _k applies, which is particularly useful during the start-up period can be, as individual monitoring and measuring components may also only be started during this period.

Fig. 15 shows simulated voltage and current profiles, which with the control method for starting up the multiphase DC converter with limited output current during the start-up process for m = 3, di, _P = cfe, p = cfe.p = 15%, Up =

700 V at different output voltages result: (a) L / i, _s = 14, s = 14, s = 0,

(b) üi, _s = 14, s = l / _3, s = 200 V and (c) l / i, s = 14, s = 14, s = 400 V. = m = nz = 1, Li = Li = Lz = 222 pH and T _s = 20 ps. Shown courses in (a), (b) and (c): above: ui, 2, _P , m, _s and current through Z_i; Middle: U2, z, p, 1/2, s and current through £ .2; below: 1/3, i _P , uz.s and current through Lz. Selected characteristics: = m = nz = 1, i = L2 = L3 = 222 pH, Ts = 20 ps.

The embodiment of the control method for the operation with low powers at all electrically isolated DC voltage connections of the multi-phase DC voltage converter is based on the fact that with low output powers it is advantageous for the precise and stable regulation of all output currents if, in the present multi-phase DC voltage converter, the between the first and second input terminals of all primary-side RF networks and between the first and second output terminals of all secondary-side RF networks, voltages present have low effective values, ie Have voltage curves that are positive or negative for a relatively long period of time in relation to the switching period, and only for a relatively short period of time. Using the control method described above for minimal routing losses of the multi-phase

DC converter, as well as with the simplified control method, this can be done for the primary-side RF networks, however, due to the used duty cycle of d _k .p = 50%, 1 <k <m, and due to the chaining of the voltages, that are present between the first and second input terminals of the primary RF networks, do not reach for all m primary RF networks simultaneously. The simultaneous reduction of the effective values of the voltages present between the first and second input terminals of all primary-side RF networks down to values close to zero can only be done because of the concatenation of the voltages present between the first and second input terminals of the primary-side RF networks ensure that the duty cycle of the control signals of the half-bridges of the HF inverter deviate from 50%, ie d _{k, P} F 50% for 1 <k <m.

Services with less than 20% or 30% of the maximum output can be considered as low outputs.

The control procedure for low-power operation at the isolated DC voltage connections of the multiphase DC converter is characterized by the following definitions, where 1 <k <m:

o all activation signals are 1.

o 0 <dk.p + dak.s <1 1 m, i.e. in each module the sum of the duty cycles on the primary and secondary side is between zero and 1 I m.

In addition, one or more of the following relationships can be implemented by the controller:

o dk, p = cfi.p, i.e. the duty cycles of the separate DC voltage connections are the same. This is useful for symmetrical loads

o dak, s = dbk.s, ie the duty cycles in the two half bridges of the HF rectifier 5 are the same, as a result of which a DC component can be eliminated. Alternatively or additionally, a DC component can also be eliminated by series capacitors o cr / _c = (k - 1) 2 TT Im, ie a phase shift between the m phases is realized on the primary side;

ßbk - ßak = 2 p / m, ie there is a phase shift between the voltage pulses of the two half bridges of the HF rectifier. The negative voltage pulse on the secondary side (of the HF rectifier) thus connects to a negative voltage pulse of the primary side (of the HF inverter) with the same delay as is the case for the positive voltage pulses. This can reduce losses (see Fig. 17). With the definition of a _k , for d _k , _P = di, _P and after the definition of di, the following curve results for the voltage i / i, 2, _P () between the first and second input terminal of the first primary-side RF network. f):

o 0 <t <di, T _s : ui, 2, p (t) «Up (referred to as time interval l _P ),

o di, _p 7s <t <Ts / / 7?: t / i, 2, p (f) «0 (referred to as time interval ll),

o Ts I m <t <(di, _P + 1 / m) 7s: i / i, 2, _P (f) «-Up (referred to as time interval IIIp), o (di, _P + 1 / m) Ts < t <T _s : ui, 2, _P (f) «0 (referred to as time interval IV _P ).

The curves of the voltages present between the first and second input terminals of the remaining, kth primary-side FIF networks are equal to the curves of m, 2, p (t— (k— 1) Ts / m).

With the definitions of ßbk - ßak and da _k .s = db _k .s and after the determination of ßak and da _k .s, the following results for the between the first and second output terminal of the / c-th secondary RF network applied voltage t / i, _s (f) (applies to 1 <k <m):

o 0 <t - Ts ßak / (2 p) <da c, s Ts. i / i, _s (f) «Us (time interval l _s ),

o dak.s T _s <t - Ts ßak / (2 p) <Ts / m: ui, _s (t) «0 (time interval lls),

o Ts I m <t - Ts ßak / (2 TT) <(dak.s + 1 / m) T _s :, s (t) «-Us (time interval llls), o (dak.s + 1 / m) T _s <t- Fs ßak / (2 TT) <T _s : i / i, s (f) «0 (time interval IV _S ).

The presented tax procedure can be used instead of

o 0 <di, _P + da _k .s <1 I m,

o dak.s ⁼ dbk.s,

o ßbk - ßak = 2 p / m,

In a first alternative, also implement the following control parameters (the same secondary voltage results here):

o 0 <di, _P + (ßbk - ßak) / (2 p) <1 / m,

O da _k .s ⁼ db _k .s ⁼ 1 / P7,

where the degrees of freedom ßb replace the originally available degrees of freedom da _k .s (for controlling the output currents of the m secondary RF rectifiers). The first alternative has no effect on the choice of ßak, ie depending on the direction of energy flow, ßak = Oa _k ± 2 pd _{k, P} is again a sensible choice.

A second alternative is given with:

o 0 <di, _P + (ßak - ßbk) / (2 TT) <1 / m, o dak.s = dbk, s = (m - 1) / m,

where the degrees of freedom ßak replace the originally available degrees of freedom dak, s (for controlling the output currents of the m secondary RF rectifiers). The second alternative therefore has an effect on the choice of ßak and, depending on the direction of energy flow, is ß k ⁼ oiak + 2 TT / /? 7 + 2 TT dk, p (supply of electrical energy) or ßbk = aak + 2 TT / m - 2 TT dk, p (electrical energy supply) a sensible choice.

To control the output currents of the m secondary RF rectifiers, degrees of freedom i, _P , dak, s, ßak and T _s remain. These can be determined, for example, as part of an optimization calculation, for example to minimize the total losses of the multiphase DC converter, under the secondary conditions of the required output currents of the m secondary RF rectifiers. In a practical implementation, however, a simple method for determining the remaining degrees of freedom is often desired. A control method for operation with low power, low switching losses and reduced computation costs can be implemented if dak, s is selected so that the current in the secondary winding of the k-th HF transformer with the expiry of the time intervals l _s and llls den Value reached zero, which provides favorable conditions for achieving low switching losses, and ßak = Gfa _c + 2 p dk, p is set. With ßak = ar _{a / c} + 2 p dk.p it is achieved that i / i, s (f) changes from zero to 14 if and only if t / i, 2, _P (f) changes from Up to zero and ui, _s (t) changes from zero to -14 if and only if t / i, 2, _P (f) changes from -Up to zero, ie the time interval l _s immediately follows the time interval l _P and the time interval lll _s immediately follows the time interval III _P.

16 shows examples of courses of c4, _p and dak, p = dbk.p for the control method for the operation with low powers, low switching losses and reduced calculation effort for a multiphase DC voltage converter with m = 3 for different operating points, under the assumptions that the primary-side RF networks are realized by RF capacitor networks and the secondary-side RF networks by RF series resonant circuit networks, that the inductance of the kth RF series resonant circuit network is designated L _k (1 <k <m) and that the capacitances of the capacitors of all RF networks are chosen so large that these capacitors only keep the DC components away from the RF transformers, but none have appreciable influences on the current profiles in the windings of the HF transformers. For 1 <k <m the following applies: the curves of dk.p and dak.p are dependent on the output current of the / c-th secondary rectifier and for different values of U i, _s = {525 V, 700 V, 1050 V} and voltages of the same size at the potentially separated DC voltage connections, 14, _s = (ii, s). The operating points selected for the determination of the curves shown in FIG. 16 are also characterized by m = m = m = 1, Li = La = Lz = 222 pH, Up = 700 V and Ts = 20 ps, which means that a constant switching period was chosen to simplify the illustration, resulting in a monotonically increasing output current of the / c-th DC converter and a constant value for Uk , _s monotonically increasing duty cycles dk.p and dak.p. For a constant output current of the / c-th DC converter and a monotonically increasing Uk, s of the / c-th DC converter, the duty cycle d increases k.p and a monotonous drop in the duty cycle dak.p. The maximum output current of the / c-th DC converter is for c / i, _P + dak.s = 1 I m, ie at the operating limit of the control method for operation with low power, low switching losses and reduced calculation effort.

17 shows simulated voltage and current profiles for 1 A output current (per HF rectifier) for m = 3, Ui, _s = Ü2, s = Uz.s = 700 V, 0 <dk.p + dak.s <1 I m, dk.p = di, _p , dak.s = dbk, s, (Xk = (k - 1) 2 tt / rn and ßbk - ßak = 2 p / m. Represented courses in (a), (b ) and (c): above: t / i, 2, _Pl ui, s and current through Li; middle: t / 2,3, p, U2, s and current through Li \ below: t / 3,1, p , t / 3, s and current through Lz. Selected characteristics: m = = re = 1, Li = L2 = L3 =

222 pH, Ts = 20 ps.

This method can also be used to regenerate electrical energy, namely for ßak <ctak. There are favorable conditions for achieving low switching losses if ßak = aak - 2 p dak.s is set. With ßak = aak - 2 p dak, s it is achieved that t / 1,2, _P (f) changes from zero to Up if and only if, _s (t) changes from U _s to zero and t / 1,2 , p (t) changes from zero to -Up if and only if t / i, _s (f) changes from -L4 to zero (and accordingly also for all other phases), ie the time interval l _p immediately follows the time interval l _s and the time interval III _P immediately follows the time interval IIIs. 18 shows a multi-phase DC voltage half-bridge converter, which provides that the circuit breakers of all second half-bridges of the m HF rectifiers of the multi-phase DC voltage converter are replaced by capacitors. An advantage of the multi-phase DC voltage half-bridge converter is that there is no additional effort (costs, additional circuitry for the control) caused by the replaced circuit breakers in the multi-phase DC voltage converter. In addition, the intermediate circuit capacitors on the output side of the multiphase DC converter can be omitted. Depending on the circumstances, a reduction in the degree of freedom of the control from 9 m to 6 m can be disadvantageous, since the activation signals, the pulse duty factors and the phase relationships βb _k are omitted for the second half bridges of the m HF rectifiers.

With a suitable choice of the primary and secondary HF networks and a suitable choice of the number of turns of the primary windings of the HF transformers, the sum of the time profiles of all magnetic fluxes in the m HF transformers disappears at all times. This property can also be realized with the implementation variants of the multi-phase DC-voltage half-bridge converter described below (implementation of a multi-phase DC-voltage half-bridge converter with primary-side star connection, a multi-phase DC-voltage half-bridge converter with primary-side star connection and secondary-side zig-zag switching and a multi-phase switching with the same function) RF transformers through a multi-phase RF transformer with m phases, as also described below.

19 shows an embodiment of the multi-phase DC voltage half-bridge converter with primary-side star connection. In comparison with the multi-phase DC voltage half-bridge converter from FIG. 18, this provides for modified connections of the primary-side HF networks. For the multi-phase DC voltage half-bridge converter with primary star connection, for 1 <k <m:

• The first input terminal of the k-th primary-side RF network is connected to the switching voltage connection of the k-th half-bridge of the RF inverter. • The second input terminal of the first primary-side RF network is connected to all second input terminals of the / c-th primary-side RF network.

These connection changes result in lower voltages between the first and second input terminals of the RF networks, which, depending on the requirements, can mean a practical advantage in comparison with the multi-phase DC voltage half-bridge converter from FIG. 18. The primary-side connection of the multi-phase DC voltage half-bridge converter with primary-side star connection can be combined with the various implementations shown here of the secondary side of the multi-phase DC voltage converter.

20 shows an embodiment of the multi-phase DC voltage half-bridge converter with primary-side star connection and secondary-side zigzag connection. In comparison to the multi-phase DC voltage half-bridge converter with primary star connection from FIG. 19, RF transformers with two secondary windings, each secondary winding having two transformer connections, and modified connections between the secondary windings of the RF transformers with two secondary windings and the input terminals of the secondary RF Networks. The winding sense of the windings of the HF transformers with two secondary windings are defined as follows: If there is a positive voltage from the first to the second transformer connection of the primary winding of the H FT transformer with two secondary windings, then the first to the second transformer connection are present the first secondary winding of the HF transformer with two secondary windings and also from the first to the second transformer connection of the second secondary winding of the HF transformer with two secondary windings positive voltages. For 1 <k <m the following applies: the transformation ratio of the / c-th HF transformer with two secondary windings (n _k ) is defined as the ratio of the number of turns of the primary winding (Nk.p) to the number of turns of the first secondary winding (Nk, a, s) of the / c-th HF transformer with two secondary windings: n _k = N _k / A // c, a, s. For the multi-phase DC voltage half-bridge converter with primary-side star connection and secondary-side zigzag connection, the following applies for 1 <k <m: the first transformer connection of the first secondary winding of the / c-th HF transformer with two secondary windings is connected to the first input terminal of the / c-th connected on the secondary side RF network, the second transformer connection of the first secondary winding of the -th HF transformer with two secondary windings is connected to the second transformer connection of the second secondary winding of the (k + 1) -th HF transformer with two secondary windings and the first transformer connection of the second secondary winding of the (+1) -th H FT transformer with two secondary windings is connected to the second input terminal of the -th secondary RF network.

Furthermore, the first transformer connection of the first secondary winding of the mth RF transformer with two secondary windings is connected to the first input terminal of the mth secondary-side RF network, the second transformer connection of the first secondary winding of the mth RF transformer with two secondary windings is connected to the second transformer connection of the second secondary winding of the first RF transformer with two secondary windings and the first transformer connection of the second secondary winding of the first RF transformer with two secondary windings is connected to the second input terminal of the mth secondary RF network connected. The number of turns of the second secondary winding of the / c-th HF transformer with two secondary windings (A // c, b, s) can be selected appropriately,

e.g. A / ic, b, s = Nk, p / ri for 1 <k <m or A / i, b, s = A / i, p / n _m .

Compared to the multi-phase DC voltage half-bridge converter with primary-side star connection from the vehicle. 19, the multi-phase DC voltage half-bridge converter with primary-side star connection and secondary-side zigzag circuit from FIG. 20 has the advantage that this circuit can be used in a technically sensible manner even with asymmetrical loading, is present when the electrical power output or received at the isolated DC voltage connections is very different.

The m RF rectifiers of the multi-phase DC voltage half-bridge converter with primary-side star connection and secondary-side zigzag connection can also be replaced by the m RF rectifiers of the multi-phase DC voltage converter from FIG. 1.

FIG. 21 shows an embodiment of the multi-phase switchable DC voltage half-bridge converter which, in comparison to the multi-phase DC voltage half-bridge converter from FIG. 18, has a relay between the RF inverter and the primary-side RF networks. The relay has: a control coil and m Changeover switch having three contacts (main contact, first changeover contact, second changeover contact). The following applies for 1 <k <m: in the case of a non-magnetized control coil, the main contact of the / c-th changeover switch is connected to the first changeover contact of the kth changeover switch, and in the case of a magnetized control coil the main contact of the kth changeover switch is connected to the second changeover contact of the k- ten changeover switch connected. The multi-phase switchable DC voltage half-bridge converter is largely identical to the multi-phase DC voltage half-bridge converter from FIG. 18, only the connections between the RF inverter and the primary-side RF networks change, as described below and shown in FIG. 21. The following applies for 1 <k <m: the switching voltage connection of the k-th half-bridge of the RF inverter is connected to the first input terminal of the k-th primary-side RF network, the main contact of the k-th change-over switch is connected to the second input terminal of the k-th connected on the primary side RF network and the first changeover contact of the kth changeover switch is connected to the switching voltage connection of the (k + 1) th half bridge of the RF inverter. The switching voltage connection of the m-th half bridge of the HF inverter is connected to the first input terminal of the m-th primary-side HF network, the main contact of the m-th change-over switch is connected to the second input terminal of the m-th primary-side HF network and the first changeover contact of the m-th changeover switch is connected to the switching voltage connection of the first half bridge of the HF inverter. The second changeover contacts of all m changeover switches are connected to each other.

The multi-phase switchable DC half-bridge converter therefore implements two different circuits, depending on whether the control coil is magnetized or not:

- Non-magnetized control coil: multiphase DC half-bridge converter according to Fig. 18

- Magnetized control coil: multiphase DC half-bridge converter with primary star connection according to Fig. 19

The m HF rectifiers of the multiphase switchable DC voltage half-bridge converter can also be replaced by the m HF rectifiers of the multiphase DC converter from FIG. 1. The multi-phase switchable DC voltage half-bridge converter can also be designed with secondary-side zig-zag circuitry as shown in FIG. 20 using HF transformers with two secondary windings.

22 shows an embodiment of the multiphase DC half-bridge converter with multiphase RF transformer. Instead of the m HF-transformers of the multiphase DC voltage half-bridge converter from FIG. 18, this provides a multiphase HF transformer with m phases. The multi-phase RF transformer with m phases can be realized using a magnetic core having k legs which are magnetically connected to one another via yokes. The following applies for 1 <k <m: the kth leg is wound with a winding package having a primary-side winding with the number of turns Np and a secondary-side winding with the number of turns N _s , k. The multi-phase RF transformer has no magnetic yoke and the technical advantage of the multi-phase RF transformer is accordingly that the total core volume of a multi-phase RF transformer with m phases is less than the sum of all core volumes of m HF -T transformers (smaller size and cost). The sensible technical use presupposes, however, that the sum of the time profiles of all magnetic fluxes in the m legs disappears at any time. With a suitable choice of the primary and secondary RF networks and a suitable choice of the number of turns of the primary and secondary windings of the HF transformers (or the HF transformers with two secondary windings), this condition is valid for all the variants explained and resulting from circuit combinations given multi-phase DC voltage half-bridge converter and the multi-phase DC voltage converter:

the multi-phase DC voltage half-bridge converter from FIG. 18,

the multiphase DC voltage converter with primary star connection from FIG. 19,

the multiphase DC converter with primary-side star connection and secondary-side zigzag circuit from FIG. 20,

- The multi-phase switchable DC voltage converter from Fig. 21 and

the multiphase DC converter from FIG. 1, - Circuit variants that can be developed by suitable combinations of the circuits shown in Figures 1, 18, 19, 20 and 21.

An example of a suitable choice of primary and secondary RF networks is the choice of RF capacitor networks as primary RF networks, the selection of RF series resonant networks as secondary RF networks and the selection of sufficiently high capacitance values in all RF networks , so that the voltages present across the capacitors of the HF networks have no appreciable influence on the voltages present on the primary and secondary windings.

Depending on the design of the primary and / or secondary RF networks, the use of resonance circuits with one or more series and / or parallel resonance frequencies enables the realization of a multi-phase resonance DC voltage converter. Each phase of the multi-phase DC resonance converter enables the same modes of operation as a resonance converter, e.g. as a “half-cycle discontinuous conduction mode” (HC-DCM) series resonant converter, with the 9 m degrees of freedom described for the multi-phase DC voltage converter.

In HC-DCM operation of the resonance DC voltage converter with a given voltage at the DC voltage connection of the HF inverter, largely load-independent voltages are established at the isolated DC voltage connections of the HF rectifiers.

The approach of developing control methods by including all degrees of freedom of the control that improve the operation of the multiphase DC converter under consideration can be applied analogously to all variants of the multiphase DC voltage converter that are explained and result from circuit combinations. Furthermore, the explanations of the control method explained above (for minimum conduction losses of the multiphase DC converter, for starting up the multiphase DC converter with a limited output current and for operation with low power at the isolated DC voltage connections of the multiphase DC converter) can be applied to all of the explanations as well as by Use circuit combinations resulting variants of the multiphase DC converter.

Claims

1. Multi-phase, electrically isolated DC voltage converter (10), for exchanging electrical energy between a primary-side DC voltage connection and m secondary-side isolated DC voltage connections, comprising the following subunits

• an HF inverter (1);

• connected to m primary RF networks (2);

• connected to it m primary windings of an HF transformer arrangement (3);

• secondary windings of the HF transformer arrangement (3);

• connected to m secondary RF networks (4);

• connected to it m HF rectifier (5); and with a control of the subunits, which is designed to determine control parameters, corresponding to degrees of freedom of the control, and to adapt them to an operating state of the DC voltage converter (10) during operation of the DC voltage converter (10), the control parameters which can be adapted in this way include at least , for 1 <k <m: dk, p duty cycle of the control signal of the kth half-bridge of the HF inverter; dak, s duty cycle of the control signal of the first half bridge of the k-th HF rectifier; dbk, s duty cycle of the control signal of the second half-bridge of the k-th HF rectifier; s _{/ c} phase angle between the rising edge of the control signal of the first half bridge of the HF inverter and the kth half bridge of the HF inverter; ßak phase angle between the rising edge of the control signal of the first half bridge of the HF inverter (si, _P ) and the rising edge of the control signal of the first half bridge of the kth HF rectifier (Sa _{k, s} ); ßbk phase angle between the rising edge of the control signal si, _p and the rising edge of the control signal of the second half-bridge of the kth HF rectifier (swc.s).

2. DC voltage converter (10) according to claim 1, wherein the control parameters that can be adjusted during operation additionally comprise at least one of

T _s switching period;

Activation signals of the half bridges.

3. DC voltage converter (10) according to claim 1 or 2, wherein the controller is designed to minimize conduction losses

• activate all half bridges;

• the phase angles cr _k, ßak, ßbk to at least approximately minimize the optimization criterion f = hiΐh

u determine, where, the effective value of the current in the primary-side winding of the k-th HF transformer and Rk represents a corresponding equivalent resistance.

4. DC voltage converter (10) according to claim 3, with m = 3 secondary-side potential-isolated DC voltage connections, the control, with symmetri cal loading of the secondary-side DC voltage connections with a power P, is designed to the phase angle cn = 0, az = 2TT / 3, = 4p / 3 to set, and to set first phase angle differences ßa-i - on = ßa 2 - 02 = ßaz - m, and to set second phase angle differences ßv \ 3ai ßpi - ß _a2 = ßm - ßaz, and in particular to set the phase angle differences depending on the ratio of the power P to one to adjust the maximum power P max by

• for P between zero and Pm _ax / 2 the first phase angle difference increases from zero and drops again to zero, and the second phase angle difference increases monotonically from 2TT / 3 to p; and

• for P between Pmax / 2 and P _ma x, the first phase angle difference increases monotonically from zero to p / 3, and the second phase angle difference remains constant at p ver.

5. DC voltage converter (10) according to claim 4, wherein the phase angle and phase angle differences run according to Table 1.

6. DC voltage converter (10) according to claim 3, 4 or 5 with m = 3 secondary-side potential-isolated DC voltage connections, wherein the control is designed for asymmetrical loading of the secondary-side DC voltage connections with powers Pi, _s , Pi, s and Pz, _s and For different voltages 1/1 _S , Ui, s and 1/3, s, adjust the phase angle and the phase angle differences according to Table 2 on the secondary-side DC voltage connections.

7. DC voltage converter (10) according to one of the preceding claims, wherein the control is designed to set the control parameters in an operating mode for starting up the DC voltage converter as follows, wherein 1 <k <m: o activation of all half bridges (11) of the HF Inverter (1);

o deactivation of all half bridges (51, 52) of the HF rectifiers (5);

and at least one of the following three groups of conditions is met:

o dk, p F 50% and optionally also dk.p = c / i _P ;

O

o 0 <d _k, <1 / m or (m - 1) / m <dk.p <1, and optionally also cfe.p = i _P.

8. DC voltage converter (10) according to claim 7, wherein the pulse duty factor dk, p is adapted in each case to control an output current of a k-th secondary-side HF rectifier.

9. DC voltage converter (10) according to one of the preceding claims, wherein the control is designed to set the control parameters during operation with low powers as follows, where 1 <k <m:

o activation of all half bridges of the HF inverter (1) and the HF rectifier (5);

od _k , p * 50%;

and in particular at least one of the following three groups of conditions:

o Sϊ = i < _{, p} <1 or S = id, _CP > m - 1;

o 0 <dk, p <1 / m or (m - 1) / m <dk.p <1;

o 0 <dk.p + dak.s <1 I m.

10. DC voltage converter (10) according to claim 9, wherein the controller is designed to set one or more of the further control parameters during operation with low powers, as follows, where 1 <k <m:

O dk.p ⁼ di .p ^' ,

O dak, s = dbk, s ^' ,

o ü _k = (k- 1) 2 TT // T ?;

O ßbk - ßak = 2 ΊT / m.

11. DC voltage converter (10) according to one of the preceding claims, wherein the controller is designed to set the control parameters as follows according to one of the three following variants in the case of operation with low powers, wherein

1 <k <m: first variant:

o 0 <di, _P + dak.s <1 fm,

O dak.s ⁼ dbk.s,

o ßbk - ßak = 2 p / m, second variant:

o 0 <i, p + (ßbk - ßak) / (2 tt) <1 / m, O dak, s— dbk, s— 1 / GP, third variant:

o 0 <di, _P + (ßak - ßbk) / (2 p) <1 / m,

o dak, s = dbk, s = (m ~ 1) / m.

12. DC voltage converter (10) according to claim 1 or 2, wherein the controller is designed to operate in a half-cycle discontinuous conduction mode, hereinafter referred to as HC-DCM, the operating mode of the DC voltage converter (10) to determine adaptable control parameters so that a characteristic curve for HC-DCM is formed in at least one element of the primary-side RF networks (2) and / or the secondary-side RF networks (4).

13. DC voltage converter (10) according to claim 12, wherein the control is designed to set the control parameters that can be adjusted during operation in the HC-DCM as follows, where 1 <k <m:

• activation of all half bridges of the HF inverter (1);

• Activation of both half bridges in at least one of the HF rectifiers (5) during time intervals in which the HF rectifier (5) has a semi-oscillatory course of its input current, and deactivation of one or both half bridges of the HF rectifier (5) during time intervals , in which the HF rectifier (5) has a vanishing input current, the input current of an HF rectifier (5) being defined as a current flowing through switching voltage connections of the HF rectifier (5) on the AC voltage side;

Setting the phase angle ak = (k-1) 2tt / iti;

• Setting the first phase angle differences ßak - ct _k = 0;

• Setting second phase angle differences ßbk - Qk = TT.

14. DC converter (10) according to claim 13, wherein

• in the case of energy transmission from the HF inverter (1) to the HF rectifiers (5), to form a current profile characteristic of HC-DCM at the output of at least one of the secondary-side HF networks (4), individual or all switches from at least one of them connected HF rectifiers (5) are deactivated, diodes present parallel to the deactivated switches permitting a natural commutation process; and or

• in the case of energy transmission from the HF rectifiers (5) to the HF inverter (1), to form a current characteristic characteristic of HC-DCM at the input of at least one of the primary-side HF networks (2), individual or all switches of the connected thereto HF inverters (1) are deactivated, diodes present parallel to the deactivated switches permitting a natural commutation process.

15. DC voltage converter (10) according to one of the preceding claims, in which the HF rectifiers (5) each have a first half bridge (51) with power switches, and a second half bridge (52b) with capacitors.

16. DC voltage converter (10) according to one of the preceding claims, wherein the RF transformer arrangement (3) has m individual RF transformers, and in particular primary-side windings of these RF transformers form a delta connection or a star connection or between a star and Delta connection are switchable.

17. DC converter (10) according to claim 16, wherein the HF-T transformers each have two secondary windings, and the secondary windings of all HF transformers form a zigzag circuit.

18. DC voltage converter (10) according to one of claims 1 to 15, wherein the HF transformer arrangement has a multi-phase HF transformer with m phases.