WO2019219136A1

WO2019219136A1 - A switched-mode power converter

Info

Publication number: WO2019219136A1
Application number: PCT/DK2019/000181
Authority: WO
Inventors: Jeppe Christian Bastholm
Original assignee: Linak A/S
Priority date: 2018-05-18
Filing date: 2019-05-20
Publication date: 2019-11-21
Also published as: CN112292805A; CN112292805B

Abstract

A switched-mode power converter (22) comprises an inductor (L₁) connected to afirst (M₁) and a second (M₂) switching element, a capacitor (C₂) connected to the second switching element (M₂) and a transformer (Tr). A control circuit (41) is configured to control the first and second switching elements with first and second duty cycles so that when one switching elements is conducting, the other one is not conducting, and to determine the duty cycles in dependence of the input voltage (V_in). The converter is further configured to continuously detect a signal (V_curr) indicative of a momentary value of a load current drawn from the input voltage; and the control circuit is further configured to determine, in each period of the switching frequency, the duty cycles in dependence of the detected signal. In this way, power factor correction is achieved with a reduced number of components and a low cost.

Description

A switched-mode power converter

Technical Field

The invention relates to a switched-mode power converter comprising an in ductor connected to a first and a second switching element, a capacitor con nected to the second switching element and a transformer. The invention also relates to a power supply comprising a switched-mode power converter and to an actuator system.

Background

Many electric and electronic systems are powered by a power supply that converts electric power from e.g. a mains AC supply net to e.g. a DC voltage level suitable to be used by the components of the electric or electronic sys tem. An example of such systems is an actuator system comprising one or more linear actuators. Such power supplies can be implemented in different ways.

Linear power supplies based on a transformer and a rectifier in the form of a rectifier bridge and a capacitor for smoothing the voltage are very robust and reliable, but they are also heavy and voluminous and have a relatively low efficiency.

Switched-mode power supplies have a higher efficiency and a considerably smaller magnetic circuit due to an operating frequency typically in the range of 30 - 100 kHz. Switched-mode power supplies exist in different circuit to pologies, such as flyback, boost, buck, SEPIC and forward converters.

Another example of a switched-mode power supply has been described in WO 2013/004232 to Linak A/S. In this power supply, the AC voltage from the mains supply is rectified in a bridge rectifier and buffered in a capacitor be fore it is fed to the converter as its input voltage. The converter comprises an inductor, two controllable switching elements, e.g. MOSFET transistors, a capacitor and a transformer. The inductor is connected between an input terminal and a connection point, and one of the switching elements between the connection point and the other input terminal, e.g. a ground terminal. The other switching element is connected between the connection point and the capacitor, whose other end is connected to ground. The primary winding of the transformer is connected between the connection point and one of the input terminals. On the secondary side, the voltage is rectified and buffered in at least one buffer capacitor. A control circuit controls the two switching ele ments to operate opposite of each other, i.e. when one of them is on the oth- er one is off. The duty cycles of the switching elements are controlled in de pendence of the input voltage, so that the output voltage can be kept sub stantially constant independently of the input voltage. In other words, the power supply adjusts itself automatically to operate on different mains AC supply voltages, such as e.g. 100 V used in Japan and 230 V used in Eu- rope. A typical switching frequency can be 50 kHz.

This is a relatively simple switched-mode converter using a low number of components, which thus also results in a low production cost. Further, it is a very robust and reliable construction, and its ability to maintain a substantially constant output voltage over a large range of input voltages allows it to be used worldwide without any need to change components or use mechanical switches to adapt to the different mains supply voltages.

However in recent years, an increasing demand for power factor correction has evolved, also for this type of power supplies.

As mentioned above, in the described power supply, as well as in most other switched-mode power supplies, the AC voltage from the mains supply is rec tified in a bridge rectifier and buffered in a capacitor before it is fed to the converter as its input voltage. This means that the power supply only draws current from the AC mains supply in short pulses at the peaks of the input waveform, where the instantaneous AC voltage exceeds the voltage across the capacitor. In the remaining part of the AC cycle, the capacitor provides the energy to the converter, and thus each current pulse must contain enough energy to sustain the load until the next peak. Thus, the waveform of the current drawn from the AC mains supply will be distorted to consist of these current pulses instead of having the shape of a sine wave correspond- ing to the voltage. In other words, the input current to the power supply will have a considerable amount of energy at harmonics of the fundamental fre quency of the mains voltage, i.e. 50 or 60 Hz. Therefore, since only the fun damental component of the current produces real power, while the other harmonics contribute to the apparent power, the power factor will be low.

A low power factor creates extra load on utility lines, which is a particular problem for power companies, because they cannot compensate for the harmonic current by adding simple capacitors or inductors. It also increases heating of building wiring and utility transformers, and it may cause stability problems in some applications and interference with other devices being powered from the same source. Further, many regulatory standards around the world now set limits on the harmonics of the AC input current.

Ideally, the power supply should present a load that emulates a pure resistor, so that the reactive power drawn by the device is zero. With the absence of input current harmonics, the current would be a perfect replica of the input voltage, i.e. a sine wave, and in phase with it.

Several ways of changing the waveform of the current drawn by a power supply to improve the power factor are known. The process of shaping the input current is commonly called power factor correction, and the purpose is to increase the power factor and decrease the harmonic content.

A relatively simple and well known way of controlling the harmonic current is the so-called passive power factor correction, where a filter is used that passes current only at line frequency (50 or 60 Hz). The filter may consist of capacitors or inductors, and it makes the power supply look more like a linear load. As an example, one inductor arranged between the bridge rectifier and the buffer capacitor can increase the power factor to some degree. An ade quate inductor may reduce the peaks of the current and spread the current out in time well enough to reduce the harmonics enough to meet the regula tions. A disadvantage of passive power factor correction is that it requires large inductors or capacitors, but in some power supplies where the large size of the inductor and its weight (due to its iron core and copper winding) are not objectionable, the method can be used. Further, it can be difficult to design an inductor that is adequate for different mains AC supply voltages, such as e.g. 100 V used in Japan and 230 V used in Europe. Thus, often a voltage range switch to switch between two inductors in dependence of the mains voltage level may be needed. Also, in practice, passive power factor correction is often less effective at improving the power factor, and the filters are typically quite expensive. Active power factor correction, where power electronics is used to change the waveform of the current drawn by a power supply to improve the power factor, is also known. In the most common example of active power factor correction, a current regulated boost converter is inserted between the bridge rectifier and the buffer capacitor. This boost converter is controlled in a man- ner to shape the input current to match the input voltage waveform, so that it attempts to maintain a constant DC voltage on its output to the buffer capaci tor while drawing a current that is in phase with and at the same frequency as the mains voltage. After the buffer capacitor, another switched-mode con verter produces the desired output voltage from the voltage on the buffer ca- pacitor. A certain ripple is unavoidable in the voltage on the buffer capacitor, but this can normally be handled by the other switched-mode converter. This approach can correct the power factor, and it permits cheaper and smaller passive components than in the passive power factor correction. However, it increases the complexity and cost of the complete power supply, because it requires additional semiconductor switches and control electronics. Actually, it is a two-stage solution, where two separate switched-mode converters are used instead of one, which of course increases the complexity and cost. Summary

Therefore, it is an object of embodiments of the invention to provide a simple, robust and reliable switched-mode power converter capable of providing a substantially constant output voltage over a large range of input voltages, where the converter uses a low number of components and has a low pro duction cost, and where the converter at the same time can correct the power factor of the converter without the use of large and heavy inductors or the considerable complexity and cost of a two-stage power supply using two separate converters to correct the power factor.

According to embodiments of the invention the object is achieved in a switched-mode power converter for converting an input voltage at a primary side of the converter to an output voltage at a secondary side of the convert er, said switched-mode power converter comprising a first inductor connect- ed between a first input terminal for said input voltage and a connection point; a first electronically controllable switching element connected between said connection point and a second input terminal for said input voltage; a second electronically controllable switching element connected between said connection point and a first capacitor having its other end connected to said second input terminal for said input voltage; a transformer having a primary winding and a secondary winding, wherein said primary winding is connected between said connection point and one of said first and second input termi nals for said input voltage; an output circuit connected to said secondary winding and arranged to rectify a voltage present on said secondary winding and buffer the rectified voltage in at least one buffer capacitor connected to a first and a second output terminal for said output voltage; and a control circuit configured to generate control signals at a switching frequency for controlling said first electronically controllable switching element with a first duty cycle and said second electronically controllable switching element with a second duty cycle so that when one of said electronically controllable switching ele ments is conducting, the other one is not conducting, wherein the control cir cuit is further configured to determine said first and second duty cycles in dependence of the input voltage. The object is achieved when the converter is further configured to continuously detect a signal indicative of a momentary value of a load current drawn by the converter from said input voltage; and the control circuit is further configured to determine, in each time period of the switching frequency, said first and second duty cycles in dependence of the continuously detected signal indicative of the momentary value of the load current.

When the duty cycles of the switching elements of the converter are continu ously regulated in dependence of the actual momentary load current, also the load current itself will be regulated to a waveform that is proportional to the waveform of the input voltage. With this waveform of the load current, the content of harmonics in the current is reduced, and the power factor is thus corrected. This power factor correction is achieved without large and heavy inductors as with passive power factor correction, and with a reduced number of components compared to the two-stage power supply that uses two sepa rate converters to correct the power factor. Thus, also the production cost is considerably reduced.

In an embodiment, each one of said first and second electronically controlla- ble switching elements comprises a field effect transistor.

The primary winding of the transformer may comprise said first inductor. In this way, the converter can be implemented with a very low number of com ponents. Alternatively, one end of the primary winding of the transformer may be connected to said second input terminal for said input voltage and the other end of the primary winding connected to said connection point through a second capacitor.

In an embodiment, the control circuit is configured to determine, in each time period of the switching frequency, said second duty cycle as being propor tional to a control signal that is the sum of a first part that is proportional to the input voltage and a second part that is proportional to the continuously detected signal indicative of the momentary value of the load current. The control circuit may further be configured to adjust said first and second duty cycles to compensate for an output voltage reduction due to a load cur rent delivered by the converter. In this way, a better regulated output voltage is achieved. In an embodiment, this is achieved when the converter further comprises a feedback circuit configured to detect said output voltage and generate a low pass filtered feedback signal in dependence of said detected output voltage, and the control circuit is further configured to determine said first and second duty cycles in dependence of said low pass filtered feedback signal.

The output circuit may comprise a third capacitor, a first diode, a second di ode and a buffer capacitor, wherein the third capacitor is connected between one end of the secondary winding and the cathode of the first diode; the an- ode of the first diode is connected to the other end of the secondary winding and to the first output terminal for said output voltage; the second diode is connected with its anode to the cathode of the first diode and its cathode to the second output terminal for said output voltage; and the buffer capacitor is connected between the first and the second output terminal for said output voltage. By using an output circuit with only one buffer capacitor, high cur rents between buffer capacitors caused by the waveform of the input voltage are avoided.

A power supply may comprise a bridge rectifier and a switched-mode power converter as described above. In this way, the power supply benefits from the described advantages of the circuit.

An actuator system may comprise at least one power supply as described above; at least one linear actuator connected to and supplied from said pow- er supply, each linear actuator comprising a reversible electric DC motor; a spindle driven by said reversible DC motor; and a spindle nut mounted on the spindle and secured against rotation, said spindle nut being arranged to be moved between two end positions; a controller; and at least one driver circuit being configured to drive the at least one linear actuator under control of the controller. In this way, also the actuator system benefits from the described advantages. Brief Description of the Drawings

Embodiments of the invention will now be described more fully below with reference to the drawings, in which

Figure 1 schematically shows an example of a linear actuator,

Figure 2 shows an example of an actuator system, where a linear actuator is powered by a power supply,

Figure 3 shows an example of a known switched-mode power converter,

Figure 4 shows an example of waveforms of voltages and currents in the converter of Figure 3,

Figures 5a and 5b show an example of how the duty cycles of the converter of Figure 3 can be controlled in dependence of the input voltage,

Figure 6 shows another example of waveforms of voltages and currents in the converter of Figure 3, Figure 7 illustrates how the duty cycles of the converter of Figure 3 can be controlled differently from Figures 5a and 5b,

Figure 8 shows an example of a modified version of the switched-mode pow er converter of Figure 3,

Figure 9 shows an example of waveforms of voltages and currents in the converter of Figure 8, Figure 10 shows another example of waveforms of voltages and currents in the converter of Figure 8,

Figure 1 1 shows the input voltage, the duty cycles, the load power, and the primary side load current as functions of the time for the converters of Fig ures 3 and 8, when the input voltage is maintained at two different levels,

Figure 12 shows an example of a control circuit for the converters of Figures 3 and 8,

Figure 13 shows waveforms of the input voltage, the duty cycles, the load power, and the primary side load current as functions of the time for the con verters of Figures 3 and 8, when the converters are supplied with a rectified sine-shaped voltage,

Figure 14 shows the converter of Figure 8 modified with power factor correc tion according to the invention,

Figure 15 shows the converter of Figure 3 modified with power factor correc- tion according to the invention,

Figure 16 shows an example of a control circuit for the converters of Figures 14 and 15, Figure 17 shows waveforms of the input voltage, the duty cycles, the load power, and the primary side load current as functions of the time for the con verters of Figures 14 and 15,

Figure 18 shows the converter of Figure 14 modified with a feedback circuit for regulating the output voltage to compensate for a voltage drop caused by the load current,

Figure 19 shows a feedback circuit for use in the converter of Figure 18, Figures 20a and 20b show an example of how the duty cycles of the convert er of Figure 18 can be controlled in dependence of the input voltage, Figure 21 shows waveforms of the input voltage, the duty cycles, the load power, and the primary side load current as functions of the time for the con verter of Figure 18,

Figure 22 shows an alternative circuit for regulating the output voltage to compensate for a voltage drop caused by the load current, and

Figure 23 shows the converter of Figure 18 with a modified output circuit.

Detailed Description

Figure 1 schematically shows an example of a linear actuator 1. The linear actuator 1 comprises a reversible electric motor 2, a transmission or reduc tion gear 3, typically with several stages, a spindle 4 having a thread 5, a spindle nut 6 engaging the thread 5 and a tube-shaped activation element 7. At the end of the activation element 7, a mounting bracket 8 for mounting the linear actuator 1 to e.g. a carrying element is arranged. The spindle nut 6 is secured against rotation. In some linear actuators, the spindle nut is con nected directly to e.g. a carrying element without the use of an activation el ement. When the spindle 4 is rotated by the motor 2, the spindle nut 6 moves along the spindle 4, thus transforming the rotation to a linear movement of the spindle nut 6 and/or the activation element 7 between two end positions. It is noted that with some motor types, the reversible electric motor 2 can drive the spindle 4 directly, so that the transmission 3 can be avoided. Alt hough other types of electric motors may be used, the reversible electric mo tor 2 is typically a reversible electric DC motor.

Typically, a linear actuator is used in an actuator system controlled by a con trol box. An example of such an actuator system 10 is illustrated in Figure 2. Via a cable 12, the linear actuator 1 is connected to a control box 13 that l l comprises at least a power supply 14, a controller 15 and a driver circuit 16 for the linear actuator 1. The cable 12 between the driver circuit 16 and the linear actuator 1 may have a length of up to two meters or more. The driver circuit 16, and thus also the electric motor 2 of the actuator 1 , is controlled by control signals from the controller 15. Typically, the controller 15 comprises a microcomputer. The control box 13 is normally placed on the equipment on which the linear actuator 1 is used. This equipment can represent any one of several different applications, such as trucks, agricultural machinery, indus trial automation equipment, hospital and care beds, leisure beds and chairs, tables or other articles of furniture with adjustable height and several other similar applications. The power supply 14 is typically connected to a mains AC supply net with a power cable 17. Finally, the control box 13 is connected to a remote control 18 allowing the operation of the linear actuator 1 to be controlled by a person in the vicinity of the actuator. The connection between the remote control 18 and the control box 13 may be a wired connection as shown in Figure 2, but a wireless communications system, such as a radio link or an infrared link, may also be used.

The speed of the electric motors of the linear actuator 1 can be controlled by adjusting the DC voltage level supplied to the motor, or it can be controlled by using pulse width modulation (PWM), where the motor speed is instead controlled by adjusting the duty cycle of the pulse width modulation.

The current consumption of the electric motor of a linear actuator of course depends of the load carried by the actuator, but typically, it will be in the range of up to 5 A. A control box may also be configured to control an actua tor system having a plurality of linear actuators. As an example, an actuator system may have three linear actuators controlled by one control box com prising the power supply 14, the controller 15 (as in Figure 2) and a driver circuit for each linear actuator. Each driver circuit, and thus also the electric motors of the actuators, are controlled individually by control signals from the controller 15, which means that some or all of the actuator motors may be running simultaneously. Thus, in this case the power supply 14 must be ca pable of delivering up to 15 A, or even more if a higher number of linear ac tuators are driven by one control box.

The power supply 14 can be implemented in different ways. It can be a con- ventional power supply based on a transformer and a rectifier in the form of a rectifier bridge and a capacitor for smoothing the voltage. Such power sup plies are very robust and reliable, but they are also heavy and voluminous and have a relatively low efficiency. Alternatively, a switched-mode power supply can be used, which has a higher efficiency and a considerably smaller magnetic circuit due to an operating frequency typically in the range of 30 - 100 kHz. Switched-mode power supplies exist in different circuit topologies, such as flyback, boost, buck, SEPIC and forward converters.

As an example of a switched-mode power supply, Figure 3 shows a diagram of the power supply 14 implemented with a converter 20 that has been de scribed in WO 2013/004232. The AC voltage from the mains supply is recti fied in a bridge rectifier Di and buffered in a capacitor C-i, before it is fed to the converter 20 as its input voltage Vin. The AC mains supply voltage differs between different parts of the world. Considering tolerances, the voltage lev- els may range from approximately 90 V to approximately 264 V. The rectified voltage at the capacitor Ci is 1.41 times higher and may thus approximately range between 125 V and 375 V.

The input voltage Vin to the converter 20 is applied to a pair of input termi- nals, and its output voltage Vout is delivered from a pair of output terminals. In this context, the word“terminal” means a point at which the converter can be or is connected to other circuitry.

The converter 20 comprises an inductor Li, two controllable switching ele- ments, here shown in the form of the two MOSFET transistors Mi and M2, two capacitors C2 and C3 and a transformer Tr having a primary winding L2 and a secondary winding L3. It is noted that instead of the MOSFET transis tors Mi and M2, other types of controllable switching elements, e.g. bipolar transistors, can be used. On the secondary side, the secondary winding l_3 is connected through two diodes D2 and D3 to two capacitors C4 and Cs. The two MOSFETs Mi and M2 are shown with their built-in body diode. A pulse width modulation control circuit 21 controls the two MOSFETs Mi and M2 to operate opposite of each other, i.e. when one of them is on, the other one is off. A small dead band may be inserted to prevent them from being on simul taneously. The duty cycle of the pulse width modulation is controlled by the control circuit 21. The duty cycle DMI is the fraction or percentage of the peri od T in which MOSFET Mi is on. Since the two MOSFETs Mi and M2 are controlled to operate opposite of each other, the duty cycle DM2, i.e. the frac tion or percentage of the period T in which MOSFET M2 is on, is 100 % - DM-I . A typical modulation frequency can be 50 kFIz corresponding to a period T of 20 ps. Figure 4 shows an example of waveforms of voltages and currents in the cir cuit of the converter 20 when the converter is operating. In the example, the input voltage Vin is 375 V and the duty cycle DMI is 25 %. During operation, the capacitor C2 will be charged to a voltage Vc2, which, as it will be de scribed, is higher than the input voltage Vin. First, the operation is described when the converter 20 is unloaded, i.e. no load current is drawn on the sec ondary side of the converter. This situation is shown in the left-hand part of Figure 4. The voltage VA in the point A in Figure 3 will change between zero (ground) when MOSFET Mi is on and Vc2 when MOSFET M2 is on. In aver age, the voltage VA equals the input voltage Vin, and therefore, the two shad- ed areas in Figure 4 are equal. Thus, in this case the voltage V 02 will be 500 V.

The current lu in the inductor Li is determined by the formula V = L dl/dt, where L is the self-inductance of the inductor Li and V is the voltage over the inductor. Thus as shown, In will increase when MOSFET Mi is on and de crease when MOSFET M2 is on. Since the converter is unloaded, the aver age value of In will be zero, and thus it will change sign in the middle of the period where MOSFET Mi is on, as well as in the middle of the period where MOSFET M2 is on. It is noted that if the average value of the voltage VA was not equal to the input voltage Vin, i.e. if the two shaded areas in Figure 4 were not equal, the average value of the current ILI would increase or de crease depending on the sign of the difference, because the average voltage across inductor Li would not be zero.

The voltage over the capacitor C3 will be equal to Vin, and therefore, the volt age VB in point B, which is also the voltage over the primary winding L2 of the transformer Tr, will be -Vin when MOSFET Mi is on and Vc2-Vin when MOSFET M2 is on. With Vin = 375 V, VB will thus switch between -375 V and 125 V. The transformer Tr can be considered as an ideal transformer with an inductance in parallel with its primary winding. This inductance represents the magnetizing current of the transformer. Therefore, the current li_2 in the pri mary winding L2 of the transformer will be the sum of the magnetizing current lL2,mg in the parallel inductor and the load current li_2,id in the primary winding of the ideal transformer. As shown in Figure 4, the magnetizing current li_2,mg will increase and decrease similarly to lu, while the load current li_2,id will be zero as long as the converter is unloaded. When MOSFET Mi is on, the cur rent IMI will be ILI -IL2, and similarly, the current Ic2 will be ILI -I L2, when MOSFET M2 is on.

When the converter 20 is loaded at the secondary side, a load current li_2,id will run in the primary winding of the ideal transformer. This situation is shown in the right-hand part of Figure 4. The waveform of this current is de termined by the voltage across the primary winding, i.e. VB, and the current IL2 will now be the sum of the magnetizing current li_2,mg and this load current li_2,id. The result is that the average value of lu will now increase to a value I load representing the load current drawn from the input voltage Vin. The value I load depends on secondary load, i.e. it is proportional to the load current on the secondary side. Flowever, as it is seen below, it also depends on the in put voltage Vin. The output voltage Vout at the secondary side of the converter 20 of course depends of the turns ratio of the transformer Tr. If, as an example, the turns ratio is 12.5:1 , capacitor C will be charged to 10 V and capacitor Cs to 30 V, so that the total output voltage Vout is 40 V. If the input voltage Vin remains approximately constant, which will often be the case, the duty cycle can also be maintained at the same value, which results in an approximately constant output voltage. If a different output voltage is wanted from the same input voltage, the duty cycle can be adjusted correspondingly. In many situations, variations in the input voltage Vin, or different input volt age levels, may occur, e.g. when a power supply should be able to be sup plied from a European mains AC supply net as well as from an American mains AC supply net. As mentioned above, the input voltage Vin will typically be in the range from 125 V to 375 V. In that case, the duty cycle can be ad- justed in dependence of the input voltage Vin as illustrated in Figure 3, where Vin is shown as an input signal to the control circuit 21. An example of this is illustrated in Figure 5a, where the duty cycle DMI is adjusted linearly from 100 % at 0 V input voltage to 0 % at a maximum input voltage, which is here set to 500 V. Thus at any input voltage in this range, the duty cycle DMI is deter- mined as DMI = (1 - Vm/500 V) 100 %. As mentioned above, since the two MOSFETs Mi and M2 are controlled to operate opposite of each other, the duty cycle DM2 is 100 % - DM-I , and thus the duty cycle DM2 is determined as DM2 = (Vin/500 V) 100 %. This is illustrated in Figure 5b. It can be seen that when the duty cycle is adjusted in this way, the voltage Vc2 at the capacitor C2 will be maintained at 500 V independently of the input voltage. This means that also the amplitude of the voltage over the primary winding L2 of the transformer Tr and the output voltage Vout at the secondary side of the converter are maintained constant for different values of the input voltage Vin. This is illustrated in Figure 6, which shows waveforms corresponding to those of Figure 4, but with an input voltage Vin = 125 V and a duty cycle DMI = 75 % according to Figure 5a. It can be seen that the voltage Vc2 at the capacitor C2 is maintained at 500 V, and also that the value load representing the load cur- rent drawn from the input voltage Vin is now three times higher than in Figure 4, which is consistent with the fact that the same amount of power should now be delivered from an input voltage of 125 V instead of 375 V. Thus, when the load on the secondary side of the converter is kept constant, the load current on the primary side will change inversely proportionally to the input voltage. It is also noted that in this situation, capacitor C will be charged to 30 V and capacitor Cs to 10 V, and thus the total output voltage Vout is still 40 V. As mentioned above, one way of controlling the duty cycle is to adjust it line arly as illustrated in Figures 5a and 5b. When looking at Figures 4 and 6, it was noted that in that case, the two shaded areas in the waveform of the voltage VA in will always be equal in size when the duty cycle is adjusted in dependence of the input voltage Vin according to Figures 5a and 5b.

Flowever, the duty cycle can also be adjusted differently, but the two shaded areas still need to be equal in size. Thus, it can be seen that if the duty cycle DMI is changed while the input voltage Vin is maintained at a given value, e.g. 375 V as in Figure 4, the size of the lower shaded area in Figure 4 will change correspondingly, and therefore, also the size of the upper shaded area must change. Consequently, it can be seen that the voltage Vc2 at the capacitor C2 will change proportionally to the duty cycle DM-I . This is also il lustrated in Figure 7, where the duty cycle can be adjusted along the vertical line at Vin = 375 V. Thus as an example, it is seen that if the duty cycle at Vin = 375 V is increased from 25 % to 37.5 %, the voltage Vc2 at the capacitor C2 will change from 500 V to 600 V, because a line from 100 % at Vin = 0 V through this point intersects the x-axis at 600 V.

Similarly, it can be seen that if the input voltage Vin is changed while the duty cycle is maintained at a given value, the voltage Vc2 at the capacitor C2 will change proportionally to the input voltage Vin. This is also illustrated in Figure 7, where the duty cycle can be adjusted along the horizontal line at DMI = 25 %. Thus, the voltage V 02 at the capacitor C2 will always be determined by the input voltage Vin and the selected duty cycle. If one of these parameters is changed, Vc2 will change correspondingly.

Figure 8 shows a diagram of the power supply 14 with a different embodi ment 22 of the converter. Looking at Figures 3, 4 and 6, it can be seen that the voltage VB across the primary winding L2 of the transformer Tr is the same as the voltage across the inductor Li. Therefore, the inductor Li can be used as the primary winding of the transformer Tr instead of L2. In this way, one inductor and the capacitor C3 can be saved without affecting the function of the circuit. This is shown in Figure 8, where the transformer Tr has the primary winding Li and the secondary winding L3. As mentioned, the function of the converter 22 of Figure 8 is the same as the function of the converter 20 of Figure 3. Waveforms of voltages and currents in the converter 22 when the converter is operating are shown in Figures 9 and 10 for Vin = 375 V and Vin

= 125 V, respectively. As it was the case for converter 20 of Figure 3, it can be seen that the load current on the primary side of converter 22 will change inversely proportionally to the input voltage, when the load on the secondary side of the converter is kept constant, provided that the duty cycle is adjusted according to Figures 5a and 5b.

The fact that the load current on the primary side of converter 20 or 22 will be inversely proportional to the input voltage, when the load on the secondary side of the converter is kept constant, is illustrated in Figure 1 1 , which shows the input voltage Vin, the duty cycles DMI and DM2, the load power Pioad, and the primary side load current lioad as functions of the time t, when the input voltage Vin is maintained at 375 V and 125 V, respectively. These values could of course also be shown for any other input voltage in between these voltage values.

The pulse width modulation control circuit 21 can be implemented in many different ways. The circuit may operate at a fixed modulation frequency and change the duty cycle in dependence of the input voltage Vin, or the modula tion frequency as well as the duty cycle may be variable. One example of a control circuit 21 , which is illustrated in Figure 12, is based on an operational amplifier 31 coupled as an inverting integrator. The non-inverting input of the amplifier 31 is connected to the input voltage Vin through a voltage divider comprising the two resistors Ri and R2. The inverting input is connected through a resistor R₄ to ground and through a resistor R3 to a controllable switch 32, which is arranged to switch between two positions. In one position, R3 is connected to ground and in the other position R3 is connected to a posi- tive voltage V₊, e.g. 12 V. A capacitor C10 connects the output of the amplifier 31 to the inverting input. The amplifier output is also connected to the input of a Schmitt trigger inverter 33, and the output of the Schmitt trigger inverter 33 is used for controlling the MOSFET Mi. Since the two MOSFETs Mi and M2 should be controlled to operate opposite of each other, i.e. when one of them is on, the other one is off, the output of the Schmitt trigger inverter 33 is also inverted in an inverter 34 and used for controlling the MOSFET M2. A driver 35 adapts the voltage levels of the inverters 33 and 34 to the voltage levels needed to drive the MOSFETs Mi and M2. The output of the Schmitt trigger inverter 33 is also used for controlling the controllable switch 32. When the output of the Schmitt trigger inverter 33 is high, the controllable switch 32 connects R3 to ground, and when it is low, the switch connects R3 to the posi tive voltage V₊.

The voltage level at the non-inverting as well as the inverting input of the op- erational amplifier 31 will be R2/(RI + R2) Vin. When R3 is connected to ground through the controllable switch 32, the output voltage of the opera tional amplifier 31 will increase linearly from the negative-going threshold voltage of the Schmitt trigger inverter 33 towards the positive-going threshold voltage. During this time, which is inversely proportional to Vin, the output of the Schmitt trigger inverter 33 will be high and ensure that the MOSFET Mi is kept in its on-state and the MOSFET M2 in its off-state. When the increasing voltage at the output of the operational amplifier 31 reaches the positive going threshold voltage of the Schmitt trigger inverter 33, the Schmitt trigger inverter output will switch to low, and R3 will now be connected to the positive voltage V₊ through the controllable switch 32. The output voltage of the oper ational amplifier 31 will now decrease linearly from the positive-going thresh old voltage of the Schmitt trigger inverter 33 towards the negative-going threshold voltage. During this time, which depends on Vin as well as V₊ and increases with Vin, the output of the Schmitt trigger inverter 33 will be low and ensure that the MOSFET M2 is kept in its on-state and the MOSFET Mi in its off-state. When the decreasing voltage at the output of the operational ampli fier 31 reaches the negative-going threshold voltage of the Schmitt trigger inverter 33, the Schmitt trigger inverter output will switch to high, and R3 will now again be connected to ground through the controllable switch 32.

Calculations of the duty cycle generated by this control circuit will show that the duty cycle DM2 is proportional to the input voltage Vin in accordance with Figure 5b, and correspondingly, the duty cycle DMI depends on the input voltage V_n in accordance with Figure 5a. The calculations also show that the duty cycle DM2 is inversely proportional to the positive voltage V₊, which, however, is here a fixed voltage. Examples of component values for the circuit of Figure 12 can be R-i = 1.2 MW, R₂ = 2.2 kQ, R₃ = 100 kQ, R = 33 kQ and C10 = 220 pF.

It is noted that with the control circuit shown in Figure 12, the duty cycle of the converters 20 and 22 is only regulated in dependence of the input voltage Vin. There is no feedback of the actual output voltage on the secondary side of the converter. The output voltage may therefore vary in dependence of the load current due to a certain inner resistance in e.g. the transformer, but in many applications, this is fully acceptable. In Figures 3 and 8, the AC voltage from the mains supply is rectified in a bridge rectifier D1 and buffered in a storage capacitor C-i, before it is fed to the converter 20 or 22. The capacitance of the capacitor C1 is selected so that the voltage across it, i.e. the input voltage Vin supplied to the converter, can be considered as a DC voltage approximately equal to the peak voltage of the rectified AC voltage. As an example, a capacitor of 330 pF can be used for a power supply designed to deliver 300 W. This means that current is drawn from the AC voltage only at the peaks of the input waveform, where capacitor Ci is being charged, and these current pulses must contain enough energy to sustain the load until the next peak. This leads to high ratios of peak-to-average input current, and thus to a low power factor because of the presence of harmonics in the input current.

However, as it will be described below, the converter described above can also be designed with power factor correction in order to improve the power factor. With power factor correction, the converter should ideally represent a load that emulates a pure resistor, so that the current drawn from the mains net has the same waveform as the input voltage, i.e. normally a sine wave, and is in phase with the voltage.

Instead of supplying a DC voltage from the capacitor Ci as the input voltage Vin to the converter 20 or 22, the converter can be supplied with a rectified AC voltage, which can be achieved by omitting the capacitor Ci or by replac ing it with a smaller high frequency bypass capacitor that allows the input voltage to follow the rectified half-sine wave. For a power supply designed to deliver 300 W, a capacitor of e.g. 1 pF or even less could be used. If this input voltage is supplied to the converter 20 or 22 having its duty cycles DMI and DM2 controlled according to Figures 5a and 5b, e.g. by the pulse width modulation control circuit 21 shown in Figure 12, the duty cycles will vary according to the waveform of the input voltage. This is illustrated in Fig ure 13, which shows waveforms of the input voltage Vin, the duty cycles DMI and DM2, the load power Pioad, and the primary side load current lioad as func tions of the time t, when the peak value of the input voltage Vin is 375 V and 125 V, respectively. Similarly to Figure 1 1 , these waveforms could of course also be shown for any other peak value in between these voltage values. The waveforms in Figure 13 are shown for the situation where the AC voltage from the mains supply has a frequency of 50 Hz, so that a half period of the sine wave has a duration of 10 ms. For a 60 Hz system, the half period will correspondingly be 8 ½ ms.

Since the pulse width modulation control circuit 21 , as described above, will maintain the voltage Vc2 at the capacitor C2, and thus also the output voltage Vout and load power Pioad, constant over the variations in Vin, the load current I load on the primary side of converter 20 or 22 will change inversely propor- tionally to the input voltage. This means that the current lioad will have a min imum at the top of the sine waveform for the input voltage Vin, and when the waveform for Vin is close to zero, the current lioad will increase to a high value, which in Figure 13 is illustrated with the dots at the waveform for lioad. This is far from the desired situation, where lioad should have a waveform more or less similar to and in phase with Vin. Below, it is described how the desired waveform can be achieved by realising that the load current can actually be controlled by small adjustments of the duty cycle.

In relation to the converter 20 in Figure 3 or the converter 22 in Figure 8, it was mentioned above that when the duty cycles DMI and DM2 are controlled according to Figures 5a and 5b, e.g. by the pulse width modulation control circuit 21 shown in Figure 12, the voltage Vc2 at the capacitor C2 will always be determined by the input voltage Vin and the selected duty cycle. If one (or both) of these parameters is changed, Vc2 will change correspondingly.

However, due to the size of the capacitor C2, there will be a certain delay be fore the voltage Vc2 follows a change in e.g. the duty cycle. This delay will at least be several periods T of the switching frequency for the converter. Thus, it is noted that it is possible to change the duty cycle for a short time, e.g. one or a few periods T, without a corresponding change in V 02, and during this short time, the two shaded areas in the waveform of the voltage VA in Figures 4, 6, 9 and 10 will not be equal in size. As a result of this, the average value of the current lu (i.e. the load current lioad) will increase or decrease corre- spondingly during this short time, and thus it is possible to affect this current by making small adjustments of the duty cycle. This current can also be seen as the current used to charge or discharge C2 towards a voltage Vc2 corre sponding to the changed duty cycle. This means that even though the duty cycles DMI and DM2 are in principle determined in each period T in depend ence of the input voltage Vin as illustrated in Figure 13, they may be adjusted a little bit up or down in order to affect the load current lioad.

This can also be seen in a different way. Looking at Figures 3 and 8, it can be seen that if the capacitor C2 is considered to be large enough to represent a fixed voltage Vc2, at least during one or a few periods T, the MOSFETs Mi and M2 and the inductor Li can be seen as forming a buck converter ar ranged in the reverse direction, i.e. a buck converter converting the voltage Vc2 to a virtual voltage at the left hand side of inductor Li. This virtual voltage is determined as DM2 Vc2. If this virtual voltage is different from the input voltage Vin, a current will run in the direction from the highest voltage to the lowest voltage. Thus if, for a given value of Vin and with a maintained value of Vc2, the duty cycle DM2 is reduced for one or a few periods T, the virtual volt age will also be reduced, and consequently the load current lioad will increase. Similarly, if the duty cycle DM2 is increased for one or a few periods T, the virtual voltage will also be increased, and consequently the load current l ioad will decrease.

In other words, this means that at a given time, e.g. within a few periods T, the size of the load current lioad can be controlled by adjusting the duty cycle correspondingly.

Figure 14 illustrates how such an adjustment can be implemented in the con verter 22 by adjusting the duty cycle in dependence of the actual load current lioad. A similar implementation can of course be made in the converter 20 of Figure 3, as it is shown in Figure 15. A current measuring resistor Rn having a low and well-defined resistance, e.g. 0.1 W, is inserted in series with the converter. The voltage drop across R11 is proportional to the current flowing through it, so that the voltage drop directly indicates the value of current lu and thus also the load current lioad. This voltage drop can thus be used as an input signal Vcurr to a modified pulse width modulation control circuit 41 to gether with the input voltage Vin.

Figure 16 shows an example of how the pulse width modulation control cir cuit 41 can be implemented. The circuit 41 is similar to the circuit 21 of Figure 1 2, except that the input signal to the amplifier 31 is now generated by an operational amplifier 36 coupled as a differential amplifier, which combines the input signals Vin and V curr.

The non-inverting input of the amplifier 36 is connected to the input voltage Vin through a voltage divider comprising the two resistors Ri and Fte as it was the case for the non-inverting input of the amplifier 31 in Figure 12. The in- verting input is connected through a resistor R13 to the voltage Vcurr repre senting the load current and to the output of the differential amplifier through a resistor RI ₄. It is noted that the current in the current measuring resistor Rn is actually the current lu shown in Figures 4, 6, 8 and 9, i.e. a current chang ing its direction two times each period T. To get a better indication on the load current lioad, which is the average value of lu, it may therefore be advan tageous to low pass filter the voltage Vcurr before it is applied to the inverting input of the amplifier 36 through the resistor R13. This low pass filter should smooth the fast variations in lu, but still allow variations in lioad over a few periods T to be detected.

The output voltage Vdiff of the differential amplifier 36 is

Since the voltage Vcurr is negative for a positive load current, Vdiff consists of a part that is proportional to the input voltage Vin plus a part that is propor tional to the actual load current. As in Figure 12, the duty cycle DM2 generated by this control circuit is propor tional to the voltage level at the non-inverting input of the amplifier 31. In Fig ure 1 2, this voltage level was proportional to the input voltage Vin. Now, this voltage level, and thus also the duty cycle DM2, is instead proportional to Vdiff, which means that it is proportional to the input voltage Vin plus a contribution from the actual (momentary) load current.

This means that if the load current I load at a given time for some reason in creases, the duty cycle DM2 is also increased, which causes the virtual volt- age from the reverse buck converter to increase and thus counteract the in crease in the load current load. This is illustrated in Figure 1 7, where the waveforms of Figure 13 are shown with a thin line for comparison. As an ex ample, at time t-i, the load current lioad would, if the duty cycle was controlled by the control circuit 21 of Figure 1 2, have a high value as illustrated with the thin waveform (corresponding to Figure 1 3) due to the input voltage Vin being at a low level at this time. Flowever, with the control circuit 41 , the load cur rent at time ti will be reduced to a much lower value that is proportional to the low voltage level of the input voltage Vin. Thus by continuously detecting the actual (momentary) load current lioad and adjusting the duty cycle not only in dependence of the waveform of the input voltage Vin, but also in dependence of the detected actual (momentary) load current lioad, it is achieved that the load current lioad is at any time adjusted to have the same waveform as the input voltage Vin. This is illustrated with the thicker waveform for l ioad in Figure 1 7. With this waveform of lioad, the content of harmonics in the current is re- duced, and the power factor is corrected.

Since the load current lioad is now controlled to have the form of a rectified sine wave, this will also be the case for the voltage Vcurr and consequently also for the voltage Vdiff and the duty cycle DM2, as it is shown in Figure 1 7.

The fact that the waveform for the load current lioad is now regulated as de scribed above of course means that the power Pioad delivered by the convert er can no longer be kept constant. Instead, it varies as shown in Figure 17. However, since the average power still needs to be the same, the peak pow er will now be twice the average power. The variation in delivered power will necessarily cause a certain ripple in the voltage Vc2 at the capacitor C2 and at the output voltage of the converter. However, this can be compensated as described later.

Figure 17 also illustrates that the delivered power will still be the same for different amplitudes of the input voltage Vin, and thus e.g. a lower amplitude of the input voltage Vin results in a higher amplitude of the load current lioad. Figure 17 shows this for Vin, peak equal to 375 V and 125 V, respectively. This means that also the adjustment of the duty cycle due to the voltage Vcurr will be relatively higher for lower input voltage levels, as it is also shown.

Further, it is noted that the amplitude of the load current l ioad of course also depends on the actual load on the secondary side of the converter. If this load is changed, the amplitude of the load current lioad will change corre spondingly.

With the control circuit 41 of Figure 16, the duty cycles DMI and DM2 are regu- lated in dependence of the input voltage Vin and the load current lioad. How ever, similarly to the control circuit 21 shown in Figure 12, there is no feed back of the actual output voltage Vout on the secondary side of the converter. The output voltage may therefore also here vary in dependence of the load current due to a certain inner resistance in e.g. the transformer.

Figure 18 shows an example of how the converter 22 can be modified to pro vide such a feedback and thus minimize the variations in the output voltage Vout of the converter in dependence of the load current. A feedback circuit 42 senses the output voltage Vout on the secondary side of the converter and provides a control signal that can be used as the positive voltage V+ in the control circuit 41. As mentioned previously, the duty cycle DM2 generated by the control circuit 41 , as well as the control circuit 21 , will be inversely propor- tional to the positive voltage V₊, and thus this voltage can be used for adjust ing the duty cycle in dependence of the output voltage V_out.

Figure 19 shows an example of how the feedback circuit 42 can be imple- mented. The circuit is based on an operational amplifier 37 coupled as an inverting integrator. The non-inverting input of the amplifier 37 is connected to a reference voltage generated by dividing a positive supply voltage VDD, e.g. 5 V or 12 V, in a voltage divider comprising the two resistors R21 and R22. The inverting input is connected through a resistor R28 to a signal that de- pends on the output voltage Vout as it will be described below. A capacitor C20 connects the output of the amplifier 37 to the inverting input.

An optocoupler 38 provides isolation between the secondary side and the primary side of the converter. A series connection of a resistor R23, a resistor R24 and a Zener diode Z21 is connected across the output voltage Vout with the light emitting diode of the optocoupler 38 arranged in parallel with the resistor R23. In this way, an increase in the output voltage Vout causes an in crease in the current through the light emitting diode, and thus also the amount of light emitted by the diode. The phototransistor of the optocoupler 38 has its collector terminal connected to the positive supply voltage VDD, or to another positive voltage, while its emitter terminal is connected to ground through a voltage divider comprising the two resistors R25 and R26. An in crease in the light received by the phototransistor increases the current con ducted by the phototransistor through the voltage divider and thus also the voltage at the midpoint between resistors R25 and R26. This midpoint voltage is then low-pass filtered in a low-pass filter comprising a resistor R27 and a capacitor C21 . The cut off frequency of the low-pass filter should be sufficient ly low to prevent voltage variations originating from the frequency of the mains voltage to occur at the capacitor C21 . The output from the low-pass filter, i.e. the voltage at the capacitor C21 , is then connected to the inverting input of the operational amplifier 37 through resistor R28. The component values are selected so that when the output voltage Vout is at its nominal value, the voltage at the capacitor C21 is equal to the voltage at the non-inverting input of the operational amplifier 37, and no current will run in the resistor R28. In this situation, the voltage over the capacitor C20 will re- main constant, and this will thus also be the case for the output of the opera tional amplifier 37 and the voltage V₊.

The function of the feedback circuit 42 can be described as follows. When the converter 22 is unloaded on the secondary side, and has been so for some time so that the voltage at the capacitor C20 has been stabilized, the current in R28 will be zero, and the circuit will have adjusted the voltage V₊ to a value that causes the control circuit 41 to adjust the duty cycle as described above in relation to Figure 16. In other words, the duty cycle is adjusted in dependence of the input voltage according to the thick line in Figures 20a and 20b, i.e. as it was also shown in Figures 5a and 5b. This is also illustrat ed in the left hand side of Figure 21 showing waveforms when the peak value of the input voltage Vin is equal to 375 V.

Flowever, when a load is then connected to the output of the converter, the output voltage Vout will decrease due to the load current and the inner re sistance in e.g. the transformer. As a result of this voltage decrease, the cur rent in the light emitting diode as well as the phototransistor of the optocou- pler 38 will decrease, and this will then also be the case for the voltage at the capacitor C21 . A current will therefore start to run in resistor R28 in the direc- tion from the inverting input of the amplifier 37 to the capacitor C21 , and the capacitor C20 will thus be charged to a higher voltage, which means that the voltage V₊ will increase. The increased voltage V₊ will cause the control cir cuit 41 to reduce the duty cycle DM2, because, as mentioned before, this duty cycle is inversely proportional to the positive voltage V₊. The change in duty cycle is illustrated in the right hand side of Figure 21 .

Due to the low-pass filter comprising R27 and C21, this change will occur suffi ciently slowly to allow the voltage Vc2 at the capacitor C2 to follow the change, and therefore, V 02 and the output voltage Vout will increase corre sponding to the reduced duty cycle DM2. This process will continue until the output voltage Vout has again reached its nominal value, and the current in R28 has decreased to zero. Since the voltage Vc2 at the capacitor C2 now is higher than before, the lines in Figures 20a and 20b according to which the duty cycle is regulated in dependence of the input voltage Vin will now have a different slope, as it is illustrated with the thin lines in Figures 20a and 20b.

Any subsequent change in the output voltage Vout due to a changed load on the secondary side of the converter will cause the feedback circuit 42 and the control circuit 41 to increase or decrease the duty cycle, and thus also change the voltage Vc2, so that the output voltage Vout is maintained at its nominal value. An alternative way of minimizing the variations in the output voltage Vout of the converter caused by the load current due to a certain inner resistance in e.g. the transformer is described below. Instead of regulating the duty cycle and thus the output voltage Vout in dependence of the actual average output voltage via the feedback circuit 42 as described above, it can be regulated in dependence of the actual average load current.

An example of this is shown as the circuit 43 in Figure 22. As for the feed back solution described above, the duty cycle DM2 generated by the control circuit 41 will be inversely proportional to the positive voltage V₊, and thus this voltage can also here be used for adjusting the duty cycle in dependence of the actual average load current.

A signal indicating the actual primary side average load current lioad can be obtained by low pass filtering the signal Vcurr as it is shown with a low pass filter comprising a resistor R31 and a capacitor C31 in Figure 22. The cut off frequency of the low-pass filter should be sufficiently low to prevent voltage variations originating from the frequency of the mains voltage to occur at the capacitor C31. However, as mentioned above, the average load current lioad on the primary side of the converter, for a given secondary side load current, depends on the peak or average value of the input voltage Vin. To deliver the same power to the secondary side, the average load current lioad will be inversely propor tional to the average value of the input voltage Vin. A signal indicative of the secondary side load current, or the delivered power, can therefore be ob tained by multiplying the low pass filtered value of Vcurr by a signal propor tional to average value of the input voltage Vin.

Thus in Figure 22, the input voltage V_n is divided in a voltage divider com prising the two resistors R32 and R33, and the divided voltage is low pass fil tered in a low pass filter comprising a resistor R34 and a capacitor C32 to ob tain a signal proportional to average value of the input voltage Vin. Also here, the cut off frequency of the low-pass filter should be sufficiently low to pre vent voltage variations originating from the frequency of the mains voltage to occur at the capacitor C32. Alternatively, a signal proportional to the peak val ue of the input voltage Vin can be obtained by replacing resistor R34 with a diode and arranging a resistor across capacitor C32 to ensure that the voltage can also follow a decreasing input voltage level.

The two signals are then multiplied in an analog multiplier 44. It is noted that since the signal Vcurr is negative, the multiplier output Vmuit will also be nega tive, and the multiplier 44 must thus be able to handle negative input signals. The multiplier output Vmuit is thus a signal indicative of the secondary side load current, or the delivered power. This signal is connected through a resis tor R35 to the inverting input of an operational amplifier 45 coupled as a dif ferential amplifier. The inverting input of the differential amplifier is also con nected to the output of the differential amplifier through a resistor R36. The non-inverting input of the amplifier 45 is connected to a reference voltage generated by dividing a positive supply voltage VDD, e.g. 5 V or 12 V, in a voltage divider comprising the two resistors R37 and R38. The output voltage V₊ of the differential amplifier 45 is

Since the voltage Vmuit is negative, the output voltage V+ consists of a fixed part determined by VDD plus a part that is proportional to the actual load cur rent. The resistor values of R35, R36, R37 and R38 are chosen so that the fixed part, which is the output voltage when Vmuit is zero, i.e. when the converter is unloaded, will be equal to the value that was used as the positive voltage V+ in the control circuit 41 in Figure 16.

The function of the circuit 43 can be described as follows. When the convert er 22 is unloaded on the secondary side, the output voltage Vmuit of the ana log multiplier 44 will be zero, and the circuit will have adjusted the voltage V+ to the value that causes the control circuit 41 to adjust the duty cycle as de scribed above in relation to Figure 16. In other words, the duty cycle is ad justed in dependence of the input voltage according to the thick line in Fig ures 20a and 20b, i.e. as it was also illustrated in the left hand side of Figure 21 showing waveforms for a peak value of the input voltage Vi_n equal to 375 V.

Flowever, when a load is then connected to the output of the converter, the load current increases. As a result of this, the output voltage Vmuit of the ana log multiplier 44 will now change to a (negative) value different from zero, which means that the voltage V+ will increase. The increased voltage V+ will cause the control circuit 41 to reduce the duty cycle DM2, because, as men tioned before, this duty cycle is inversely proportional to the positive voltage V+. The reduction in duty cycle is illustrated in the right hand side of Figure 21. Due to the low-pass filter comprising R31 and C31 , this change will occur sufficiently slowly to allow the voltage Vc2 at the capacitor C2 to follow the change, and therefore, Vc2 and the output voltage Vout will increase corre sponding to the reduced duty cycle DM2. In this way, the reduction in the out put voltage Vout of the converter caused by the load current due to a certain inner resistance in e.g. the transformer is compensated. Since the voltage Vc2 at the capacitor C2 now is higher than before, the lines in Figures 20a and 20b according to which the duty cycle is regulated in dependence of the input voltage Vin will now have a different slope, as it is illustrated with the thin lines in Figures 20a and 20b.

Any subsequent change in the load on the secondary side of the converter will cause the circuit 43 and the control circuit 41 to increase or decrease the duty cycle correspondingly, and thus also change the voltage Vc2 and the output voltage Vout to compensate for the effects of the changed load.

It was mentioned above that the voltages across the capacitors C4 and C5 in the output circuit depend on the duty cycles. Thus, in the example described above, it was mentioned that when the input voltage Vin is 375 V and the duty cycle DMI is 25 %, capacitor C4 will be charged to 10 V and capacitor C5 to 30 V, so that the total output voltage Vout is 40 V. Similarly, when the input voltage Vin is 125 V and the duty cycle DMI is 75 %, capacitor C4 will be charged to 30 V and capacitor C5 to 10 V. In other words, this means that the voltage of capacitor C4 will be proportional to the duty cycle DMI and the volt- age of capacitor C5 proportional to the duty cycle DM2.

Flowever, when the input voltage and the duty cycles now continuously change with the mains frequency, e.g. as shown in Figures 17 and 21 , this will also be the case for the voltages of the capacitors C4 and C5. Thus as an example, when the peak value of the input voltage Vin is equal to 375 V, ca pacitor C4 will be charged to 10 V and capacitor C5 to 30 V at the top of the input voltage waveform, while capacitor C4 will be charged to 40 V and ca pacitor C5 to 0 V a few ms later, when the input voltage is at zero. This implies that high currents will continuously circulate between the two capacitors, which can be avoided by modifying the output circuit as shown in Figure 23. In Figure 23, one end of the secondary winding l_3 of the transformer Tr is connected to the anode of a diode D₄2 and to the negative terminal of the output voltage V_out. The other end of the secondary winding is connected through a capacitor C41 to the cathode of diode D₄2 and the anode of a diode D_4I . The cathode of diode D_4I is connected to the positive terminal of the output voltage Vout. A capacitor C42 is connected between the positive and the negative terminal of the output voltage Vout.

The voltage at the secondary winding L3 of the transformer Tr is a square wave equal to the voltage at the primary winding divided by the turns ratio of the transformer Tr. The voltage at the primary winding is illustrated as VB in Figures 4 and 6 and Vu in Figures 9 and 10. When the voltage at the sec ondary winding L3 is negative, a current will circulate through L3, the diode D42 and the capacitor C41 , and this current will charge capacitor C41 to the voltage at the secondary winding L3. When the voltage at the secondary winding L3 is positive, a current will circulate through L3, the capacitor C41 , the diode D41 and the capacitor C42, and this current will now discharge ca pacitor C41 into the capacitor C42, which will thus be charged to the peak-to- peak voltage of the secondary winding L3. The voltage to which the capacitor C41 is charged when the voltage at the secondary winding L3 is negative will vary with the duty cycle of the converter, but no high currents occur due to the changes in the duty cycle with the mains frequency. The capacitor C41 can now be a relatively small high frequency capacitor, while C42 will typically be an electrolytic capacitor, whose capacitance is selected to ensure a suffi- ciently low ripple caused by the mains frequency in the output voltage Vout. As an example, for a power supply designed to deliver 300 W, a capacitor of 47 pF can be used as capacitor C41 and a capacitor of 6800 pF can be used as capacitor C42. In other words, there is disclosed a switched-mode power converter 20; 22 for converting an input voltage V_n at a primary side of the converter to an output voltage Vout at a secondary side of the converter, said switched-mode power converter comprising a first inductor Li connected between a first input terminal for said input voltage Vin and a connection point A; a first electroni cally controllable switching element Mi connected between said connection point A and a second input terminal for said input voltage Vi_n; a second elec tronically controllable switching element M2 connected between said connec- tion point A and a first capacitor C2 having its other end connected to said second input terminal for said input voltage Vi_n; a transformer Tr having a primary winding Li ; L2 and a secondary winding L3, wherein said primary winding is connected between said connection point A and one of said first and second input terminals for said input voltage Vi_n; an output circuit con- nected to said secondary winding L3 and arranged to rectify a voltage present on said secondary winding and buffer the rectified voltage in at least one buffer capacitor C4, Cs; C42 connected to a first and a second output terminal for said output voltage V₀ut; and a control circuit 41 configured to generate control signals at a switching frequency for controlling said first electronically controllable switching element Mi with a first duty cycle DMI and said second electronically controllable switching element M2 with a second duty cycle DM2 so that when one of said electronically controllable switching elements is conducting, the other one is not conducting, wherein the control circuit 41 is further configured to determine said first and second duty cycles DM-I , DM2 in dependence of the input voltage Vin. The converter 20; 22 is further config ured to continuously detect a signal V_CUrr indicative of a momentary value of a load current lioad drawn by the converter from said input voltage Vi_n; and the control circuit 41 is further configured to determine, in each time period T of the switching frequency, said first and second duty cycles DM-I , DM2 in de- pendence of the continuously detected signal Vcurr indicative of the momen tary value of the load current lioad.

In an embodiment, each one of said first and second electronically controlla ble switching elements Mi , M2 comprises a field effect transistor.

The primary winding of the transformer Tr may comprise said first inductor Li . In this way, the converter can be implemented with a very low number of components. Alternatively, one end of the primary winding L2 of the trans former Tr may be connected to said second input terminal for said input volt age Vin and the other end of the primary winding L2 connected to said con nection point A through a second capacitor C3.

In an embodiment, the control circuit 41 is configured to determine, in each time period T of the switching frequency, said second duty cycle DM2 as be ing proportional to a control signal Vdiff that is the sum of a first part that is proportional to the input voltage Vin and a second part that is proportional to the continuously detected signal Vcurr indicative of the momentary value of the load current I load.

The control circuit 41 may further be configured to adjust said first and sec ond duty cycles DM-I , DM2 to compensate for an output voltage reduction due to a load current delivered by the converter. In this way, a better regulated output voltage is achieved. In an embodiment, this is achieved when the con verter further comprises a feedback circuit 42 configured to detect said output voltage Vout and generate a low pass filtered feedback signal V+ in depend ence of said detected output voltage, and the control circuit 41 is further con- figured to determine said first and second duty cycles D M-I , DM2 in depend ence of said low pass filtered feedback signal V+. The output circuit may comprise a third capacitor C41 , a first diode D₄2, a second diode D_4I and a buffer capacitor C42, wherein the third capacitor C41 is connected between one end of the secondary winding L3 and the cathode of the first diode D42; the anode of the first diode D42 is connected to the other end of the secondary winding L3 and to the first output terminal for said out put voltage V₀ut; the second diode D41 is connected with its anode to the cathode of the first diode D42 and its cathode to the second output terminal for said output voltage V₀ut; and the buffer capacitor C42 is connected be tween the first and the second output terminal for said output voltage V_out. By using an output circuit with only one buffer capacitor, high currents between buffer capacitors caused by the waveform of the input voltage are avoided.

A power supply may comprise a bridge rectifier D1 and a switched-mode power converter 20; 22 as described above. In this way, the power supply benefits from the described advantages of the circuit.

An actuator system may comprise at least one power supply 14 as described above; at least one linear actuator 1 connected to and supplied from said power supply, each linear actuator comprising a reversible electric DC motor 2; a spindle 4 driven by said reversible DC motor 2; and a spindle nut 6 mounted on the spindle 4 and secured against rotation, said spindle nut 6 being arranged to be moved between two end positions; a controller 15; and at least one driver circuit 16 being configured to drive the at least one linear actuator 1 under control of the controller 15. In this way, also the actuator system benefits from the described advantages.

Although various embodiments of the present invention have been described and shown, the invention is not restricted thereto, but may also be embodied in other ways within the scope of the subject-matter defined in the following claims.

Claims

C l a i m s :

1. A switched-mode power converter (20; 22) for converting an input voltage (Vin) at a primary side of the converter to an output voltage (V_out) at a sec ondary side of the converter, said switched-mode power converter compris ing:

• a first inductor (Li) connected between a first input terminal for said input voltage (Vin) and a connection point (A);

• a first electronically controllable switching element (Mi) connected be tween said connection point (A) and a second input terminal for said input voltage (Vin) ;

• a second electronically controllable switching element (M2) connected between said connection point (A) and a first capacitor (C2) having its other end connected to said second input terminal for said input volt age (Vin);

• a transformer (Tr) having a primary winding (Li ; L2) and a secondary winding (L3), wherein said primary winding is connected between said connection point (A) and one of said first and second input terminals for said input voltage (Vin);

• an output circuit connected to said secondary winding (L3) and ar ranged to rectify a voltage present on said secondary winding and buffer the rectified voltage in at least one buffer capacitor (C4, Cs; C42) connected to a first and a second output terminal for said output volt age (Vout) ; and

• a control circuit (41 ) configured to generate control signals at a switch ing frequency for controlling said first electronically controllable switch ing element (Mi) with a first duty cycle (DM-I ) and said second electron ically controllable switching element (M2) with a second duty cycle (DM2) SO that when one of said electronically controllable switching el ements is conducting, the other one is not conducting, wherein the control circuit (41 ) is further configured to determine said first and sec ond duty cycles (DM-I , DM2) in dependence of the input voltage (Vin), c h a r a c t e r i z e d in that

the converter (20; 22) is further configured to continuously detect a signal (Vcurr) indicative of a momentary value of a load current (load) drawn by the converter from said input voltage (Vin); and

the control circuit (41) is further configured to determine, in each time period (T) of the switching frequency, said first and second duty cycles (DM-I, DM2) in dependence of the continuously detected signal (Vcurr) indicative of the mo mentary value of the load current (lioad).

2. A switched-mode power converter according to claim 1, c h a r a c t e r i z e d in that each one of said first and second electronical ly controllable switching elements (Mi, M2) comprises a field effect transistor.

3. A switched-mode power converter according to claim 1 or 2, c h a r a c t e r i z e d in that the primary winding of the trans former (Tr) comprises said first inductor (Li).

4. A switched-mode power converter according to claim 1 or 2, c h a r a c t e r i z e d in that one end of the primary winding (L2) of the transformer (Tr) is connected to said second input terminal for said input voltage (Vin) and the other end of the primary winding (L2) is connected to said connection point (A) through a second capacitor (C3).

5. A switched-mode power converter according to any one of claims 1 to 4, c h a r a c t e r i z e d in that the control circuit (41) is configured to determine, in each time period (T) of the switching frequency, said second duty cycle (DM2) as being proportional to a control signal (Vdiff) that is the sum of a first part that is proportional to the input voltage (Vin) and a second part that is proportional to the continuously detected signal (Vcurr) indicative of the momentary value of the load current (lioad).

6. A switched-mode power converter according to any one of claims 1 to 5, c h a r a c t e r i z e d in that the control circuit (41) is further con- figured to adjust said first and second duty cycles (DM-I , DM2) to compensate for an output voltage reduction due to a load current delivered by the con verter.

7. A switched-mode power converter according to claim 6, c h a r a c t e r i z e d in that the converter further comprises a feedback circuit (42) configured to detect said output voltage (V_out) and generate a low pass filtered feedback signal (V₊) in dependence of said detected output volt age, and that the control circuit (41 ) is further configured to determine said first and second duty cycles (DM-I , DM2) in dependence of said low pass fil tered feedback signal (V₊).

8. A switched-mode power converter according to any one of claims 1 to 7, c h a r a c t e r i z e d in that said output circuit comprises a third capacitor (C41), a first diode (D₄₂), a second diode (D₄-i) and a buffer capaci tor (C42), wherein

• the third capacitor (C41) is connected between one end of the second ary winding (L3) and the cathode of the first diode (D42);

• the anode of the first diode (D42) is connected to the other end of the secondary winding (L3) and to the first output terminal for said output voltage (V out) ;

• the second diode (D41) is connected with its anode to the cathode of the first diode (D42) and its cathode to the second output terminal for said output voltage (Vout) ; and

• the buffer capacitor (C42) is connected between the first and the sec ond output terminal for said output voltage (Vout) .

9. A power supply (14) comprising a bridge rectifier (D-i) and a switched- mode power converter (20; 22) according to any one of claims 1 to 8.

10. An actuator system (10) comprising:

• at least one power supply (14) according to claim 9; at least one linear actuator (1 ) connected to and supplied from said power supply, each linear actuator comprising:

o a reversible electric DC motor (2);

o a spindle (4) driven by said reversible DC motor (2); and o a spindle nut (6) mounted on the spindle (4) and secured against rotation, said spindle nut (6) being arranged to be moved between two end positions;

a controller (15); and

at least one driver circuit (16) being configured to drive the at least one linear actuator (1 ) under control of the controller (15).