WO2018076192A1 - 一种天线和无线路由器 - Google Patents

一种天线和无线路由器 Download PDF

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Publication number
WO2018076192A1
WO2018076192A1 PCT/CN2016/103335 CN2016103335W WO2018076192A1 WO 2018076192 A1 WO2018076192 A1 WO 2018076192A1 CN 2016103335 W CN2016103335 W CN 2016103335W WO 2018076192 A1 WO2018076192 A1 WO 2018076192A1
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WIPO (PCT)
Prior art keywords
substrate
antenna
radiator
antenna according
screen
Prior art date
Application number
PCT/CN2016/103335
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English (en)
French (fr)
Inventor
苏华峰
张利鹏
陈开宏
王照
何丽娥
Original Assignee
深圳鲲鹏无限科技有限公司
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
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Publication date
Application filed by 深圳鲲鹏无限科技有限公司 filed Critical 深圳鲲鹏无限科技有限公司
Priority to PCT/CN2016/103335 priority Critical patent/WO2018076192A1/zh
Publication of WO2018076192A1 publication Critical patent/WO2018076192A1/zh

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Classifications

    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/36—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith

Definitions

  • the present invention belongs to the field of communications, and in particular, to an antenna and a wireless router.
  • Prior art antennas include radiators, reflectors, directors, isolation elements, and the like.
  • the radiator mainly uses a dipole or a monopole to provide horizontal omnidirectional and vertically polarized radiation performance, and a dipole or a monopole is a radiation having an electrical length of one-half or four-quarters of the wavelength of the antenna.
  • the radiation of the radiation is omnidirectional in the horizontal plane, and the direction of polarization of the radiated electric field is perpendicular to the earth, that is, vertical polarization.
  • This type of radiator is placed outside the antenna when it is used, generally perpendicular to the horizontal plane.
  • the radiator has a narrow bandwidth, low radiation efficiency, high profile, and cannot be planarized, so that the prior art antenna has a large size, a narrow antenna band width, low radiation efficiency, and a high profile.
  • An object of the present invention is to provide an antenna and a wireless router, which are intended to solve the problem that the radiator of the antenna of the prior art has a narrow bandwidth, low radiation efficiency, high profile, and cannot be planarized, thereby making the antenna size of the prior art. Large, narrow antenna bandwidth, low radiation efficiency, and high profile.
  • an antenna comprising a radiator, a director, at least one absorber, and a reflector, the director being located in front of the radiator with respect to a radiation direction of the radiator, The reflector is located behind the radiator, and the absorber is located at the side of the radiator; the radiator comprises a substrate, a composite left and right hand transmission line unit attached to the front surface of the substrate, an electromagnetic field band gap EBG structure attached to the bottom surface of the substrate, and an EBG structure.
  • a metal plate as a ground connected to the composite left and right hand transmission line unit.
  • a wireless router including the antenna described above.
  • the radiator of the antenna includes a composite left and right hand transmission line attached to the front surface of the substrate
  • the element and the EBG structure attached to the bottom surface of the substrate and the metal plate on the EBG structure side connected to the composite right and left hand transmission line unit, so that the radiator has a low profile, and the antenna is miniaturized, and the EBG structure can be eliminated.
  • the surface wave between the left and right hand transmission line unit and the ground is combined to improve the efficiency of the antenna and increase the bandwidth of the antenna.
  • FIG. 1 is a schematic structural view of an antenna provided by an embodiment of the present invention.
  • FIG. 2 is a schematic structural view of an isolator of an antenna according to an embodiment of the present invention.
  • FIG 3 is a schematic structural view of a radiator of an antenna according to an embodiment of the present invention.
  • FIG. 4 is a structure of a composite right and left hand transmission line unit in a radiator of an antenna according to an embodiment of the present invention.
  • FIG. 5 is a schematic structural view of an EBG structure in a radiator of an antenna according to an embodiment of the present invention.
  • FIG. 6 is a schematic structural view of a ground in a radiator of an antenna according to an embodiment of the present invention.
  • FIG. 7 is a schematic structural view showing a composite left and right hand transmission line unit and an EBG structure attached to a front surface of a substrate in a radiator of an antenna according to an embodiment of the present invention.
  • Embodiment 8 is a schematic structural diagram of a director of an antenna according to Embodiment 1 of the present invention.
  • FIG. 9 is a front view of a first capacitive screen and a second capacitive screen of a director of an antenna according to Embodiment 1 of the present invention.
  • FIG. 10 is a bottom view of a first capacitive screen and a second capacitive screen of a director of an antenna according to Embodiment 1 of the present invention.
  • FIG. 11 is a front elevational view of a resonant screen of a director of an antenna according to an embodiment of the present invention.
  • FIG. 12 is a schematic bottom view of a resonant screen of a director of an antenna according to an embodiment of the present invention.
  • FIG. 13 is a front elevational view of a fractal capacitive screen employed in a director of an antenna according to an embodiment of the present invention.
  • FIG. 14 is a front elevational view showing a Jerusalem cross resonant screen employed by the director of the antenna according to the first embodiment of the present invention.
  • FIG. 15 is a schematic structural diagram of an antenna according to Embodiment 2 of the present invention.
  • 16 is a schematic structural diagram of a director of an antenna according to Embodiment 2 of the present invention.
  • 17 is a front elevational view of a zero index lens of a director of an antenna according to an embodiment of the present invention.
  • FIG. 18 is a schematic bottom view of a zero-refractive-index lens of a director of an antenna according to Embodiment 2 of the present invention. [0024] FIG.
  • FIG. 19 is a front elevational view of a capacitive screen of a director of an antenna according to Embodiment 2 of the present invention. [0025] FIG.
  • FIG. 20 is a schematic bottom view of a capacitive screen of a director of an antenna according to Embodiment 2 of the present invention.
  • FIG. 21 is a front elevational view of a resonant screen of a director of an antenna according to Embodiment 2 of the present invention. [0027] FIG.
  • FIG. 22 is a schematic bottom view of a resonant screen of a director of an antenna according to Embodiment 2 of the present invention. [0028] FIG.
  • FIG. 23 is a schematic structural diagram of a reflector of an antenna according to Embodiment 2 of the present invention.
  • 24 is a front elevational view of an absorber of an antenna according to an embodiment of the present invention.
  • 25 is a side view of a microwave absorber of an antenna according to an embodiment of the present invention.
  • 26 is a schematic bottom view of an absorber of an antenna according to an embodiment of the present invention.
  • 27 is a schematic diagram showing the performance of an absorber of an antenna according to an embodiment of the present invention.
  • FIG. 28 is an operation of the 2.4 GHz band between the antennas provided by the first embodiment of the present invention.
  • 29 is a front elevational view showing another type of absorber of an antenna according to an embodiment of the present invention.
  • Embodiment 1 is a diagrammatic representation of Embodiment 1:
  • an antenna according to Embodiment 1 of the present invention includes a radiator 1, a director 2, one or two absorbers 3, and a reflector 4, which are directed to a radiation direction of the radiator 1.
  • the device 2 is located in front of the radiator 1, the reflector 4 is located behind the radiator 1, and the two absorbers 3 are located on both sides of the radiator 1, respectively.
  • the antenna provided by the first embodiment of the present invention may further include one or two isolators 5, and the two isolators 5 are respectively located behind the two absorbers 3 facing away from the radiator 1, or located in two absorbers. 3 faces the front of the radiator 1.
  • the isolator 5 can be made of an I-shaped single negative dielectric constant material (as shown in Figure 2).
  • the reflector 4 in the antenna provided in the first embodiment of the present invention may be a metal flat plate.
  • the radiator of the antenna according to the first embodiment of the present invention includes a substrate 61 , a composite left and right hand transmission line unit 62 attached to the front surface of the substrate 61 , and an electromagnetic field band gap attached to the bottom surface of the substrate 61 .
  • a microstrip feed line 622 is attached to the front surface of the substrate 61 near the composite left and right hand transmission line unit 62.
  • the composite left and right hand transmission line unit 62 includes a plurality of periodic metal patches 621.
  • Each of the metal patches 621 may be formed with a hollow pattern 6211 formed by an inductive metal line loading, and the metal lines of the hollow pattern 6211 form an equivalent inductance.
  • the hollowed out pattern 6211 reduces the equivalent parallel capacitance to ground required for the resonant ⁇ and improves the equivalent parallel inductance to ground, thereby reducing the antenna area and miniaturizing the antenna.
  • the metal patch 621 is square, the hollow pattern 6211 formed by each metal patch 621 is located at the center, the hollow pattern 6211 is connected to the edge 6212 of the metal patch 621, and the metal line of the hollow pattern 6211 passes through the metal probe.
  • the needle 6214 is connected to the metal plate 64.
  • the center of the hollow pattern 6211 and the corresponding substrate position are respectively provided with a through hole 6213.
  • the metal probe 6214 is soldered through the hollow pattern and the through hole 6213 of the substrate to the metal wire of the hollow pattern 6211 beside the through hole.
  • the aperture of the through hole of the hollow pattern is slightly larger than the aperture of the through hole of the substrate and the diameter of the metal probe, thereby facilitating the soldering of the metal probe to the metal line of the hollow pattern beside the through hole.
  • the metal line of the hollow pattern 6211 forms an equivalent grounding inductance, thereby reducing the antenna area, and the inductance is connected in parallel with the equivalent capacitance of the metal patch itself to form a loop radiator of the composite left and right hand transmission line unit.
  • the metal patch 621 can also have any shape such as a circular shape or a fan shape.
  • the cutout pattern can take a variety of styles, such as a bend line style, an arc style, a triangle style, and the like.
  • the EBG structure 63 is equivalent to an ideal artificial magnetic conductor (artificial magnetic conductors).
  • the EBG structure 63 can eliminate the surface wave between the composite left and right hand transmission line unit 62 and the ground, thereby improving the efficiency of the antenna and increasing the bandwidth of the antenna.
  • the EBG structure 63 includes a plurality of periodic unit junctions.
  • Structure 633, unit structure 633 may have a hollow pattern, and the hollow pattern may adopt various styles, such as a bend line style, an arc style, a triangle style, and the like.
  • the EBG structure 63 is provided with a through hole 631 corresponding to the through hole 6213 of the substrate, and the through hole diameter of the EBG structure is larger than the through hole diameter of the substrate to prevent the EBG structure 63 from being connected to the metal flat plate 64.
  • the EBG structure 63 and the metal flat plate 64 have an air layer gap 634.
  • the air gap reduces the equivalent dielectric constant of the antenna. The lower the equivalent dielectric constant, the higher the radiation efficiency and the wider the bandwidth. Therefore, the air layer gap improves the antenna efficiency and bandwidth.
  • the metal flat plate 64 may also be attached to the EBG structure 63 through an insulating medium.
  • the radiator of the antenna according to the first embodiment of the present invention may adopt another structure, and the difference from the radiator is that an EBG structure 66 is attached to the front surface of the substrate, and the EBG structure 66 is located in the composite.
  • the circumference of the right and left hand transmission line unit 65 This allows the EBG structure to reach two layers, thereby widening the bandwidth of the EBG structure.
  • the radiator includes a composite right and left hand transmission line unit attached to the front surface of the substrate, and an EBG structure attached to the bottom surface of the substrate, and a grounded and connected to the composite left and right hand transmission line unit on the EBG structure side.
  • the metal plate so the radiator has a low profile, and the antenna is miniaturized.
  • the EBG structure can eliminate surface waves between the composite left and right hand transmission line units and the ground, thereby improving the efficiency of the antenna and increasing the bandwidth of the antenna; Since the composite left and right hand transmission line unit includes a plurality of periodic metal patches, each metal patch is formed with a hollow pattern formed by inductive metal wire loading, and the metal line of the hollow pattern forms an equivalent inductance, thereby reducing the need for resonance The equivalent parallel capacitance to ground and the equivalent parallel inductance to the ground, thereby reducing the antenna area and miniaturizing the antenna; and because of the air layer gap between the EBG structure and the metal plate, the air layer gap is reduced. The equivalent dielectric constant of the antenna, thereby improving antenna efficiency and width.
  • a director of an antenna according to Embodiment 1 of the present invention includes a frequency selective surface (Fre quency Selective).
  • FSS lens includes a first capacitive screen 41, a second capacitive screen 42 and a resonant screen 43 between the first capacitive screen 41 and the second capacitive screen 42, a first capacitive screen 41 and a second capacitive screen 42 may be a capacitive screen having the same structure or completely different, and the first capacitive screen 41 and the second capacitive screen 42 are each equivalent to one capacitor.
  • the first capacitive screen 41 faces the radiator, and the second capacitive screen 42 faces the outside of the antenna.
  • the first capacitive screen and the second capacitive screen of the director of the antenna provided by the first embodiment of the present invention each include a substrate and a plurality of periodic capacitive metal stickers attached to the front surface of the substrate.
  • the sheet 4011, the surface of the plurality of periodic capacitive metal patches attached to the front surface of the substrate is used as the front surface 401 of the first capacitive screen and the second capacitive screen, and the bottom surface of the substrate serves as the bottom surface of the first capacitive screen and the second capacitive screen. 402.
  • the capacitive metal patch may be a square or circular patch.
  • a resonant screen of a director of an antenna includes a substrate and a plurality of periodic unit structures 4311 attached to a front surface of the substrate, each unit structure 4311 including an outer conductor. 431 2.
  • the inner conductor 4313 and the bending groove 4314 between the outer conductor 4312 and the inner conductor 4313, and the outer conductor 4312 is an equivalent ground.
  • An equivalent capacitance is formed between the bending groove 4314 and the outer conductor 4312, and the equivalent capacitance forms an equivalent parallel resonant circuit with the equivalent inductance introduced by the outer conductor 4312 and the inner conductor 4313.
  • the inner conductor 4313 has a square slot 4315 in the center to reduce energy coupling between the frequency selective surfaces.
  • the surface of the plurality of periodic unit structures 4311 attached to the front surface of the substrate serves as the front surface 431 of the resonant screen.
  • the bottom surface of the substrate may include a metal flat plate or may not include a metal flat plate. When the metal flat plate is included, the metal flat plate serves as a resonant screen.
  • the bottom surface 432 when the metal plate is not included, the bottom surface of the substrate serves as the bottom surface of the resonant screen.
  • the bottom surface 432 of the resonant screen may have a periodic groove 4321.
  • the groove 4321 may be circular, square or any other shape. The periodic groove can effectively reduce the coupling between the resonant screen and the first capacitive screen and the second capacitive screen. The thickness of the director is reduced.
  • the front faces of the first capacitive screen and the second capacitive screen may be replaced by the FSS capacitive unit adopting the fractal technique in FIG.
  • the front side of the resonant screen can be replaced by the Jerusalem cross-shaped unit in Figure 14.
  • the bottom surface of the first capacitive screen 41 faces the bottom surface of the resonant screen 43, and the second capacitive screen
  • the bottom surface of 42 faces the front side of the resonant screen 43.
  • the director of the antenna provided in Embodiment 1 of the present invention can be applied to an antenna operating in a frequency band of 2.4 GHz or 5.8 GHz.
  • the period of the resonant screen unit structure of the FSS lens in the director of the antenna operating in the 5.8 GHz band and the period of the capacitive metal patch of the first capacitive screen and the second capacitive screen can be compared to the FSS lens operating in the 2.4 GHz band The cycle is less, depending on the actual situation.
  • electromagnetic waves can be controlled through each unit structure. Phase The position, the electromagnetic wave of different paths is realized in the same phase after passing through the FSS lens, and the transformation of the spherical wave into a plane is completed, thereby achieving the compression lobe width, and finally the effect of the 3dB lobe width of the electromagnetic wave radiated by the compression radiator is achieved.
  • the radiation of the antenna toward both sides becomes less, and the antenna isolation is correspondingly increased.
  • the lens is of a band-pass type, electromagnetic waves can be directly transmitted, so that the antenna front-to-back ratio can be improved, which is advantageous for improving the isolation of the antenna.
  • the director of the antenna since the director of the antenna includes a frequency selective surface FSS lens, the FS S lens includes a first capacitive screen, a second capacitive screen, and the first capacitive screen and the second capacitive screen.
  • the resonant screen between. Therefore, the electromagnetic wave will be transmitted directly from the FSS lens, and will not affect the front-to-back ratio of the antenna, and the compression amplitude of the 3dB lobe width of the antenna is large, and the 3dB lobe width of the antenna can be compressed by 30° or more, and the lobe width is compressed.
  • the performance is much better than the traditional antenna director; after the electromagnetic wave passes through the FSS lens, the spherical wave is quickly converted into a plane wave, and the energy of the electromagnetic wave is no longer radiated to both sides of the radiation source, thereby improving the use of the present invention.
  • the inter-antenna isolation of the MIMO system of the antenna director Since the director of the antenna includes a zero-refractive-index lens and an FSS lens, or a zero-refractive-index lens and two FSS lenses, the two types of lenses act simultaneously, compressing the lobe width and improving the antenna. Inter-isolation.
  • the main function of the absorber of the antenna provided by the first embodiment of the present invention is to absorb and reflect the energy radiated by the radiator to both sides, thereby reducing the coupling between adjacent antennas and improving the antenna isolation.
  • Embodiment 1 of the present invention provides an absorber for an antenna, the absorber comprising a substrate, a metamaterial layer attached to a front surface of the substrate, and a metal flat layer attached to a bottom surface of the substrate, wherein the metamaterial layer has a plurality of layers A periodic cell structure, each cell structure is formed with a hollow pattern, the hollow pattern is formed by an inductive metal line loading, and the hollow pattern forms an equivalent inductance.
  • the absorber of the antenna according to the first embodiment of the present invention includes a substrate 11, a metamaterial layer 12 attached to the front surface of the substrate 11, and a metal plate layer 13 attached to the bottom surface of the substrate 11.
  • the metamaterial layer 12 refers to a composite material layer having an artificially designed structure and exhibiting extraordinary physical properties not possessed by the natural material, and the metamaterial layer 12 may be a soft and hard surface (soft And hard surfaces), photonic crystals, electromagnetic band-gap structures, double negative
  • the metamaterial layer 12 has a plurality of periodic unit structures 121, and the unit structure 121 shown in FIG. 24 is a metal patch. As shown in FIG. 24, the metamaterial layer 12 of the absorber of the antenna operating in the 2.4 GHz band provided by the first embodiment of the present invention has nine periodic cell structures 121.
  • Each of the unit structures 121 is formed with a hollow pattern 1211 formed by an inductive metal line loading, and the hollow pattern 1211 forms an equivalent inductance.
  • the function of the hollow pattern 1211 is to increase the equivalent inductance of the unit structure 121, miniaturize the entire unit structure, thereby miniaturizing the entire absorber, so that a larger number of periods and unit structures can be accommodated by the metamaterial layer under the same area. , to make the absorption rate of the absorber better.
  • the unit structure 121 is square, and the hollow pattern 1211 formed by each unit structure 121 is located at the center, and the outer contour of the hollow pattern 1211 may be any shape such as a square, a circle, a fan, or the like.
  • Cutout patterns can take a variety of styles, such as bend line styles, arc styles, triangle styles, and more.
  • An equivalent capacitance C is formed between each side of the unit structure 121 and an adjacent side of the adjacent unit structure, and the hollow pattern 1211 forms an equivalent inductance L, and the equivalent capacitance C is connected in parallel with the equivalent inductance L.
  • a resistor R is soldered between each side of each unit structure 121 and an adjacent side of an adjacent unit structure (a, b, c, d in FIG. 24), thereby making the resistance R and the equivalent capacitance C and equivalent Inductor L is connected in parallel.
  • the function of the resistor R is to match the air impedance, and to absorb the electromagnetic wave energy reflected by the bottom metal plate layer 13, and to reduce the effect of reflection.
  • the resistor used may be a 230 ohm resistor.
  • each side of each unit structure 121 it is also possible to solder between each side of each unit structure 121 and an adjacent side of an adjacent unit structure (next to a, b, c, d in FIG. 24).
  • the capacitor in turn, connects the resistor and capacitor in parallel to further miniaturize the entire absorber.
  • FIG. 29 it is a front view of another type of absorber of an antenna according to Embodiment 1 of the present invention.
  • the hollow pattern 2211 formed by each of the unit structures 221 shown in FIG. 29 is different from the hollow pattern shown in FIG. 24, and each unit structure 221 has a square hole 2212 at the center thereof, and a hollow pattern 2211 is formed on the outer periphery of the square hole 2212.
  • the periphery of the hollow pattern 2211 is the ground 2213.
  • the hollow pattern 2211 is equivalent to the inner conductor, and the equivalent capacitance C is formed between the hollow pattern 2211 and the ground 2213, and the hollow pattern 2211 forms an equivalent inductance L, and the equivalent capacitance C is connected in parallel with the equivalent inductance L.
  • a resistance R is soldered between the hollow pattern of each side of each unit structure 221 and the ground (&, b, c, d in Fig. 29), thereby paralleling the resistance R with the equivalent capacitance C and the equivalent inductance L.
  • the metal plate layer 13 serves to reflect electromagnetic waves reaching the metal plate layer 13 to the resistor and the substrate 11, and is absorbed by the resistor and the substrate 11.
  • FIG. 27 is a schematic diagram showing the performance of the absorber of the antenna operating in the 2.4 GHz band according to the first embodiment of the present invention
  • FIG. 28 is the presence or absence between the antennas operating in the 2.4 GHz band.
  • the substrate of the first embodiment of the present invention may be an FR4 substrate.
  • the FR4 substrate is a currently the cheapest substrate.
  • the price is extremely low, and the ultra-thin layer has the ultra-thin thickness, so that the antenna of the first embodiment of the present invention absorbs the wave.
  • the hollow pattern of the unit structure is of the same pattern, and as long as the size of the metamaterial layer is different, the absorber of the antenna can be operated in different frequency bands.
  • the absorber of the antenna provided by the first embodiment of the present invention includes a metamaterial layer attached to the front surface of the substrate and a metal plate layer attached to the bottom surface of the substrate, thereby widening the bandwidth of the antenna and reducing a large amount of electromagnetic Reflection, significantly improve the isolation between the antennas, and has no effect on other antenna performance. Applying it to multi-antenna ⁇ system can improve the isolation between antennas, improve system communication capacity, and increase the anti-interference ability of the system. . Moreover, since the metamaterial layer has a plurality of periodic cell structures, each cell structure is formed with a hollow pattern, and the hollow pattern is formed by inductive metal wire loading, and the hollow pattern forms an equivalent inductance, thereby improving the equivalent of the cell structure.
  • Inductance miniaturizes the entire cell structure, thereby miniaturizing the entire absorber, allowing the superabsorbent layer to accommodate a greater number of cycles and cell structures in the same area, allowing the absorber to absorb better.
  • each side of the unit structure is soldered with an electric resistance between adjacent sides of the adjacent unit structure, or a resistance pattern is formed between the hollow pattern of each side of each unit structure and the ground, the air impedance can be matched. The same absorbing the electromagnetic wave energy reflected by the bottom metal plate layer, thereby reducing the reflection.
  • each side of the unit structure is soldered with a capacitor between adjacent sides of the adjacent unit structure, or a capacitor is soldered between the hollow pattern of each side of each unit structure and the ground, thereby forming a resistor and a capacitor Parallel to further miniaturize the entire absorber.
  • the antenna provided in Embodiment 2 of the present invention is different from the antenna provided in Embodiment 1 of the present invention in that a director 6 of another structure and a reflector 7 of another structure are used.
  • the director of the antenna working in the 5.8 GHz band provided by the second embodiment of the present invention includes a zero-refractive-index lens 51 and one or two antennas provided by Embodiment 1 of the present invention.
  • the FSS lens 52 in the director The period of the resonant screen unit structure of the FSS lens 52 in the second embodiment of the present invention and the period of the capacitive metal patch of the first capacitive screen and the second capacitive screen may be compared to the operation in the 2.4 GHz band in the first embodiment of the present invention.
  • the cycle of the FSS lens is less, depending on the actual situation.
  • the zero index lens 51 faces the radiator of the antenna, and the FSS lens 52 faces the outside of the antenna.
  • the zero-refractive-index lens 51 has a plurality of periodic unit structures 511, which have a ring-shaped structure and may be square rings, circular rings or other shaped rings.
  • Two adjacent cell structures form an equivalent series resonance, forming an equivalent negative dielectric constant effect at the resonance, but an equivalent near-zero refractive index characteristic occurs near the resonance, and electromagnetic waves of different paths pass through the zero-refractive-index lens.
  • the phase of the electromagnetic wave transmitted in different paths is realized in phase, thereby completing the conversion of the spherical wave to the plane wave, and the effect of compressing the width of the antenna lobe is realized.
  • the zero-refractive-index lens 51 has a face of a plurality of periodic unit structures 511 as a front face of the zero-refractive-index lens 51.
  • the back side of the zero index lens 51 faces the FSS lens 52.
  • the FS S lens since the director of the antenna includes a frequency selective surface FSS lens, the FS S lens includes a first capacitive screen, a second capacitive screen, and the first capacitive screen and the second capacitive screen.
  • the resonant screen between. Therefore, the electromagnetic wave will be transmitted directly from the FSS lens, and will not affect the front-to-back ratio of the antenna, and the compression amplitude of the 3dB lobe width of the antenna is large, and the 3dB lobe width of the antenna can be compressed by 30° or more, and the lobe width is compressed.
  • the performance is much better than the traditional antenna director; after the electromagnetic wave passes through the FSS lens, the spherical wave is quickly converted into a plane wave, and the energy of the electromagnetic wave is no longer radiated to both sides of the radiation source, thereby improving the use of the present invention.
  • the inter-antenna isolation of the MIMO system of the antenna director Since the director of the antenna includes a zero-refractive-index lens and an FSS lens, or a zero-refractive-index lens and two FSS lenses, the two types of lenses act simultaneously, compressing the lobe width and improving the antenna. Inter-isolation.
  • a reflector of an antenna provided by Embodiment 2 of the present invention includes a substrate and a plurality of periodic metal patches attached to a front surface of the substrate.
  • the embodiment of the present invention further provides a wireless router including the antenna provided by Embodiment 1 of the present invention and/or Embodiment 2 of the present invention.

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Abstract

本发明适用于通信领域，提供了一种天线和无线路由器。所述天线包括辐射器、引向器、至少一个吸波器和反射器，相对于辐射器的辐射方向，引向器位于辐射器的前方，反射器位于辐射器的后方，吸波器位于辐射器的侧面；所述辐射器包括基板、附着于基板的正面的复合左右手传输线单元、附着于基板的底面的电磁场带隙EBG结构和位于EBG结构侧的、与复合左右手传输线单元相连的作为地的金属平板。本发明的天线尺寸小、低剖面、水平全向辐射、高效率、宽带宽。

Description

一种天线和无线路由器
技术领域
[0001] 本发明属于通信领域， 尤其涉及一种天线和无线路由器。
背景技术
[0002] 现有技术的天线包括辐射器、 反射器、 引向器和隔离元件等。 其中辐射器主要 采用偶极子或者单极子提供水平全向且垂直极化辐射性能， 偶极子或者单极子 是一种天线电长度为波长的二分之一或者四分之长度的辐射器， 其辐射在水平 面上是全向辐射， 并且辐射电场的极化方向与大地垂直， 即为垂直极化， 该类 型辐射器在使用吋， 一般垂直于水平面外置于天线外。 然而该辐射器的带宽窄 、 辐射效率低、 剖面高且无法平面化， 从而使现有技术的天线尺寸大、 天线带 宽窄、 辐射效率低、 剖面高。
技术问题
[0003] 本发明的目的在于提供一种天线和无线路由器， 旨在解决现有技术的天线的辐 射器的带宽窄、 辐射效率低、 剖面高且无法平面化， 从而使现有技术的天线尺 寸大、 天线带宽窄、 辐射效率低、 剖面高的问题。
问题的解决方案
技术解决方案
[0004] 第一方面， 提供了一种天线， 所述天线包括辐射器、 引向器、 至少一个吸波器 和反射器， 相对于辐射器的辐射方向， 引向器位于辐射器的前方， 反射器位于 辐射器的后方， 吸波器位于辐射器的侧面； 所述辐射器包括基板、 附着于基板 的正面的复合左右手传输线单元、 附着于基板的底面的电磁场带隙 EBG结构和 位于 EBG结构侧的、 与复合左右手传输线单元相连的作为地的金属平板。
[0005] 第二方面， 提供了一种无线路由器， 所述无线路由器包括上述的天线。
发明的有益效果
有益效果
[0006] 在本发明中， 由于天线的辐射器包括附着于基板的正面的复合左右手传输线单 元和附着于基板的底面的 EBG结构和位于 EBG结构侧的、 与复合左右手传输线 单元相连的作为地的金属平板， 因此该辐射器剖面低， 实现了天线的小型化， 另外， EBG结构可消除复合左右手传输线单元与地之间的表面波， 进而提高天 线的效率， 并增加天线的带宽。
对附图的简要说明
附图说明
[0007] 图 1是本发明实施例- -提供的天线的结构示意图。
[0008] 图 2是本发明实施例- -提供的天线的隔离器的结构示意图。
[0009] 图 3是本发明实施例- -提供的天线的辐射器的结构示意图。
[0010] 图 4是本发明实施例- -提供的天线的辐射器中的复合左右手传输线单元的结构
[0011] 图 5是本发明实施例- -提供的天线的辐射器中的 EBG结构的结构示意图。
[0012] 图 6是本发明实施例- -提供的天线的辐射器中的地的结构示意图。
[0013] 图 7是本发明实施例- -提供的天线的辐射器中的复合左右手传输线单元和 EBG 结构附着于基板正面的结构示意图。
[0014] 图 8是本发明实施例一提供的天线的引向器的结构示意图。
[0015] 图 9是本发明实施例一提供的天线的引向器的第一电容屏和第二电容屏的正面
[0016] 图 10是本发明实施例一提供的天线的引向器的第一电容屏和第二电容屏的底面
[0017] 图 11是本发明实施例- -提供的天线的引向器的谐振屏的正面示意图。
[0018] 图 12是本发明实施例- -提供的天线的引向器的谐振屏的底面示意图。
[0019] 图 13是本发明实施例- -提供的天线的引向器的采用的分形电容屏的正面示意图
[0020] 图 14是本发明实施例一提供的天线的引向器的采用的耶路撒冷十字架谐振屏的 正面示意图。
[0021] 图 15是本发明实施例二提供的天线的结构示意图。
[0022] 图 16是本发明实施例二提供的天线的引向器的结构示意图。 [0023] 图 17是本发明实施例」二提供的天线的引向器的零折射率透镜的正面示意图。
[0024] 图 18是本发明实施例」二提供的天线的引向器的零折射率透镜的底面示意图。
[0025] 图 19是本发明实施例」二提供的天线的引向器的电容屏的正面示意图。
[0026] 图 20是本发明实施例」二提供的天线的引向器的电容屏的底面示意图。
[0027] 图 21是本发明实施例」二提供的天线的引向器的谐振屏的正面示意图。
[0028] 图 22是本发明实施例」二提供的天线的引向器的谐振屏的底面示意图。
[0029] 图 23是本发明实施例」二提供的天线的反射器的结构示意图。
[0030] 图 24是本发明实施例-一提供的天线的吸波器的正面示意图。
[0031] 图 25是本发明实施例-一提供的天线的吸波器的侧面示意图。
[0032] 图 26是本发明实施例-一提供的天线的吸波器的底面示意图。
[0033] 图 27是本发明实施例-一提供的天线的吸波器的性能示意图。
[0034] 图 28是工作于 2.4GHz频段的天线间有无采用本发明实施例一提供的工作于 2.4G
Hz频段的天线的吸波器的隔离对比示意图。
[0035] 图 29是本发明实施例-一提供的天线的另一种吸波器的正面示意图。
实施该发明的最佳实施例
本发明的最佳实施方式
[0036] 为了使本发明的目的、 技术方案及有益效果更加清楚明白， 以下结合附图及实 施例， 对本发明进行进一步详细说明。 应当理解， 此处所描述的具体实施例仅 仅用以解释本发明， 并不用于限定本发明。
[0037] 为了说明本发明所述的技术方案， 下面通过具体实施例来进行说明。
[0038] 实施例一：
[0039] 请参阅图 1， 本发明实施例一提供的天线包括辐射器 1、 引向器 2、 一个或两个 吸波器 3和反射器 4， 相对于辐射器 1的辐射方向， 引向器 2位于辐射器 1的前方， 反射器 4位于辐射器 1的后方， 两个吸波器 3分别位于辐射器 1的两个侧面。
[0040] 本发明实施例一提供的天线还可以包括一个或两个隔离器 5， 两个隔离器 5分别 位于两个吸波器 3背向辐射器 1的后面， 或者位于两个吸波器 3朝向辐射器 1的前 面。 隔离器 5可以采用工字形的单负介电常数材料 （如图 2所示） 。 本发明实施 例一提供的天线中的反射器 4可以是金属平板。 [0041] 请参阅图 3至图 6， 本发明实施例一提供的天线的辐射器包括基板 61、 附着于基 板 61的正面的复合左右手传输线单元 62、 附着于基板 61的底面的电磁场带隙 (Ele ctromagnetic Band Gap, EBG)结构 63和位于 EBG结构 63侧的、 与复合左右手传输 线单元 62相连的作为地的金属平板 64。 由于复合左右手传输线单元 62直接与金 属平板 64相连， 将电流导入地中， 因此 EBG结构 63在复合左右手传输线单元 62 和金属平板 64之间不会造成复合左右手传输线单元 62与地之间的电磁场分布的 改变， 且能消除表面波。
[0042] 在本发明实施例一中， 基板 61的正面在靠近复合左右手传输线单元 62的位置还 附着有一个微带馈电线 622。 复合左右手传输线单元 62包括多个周期性的金属贴 片 621。 每个金属贴片 621可以形成有镂空图案 6211， 镂空图案 6211是由感性的 金属线加载形成的， 镂空图案 6211的金属线形成等效电感。 镂空图案 6211减少 了谐振吋需要的对地等效并联电容并且提高了对地等效并联电感， 从而减小了 天线面积， 实现了天线的小型化。
[0043] 请参阅图 4， 金属贴片 621呈方形， 每个金属贴片 621形成的镂空图案 6211位于 中央， 镂空图案 6211连接金属贴片 621的边缘 6212， 镂空图案 6211的金属线通过 金属探针 6214与金属平板 64连接， 镂空图案 6211中央和对应的基板位置均设有 通孔 6213， 金属探针 6214穿过镂空图案和基板的通孔 6213焊接在通孔旁边的镂 空图案 6211的金属线上； 镂空图案的通孔的孔径略大于基板的通孔的孔径和金 属探针的直径， 从而方便金属探针焊接在通孔旁边的镂空图案的金属线上。 镂 空图案 6211的金属线形成了等效的对地电感， 从而减少了天线面积， 且该电感 与金属贴片自身的对地等效电容并联， 最终形成复合左右手传输线单元的幵路 辐射器。 金属贴片 621也可以呈圆形、 扇形等任意形状。 镂空图案可以采用多种 样式， 例如弯折线样式、 弧线样式、 三角形样式等等。
[0044] 请参阅图 5， EBG结构 63等效于理想的人工磁导体 (artificial magnetic conductors
， AMC)， 当电磁波在金属平面传输吋， 表面波会被激励出现， 表面波的出现引 起大量电磁波能量的损耗， 造成天线辐射效率低下， 并且带宽变窄。 EBG结构 6 3可消除复合左右手传输线单元 62与地之间的表面波， 进而提高天线的效率， 并 增加天线的带宽。 在本发明实施例一中， EBG结构 63包括多个周期性的单元结 构 633， 单元结构 633可以具有镂空图案， 镂空图案可以采用多种样式， 例如弯 折线样式、 弧线样式、 三角形样式等等。 EBG结构 63于与基板的通孔 6213对应 处设有通孔 631， 且 EBG结构的通孔孔径大于基板的通孔孔径， 以避免 EBG结构 63与金属平板 64连接。
[0045] 在本发明实施例一中， EBG结构 63与金属平板 64之间具有空气层间隙 634。 空 气层间隙减少了天线的等效介电常数， 由于越低的等效介电常数， 辐射效率就 越高， 带宽就越宽， 因此该空气层间隙提高了天线效率与带宽。
[0046] 在本发明实施例一中， 金属平板 64也可以通过绝缘介质附着在 EBG结构 63上。
[0047] 请参阅图 7， 本发明实施例一提供的天线的辐射器还可以采用另一种结构， 与 上述辐射器的区别在于， 基板的正面还附着有 EBG结构 66， EBG结构 66位于复 合左右手传输线单元 65的周围。 使得 EBG结构达到了两层， 从而能展宽 EBG结 构的带宽。
[0048] 在本发明实施例一中， 由于辐射器包括附着于基板的正面的复合左右手传输线 单元和附着于基板的底面的 EBG结构和位于 EBG结构侧的、 与复合左右手传输 线单元相连的作为地的金属平板， 因此该辐射器剖面低， 实现了天线的小型化 ， 另外， EBG结构可消除复合左右手传输线单元与地之间的表面波， 进而提高 天线的效率， 并增加天线的带宽； 另外， 由于复合左右手传输线单元包括多个 周期性的金属贴片， 每个金属贴片形成有由感性的金属线加载形成的镂空图案 ， 镂空图案的金属线形成等效电感， 因此， 减少了谐振吋需要的对地等效并联 电容并且提高了对地等效并联电感， 从而减小了天线面积， 实现了天线的小型 化； 又由于 EBG结构与金属平板之间具有空气层间隙， 空气层间隙减少了天线 的等效介电常数， 从而提高了天线效率与带宽。
[0049] 请参阅图 8， 本发明实施例一提供的天线的引向器包括一个频率选择表面 （Fre quency Selective
Surface , FSS) 透镜， FSS透镜包括第一电容屏 41、 第二电容屏 42和位于第一电 容屏 41和第二电容屏 42之间的谐振屏 43， 第一电容屏 41和第二电容屏 42可以是 结构相同或完全不相同的电容屏， 第一电容屏 41和第二电容屏 42均等效为一个 电容。 第一电容屏 41朝向辐射器， 第二电容屏 42朝向天线的外部。 [0050] 请参阅图 9和 10， 本发明实施例一提供的天线的引向器的第一电容屏和第二电 容屏均包括基板和附着于基板正面的多个周期性的电容性金属贴片 4011， 附着 于基板正面的多个周期性的电容性金属贴片所在的面作为第一电容屏和第二电 容屏的正面 401， 基板的底面作为第一电容屏和第二电容屏的底面 402。
[0051] 在本发明实施例一中， 电容性金属贴片可以是方形或圆形的贴片。
[0052] 请参阅图 11和 12， 本发明实施例一提供的天线的引向器的谐振屏包括基板和附 着于基板正面的多个周期性的单元结构 4311， 每个单元结构 4311包括外导体 431 2、 内导体 4313和位于外导体 4312和内导体 4313之间的弯折槽 4314， 外导体 4312 为等效的地。 弯折槽 4314和外导体 4312之间形成等效电容， 该等效电容与由外 导体 4312和内导体 4313引入的等效电感形成等效并联谐振回路。 内导体 4313的 中央具有方形槽 4315， 用以减少频率选择性表面间的能量耦合。 附着于基板正 面的多个周期性的单元结构 4311所在的面作为谐振屏的正面 431， 基板的底面可 以包括金属平板也可以不包括金属平板， 当包括金属平板吋， 该金属平板作为 谐振屏的底面 432， 当不包括金属平板吋， 基板的底面作为谐振屏的底面。 谐振 屏的底面 432可以具有周期性的槽 4321， 槽 4321可以是圆形、 方形或其他任意形 状， 周期性的槽可有效减少谐振屏与第一电容屏和第二电容屏之间的耦合， 降 低了引向器的厚度。
[0053] 在本发明实施例一中， 第一电容屏和第二电容屏的正面可以被图 13中的采用分 形技术的 FSS电容性单元代替。 谐振屏的正面可以被图 14中的耶路撒冷十字形单 元代替。
[0054] 在本发明实施例一中， 第一电容屏 41的底面朝着谐振屏 43的底面， 第二电容屏
42的底面朝着谐振屏 43的正面。
[0055] 本发明实施例一提供的天线的引向器可以应用在工作于 2.4GHz或者 5.8GHz频 段的天线。 工作于 5.8GHz频段的天线的引向器中的 FSS透镜的谐振屏单元结构的 周期和第一电容屏和第二电容屏的电容性金属贴片的周期可以比工作于 2.4GHz 频段的 FSS透镜的周期少一些， 具体根据实际情况而定。
[0056] 在本发明实施例一中， 通过调整 FSS透镜的第一电容屏和第二电容屏的金属贴 片的尺寸和谐振屏上的弯折槽的尺寸， 可以控制电磁波通过每个单元结构的相 位， 实现不同路径电磁波经过 FSS透镜后同相， 完成球面波转变为平面的转换， 从而达到压缩波瓣宽度， 最终实现了压缩辐射器辐射出的电磁波的 3dB波瓣宽度 的效果。 此外， 由于对电磁波束的压缩， 天线朝向两侧的辐射变少， 天线隔离 度也会相应提高。 且由于该透镜是带通型的， 电磁波可以直接透射而过， 因此 能改善天线前后比， 有利于天线隔离度的提高。
[0057] 在本发明实施例一中， 由于天线的引向器包括一个频率选择表面 FSS透镜， FS S透镜包括第一电容屏、 第二电容屏和位于第一电容屏和第二电容屏之间的谐振 屏。 因此电磁波将直接从 FSS透镜透射而过， 不会对天线的前后比造成影响， 且 对天线 3dB波瓣宽度的压缩幅度大， 能将天线 3dB波瓣宽度压缩 30°以上， 压缩波 瓣宽度的性能远好于传统的天线引向器； 同吋电磁波经过 FSS透镜后， 球面波很 快转换成了平面波， 电磁波的能量不会再向辐射源的两侧辐射， 因此， 提高了 采用本发明的天线引向器的 MIMO系统的天线间隔离度。 又由于天线的引向器包 括一个零折射率透镜和一个 FSS透镜， 或者， 包括一个零折射率透镜和两个 FSS 透镜， 因此两类透镜同吋作用， 既压缩波瓣宽度， 又提高了天线间隔离度。
[0058] 本发明实施例一提供的天线的吸波器的主要作用是将辐射器辐射向两侧的能量 吸收和反射， 从而降低临近天线间的耦合， 改善天线隔离度。
[0059] 本发明实施例一提供了一种天线的吸波器， 所述吸波器包括基板、 附着于基板 正面的超材料层和附着于基板底面的金属平板层， 超材料层具有多个周期性的 单元结构， 每个单元结构形成有镂空图案， 镂空图案是由感性的金属线加载形 成的， 镂空图案形成等效电感。
[0060] 请参阅图 24至图 26， 本发明实施例一提供的天线的吸波器包括基板 11、 附着于 基板 11正面的超材料层 12和附着于基板 11底面的金属平板层 13。
[0061] 在本发明实施例一中， 超材料层 12是指具有人工设计的结构并呈现出天然材料 所不具备的超常物理性质的复合材料层， 超材料层 12可以是软硬表面 (soft and hard surfaces)层、 光子晶体 (photonic crystals)层、 电磁带隙结构 (electromagnetic band-gap structures)层、 双负材料 (double negative
materials)层、 左手材料 (left-handed materials)层、 人工磁导体 (artificial magnetic conductors , AMC)层等。 AMC层是一种人工制作的等效理想磁面层。 [0062] 超材料层 12具有多个周期性的单元结构 121， 图 24所示的单元结构 121是金属贴 片。 如图 24所示， 本发明实施例一提供的工作于 2.4GHz频段的天线的吸波器的 超材料层 12具有 9个周期性的单元结构 121。
[0063] 每个单元结构 121形成有镂空图案 1211， 镂空图案 1211是由感性的金属线加载 形成的， 镂空图案 1211形成等效电感。 镂空图案 1211的作用在于提高单元结构 1 21的等效电感， 小型化整个单元结构， 从而小型化整个吸波器， 使得在相同的 面积下超材料层能容纳的更多数量的周期和单元结构， 让吸波器的吸收率更佳 。 请参阅图 24， 单元结构 121呈方形， 每个单元结构 121形成的镂空图案 1211位 于中央， 镂空图案 1211的外轮廓可以是方形、 圆形、 扇形等任意形状。 镂空图 案可以采用多种样式， 例如弯折线样式、 弧线样式、 三角形样式等等。 单元结 构 121的每条边与相邻单元结构的相邻边之间形成等效电容 C， 镂空图案 1211形 成等效电感 L， 等效电容 C与等效电感 L并联。 每个单元结构 121的每条边与相邻 单元结构的相邻边之间 （如图 24中的 a、 b、 c、 d) 焊接电阻 R， 从而将电阻 R与 等效电容 C和等效电感 L并联。 电阻 R的作用在于匹配空气阻抗， 同吋吸收被底面 金属平板层 13反射的电磁波能量， 并减小反射的作用。 在本发明实施例一中， 所用的电阻可以为 230欧姆的电阻。
[0064] 在本发明实施例一中， 还可以在每个单元结构 121的每条边与相邻单元结构的 相邻边之间 （如图 24中的 a、 b、 c、 d旁边） 焊接电容， 从而将电阻和电容并联， 以进一步小型化整个吸波器。
[0065] 请参阅图 29， 是本发明实施例一提供的天线的另一种吸波器的正面示意图。 图 29所示的每个单元结构 221形成的镂空图案 2211不同于图 24所示的镂空图案的样 式， 每个单元结构 221的中央具有方形孔 2212， 镂空图案 2211形成于方形孔 2212 的外围， 镂空图案 2211的外围是地 2213。 镂空图案 2211相当于内导体， 镂空图 案 2211与地 2213之间形成等效电容 C， 镂空图案 2211形成等效电感 L， 等效电容 C 与等效电感 L并联。 每个单元结构 221的每条边的镂空图案与地之间 （如图 29中 的&、 b、 c、 d) 焊接电阻 R， 从而将电阻 R与等效电容 C和等效电感 L并联。
[0066] 在本发明实施例一中， 还可以在每个单元结构 221的每条边的镂空图案与地之 间 （如图 29中的 a、 b、 c、 d旁边） 焊接电容， 从而将电阻和电容并联， 以进一步 小型化整个吸波器。
[0067] 金属平板层 13用于将到达金属平板层 13处的电磁波反射至电阻和基板 11， 并被 电阻和基板 11吸收。
[0068] 请参阅图 27和图 28， 其中图 27是本发明实施例一提供的工作于 2.4GHz频段的天 线的吸波器的性能示意图， 图 28是工作于 2.4GHz频段的天线间有无采用本发明 实施例一提供的工作于 2.4GHz频段的天线的吸波器的隔离对比示意图。 根据对 比可知， 工作于 2.4GHz频段的天线间有采用本发明实施例一提供的工作于 2.4GH z频段的天线的吸波器的隔离度比没有采用的隔离度高出接近 10dB。
[0069] 本发明实施例一的基板可以是 FR4基板。 FR4基板是一种当前最廉价的基板， 当超材料层制作于 FR4基板上吋， 价格极低， 又由于超材料层具备超薄厚度的 特性， 使得本发明实施例一提供的天线的吸波器非常适合应用于消费类电子产
P
[0070] 对于单元结构的镂空图案是同一个样式的， 只要超材料层的尺寸大小不一样， 则可以使天线的吸波器工作在不同的频段。
[0071] 本发明实施例一提供的天线的吸波器， 由于包括附着于基板正面的超材料层和 附着于基板底面的金属平板层， 因此使天线的带宽变宽， 且减少了大量的电磁 反射， 显著提高天线间的隔离度， 对天线其他的性能并未造成任何影响， 将其 应用于多天线构成 ΜΙΜΟ系统中， 可提高天线间隔离度， 改善系统通信容量， 增 加系统的抗干扰能力。 又由于超材料层具有多个周期性的单元结构， 每个单元 结构形成有镂空图案， 镂空图案是由感性的金属线加载形成的， 镂空图案形成 等效电感， 因此提高了单元结构的等效电感， 小型化整个单元结构， 从而小型 化整个吸波器， 使得在相同的面积下超材料层能容纳的更多数量的周期和单元 结构， 让吸波器的吸收率更佳。 又由于单元结构的每条边与相邻单元结构的相 邻边之间焊接有电阻， 或者， 每个单元结构的每条边的镂空图案与地之间焊接 有电阻， 因此能匹配空气阻抗， 同吋吸收被底面金属平板层反射的电磁波能量 ， 从而起到减小反射的作用。 另外， 由于单元结构的每条边与相邻单元结构的 相邻边之间焊接有电容， 或者， 每个单元结构的每条边的镂空图案与地之间焊 接有电容， 从而将电阻和电容并联， 以进一步小型化整个吸波器。 [0073] 例二：
[0072] 请参阅图 15， 本发明实施例二提供的天线与本发明实施例一提供的天线的区别 在于采用另一种结构的引向器 6和另一种结构的反射器 7。
[0073] 请参阅图 16至图 22， 本发明实施例二提供的工作于 5.8GHz频段的天线的引向器 包括一个零折射率透镜 51和一个或两个本发明实施例一提供的天线的引向器中 的 FSS透镜 52。 本发明实施例二中的 FSS透镜 52的谐振屏单元结构的周期和第一 电容屏和第二电容屏的电容性金属贴片的周期可以比本发明实施例一中的工作 于 2.4GHz频段的 FSS透镜的周期少一些， 具体根据实际情况而定。 零折射率透镜 51朝向天线的辐射器， FSS透镜 52朝向天线的外部。
[0074] 零折射率透镜 51具有多个周期性的单元结构 511， 单元结构 511呈环形结构， 可 以是方形的环、 圆形的环或其他形状的环。 两相邻单元结构形成等效的串联谐 振， 在谐振处形成等效负介电常数效应， 但是在临近谐振处则会出现等效近零 折射率特性， 不同路径的电磁波经过零折射率透镜后， 实现不同路径传输电磁 波相位同相， 从而完成了球面波到平面波的转换， 实现了压缩天线波瓣宽度的 效果。
[0075] 零折射率透镜 51具有多个周期性的单元结构 511的面作为零折射率透镜 51的正 面。 零折射率透镜 51的背面朝向 FSS透镜 52。
[0076] 在本发明实施例二中， 由于天线的引向器包括一个频率选择表面 FSS透镜， FS S透镜包括第一电容屏、 第二电容屏和位于第一电容屏和第二电容屏之间的谐振 屏。 因此电磁波将直接从 FSS透镜透射而过， 不会对天线的前后比造成影响， 且 对天线 3dB波瓣宽度的压缩幅度大， 能将天线 3dB波瓣宽度压缩 30°以上， 压缩波 瓣宽度的性能远好于传统的天线引向器； 同吋电磁波经过 FSS透镜后， 球面波很 快转换成了平面波， 电磁波的能量不会再向辐射源的两侧辐射， 因此， 提高了 采用本发明的天线引向器的 MIMO系统的天线间隔离度。 又由于天线的引向器包 括一个零折射率透镜和一个 FSS透镜， 或者， 包括一个零折射率透镜和两个 FSS 透镜， 因此两类透镜同吋作用， 既压缩波瓣宽度， 又提高了天线间隔离度。
[0077] 请参阅图 23， 本发明实施例二提供的天线的反射器包括基板和附着于基板正面 的多个周期性的金属贴片。 [0078] 本发明实施例还提供了一种包括本发明实施例一和 /或本发明实施例二提供的 天线的无线路由器。
[0079] 以上所述仅为本发明的较佳实施例而已， 并不用以限制本发明， 凡在本发明的 精神和原则之内所作的任何修改、 等同替换和改进等， 均应包含在本发明的保 护范围之内。

Claims

权利要求书
一种天线， 其特征在于， 所述天线包括辐射器、 引向器、 至少一个吸 波器和反射器， 相对于辐射器的辐射方向， 引向器位于辐射器的前方 ， 反射器位于辐射器的后方， 吸波器位于辐射器的侧面； 所述辐射器 包括基板、 附着于基板的正面的复合左右手传输线单元、 附着于基板 的底面的电磁场带隙 EBG结构和位于 EBG结构侧的、 与复合左右手传 输线单元相连的作为地的金属平板。
如权利要求 1所述的天线， 其特征在于， 所述天线还包括至少一个隔 离器， 隔离器位于吸波器背向辐射器的后面， 或者位于吸波器朝向辐 射器的前面。
如权利要求 2所述的天线， 其特征在于， 所述隔离器采用工字形的单 负介电常数材料； 反射器是金属平板或者是反射器包括基板和附着于 基板正面的多个周期性的金属贴片。
如权利要求 1所述的天线， 其特征在于， 所述复合左右手传输线单元 包括多个周期性的金属贴片， 每个金属贴片形成有由感性的金属线加 载形成的镂空图案， 镂空图案的金属线形成等效电感。
如权利要求 1所述的天线， 其特征在于， 所述 EBG结构包括多个周期 性的单元结构， 所述 EBG结构的单元结构具有镂空图案， EBG结构的 镂空图案采用弯折线样式、 弧线样式或三角形样式。
如权利要求 1所述的天线， 其特征在于， 所述 EBG结构与金属平板之 间具有空气层间隙， 或者， 金属平板通过绝缘介质附着在 EBG结构上
[权利要求 7] 如权利要求 1所述的天线， 其特征在于， 所述基板的正面还附着有位 于复合左右手传输线单元的周围的 EBG结构。
[权利要求 8] 如权利要求 1所述的天线， 其特征在于， 所述引向器包括一个频率选 择表面 FSS透镜， FSS透镜包括第一电容屏、 第二电容屏和位于第一 电容屏和第二电容屏之间的谐振屏， 第一电容屏和第二电容屏均等效 为一个电容； 第一电容屏和第二电容屏均包括基板和附着于基板正面的多个周期性 的电容性金属贴片， 附着于基板正面的多个周期性的电容性金属贴片 所在的面作为第一电容屏和第二电容屏的正面， 基板的底面作为第一 电容屏和第二电容屏的底面；
谐振屏包括基板和附着于基板正面的多个周期性的单元结构， 每个单 元结构包括外导体、 内导体和位于外导体和内导体之间的弯折槽； 附 着于基板正面的多个周期性的单元结构所在的面作为谐振屏的正面； 第一电容屏的底面朝着谐振屏的底面， 第二电容屏的底面朝着谐振屏 的正面；
第一电容屏朝向辐射器， 第二电容屏朝向天线的外部。
如权利要求 8所述的天线， 其特征在于， 所述谐振屏的基板的底面包 括金属平板， 金属平板具有周期性的槽。
如权利要求 8所述的天线， 其特征在于， 所述引向器还包括一个零折 射率透镜， 或者， 所述引向器还包括一个零折射率透镜和另一个 FSS 透镜； 零折射率透镜具有多个周期性的单元结构， 单元结构呈环形结 构， 零折射率透镜具有多个周期性的单元结构的面作为零折射率透镜 的正面， 零折射率透镜的背面朝向 FSS透镜； 零折射率透镜朝向辐射 器， FSS透镜朝向天线的外部。
如权利要求 1所述的天线， 其特征在于， 所述吸波器包括基板、 附着 于基板正面的超材料层和附着于基板底面的金属平板层， 超材料层具 有多个周期性的单元结构， 每个单元结构形成有镂空图案， 镂空图案 是由感性的金属线加载形成的， 镂空图案形成等效电感。
如权利要求 11所述的天线， 其特征在于， 所述单元结构是金属贴片， 每个单元结构形成的镂空图案位于中央， 所述单元结构呈方形， 单元 结构的每条边与相邻单元结构的相邻边之间形成等效电容， 等效电容 与等效电感并联， 每个单元结构的每条边与相邻单元结构的相邻边之 间焊接有电阻， 从而将电阻与等效电容和等效电感并联， 每个单元结 构的每条边与相邻单元结构的相邻边之间焊接有电容， 从而将电阻和 电容并联。
[权利要求 13] 如权利要求 11所述的天线， 其特征在于， 每个单元结构的中央具有方 形孔， 镂空图案形成于方形孔的外围， 镂空图案的外围是地， 镂空图 案与地之间形成等效电容， 等效电容与等效电感并联， 每个单元结构 的每条边的镂空图案与地之间焊接有电阻， 从而将电阻与等效电容和 等效电感并联， 每个单元结构的每条边的镂空图案与地之间焊接有电 容， 从而将电阻和电容并联。
[权利要求 14] 一种无线路由器， 其特征在于， 所述无线路由器包括权利要求 1至 13 任一项所述的天线。
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