WO2017082515A1 - Method for processing reception signal from base station by terminal in wireless communication system - Google Patents

Abstract

Disclosed are method and terminal for processing a reception signal, the method in which channel information about some REs in a subframe is obtained by means of a DMRS transmitted through a downlink subframe, a parameter for obtaining channel information about remaining REs in the subframe is determined by means of the channel information about some REs, an initial value is compensated until the error of the initial value to which the parameter is applied is below a threshold, and the channel information about the remaining REs is determined by means of a compensation result.

Description

Method in which a terminal processes a received signal from a base station in a wireless communication system

The following description relates to a wireless communication system, and more particularly, to a method and apparatus for processing a received signal from a base station by a terminal in a wireless communication system.

Wireless communication systems are widely deployed to provide various kinds of communication services such as voice and data. In general, a wireless communication system is a multiple access system capable of supporting communication with multiple users by sharing available system resources (bandwidth, transmission power, etc.). Examples of multiple access systems include code division multiple access (CDMA) systems, frequency division multiple access (FDMA) systems, time division multiple access (TDMA) systems, orthogonal frequency division multiple access (OFDMA) systems, and single carrier frequency (SC-FDMA). division multiple access (MCD) systems and multi-carrier frequency division multiple access (MC-FDMA) systems.

The present invention has been made to solve the above problems, an object of the present invention is to improve the process of processing the downlink received signal by the terminal to improve the received signal processing performance.

Another object of the present invention is to improve the efficiency of accuracy to the maximum with only low complexity in the process of improving the reception signal processing performance.

Another object of the present invention is to minimize the increase in complexity required through the proposed invention.

Technical objects to be achieved in the present invention are not limited to the above-mentioned matters, and other technical problems not mentioned above are provided to those skilled in the art from the embodiments of the present invention to be described below. May be considered.

In accordance with one aspect of the present invention, there is provided a method of processing a received signal, the method comprising: obtaining channel information on a part of a resource element (RE) in a subframe by using a DeModulation Reference Signal (DMRS) transmitted through a downlink subframe, and part of the RE Parameter for obtaining channel information on the remaining REs in the subframe using the channel information for the

Determining the parameters

Compensating the initial value until the error of the applied initial value is less than the threshold value, and determining the channel information for the remaining RE using the compensated result.

The received signal processing method may further include decoding downlink data included in a subframe using channel results for some REs and channel results for the remaining REs.

The compensating step may be performed using a conjugate gradient algorithm or a Newton method.

The initial filter and acceleration filter applied to the CG algorithm are parameters

Can be.

parameter

Is

,

And

Is selected from a lookup table based on

Represents the maximum delay spread,

Represents the maximum Doppler shift,

May represent the power of noise.

Determining the channel information for the remaining RE, the cross-covariance matrix

By applying to the cross-covariance matrix

May be a matrix representing a correlation between some REs and others.

parameter

Is a covariance matrix representing the correlations between some REs.

Is determined by using an autocovariance matrix.

silver

Can be calculated using an approximated formula of the function.

The terminal for solving the above technical problem includes a transmitter, a receiver, and a processor operating in connection with the transmitter and the receiver, wherein the processor is configured for some REs in the subframe using DMRS transmitted through a downlink subframe. Parameter for acquiring channel information and using the channel information of some REs to acquire channel information of the remaining REs in a subframe

To determine the parameters

Compensate for the initial value until the error of the applied initial value is less than the threshold value, and determine the channel information for the remaining RE using the compensated result.

According to embodiments of the present invention, the following effects can be expected.

First, in a wireless communication system, accurate downlink signal processing is possible by improving the accuracy of channel estimation in the process of the received signal by the terminal.

Secondly, according to the proposed invention, the added complexity can be minimized while improving the accuracy of downlink channel estimation.

Effects obtained in the embodiments of the present invention are not limited to the above-mentioned effects, and other effects not mentioned above are commonly known in the art to which the present invention pertains from the description of the embodiments of the present invention. Can be clearly derived and understood by those who have In other words, unintended effects of practicing the present invention may also be derived by those skilled in the art from the embodiments of the present invention.

BRIEF DESCRIPTION OF THE DRAWINGS The accompanying drawings are provided to help understand the present disclosure, and provide embodiments of the present disclosure with the detailed description. However, the technical features of the present invention are not limited to the specific drawings, and the features disclosed in the drawings may be combined with each other to constitute a new embodiment. Reference numerals in each drawing refer to structural elements.

1 is a diagram illustrating a Doppler spectrum.

2 is a diagram illustrating narrow beamforming according to the invention.

3 is a diagram illustrating Doppler spectra when narrow beamforming is performed.

4 is a diagram illustrating an example of a synchronization signal service zone of a base station.

5 is an example of a frame structure proposed in a communication environment using mmWave.

6 shows the structure of an Orthogonal Variable Spreading Factor (OVSF) code.

7 is a diagram illustrating an example of an arrangement of terminals.

8 illustrates a channel estimation process using a DeModulation Reference Signal (DMRS).

9 is a flowchart illustrating a downlink signal processing method according to an exemplary embodiment.

10 is a flowchart illustrating a channel interpolation process according to an exemplary embodiment.

11 is a diagram illustrating a configuration of a terminal and a base station related to the proposed embodiment.

The terms used in the present invention have been selected as widely used general terms as possible in consideration of the functions in the present invention, but this may vary according to the intention or precedent of the person skilled in the art, the emergence of new technologies and the like. In addition, in certain cases, there is also a term arbitrarily selected by the applicant, in which case the meaning will be described in detail in the description of the invention. Therefore, the terms used in the present invention should be defined based on the meanings of the terms and the contents throughout the present invention, rather than the names of the simple terms.

The following embodiments combine the components and features of the present invention in a predetermined form. Each component or feature may be considered to be optional unless otherwise stated. Each component or feature may be embodied in a form that is not combined with other components or features. In addition, some of the components and / or features may be combined to form an embodiment of the present invention. The order of the operations described in the embodiments of the present invention may be changed. Some components or features of one embodiment may be included in another embodiment, or may be replaced with corresponding components or features of another embodiment.

In the description of the drawings, procedures or steps which may obscure the gist of the present invention are not described, and procedures or steps that can be understood by those skilled in the art are not described.

Throughout the specification, when a part is said to "comprising" (or including) a component, this means that it may further include other components, except to exclude other components unless specifically stated otherwise. do. In addition, the terms "... unit", "... group", "module", etc. described in the specification mean a unit for processing at least one function or operation, which is hardware or software or a combination of hardware and software. It can be implemented as. Also, "a or an", "one", "the", and the like are used differently in the context of describing the present invention (particularly in the context of the following claims). Unless otherwise indicated or clearly contradicted by context, it may be used in the sense including both the singular and the plural.

In the present specification, embodiments of the present invention have been described based on data transmission / reception relations between a base station and a mobile station. Here, the base station is meant as a terminal node of a network that directly communicates with a mobile station. The specific operation described as performed by the base station in this document may be performed by an upper node of the base station in some cases.

That is, various operations performed for communication with a mobile station in a network consisting of a plurality of network nodes including a base station may be performed by the base station or network nodes other than the base station. In this case, the 'base station' may be replaced by terms such as a fixed station, a Node B, an eNode B (eNB), an advanced base station (ABS), or an access point.

In addition, a 'mobile station (MS)' may be a user equipment (UE), a subscriber station (SS), a mobile subscriber station (MSS), a mobile terminal, an advanced mobile station (AMS), a terminal. (Terminal) or a station (STAtion, STA) and the like can be replaced.

Also, the transmitting end refers to a fixed and / or mobile node that provides a data service or a voice service, and the receiving end refers to a fixed and / or mobile node that receives a data service or a voice service. Therefore, in uplink, a mobile station may be a transmitting end and a base station may be a receiving end. Similarly, in downlink, a mobile station may be a receiving end and a base station may be a transmitting end.

In addition, the description that the device communicates with the 'cell' may mean that the device transmits and receives a signal with the base station of the cell. That is, a substantial target for the device to transmit and receive a signal may be a specific base station, but for convenience of description, it may be described as transmitting and receiving a signal with a cell formed by a specific base station. Similarly, the description of 'macro cell' and / or 'small cell' may not only mean specific coverage, but also 'macro base station supporting macro cell' and / or 'small cell supporting small cell', respectively. It may mean 'base station'.

Embodiments of the present invention may be supported by standard documents disclosed in at least one of the wireless access systems IEEE 802.xx system, 3GPP system, 3GPP LTE system and 3GPP2 system. That is, obvious steps or parts which are not described among the embodiments of the present invention may be described with reference to the above documents.

In addition, all terms disclosed in the present document can be described by the above standard document. In particular, embodiments of the present invention may be supported by one or more of the standard documents P802.16e-2004, P802.16e-2005, P802.16.1, P802.16p, and P802.16.1b standard documents of the IEEE 802.16 system. have.

Hereinafter, exemplary embodiments of the present invention will be described in detail with reference to the accompanying drawings. The detailed description, which will be given below with reference to the accompanying drawings, is intended to explain exemplary embodiments of the present invention and is not intended to represent the only embodiments in which the present invention may be practiced.

In addition, specific terms used in the embodiments of the present invention are provided to help the understanding of the present invention, and the use of the specific terms may be changed to other forms without departing from the technical spirit of the present invention.

1. Communication system using ultra high frequency band

In LTE (Long Term Evolution) / LTE-A (LTE Advanced) system, the error value of the oscillator of the terminal and the base station is defined as a requirement, and is described as follows.

UE side frequency error (in TS 36.101)

The UE modulated carrier frequency shall be accurate to within ± 0.1 PPM observed over a period of one time slot (0.5 ms) compared to the carrier frequency received from the E-UTRA Node B

eNB side frequency error (in TS 36.104)

Frequency error is the measure of the difference between the actual BS transmit frequency and the assigned frequency.

Meanwhile, the oscillator accuracy according to the type of base station is shown in Table 1 below.

BS class	Accuracy
Wide Area BS	± 0.05 ppm
Local Area BS	± 0.1 ppm
Home BS	± 0.25 ppm

Therefore, the maximum difference of the oscillator between the base station and the terminal is ± 0.1ppm, and when an error occurs in one direction, a maximum offset value of 0.2 ppm may occur. This offset value is multiplied by the center frequency and converted into Hz units for each center frequency.

On the other hand, in the OFDM system, the CFO value is shown differently by subcarrier spacing, and in general, even if the CFO value is large, the effect of the OFDM system with a sufficiently large subcarrier spacing is relatively small. Therefore, the actual CFO value (absolute value) needs to be expressed as a relative value affecting the OFDM system, which is called a normalized CFO. The normalized CFO is expressed by dividing the CFO value by the subcarrier spacing. Table 2 below shows the CFO and normalized CFO for each center frequency and oscillator error value.

Center frequency (subcarrier spacing)	Oscillator offset
	± 0.05ppm	± 0.1ppm	± 10 ppm	± 20ppm
2 GHz (15 kHz)	± 100 Hz (± 0.0067)	± 200 Hz (± 0.0133)	± 20 kHz (± 1.3)	± 40 kHz (± 2.7)
30 GHz (104.25 kHz)	± 1.5 kHz (± 0.014)	± 3 kHz (± 0.029)	± 300 kHz (± 2.9)	± 600 kHz (± 5.8)
60 GHz (104.25 kHz)	± 3 kHz (± 0.029)	± 6 kHz (± 0.058)	± 600 kHz (± 5.8)	± 1.2 MHz (± 11.5)

In Table 2, a subcarrier spacing (15 kHz) is assumed for a center frequency of 2 GHz (for example, LTE Rel-8 / 9/10), and a subcarrier spacing of 104.25 kHz for a center frequency of 30 GHz or 60 GHz. This prevents performance degradation considering the Doppler effect for each center frequency. Table 2 above is a simple example and it is apparent that other subcarrier spacings may be used for the center frequency.

On the other hand, in a situation where the terminal moves at a high speed or in a high frequency band, a Doppler spread phenomenon greatly occurs. Doppler dispersion causes dispersion in the frequency domain, resulting in distortion of the received signal at the receiver's point of view. Doppler dispersion

It can be expressed as. In this case, v is the moving speed of the terminal, and λ means the wavelength of the center frequency of the transmitted radio waves. θ means the angle between the received radio wave and the moving direction of the terminal. The following description is based on the assumption that θ is zero.

In this case, the coherence time is in inverse proportion to the Doppler variance. If the coherence time is defined as a time interval in which the correlation value of the channel response in the time domain is 50% or more,

It is expressed as In a wireless communication system, Equation 1 below is mainly used which represents a geometric mean between the equation for Doppler variance and the equation for coherence time.

[Equation 1]

1 is a diagram illustrating a Doppler spectrum.

The Doppler spectrum, or Doppler power spectrum density, which represents a change in Doppler value according to the frequency change, may have various shapes according to a communication environment. In general, in an environment where scattering occurs a lot, such as downtown, if the received signal is received at the same power in all directions, the Doppler spectrum appears in the U-shape as shown in FIG. 1 shows the center frequency

And the maximum Doppler variance

U-shaped Doppler spectra are shown.

FIG. 2 is a diagram showing narrow beamforming according to the present invention, and FIG. 3 is a diagram showing Doppler spectrum when narrow beamforming is performed.

Since the ultrahigh frequency wireless communication system is located in a band having a very high center frequency, an antenna array including a plurality of antennas may be installed in a small space with a small antenna. This feature enables pin-point beamforming, pencil beamforming, narrow beamforming, or thin beamforming using tens to hundreds of antennas. This narrow beamforming means that the received signal is received only at a certain angle, not in the same direction.

FIG. 2A illustrates a case where the Doppler spectrum is U-shaped according to a signal received in an equal direction, and FIG. 2B illustrates a case where narrow beamforming using a plurality of antennas is performed.

As such, when the narrow beamforming is performed, the Doppler spectrum also appears narrower than the U-shape due to the reduced angular spread. As shown in FIG. 3, it can be seen that Doppler variance appears only in a certain band when the narrow beamforming is performed.

In the wireless communication system using the ultra-high frequency band described above, the center frequency operates in the band of several GHz to several tens of GHz. This characteristic of the center frequency makes the influence of the CFO due to the Doppler effect or the oscillator difference between the transmitter / receiver caused by the movement of the terminal more serious.

4 is a diagram illustrating an example of a synchronization signal service zone of a base station.

The terminal performs synchronization with the base station by using a downlink (DL) synchronization signal transmitted by the base station. In this synchronization process, timing and frequency are synchronized between the base station and the terminal. In the synchronization process, the base station transmits the synchronization signal by configuring the beam width as wide as possible so that terminals in a specific cell can receive and use the synchronization signal.

On the other hand, in the mmWave communication system using a high frequency band, path loss is greater than that of a low frequency band in synchronizing signal transmission. That is, in the case of a system using a high frequency band, a cell radius that can be supported compared to a conventional cellular system (for example, LTE / LTE-A) using a relatively low frequency band (for example, 6 GHz or less). This is greatly toned.

As one method for solving such a reduction in cell radius, a synchronization signal transmission method using beamforming may be used. When beamforming is used, the cell radius is increased, but the beam width is reduced. Equation 2 below shows the change in the received signal SINR according to the beam width.

[Equation 2]

Equation 2 is the beam width according to the beamforming

If received decreases, the received SINR is

Fold improvement.

In addition to the beamforming method, another method for solving the reduction of the cell radius may be considered to repeatedly transmit the same sync signal. This method requires additional resource allocation on the time axis, but has the advantage of increasing the cell radius without reducing the beam width.

Meanwhile, the base station allocates resources to each terminal by scheduling frequency resources and time resources located in a specific area. Hereinafter, this specific zone is defined as a sector. In the sectors shown in FIG. 4, A1, A2, A3, and A4 represent sectors having a radius of 0 to 200 m and widths of 0 to 15 ', 15 to 30', 30 to 45 ', and 45 to 60', respectively. B1, B2, B3, and B4 represent sectors having a radius of 200 to 500 m and widths of 0 to 15 ', 15 to 30', 30 to 45 ', and 45 to 60', respectively. Based on the contents shown in FIG. 4, sector 1 is defined as {A1, A2, A3, A4}, and sector 2 is defined as {A1, A2, A3, A4, B1, B2, B3, B4}. In addition, if the synchronization signal service area of the current base station is sector 1, it is assumed that an additional power of 6 dB or more is required for transmission of the synchronization signal in order for the base station to service the synchronization signal in sector 2.

First, the base station can obtain an additional gain of 6 dB using the beamforming technique to serve sector 2. Through this beamforming process, the service radius can be increased from A1 to B1. However, since beam width is reduced through beamforming, A2, A3, and A4 cannot be serviced at the same time. Therefore, when beamforming is performed, a synchronization signal should be separately transmitted to the A2 to B2, A3 to B3, and A4 to B4 sectors. In other words, the base station must transmit a synchronization signal four times beamforming to serve sector 2.

On the other hand, considering the repetitive transmission of the synchronization signal described above, the base station can transmit the synchronization signal to all sectors 2, but must transmit the synchronization signal four times on the time axis. As a result, the resources required to service sector 2 are the same for both beamforming and iterative transmission.

However, in the beamforming method, since the beam width is narrow, it is difficult for a terminal moving at a high speed or a terminal at the boundary of a sector to stably receive a synchronization signal. Instead, if the ID of the beam in which the terminal is located can be distinguished, there is an advantage that the terminal can determine its own position through a synchronization signal. On the other hand, in the repetitive transmission scheme, since the beam width is wide, it is very unlikely that the terminal misses the synchronization signal. Instead, the terminal cannot determine its location.

5 is an example of a frame structure proposed in a communication environment using mmWave.

First, one frame consists of Q subframes and one subframe consists of P slots. One slot consists of T OFDM symbols. At this time, unlike the other subframes, the first subframe in the frame uses the 0 th slot (slot indicated by 'S') for synchronization purposes. The 0 th slot is composed of A OFDM symbols for timing and frequency synchronization, B OFDM symbols for beam scanning, and C OFDM symbols for informing the UE of system information. The remaining D OFDM symbols are used for data transmission to each terminal.

Meanwhile, such a frame structure is merely a mere example, and Q, P, T, S, A, B, C, and D may each be arbitrary values and may be values set by a user or automatically set on a system.

Hereinafter, a timing synchronization algorithm between a base station and a terminal will be described. Consider a case in which the base station repeatedly transmits the same synchronization signal A times in FIG. 5. The terminal performs timing synchronization using an algorithm of Equation 3 based on the synchronization signal transmitted from the base station.

[Equation 3]

In equation (3)

, N _g , i represent the length of an OFDM symbol, the length of a cyclic prefix (CP), and the index of an OFDM symbol, respectively.

Denotes a vector of the received signal at the receiver. At this time,

Cold signal vector

of

From the first

Vector defined by the first element.

The algorithm of Equation 3 operates under the condition that two adjacent OFDM received signals in time are the same. Such an algorithm can use a sliding window method, which can be implemented with low complexity, and has a strong characteristic of frequency offset.

Meanwhile, Equation 4 below represents an algorithm for performing timing synchronization by using a correlation between a received signal and a signal transmitted by a base station.

[Equation 4]

In Equation 4, s denotes a signal transmitted by the base station and is a signal vector previously promised between the terminal and the base station. Equation 4 may produce better performance than Equation 3, but may not be implemented as a sliding window method, and thus requires high complexity. It also has a feature that is vulnerable to frequency offset.

Following the description of the timing synchronization method, the beam scanning process will be described. Beam scanning refers to the operation of the transmitter and / or receiver to find the direction of the beam that maximizes the receiver's received SINR. For example, the base station determines the direction of the beam through beam scanning before transmitting data to the terminal.

Referring to FIG. 4 as an example, FIG. 4 illustrates a sector served by one base station divided into eight regions. At this time, the base station transmits beams in the areas (A1 + B1), (A2 + B2), (A3 + B3), and (A4 + B4), respectively, and the terminal can distinguish beams transmitted by the base station. In this condition, the beam scanning process can be embodied in four processes. First, i) the base station transmits a beam in four areas in sequence. ii) The terminal determines the beam that is determined to be the most suitable among the beams in view of the received SINR. iii) The terminal feeds back information on the selected beam to the base station. iv) The base station transmits data using the beam having the feedback direction. Through the above beam scanning process, the UE can receive downlink data through the beam with optimized reception SINR.

Hereinafter, the Zadoff-Chu sequence will be described. The Zadoff-Chu sequence is called a chu sequence or ZC sequence and is defined by Equation 5 below.

[Equation 5]

In Equation 5, N is the length of the sequence, r is the root value,

Represents the n th element of the ZC sequence. A characteristic of the ZC sequence is that all elements have the same size (constant amplitude). In addition, the DFT results of the ZC sequence also appear the same for all elements.

Next, the ZC sequence and the cyclic shifted version of the ZC sequence have a correlation as shown in Equation (6).

[Equation 6]

In equation (6)

Is

Is a sequence of circular shifted by i, and represents 0 except that the autocorrelation of the ZC sequence is i = j. In addition, the ZC sequence also has a zero auto-correlation property, it is also expressed as having a constant Amplitude Zero Auto Correlation (CAZAC).

As a final feature of the ZC sequence, there is a correlation between the length N of the sequence and the ZC sequences having root values that are mutually smaller than Equation 7 below.

[Equation 7]

In equation (7)

Is mutually different from N. For example, if N = 111

Always satisfies equation (7). Unlike the autocorrelation of Equation 6, the cross-correlation of the ZC sequence is not completely zero.

Following the ZC sequence, the Hadamard matrix is described. Hadamard matrix is defined as Equation 8 below.

[Equation 8]

In equation (8)

Denotes the size of the matrix. Hadamard matrices are always independent of size n

Is a unitary matrix that satisfies. Also, in the Hadamard matrix, all columns and all rows are orthogonal to each other. For example, when n = 4, the Hadamard matrix is defined as in Equation 9.

[Equation 9]

It can be seen from Equation 9 that the columns are orthogonal to each other.

6 shows the structure of an Orthogonal Variable Spreading Factor (OVSF) code. The OVSF code is generated based on the Hadamard matrix and has a specific rule.

First, when branching to the right side of the OVSF code (lower branch), the first code repeats the upper code on the left side twice (mother code), and the second code repeats the high code code once and inverts it once. Is generated. 6 shows a tree structure of the OVSF code.

All of these OVSF codes are orthogonal except for the relationship between adjacent higher and lower codes on the code tree. For example, in FIG. 6, the code [1 -1 1 -1] is orthogonal to [1 1], [1 1 1 1], and [1 1 -1 -1]. In addition, the OVSF code has the same length as the code length. That is, in FIG. 6, it can be seen that the length of a specific code is equal to the total number of branches to which the corresponding code belongs.

7 is a diagram illustrating an example of an arrangement of terminals. In FIG. 7, a random access channel (RACH) will be described.

In the LTE system, when the RACH signals transmitted by the terminals arrive at the base station, the RACH signal power of the terminals received by the base station should be the same. To this end, the base station defines a parameter called 'preambleInitialReceivedTargetPower', and broadcasts the parameter to all terminals in the cell through SIB (System Information Block) 2. The UE calculates a path loss using a reference signal, and determines the transmission power of the RACH signal by using the calculated path loss and the 'preambleInitialReceivedTargetPower' parameter as shown in Equation 10 below.

[Equation 10]

In Equation 10, P_PRACH_Initial, P_CMAX, and PL represent the transmission power of the RACH signal, the maximum transmission power of the terminal, and the path loss, respectively.

Referring to Equation 10 as an example, it is assumed that the maximum transmit power of the terminal is 23 dBm and the RACH reception power of the base station is -104 dBm. In addition, it is assumed that the terminal is arranged as shown in FIG.

First, the terminal calculates a path loss using the received synchronization signal and the beam scanning signal, and determines the transmission power based on this. Table 3 below shows the path loss of the terminal and its transmission power.

Terminal	preambleInitialReceivedTargetPower	Path loss	Transmission power required	Transmission power	Extra power needed
K1	-104dBm	60 dB	-44 dBm	-44 dBm	0 dBm
K2	-104dBm	110 dB	6 dBm	6 dBm	0 dBm
K3	-104dBm	130 dB	26 dBm	23 dBm	3 dBm

In Table 3, although the path loss is very small in the case of the K1 terminal, the RACH signal must be transmitted with a very small power (-44 dBm) to match the RACH reception power. Meanwhile, in the case of the K2 terminal, the path loss is large, but the required transmission power is 6 dBm. However, in the case of the K3 terminal, the path loss is so large that the required transmission power exceeds the P_CMAX = 23 dBm of the terminal. In this case, the terminal should transmit at 23 dBm, which is the maximum transmission power, and the RACH access success rate of the terminal is degraded by 3 dB.

2. Proposed Downlink Signal Processing Method

Hereinafter, a method of processing a downlink signal received by a terminal from a base station will be described. 8 illustrates a channel estimation process using a DeModulation Reference Signal (DMRS). The UE estimates the downlink channel using the DMRS included in the downlink subframe, and decodes downlink data according to the estimated channel.

Meanwhile, the UE directly calculates a channel for a RE (Resource Element) loaded with a DMRS in a subframe, but interpolates a channel based on a channel of the DMRS RE for other REs. Estimate. Various methods can be used for this interpolation process, and a method of using a 2D Weiner filter will be described.

In the channel interpolation method using the 2D Winer filter, channel interpolation is simultaneously performed on the frequency axis and the time axis as in Equation 11 below.

[Equation 11]

In equation (11)

Denotes a channel value estimated from the DMRS of the antenna port p, and its size is 12 × 1. Covariance matrix

Denotes auto-covariance between DMRSs of antenna port p,

The (m, n) th element of the matrix is expressed by Equation 12 below.

[Equation 12]

In equation (12)

Denotes a maximum Doppler shift, an OFDM symbol length, a maximum delay spread, and a subcarrier spacing, respectively. And,

Denotes the time axis position and the frequency axis position of the m th RE, respectively. For example, in boxes 810 and 820 indicated by dotted lines in FIG. 8, 12 REs defined by hatched lines are defined, and the 12 REs have the same structure as the DMRS of the LTE system. Since the REs are code division multiplexed (CDM), the receiver can estimate one channel value from the two REs using an orthogonal cover code (OCC). In FIG. 8, the index of the RE is indexed by increasing from left to right along the time axis, and then moved down along the frequency axis to increase again from left to right, and indexed by 0, 1, 2, ..., 5. In this case, t (0) -t (1) = 5 and f (0) -f (1) = 0. If m = 0 and n = 3, then t (0) -t (3) = 5 and f (0) -f (3) 5 =.

Meanwhile, in Equation 11

Denotes a cross-covariance representing the correlation between the DMRS RE and the remaining REs.

The (m, n) th element of the matrix is defined as in Equation 13 below.

[Equation 13]

In Equation 11,

Is

Is the result generated by passing the interpolation filter, and its size is 168 × 1. In equation (11)

Represents the power of noise.

Meanwhile, in Equation 11

Computing requires a lot of complexity due to the inverse computation. Accordingly, the terminal is inverse matrix

Is stored in the form of a look-up table and used according to Equation 14 below.

[Equation 14]

According to Equation 14, the terminal is

As the value is determined, the corresponding inverse is selected. At this time,

Is an approximated inverse value selected using the lookup table,

Represents the exact inverse obtained through the calculation. That is, the terminal selects an inverse matrix matching the parameters, instead of directly calculating the inverse matrix.

On the other hand, since the lookup table method requires a storage space, the resolution of the lookup table is inevitably limited in a terminal having a limited storage space. The lower the resolution of the lookup table, the lower the performance of channel estimation using the 2D Wiener filter, which is a problem. Hereinafter, an embodiment of improving the performance degradation described above will be described.

9 is a flowchart illustrating a downlink signal processing method according to an exemplary embodiment. According to the proposed embodiment, the accuracy of the channel estimation result can be improved by combining the method using the 2D Wiener filter and the method using the numerical analysis algorithm.

Before describing an embodiment, the numerical analysis algorithm is an algorithm that calculates a numerical approximation value from a certain value, and examples thereof include the Newton method and the conjugate gradient method. The principle of the numerical analysis algorithm may be expressed by Equation 15 below.

[Equation 15]

In equation (15)

Represents a matrix,

Denote an input vector and an output vector, respectively.

If there is an inverse of the matrix, the simplest way to get the input vector from the output vector is

[Equation 16]

On the other hand, the numerical algorithm is inverse matrix

Instead of calculating directly, the iterative calculation process is performed as shown in Equation 17 below.

from

Find it.

[Equation 17]

In equation (17)

Is an estimate of the input vector found in the nth iteration,

Is

Is an additional variable used in the process of calculating. As the number of repetitions in Equation 17 increases

Is

Converges to the error value of the inverse matrix

Means smaller.

An embodiment using the above-described numerical analysis algorithm will be described in detail with reference to FIG. 9.

When the UE receives the downlink signal from the base station (S910), the UE estimates a downlink channel of some REs (that is, the RE where the DMRS is located) using the DMRS included in the downlink subframe (S920). Subsequently, the UE estimates channels of the remaining REs except for the DMRS using the estimated result value for the DMRS RE (S930). This process may be understood as a process of interpolating the calculated result values for the DMRS RE. If the channels for all REs are estimated, the terminal decodes downlink data using the channel estimation result (S940).

In the above-described series of processes, the S930, that is, the 2D Wiener filter may be used in the process of interpolating the result of the channel calculated for the DMRS RE, as described above. Hereinafter, an exemplary embodiment of a channel interpolation process will be described with reference to FIG. 10.

First, the terminal from the information on the channel calculated for the DMRS RE

To calculate,

from

Are respectively calculated (S1010). This calculation process may be performed according to the process described in Equations 11 to 13. Subsequently, the terminal determines parameters in S1010.

Using

In step S1020, this process may be performed by using a lookup table as described in Equation 14.

On the other hand, the value obtained in S1020

instead

Expressed as

It means that the value can be used as the initial value of the numerical analysis algorithm to be described later. The terminal is an initial interpolation value based on the result values in S1010 and S1020

To calculate (S1030).

Subsequently, the terminal

And

Based on the numerical algorithm,

It is obtained (S1040). In the illustrated embodiment, a case where the CG algorithm is applied as an example of a numerical analysis algorithm will be described. The resulting value of the numerical algorithm

Cross-covariance

Is applied to the final result value

Is obtained.

Is the final result of the interpolation process and means the output for the channel.

Equation 18 more specifically illustrates a process of applying a numerical analysis algorithm (CG algorithm).

Equation 18

In equation (18)

Denotes an initial filter and an acceleration filter applied to the CG algorithm, respectively. According to one embodiment, the initial filter and the acceleration filter applied to the CG algorithm of FIG.

Calculated by

Can be (i.e.

). In equation (18)

Values are auxiliary variables used to solve solutions in numerical algorithms and have no special meaning.

The numerical algorithm of Equation 18 is an initial value.

The process of compensating is repeatedly performed until it is smaller than the predetermined threshold. That is, in Equation 18

Is not satisfied (that is,

Is the initial value

If it is small enough compared to the) the end of the repetition of the compensation process. At this time, the variable to control the accuracy of the result value

Can be set to 0.01,

The larger the value, the faster the algorithm terminates, resulting in a faster result, but less accurate.

The smaller the value, the better the accuracy of the calculated result, instead of the algorithm terminating slowly.

According to the embodiment described above, the terminal can improve the accuracy of the result value by applying a numerical analysis algorithm to the result value obtained using the lookup table. In addition, since the initial filter and the acceleration filter of the numerical algorithm application process need not be implemented or generated separately, the computational complexity required for driving the numerical algorithm is minimized. As a result, the terminal can obtain an improved channel interpolation result value with only minimal complexity, and accordingly, a disadvantage of the method of using the lookup table can be improved.

Hereinafter, the covariance matrix

An embodiment of calculating the approximation to is proposed. As previously described in Equation 12, the covariance matrix

Each element of

Is defined as

Is

It is expressed as a function. At this time,

The complexity of the function is not low, and it is important to calculate one covariance matrix.

The function's call is made twice as large as the total matrix. That is, if the complexity of the calculation process of the covariance matrix is improved, the complexity of the above-described embodiments may be further reduced.

Meanwhile,

The function may be redefined as in Equation 19 below.

[Equation 19]

If c is much smaller than 1 in Equation 19, Equation 19 is approximated as Equation 20 below.

[Equation 20]

Substantially covariance matrix

The value of c used in is very small and there is no big error even when Q = 3. As a result,

When the function-based covariance matrix is approximated and applied as in Equation 20, an increase in the complexity of the covariance matrix calculation process according to the present embodiment can be minimized.

3. Device Configuration

11 is a diagram illustrating a configuration of a terminal and a base station according to an embodiment of the present invention. In FIG. 11, the terminal 100 and the base station 200 may include radio frequency (RF) units 110 and 210, processors 120 and 220, and memories 130 and 230, respectively. 11 illustrates only a 1: 1 communication environment between the terminal 100 and the base station 200, a communication environment may be established between a plurality of terminals and a plurality of base stations. In addition, the base station 200 illustrated in FIG. 11 may be applied to both the macro cell base station and the small cell base station.

Each RF unit 110, 210 may include a transmitter 112, 212 and a receiver 114, 214, respectively. The transmitting unit 112 and the receiving unit 114 of the terminal 100 are configured to transmit and receive signals with the base station 200 and other terminals, and the processor 120 is functionally connected with the transmitting unit 112 and the receiving unit 114. In connection, the transmitter 112 and the receiver 114 may be configured to control a process of transmitting and receiving signals with other devices. In addition, the processor 120 performs various processes on the signal to be transmitted and transmits the signal to the transmitter 112, and performs the process on the signal received by the receiver 114.

If necessary, the processor 120 may store information included in the exchanged message in the memory 130. With such a structure, the terminal 100 can perform the method of various embodiments of the present invention described above.

The transmitter 212 and the receiver 214 of the base station 200 are configured to transmit and receive signals with other base stations and terminals, and the processor 220 is functionally connected to the transmitter 212 and the receiver 214 to transmit the signal. 212 and the receiver 214 may be configured to control the process of transmitting and receiving signals with other devices. In addition, the processor 220 may perform various processing on the signal to be transmitted, transmit the signal to the transmitter 212, and may perform processing on the signal received by the receiver 214. If necessary, the processor 220 may store information included in the exchanged message in the memory 230. With such a structure, the base station 200 may perform the method of the various embodiments described above.

Processors 120 and 220 of the terminal 100 and the base station 200 respectively instruct (eg, control, coordinate, manage, etc.) the operation in the terminal 100 and the base station 200. Respective processors 120 and 220 may be connected to memories 130 and 230 that store program codes and data. The memories 130 and 230 are coupled to the processors 120 and 220 to store operating systems, applications, and general files.

The processor 120 or 220 of the present invention may also be referred to as a controller, a microcontroller, a microprocessor, a microcomputer, or the like. The processors 120 and 220 may be implemented by hardware or firmware, software, or a combination thereof.

When implementing an embodiment of the present invention using hardware, application specific integrated circuits (ASICs) or digital signal processors (DSPs), digital signal processing devices (DSPDs), and programmable logic devices (PLDs) configured to perform the present invention. Field programmable gate arrays (FPGAs) may be provided in the processors 120 and 220.

Meanwhile, the above-described method may be written as a program executable on a computer, and may be implemented in a general-purpose digital computer which operates the program using a computer readable medium. In addition, the structure of the data used in the above-described method can be recorded on the computer-readable medium through various means. Program storage devices that may be used to describe storage devices that include executable computer code for performing the various methods of the present invention should not be understood to include transient objects, such as carrier waves or signals. do. The computer readable medium includes a storage medium such as a magnetic storage medium (eg, a ROM, a floppy disk, a hard disk, etc.), an optical reading medium (eg, a CD-ROM, a DVD, etc.).

It will be understood by those skilled in the art that embodiments of the present invention can be implemented in a modified form without departing from the essential characteristics of the above description. Therefore, the disclosed methods should be considered in descriptive sense only and not for purposes of limitation. The scope of the present invention is shown in the claims rather than the detailed description of the invention, and all differences within the equivalent scope should be construed as being included in the scope of the present invention.

The received signal processing method as described above may be applied to various wireless communication systems including not only 3GPP LTE and LTE-A systems, but also IEEE 802.16x and 802.11x systems. Furthermore, the proposed method can be applied to mmWave communication system using ultra high frequency band.

Claims (14)

1. In the method for the terminal to process the received signal received from the base station,

   Acquiring channel information on some resource elements (REs) in the subframe using a DeModulation Reference Signal (DMRS) transmitted through a downlink subframe;

   A parameter for obtaining channel information on the remaining REs in the subframe using the channel information on the some REs

   

   Determining;

   The parameter

   

   Compensating the initial value until the error of the initial value to which the applied value is less than the threshold value; And

   And determining channel information for the remaining REs using the compensated results.
2. The method of claim 1,

   The received signal processing method further includes decoding downlink data included in the subframe using the channel result for the partial RE and the channel result for the remaining RE.
3. The method of claim 1,

   The compensating step is performed using a Conjugate Gradient (CG) algorithm or a Newton method.
4. The method of claim 3,

   The initial filter and the acceleration filter applied to the CG algorithm are the parameters

   

   The received signal processing method.
5. The method of claim 1,

   The parameter

   

   Is

   

   ,

   

   And

   

   Is selected from a lookup table based on

   

   Represents a maximum delay spread, and

   

   Represents the maximum Doppler shift, and

   

   Is a signal representing the power of noise.
6. The method of claim 1,

   Determining the channel information for the remaining RE, cross-covariance matrix of the compensated result

   

   Performed by applying to the cross-covariance matrix

   

   Is a matrix representing a correlation between the some REs and the remaining REs.
7. The method of claim 1,

   The parameter

   

   Is a covariance matrix representing the correlation between some REs

   

   Determined by using the self-covariance matrix

   

   silver

   

   The received signal processing method, which is calculated using an approximated formula of the function.
8. A terminal for processing a received signal received from a base station,

   A transmitter;

   Receiver; And

   A processor operating in connection with the transmitter and the receiver,

   The processor,

   Obtaining channel information on some REs in the subframe using DMRS transmitted through a downlink subframe,

   A parameter for obtaining channel information on the remaining REs in the subframe using the channel information on the some REs

   

   To determine,

   The parameter

   

   Compensates for the initial value until the error of the applied initial value is below the threshold,

   And determining channel information on the remaining REs using the compensated result.
9. The method of claim 8,

   The processor decodes downlink data included in the subframe using the channel result for the partial RE and the channel result for the remaining RE.
10. The method of claim 8,

    The processor is to compensate for the initial value using a CG algorithm or Newton scheme.
11. The method of claim 10,

    The initial filter and the acceleration filter applied to the CG algorithm are the parameters

    

    It is a terminal.
12. The method of claim 8,

    The parameter

    

    Is

    

    ,

    

    And

    

    Is selected from a lookup table based on

    

    Represents a maximum delay spread, and

    

    Represents the maximum Doppler shift, and

    

    Denotes the power of noise.
13. The method of claim 8,

    The processor cross-compensates the compensated result.

    

    Determine channel information for the remaining REs by applying to the cross covariance matrix.

    

    Is a matrix representing a correlation between the some REs and the remaining REs.
14. The method of claim 8,

    The parameter

    

    Is a covariance matrix representing the correlation between some REs

    

    Determined by using the self-covariance matrix

    

    silver

    

    Terminal, which is calculated using an approximated formula of the function.

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