WO2017054658A1

WO2017054658A1 - Transceiver devices

Info

Publication number: WO2017054658A1
Application number: PCT/CN2016/099333
Authority: WO
Inventors: Thomas Winiecki
Original assignee: Jrd Communication Inc.
Priority date: 2015-09-28
Filing date: 2016-09-19
Publication date: 2017-04-06
Also published as: GB2542625B; CN108141241A; GB201517090D0; CN108141241B; CN110971257B; CN110971257A; GB2542625A

Abstract

A transceiver for a wireless communication device which meets noise performance and intermodulation component emission requirements without the need for high cost surface acoustic wave filters in front-end circuitry is provided. Low-cost, LC filters (211-216) are used instead and tolerance to blockers is achieved by deliberately degrading second order distortion tolerance in the receive path (224) and taking measurements of received signal strength while switching between transmit and receive modes and adjusting the low noise amplifier (226) gain depending on the measurement. In transmitter circuitry, introducing a pre-chosen baseband frequency shift in the digital domain and compensating for the shift by an adjustment of the local oscillator (231) signal forces intermodulation frequencies generated by nonlinear baseband components (235, 236, 237) to fall outside rather than inside protected transmission bands.

Description

TRANSCEIVER DEVICES

TECHNICAL FIELD

Embodiments of the present invention generally relate to transceiver devices which have particular application in wireless communication systems using Time Division Duplex (TDD) or a combination of time and frequency division duplexing.

BACKGROUND

Wireless communication systems, such as the third-generation (3G) of mobile telephone standards and technology are well known. Such 3G standards and technology have been developed by the Third Generation Partnership Project. Communication systems and networks have further developed towards a broadband and mobile system. The Third Generation Partnership Project has developed a Long Term Evolution (LTE) solution. The Third Generation Partnership Project Release 12 and Release 13 specify certain requirements for a mobile terminal (or user equipment) . A category 0 user equipment (UE) may access the air interface using the so-called half duplex frequency division duplexing (HD-FDD) whereby transmitted and received signals occupy different channels associated with paired frequency spectrum and additionally, are sent and received in separate timeslots.

A known transceiver device which operates using half duplex frequency division duplexing (HD-FDD) implements surface acoustic wave (SAW) filters, with their sharp cut-off characteristics, for signal conditioning. For each supported frequency band, a dedicated SAW filter is required unless two supported bands overlap as is the case for bands 5 and 26 or bands 2 and 25, for example. If several frequency bands are supported by the UE but the number of receive inputs of the transceiver device is limited, RF band switches are also required. This increases cost and receiver insertion loss. .Figure 1 shows the basic architecture for a known HD-FDD transceiver for a single frequency band. The transceiver comprises a module 100 which includes base-band and RF circuitry implemented in an integrated circuit. The transceiver also comprises a switch arrangement 101 which switches a single antenna 102 between transmit and receive

lines

103 and 104 respectively. The transmit line 103 shows a power amplifier 105 and a transmit band-pass SAW filter 106. The receive line 104 includes a receive band-pass SAW 107. The components of Figure 1 are typically mounted on a printed circuit board (PCB) . The requirements of the transmit filter 106 may vary from band to band. In some bands, stringent emission specifications apply near the transmit frequency. This necessitates dedicated filtering and in such cases a SAW filter is typically used to keep the unwanted emissions low. The receive SAW filter 107 is used to provide rejection of out-of-band blockers (or jamming signals) and the Third Generation Partnership Project standard defines a number of test cases that the receiver must comply with. Typically, SAW filters are employed in order to meet these requirements. For UEs operating using HD-FDD in particular, another challenge is that they need to co-exist with other UEs accessing the same band but operating in either FDD mode or in HD-FDD mode but occupying different timeslots for transmission and reception. For example, a signal transmitted by HDD-FDD UE can represent a potentially very large interfere to other similar UEs operating in close proximity.

SAW filters take up a large area on the PCB and are expensive to manufacture. The necessary switches that are required in a transceiver employing SAW filters also add to size and cost. Therefore it would be advantageous to provide a transceiver which did not require the use of SAW filters yet would still perform satisfactorily. As different countries specify different frequency bands for mobile communications, it would also be advantageous to provide a transceiver for a UE that could be used worldwide.

A problem presented to the designer of a receiver for mobile communications is that of tolerance to blockers (or jamming signals) . In known devices, the detection of blockers is performed using RMS power detectors or envelope detectors. Ideally, the detector is placed between the first and second stage of a low noise amplifier at which point the circuit is not overly sensitive but the signal is still wide-band. However such circuits occupy significant silicon area, consume power and are difficult to design for accurate power measurement across a wide range of conditions. Moreover, averaging power is not a trivial exercise and typically employs a combination of analogue and digital averaging which is difficult to implement. Hence it would be advantageous to provide a simplified means for detecting and compensating for the presence of blockers in a mobile communications receiver.

A problem presented to the designer of a transmitter for mobile communications is the rejection of unwanted transmitted frequency products (such as intermodulation components) which fall into protected bands. Typical requirements for a transmitter filter in a UE may vary from band to band. For most bands a simple low pass filter may be sufficient to suppress power amplifier harmonic output, for example, at twice or three times the desired carrier frequency. In some bands however, stringent emission specifications apply near the transmit frequency which may necessitates dedicated filtering. Conventionally, in such cases a SAW filter is used to keep emissions low. It would be advantageous to provide a transmitter which used lower cost filters yet could still meet emission specifications.

Embodiments of the present invention provide a transceiver architecture which does not require the use of SAW filters or their associated switching components yet whose performance is comparable with a transceiver which does include SAW filters. Further embodiments provide a receiver having a simplified means for detecting blocker and a transmitter having a means for rejecting unwanted transmitted frequency products.

The embodiments described below are not limited to implementations which solve any or all of the disadvantages of known systems.

SUMMARY

This Summary is provided to introduce a selection of concepts in a simplified form that are further described below in the Detailed Description. This Summary is not intended to identify key features or essential features of the claimed subject matter, nor is it intended to be used as an aid in determining the scope of the claimed subject matter.

According to a first aspect of the present invention, there is provided a transceiver having a front end module including N transmit paths and N receive paths, where N is an integer, each of said transmit and receive paths including a filter, wherein a pass-band of each filter is chosen such that harmonics of in-band signals are filtered out and wherein Ni is chosen such that a total pass-band of the transceiver extends across a plurality of radio access networks.

The transceiver may be incorporated in a user equipment or any other form of wireless communication device or in a base station arranged for wireless communication with a wireless communication device.

Preferably, the choice of frequency pass bands of the filters is such that any user equipment incorporating the transceiver may be used worldwide. The filters may comprise low-pass filters or band-pass filters or a combination of the two. The filters may be of simple construction of a type which is cheap to produce and which does not take up a large area compared with SAW filters. Advantageously, the transceiver does not require the RF band switches which are required when using SAW filters, thus providing further cost and size savings. In one embodiment, the filters in the receive paths comprise fifth order LC filters using a Chebyshev 1 topology. Removal of SAW filters and RF switches from a transceiver also reduces receiver insertion losses. This results in an improved sensitivity.

Preferably, the bandwidth of each transmit and receive path is chosen to be low enough to allow impedance matching and band tuning.

Preferably, the frequency range of each receive port in each receive path is set to be well below an octave in order to allow efficient filtering of harmonics.

Preferably, the frequency range of each transmit and receive path is chosen to be sufficiently broad to include all bands of interest yet keep the number of receive ports needed for worldwide band coverage low. For example, for LTE category 0 use, three pass bands, and therefore three filters in each of the receive and transmit paths may be employed. The signal path selected does not depend on the actual LTE frequency band chosen but many on the broad frequency range in which the used LTE band falls

In one embodiment, an ISM band rejection filter is provided at an antenna port in order to suppress both transmitter noise into the ISM band as well as blockers. This refinement is particularly useful in cases where high band LTE reception and transmission may have to co-exist with Wi-Fi radios operating in close proximity.

According to a second aspect of the present invention, there is provided a method for optimising the gain of an amplifier in a receiver, the method comprising: introducing a second-order distortion into the receiver, ； measuring a magnitude of received signal strength； adjusting the gain of said amplifier by an amount depending on the magnitude of the measured signal strength； and removing the second order distortion..

According to a third aspect of the present invention, there is provided a receiver comprising a variable gain amplifier and a processor, wherein the processor is adapted to introduce a second order distortion into the receiver, measure a magnitude of received signal strength, adjust the gain of said amplifier by an amount depending on the magnitude of the measured signal strength, and remove the second order distortion.

Introducing, increasing or adding a second order distortion into the receiver may be done, in one example, by introducing a gain offset between positive and negative constituents of a differential signal of the receiver. A typical receiver includes differential circuitry for handling signals having positive and negative paths) for the purpose of tuning out unwanted second order distortion components. The invention exploits the presence of this circuitry by deliberately introducing second order distortion for a period of time, during which, measurements of total received signal power can be made. For such a receiver incorporated in a UE, the period of time during which the measurements are made may be arranged to coincide with the period during which the UE is making a transmit-to-receive signal transition. The presence of an out-of-band blocker signal during the degradation of the receiver’s performance (resulting from the deliberate introduction of second order distortion) , results in an increase in in-band noise power. The measured total received signal power is then the combined power of the desired signal and out-of-band power. In one embodiment, if the magnitude of this total received power is measured to be at or below the design point of the receiver, then the gain of the amplifier (which, typically, may be a low noise amplifier, LNA) is held as high as possible in order to optimise its noise figure. If the magnitude of the total received power is measured to be above a pre-set threshold then the gain of the amplifier may be gradually lowered which would gracefully reduce receiver sensitivity. In cases where the magnitude of the total received power is measured to be above some pre-set maximum level, reception of signals may be abandoned and rescheduled for a later time. An example of a blocker is a TV transmission which to some extent, may be suppressed by high pass filtering in the low-band receive path.

According to a fourth aspect of the invention, there is provided a method for reducing the effects of emissions of intermodulation components in a transmitter, the method comprising: introducing a frequency shift in a baseband signal and compensating fro said frequency shift by introducing a shift in local oscillator frequency by and equal and opposite amount.

According to a fifth aspect of the invention, there is provided a transmitter comprising a means for introducing a frequency shift in a digital baseband signal, and a controller for adjusting a frequency of a local oscillator signal in order to compensate for said frequency shift.

In one embodiment, the means for introducing the frequency shift in the digital baseband signal comprises a digital rotator which multiplies in-phase and out-of-phase components of the digital baseband signal by the cosine and sine components respectively of a time-varying signal respectively to produce frequency-shifted digital baseband signals, and wherein the transmitter further comprises a digital-to-analog converter for converting the frequency-shifted digital baseband signals to frequency-shifted analogue baseband signals and a mixer for mixing the frequency-shifted analog baseband signals with a programmable local oscillator signal to produce a desired carrier frequency.

Advantageously, a transmitter in accordance with an embodiment of the invention may be configured to cause undesired intermodulation frequency components to fall closer to a band where emission limits are less stringent. Undesirable, intermodulation components typically arise due to nonlinearities in baseband components such as digital-to-analogue-converters, mixers and filters. Consider an example where transmission is to occur in band 13 (782 MHz) and the wide-band Public Safety region is at 763-768 MHz. In certain resource block allocations, the fifth order intermodulation components generated in the baseband transmit chain will fall into the protected band. The transmitter can ameliorate the effects of such intermodulation components by shifting the baseband frequency. To compensate for this shift, the local oscillator frequency is shifted by an equal and opposite amount. The desired signal appears in the correct location because the frequency shifts cancel each other out. However, the undesired frequency components now fall much closer to the band were the emission limits are less stringent. Therefore, no transmit filtering, such as SAW filters, is needed to reject undesired out of band components.

Preferably, these frequency offsets are carried out at higher transmit power where transmit carrier leakage (local oscillator feed through) is low enough not to cause unwanted in band emissions. At lower power, where intermodulation products are much lower, the signal may be transmitted in the conventional manner.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the invention will be described, by way of example, with reference to the following drawings, in which:

Figure 1 is a schematic block diagram of a known transceiver device；

Figure 2 is a schematic block diagram of a transceiver according to an embodiment of the invention.

Figure 3 is a simplified flowchart illustrating a method of adjusting an amplifier gain in a receiver in accordance with an embodiment of the invention； and

Figure 4 is a simplified flowchart illustrating a method of controlling emissions of intermodulation components in a transmitter in accordance with an embodiment of the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Embodiments of the present invention are described below by way of example only. These examples represent the best ways of putting the invention into practice that are currently known to the Applicant although they are not the only ways in which this could be achieved. The description sets forth the functions of the example and the sequence of steps for constructing and operating the example. However, the same or equivalent functions and sequences may be accomplished by different examples.

Referring to figure 2, a transceiver 200 comprises an RF and baseband module 201 and a front-end module 202. The transceiver may be incorporated in a user equipment. A switching module 203 switches an antenna 204 between one of three receive

paths

205, 206, 207 and three transmit

paths

208, 209, 210. Each receive and transmit path 205-210 includes a filter 211-216, respectively, and each filter is chosen to have a particular bandwidth. The frequency range of each of the filters is chosen to allow for worldwide coverage and for filtering out of unwanted harmonics of in-band signals, particularly in the receive path.

In one embodiment the receive path 205 and the transmit path 214 are low band paths and of their

respective filters

211 and 216 have a pass band covering a first plurality of E-UTRAN (Evolved Universal Mobile Telecommunication System Territorial Radio Access Network) bands and having a bandwidth of 694-960 MHz. The receive path 206 and the transmit path 215 are mid-band receive paths and their

respective filters

212 and 215 have a pass band covering a second plurality of E-UTRAN bands and having a bandwidth of 1710-2200 MHz. The receive path 207 and the transmit path 210 are high band paths and their

respective filters

213 and 216 have a pass band covering a third plurality of E-UTRAN bands and having a bandwidth of 2300-2690 MHz. This choice of bandwidths permits worldwide coverage of the transceiver 200 and also ensures that in-band second harmonics and above are filtered out. In a further embodiment a very low band path is added to cover E-UTRAN band 31 and each transmit and receive path may incorporate a filter having a pass band of 380-470 MHz

In an alternative embodiment the filters 211-216 are all lowpass filters. In such an embodiment, the

filters

211 and 214 in the low band pass have a cut-off frequency of 960 MHz, the

filters

212 and 215 in the mid-band paths have a cut-off frequency of 2200 MHz and the

filters

213 and 216 in the high band paths have a cut-off frequency of 2960 MHz. This choice of bandwidths again permits worldwide coverage of the transceiver 200 and also ensures that in-band third harmonics and above are filtered out.

In another embodiment the mid-band path is arranged to have a pass band from 1428-2200 MHz in order to include Japanese LTE bands 11 and 21.

Each transmit

path

208, 209, 210 in the front end module 202 includes an

amplifier

217, 218, 219 respectively. The outputs of each amplifier are connected respectively to one of the

filters

214, 215, 216 and each

amplifier

217, 218, 219 receives a respective input from the RF and baseband module 201.

The filters 211-216 in this example are passive LC filters.

The RF and baseband module 201 includes receive circuitry 220 and transmit circuitry 221. The RF and baseband module 201 also includes a digital signal processor 222 which has a first output 223 which is connected to a module 224. The module 224 schematically represents a differential circuit of a receive mixer module 225 of the receive circuitry 220. The digital signal processor 222 is arranged to provide a control signal at its output 223 for controlling the differential circuit 224 in a manner to be described below. The receive mixer module 225 of the receive circuitry 220 receives inputs from a low noise amplifier module 226 which in turn receives signals from the low-band path 205, the mid-band path 206 and the high band path 207 once the signals have passed through the

respective filters

211, 212 and 213. A second output of the digital signal processor 222 is connected to the low noise amplifier module 226 and is used to adjust the gain of at least one low noise amplifier comprising the module 226 in a manner to be described below. An output of the receive mixer module 225 of the receive circuitry 220 is fed through a low-pass filter 227 and thence through an amplifier 228 and thence through an analogue to digital converter 229 whose output is fed into the digital signal processor 222.

The transmit circuitry 221 includes a transmit mixer module 230 which has a first input connected to a local oscillator 231 and three outputs connected through an amplifier module 232 to the

respective filters

214, 215, 216 of the low band, mid-band and high band paths of the front-end module 202. A third output of the digital signal processor 222 is connected to a local oscillator control module 233. An output of the transmit carrier offsets control module is connected to the local oscillator 231. A fourth output of the digital signal processor 222 is connected to a digital rotator module 234.. An output of the digital rotator module 234 is connected to a digital to analogue converter 235 whose output in turn is connected to a low-pass filter 236 whose output is fed through an amplifier 237 to an input of the transmit mixer module 230.

A method for measuring and compensating for blockers (jamming signals) will now be described. When receiving a communication signal via the antenna 204, as is customary, the RF signal received on one of the receive

paths

205, 206, 207 is converted to baseband (by mixing with the signal from the local oscillator 231) in the receive mixer module 225, filtered and converted to digital signals by the analogue to digital converter 229 for reception and processing by the digital signal processor 222. Each time the transceiver switches between transmit and receive modes the second order distortion toleration of the receive circuitry 220 is deliberately degraded. During this switching between modes, the digital signal processor 222 generates a signal on line 223 which causes an offset between mixer components, illustrated schematically by the introduction of an offset into the differential circuit 224. For example, a gain offset between positive and negative constituent components of a differential signal in the circuit 224 is introduced under the control of the digital signal processor 222. This offset creates intermodulation components due to out of band blockers that fall in-band and can be measured in the digital signal processor 222 during this period between switching from transmit to receive mode. Once the measurement is done, the offset is removed. Depending on the results of the measurements the digital signal processor 222 adjusts the gain in the receive circuitry amplifier block 226. During the next period of switching between transmit and receive modes, the digital signal processor 222 again degrades the second order distortion tolerance, measures total received power, then removes the offset and again adjusts the amplifier gain, if appropriate, for optimising performance. This process can be repeated during each period of switching between transmit and receive modes. The measured total received power can be compared with a design criterion of the receiver and the gain adjusted in order to optimise performance. One way of introducing the necessary offset is by shifting the biasing point of a mixer clock signal. The average DC level or total average in band noise created due to the offset is a direct measure of out of band noise entering the receiver. Typically, the dominant intermodulation product is a DC term which can be averaged by the digital signal processor 222. It will be understood that any RF filtering related to tuned circuitry is automatically accounted for. It will also be noted that a far-out-of-band blocker will have a comparatively lower amplitude at the mixer plane than a close-n interferer and will therefore yield lower reading. It will also be appreciated that other known methods for increasing or adding a second order distortion are applicable.

The gain of the receiver amplifiers 226 can be controlled, by way of the signal from the digital signal processor 222, by switching in dummy loads and changing current biasing. As measured blocker power increases, the gain of the amplifiers 226 can be gradually lowered. If measured blocker power is low, gain can be increased.

A method for reducing the effects of emissions of intermodulation components in the transmitter circuit 221 will now be described. When transmitting a modulated signal from the antenna 204 digital I/O and Q signals from the digital signal processor 222 are converted to baseband analogue signals (at a pre-set frequency) by the digital to analog converter 235 and up-converted to the desired carrier frequency by mixing with a signal from the local oscillator. Conventionally, a baseband signal is up-converted using a fixed local oscillator signal that is generated in a phase locked loop module of the transmitter. In the figure 2, the in phase (I signal) output by the amplifier 237 can be multiplied by the cosine of the local oscillator signal and the quadrature signal (Q signal output by the amplifier 237 can be multiplied by the sine of the local oscillator signal. Quadrature generation can be done typically using divide by two circuits or polyphase filters. Additionally, in the transceiver of figure 2 a frequency shift is added in the digital domain by the digital rotator block 234 which performs a frequency conversion on the digital signal samples. The digital to analog converter 235, filter 236 and amplifier 237 which form the analog components of the baseband circuit in the transmit circuit 221 typically produce unwanted intermodulation frequency components because they do not necessarily function in a perfectly linear fashion. These intermodulation components may fall within a protected band and so it would be advantageous if their emissions could be prevented. The transceiver of figure 2 enables the shifting of these unwanted and intermodulation frequency components into a band which is removed from the protected band. This is done by shifting the frequency of the baseband signal so that the intermodulation frequency components generated by the nonlinear baseband components fall outside the protected band. To ensure that the transmitted carrier frequency is as it should be, a compensating frequency offset of equal magnitude yet opposite to the shift imposed on the baseband signal is applied to the local oscillator 231 under the control of the control module 233 in response to signal from the digital signal processor 222.

An appropriate baseband frequency shift is performed in the digital domain by the digital rotator module 234. The technique of digital rotation is known and essentially comprises multiplying the cosine and sine components of a time varying signal with the I and Q digital samples, respectively, which are output from the digital signal processor 222. Hence the I and Q digital samples arrive at the digital to analogue converter 235 at a translated frequency. The digital signal processor 222 applies a control signal to the control module 233 such that the control module 233 programmes the local oscillator 231 to adjust its frequency output in order to compensate for the frequency offset introduced into the baseband signal. Hence the output of the transmit mixer module 230 is at the correct carrier frequency.

Depending on the choice of frequency shift of the baseband signal (and corresponding opposing shift in the local oscillator frequency) the non-linear components comprising the digital to analogue converter 235, filter 236 and the mixers the mixer module 230 create frequency products (intermodulation components) at multiples of the baseband frequency that are away from the desired transmission band. It will be appreciated that the widths of the higher-order intermodulation products scale with the modulation order. In order to meet spectral emission requirements conventional transmitters employ a post-power amplifier filter in order to suppress the unwanted frequency components (for example components falling into the public safety bands) . However, by using the transceiver of figure 2, introducing the frequency shifts as described above can make unwanted modulation products fall much closer to the channel where they can be tolerated. It is possible to choose a frequency shift such that all intermodulation products lie close to the region of the desired transmit carrier frequency. This is not generally a problem as long as their power level is sufficiently low

Referring now to the simplified flowchart of figure 3, a method for optimising low noise amplifier gain in a receiver of a wireless communication device is described. At 301, the digital signal processor 222 determines whether the transceiver is switching between transmit and receive modes and if so, at 302 generates a signal on line 223 in order to introduce an offset in differential signal paths 224 so that tolerance to second order distortion components in the receiver mixer circuit 225 is degraded. At 303, the digital signal processor 222 measures total received signal strength via the receive circuitry 220. At 304, the measured value is compared with a pre-set value such as the receiver’s design point or the received signal strength last measured without offset applied (that is, during normal operation) . The difference in power levels measured before and after applying the offset is a measure of out-of-band interferer power. The power level without offset applied will be known as this is measured during normal operation. At 305 the digital signal processor 222 generates a signal for adjusting the gain of the receiver amplifiers in module 226. If the measured difference is at or below the design point, then the gain is increased. Otherwise the gain is decreased. At 306, the offset which was applied at 302 is removed. All steps from 302 to 306 inclusive performed during the period in which the transceiver is switching from a transmit mode to a receive mode. It will be understood that the order of the steps may be changed as long as the desired outcome of the procedure is not affected. For example, step 306 can be performed before

steps

304 or 305.

Referring now to the simplified flowchart of figure 4, a method for reducing the effects of emissions of intermodulation components in a transmitter is described. At 401, a frequency shift in the digital baseband signals being output by the digital signal processor 222 is introduced by the action of the digital rotator module 234. At 402, the resulting frequency-shifted digital baseband signals are converted to frequency-shifted analogue signals by the baseband components； digital-to-analogue converter 235, filter 236 and amplifier 237. At 403 the frequency of the local oscillator 231 is adjusted by the control module 233 in order to compensate for the frequency shift introduced in step 402. At 404, the adjusted local oscillator signal is mixed with the frequency-shifted baseband signals in an up-conversion process to produce a carrier frequency signal at the desired frequency.

Those skilled in the art will recognise that the boundaries between functional modules are merely illustrative and that alternative embodiments may merge functional modules or circuit elements or impose an alternate decomposition of functionality upon various functional modules or circuit elements. Thus, it is to be understood that the architectures depicted herein are merely exemplary, and that in fact many other architectures can be implemented which achieve the same functionality.

It will be understood that the benefits and advantages described above may relate to one embodiment or may relate to several embodiments. The embodiments are not limited to those that solve any or all of the stated problems or those that have any or all of the stated benefits and advantages.

Any reference to 'an' item refers to one or more of those items. The term 'comprising' is used herein to mean including the method blocks or elements identified, but that such blocks or elements do not comprise an exclusive list and a method or apparatus may contain additional blocks or elements.

The steps of the methods described herein may be carried out in any suitable order, or simultaneously where appropriate. Aspects of any of the examples described above may be combined with aspects of any of the other examples described to form further examples without losing the effect sought.

It will be understood that the above description of a preferred embodiment is given by way of example only and that various modifications may be made by those skilled in the art. Although various embodiments have been described above with a certain degree of particularity, or with reference to one or more individual embodiments, those skilled in the art could make numerous alterations to the disclosed embodiments without departing from the scope of this invention.

Claims

A transceiver having a front end module including N transmit paths and N receive paths, where N is an integer, each of said transmit and receive paths including a filter, wherein a pass-band of each filter is chosen such that harmonics of in-band signals are filtered out and wherein N is chosen such that a total pass-band of the transceiver extends across a plurality of radio access networks.
The transceiver of claim 1 wherein at least one filter is a band-pass filter.
The transceiver of claim 1 or claim 2 wherein at least one filter is a low-pass filter.
The transceiver of any preceding claim wherein the filters are LC filters
A receiver comprising a variable gain amplifier and a processor, wherein the processor is adapted to introduce a second order distortion into the receiver, measure a magnitude of received signal strength, adjust the gain of said amplifier by an amount depending on the magnitude of the measured signal strength, and remove the second order distortion.
A method for optimising the gain of an amplifier in a receiver, the method comprising: introducing a second order distortion into the receiver； measuring a magnitude of received signal strength； adjusting the gain of said amplifier by an amount depending on the magnitude of the measured signal strength； and removing the second order distortion.
The method of claim 6 comprising； measuring a magnitude of received signal strength prior to introducing the second order distortion, comparing the magnitudes of measured received signal strengths without and with the second order distortion being introduced, and adjusting the gain of said amplifier depending on the result of the comparison.
The method of claim 6 or 7 wherein the second order distortion is introduced by offsetting a gain between positive and negative signal components of a differential signal in the receiver.
The method of claim, 7 or 8 wherein the gain is increased if the magnitude of the difference between measured received signal strengths is below or equal to a pre-set threshold.
The method of claim 7 or claim 8 wherein the gain is decreased if the magnitude of the difference between measured received signal strengths is above a pre-set threshold.
The method of any of claims 6 to 10, wherein the receiver is included in a transceiver of a wireless communication device and said offsetting the gain is initiated during a period when the transceiver switches between transmit and receive modes.
A transmitter comprising a means for introducing a frequency shift in a digital baseband signal, and a controller for adjusting a frequency of a local oscillator signal in order to compensate for said frequency shift.
The transmitter of claim 12 wherein the means for introducing a frequency shift is a digital rotator which multiplies in-phase and out-of-phase components of the digital baseband signal by the cosine and sine components respectively of a time-varying signal respectively to produce frequency-shifted digital baseband signals, and wherein the transmitter further comprises a digital-to-analogue converter for converting the frequency-shifted digital baseband signals to frequency-shifted analogue baseband signals and a mixer for mixing the frequency-shifted analogue baseband signals with a programmable local oscillator signal to produce a desired carrier frequency.
A method for reducing the effects of emissions of intermodulation components in a transmitter, the method comprising: introducing a frequency shift in a digital baseband signal and compensating for said frequency shift by introducing a shift in local oscillator frequency by and equal and opposite amount.
The method of claim 14 comprising multiplying in-phase and out-of-phase components of the digital baseband signal by the cosine and sine components respectively of a time-varying signal respectively to produce frequency-shifted digital baseband signals, converting the frequency-shifted digital baseband signals to frequency-shifted analogue baseband signals, and mixing the frequency-shifted analogue baseband signals with a programmable local oscillator signal to produce a desired carrier signal.