WO2016197823A1 - 圆极化介质谐振天线及其参数确定方法和通信设备 - Google Patents

圆极化介质谐振天线及其参数确定方法和通信设备 Download PDF

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Publication number
WO2016197823A1
WO2016197823A1 PCT/CN2016/083349 CN2016083349W WO2016197823A1 WO 2016197823 A1 WO2016197823 A1 WO 2016197823A1 CN 2016083349 W CN2016083349 W CN 2016083349W WO 2016197823 A1 WO2016197823 A1 WO 2016197823A1
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Prior art keywords
frequency band
dielectric resonator
dielectric
slit
circularly polarized
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PCT/CN2016/083349
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English (en)
French (fr)
Inventor
王小明
张铮
焦磊
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中兴通讯股份有限公司
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Publication of WO2016197823A1 publication Critical patent/WO2016197823A1/zh

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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/36—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/36—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith
    • H01Q1/38—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith formed by a conductive layer on an insulating support
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/48—Earthing means; Earth screens; Counterpoises
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/50—Structural association of antennas with earthing switches, lead-in devices or lightning protectors
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q13/00—Waveguide horns or mouths; Slot antennas; Leaky-waveguide antennas; Equivalent structures causing radiation along the transmission path of a guided wave
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/10—Resonant antennas
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/30—Arrangements for providing operation on different wavebands
    • H01Q5/307—Individual or coupled radiating elements, each element being fed in an unspecified way

Definitions

  • the present application relates to, but is not limited to, the field of communications, and in particular, to a circularly polarized dielectric resonant antenna, a parameter determining method thereof, and a communication device.
  • the working frequency band of the communication equipment is mainly concentrated below 3 GHz, which makes the spectrum resources very crowded, and the high frequency bands (such as the millimeter wave and the centimeter wave band) higher than 3 GHz are available with abundant spectrum resources, which can effectively alleviate the spectrum.
  • the current situation of resource shortage can realize high-speed short-distance communication and support the requirements of the capacity and transmission rate of the fifth-generation mobile communication system (5G).
  • the conductor loss of the metal antenna is severe, resulting in a low radiation efficiency of the antenna. Since there is no conductor loss and surface wave loss, the dielectric resonant antenna has a high radiation efficiency and can reach more than 95% in the millimeter wave band.
  • Dielectric Resonator Antenna usually has a narrow working bandwidth, especially a single-feed circular polarized dielectric resonant antenna. Bandwidth improvement is a major research direction of dielectric resonant antennas.
  • This paper provides a circularly polarized dielectric resonant antenna and its parameter determining method and communication device to at least partially solve the problem of small bandwidth of the circularly polarized dielectric resonant antenna.
  • the embodiment of the invention provides a circularly polarized dielectric resonant antenna, and the circularly polarized dielectric resonant antenna includes:
  • a dielectric substrate comprising a first surface and a second surface on the back surface of the first surface; the first surface is provided with a metal floor;
  • a dielectric resonator in contact with the metal floor, configured to: radiate a wireless signal and operate at First frequency band;
  • a slit located on the metal floor, configured to: radiate a wireless signal, and operate in a second frequency band; wherein the first frequency band and the second frequency band are at least partially different, and together comprise a third frequency band;
  • the three frequency bands are continuous frequency bands; the return loss of the circularly polarized dielectric resonant antenna when operating in the third frequency band is greater than a specified value;
  • a feed line located on the second surface, coupled to the dielectric resonator and the gap, respectively, is configured to cause circular polarization radiation of the dielectric resonator and the slit, respectively.
  • the dielectric resonator is a resonator having a degenerate mode of operation.
  • the dielectric resonator is in the form of a cylinder or a rectangular parallelepiped.
  • the slit is an annular slit.
  • the annular gap is an axisymmetric ring gap.
  • the antenna further includes: a feeding port and a matching network
  • the feed port is connected to the feeder through the matching network
  • the matching network is configured to perform impedance matching on the circularly polarized dielectric resonant antenna.
  • the feed line comprises an L-shaped microstrip line.
  • a geometric center of a side of the dielectric resonator in contact with the metal floor overlaps a geometric center of the gap.
  • the embodiment of the invention further provides a communication device, comprising: a circularly polarized dielectric resonant antenna as described above.
  • the embodiment of the invention further provides a method for determining a parameter of a circularly polarized dielectric resonant antenna, the method comprising:
  • Adjusting physical parameters of at least one of the dielectric resonator and the slit causing the dielectric resonator to operate in a first frequency band and causing the slit to radiate a wireless signal and operate in a second frequency band; wherein the first frequency band and the second frequency band
  • the frequency bands collectively comprise a third frequency band, wherein the third frequency band is a continuous frequency band; and the return loss of the circularly polarized dielectric resonant antenna when operating in the third frequency band is greater than a specified value;
  • the physical parameters of the feeder are adjusted to cause circular polarization radiation of the dielectric resonator and the slit, respectively.
  • the gap is an annular gap
  • the adjusting a physical parameter of at least one of the dielectric resonator and the slit, causing the dielectric resonator to operate in the first frequency band and causing the slit to radiate a wireless signal and operate in the second frequency band includes:
  • the first resonant frequency is a resonant frequency band of the first frequency band
  • the adjusting the physical parameters of the feed line, respectively, the circularly polarizing radiation of the dielectric resonator and the slit comprises: adjusting physical parameters of the L microstrip line, respectively, causing the dielectric resonator and the gap respectively Perform circularly polarized radiation.
  • the embodiment of the invention further provides a computer readable storage medium storing computer executable instructions, which are implemented to implement a parameter determination method of the circularly polarized dielectric resonant antenna.
  • the circularly polarized dielectric resonant antenna and the parameter determining method and the communication device provided by the embodiments of the present invention simultaneously use the dielectric resonator and the slit as a radiator to perform signal radiation, and corresponding to the first frequency band and the gap corresponding to the dielectric resonator
  • the second frequency band is at least partially different, and the first frequency band and the second frequency band together form a continuous frequency band in which the return loss is less than a specified value, that is, the third frequency band.
  • the bandwidth corresponding to the third frequency band is large, and the circularly polarized dielectric resonant antenna thus formed has the characteristics of simple structure and simple fabrication.
  • FIG. 3 are schematic structural diagrams of a circularly polarized dielectric resonant antenna according to an embodiment of the present invention
  • FIG. 4 is a frequency response curve of return loss of a circularly polarized dielectric resonant antenna according to an embodiment of the present invention. Simulation and actual measurement diagram;
  • FIG. 5 is a schematic diagram of simulation and actual measurement of an axial ratio frequency response curve of a circularly polarized dielectric resonant antenna according to an embodiment of the present invention
  • FIG. 6 is a schematic diagram of simulation and actual measurement of radiation directions of circularly polarized dielectric resonant antennas at three different frequency points according to an embodiment of the present invention
  • FIG. 7 is a schematic flowchart of determining a parameter of a circularly polarized dielectric resonant antenna according to an embodiment of the present disclosure
  • FIG. 8 is a schematic flowchart of adjusting physical parameters of a dielectric resonator and a slot according to an embodiment of the present disclosure
  • FIG. 9 is a schematic diagram of a parameter determining process of another circularly polarized dielectric resonant antenna according to an embodiment of the present invention.
  • FIG. 10 is a schematic diagram showing the relationship between the inner diameter R of a circular slot and the axial ratio of a circularly polarized antenna in a frequency band according to an embodiment of the present invention.
  • the embodiment provides a circularly polarized dielectric resonant antenna
  • the circularly polarized dielectric resonant antenna includes:
  • the dielectric substrate 1 includes a first surface and a second surface on the back surface of the first surface; the first surface is provided with a metal floor 2;
  • a dielectric resonator 3 in contact with the metal floor 2, configured to: radiate a wireless signal and operate in a first frequency band;
  • a slot 4 located on the metal floor 2, configured to: radiate a wireless signal and operate in a second frequency band; wherein the first frequency band and the second frequency band together comprise a third frequency band; a continuous frequency band; the return loss of the circularly polarized dielectric resonant antenna operating in the third frequency band is greater than a specified value;
  • the slit 4 may be a slit of various shapes such as a strip slit, in this embodiment
  • the ground selection may be an annular slit as shown in FIGS. 1 and 2;
  • the feed line 5, located on the second surface, is coupled to the dielectric resonator 3 and the slit 4, respectively, and is configured to cause circular polarization radiation of the dielectric resonator 3 and the slit 4, respectively.
  • the dielectric substrate 1 has a plate shape.
  • the dielectric substrate 1 may be a rectangular plate or a square plate.
  • the two faces having a larger surface area are generally the first surface and the second surface described in the embodiment.
  • the first surface and the second surface may be positive and negative sides of each other.
  • a metal floor 2 is provided on the first surface of the dielectric substrate 1.
  • the metal floor 2 can be composed of a metal layer attached or plated on the dielectric substrate 1 and can communicate with a ground point in the antenna or device, so it is also called a ground plate.
  • the side of the metal floor 2 that is in contact with the second surface may be equal in area to the second surface and aligned with each other.
  • the dielectric substrate 1 herein may be composed of a medium having a dielectric constant of a predetermined value.
  • the dielectric constant of the medium constituting the dielectric substrate 1 of the present embodiment can be referred to the dielectric constant of the medium of the dielectric substrate in the related art, and will not be described in detail herein.
  • the dielectric resonator 3 is a radiator capable of radiating a wireless signal, and is usually composed of a medium having a relatively high dielectric constant.
  • the dielectric resonator has a relative dielectric constant of 6 to 140, and the dielectric constant of the dielectric resonator 3 in the present embodiment may be 10.2, but is not limited to 10.2 or 6 to 140.
  • a slit exists as a coupling feed member in a circularly polarized dielectric resonator antenna, and is disposed to feed the dielectric resonator 3.
  • the slit is set to perform radiation of a wireless signal.
  • the slit provided as the radiation signal in the embodiment may be a slit on the metal floor 2. A metal portion of the metal floor 2 may also remain inside the slit 4.
  • the feed line 5 may be a variety of feed structures that can be electromagnetically coupled to the dielectric resonator 3 and the slit 4, respectively, for feeding.
  • the feeder 5 can adopt a single feed line, and transmits signals to the dielectric resonator 3 and the slit 4 by electromagnetic coupling in one feeding region to realize feeding.
  • the feed line 5 can also realize the circularly polarized radiation of the dielectric resonator 3 and the slit 4 by adjusting the physical parameters of the feed line 5, thereby realizing the circularly polarized dielectric resonant antenna provided by the embodiment. Circularly polarized radiation of the antenna.
  • the dielectric resonator 3 and the slit 4 are simultaneously used as radiation radiators for radiating wireless signals to perform wireless signal radiation. Further, by adjusting the physical parameters of the dielectric resonator 3 and the slit 4, the resonance frequencies of the dielectric resonator 3 and the slit 4 can be made close, thereby achieving the approach of the dielectric resonator 3 and the operating band of the slit 4.
  • the operating band of the dielectric resonator is the first frequency band
  • the operating frequency band of the slot 4 is the second frequency band.
  • the first frequency band and the second frequency band may be at least partially different in this embodiment.
  • the first frequency band and the second frequency band are at least partially different and can be divided into two cases, one case where the first frequency band and the second frequency band partially overlap; in the second case, the first frequency band and the second frequency band are adjacent.
  • the first frequency band and the second frequency band jointly form a third frequency band
  • the third frequency band is an operating frequency band of the circularly polarized dielectric resonant antenna described in the embodiment, and the circularly polarized dielectric resonant antenna operates in the first
  • the return loss at the time of the three-band is large, and is required to be larger than the specified value in this embodiment.
  • the specified value may be 10 dB.
  • the slot 4 is used to participate in the radiation of the wireless signal, so that the working bandwidth of the circularly polarized dielectric resonant antenna is the third frequency band, and the third frequency band is obviously larger than the first frequency band, so the circularly polarized dielectric resonant antenna is improved.
  • the working bandwidth solves the problem that the working bandwidth of the circularly polarized dielectric resonant antenna in the related art is narrow.
  • Table 1 is an alignment of the axial ratio relative bandwidths of the circularly polarized dielectric resonator antenna of the present embodiment and the circularly polarized dielectric resonator antenna of the related art.
  • the dielectric resonator 3 is a resonator having a degenerate operation mode. And generally, when a resonator having a degenerate mode of operation is applied to a circularly polarized dielectric resonator antenna according to an embodiment of the present invention, it generally operates in a degenerate mode of operation.
  • the dielectric resonator 3 operating in the degenerate mode of operation generally has two mutually orthogonal polarization modes. The dielectric resonator 3 will also operate in these two orthogonal polarization modes.
  • the dielectric resonator 3 may be in the form of a cylinder or a rectangular parallelepiped. In the embodiment, the dielectric resonator 3 may alternatively be a cylinder or a rectangular parallelepiped to simplify fabrication and Control of accuracy.
  • the slit 4 can optionally be an annular slit, for example an axially symmetric ring gap. In the present embodiment, it has been proved that the slit 4 has an excellent axial radiation effect. In this embodiment the axis of symmetry of the slot 4 will generally pass through the geometric center of the slot 4.
  • the slit 4 may be an annular gap having an axis symmetry such as a circular ring slit, a rectangular ring slit, an elliptical ring slit, or a hexagonal ring slit.
  • the annular slot may also be a central symmetric slot, and the center of symmetry is the geometric center of the slot 4.
  • the centrally symmetrical slit 4 here may comprise a circular slit, the center of symmetry being the toroid of the annular gap.
  • the centrally symmetrical slit 4 may further comprise a square ring slit, the center of symmetry being the center of the square in which the square ring gap is located.
  • the vertical projection of the dielectric resonator 3 on the metal floor 2 is generally covered by the gap 4.
  • the bottom surface of the dielectric resonator 3 in contact with the metal floor 2 completely covers the gap 4, that is, the area of the bottom surface of the dielectric resonator 3 in contact with the metal floor 2 is not less than that surrounded by the gap 4. area.
  • the antenna further includes: a feed port 7 and a matching network 6; the feed port 7 is connected to the feed line 5 through the matching network 6;
  • the matching network 6 is configured to perform impedance matching on the circularly polarized dielectric resonant antenna.
  • the feed port 7 is connected to a feed network in the device and is configured to receive an electrical signal supplied by the feed network.
  • the matching network 6 may include various structures capable of achieving impedance matching, such as an impedance transformation line or a lumped element network.
  • the lumped elements can include components such as capacitors or inductors.
  • impedance matching can be achieved by changing the line width of the impedance conversion line, changing the electrical characteristic parameters of the components in the lumped element network, and adjusting the impedance of the matching network.
  • Adjusting the electrical characteristic parameters of the components in the lumped component network may include adjusting one or more of the capacitance value of the capacitor, adjusting the inductance value of the inductor, and adjusting the resistance value of the resistor to adjust the electrical characteristic parameter.
  • the feed line may alternatively be an L-shaped feed line.
  • the L-shaped feeder can be divided into two parts, and the angle between the two parts can be 90°.
  • it may be an L-shaped microstrip line in this embodiment.
  • the microstrip line is a microwave transmission line composed of a single conductor that can be mounted on a dielectric substrate, and has the characteristics of low transmission impedance and good signal transmission efficiency; and directly adopts an L-shaped microstrip line for feeding, and has a simple structure and cost. Low and simple and efficient.
  • the geometry of the side of the dielectric resonator 3 in contact with the metal floor 2 The center overlaps the geometric center of the slit 4.
  • the dielectric resonator 3 described in Figs. 1 and 2 has a cylindrical shape.
  • the slit 4 is a circular slit 4.
  • the center of the circular bottom surface of the cylindrical dielectric resonator 3 in contact with the metal floor 2 coincides with the center of the ring gap.
  • Antenna polarization is a parameter that describes the spatial orientation of the radiated electromagnetic wave vector. Since the electric field has a constant relationship with the magnetic field, the spatial direction of the electric field vector is generally used as the polarization direction of the electromagnetic wave radiated by the antenna. In the embodiment, the polarization mode may be a manner of antenna polarization.
  • the phase difference of the dielectric resonator 3 and the slit 4 operating at the resonance frequencies of the respective polarization modes is maintained at about 90 degrees, so that circularly polarized radiation can be realized.
  • the preset phase value is usually a small value, such as plus or minus 3 degrees, plus or minus 2 degrees, and the like.
  • the phase difference of the dielectric resonator 3 and the slit 4 when operating at the resonance frequency of the respective polarization modes may be a positive phase value greater than 0 degrees, or may be a negative phase value less than 0 degrees.
  • the absolute value of the phase difference is about 90 degrees.
  • FIG. 3 is a schematic diagram showing parameters of a plurality of components in a dielectric resonant antenna according to an embodiment of the present invention.
  • the slot width of the annular slot is indicated by W slot ;
  • the R is the radius of the inner ring of the annular slot;
  • the L s is the length of the open end of the L-shaped microstrip line;
  • the L 1 is L-shaped The length of the connection end of the microstrip line and the impedance conversion line;
  • the L t is the length of the impedance transformation line;
  • the W t is the width of the impedance transformation line.
  • the x-axis, the y-axis, and the z-axis together form a spatial rectangular coordinate system.
  • FIG. 4 to FIG. 6 are simulation and actual measurement diagrams of the circularly polarized dielectric resonant antenna provided by the embodiment.
  • the measured 10dB impedance bandwidth of the antenna covers 5.85GHz-7.20GHz and the relative bandwidth is 20.7%.
  • FIG. 5 is a simulation and actual measurement diagram of an axial ratio frequency response curve according to an embodiment of the present invention.
  • the measured 3dB impedance bandwidth of the antenna covers 6.22GHz-7.20GHz and the relative bandwidth is 14.6%, which is included in the 10dB impedance bandwidth.
  • the axial ratio of the antenna is defined as: the end point trajectory of the instantaneous electric field vector of the arbitrary polarized wave is an ellipse, and the ratio of the long axis 2A and the short axis 2B of the ellipse is called an axial ratio AR (Axial Ratio).
  • the axial ratio is one of the circularly polarized antennas
  • An important performance indicator which represents the purity of circular polarization, with an axial ratio of no more than 3 dB, defined as the circular polarization bandwidth of the antenna. It is an important indicator to measure the difference in signal gain between antennas in different directions.
  • FIG. 6 is a simulation diagram and a measured diagram of radiation patterns of three frequency points in the working bandwidth according to an embodiment of the present invention.
  • FIG. 6 is a diagram showing the simulation and measurement of the radiation direction of the circularly polarized dielectric resonator antenna operating at 6.22 GHz according to an embodiment of the present invention.
  • FIG. 6b is a simulation and actual measurement diagram of the radiation direction of the circularly polarized dielectric resonant antenna operating at 6.7 GHz according to an embodiment of the present invention.
  • FIG. 6 is a simulation and measurement diagram of the radiation direction of the circularly polarized dielectric resonator antenna operating at 7.1 GHz according to an embodiment of the present invention.
  • the feed line is an L-shaped microstrip line bent 90° to the right
  • a left-hand circular polarization characteristic will be formed.
  • the feeder is an L-shaped microstrip line bent 90° to the left
  • a right-hand circular polarization characteristic is formed.
  • the bending direction of the L-shaped microstrip line can be selected according to the polarization required for the actual work.
  • An embodiment of the present invention provides a communication device, where the communication terminal includes at least one of a circularly polarized dielectric resonant antenna as described above.
  • the communication device herein can be a variety of mobile communication devices, such as mobile phones, tablets, or various in-vehicle mobile communication devices.
  • the communication device provided in this embodiment may include a feed network and a communication chip, and the feed network is connected to the feed port, and the operation of the circularly polarized dielectric resonant antenna is controlled according to the communication chip. For example, by controlling the electrical signal transmitted by the feed network to the feeder, the radiation of the wireless signal of the dielectric resonator and the ring feeder is controlled.
  • the communication device in this embodiment adopts the circularly polarized dielectric resonant antenna described above, and has the characteristics of simple antenna structure, low cost, large working bandwidth and good communication quality.
  • the embodiment provides a parameter determining method for a circularly polarized dielectric resonant antenna, and the method includes:
  • Step S110 adjusting physical parameters of at least one of the dielectric resonator and the slit, causing the dielectric resonator to operate in a first frequency band and causing the slit to radiate a wireless signal and operate in a second frequency band; wherein the first frequency band And the second frequency band is at least partially different, and together form a third frequency band, wherein the third frequency band is a continuous frequency band; and the return loss of the circularly polarized dielectric resonant antenna when operating in the third frequency band is greater than a specified value ;
  • Step S120 adjusting physical parameters of the feeder, so that the dielectric resonator and the gap are separately Row circularly polarized radiation.
  • the physical parameters of at least one of the dielectric resonator and the slit are adjusted such that both the dielectric resonator and the annular slit participate in the radiation of the wireless signal.
  • the working frequency band of the dielectric resonator and the slot are at least partially different, so that the working bandwidth of the antenna is widened by the working frequency band of the slot; thus the circularly polarized resonant antenna obtained by determining the parameter in this way has the characteristics of high working bandwidth.
  • the step S110 in the embodiment may include: fixing physical parameters of the dielectric resonator, adjusting physical parameters of the slot; fixing physical parameters of the slot, adjusting physical parameters of the dielectric resonator; and may also adjusting the dielectric resonator at the same time And the physical parameters of the gap.
  • the physical parameters of the dielectric resonator herein may include: a dimensional parameter of the dielectric resonator, a shape parameter, and a dielectric constant.
  • the dimensional parameters herein may include the values of the length, width, and height of the dielectric resonator.
  • the shape parameter may include changing the shape of the dielectric resonator, for example, changing the dielectric resonator of the cylinder to a dielectric resonator of a rectangular body.
  • the dielectric parameter may include a dielectric constant or the like of a medium of the dielectric resonator.
  • the physical parameters of the feeder are adjusted. Adjusting the physical parameters of the feeder here may include adjusting the size parameter of the feeder, the feeding parameter and the material parameter, the shape parameter, and the like. For example, adjusting the line length of the L-shaped feeder can be one of adjusting the size parameters of the feeder. Adjusting the feed parameters may include adjusting parameters such as the feed position and the size of the feed area. Adjusting the material parameters can include adjusting the constituent materials of the feeder.
  • the dielectric resonator and the annular slit are respectively polarized in two orthogonal directions, and circularly polarized radiation can be easily realized to form a circularly polarized antenna.
  • the parameter determining method of the circularly polarized dielectric resonant antenna described in this embodiment can be applied to an automatic design scene of a circularly polarized dielectric resonant antenna in various electronic devices having information processing functions. By performing the above method, the electronic device can design a circularly polarized dielectric resonant antenna that satisfies a large working bandwidth.
  • the slit of the circularly polarized dielectric resonant antenna is an annular slit.
  • the step S110 may include:
  • Step S111 determining physical parameters of the dielectric resonator and the first resonant frequency
  • Step S112 estimating, according to the first resonant frequency, a second resonance estimation frequency of the annular slot;
  • Step S113 determining an estimation parameter of the annular slot based on the second resonance estimation frequency
  • Step S114 Simulating the circularly polarized dielectric resonant antenna based on the physical parameters of the dielectric resonator and the estimated parameters, and obtaining a simulation result;
  • Step S115 Determine physical parameters of the annular slot according to the simulation result.
  • the determining the dielectric resonator in the embodiment may include: an electronic device performing parameter determination, receiving physical parameters of the dielectric resonator from a human-machine interaction interface or from other devices.
  • the second resonance estimation frequency is then estimated from the physical parameters of the dielectric resonator.
  • the dielectric waveguide model can be approximated to calculate the resonant frequency of the dielectric resonator, which is referred to as the first resonant frequency in this embodiment.
  • the second resonance frequency can be obtained by calculating the first resonance frequency band and the estimation constant according to a preset function relationship according to an estimation constant obtained by an empirical value or the like.
  • the second resonance estimation frequency can be obtained by adding the first resonance frequency to the estimation parameter or subtracting the estimation parameter.
  • the physical parameter of the annular gap is reversed by using the second resonance estimation frequency, and the physical parameter obtained by the inverse is the aforementioned estimation parameter. For example, taking the ring gap as an example, the following formula can be used to derive the estimated parameters:
  • the f slot is the second resonance estimation frequency
  • c is the speed of light in the vacuum
  • R is the inner diameter of the ring gap
  • W slot is the width of the ring gap
  • ⁇ eff is calculated by:
  • ⁇ d is the relative dielectric constant of the dielectric resonator
  • ⁇ r is the relative dielectric constant of the dielectric substrate 1.
  • step S114 simulation results are obtained according to the physical parameters of the dielectric resonator and the estimation parameters. If the simulation result indicates that the preset condition is met, the estimated parameter can be directly determined as the physical parameter of the annular gap. If the simulation result does not satisfy the preset condition, the simulation may not be considered, and the estimation parameter of the annular gap will be adjusted according to the simulation result, and the process returns to step 114 again until the simulation passes.
  • the simulation here satisfies the preset condition, which may include: in the simulated return loss frequency response simulation curve, there are two valleys of return loss in a continuous frequency band in which the return loss is less than the specified value.
  • the preset conditions herein may not be limited to the above conditions.
  • the method may further include:
  • Step S116 After the simulation is passed, the method further includes: fabricating a circularly polarized dielectric resonant antenna according to physical parameters and estimated parameters of the dielectric resonator;
  • Step S117 Performing actual measurement on the circularly polarized dielectric resonant antenna to form a measured result; and determining the final physical parameter of the annular slit according to the measured result.
  • the measured result indicates that the preset condition is met
  • the corresponding estimated parameter can be directly used as the final physical parameter of the annular slot.
  • the estimated parameter can be adjusted according to the measured result, and the process returns to step S114 or Step S115, until the actual measurement is passed.
  • the present example provides a broadband circularly polarized dielectric resonant antenna including a dielectric substrate 1, a metal floor 2, a dielectric resonator 3, a ring gap 4, and an L-shaped microstrip line 5.
  • the metal floor 2, the ring gap 4, and the dielectric resonator 3 are located on the same side of the dielectric substrate 1, and the L-shaped microstrip line 5 and the impedance conversion line 6 are located on the other side of the dielectric substrate 1.
  • the dielectric resonator 3 may be a cylindrical dielectric resonator formed of a dielectric material, which generally has a relatively high relative dielectric constant (6-140).
  • the relative dielectric constant in this embodiment was 10.2.
  • the dielectric resonator 3 is fixed above the metal floor 2, and the center of the lower surface thereof coincides with the center of the slit 4 on the metal floor 2.
  • the operating frequency of the dielectric resonator 3 radiation mode can be approximated by a Dielectric Waveguide Model (DWM).
  • DWM Dielectric Waveguide Model
  • the HEM 11 ⁇ mode is selected as the operating mode of the cylindrical dielectric resonator 3, and its resonant frequency can be approximated by:
  • ⁇ d is the relative dielectric constant of the dielectric resonator
  • a is the radius of the cylindrical dielectric resonator
  • x a / (2d)
  • d is the height of the cylindrical dielectric resonator
  • the resonant frequency of the ring gap 4 in this example can be approximated by:
  • ⁇ d is the relative dielectric constant of the dielectric resonator 3 and ⁇ r is the relative dielectric constant of the dielectric substrate 1.
  • the size of the dielectric resonator 3 and the size of the slit 4 are adjusted such that the operating frequencies of the two radiation modes are close to each other to broaden the bandwidth.
  • FIG. 2 is a schematic top view of the antenna structure of the example.
  • the plane geometric center of the dielectric resonator 3 and the slit 4 coincides with the position 8.
  • the width of the L-shaped microstrip line 5 is set to satisfy the 50 ohm impedance.
  • the L-shaped microstrip line 5 is bent at 90 degrees at the position 8 to form an L shape, and the dielectric resonator 3 and the slit 4 are excited.
  • the dielectric resonator 3 produces a degenerate mode of two orthogonal polarizations, and the same slit 4 also produces two orthogonally polarized radiation modes.
  • both the dielectric resonator 3 and the slit 4 can be subjected to circularly polarized radiation.
  • the length of the open end optionally 0.4 ⁇ g, wherein, ⁇ g the wavelength of the electromagnetic wave dielectric substrate is at a 6.5GHz frequency.
  • broadband impedance matching can be achieved by adding a matching network between the L-shaped microstrip line 5 and the feed port 7.
  • Matching networks can use impedance transform lines or lumped components (capacitor / inductor) network.
  • the antenna is impedance matched using the impedance transform line 6. Changing the line width of the impedance conversion line 6 changes its characteristic impedance, and controlling its length enables impedance matching in the desired operating frequency band.
  • the dielectric resonators in this example may include cylinders, cuboids, toroids, and spheres and other dielectric resonators having a degenerate mode of operation.
  • the annular gap described in this example may also be replaced by a square ring slit and other annular slits having a symmetrical structure.
  • the matching network of this example may include a matching network such as an impedance transformation line, a lumped element network, or the like that can achieve impedance matching.
  • the present example provides a parameter determining method for a broadband circularly polarized dielectric resonant antenna, which may include the following steps:
  • Step S210 Adjust the size of the dielectric resonator and the annular slit so that the resonance frequencies of the two are close to each other to expand the antenna bandwidth.
  • Step S220 The L-shaped microstrip line is used to provide excitation to the dielectric resonator and the annular slit to generate an operation mode in which the orthogonal polarization phase difference is 90°, thereby realizing circularly polarized radiation.
  • Step 230 Perform impedance matching on the antenna by using a matching network to broaden the impedance bandwidth.
  • FIG. 10 is a view showing the axial ratio of the different inner diameters R of the annular slits at a plurality of frequency points in the circularly polarized dielectric resonator antenna provided by the first example or the circularly polarized dielectric resonant antenna fabricated by the method according to the second embodiment.
  • the bandwidth corresponding to the axial ratio less than 3 dB is the bandwidth of the circularly polarized dielectric resonant antenna
  • different R causes a change in the axial ratio bandwidth of the circularly polarized dielectric resonant antenna.
  • the axial ratio at a plurality of frequency points when R is 2.9 mm, 3.0 mm, and 3.1 mm is given in Fig. 10, respectively.
  • the embodiment of the invention further provides a computer readable storage medium storing computer executable instructions, which are implemented to implement a parameter determination method of the circularly polarized dielectric resonant antenna.
  • the disclosed apparatus and method can be implemented in other manners.
  • the device embodiments described above may only be illustrative.
  • the division of the unit may only be a logical function division.
  • there may be another division manner for example, multiple units or components may be combined. Or can be integrated into another system, or one These features can be ignored or not executed.
  • the coupling, or direct coupling, or communication connection of the various components shown or discussed may be indirect coupling or communication connection through some interface, device or unit, and may be electrical, mechanical or otherwise. of.
  • the units described above as separate components may or may not be physically separated, and the components displayed as the unit may or may not be physical units, that is, may be located in one place or distributed to multiple network units; Some or all of the units may be selected according to actual needs to achieve the purpose of the solution of the embodiment.
  • the functional units in the embodiments of the present invention may all be integrated into one processing module, or different units may be separately used as one unit, or two or more units may be integrated into one unit; the above integrated unit It can be implemented in the form of hardware or in the form of hardware plus software functional units.
  • the foregoing program may be stored in a computer readable storage medium, and the program is executed when executed.
  • the foregoing storage medium may include: a mobile storage device, a read-only memory (ROM), a random access memory (RAM), a magnetic disk, or an optical disk.
  • ROM read-only memory
  • RAM random access memory
  • magnetic disk or an optical disk.
  • optical disk A medium that can store program code.
  • all or part of the steps of the above embodiments may also be implemented by using an integrated circuit. These steps may be separately fabricated into individual integrated circuit modules, or multiple modules or steps may be fabricated into a single integrated circuit module. achieve.
  • the devices/function modules/functional units in the above embodiments may be implemented by a general-purpose computing device, which may be centralized on a single computing device or distributed over a network of multiple computing devices.
  • the device/function module/function unit in the above embodiment is implemented in the form of a software function module. And when sold or used as a stand-alone product, it can be stored on a computer readable storage medium.
  • the above mentioned computer readable storage medium may be a read only memory, a magnetic disk or an optical disk or the like.
  • the circularly polarized dielectric resonant antenna and the parameter determining method and the communication device provided by the embodiments of the present invention simultaneously use the dielectric resonator and the slit as a radiator to perform signal radiation, and corresponding to the first frequency band and the gap corresponding to the dielectric resonator
  • the second frequency band is at least partially different, and the first frequency band and the second frequency band together form a continuous frequency band in which the return loss is less than a specified value, that is, the third frequency band.
  • the bandwidth corresponding to the third frequency band is large, and the circularly polarized dielectric resonant antenna thus formed has the characteristics of simple structure and simple fabrication.

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Abstract

一种圆极化介质谐振天线包括：介质基板，包括第一表面和位于所述第一表面背面的第二表面；所述第一表面设置有金属地板；介质谐振器，与所述金属地板接触，设置为：辐射无线信号，且工作在第一频带；缝隙，位于所述金属地板上，设置为：辐射无线信号，且工作在第二频带；其中，所述第一频带和所述第二频带至少部分不同，且共同形成第三频带；所述第三频带为连续频带；所述圆极化介质谐振天线工作在所述第三频带时的回波损耗，大于指定值；馈线，位于所述第二表面，分别与所述介质谐振器和所述环状缝隙耦合，设置为：分别使所述介质谐振器和所述缝隙进行圆极化辐射。

Description

圆极化介质谐振天线及其参数确定方法和通信设备 技术领域
本申请涉及但不限于通信领域，尤其涉及一种圆极化介质谐振天线及其参数确定方法和通信设备。
背景技术
现有移动通信中，通信设备的工作频段主要集中在3GHz以下，这使得频谱资源十分拥挤，而在高于3GHz的高频段(如毫米波、厘米波频段)可用频谱资源丰富，能够有效缓解频谱资源紧张的现状，可以实现高速短距离通信，支持第五代移动通信系统(5G)容量和传输速率等方面的需求。然而在毫米波和亚毫米波频段，金属天线的导体损耗很严重，导致天线的辐射效率很低。由于不存在导体损耗和表面波损耗，介质谐振天线的辐射效率很高，在毫米波段可达到95％以上。因此，介质谐振天线受到了广泛的关注和研究。介质谐振天线(Dielectric Resonator Antenna，简称DRA)通常工作带宽较窄，尤其是单馈电方式的圆极化介质谐振天线。带宽提升是介质谐振天线的一个主要研究方向。
发明内容
以下是对本文详细描述的主题的概述。本概述并非是为了限制权利要求的保护范围。
本文提供一种圆极化介质谐振天线及其参数确定方法和通信设备，以至少部分解决圆极化介质谐振天线带宽小的问题。
本发明实施例提供一种圆极化介质谐振天线，所述圆极化介质谐振天线包括：
介质基板，包括第一表面和位于所述第一表面背面的第二表面；所述第一表面设置有金属地板；
介质谐振器，与所述金属地板接触，设置为：辐射无线信号，且工作在 第一频带；
缝隙，位于所述金属地板上，设置为：辐射无线信号，且工作在第二频带；其中，所述第一频带和所述第二频带至少部分不同，且共同组成第三频带；所述第三频带为连续频带；所述圆极化介质谐振天线工作在所述第三频带时的回波损耗，大于指定值；
馈线，位于所述第二表面，分别与所述介质谐振器和所述缝隙耦合，设置为：分别使所述介质谐振器和所述缝隙进行圆极化辐射。
可选地，所述介质谐振器为具有简并工作模式的谐振器。
可选地，所述介质谐振器的形态为圆柱体或长方体。
可选地，所述缝隙为环状缝隙。
可选地，所述环状缝隙为轴对称环缝隙。
可选地，所述天线还包括：馈电端口和匹配网络；
所述馈电端口通过所述匹配网络与所述馈线连接；
其中，所述匹配网络设置为：对所述圆极化介质谐振天线进行阻抗匹配。
可选地，所述馈线包括L形微带线。
可选地，所述介质谐振器与所述金属地板接触的一面的几何中心和所述缝隙的几何中心重叠。
本发明实施例还提供一种通信设备，所述通信设备包括：如上所述的圆极化介质谐振天线。
本发明实施例还提供一种圆极化介质谐振天线的参数确定方法，所述方法包括：
调整介质谐振器和缝隙至少其中之一的物理参数，使所述介质谐振器工作在第一频带且使所述缝隙辐射无线信号并工作在第二频带；其中，所述第一频带和第二频带共同组成第三频带，所述第三频带为连续频带；所述圆极化介质谐振天线工作在所述第三频带时的回波损耗，大于指定值；
调整馈线的物理参数，分别使所述介质谐振器和所述缝隙进行圆极化辐射。
可选地，所述缝隙为环状缝隙；
所述调整介质谐振器和缝隙至少其中之一的物理参数，使所述介质谐振器工作在第一频带且使所述缝隙辐射无线信号并工作在第二频带包括：
确定所述介质谐振器的物理参数及所述第一谐振频率；其中，所述第一谐振频率为所述第一频带的谐振频带；
根据所述第一谐振频率，估算得到环状缝隙的第二谐振估算频率；
基于所述第二谐振估算频率，确定所述环状缝隙的估算参数；
基于所述介质谐振器的物理参数及所述估算参数，对所述圆极化介质谐振天线进行仿真，得到仿真结果；
根据所述仿真结果，确定所述环状缝隙的物理参数。
可选地，所述调整馈线的物理参数，使所述介质谐振器和所述缝隙分别进行圆极化辐射包括：调整L微带线的物理参数，使所述介质谐振器和所述缝隙分别进行圆极化辐射。
本发明实施例还提供了一种计算机可读存储介质，存储有计算机可执行指令，所述计算机可执行指令被执行时实现上述圆极化介质谐振天线的参数确定方法。
本发明实施例提供的圆极化介质谐振天线及其参数确定方法和通信设备，同时将介质谐振器和缝隙作为辐射体，进行信号的辐射，且使介质谐振器对应的第一频带和缝隙对应的第二频带至少部分不同，且使第一频带和第二频带共同形成回波损耗均小于指定值的连续频带，即所述第三频带。第三频带对应的带宽大，且这样形成的圆极化介质谐振天线具有结构简单及制作简单的特点。
在阅读并理解了附图和详细描述后，可以明白其他方面。
附图概述
图1至图3为本发明实施例提供的圆极化介质谐振天线的结构示意图；
图4为本发明实施例提供的圆极化介质谐振天线回波损耗频率响应曲线 仿真和实测示意图；
图5为本发明实施例提供的圆极化介质谐振天线的轴比频率响应曲线仿真和实测示意图；
图6为本发明实施例提供的圆极化介质谐振天线在三个不同频点的辐射方向仿真和实测示意图；
图7为本发明实施例提供的一种圆极化介质谐振天线的参数确定流程示意图；
图8为本发明实施例提供的一种调整介质谐振器和缝隙的物理参数的流程示意图；
图9为本发明实施例提供的另一种圆极化介质谐振天线的参数确定流程示意图；
图10为本发明实施例提供的一种圆环缝隙的内径R与圆极化天线在频带的轴比的关系示意图。
本发明的较佳实施方式
下面结合附图对本发明的实施方式进行描述。需要说明的是，在不冲突的情况下，本申请中的实施例及实施例中的各种方式可以相互组合。
如图1和图2所示，本实施例提供了一种圆极化介质谐振天线，所述圆极化介质谐振天线包括：
介质基板1，包括第一表面和位于所述第一表面背面的第二表面；所述第一表面设置有金属地板2；
介质谐振器3，与所述金属地板2接触，设置为：辐射无线信号，且工作在第一频带；
缝隙4，位于所述金属地板2上，设置为：辐射无线信号，且工作在第二频带；其中，所述第一频带和所述第二频带共同组成第三频带；所述第三频带为连续频带；所述圆极化介质谐振天线工作在所述第三频带时的回波损耗，大于指定值；所述缝隙4可为条状缝隙等各种形状的缝隙，在本实施例中可选地可为如图1和图2所示的环状缝隙；
馈线5，位于所述第二表面，分别与所述介质谐振器3和所述缝隙4耦合，设置为：分别使所述介质谐振器3和所述缝隙4进行圆极化辐射。
所述介质基板1呈板状，例如，所述介质基板1可为矩形板，或方形板。所述介质基板1的6个表面中，表面积较大的两个面通常为本实施例中所述的第一表面和第二表面。且所述第一表面和第二表面可互为正反面。在本实施例中在介质基板1的第一表面设置有金属地板2。所述金属地板2可由贴合或电镀在所述介质基板1上的金属层构成，且可与天线或设备内的接地点连通，故又称接地板。在本实施例中所述金属地板2的与第二表面接触的一面可与所述第二表面等面积，且相互对齐。这里的介质基板1可为由介电常数为预设值的介质构成。本实施例构成所述介质基板1的介质的介电常数，可以参见相关技术中介质基板的介质的介电常数，在此就不详细描述了。
所述介质谐振器3为能够辐射无线信号的一种辐射体，通常由相对介电常数相对较高的介质构成。一般情况下所述介质谐振器的相对介电常数为6到140，在本实施例中所述介质谐振器3的相对介电常数可为10.2，但不局限于10.2或6到140。
在相关技术中，通常缝隙在圆极化介质谐振天线中都是作为耦合馈电部件存在，设置为向介质谐振器3进行馈电的，在本实施例中缝隙将设置为进行无线信号的辐射。且在本实施例中设置为辐射信号的缝隙可为所述金属地板2上的缝隙。在该缝隙4内侧还可保留有金属地板2的金属部分。
所述馈线5可为各种能够分别与所述介质谐振器3和缝隙4进行电磁耦合，以进行馈电的馈电结构。在本实施例中所述馈线5可采用单馈线，在一个馈电区域内通过电磁耦合作用，向介质谐振器3和缝隙4传输信号，以实现馈电。在本实施例中所述馈线5还可通过调整自身的物理参数，可以使所述介质谐振器3和缝隙4分别实现圆极化辐射，从而使得本实施例提供的圆极化介质谐振天线实现天线的圆极化辐射。
显然，在本实施例中所述的圆极化介质谐振天线中，同时将介质谐振器3和缝隙4作为辐射无线信号的辐射体，进行无线信号辐射。且可以通过调整介质谐振器3和缝隙4的物理参数，可以使介质谐振器3和缝隙4的谐振频率靠近，从而实现介质谐振器3和缝隙4的工作频带的靠近。在本实施例 中假设介质谐振器的工作频带为第一频带，缝隙4的工作频带为第二频带。在本实施例中第一频带和第二频带可至少部分不同。这里第一频带和第二频带至少部分不同可分为两种情况，一种情况为第一频带和第二频带有部分重叠；第二种情况，第一频带和第二频带相邻。在本实施例中第一频带和第二频带共同形成了第三频带，所述第三频带为本实施例中所述圆极化介质谐振天线的工作频带，圆极化介质谐振天线工作在第三频带时的回波损耗较大，在本实施例中要求大于指定值。在实现过程中，所述指定值可为10dB。
显然在本实施例中利用缝隙4参与无线信号的辐射，使得圆极化介质谐振天线的工作带宽为第三频带，而第三频带显然是大于第一频带，故提升了圆极化介质谐振天线的工作带宽，解决了相关技术中圆极化介质谐振天线的工作带宽窄的问题。
表1为本实施例中所述圆极化介质谐振天线与相关技术中的圆极化介质谐振天线的轴比相对带宽的比对。
表1
Figure PCTCN2016083349-appb-000001
在本实施例中，所述介质谐振器3为具有简并工作模式的谐振器。且通常当具有简并工作模式的谐振器，应用于本发明实施例所述的圆极化介质谐振天线时，其一般都工作在简并工作模式。工作在简并工作模式的介质谐振器3，通常都有两种相互正交的极化模式。所述介质谐振器3还将工作在这两种正交极化模式。可选地，所述介质谐振器3形态可为圆柱体或长方体。在本实施例中所述介质谐振器3可选地可为圆柱体或长方体，以简化制作及 对精确度的控制。
所述缝隙4可选地可为环状缝隙，例如可为轴对称环缝隙。在本实施例中实践证明所述缝隙4为轴对称缝隙的辐射效果好。在本实施例中缝隙4的对称轴一般将通过缝隙4的几何中心。在本实施例中所述缝隙4可为圆环缝隙、矩形环缝隙、椭圆环缝隙、六边形环缝隙等具有轴对称的环状缝隙。在本实施例中，可选地，所述环状缝隙还可为中心对称缝隙，对称中心即为所述缝隙4的几何中心。这里的中心对称的缝隙4可包括圆环缝隙，对称中心为该圆环缝隙的环心。所述中心对称的缝隙4还可包括方形环缝隙，对称中心为该方形环缝隙所在正方形的中心。
在实现时，通常所述介质谐振器3在所述金属地板2上的垂直投影是覆盖所述缝隙4的。一般情况下，所述介质谐振器3与金属地板2接触的底面是完全覆盖住所述缝隙4的，即所述介质谐振器3与金属地板2接触的底面的面积将不小于缝隙4所包围的面积。
如图1至图2所示，在本实施例中所述天线还包括：馈电端口7和匹配网络6；所述馈电端口7通过所述匹配网络6与所述馈线5连接；其中，所述匹配网络6设置为：对所述圆极化介质谐振天线进行阻抗匹配。所述馈电端口7与设备内的馈电网络连接，设置为：接收馈电网络供给的电信号。所述匹配网络6可包括阻抗变换线或集总元件网络等各种能够实现阻抗匹配的结构。所述集总元件可包括电容或电感等元件。在本实施例中可以通过改变所述阻抗变换线的线宽、改变集总元件网络中元器件的电特性参数，调整匹配网络的阻抗，从而实现阻抗匹配。调整集总元件网络中元器件的电特性参数，可包括调整电容的电容值、调整电感的电感值及调整电阻的电阻值中的一个或多个等以实现电特性参数的调整。
在本实施例中所述馈线可选地可为L形馈线。L形馈线可分为两个部分，这两个部分的夹角可为90°。在本实施例中可选地可为L形微带线。微带线是一种能够安装在介质基板上的单一导体构成的微波传输线，具有传输阻抗低，信号传输效率好的特点；且直接采用L形微带线进行馈电，还具有结构简单、成本低及制作简单效率高的特点。
在本实施例中，所述介质谐振器3与所述金属地板2接触的一面的几何 中心和所述缝隙4的几何中心重叠。在图1和图2中所述介质谐振器3的形态为圆柱体。所述缝隙4为圆环缝隙4。所述圆柱体的介质谐振器3与金属地板2接触的圆形底面的圆心与圆环缝隙的环心重合。
天线极化是描述天线辐射电磁波矢量空间指向的参数。由于电场与磁场有恒定的关系，故一般都以电场矢量的空间指向作为天线辐射电磁波的极化方向。在本实施例中所述极化模式可为天线极化的方式。
在本实施例中所述介质谐振器3和缝隙4在各自的极化模式的谐振频率工作时的相位相差保持在90度左右，这样就能实现圆极化辐射。在本实施例中所述预设相位值通常是一个很小的值，例如正负3度、正负2度等取值。在本实施例中所述介质谐振器3和缝隙4在各自极化模式的谐振频率工作时的相位相差可能为大于0度的正相位值，也可能为小于0度的负相位值。总之所述相位相差的绝对值在90度左右。
图3所示的为本发明实施例提供的介质谐振天线中多个部件的参数示意图。在图3中用Wslot表示环状缝隙的缝宽；所述R为环状缝隙的内环的半径；所述Ls为L形微带线的开路端的长度；所述L1为L形微带线与阻抗变换线连接端的长度；所述Lt为阻抗变换线的长度；所述Wt为阻抗变换线的宽度。所述h为介质基板的厚度；所述d为介质谐振器在z轴上的高度；所述2a为介质谐振器的径长。W50为馈电端口的在x轴上的宽度；所述L2为馈电端口的在y轴上的宽度。x轴、y轴及z轴共同形成空间直角坐标系。
图4至图6为本实施例提供的圆极化介质谐振天线的仿真和实测图。该圆极化介质谐振天线的参数如下：L1＝17mm,L2＝6.5mm,Lt＝5mm,R＝3mm,Wslot＝0.25mm，Wt＝W50＝3mm，W＝L＝60mm，a＝5mm，d＝12mm，Wt＝1.5mm，Ls＝11.5mm。
图4为本发明实施例回波损耗频率响应曲线仿真和实测图。天线实测10dB阻抗带宽覆盖5.85GHz-7.20GHz，相对带宽为20.7％。
图5为本发明实施例轴比频率响应曲线仿真和实测图。天线实测3dB阻抗带宽覆盖6.22GHz-7.20GHz，相对带宽为14.6％，包含在10dB阻抗带宽内。天线的轴比定义：任意极化波的瞬时电场矢量的端点轨迹为一椭圆，椭圆的长轴2A和短轴2B之比称之为轴比AR(Axial Ratio)。轴比是圆极化天线的一 个重要的性能指标，它代表圆极化的纯度，轴比不大于3dB的带宽，定义为天线的圆极化带宽。它是衡量天线对不同方向的信号增益差异性的重要指标。
图6为本发明实施例工作带宽内三个频点的辐射方向图仿真和实测图。图6中a图为本发明实施例圆极化介质谐振天线工作在6.22GHz的辐射方向仿真和实测图。图6中b图为本发明实施例圆极化介质谐振天线工作在6.7GHz的辐射方向仿真和实测图。图6中c图为本发明实施例圆极化介质谐振天线工作在7.1GHz的辐射方向仿真和实测图。
观测图6中每个辐射方向图可知，轴向(θ＝0°)的主极化(左旋圆极化)比交叉极化(右旋圆极化)高出15dB以上，具有较好的左旋圆极化特性。
在本发明实施例中，若馈线为向右弯折90°的L形微带线，将形成左旋圆极化特性。若馈线为向左弯折90°的L形微带线，形成右旋圆极化特性。可根据实际工作所需极化选择L形微带线的弯折方向。
本发明实施例提供一种通信设备，所述通信终端包括：如上所述的圆极化介质谐振天线的至少其中之一。这里的通信设备可为各种移动通信设备，例如，手机、平板电脑或各种车载的移动通信设备。在本实施例提供的通信设备中可包括馈电网络及通信芯片，该馈电网络连接馈电端口，根据通信芯片控制所述圆极化介质谐振天线的工作。例如，通过控制馈电网络向馈线传输的电信号，控制介质谐振器和环形馈线的无线信号的辐射等。
总之，本实施例所述通信设备采用了以上所述的圆极化介质谐振天线，具有天线结构简单、成本低、工作带宽大通信质量好的特点。
如图7所示，本实施例提供一种圆极化介质谐振天线的参数确定方法，所述方法包括：
步骤S110：调整介质谐振器和缝隙至少其中之一的物理参数，使所述介质谐振器工作在第一频带且使所述缝隙辐射无线信号并工作在第二频带；其中，所述第一频带和所述第二频带至少部分不同，且共同形成第三频带，所述第三频带为连续频带；所述圆极化介质谐振天线工作在所述第三频带时的回波损耗，大于指定值；
步骤S120：调整馈线的物理参数，使所述介质谐振器和所述缝隙分别进 行圆极化辐射。
在本实施例中将通过调整介质谐振器和缝隙两者中的至少一个的物理参数，使得介质谐振器和环状缝隙都参与无线信号的辐射。且介质谐振器和缝隙的工作频带至少部分不同，这样通过缝隙的工作频带来拓宽天线的工作带宽；从而采用这种方式确定参数得到的圆极化谐振天线具有工作带宽高的特点。本实施例中所述指定值可以参见前述实施例中的相关描述，在此就不在重复了。
在本实施例中所述步骤S110可包括：固定所述介质谐振器的物理参数，调整缝隙的物理参数；固定缝隙的物理参数，调整介质谐振器的物理参数；也可包括同时调整介质谐振器和缝隙的物理参数。这里的介质谐振器的物理参数可包括：介质谐振器的尺寸参数、形状参数和介电常数。这里的尺寸参数可包括介质谐振器的长、宽和高的取值。所述形状参数可包括改变介质谐振器的形状，例如，将圆柱体的介质谐振器变更成长方体的介质谐振器。所述介电参数可包括所述介质谐振器的介质的介电常数等。所述缝隙的物理参数可包括缝隙的形状参数和尺寸参数。调整所述形状参数可包括：将圆环缝隙变更为椭圆环缝隙或方形环缝隙。调整所述缝隙的尺寸参数可包括：调整缝宽和调整缝隙的环径。例如，调整圆环缝隙所在圆环的半径或直径可为调整缝隙的环径的一种。
在步骤S120中确定了介质谐振器和环状缝隙的物理参数之后，将调整馈线的物理参数。这里调整馈线的物理参数可包括调整馈线的尺寸参数、馈电参数及材料参数、形状参数等。例如，调整L形馈线的线长可为调整馈线的尺寸参数的一种。调整馈电参数可包括调整馈电位置、馈电区域的大小等参数。调整材料参数可包括调整馈线的组成材料。
总之，在本实施例中通过调整馈线的物理参数，使介质谐振器和环状缝隙分别向两个正交方向极化，就能简便地实现圆极化辐射，形成圆极化天线。本实施例所述的圆极化介质谐振天线的参数确定方法，可应用于各种具有信息处理功能的电子设备中的圆极化介质谐振天线的自动设计场景中。电子设备通过执行上述方法，可以设计出一个满足较大工作带宽的圆极化介质谐振天线。
在本实施例中所述圆极化介质谐振天线的缝隙为环状缝隙。如图8所示，所述步骤S110可包括：
步骤S111：确定所述介质谐振器的物理参数及所述第一谐振频率；
步骤S112：根据所述第一谐振频率，估算得到所述环状缝隙的第二谐振估算频率；
步骤S113：基于所述第二谐振估算频率，确定所述环状缝隙的估算参数；
步骤S114：基于所述介质谐振器的物理参数及所述估算参数，对所述圆极化介质谐振天线进行仿真，得到仿真结果；
步骤S115：根据所述仿真结果，确定环状缝隙的物理参数。
在本实施例中所述确定介质谐振器可包括：进行参数确定的电子设备，从人机交互接口或从其他设备接收所述介质谐振器的物理参数。再根据介质谐振器的物理参数估算出所述第二谐振估算频率。例如，可利用介质波导模型近似计算出所述介质谐振器的谐振频率，在本实施例中称为第一谐振频率。
在步骤S112中，可以根据第一频带与第二频带至少部分不重合的特点，确定出第一谐振频率与环状缝隙的谐振频率相互靠近，但是不重合。在本实施例中可以将根据经验值得到的估算常数等，将所述第一谐振频带和估算常数按照预设函数关系计算之后，得到所述第二谐振估算频率。例如，将所述第一谐振频率加上所述估算参数或减去所述估算参数，即可得到所述第二谐振估算频率。再利用所述第二谐振估算频率反推环状缝隙的物理参数，这里反推得到的物理参数即为前述的估算参数。例如，以下以圆环缝隙为例，可利用如下公式推导估算参数：
Figure PCTCN2016083349-appb-000002
所述fslot为所述第二谐振估算频率，c为真空中光速，R为圆环缝隙内径，Wslot为圆环缝隙宽度，εeff由下式计算：
Figure PCTCN2016083349-appb-000003
其中εd为介质谐振器的相对介电常数，而εr为介质基板1的相对介电常 数。
显然通过上述公式(1)和(2)，将得到至少一组Wslot和R。
在步骤S114中，将根据介质谐振器的物理参数和估算参数，进行仿真得到仿真结果。若仿真结果表明满足预设条件，该估算参数可直接确定为环状缝隙的物理参数。若仿真结果不满足所述预设条件，则可认为仿真不通过，将根据仿真结果调整环状缝隙的估算参数，再次返回步骤114，直至仿真通过。这里的仿真满足预设条件可包括：仿真得到的回波损耗频率响应仿真曲线中，在回波损耗小于所述指定值的连续频带内有两个回波损耗的波谷。当然这里的预设条件还可以不限于上述条件。
当然在实现时，所述方法还可包括：
步骤S116：仿真通过之后，还可以包括根据介质谐振器的物理参数和估算参数，制作圆极化介质谐振天线；
步骤S117：对圆极化介质谐振天线进行实测，形成实测结果；再根据实测结果，确定环状缝隙的最终物理参数。这里若实测结果表明满足预设条件，则可以直接将对应的估算参数作为环状缝隙的最终物理参数，若实测结果表明不满足预设条件，则可以根据实测结果调整估算参数，返回步骤S114或步骤S115，直至实测通过。
以下根据上述实施例提供的任意一个技术方案，提供几个示例：
示例一：
如图1和图2所示，本示例提供一种宽带圆极化介质谐振天线，该天线包括介质基板1、金属地板2、介质谐振器3、圆环缝隙4、L形微带线5、阻抗变换线6和馈电端口7。金属地板2、圆环缝隙4和介质谐振器3位于介质基板1的同一侧，L形微带线5和阻抗变换线6位于介质基板1的另一侧。
介质谐振器3可为由介质材料形成的圆柱体介质谐振器，其一般具有较高的相对介电常数(6-140)。在本实施例中的相对介电常数为10.2。
介质谐振器3固定在金属地板2上方，其下表面中心可与位于金属地板2上的缝隙4的中心重合。
介质谐振器3辐射模式的工作频率，可通过介质波导模型(Dielectric  Waveguide Model,DWM)近似计算得出。在本示例中，选取HEM11δ模式作为圆柱体介质谐振器3的工作模式，其谐振频率可由下式近似得出：
Figure PCTCN2016083349-appb-000004
其中，c为真空中光速，εd为介质谐振器的相对介电常数，a为圆柱介质谐振器半径，x＝a/(2d)，d为圆柱介质谐振器高度。
本示例中圆环缝隙4的谐振频率可由下式近似得出：
Figure PCTCN2016083349-appb-000005
其中c为真空中光速，R为圆环缝隙内径，wslot为圆环缝隙宽度，εeff由下式计算：
Figure PCTCN2016083349-appb-000006
其中εd为介质谐振器3的相对介电常数，而εr为介质基板1的相对介电常数。
调整介质谐振器3尺寸和缝隙4尺寸，使两者辐射模式的工作频点相靠近，以展宽带宽。
图2可为本示例天线结构俯视示意图。介质谐振器3和缝隙4的面几何中心重合于位置8。在本发明实施例中，设置L形微带线5的宽度以满足50欧姆阻抗。L形微带线5在位置8处弯折90度形成L形，对介质谐振器3和缝隙4进行激励。通过这种激励方式，介质谐振器3会产生两种正交极化的简并模式，同样缝隙4也会产生两种正交极化的辐射模式。通过调整L形微带线5的开路端长度，可以使介质谐振器3和缝隙4都进行圆极化辐射。在本示例中可选择开路端长度为0.4λg，其中，λg为6.5GHz频点处介质基板中的电磁波波长。
根据天线的阻抗匹配情况，可通过在L形微带线5和馈电端口7之间增加匹配网络来实现宽带阻抗匹配。匹配网络可采用阻抗变换线或者集总元件 (电容/电感)网络。在本示例中，采用阻抗变换线6对天线进行阻抗匹配。改变阻抗变换线6的线宽可改变其特性阻抗，控制其长度可实现在所需工作频段实现阻抗匹配。
本示例中所述介质谐振器可包括圆柱体、长方体、圆环形和球形和其它具有简并工作模式的介质谐振器。本示例所述圆环缝隙还可由方环缝隙和其它具有对称结构的环状缝隙替代。本示例所述匹配网络可包括阻抗变换线、集总元件网络等能够实现阻抗匹配的匹配网络。
示例二：
参见图9，本示例提供宽带圆极化介质谐振天线的参数确定方法，可包括如下步骤：
步骤S210：调整介质谐振器和环状缝隙的尺寸，以使两者的谐振频率相靠近，以扩展天线带宽。
步骤S220：采用L形微带线对介质谐振器和环状缝隙提供激励，以产生正交极化相位差为90°的工作模式，从而实现圆极化辐射。
步骤230：采用匹配网络对天线进行阻抗匹配，以展宽阻抗带宽。
图10所示为示例一提供的圆极化介质谐振天线或基于示例二提供的方法制作的圆极化介质谐振天线中，环状缝隙的不同内径R在多个频点的轴比。若轴比小于3dB所对应的带宽为圆极化介质谐振天线的带宽，则可知不同的R会导致圆极化介质谐振天线的轴比带宽的变化。在图10中分别给出了R为2.9mm、3.0mm和3.1mm时在多个频点的轴比。显然，当R＝3.0时，对应圆极化介质谐振天线的轴比带宽最大。
本发明实施例还提供了一种计算机可读存储介质，存储有计算机可执行指令，所述计算机可执行指令被执行时实现上述圆极化介质谐振天线的参数确定方法。
在本申请所提供的实施例中，可以理解到，所揭露的设备和方法，可以通过其它的方式实现。以上所描述的设备实施例仅仅可以是示意性的，例如，所述单元的划分，仅仅可以为一种逻辑功能划分，实际实现时可以有另外的划分方式，如：多个单元或组件可以结合，或可以集成到另一个系统，或一 些特征可以忽略，或不执行。另外，所显示或讨论的不同组成部分相互之间的耦合、或直接耦合、或通信连接可以是通过一些接口、设备或单元的间接耦合或通信连接，可以是电性的、机械的或其它形式的。
上述作为分离部件说明的单元可以是、或也可以不是物理上分开的，作为单元显示的部件可以是、或也可以不是物理单元，即可以位于一个地方，也可以分布到多个网络单元上；可以根据实际的需要选择其中的部分或全部单元来实现本实施例方案的目的。
另外，在本发明实施例中的功能单元可以全部集成在一个处理模块中，也可以是不同单元分别单独作为一个单元，也可以两个或两个以上单元集成在一个单元中；上述集成的单元既可以采用硬件的形式实现，也可以采用硬件加软件功能单元的形式实现。
本领域普通技术人员可以理解：实现上述方法实施例的全部或部分步骤可以通过程序指令相关的硬件来完成，前述的程序可以存储于一计算机可读取存储介质中，该程序在执行时，执行包括上述方法实施例的步骤；而前述的存储介质可包括：移动存储设备、只读存储器(ROM，Read-Only Memory)、随机存取存储器(RAM，Random Access Memory)、磁碟或者光盘等各种可以存储程序代码的介质。
本领域普通技术人员可以理解上述实施例的全部或部分步骤可以使用计算机程序流程来实现，所述计算机程序可以存储于一计算机可读存储介质中，所述计算机程序在相应的硬件平台上(如系统、设备、装置、器件、处理器等)执行，在执行时，包括方法实施例的步骤之一或其组合。
可选地，上述实施例的全部或部分步骤也可以使用集成电路来实现，这些步骤可以被分别制作成一个个集成电路模块，或者将它们中的多个模块或步骤制作成单个集成电路模块来实现。
上述实施例中的装置/功能模块/功能单元可以采用通用的计算装置来实现，它们可以集中在单个的计算装置上，也可以分布在多个计算装置所组成的网络上。
上述实施例中的装置/功能模块/功能单元以软件功能模块的形式实现 并作为独立的产品销售或使用时，可以存储在一个计算机可读取存储介质中。上述提到的计算机可读取存储介质可以是只读存储器，磁盘或光盘等。
本领域的普通技术人员可以理解，可以对本申请的技术方案进行修改或者等同替换，而不脱离本申请技术方案的精神和范围。本申请的保护范围以权利要求所定义的范围为准。
工业实用性
本发明实施例提供的圆极化介质谐振天线及其参数确定方法和通信设备，同时将介质谐振器和缝隙作为辐射体，进行信号的辐射，且使介质谐振器对应的第一频带和缝隙对应的第二频带至少部分不同，且使第一频带和第二频带共同形成回波损耗均小于指定值的连续频带，即所述第三频带。第三频带对应的带宽大，且这样形成的圆极化介质谐振天线具有结构简单及制作简单的特点。

Claims (12)

  1. 一种圆极化介质谐振天线，所述圆极化介质谐振天线包括：
    介质基板，包括第一表面和位于所述第一表面背面的第二表面；所述第一表面设置有金属地板；
    介质谐振器，与所述金属地板接触，设置为：辐射无线信号，且工作在第一频带；
    缝隙，位于所述金属地板上，设置为：辐射无线信号，且工作在第二频带；其中，所述第一频带和所述第二频带至少部分不同，且共同组成第三频带；所述第三频带为连续频带；所述圆极化介质谐振天线工作在所述第三频带时的回波损耗，大于指定值；
    馈线，位于所述第二表面，分别与所述介质谐振器和所述缝隙耦合，设置为：使所述介质谐振器和所述缝隙分别进行圆极化辐射。
  2. 根据权利要求1所述的天线，其中，
    所述介质谐振器为具有简并工作模式的谐振器。
  3. 根据权利要求2所述的天线，其中，
    所述介质谐振器的形态为圆柱体或长方体。
  4. 根据权利要求1所述的天线，其中，
    所述缝隙为环状缝隙。
  5. 根据权利要求4所述的天线，其中，
    所述环状缝隙为轴对称环缝隙。
  6. 根据权利要求1至5任一项所述的天线，
    所述天线还包括：馈电端口和匹配网络；
    其中，所述馈电端口通过所述匹配网络与所述馈线连接；
    其中，所述匹配网络设置为：对所述圆极化介质谐振天线进行阻抗匹配。
  7. 根据权利要求1至5任一项所述的天线，其中，
    所述馈线包括L形微带线。
  8. 根据权利要求1至5任一项所述的方法，其中，
    所述介质谐振器与所述金属地板接触的一面的几何中心和所述缝隙的几何中心重叠。
  9. 一种通信设备，所述通信设备包括：如权利要求1至8中任一项所述的圆极化介质谐振天线。
  10. 一种圆极化介质谐振天线的参数确定方法，所述方法包括：
    调整介质谐振器和缝隙至少其中之一的物理参数，使所述介质谐振器工作在第一频带且使所述缝隙辐射无线信号并工作在第二频带；其中，所述第一频带和第二频带共同组成第三频带，所述第三频带为连续频带；所述圆极化介质谐振天线工作在所述第三频带时的回波损耗，大于指定值；
    调整馈线的物理参数，使所述介质谐振器和所述缝隙分别进行圆极化辐射。
  11. 根据权利要求10所述的方法，其中，
    所述缝隙为环状缝隙；
    所述调整介质谐振器和缝隙至少其中之一的物理参数，使所述介质谐振器工作在第一频带且使所述缝隙辐射无线信号并工作在第二频带包括：
    确定所述介质谐振器的物理参数及所述第一谐振频率；其中，所述第一谐振频率为所述第一频带的谐振频带；
    根据所述第一谐振频率，估算得到环状缝隙的第二谐振估算频率；
    基于所述第二谐振估算频率，确定所述环状缝隙的估算参数；
    基于所述介质谐振器的物理参数及所述估算参数，对所述圆极化介质谐振天线进行仿真，得到仿真结果；
    根据所述仿真结果，确定所述环状缝隙的物理参数。
  12. 根据权利要求10所述的方法，其中，
    所述调整馈线的物理参数，使所述介质谐振器和所述缝隙分别进行圆极化辐射包括：
    调整L微带线的物理参数，使所述介质谐振器和所述缝隙分别进行圆极化辐射。
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