WO2016056951A1 - Amplificateur de puissance pour l'amplification d'un signal d'entrée en un signal de sortie - Google Patents

Amplificateur de puissance pour l'amplification d'un signal d'entrée en un signal de sortie Download PDF

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Publication number
WO2016056951A1
WO2016056951A1 PCT/SE2014/051155 SE2014051155W WO2016056951A1 WO 2016056951 A1 WO2016056951 A1 WO 2016056951A1 SE 2014051155 W SE2014051155 W SE 2014051155W WO 2016056951 A1 WO2016056951 A1 WO 2016056951A1
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Prior art keywords
signal
input
sub
power amplifier
amplifier
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PCT/SE2014/051155
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English (en)
Inventor
Richard Hellberg
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Telefonaktiebolaget L M Ericsson (Publ)
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Priority to PCT/SE2014/051155 priority Critical patent/WO2016056951A1/fr
Publication of WO2016056951A1 publication Critical patent/WO2016056951A1/fr

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/60—Amplifiers in which coupling networks have distributed constants, e.g. with waveguide resonators
    • H03F3/602—Combinations of several amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H03F1/0288—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers using a main and one or several auxiliary peaking amplifiers whereby the load is connected to the main amplifier using an impedance inverter, e.g. Doherty amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H03F1/0294—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers using vector summing of two or more constant amplitude phase-modulated signals
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/189—High-frequency amplifiers, e.g. radio frequency amplifiers
    • H03F3/19—High-frequency amplifiers, e.g. radio frequency amplifiers with semiconductor devices only
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F3/211—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only using a combination of several amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/24—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages
    • H03F3/245—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages with semiconductor devices only
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/111—Indexing scheme relating to amplifiers the amplifier being a dual or triple band amplifier, e.g. 900 and 1800 MHz, e.g. switched or not switched, simultaneously or not
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/411—Indexing scheme relating to amplifiers the output amplifying stage of an amplifier comprising two power stages

Definitions

  • Embodiments herein relate to power amplifiers for wireless communication systems, such as telecommunication systems.
  • a power amplifier for amplification of an input signal into an output signal is disclosed.
  • a radio network node, comprising the power amplifier, and a user equipment, comprising the power amplifier, are disclosed.
  • Power amplifiers are widely used in communication systems, for example in radio base stations and cellular phones of a cellular radio network.
  • power amplifiers typically amplify signals of high frequencies for providing a radio transmission signal.
  • a consideration in the design of power amplifiers is the efficiency thereof. High efficiency is generally desirable so as to reduce the amount of power that is dissipated as heat.
  • the amount of power that is available may be limited due to powering by a battery, included in e.g. the satellite.
  • An increase in efficiency of the power amplifier would allow an increase of operational time between charging of the battery.
  • a conventional Power Amplifier such as class B, AB, F, has a fixed Radio Frequency (RF) load resistance and a fixed voltage supply.
  • Class B or AB bias causes the output current to have a form close to that of a pulse train of half wave rectified sinusoid current pulses.
  • the Direct Current (DC), and hence DC power is largely proportional to the RF output current amplitude, and voltage.
  • the output power is proportional to the RF output current squared.
  • An efficiency of the conventional power amplifier i.e. output power divided by DC power, is therefore also proportional to the output amplitude. The average efficiency is consequentially low when amplifying signals that on average have a low output amplitude, or power, compared to the maximum required output amplitude.
  • Known RF power amplifiers include both Doherty and Chireix type power amplifiers. These kinds of RF PAs are generally more efficient than the conventional amplifier described above for amplitude-modulated signals with high Peak-to- Average Ratio (PAR), since they have a lower average sum of output currents from the transistors. Reduced average output current means high average efficiency.
  • PAR Peak-to- Average Ratio
  • the reduced average output current is obtained by using two transistors that influence each other's output voltages and currents through a reactive output network, which is coupled to a load.
  • a reactive output network which is coupled to a load.
  • the constituent transistors By driving the constituent transistors with the right amplitudes and phases, the sum of RF output currents is reduced at all levels except the maximum. Also for these power amplifiers the RF voltage at one or both transistor outputs is increased.
  • RF power amplifier can be driven in a so called backed off operation. This means that the power amplifier is operated a certain level, e.g.
  • Backed off operation may also refer to that an instantaneous output power is relatively low.
  • WO03/0611 1 discloses a composite power amplifier including a first and a second power amplifier connected to an input signal over an input network and to a load over an output network.
  • the composite power amplifier may be operated, typically over a 3 to 1 bandwidth, in Doherty mode, in Chireix mode or in other intermediate modes between the Doherty and Chireix modes.
  • the 3 to 1 bandwidth of high efficiency is achieved by devising an output network 13 that has both suitable impedance transformation characteristics and full power output capacity over the bandwidth.
  • This kind of composite power amplifiers are usually driven by complex digital adaptive control systems for providing multiple drive signals to be fed to sub-amplifiers of the composite power amplifier.
  • the drive signals are typically created by individual digital-to-analog converters.
  • US7145387 discloses a system including a 2-stage high-efficiency power amplifier with increased robustness against circuit variations and with radically increased bandwidth of high efficiency. This system is operable in high-efficiency modes, for example at or near a Chireix mode but using Doherty-like drive signals.
  • US7145387 discloses a composite power amplifier including a first and a second power amplifier connected to an input signal over an input network and to a load over an output network. The output network includes phase shifting elements for generating different phase shifts from each power amplifier output to the load.
  • the input network includes means for driving both power amplifiers to produce first output current components having an amplitude that increases linearly with increasing output signal amplitude below a transition point and decreases monotonically with increasing output signal amplitude above said point, and second output current components having an amplitude that increases linearly with increasing output signal amplitude both below and above the transition point.
  • a problem with this system is that it is adapted for operation at one frequency only. Additionally, the drive signal control structure described in US7145387 is based on output network
  • transimpedance relations in 2-stage amplifiers This can be implemented as an analog RF-in circuit.
  • the analog RF-in circuit is difficult to operate automatically, i.e. to obtain the transimpedance relations, or over wide bandwidths.
  • a problem is that these transimpedance relations become severely more difficult to obtain for 3-stage amplifiers and further multi-stage amplifiers.
  • An object is to provide a power amplifier of the above mentioned kind with improved analogue operation.
  • the object is achieved by a power amplifier comprising a first and a second sub-amplifier for amplification of an input signal into an output signal.
  • the first and second sub-amplifiers are connected to an input network for receiving the input signal at an input port of the input network, and the first and second sub-amplifiers are connected to an output network for providing the output signal at an output port of the output network.
  • the input network comprises a first input branch configured to provide a first signal, derived from the input signal.
  • An average rate of amplitude change of the first signal versus amplitude change of the input signal in a first amplitude range is less than an average rate of amplitude change of the first signal versus amplitude change of the input signal in a second amplitude range.
  • the input network further comprises a second input branch configured to provide a second signal, derived from the input signal.
  • the second signal has a triangularly shaped amplitude envelope which increases in the first amplitude range and which decreases in the second amplitude range.
  • the power amplifier is configured to provide the first signal to the first and second sub-amplifiers, and provide the second signal to the first sub-amplifier.
  • the object is achieved by a radio network node, comprising the power amplifier.
  • the object is achieved by a user equipment, comprising the power amplifier.
  • a universal way of controlling and driving a high-efficiency multi-stage power amplifier based on passive output combining networks is provided.
  • the power amplifier according to embodiments herein may be used for single-frequency amplifiers as in prior art, but may be especially well suited for so called wideband power amplifiers.
  • embodiments of the power amplifier makes use of two nonlinear amplitude shaped signals, i.e. the first signal and the second signal.
  • the second signal rises at low amplitudes, e.g. in the first amplitude range, and decreases at high amplitudes, e.g. in the second amplitude range.
  • the second signal is fed to at least one sub-amplifier, in preferred embodiments to a single sub- amplifier.
  • the second signal is controlled in phase and amplitude in accordance with the transimpedance from a single sub-amplifier to the output.
  • the first signal may be low, e.g. zero, in the first amplitude range and increasing in the second amplitude range.
  • the second signal may be fed to all sub-amplifiers based on an in-phase combining criteria at the output port.
  • all sub-amplifiers based on an in-phase combining criteria at the output port.
  • in-phase combining may be achieved by a wideband passive distribution network of transmission lines whose length differences are the mirror image of the ones in the output network.
  • the second signal may in preferred embodiments be routed to a single sub-amplifier by passive combining and gain setting/switching.
  • a wideband approximation to the ideal phase of the first signal may in most cases be provided by inserting this signal at a point in the transmission line distribution network at the input side, possibly also with a static phase shift.
  • the input network is therefore passive, except for switching used to select and adjust which sub-amplifier to be operated at low amplitudes, i.e. in the first and second amplitude ranges.
  • FIG. 1 is a schematic overview of the power amplifier according to embodiments herein,
  • Figure 2 is a diagram illustrating amplitude envelopes for the first and second signal
  • Figure 3a is an illustration of electrical lengths of transmission lines at center frequency
  • Figures 3b-3d are diagrams including graphs over efficiency, transition point amplitude and relative phase versus frequency
  • Figure 4 is another schematic overview of an exemplifying power amplifier according to embodiments herein,
  • Figure 5a is another illustration of electrical lengths of transmission lines at center frequency
  • Figures 5b-5d are other diagrams including graphs over efficiency, transition point amplitude and relative phase versus frequency
  • Figure 6a is a further illustration of electrical lengths of transmission lines at center frequency
  • Figures 6b-6d are further diagrams including graphs over efficiency, transition point amplitude and relative phase versus frequency
  • FIG. 7 is a further schematic overview of another exemplifying power amplifier according to embodiments herein,
  • Figure 8 illustrates an exemplifying radio network node according to embodiments herein.
  • Figure 9 illustrates an exemplifying user equipment according to
  • Figure 1 depicts an exemplifying power amplifier 100, such as a composite power amplifier, according to embodiments herein.
  • the power amplifier 100 comprises a first and a second sub-amplifier 111 , 112 for amplification of an input signal into an output signal.
  • the first and second sub-amplifiers 11 1 , 1 12 may be operated to amplify the input signal into the output signal.
  • the first and second sub-amplifiers 1 11 , 1 12 are connected to an input network 120 for receiving the input signal at an input port 150 of the input network 120. Furthermore, the first and second sub-amplifiers 11 1 , 112 are connected to an output network 130 for providing the output signal at an output port 140 of the output network 130.
  • the input network 120 comprises a first input branch 101 configured to provide a first signal, derived from the input signal. An average rate of amplitude change of the first signal versus amplitude change of the input signal in a first amplitude range is less than an average rate of amplitude change of the first signal versus amplitude change of the input signal in a second amplitude range. See e.g. the dashed line in Figure 2.
  • the first signal may have an amplitude envelope that is below 30% of a maximum value of an operational amplitude range of the power amplifier in the first amplitude range of the operational amplitude range of the power amplifier 100
  • the amplitude envelope of the first signal is below 20%, below 10%, about zero, or zero, depending on desired efficiency of the power amplifier. This means that the lower the amplitude envelope is in the first amplitude range, the higher - thus better- becomes the efficiency of the power amplifier.
  • the highest efficiency is achieved when the amplitude envelope of the first signal is zero in the first amplitude range.
  • the amplitude envelope of the first signal is further increasing in the second amplitude range of the operational amplitude range, wherein amplitudes of the input signal in the first amplitude range are less than amplitudes of the input signal in the second amplitude range.
  • the second amplitude range follows, directly or indirectly, after the first amplitude range.
  • the second amplitude range is subsequent to the first amplitude range with respect to magnitude of the amplitudes in the respective ranges.
  • the amplitude envelope may refer to the power amplifier's response to a test signal, such as the input signal, for which the amplitude is increasingly ramped up.
  • the input network 120 further comprises a second input branch 102 configured to provide a second signal, derived from the input signal.
  • the second signal has a triangularly shaped amplitude envelope which increases in the first amplitude range and which decreases in the second amplitude range.
  • the power amplifier 100 is configured to provide the first signal to the first and second sub- amplifiers 1 11 , 1 12, and provide the second signal to the first sub-amplifier 1 11.
  • the second signal may be based on the first signal.
  • the second signal may for example be achieved by subtracting the first signal from the input signal.
  • the second signal may also be generated by subtracting the first signal from a copy of the input signal. In this manner, generation of the first and second signals is independent from each other.
  • An advantage may hence be that amplitudes of the first and second signals also may be independently adjusted.
  • the first signal may sometimes be referred to as a "Class C"-signal due to its characteristic amplitude response.
  • the second signal may also be generated in many other ways. For example, by means of variable gain amplifiers controlled by respective non-linearly processed peak detector signals.
  • the power amplifier 100 may comprise means configured to enable a concurrent super-positioning of the first and second signals with respect to phase and group delay.
  • the means may be embodied in the form of a phase splitting network. Alternatively or additionally, the means may be realized by means of an offset insertion point as described herein.
  • the means may be part of the input network 120, i.e. the input network 120 is configured to enable concurrent super-positioning of the first and second signals with respect to phase and group delay.
  • the means may be configured to provide the first signal to the first and second sub-amplifiers 1 11 , 1 12 via a first sub-branch 105 and a second sub- branch 106, respectively.
  • the power amplifier 100 may be configured to insert the second signal into the first sub-branch 105 at an insertion point of the first sub-branch 105.
  • Phase and group delay of the triangularly shaped amplitude envelope due to the insertion point may enable the concurrent super-positioning of the first and second signals with respect to phase and group delay. Thanks to the insertion point a difference in electrical length for the first and second signals is achieved.
  • the output network 130 may comprise a first output transmission line 131 , connected to the first sub-amplifier 1 11 , and a second output transmission line 132, connected to the second sub-amplifier 112, wherein the input network 120 may comprise a first input transmission line 121 , connected to the second input branch 102, wherein electrical length of the first input transmission line 121 may be twice as long as electrical length of the first output transmission line 131 , wherein the insertion point may be located at an output 123 of the first input transmission line 121 .
  • the input network 120 may comprise a second input transmission line 122, connected to the second input branch 102, wherein electrical length of the second input transmission line 122 may be twice as long as electrical length of the second output transmission line 132, wherein a further insertion point at the second sub- branch 106 may be located at an output 124 of the second input transmission line 122, wherein phase and group delay of the triangularly shaped amplitude envelope due to the further insertion point may enable concurrent super-positioning of the first and second signals with respect to phase and group delay.
  • the means may comprise a phase splitting network configured to enable the concurrent super-positioning of the first and second signals with respect to phase and group delay.
  • the phase splitting network may be configured to provide the first and second signals at a common insertion point at which electrical length to the first and second sub-amplifiers 111 , 1 12 is equal.
  • the means may comprise a constant phase difference network configured to enable the concurrent super-positioning of the first and second signals with respect to phase and group delay.
  • the power amplifier 100 may be configured to provide the second signal to the first sub-amplifier 11 1 for frequencies in a first frequency range, and to provide the second signal to the second sub-amplifier 1 12 for frequencies in a second frequency range, wherein the first frequency range may be different from the second frequency range.
  • the power amplifier 100 may have a transition interval whose length between the first and second amplitude ranges is configurable, whereby the first and/or second amplitude range is configurable.
  • the transition point is located at the end of the first amplitude range and at the beginning of the second amplitude range.
  • the input network 120 may comprise a first attenuator 161 , connected to the second input branch 102 and the first sub-amplifier 1 11 , and a second attenuator 162, connected to the second input branch 102 and the second sub-amplifier 112, wherein the power amplifier 100 may be configured to activate the first attenuator 161 when the second signal is in the first frequency range, and wherein the power amplifier 100 may be configured to activate the second attenuator 162 when the second signal is in the second frequency range.
  • the input network 120 may comprise a third attenuator 163, connected to an output 104 of the first input branch 101 and an input 103 of the second input branch 102, whereby the second input branch 102 is configured to provide the second signal.
  • the third attenuator 163 controls mix of the first signal and the input signal.
  • High-efficiency multi-stage power amplifiers aka composite power amplifiers, based on passive output combining networks come in many varieties. They are very different in their output network design and the shapes of drive signals to the sub- amplifiers, or contain this diversity within the wide usage band. Different as they are, they also share two characteristics.
  • a first characteristic is that the sum of the maximum output power of all sub- amplifiers should be able to combine at the output port 140.
  • a second characteristic is that there is at least one sub-amplifier that has higher transimpedance to the output port 140 than the other sub-amplifiers. That sub- amplifier is the one that should be active alone in the first amplitude range to increase efficiency. See Figure 2.
  • the in-phase power combining of all sub-amplifiers to full power is achieved by providing the input nodes (gates) of the sub-amplifier transistors with drive signals of the correct phase differences and the correct amplitudes.
  • the maximum drive voltage amplitudes are generally equal if the same types of transistors are used, but the drive currents are proportional to the sizes of the transistors, so the driver amplifiers must be scaled accordingly.
  • the correct phase differences are such that the different phase shifts due to the line lengths, i.e. from amplifier to output, in the output network are cancelled. In this manner, the signal - after going through the different paths - has the same phase at the output.
  • any sub-amplifier 11 1 , 1 12 operating alone at low output amplitudes, i.e. in the first amplitude range, is proportional to the transimpedance from that transistor to the load.
  • Transimpedance means voltage generated at a second node divided by the current input at the first node.
  • High transimpedance to the load therefore means getting more RF voltage at the load for less RF current from the sub-amplifier's transistor.
  • the one with a suitable transimpedance to the output may be chosen as the active one at the lowest output amplitudes.
  • the relative phase of the drive signal to this single sub- amplifier may be chosen so that it has about the same phase at the output as the collective in-phase combined full-amplitude signal.
  • a triangular amplitude drive function i.e. the second signal
  • the second signal has a peak at a transition point at a medium output amplitude and may preferably be zero at zero and full output amplitude. As mentioned above, zero is understood to have some margin, such as 30% of the maximum operational amplitude.
  • the transition point may separate, or may be located between, the first amplitude range and the second amplitude range.
  • a drive amplitude function, i.e. the first signal that may start at zero at the transition point and ends at the highest point is used for the collective drive of all amplifiers to the maximum output.
  • Correct amplitude may be defined by a linear drive signal - as a reference - for one sub-amplifier which drive signal goes from zero to an amplitude at full power that for convenience is
  • the first signal then also goes to 1 at full power.
  • the second signal will have a linear slope in the low amplitude region, just as the linear reference signal, but with a voltage that is scaled down by the transimpedance, but also divided by the sub-amplifiers part of the output power, i.e. an increase since only one sub-amplifier is responsible for the first part, but all sub-amplifiers contribute to the full power ouptut.
  • the correct phase is the one such that the output phase with single-amplifier drive is the same as the phase of the collectively driven, in-phase combined, output.
  • the power amplifier 100 may be a 3:1 bandwidth wideband amplifier provided with an analog signal processing network, such as the input network 120, for controlling it.
  • the transmission line lengths are given as the electrical length in wavelengths at center frequency of the power amplifier 100, i.e. the center frequency of the operational bandwidth of the power amplifier 100. All parts except the transmission lines are assumed to have zero time delay, or have corresponding matched delay in parallel sub-branches.
  • the first input transmission line 121 has an electrical length of one wavelength at center frequency
  • the second input transmission line 122 has an electrical length of one half wavelength at center frequency
  • the first output transmission line 131 has an electrical length of one half wavelength at center frequency
  • the second output transmission line 132 has an electrical length of one quarter wavelength at center frequency.
  • the first signal may be provided by a class C amplifier, e.g. an amplifier 110, for the collective drive to all sub-amplifiers 1 11 , 112.
  • the power amplifier 100 may be resistively loaded in the full frequency range of interest.
  • the correct phases for in- phase combining is achieved over the full frequency range by providing delay networks or lengths of transmission line in the sub-branches 105, 106, which thus may include the first and second input transmission lines 121 , 122, to the sub- amplifiers 1 11 , 112.
  • the total length, i.e. counted as time delay, from a signal splitting point after the class C amplifier, i.e. after the output 104 of the first input branch 101 , to the output port 140, which may be connected to an antenna (not shown) should be the same for the sub-branches 105, 106.
  • the second signal is obtained from the linear input signal minus the class C signal.
  • This second signal, or drive signal is preferably added only to the best sub-amplifier sub-branch with the correct amplitude adjusted by an attenuator 161 or 162, and the other sub-branch gets no signal at all, or a very small signal.
  • the best sub-amplifier sub-branch is generally the one with the highest transimpedance from the sub-amplifier to the output 140. In the present case of an output network 130 with transmission lines directly coupled to the output 140, this is generally the sub-amplifier whose transmission line is closest to an odd multiple of a quarter wavelengths at the operating frequency in question. The higher
  • transimpedance occurs due to the impedance transforming properties of the quarterwave line.
  • phase shift over frequency also called group delay
  • group delay the phase shift over frequency
  • This phase shift behaves largely as the phase shift due to a length of transmission line twice that of the transmission line in the output network.
  • An effective solution is therefore to compensate by inserting the second signal at a point further into the sub-amplifier sub-branch. The point may be inserted at the transmission line sub-branch corresponding to that delay, giving effectively a negative relative delay.
  • Figure 3a an exemplifying configuration of the power amplifier 100 is shown.
  • the first output transmission line 131 has an electrical length of one half wavelength at center frequency and the second output transmission line 132 has an electrical length of one quarter wavelength at center frequency.
  • first sub-amplifier 11 1 ,1 is plotted with a dotted line and the second sub-amplifier 112, 2 is plotted with a solid line. See also legend in Figure 3a, which shows that reference numerals 1 and 2 are associated to dotted and solid lines, respectively.
  • the transition point is in the described implementation adjusted for the frequency of operation by the setting of gate bias voltage to the class C amplifier. It can also be achieved automatically in wide bandwidths by having frequency-dependent gain (e.g. by filters) before and after the class C stage (the second filter equalizing the maximum amplitude across the bandwidth).
  • frequency-dependent gain e.g. by filters
  • phase of the second signal relative to the in-phase combining phase i.e. the first signal
  • the frequency regions are coded with dotted and solid lines according to which sub- amplifier is driven, i.e. operated or active, in the low amplitude range, e.g. the first amplitude range.
  • An ideal relative phase of the second signal, illustrated by the dotted and solid lines, differs somewhat from that achieved by the simple
  • the difference is less than 0.25 radians for the worst frequency in the central frequency region, which should affect efficiency only marginally.
  • the insertion point is not used, i.e. the second signal is inserted at the same point as the first signal, would give zero radians offset, which in turn would give a maximum deviation of about one radian from the ideal phase.
  • the insertion point provides an improvement.
  • a 90 degree phase shift and a quarter wavelength offset insertion point for the middle part of the range may be used. It is also possible to have a filter with a response that tracks the phase shift over frequency. Adding phase adjustment means for removing the residual phase difference is yet another possibility.
  • the combination of the first and second signal may be applied for infinite bandwidths by using resistive combiners. This solution is preferred for low power level. If loss of the resistive combiners is considered to be too high, for example if the combination of the first and second signal is applied at higher power levels, a Wilkinson combiner may be used.
  • Figure 4 depicts an embodiment of the power amplifier 100.
  • the power amplifier 100 is extended by adding a stage, i.e. by adding a sub-branch (lowest sub-branch in Figure 4) including a third sub-amplifier 113.
  • the modified output network 130 comprises only of transmission lines, including a third output transmission line 133, directly coupled to the output port 140.
  • the third output transmission line 132 is connected to the third sub- amplifier 1 13.
  • the input network 120 may comprise a third input transmission line 123, connected to the second input branch 102 (not shown in Figure 4). Electrical length of the third input transmission line 123 may be twice as long as electrical length of the third output transmission line 133.
  • the power amplifier 100 of Figure 4 includes a fourth attenuator 164. One attenuator from among the first, second and fourth attenuator 161 , 162, 164 is activated at a time. Thus, the other attenuators are blocking when one is active.
  • Figure 4 illustrates combiners 181 , 182, 183 to be used for insertion of the second signal into respective sub-branches at respective amplitude ranges.
  • Each of the respective sub-branches may comprise intermediate
  • the electrical length of the intermediate transmission lines 171 , 172, 173 may be calculated as a dummy length X (same for all) minus the lengths of the line from the amplifier to the output and the input transmission line lengths.
  • the dummy length X is chosen so that no intermediate transmission line gets a negative length.
  • Figure 4 shows a 3-stage power amplifier with a 9: 1 bandwidth of high efficiency.
  • the electrical lengths of the first, second and third output transmission lines 131 , 132, 133 in the output network have electrical lengths of a whole, a half and a quarter wavelength at center frequency, respectively.
  • the addition of the second signal in its sub-branch is at a point 2 wavelengths at center frequency from the common split point of the first signal.
  • the first input transmission line 121 has an electrical length of 2 wavelengths at center frequency in this example.
  • the maximum deviation from the ideal phase is then a little more than twice as much as in the two-stage example of Figure 1. In this case, it may be beneficial to add means for phase adjustment or phase tracking filters as outlined above.
  • Figure 5a an exemplifying configuration of the power amplifier 100 of
  • Figure 4 is shown.
  • the first output transmission line 131 has an electrical length of one wavelength at center frequency
  • the second output transmission line 132 has an electrical length of one half wavelength at center frequency
  • the third output transmission line 133 has an electrical length of one quarter wavelength at center frequency.
  • the first sub-amplifier 11 1 is plotted with a dotted line
  • the second sub-amplifier 1 12 is plotted with a solid line
  • the third sub-amplifier 113 is plotted with a dashed line.
  • Figure 5b the average efficiency with assumed ideal class B operation for a 7 dB PAR Rayleigh signal amplitude distribution is shown.
  • the first sub-amplifier 1 11 is operated in four intervals (dotted lines)
  • the second sub-amplifier 112 is operated in two intervals (solid lines)
  • the third sub-amplifier 1 13 is operated in only one interval (dashed line) at an intermediate frequency of the operational bandwidth.
  • phase of the second signal relative to the in-phase combining phase, i.e. the first signal, of the same sub-amplifier is illustrated.
  • the frequency regions are coded as above according to which sub-amplifier is driven in the first amplitude range.
  • Figure 6a shows an exemplifying configuration of the power amplifier 100 in which the second and third sub-amplifier are connected to a common output transmission line of length 1 ,25 wavelengths at center frequency.
  • the second and third sub-amplifiers are also connected to respective transmission lines of length one half wavelength and one quarter wavelength, respectively.
  • the deviations are greater than for the cases illustrated in Figure 3d and 5d.
  • the solid thin black lines represent the phase shift over frequency, based in this case on the lengths to the nearest common junction.
  • the first sub-amplifier 11 1 in the three quarter wavelength line sub-branch closely follows the think black line (distinguishable in the Figure), since it is coupled directly to the output.
  • the other two sub-amplifiers 1 12, 1 13 are more off, and possibly needs one of the aforementioned phase compensations.
  • Figure 7 further embodiments are illustrated.
  • One embodiment variant is to use only some of the frequency intervals illustrated in Figures 3b, 5b and 6b.
  • One example is a single or dual-band implementation. In those cases not all of the attenuators and combiners of the full implementation are needed.
  • the input side, just after the insertion of the first signal but before the insertion point may also sometimes be made shorter. This is because only the transmission line lengths that precede the insertion point for the actually used intervals are needed. If the intervals associated with the longest transmission line on the output side, i.e. after the insertion point, are not used, i.e. the one wavelength line in this example, the line preceding its insertion point may be disregarded
  • the class C function may be implemented directly in the sub- amplifiers 1 11 , 1 12 and/or 1 13 for sub-branches that do not need the second signal. In such cases, the amplified input signal is fed directly to the sub-amplifiers 1 11 , 1 12 and/or 1 13 without prior class C processing. Combinations of final stage class C biasing and class C pre-processing are also feasible.
  • Another variant that is to have a common transmission line that is split into lines 121 ,122, 123 at a point within the length of the shortest of these, i.e. in this case 122.
  • An advantage may be that the splitter may be made electrically long without adding extra length, e.g. Wilkinson variants.
  • a fifth attenuator 165 may be connected to the second input transmission line 122 and the second sub- amplifier 1 12.
  • the fifth attenuator 165 may be set to match insertion loss of the combiner 181 in the first sub-branch 105.
  • a sixth attenuator 166 may be connected to the third input transmission line 123 and the third sub-amplifier 1 13.
  • the sixth attenuator 166 may also be set to match insertion loss of the combiner 181 in the first sub-branch 105.
  • the transition point amplitude is fairly consistent over the design frequency band or bands.
  • the gate bias setting and attenuator setting, etc. can then be fixed in production, by a priori design or by trimming.
  • Figure 8 shows an exemplifying radio network node 200.
  • radio network node may refer to is a piece of equipment that facilitates wireless communication between user equipment (UE) and a network. Accordingly, the term “radio network node” may refer to a Base Station (BS), a Base Transceiver Station (BTS), a Radio Base Station (RBS), a NodeB in so called Third Generation (3G) networks, evolved Node B, eNodeB or eNB in Long Term Evolution (LTE) networks, or the like.
  • BS Base Station
  • BTS Base Transceiver Station
  • RBS Radio Base Station
  • NodeB in so called Third Generation (3G) networks
  • evolved Node B, eNodeB or eNB in Long Term Evolution (LTE) networks or the like.
  • UTRAN UMTS Terrestrial Radio Access Network
  • radio network node may also refer to a Radio Network Controller.
  • GSM Global System for Mobile Communications
  • GERAN EDGE Radio Access Network
  • BSC Base Station Controller
  • the radio network node 200 comprises a power amplifier 210, e.g. the power amplifier 100, according to the embodiments described above.
  • the radio network node 200 may comprise a processing circuit 220 and/or a memory 230.
  • the radio network node 200 may further comprise additional transceiver circuitry (not shown) for facilitating transmission and reception of data, e.g. in the form of radio signals.
  • Figure 9 shows an exemplifying user equipment 300.
  • the term "user equipment” may refer to a mobile phone, a cellular phone, a Personal Digital Assistant (PDA) equipped with radio
  • PDA Personal Digital Assistant
  • the sensor may be any kind of weather sensor, such as wind, temperature, air pressure, humidity etc.
  • the sensor may be a light sensor, an electronic switch, a microphone, a loudspeaker, a camera sensor etc.
  • the user equipment 300 comprises a power amplifier 310, e.g. the power amplifier 100, according to the embodiments described above.
  • the user equipment 300 may comprise a processing circuit 320 and/or a memory 330.
  • processing circuit and/or a memory 330.
  • the user equipment 300 may further comprise additional transceiver circuitry (not shown) for facilitating transmission and reception of data, e.g. in the form of radio signals.
  • processing circuit may be a processing unit, a processor, an application specific integrated circuit (ASIC), a field-programmable gate array (FPGA) or the like.
  • ASIC application specific integrated circuit
  • FPGA field-programmable gate array
  • a processor, an ASIC, an FPGA or the like may comprise one or more processor kernels.
  • the processing circuit may be embodied by a software or hardware module. Any such module may be a determining means, estimating means, capturing means, associating means, comparing means, identification means, selecting means, receiving means, transmitting means or the like as disclosed herein.
  • the expression “means” may be a unit, such as a determining unit, selecting unit, etc.
  • memory may refer to a hard disk, a magnetic storage medium, a portable computer diskette or disc, flash memory, random access memory (RAM) or the like. Furthermore, the term “memory” may refer to an internal register memory of a processor or the like.
  • number may be any kind of digit, such as binary, real, imaginary or rational number or the like. Moreover, “number”, “value” may be one or more characters, such as a letter or a string of letters, “number”, “value” may also be represented by a bit string.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Amplifiers (AREA)

Abstract

La présente invention concerne un amplificateur de puissance (100) comprenant un premier et un second sous-amplificateur (111, 12) pour l'amplification d'un signal d'entrée en un signal de sortie. Un réseau d'entrée (120) de l'amplificateur de puissance (100) comprend une première dérivation d'entrée (101) conçu pour fournir un premier signal, dérivé du signal d'entrée. Un taux moyen de changement d'amplitude du premier signal par rapport au changement d'amplitude du signal d'entrée dans une première plage d'amplitude est inférieur à un taux moyen de changement d'amplitude du premier signal par rapport au changement d'amplitude du signal d'entrée dans une seconde plage d'amplitude. Le réseau d'entrée (120) comprend en outre une seconde dérivation d'entrée (102) conçu pour fournir un second signal dérivé du signal d'entrée. Le second signal présente une enveloppe d'amplitude de forme triangulaire qui augmente dans la première plage d'amplitude et qui diminue dans la seconde plage d'amplitude. L'amplificateur de puissance (100) est conçu pour fournir le premier signal aux premier et second sous-amplificateurs (111, 112), et fournir le second signal au premier sous-amplificateur (111).
PCT/SE2014/051155 2014-10-06 2014-10-06 Amplificateur de puissance pour l'amplification d'un signal d'entrée en un signal de sortie WO2016056951A1 (fr)

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US10389309B2 (en) 2014-10-06 2019-08-20 Telefonaktiebolaget Lm Ericsson (Publ) Amplifier circuit and method
US10554176B2 (en) 2015-10-27 2020-02-04 Telefonaktiebolaget Lm Ericsson (Publ) Distributed power amplifiers

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Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US10389309B2 (en) 2014-10-06 2019-08-20 Telefonaktiebolaget Lm Ericsson (Publ) Amplifier circuit and method
US10554176B2 (en) 2015-10-27 2020-02-04 Telefonaktiebolaget Lm Ericsson (Publ) Distributed power amplifiers

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