WO2016019354A1 - Partially synchronized multilateration/trilateration method and system for positional finding using rf - Google Patents

Partially synchronized multilateration/trilateration method and system for positional finding using rf Download PDF

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Publication number
WO2016019354A1
WO2016019354A1 PCT/US2015/043321 US2015043321W WO2016019354A1 WO 2016019354 A1 WO2016019354 A1 WO 2016019354A1 US 2015043321 W US2015043321 W US 2015043321W WO 2016019354 A1 WO2016019354 A1 WO 2016019354A1
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WO
WIPO (PCT)
Prior art keywords
signal
location
reference signals
signals
management unit
Prior art date
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PCT/US2015/043321
Other languages
French (fr)
Inventor
Felix Markhovsky
Truman Prevatt
Russ Markhovsky
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Invisitrack, Inc.
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
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Publication date
Priority to US15/501,169 priority Critical patent/US10281557B2/en
Priority to KR1020177005271A priority patent/KR102166578B1/en
Priority to CN201580053072.5A priority patent/CN106922219B/en
Priority to EP15827815.0A priority patent/EP3175668A4/en
Priority to JP2017505477A priority patent/JP6557849B2/en
Application filed by Invisitrack, Inc. filed Critical Invisitrack, Inc.
Priority to US14/923,299 priority patent/US9913244B2/en
Publication of WO2016019354A1 publication Critical patent/WO2016019354A1/en
Priority to US15/442,277 priority patent/US10117218B2/en
Priority to US15/595,702 priority patent/US11835639B2/en
Priority to US15/900,654 priority patent/US11131744B2/en
Priority to US16/164,724 priority patent/US10863313B2/en
Priority to US16/367,014 priority patent/US10845453B2/en
Priority to US16/389,827 priority patent/US11474188B2/en
Priority to US16/398,121 priority patent/US10873830B2/en
Priority to US16/734,205 priority patent/US11125850B2/en
Priority to US17/090,397 priority patent/US11395105B2/en
Priority to US17/090,247 priority patent/US11375341B2/en
Priority to US17/090,486 priority patent/US11388554B2/en
Priority to US17/837,944 priority patent/US11917493B2/en

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Classifications

    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S5/00—Position-fixing by co-ordinating two or more direction or position line determinations; Position-fixing by co-ordinating two or more distance determinations
    • G01S5/02—Position-fixing by co-ordinating two or more direction or position line determinations; Position-fixing by co-ordinating two or more distance determinations using radio waves
    • G01S5/0205—Details
    • G01S5/0221—Receivers
    • G01S5/02213—Receivers arranged in a network for determining the position of a transmitter
    • G01S5/02216—Timing or synchronisation of the receivers
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S1/00—Beacons or beacon systems transmitting signals having a characteristic or characteristics capable of being detected by non-directional receivers and defining directions, positions, or position lines fixed relatively to the beacon transmitters; Receivers co-operating therewith
    • G01S1/02—Beacons or beacon systems transmitting signals having a characteristic or characteristics capable of being detected by non-directional receivers and defining directions, positions, or position lines fixed relatively to the beacon transmitters; Receivers co-operating therewith using radio waves
    • G01S1/08—Systems for determining direction or position line
    • G01S1/20—Systems for determining direction or position line using a comparison of transit time of synchronised signals transmitted from non-directional antennas or antenna systems spaced apart, i.e. path-difference systems
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S5/00—Position-fixing by co-ordinating two or more direction or position line determinations; Position-fixing by co-ordinating two or more distance determinations
    • G01S5/02—Position-fixing by co-ordinating two or more direction or position line determinations; Position-fixing by co-ordinating two or more distance determinations using radio waves
    • G01S5/0205—Details
    • G01S5/021—Calibration, monitoring or correction
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S5/00—Position-fixing by co-ordinating two or more direction or position line determinations; Position-fixing by co-ordinating two or more distance determinations
    • G01S5/02—Position-fixing by co-ordinating two or more direction or position line determinations; Position-fixing by co-ordinating two or more distance determinations using radio waves
    • G01S5/0205—Details
    • G01S5/0218—Multipath in signal reception
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04W—WIRELESS COMMUNICATION NETWORKS
    • H04W64/00—Locating users or terminals or network equipment for network management purposes, e.g. mobility management

Definitions

  • Patent Application No. 12/502,809 filed on July 14, 2009, now U.S. Patent No. 7,872,583, issued January 8, 2011, entitled METHODS AND SYSTEM FOR REDUCED
  • the present embodiment relates to wireless communications and wireless networks systems and systems for a Radio Frequency (RF)-based identification, tracking and locating of objects, including RTLS (Real Time Locating Service) and LTE based locating services.
  • RF Radio Frequency
  • RE-based identification and location-finding systems for determination of relative or geographic position of objects are generally used for tracking single objects or groups of objects, as well as for tracking individuals.
  • Conventional location-finding systems have been used for position determination in an open, outdoor environment.
  • RF-based, Global Positioning System (GPS)/Global Navigation Satellite System (GNSS), and assisted GPSs/GNSSs are typically used.
  • GPS Global Positioning System
  • GNSS Global Navigation Satellite System
  • assisted GPSs/GNSSs are typically used.
  • conventional location-finding systems suffer from certain inaccuracies when locating the objects in closed (i.e., indoor) environments, as well, as outdoors.
  • Cellular wireless communication systems provide various methods of locating user equipment (UE) position indoors and in environments that are not well suited for GPS. The most accurate methods are positioning techniques that are based on the
  • LTE Long Term Evolution
  • U-TDOA Uplink Time Difference of Arrival
  • the fundamental requirement for multiiateration/triiateration based systems is the complete and precise time synchronization of the system to a single common reference time, in cellular networks, the DL- OTDOA and the U-TDOA locating methods also require, in the case of DL-OTDOA, that transmissions from multiple antennas be time synchronized, or in the case of U-TDOA, thai multiple receivers be time synchronized.
  • FCC directive NG 91.1 specifies locate accuracy requirements of 50 meters and 100 meters.
  • LBS Location Based Services
  • the indoors location requirements are much more stringent - 3 meters @67% reliability.
  • the ranging and locate error introduced by the time synchronization error of 1 0 ns is much larger than the 3 meters ranging error (standard deviation of 1.0 ns).
  • the present disclosure relates to methods and systems for Radio Frequency (RF)-based identification, tracking and locating of objects, including Real Time Locating Service (RTLS) systems that substantially obviate one or more of the disadvantages associated with existing systems.
  • RF Radio Frequency
  • RTLS Real Time Locating Service
  • the methods and systems can use partially synchronized (in time) receivers and/or transmitters.
  • RF-based tracking and locating is
  • the proposed system can use software implemented digital signal processing an ! software defined radio technologies (SDR). Digital signal processing (DSP) can be used as well.
  • SDR software implemented digital signal processing
  • DSP Digital signal processing
  • One approach described herein employs clusters of receivers and/or transmitters precisely time synchronized within each cluster, while the inter-cluster time synchronization can be much less accurate or not required at all.
  • the present embodiment can be used in all wireless systems/networks and include simplex, half duplex and full duplex modes of operation.
  • the embodiment described below operates with wireless networks that employ various modulation types, including OFDM modulation and/or its derivatives.
  • the embodiment described below operates with LTE networks and it is also applicable to other wireless systems/networks.
  • RF-based .tracking and locating is
  • the proposed system can use software- and/or hardware-implemented digital signal processing.
  • FIG. 1 and FIG. 1 A illustrate narrow bandwidth ranging signal frequency components, in accordance with an embodiment:
  • FIG. 2 illustrates exemplary wide bandwidth ranging signal frequency components
  • FIG. 3 A, FIG. 3B and FIG. 3C illustrate block diagrams of master and slave units of an RF mobile tracking and locating system, in accordance with an embodiment
  • FIG . 4 illustrates an embodiment synthesized wideband base band ranging signal
  • FIG, 5 illustrates elimination of signal precursor by cancellation, in accordance with an embodiment
  • FIG. 6 Illustrates precursor cancellation with fewer carriers, in accordance with an embodiment
  • FIG. 7 illustrates an embodiment of one-way transfer function phase
  • F3CJ- 8 illustrates an embod iment of a location method
  • FIG. 9 illustrates LIE reference signals mapping
  • FIG. 10 illustrates an embodiment of an enhanced Cell ID + RTT locating technique
  • FIG. 1 1 illustrates an embodiment of an OTDOA locating technique
  • FIG. 1.2 illustrates the operation of a Time Observation Unit (TMO) installed at an operator's e B facility, in accordance with an embodiment
  • FIG. 13 illustrates an embodiment of a wireless network locate equipment diagram
  • FIG. 14 illustrates an embodiment of a wireless network locate downlink ecosystem for enterprise applications
  • FIG. 15 illustrates an embodiment of a wireless network locate downlink ecosystem for network wide applications
  • FIG. 16 illustrates an embodiment of a wireless network locate uplink ecosystem for enterprise applications
  • FIG. 17 illustrates an embodiment of a wireless network locale uplink
  • FIG. 18 illustrates an embodiment of an UL-TDOA environment that may include one or more DAS and/or femto/small cell antennas;
  • FIG. 19 illustrates an embodiment of an UL-TDOA like that of FIG. 1.8 that ma include one or more cell towers that can be used in lieu of DAS base stations and/or femto/small ceils;
  • FIG. 20 illustrates an embodiment of cell level locating
  • FIG. 21 illustrates an embodiment of serving ceil and sector ID locating
  • FIG. 22 illustrates an embodiment of E-O ' D plus AoA locating
  • FIG. 23 illustrates an embodiment of AoA locating
  • FIG. 24 illustrates an embodiment of TDOA with wide and close distances between receiving antenna
  • FIG, 25 illustrates an embodiment of a three sector deployment
  • FIG. 26 illustrates an embodiment of antenna potts mapping
  • FIG. 27 illustrates an embodiment of an LTE Release 11 U-TDOA locating technique
  • FIG. 28 illustrates an embodiment of a multichannel Location Management Unit (LMU) high level block diagram
  • FIG. 29 illustrates an embodiment of a DL-OTDOA technique in
  • FIG. 30 illustrates an embodiment of a U-TDOA technique in wireless/cellular network with a location Server
  • FIG. 31 illustrates an embodiment of a depiction of a rackmount enclosure
  • FIG. 32 illustrates an embodiment of a high level block diagram of multiple single channel LMUs clustered (integrated) in a rackmount enclosure
  • FIG. 33 illustrates an embodiment of a high level block diagram of multiple small cells with integrated LMU clustered (integrated) in a rackmount enclosure (one-to-one antenna connection mapping);
  • FIG. 34 illustrates an embodiment of a high level block diagram of LMUs and DAS integration.
  • FIG. 35 illusirates an embodiment of a high level block diagram of LMUs and WiFi infrastructure integration.
  • the present embodiments relate to a method and system for RF-based identification, tracking and locating of objects, including RTLS.
  • the method and system employs a narrow' bandwidth ranging signal.
  • the embodiment operates in VHF band, but can be also used in HF, LF and VLF bands as well as UHF band and higher frequencies. It employs multi-path mitigation processor. Employing multi-path mitigation processor increases the accuracy of tracking and locating implemented by a system.
  • the embodiment includes small, highly portable base units that allow users to track, locate and monitor multiple persons and objects.
  • Each unit has its own ID.
  • Each unit broadcasts an RF signal with its ID, and each unit is able to send back a return signal, which can include its ID as well as voice, data and additional information.
  • Each unit processes the returned signals from the other units and, depending on the triangulation or trilateration and/or other methods used, continuously determines their relative and/or actual locations.
  • the preferred embodiment can also be easily integrated with products such as GPS devices, smart phones, two- way radios and PDAs. The resulting product will have all of the functions of the stand-alone devices while leveraging the existing display, sensors (such as altimeters, GPS, accelerometers and compasses) and processing capacity of its host.
  • a GPS device with the device technology describe herein will be able to provide the user's location on a map as well as to ma the locations of the other members of the group.
  • the size of the preferred embodiment based on an FPGA implementation is between approximately 2x4x1 inches and 2x2x0.5 inches, or smaller, as integrated circuit technology improves.
  • the antenna will be either integrated into the device or protrude through the device enclosure.
  • An ASIC (Application Specific Integrated Circuit) based version of the device will be able to incorporate the functions of the FPGA and most of the other electronic components in the unit or Tag.
  • the ASIC -based standalone version of the product will result in the device size of 1x0.5x0.5 inches or smaller.
  • the antenna size will be determined by the frequency used and part of the antenna can be integrated into the enclosure.
  • the ASIC based embodiment is designed to be integrated into products can consist of nothing more than a chipset.
  • the devices can use standard system components (off-the-shelf components) operating at multiple frequency ranges (bands) for processing of multi-path mitigation algorithms.
  • the software for digital signal processing and software-defined radio can be used.
  • the signal processing software combined with minimal hardware, allows assembling the radios that have transmitted and received waveforms defined by the software.
  • J0057J U.S. Patent No, 7,561 ,048 discloses a narrow-bandwidth ranging signal system, whereby ike narrow -bandwidth ranging signal is designed to fit into a low-bandwidth channel, for example using voice channels that are only several kilohertz wide (though some of low- bandwidth channels may extend into a few tens of kilohertz). This is in contrast to conventional location- finding systems that use channels from hundreds of kilohertz to tens of megahertz w ide.
  • 7,561,048 can be successfully deployed on LF, VLF and other bands because its ranging signal bandwidth is far below the carrier frequency value; 2) at lower end of RF spectrum (some VLF, LF, HF and VHF bands), e.g., up to UHF band, conventional location-finding systems cannot be used because the FCC severely limits the allowable channel bandwidth (12 - 25 kHz), which makes it impossible to use conventional ranging signals. Unlike conventional location- finding systems, the narrow-bandwidth ranging signal system's ranging signal bandwidth is fully compliant with FCC regulations and other international spectrum regulatory bodies; and 3) it is well known (see MRT. the basics, by Ray H. Hashemi, William G. Bradley ...
  • a narrow-bandwidth signal has inherently higher SNR (Signal-to-Noise-Ratio) as compared to a wide-bandwidth signal. This increases the operating range of the narrow-bandwidth ranging signal location-finding system independently of the frequency/band it operates, including UHF band .
  • the narrow-bandwidth ranging signal location- finding system can be deployed on lower end of the RF spectrum - for example VHF and lower frequencies bands, down to LF/VLF bands, where the multipath phenomena is less pronounced.
  • the narrow-bandwidth ranging location-finding system can be also deployed on UHF band and beyond, improving the ranging signal SNR and, as a result, increasing the location-finding system operating range. JOO60f To mi imize multipath, e.g., RF energy reflections, it is desirable to operate on VLF/LP bands.
  • RF frequencies/bands may be used, for example HP, VHF, UHF and UWB.
  • the noise level from natural and manmade sources is significantly lower compared to VLF, LF and HF bands; and at VHF and HF frequencies the multi-path phenomena (e.g., RF energy reflections) is less severe than at UHF and higher frequencies.
  • the antenna efficiency is significantly better, than on HF and lower frequencies, and at VHF the RF penetration capabilities are much better than at UHF.
  • the VHF band provides a good compromise for mobile/portable applications.
  • the UHF can be a good choice.
  • the narrow-bandwidth ranging signal system will have advantages over the conventional wide-bandwidth ranging signal location-finding systems.
  • Narrow bandwidth ranging allows the user to either receive licenses or receive exemption from licenses, or use unlicensed bands as set forth in the FCC because narrow band ranging allows for operation on many different bandwidths/frequencies, including the most stringent narrow bandwidibs: 6.25kpi3 ⁇ 4, 1 1.25kHz, 12.5kHz, 25kHz and 50kHz set forth in the FCC and comply with the corresponding technical requirements for the appropriate sections. As a result, multiple FCC sections and exemptions within such sections will be applicable.
  • the primary FCC Regulations thai are applicable are: 47 CFR Part 90- Private Land Mobile Radio Services, 47 CFR Part 94 personal Radio Services, 47 CFR Part 15 - Radio Frequency Devices, (By comparison, a wideband signal in this context is from several hundred KHz up to 10-20 MHz.)
  • VHP implementations allow the user to operate the device np to lOOmW under certain exemptions (Low Power Radio Service bein an example).
  • the allowable transmitted power at VHF band is between 2 and 5 Watts.
  • UHF a d
  • the allowable transmitted power is I Watt
  • Narrow band ranging can comply with many if not all of the different spectrum allowances and allows for accurate ranging while still complying with the most stringent regulatory requirements. This holds true not just for the FCC, but for other international organizations that regulate the. use of spectrum throughout the world, including Europe, Japan and Korea.
  • the proposed system works at VHF frequencies and employs a proprietary method for sending and processing the RF signals. More specifically, it uses DSP techniques and software-defined radio (SDR) to overcome the limitations of the narrow bandwidth requirements at VHF frequencies.
  • SDR software-defined radio
  • Battery consumption is a function of design, tran smitted power and the duty cycle of the device, e.g., the time interval between two consecutive distance (location) measurements.
  • the duty cycle is large, l OX to S 000X.
  • an FPGA version that franstniis 100 mW of power will have an up time of approximately three weeks.
  • An ASIC based version is expected to increase the up time by 10X.
  • ASICs have inherently lower noise level.
  • the ASIC -based version may also increase the operating range by about 40%.
  • tracking and location systems employ Track-Locate-Navigate methods. These methods include Time-Of-Arrival (TO A), Differential-Time-Of-Arrival (DTOA) and combination of TOA and DTOA. Time-Of-Arrival (TO A) as the distance measurement technique is generally described in U.S. Patent No. 5,525,967. A . TOA/DTOA- based system measures the RF ranging signal Direct-Line-Of-Site (DLOS) iirae-of-Hight, e.g., time-delay, which is then converted to a distance range.
  • DLOS Direct-Line-Of-Site
  • the embodiment advantageously uses the multi-path mitigation processor to separate the DLOS signal and reflected signals.
  • the embodiment significantly lowers the error in the estimated ranging signal DLOS time-of-flight.
  • the proposed multi-path mitigation method can be used on all RF bands. It can also be used with wide bandwidth ranging signal location-finding systems. And it can support various modulation/demodulation techniques, including Spread Spectrum techniques, such as DSS (Direct Spread Spectrum) and FH
  • noise reduction methods can be applied in order to further improve the method's accuracy.
  • These noise reduction methods can include, but are not limited to, coherent summing, non-coherent summing, Matched filtering, temporal diversity techniques, etc.
  • the remnants of the multi-path interference error can be further reduced by applying the postprocessing techniques, such as, maximum likelihood estimation (like.g., Viterbi Algorithm), minimal variance estimation (Kalman Filter), etc. [0 ⁇ 75
  • the embodiment cm be used m systems with, simplex, half-duplex and full duplex modes of operation. Full-duplex operation is very demanding in terms of complexity, cost and logistics on the RF transceiver, which limits the system operating range in
  • the reader (often referred to as the "master") and the tags (sometimes also referred to as “slaves” or “targets") are controlled by a protocol that only allows the master or the slave to transmit at any given time.
  • the narrow bandwidth ranging signal multi-path mitigation processor does not increase the ranging signal bandwidth. It uses different frequency components, advantageously, to allow propagation of a narrow bandwidth ranging signal.
  • Further ranging signal processing can be carried out in the frequency domain by way of employing super resolution spectrum estimation algorithms (MUSIC, rootMUSIC, ESPRIT) and/or statistical algorithms like RELAX, or in time-domain by assembling a synthetic ranging signal with a relatively large bandwidth and applying a further processing to this signal.
  • the different frequency component of narrow bandwidth ranging signal can be pseudo randomly selected, it can also be contiguous or spaced apart in frequency, and it can have uniform and/or non-uniform spacing in frequency.
  • the embodiment expands multipath mitigation technology.
  • the signal model for the narrowband ranging is a complex exponential (as introduced elsewhere in this document) whose frequency is directly proportional to the delay defined by the range plus similar terms whose delay is defined by the time delay related to the multipath.
  • the model is independent of the actual implementation of the signal structure, e.g., stepped frequency, Linear Frequency Modulation, etc.
  • the frequency separation between the direct path and multipath is nominally extremely small and normal frequency domain processing is not sufficient to estimate the direc path, range.
  • a stepped frequency ranging signal at a lOO Hz stepping rate over 5 MHz at a. range of 30 meters (100.07 nanoseconds delay) results in a frequency of 0.062875 radians/sec.
  • a multipath reflection with a path length of 35 meters would result in a frequency of 0.073355.
  • the separation is 0.0104792.
  • Frequency resolution of the 50 sample observable has a native frequency resolution of 0.12566 Hz. Consequently it is not possible to use conventional .frequency estimation techniques for the separation of the direct path from the reflected path and accurately estimate the direct path range.
  • the embodiments use a unique combination of implementations of subspace decomposition high resolution spectral estimation methodologies and multimodal cluster analysis.
  • the subspace decomposition technology relies on breaking the estimated covariance matrix of the observed data into two orthogonal subspaces, the noise subspace and the signal subspace.
  • the theory behind the subspace decomposition methodology is that the projection of the obser able onto the noise subspace consists of only the noise and the projection of the observable onto the signal subspace consists of only the signal.
  • the super resolution spectrum estimation algorithms and RELAX algorithm are capable of distinguishing closely placed frequencies (sinusoids) in spectrum in presence of noise.
  • the frequencies do not have to be harmonically related and, unlike the Digital Fourier Transfonn (DFT), the signal model does not introduce any artificial periodicity.
  • DFT Digital Fourier Transfonn
  • these algorithms provide significantly higher resolution than Fourier Transform.
  • DLOS Direct Line Of Sight
  • MP multi-paths
  • applying the thresholded method which will be explained later, to the artificially produced synthetic wider bandwidth ranging signal makes it possible to reliably distinguish DLOS from other paths with high accuracy.
  • the Digital signal processing can be employed by the multi-path mitigation processor to reliably distinguish the DLOS from other MP paths.
  • the noted super-resolution algorithms work on the premise that the signals impinging on the antennas are not fully correlated. Thus, the performance degrades severely in a highly correlated signal environment as may be encountered in multipath propagation.
  • Multipath mitigation techniques may involve a preprocessing scheme called spatial smoothing. As a result, the multipath mitigation process may become computationally intensive, complicated, i.e., increases the complexity of the system implementation. Multipath mitigation with lower system computational costs and implementation complexity may be achieved hy using the super - resolution Matrix Pencil (MP) algorithm.
  • MP Matrix Pencil
  • the MP algorithm is classified as a non-search procedure. Therefore, it is computationally less complicated and eliminates problems encountered in search procedures used in other super-resolution algorithms.
  • the MP algorithm is not sensitive to correlated signals and only requires a single channel estimate and can also estimate the delays associated with coherent multipath components.
  • the incoming (i.e., received) signal is modeled as a linear combination of complex exponentials and their complex amplitudes of frequencies;
  • the received signal will he as follows:
  • Parameters a K and ⁇ ⁇ are random time-variant functions reflecting motions of people and equipment in and around buildings. However, since the rate of their variations is very slow as compared to the measurement time interval, these parameters can be treated as time-invariant random variables within a given measurement cycle.
  • Equation ( I ⁇ can be presented in frequency domain as:
  • the reader (often referred to as the "master") and the tags (also referred to as “slaves” or “targets”) are controlled by a protocol that only allows the master or the slave to transmit at any given time.
  • the tags serve as Transponders.
  • the tags receive the ranging signal from a reader (master device), store it in the memory and then, after certain time (delay), re-transmit the signal back to the master.
  • FIG. 1 and FIG. 1A An example of rang ing signal is shown in FIG. 1 and FIG. 1A.
  • the exemplary ranging signal employs different frequency components that are contiguous.
  • Other waveforms, including pseudo random, spaced in frequency and/or time or orthogonal, etc. can be also used for as long as the ranging signal bandwidth remains narrow.
  • the time duration T f for every frequency component is long enough to obtain the ranging signal narrow-bandwidth property.
  • FIG. 2 Another variation of a ranging signal with different frequency components is shown on FIG. 2. It includes multiple frequencies (ft, f>, ⁇ £*, Q transmitted over long period of time to make individual frequencies narrow-band. Such signal is more efficient, but it occupies in a wide bandwidth and a wide bandwidth ranging signal impacts the SNR, which, in turn, reduces the operating range. Also, such wide bandwidth ranging signal will violate FCC requirements on the VHF band or lower frequencies bands. However, in certain applications this wide-bandwidth ranging signal allows an easier integration into existing signal and transmission protocols. Also, such a signal decreases the track-locate time.
  • These multiple-frequency (fi, f>, 3 ⁇ 4, 3 ⁇ 4, f detox) bursts may be also contiguous and/or pseudo random, spaced in frequency and/or time or orthogonal, etc.
  • the narrowband ranging mode will produce the accuracy in the form of instantaneous wide band ranging while increasing the range at wfiich this accuracy can be realized, compared to wide band ranging.
  • This performance is achieved because at a fixed transmit power, the SNR (in the appropriate signal bandwidths) at the receiver of the narrow band ranging signal is greater man the SNR at the receiver of a wideband ranging signal.
  • the SNR gain is on the order of the ratio of the total bandwidth of the wideband ranging signal and the bandwidth of each channel of the narrow band ranging signal. This provides a good trade-off when very rapid ranging is not required, e.g., for stationary and slow-moving targets, such as a person walking or running.
  • [OlOOf Master devices and Tag devices are identical and can operate either in Master or Transponder mode. All devices include data/remote control communication channels. The devices can exchange the information and master device(s) can remotely control tag devices.
  • multi-path mitigation processor originates the ranging signal to tag(s) arid, after a certain delay, the.
  • master/reader receives the repeated ranging signal from the tag(s).
  • master's multi-path mitigation, processor compares the received ranging signal with the one that was originally sent from the master and determines the A ( f n ) estimates in form of an amplitude and a phase for every frequency component f n .
  • a ( , ) is defined for one-w ay ranging signal trip.
  • the . ranging signal makes a round-trip, in other words, it travels both ways: from a master/reader to a target/slave and from the target/slave back to the master/reader.
  • this round-trip signal complex amplitude, which is received back by the master cart be calculated as follows:
  • a complex amplitude determination is based on
  • the phase values Z A RT (/ n ) are obtained by comparing the received by a reader/master returned baseband ranging signal phase and the original (i.e., sent by reader/master) base band ranging signal phase.
  • master and tag devices have independent clock systems a detailed explanation of devices operation is augmented by analysis of the clock accuracy impac t on the phase estimation error.
  • values are directly obtainable from target slave device. However ; . the one-way phase Z A ⁇ f n ) values cannot be measured directly.
  • FIG. 3A depicts block diagrams of a master or a slave unit (tag) of an RF mobile tracking and locating system
  • Fosc refers to the frequency of the device system clock (crystal oscillator 20 in FIG. 3A). All frequencies generated within the device are generated from this system clock crystal oscillator.
  • M is a master device (unit);
  • AM is a tag (target) device (unit).
  • the tag device is operating in the transponder mode and is referred to as transponder (AM) unit.
  • the device consists of the RF front-end and the RF back-end, base-band and the multi-path mitigation processor.
  • the RF back-end, base-band and the multi-path mitigation processor are implemented in the FPGA 150 (see FIGs. 3B and 3C).
  • the system clock generator 20 (see FIG. 3A) oscillates at; F yS(;: - 20 MHz; or
  • the base band ranging signal is generated in digital format by the master' FPGA 150, blocks 155 - 180 (see FIG. 2B). It consists of two frequency components each containing multiple periods of cosine or sine waves of different frequency.
  • t 0
  • the FPGA 150 in a master device (FIG, 38) outputs the digital base-band ranging signal to its up-converter 50 via I/Q DACs 120 and 125.
  • the FPGA 150 starts with F l frequency and after time T start generating F 2 frequency for time duration of
  • the master' base-band I/Q DAC(s) 120 and 125 outputs are as follows:
  • the master (M) and the transponder (AM) work in a half-duplex mode.
  • Master's RF front-end up-converts the base-band ranging signal, generated by the multi-path mitigation processor, using quadrature up-converter (i.e., mixer) 50 and transmits this up- converted signal.
  • quadrature up-converter i.e., mixer
  • the master switches from TX to RX mode using RF Front-end TX/RX Switch 15.
  • the transponder receives and down-converts the received signal back using its RF Front-end mixer 85 (producing First IF) and ADC 140
  • this second IF signal is digitally filtered in the Transponder RF back- end processor using digital filters 190 and further down-converted to the base-band ranging signal using the RF back-end quadrature mixer 200, digital I/Q filters 210 and 230, a digital quadrature oscillator 220 and a summer 270.
  • This base-band ranging signal is stored in the transponder's memory 170 using Ram Data Bus Controller 195 and control logic 180.
  • the transponder switches from RX to TX mode using RF front- end switch 15 and after certain delay t RTX begins re-transmitting the stored base-band signal.
  • the master receives the transponder transmission and down-converts the received signal back to the base-band signal using its RF back-end quadrature mixer 200, the digital I and Q fi iters 210 and 230, the digital quadrature oscillator 220 (see FIG. 3C).
  • the master calculates the phase difference between /' and , in the received (i.e., recovered) base-band signal using multi-path mitigation processor arctan block 250 and phase compare block 255.
  • the amplitude values are derived from the RF back-end RSSS block 240, [01141
  • the multi-path mitigation processor calculates amplitude and phase difference estimates for many time instances over the ranging signal frequency component duration ( .?' f ⁇ . These values, when averaged, improve SNR.
  • the SNR improvement can be in an order thai is proportional to , where N is a number of instances when amplitude and phase difference values were taken (i.e., determined).
  • 3 ⁇ 4 00 r" x 3 ⁇ 4 ⁇ x C ) f y / ⁇ ⁇ ] ⁇ ⁇ ;
  • phase at master's DAC(s) 1.20 and 125 ontpnts are as follows;
  • DACs 120 and 125 have internal propagation delay, t sc , that does not depend upon the system clock.
  • the transmitter circuitry components 15, 30, 40 and 50 will introduce additional delay, t l x , that does not depend uport the system clock.
  • phase of the transmitted RF signal by the master can be calculated as follows;
  • the #> Ml,LT values depend upon. the transmitted frequencies, e.g. I d P .
  • the transponder (AM) receiver' is not able to resolve each path because of limited (i.e., narrow) bandwidth, of the RF portion of the receiver. Thus, after a certain time, for example, 1
  • ⁇ 3 ⁇ 4 ;.' : xi3 ⁇ 4 c , xfc;(i;? M ) - 3 ⁇ 4. (/ ⁇ - - c-' ) + ⁇ -- ⁇ 4 ⁇ + ⁇ ⁇ + ⁇ 3 ⁇ 4 )+ «P& A
  • an output e.g. first IF
  • the phase of the signal is as follows:
  • the propagation delay 4 in the receiver RF section does not depend upon the system clock.
  • the first IF signal is sampled by the RF Back-end ADC 140.
  • ADC 140 is under-sampling the input signal (e.g., first IF).
  • the ADC also acts like a down-converter producing the second IF.
  • the first IF filters, amplifiers and the ADC add propagation delay time.
  • the second IF signal (from roe ADC output) is filtered by the RF Back-end digital filters 190 and further down-converted back to base-band ranging signal by the third down-converter (i.e., quadrature mixer 200, digital filters 230 and 210 and digital quadratare oscillator 220), summed in the summer 270 and is stored in the memoiy 170.
  • the third down-converter output i.e., quadrature mixer
  • propagation delay f f nR ⁇ -A in the FIR section 190 does not depend upon the system clock.
  • the signal from transponder goes through the same down- conversion process as in the transponder receiver. The result is the recovered base-band ranging signal that was originally sent by the master.
  • T n is the propagation delay through roaster (M) and transponder (AM) circuitry.
  • T ls u and j:V/ are propagation delays through the master (M) and transponder
  • the initial phase value 2 x ⁇ p ' :JLT can be assumed to be equal zero because the subspace algorithms are not sensitive to a constant phase offset. If necessary, the 2 ⁇ ⁇ 1 value (phase initial value) can be found by determining the TOA (Time Of Arrival) using the narrow-bandwidth ranging signal method as described in U.S. Patent- No.7,561,048, incorporated herein by reference in its entirety. This method estimates the ranging signal round trip delay, which is equal to 2 x ⁇ ⁇ ' and the 2 ⁇ ⁇ p Ji:i value can be found from the following equation:
  • the returned base-band ranging signal phase values , ⁇ f( o ( ) are calculated by the multi-path processor's arctan block 250.
  • the multi-path mitigation processor phase compare block 255 calculates
  • each component is consists of multiple periods of a sinusoid, it is also possible to estimate the phase and the amplitude of the received ranging signal by sampling the received base-band signal individual component frequency with the corresponding original (i.e., sent by the master) base-band signal individual frequency component and integrating the resulting signal over period T ⁇ ⁇ .
  • This operation generates complex amplitude values i ( ; ) of recei ed ranging signal in the I/Q format. Note that each base-band signal individual frequency component that was sent by the master has to be shifted in time b the 7 , is .. w .
  • the integration operation produces effect of averaging out the multiple instances of the amplitude and the phase (e.g., increasing the SNR). Note that the phase and the amplitude values can be translated from
  • the ranging signal bandwidth is narrow, the frequency difference / admir - /, can be relatively large, for example, in an order of several megahertz. As a result, the receiver's bandwidth has to be kept wide enough to pass all of the . j : f n ranging signal frequencies components. This wide receiver bandwidth impacts the SNR.
  • the received ranging signal base-band frequency components can be filtered by the RF back-end processor in FPGA. 150 by the digital narrow bandwidth filters tuned for each individual frequency component of the received baseband ranging signal.
  • this large number of digital filters puts additional burden on the FPGA resources, increasing its cost, size and power consumption.
  • the master and the transponder units are capable of synchronizing clocks with any of the devices.
  • a master device can serve as a reference. Clock synchronization is accomplished by using the remote control
  • the frequency of temperature compensated crystal oscillator TCXO 20 is adjusted.
  • the frequency difference is measured at the output of the summer 270 of the master device while the selected transponder device is transmitting a carrier signal. [0152 j Thereafter, the master sends a command io the transponder to increase/decrease TCXO frequency. This procedure may be repeated several times to achieve greater accuracy by minimizing frequency at the summer 270 output. Please note that in an ideal case the frequency at the summer 270 output should become equal to zero.
  • An alternative method is to measure the frequency difference and make a correction of the estimated phase without adjusting the transponder * TCXO frequency.
  • TCXO components have high degree of accuracy and stability. Specifically, TCXO components for the GPS commercial applications are very accurate. With these devices, the phase error impact on locating accuracy can be less than one meter without the need for frequent clock synchronization.
  • narrow bandwidth ranging signal multi-path mitigation processor After narrow bandwidth ranging signal multi-path mitigation processor obtains the returned narrow bandwidth ranging signal complex amplitude A RT (j ls ) , the further processing (i.e., execution of super-resolution algorithms), is implemented in the software-based component, which is a part of the multi-path mitigation processor.
  • This software component can be implemented in the master (reader) host computer CPU and/or the microprocessor that is embedded in the FPO A 1 50 (not shown).
  • the multi-path mitigation aigorithm(.s) software component is executed by the master host computer CPU.
  • the multi-path mitigation processor selects t with the smallest value (i.e., the DLOS delay time),
  • the DLOS path can be separated from MP paths by employing a continuous (in time) chirp.
  • this continuous chirp is Linear Frequency
  • each recei ved single chirp is aligned so thai the returned chirp is from the middle of the area of interest
  • the relumed signal (dip) is sii - r) .
  • the multi-path mi ligation processor then "deramps" the ,v(/- r) by performing complex, conjugate mix with the originally transmitted chirp.
  • the resulting signal is a complex sinusoid:
  • f r ⁇ /) exp( ⁇ r)exp ⁇ ⁇ 2 ⁇ )& ⁇ ⁇ . ⁇ 2 ) , (8) where ⁇ ⁇ ( ⁇ % ⁇ is the amplitude and 2 ⁇ is the frequency and 0 ⁇ t ⁇ l Note thai the last terra is a phase and it is negligible.
  • the composi te decamped signal consists of multiple complex sinusoids :
  • L is the number of ranging signal paths, including the DLOS path and 0 ⁇ i ⁇ l
  • the number of samples can be a multiple of N, e.g. ⁇ xN;a 1,2,
  • the multi-path mitigation processor produces a ⁇ complex amplitude samples in time domain that are used in further processing (i.e., execution of super-resolution algorithms).
  • This further processing is implemented in the software component, which is a part of the multi-path mitigation processor.
  • This software component can be executed by the master (reader) host computer CPU and/or by the
  • the multi-path mitigation algorithm(s) software is executed by the master host computer CPU.
  • the super-resolution algorithm(s) produce estimation of 2 ⁇ ⁇ ; "frequencies", e.g. T K values.
  • the multi-path mitigation processor selects ⁇ with the smallest value, i.e. the DLOS delay time,
  • threshold technique which can serve as an alternative to the super-resolution algorithms, in other words, it is used to enhance reliability and accuracy in distinguishing DLOS path from other MP paths using the artificially generated synthetic wider bandwidth ranging signal.
  • the frequency domain base-band ranging signal shown in FIG. 1 and FIG, 1 A can be
  • FIG, 4 shows two periods of ' ⁇ ' for the case where ⁇ 11 and ⁇ > ⁇ 250 kHz.
  • the bandwidth extends from zero frequency to ' ::: 2.75 MHz.
  • the basic idea of the thresholded method that is used hi the preferred embodiment is to enhance the artificially generated synthetic wider bandwidth ranging reliability and accuracy in distinguishing DLOS path from other MP paths.
  • the threshold method detects when the start of the leading edge of a wideband pulse arrives at a receiver. Because of filtering in the transmitter and receiver, the leading edge does not rise instantaneously, but rises out of the noise with smoothly increasing slope.
  • the TOA of the leading edge is measured by detecting when the leading edge crosses a predetermined threshold T.
  • a small threshold is desirable because it gets crossed sooner and the error delay ⁇ between the true start of the pulse and the threshold crossing is small.
  • any pulse replica arriving due to multi-path has no effect if the start of the replic a having a delay greater than r .
  • the presence of noise places a limit on how small the threshold T can be.
  • One way to decrease the delay ⁇ is to use the derivative of the received pulse instead of the pulse itself, because the derivative rises faster. The second derivative has an even faster rise. Higher order derivatives might be used, but in practice they can raise the noise level to an unacceptable value, so the thresholded second derivative is used.
  • the 2.75 Mffcs wide signal depicted in FIG. 4 has a fairly wide bandwidth, it is not suitable for measuring range by the abovementioned method. That method requires transmitted pulses each having a zero-signal precursor. However, it is possible to achieve that goal by modifying the signal so that the sinusoidal waveform between the pulses is essentially cancelled out. In the preferred embodiment it is done by constructing a waveform which closely approximates the signal on a chosen interval between the tall pulses, and then subtracting it from the original signal.
  • the technique can be illustrated by applying it to the signal in FIG. 1.
  • the two black dots shown on the waveform are the endpoints of an interval I centered between the first two pulses.
  • the left and right endpoints of the interval I which have been experimentally determined to provide the best results, are respectively at:
  • h (t) is ⁇ generated by the following sum;
  • FIG. 5 shows the resulting signal r(t) for the original signal s(t) shown in FIG. 1, where N - 11. In this case the construction of r(i) requires 25 carriers (including the DC terra 3 ⁇ 4 ).
  • Aii carriers are cosine functions (including DC) spaced Af apart, except for one carrier, which is a sine function located at frequency ( ⁇ ? )4/ ,
  • the length of the cancelled portion of s (t ) is about 3.7 microseconds or 1 , 1 10 meters. This is more than enough to eliminate any residual signal from previous non-zero portions of r (t) due to the multi-path.
  • the main peak has value of approximately 35, and the largest magnitude in the precursor (i.e., cancellation) region is about 0.02, which is 65 dB below the main peak. This is desirable for getting good performance using the TOA measurement thresholded technique as described above.
  • the period of the signal is only 2 Af ⁇ 2,35 microseconds as compared, to the signal in FIG. 5, where the period is 8 microseconds. Since this example has more periods per unit time, one might expect that more processing gain could be achieved.
  • the amplitude of the main peak is about 1/3 as large as before, which tends to cancel the expected extra processing gain.
  • the length of the zero-signal precursor segments is shorter, about 0.8 microseconds or 240 meters. This should still be enough to eliminate any residual signal from previous non-zero portions of r ( t) due to the multi-path.
  • the total bandwidth of (2N ' + 1 ⁇ ⁇ / ⁇ 5.95 MHz is about the same as before, and that the width of the half-cycle of the main peak is also roughly the same. Since fewer carriers are used, there should be some extra processing gain when each carrier is narrowband filtered at the receiver.
  • the largest magnitude in the precursor (i.e., cancellation) region is now about 75 dB below the main peak, a 10 dB improvement from the previous example,
  • ⁇ n94 Transmission at RF Frequencies ⁇ to this point r (t) has been described as a base-band signal for purposes of simplicity. However, it can be translated up to RF, transmitted, received, and then reconstituted as a base-band signal at the receiver. To illustrate, consider what happens to one of the frequency components ⁇ & k in the base-band signal r (/) traveling via one of the multi-path propagation paths having index j (radian/sec frequencies are used for notational simplicity):
  • the parameter b k is the k m coefficient in expression (21) for r(f) .
  • the parameters r and ⁇ are respectively the path delay and phase shift (due to dielectric properties of a reflector) of the h propagation path.
  • the parameter ⁇ is the phase shift occurring in the down-conversion to baseband in the receiver.
  • This method is not restricted to 1 -millisecond transmissions, and the length of the transmissions may be increased or decreased. However, the total time for all transmissions should be short enough to freeze any motion of the receiver or transmitter.
  • the transmitter can then transmit all frequencies with the same amplitude, which simplifies its design. It should be noted, that this method also weights the noise at each frequency, the effect of which should be considered . It should also be noted that coefficient weighting should be done at the receiver in order to effect the polarity reversal of g (?) to get twice as many useable main peaks,
  • F Radio Frequency
  • the ranging signal(s) are subject to the extensive sophisticated signal processing techniques, including the multi-path mitigation.
  • a narrow bandwidth ranging signal or base-band narrow bandwidth ranging signal several individual frequency components are modulated with the identical data/control signals and in case of voice with digitized voice packets data.
  • the indmdual frequency components that have the highest signal strength are demodulated and the obtained information reliability may be further enhanced by performing ''voting" or other signal processing techniques that utilize the information red undancy .
  • This method allows to avoid the "null" phenomena, wherein the incoming RF signals from multiple paths are destmctively combining with the DLOS path and each other, thus significantly reducing the received signal strength and associated with it SNR, Moreover, such method allows to find a set of frequencies at which the incoming signals from multiple paths are constructively combining with DLOS path am! each other, thus increasing the received signal strength and associated with it SNR.
  • the model size estimation is accomplished using an "F" statistic.
  • F the singular value decomposition of the estimate of the covariance matrix (with forward/backward correlation smoothing) is ordered in ascending order. Thereafter, a division is made whereby the (n ⁇ 1.) eigenvalue is divided by the n-th eigenvalue. This ratio is an "F” random variable. The worst case is an "F” random variable of (1 ,1) degree of freedom. The 95% confidence interval for a "F” random variable with (1,1) degrees of freedom is 161. Setting that value as a threshold determines the model size. Note also that for the noise subspace, the eigenvalues represent an estimate of the noise power.
  • the frequencies that are estimated by super-resolution algorithms are artificial frequencies (equation 2). In fact, these frequencies are individual paths delays of the multipath environment. As a result, there should be no negative frequencies and all negative frequencies that are produced by a super-resolution algorithm are spurious frequencies to be rejected.
  • the ranging signal makes a round-trip. In other words, it travels both ways: from a master/reader to a target/slave and from the target/slave back to the master/reader:
  • [0224 ⁇ Master transmits a tone: x e " , where ⁇ is an operating frequency in the operating band and a is the tone signal amplitude.
  • the received signal (one-way) is as follows:
  • N is number of signal paths in the multipath environment; . K0 and ' ⁇ are amplitude and time-of-flig ' ht of the DLOS signal; , ⁇ ⁇ > > t! , ari ⁇ ** 0 can be positive or negative.
  • N paths delays includes combinations of these paths delays, for example:
  • the one-way amplitude values are directlv obtainable from target/slave device.
  • the owe-way phase values ' "" " / « ⁇ ' ' cannot be measured directly. It is possible to determine the phase of the one- way from the roundtrip phase measurements observation: ⁇ K - * e
  • ( ⁇ ) be a one-way function that generates ⁇ l ⁇ ) on the interval ( 6 ⁇ , m ⁇ t and assume that has at least one zero on * - ⁇ .
  • the zeros or ' will separate ( ⁇ ⁇ , ⁇ 2 ) - nto ⁇ fi n it e number of abutting open frequency intervals J L , J, , ... , J Vogel .
  • the solution ⁇ ( ⁇ ) or ⁇ ( ⁇ ) wu j De f oun( j using either Theorem 1 or Theorem 3.
  • H ( "' ( ⁇ W j ) or -//' ⁇ "' (es/ j ) according to whether our solution in z is H( «> ⁇ or -H ( ⁇ 2>) . Since H : (fi>, ) ⁇ 0 , the two limits will be equal if and only if the solutions in J x and J, are both ⁇ m) or both -H( «>) . If the left and right hand limits are unequal we invert the solution in subinterval J . Otherwise, we don't.
  • the super (high) resolution spectral estimation methods are designed to distinguish closely-placed frequencies in the spectrum and used for estimating the individual frequencies of multiple harmonic signals, e.g., paths delays. As a result, path delays can be accurately estimate .
  • the super resolution spectral estimation makes use of the eigen-structure of the covariance matrix of the baseband ranging signal samples and covariance matrix intrinsic properties to provide a solution to an underlying estimation of individual frequencies, e.g. paths delays.
  • One of the eigen-structure properties is thai the eigenvalues can. be combined and consequently divided into orthogonal noise and signal eigenvectors, aka subspaces.
  • Another eigen-structure property is the rotation-invariant signal subspaces property.
  • the subspace decomposition technology (MUSIC, rootMUSIC, ESPRIT, etc.) relies on breaking the estimated covariance matrix of the observed data into two orthogonal subspaces, the noise subspace and the signal subspace.
  • the theory behind the subspace decomposition methodology is that the projection of the observable onto the noise subspace consists of only the noise and the projection of the observable onto the signal subspace consists of only the signal
  • the spectral estimation methods assume that signals are narrow-band, and the number of harmonic signals is also known, i.e., the size of the signal subspace needs to be known.
  • the size of the signal subspace is called as the model size.
  • it cannot be known in any detail and can change rapidly - particularly indoors - as the environment changes.
  • One of the most difficult and subtle issues when applying any subspace decomposition algorithm is the dimension of the signal subspace that can be taken as the number of frequency components present, and which is the number multipath reflections plus the direct path. Because of real-w orld measurement imperfections there always will be an error in the model size estimation, which in turn will result in loss of accuracy of frequencies estimation, i.e., distances.
  • one embodiment includes six features that advance the state of the art in the methodology of subspace decomposition high resolution estimation. Included is combining two or more algorithms estimating individual frequencies by using different eigen-structure properties that further reduces the delay path determination ambiguity.
  • [0279J Root Music finds the individual frequencies, that when the observable is projected onto the noise subspace, minimizes the energy of the projection.
  • the Esprit algorithm determines the individual frequencies from the rotation operator. And in many respects this operation is the conjugate of Music in that it finds the frequencies thai, when the observable is projected onto the signal subspace, maximizes the energ of the projection.
  • the model size is the key to both of these algorithms, and in practice, in a complex signal environment such as seen in indoor ranging - the model size which provides the best performance for Music nd Esprit are in general not equal, for reasons that will be discussed below,
  • [0 811 for Music it is preferable to err on the side of identifying a basis element of the decomposition as a "signal eigen value" (Type I Error). This will minimize the amount of signal energy that is projected on the noise subspace and improve the accuracy.
  • a signal eigen value Type I Error
  • the first feature of the embodiment is the use of the F-statistic to estimate the model size (see above) .
  • the second feature is the use of different Type I Error probabilities in the F-statistic for Music and Esprit. This implements the Type I Error differences between Music and Esprit as discussed above.
  • the third feature is the use of a base model size and a window in order to maximize the probability of detecting the direct path.
  • the fourth feature of the embodiment is the use of multiple measurements.
  • the probability distribution of the multiple answers resulting from multiple measurements, each using multiple model sizes from both a Music and Esprit implementation, will be multimodal. Conventional cluster analysis will not be sufficient for this application.
  • the fifth feature is the development of multimodal cluster analysis to estimate the direct range and equivalent range of the reflected muitipath components.
  • the sixth feature is the analysis of the statistics of the range estimates provided by the cluster analysis (range and standard deviation and combing those estimates that are statistically
  • the present embodiments relate to a positioning/locating method m wireless communication and other wireless networks that substantially obviate one or more of the disadvantages of the related art.
  • the present embodiments advantageoosl improve the accuracy of tracking and locating functionality in multiple types of wireless network by utilizing rauiii- path mitigation processes, techniques and algorithms, described in U.S.
  • Wireless Personal Area Networks such as ZigBee and Blue Tooth
  • WLAN wireless local area network
  • WMAN Wireless Metropolitan Area Networks
  • W Max being the primary example
  • W AN wireless Wide Area Networks
  • W AN Wireless Wide Area Networks
  • MDN Mobile Devices Networks
  • wireless network typically comprised of a combination of devices, including base stations, desktop, tablet and laptop computers, handsets, smartphones, actuators, dedicated tags, sensors as well as other communication and data devices (generally, all these devices are referred to as "wireless network, devices").
  • Cell ID location technique allows estimating the position of the user (UE- User Equipment) with the accuracy of the particular sector coverage area.
  • the attainable accuracy depends on the cell (base station) sectoring scheme and antenna beam-width.
  • RTT Random Trip Time
  • the RTT constitutes the difference between
  • DPDCH Downlink Physical Channel
  • DPCCH Dedicated Physical Data Channel/Dedicated Physical Control Channel
  • the abovementioned frame(s) act as a ranging signal.
  • the distance from eNB can be calculated (see Figure 10). [02981 ⁇ the Observed Time Difference of Arrival (OTDOA) technique the time of arrival, of the signal earning from neighboring base stations (eNB) is calculated
  • OTDOA Observed Time Difference of Arrival
  • the UE position can be estimated in the handset (UE-based method) or in the network (NT-based, UE-assisted method) once the signals from three base stations are received.
  • the measured signal is the CPICH (Common Pilot Channel).
  • the propagation time of signals is correlated with a locally generated replica.
  • the peak of correlation indicates the observed time of propagation of the measured signal.
  • Time difference of arrival values between two base stations determines a hyperbola. At least three reference points are needed to define two hyperbolas. The location of the UE is in the intersection of these two hyperbolas (see Figure i i).
  • Idle Period Downlink is further OTDOA enhancement.
  • the OTDOA- IPDL technique is based on the same measurements as the regular OTDOA Time measurements are taken during idle periods, in which serving eNB ceases its transmissions and allows the UE within the coverage of this cell to hear pilots coming from distant eNB(s).
  • Serving eNB provides idle periods in continuous or burst mode. In the continuous mode, one idle period is inserted in every downlink physical frame (10 ms). In the burst mode, idle periods occur in a pseudorandom way. Further improvement is obtained via Time Aligned IPDL (TA-IPDL). Time alignment creates a common idle period, during which, each base station will either cease its transmission or transmit the common pilot.
  • TA-IPDL Time Aligned IPDL
  • the pilot signal measurements will occur in idle period.
  • One Multi-path mitigation, technique uses detections/measurements from excess number of e B(s) or Radio Base Stations (RBS). The minimum is three, but for multipath mitigation the number of RBS's required is at least six to eight (see METHOD AND
  • ⁇ he probability of .an UE hearing from this large number of eNB(s) is much lower than from three eNB(s). This is because with large number of RBS (eNBs) there will be several ones that are far away from the UE and the received signal from these RBS (es) may fall below the UE receiver sensitivity level or the received signal will have low SNR.
  • 3 G TS 25.515 v.3.0.0 (199-10) standards define the RTT as ".... the difference between transmission of a downlink DPCH frame (signal) and the reception of the beginning (first significant path) of the corresponding uplink DPCCH/DPDCH frame (signal) from UE".
  • the standard does not define what constitutes this "first significant path”.
  • the standard goes on noting that "The definition of the first significant path needs further elaboration". For example, in heavy multipath environment it is a common occurrence whereby the DLOS signal, which is the first significant path, is severely attenuated (10 dB - 20 dB) relatively to one or more reflected signal(s).
  • the "first significant path" is detennined by measuring the signal strength, it may be one of the reflected signal(s) and not the DLOS signal. This will result in erroneous TOA/DTOA/RTT measurement s) and loss of locating accuracy.
  • the embodiment can be used in all wireless networks that employ reference and/or pilot signals, and/ or synchronization signals, including simplex, half-duplex and full duplex modes of operation.
  • the embodiment operates with wireless networks that employ OFDM modulation and/or its derivatives.
  • the embodiment operates with LTE networks.
  • wireless networks including WiMax, WiFi, and White Space.
  • Other wireless networks that do not use reference and/or pilot or synchronization signals may employ one or more of the following types of alternate modulation embodiments as described in U.S. Patent No, 7,872,583: 1 ) where a portion of frame is dedicated to the ranging signal/ranging signal elements as described in U.S. Patent No. 7,872,583; 2) where the ranging signal elements (U.S. Patent No. 7,872,583) are embedded into transmit/receive signals frame(s); and 3) where the ranging signal elements (described in U.S. Patent No. 7,872,583) are embedded with the data.
  • a smart phone can have Blue Tooth, WiFi, GSM and LTE functionality with the capability of operating on multiple networks at the same time.
  • different wireless networks can be utilized to provide positioning/locating information.
  • the proposed embodiment method and system leverages the wireless network reference/pilot and/ or synchronization signals. Furthermore, the reference/pilot signal/ synchronization signals measurements might be combined with RTT (Round Trip Time) measurements or system timing, According to an embodiment, RF -based tracking and locating is implemented on 3 GPP LTE cellular networks, but could be also implemented on other wireless networks, for example WiMax, Wi-Fi, LTE, sensors networks, etc. that employ a variety of signaling techniques. Both the exemplary and mentioned above alternative embodiments employ multi-path mitigation method/techniques and algorithms that are described in U.S. Patent No. 7,872,583. The proposed system can use software implemented digital signal processing. [033 J 1 The system of the embodiment leverages User Equipment (UE),. e.g.
  • UE User Equipment
  • a base station generally consists of transmitters and receivers in a cabin or cabinet connected to antennas by feeders. These base stations include, Micro Cell, Pico Cell, Macro Cell, Umbrella Cell, Cell Phone towers, Routers and Femtocells. As a result, there will be little or no incremental cost to the UE device and overall system. At the same time the locate accuracy will be significantly improved.
  • the improved accuracy comes from the multipara mitigation that is provided by the present embodiments md U.S. Patent No. 7,872,583.
  • the embodiments use multi-path mitigation algorithms, network reference/ pilot and/ or synchronization signals and network node (eNB). These might be supplemented with RTT (Round Time Trip) measurements.
  • the multi- path mitigation algorithms are implemented in UE and/or base station (eNB), or both: UE and eNB
  • processor/algorithms that allow separating the DLOS signal and reflected signals, even when DLOS signal is significantly attenuated (10 dB - 20 dB lower) relatively to one or more reflected signals.
  • the embodiments significantly lower the error in the estimated ranging signal DLOS time-of-fhght and consequently TOA, RTT and DTOA measurements.
  • the proposed multi-path mitigation and DLOS differentiating (recognizing) method can be used on all RF bands and wireless systems/networks. And it can support various modulation/demodulation techniques, including Spread Spectrum techniques, such as DSS (Direct Spread Spectrum) and FH (Frequency Hopping).
  • noise reduction methods can be applied in order to further improve the method's accuracy.
  • These noise reduction methods can include, but are not limited to, coherent summing, non-coherent summing, Matched filtering, temporal diversity techniques, etc.
  • the remnants of the multi-path interference error can be further reduced by applying the postprocessing techniques, such as, maximum likelihood estimation (e.g., Viterbi Algorithm), minimal variance estimation (Kalman Filter), etc..
  • the multi-path mitigation processor and multi-path mitigation techniques/algorithms do not change the RTT, CPCIH and other signals and/or frames.
  • the present embodiments leverage wireless network reference, pilot and; or
  • the invention uses the channel estimation statistics that is generated by UE and/or eNB (see Iwamatsu et aL APPARATUS FOR ESTIMATING PROPAGATION PATH CHARACTERISTICS, US
  • LTE networks use specific (non-data) reference/ pilot and/ or synchronization s signals (known signals) that are transmitted in every downlink and uplink subframe, and might span entire cell bandwidth.
  • reference/ pilot and synchronization signals are reference signals.
  • An example of the LTE reference signals is in FIG. 9 (these signals are interspersed among LTE resource elements). From FIG. 2, reference signals (symbols) are transmitted every sixth subcarrier. Further, reference signals (symbols) are staggered in both time and frequency. In total, reference signals are covering every third subcarrier.
  • reference signals are used in the initial cell search by the UE, downlink signal strength measurements, scheduling and handover, etc. Included in the reference signals are UE-specific reference signals for channel estimation (response determination) for coherent demodulation. In addition to the UE-specific reference signals, other reference signals may be also used for channel estimation purposes, (see Chen et al., US patent publication No.
  • LTE employs the OFDM (Orthogonal Frequency Division Multiplexing) modulation (technique), in. LTE the 1ST (inter Symbol Interference) caused by multipath is handled by inserting Cyclic prefix (CP) at the beginning of each OFDM symbol The CP provides enough delay so that delayed reflected signals of the previous OFDM symbol will die out before reaching the next OFDM symbol,
  • OFDM Orthogonal Frequency Division Multiplexing
  • 1ST Inter Symbol Interference
  • An OFDM symbol consists of multiple very tightly spaced subcarriers. Inside the OFDM symbol time-staggered copies of the current symbol (caused by multipath.) result in Inter Carrier Interference (ICI). In LTE the ICI is handled (mitigated) by determining the multipath channel response and correcting the channel response in the receiver.
  • ICI Inter Carrier Interference
  • the multipath channel response (estimation) is computed in the receiver from subcarriers bearing the reference symbols. Interpolation is used to estimate the channel response on the remaining subcarriers. The channel response is calculated (estimated) in form of channel amplitude and phase. Once the channel response is determined (by periodic transmission of known reference signals), the channel distortion caused by multipath is mitigated by applying an amplitude and phase shift on a subcarrier-by-subcarrier basis (see Jim Zyren, Overview of the 3 GPP Long Term Evolution Physical Layer, white paper). [03211 LTE multipath mitigation is designed Co remove the ISI (by inserting a Cyclic Prefix) and ICl, but not to separate the DLOS signal from reflected signals.
  • DFT Digital Fourier Transformation
  • DFT technique(s) can resolve (remove) only copies of signal(s) that are delayed for times that are longer than or equal to the time that is inversely proportional to the signal and/or channel bandwidth.
  • This method accuracy may be adequate for an efficient data transfer, but not accurate enough for precise distance measurement in a heavy multipath environment.
  • the signal and receiver channel bandwidths should be wider - one hundred megahertz for three meters.
  • CPJCTi uplink DPCCH DPDCH and other signals that are used in various CELL ID and OTDOA methods techniques, including the RTT measurements, as well as the LTE received signal subcarriers have bandwidths that are significantly lower than ten megahertz.
  • the currently employed (in LTE) method/technique will produce locating errors in the range of 100 meters.
  • the embodiments use a unique combination of implementations of subspace decomposition high resolution spectral estimation methodologies and multimodal cluster analysis.
  • This analysis and related multi-path mitigation method/techniques and algorithms described in U.S. Patent No. 7,872,583, allow a reliable and accurate separation of DLOS path from other reflected signals paths.
  • the ranging signal complex amplitude is the channel response that is calculated (estimated) by the LTE receiver in form of amplitude and phase.
  • the channel response statistics that is calculated (estimated) by the LTE receiver can provide complex amplitude Information that is required by the method/techniques and algorithms described in U.S. Patent No. 7,872,583.
  • phase change of the received signal e.g. channel response phase
  • the phase change of the received signal will be directly proportional to the signal's frequency (a straigh line); and the RF signal time-of-flight
  • phase vs. frequency dependency by computing fast derivative of the phase vs. frequency dependency.
  • the result will be the propagation delay constant.
  • phase and frequency synchronization phase coherency
  • multipath mitigation processor and method(s)/techniques and algorithms described in U.S. Patent No. 7,872,583 will accurately separate DLOS path from odier reflected signals paths and determine this DLOS path length (time-of-ilighi).
  • the amplitude and phase vs. frequency dependency that is computed by the LIE receiver does not include an actual phase value because all amplitude and phase valises are computed from the downlink uplink reference signals, e.g. relative to each other.
  • the amplitude and phase of the channel response that is calculated (estimated) by the LTE receiver needs actual phase value at least at one frequency (subcarrier frequency).
  • this actual phase value can be determined from one or more RTT measurement(s), TOA measurements; or
  • All of the above methods provide the time-of-flight values of one or more reference signals. From the time-of-flight values and frequencies of these reference signals actual phase values at one or more frequencies can be calculated.
  • RTT/TOA/TDOA/OTDOA including DL-OTDOA, U-TDOA, UL- TDOA, etc.
  • measurements can be carried out with the resolution of 5 meters.
  • TOA measurements are carried during dedicated connections. Thus, multiple simultaneous measurements are possible when UE is in handover state and times when UE periodically collects and reports measurements back to the UE, in which the DPCH frames are exchanged between the UE and different networks (base stations). Similar to RTT, TOA measurements provide the signal's time-of-ffight (propagation delay), but TOA measurements cannot be made simultaneously (Jakub Marek Borkowskt: Performance of Cell ID+R1T Hybrid Positioning Method for UMTS).
  • FIG. 1 An example of UE positioning method is shown in FIG. 1.
  • the Cell ID ⁇ RTT track-locate method accuracy is impacted by the multipath (RTT measurements) and the eNB (base station) antenna beamwidth.
  • Base stations antennas beamwidths are between 33 and 65 degrees. These wide beamwidths results in locating error of 50 - 150 meters in urban areas (Jakub Marek Borkowski: Performance of Cell ID+RTT Hybrid Positioning Method for UMTS).
  • the current LTE RTT distance measurement average error is approximately 100 meters
  • the overall expected average locate error of the currently employed by LTE Ceil ID ⁇ RTT method is approximately 150 meters.
  • One of the embodiments is the UE locating based on the AOA method, whereby one or more reference signals from the UE is used for the UE locate purposes, f involves an AOA determination device location for determining the DLOS AOA.
  • the device can be collocated with the base station and/or installed at another one or more locations independent from the base station location. The coordinates of these locations are presumably known. No changes are required on the UE side.
  • This device includes a small antenna arm and is based on a variation of the same multipath mitigation processor, method(s)/teehniqites and algorithms described in U.S. Patent No. 7,872,583.
  • This one possible embodiment has the advantage of precise determination (very narrow beamwidth) of the AO of the DLOS RF energy from, an UE unit,
  • an accurate DLOS AOA determination can be made will greatly improve the Cell ID + RTT track- locate method precision - 10X or greater.
  • Another advantage of this approach is that the UE location can be determined at any moment with a single tower, (does not require placing UE in soft handover mode). Because an accurate location fix can be obtained with a single tower there is no need to synchronize multiple ceil towers.
  • Another option of determining the DLOS AOA is to use the existing eNB antenna array and the eNB equipment. This option may further lower the cost of implementation of the improved Ceil ID + RTT method. However, because eNB antennas are not designed for the locating applications, the positioning accuracy may be degraded. Also, network operators may be unwilling to implement required changes in base station (software/hardware).
  • the U-TDOA/ UL-TDOA are in a study phase; to be standardized in 20 i 1 .
  • 2011/0124347 Al (Method and Apparatus for UE positioning in LTE networks, Chen, at al.).
  • the Release 9 DL-OTDOA suffers from the multipath.
  • Some of the multipath mitigation can be achieved via increased PRS signal bandwidth.
  • the hade-off is increased scheduling comple ity and longer times between UE positions fixes.
  • the best possible accuracy is 100 meters, see Chen, Table 1.
  • Embodiments described herein allow for up t,q 50X ranging/locate accuracy improvement for a given signal bandwidth over the performance achieved by the Release 9 DL- OTDOA method and the UL-PRS method of Chen e al. described in the Background section.
  • applying embodiments of the methods described herein to the Release 9 PRS processing reduces the locale error down to 3 meters or better in 95% of all possible cases .
  • this accuracy gain will reduce the scheduling complexity and the time between HE position fixes.
  • the ranging to the serving cell can be determined from other serving cells' signals, thus improvin the neighboring cells hearability and reducing the.
  • Embodiments also enable the acc uracy of the U-TDOA method and UL-TDOA from. Chen et al (described in the Background) to be improved up to 50 times. Applying
  • embodiments to the Chen's UL-TDOA variant reduces the locate error down to 3 meters or better in 95% of all possible cases. Moreover, this accuracy gain further reduces the scheduling complexity and the time between UE positions fixes.
  • Chen's UL-TDOA method accuracy can be improved up to 5 X,
  • applying the present embodiments io the Chen's U-TDO variant will reduce the locate error down to 3 meters or belter in 95% of all possible cases.
  • this accuracy gain will further reduce the scheduling complexity and the time between UE position fixes.
  • the PRS and/or other signals used in the proces of one-way ranging would be frequency and phase coherent.
  • the OFDM based systems like LTE. ate frequency coherent.
  • the UE units and e B(s) are not phase or time synchronized by a common source - like UTC, to a couple nanoseconds, e.g. there exists a random phase adder.
  • the embodiment of the multipath processor calculates the differential phase between the ranging signal(s), e.g. reference signals, individual components (subcarriers). This eliminates the random phase term adder.
  • the embodiment described below operates with wireless networks that employ OFDM modulation and/or its derivatives; and reference/ pilot/ and or synchronization signals.
  • the embodiment described below operates with LTE networks and it is also applicable to other wireless systems and other wireless networks, including other types of modulation, with or without reference/ pilot/ and/or synchronization signals.
  • wireless networks including WiMax, WiFi, and White Space.
  • Other wireless networks that do not use reference / pilot and/ or synchronization signals may employ one or more of the following types of alternate modulation embodiments as described in U.S. Patent No. 7,872,583: 1 ) where a portion of frame is dedicated to the ranging signal/ranging signal elements; 2) where the ranging signal elements are embedded into transmit receive signals frame(s); and 3) where the ranging signal elements are embedded with the data.
  • Embodiments of the muitipath mitigation range estimation algorithm described herein (also described in U.S. Patent Nos. 7,969,33 ! and 8,305,215) works by providing estimates of the ranges in the ensemble made up of the direct path (DLOS) of a signal plus the mul tipa th reflections.
  • DLOS direct path
  • the LTE DAS system produces multiple copies of the same signal seen at various time offsets to a mobile receiver (UE .
  • the delays are used to uniquely determine geometric relationships between the antennas and the mobile receiver.
  • the signal seen by the receiver resembles that seen in a muitipath environment - except the major "muitipath" components result from the sum of the offset signals from the multiple DAS antennas.
  • the signal ensemble seen by the receiver is identical to the type of signal ensemble embodiments are designed to exploit - except that in this case the major muitipath components are not traditional muitipath.
  • algorithms is capable of determining the attenuation and propagation delay of the DLOS and each path, e.g. reflection, (see equations 1 - 3 and associated descriptions). While muitipath can be present because of the dispersive RF channel (environment), the major muitipath components in this signal ensemble are associated w ith transmissions from multiple antennas. Embodiments of the present muitipath algorithm can estimate these muitipath components, isolate the ranges of the DAS antennas to the receiver, and provide range data to the location processor (implemented in software). Depending on the antenna placing geometry, this solution can provide both X, Y and X, Y, Z location coordinates.
  • present embodiments do not require any hardware and/or new network signals) additions.
  • the positioning accuracy can. be significantly improved by 1) mitigating the multipath and 2) in case of active DAS the lower bound of positioning error can be drastically reduced, such as reducing from approximately 50 meters to approximately 3 meters.
  • the signal propagation delay may be determined automatically, using the loopback techniques, whereby the known signal is sent round trip and this round trip time is measured. This loopback technique also eliminates the signal propagation delay changes (drift) with temperature, time, etc.
  • Pico cells and micro cells further enhance the resolution by providing additional reference points.
  • the embodiment described above of individual range estimates in a signal ensemble of multiple copies from multiple antenna can be further enhanced by changes to the signal transmit structure in the following two ways.
  • the first is to time multiplex the transmissions from each antenna.
  • the second approach is to frequency multiplex for each of the antennas.
  • time and frequency multiplexing simultaneously further improve the ranging and location accuracy of the system.
  • Another approach is to add a propagation delay to each antenna. The delay values would be chosen to be large enough to exceed the delay spread in a particular DAS environment (channel), but smaller than the Cyclic Prefix (CP) length so that the multipath caused by additional delays will not result in ISI (Inter Symbol Interference).
  • CP Cyclic Prefix
  • one or more reference signal(s) subcarriers are used to determine subcarriers phase and amplitude that are in turn applied to the multi-path processor for multipath interference mitigation and generation of range based location observables and locate estimate using multilateration and location consistency algorithms to edit out wild points.
  • the LTE uplink signaling also includes reference signals, mobile devic to base, which also contains reference subcarriers.
  • reference signals mobile devic to base
  • additional UL reference signals might be added in the upcoming and future standard releases.
  • the uplink signal is processed by multiple base units (eNB) using the same range to phase, multipath mitigation processing to generate range related observables.
  • location consistency algorithms are used as established by the multilateration algorithm to edit wild point observables and generate a location estimate.
  • relevant one or more reference (including pilot and/ or synchronization) subearriers of both the LTE d ownlink and LTE uplink are collected, the range to phase mapping is applied, multipath mitigation is applies and the range associated observable is estimated.
  • These data would then be fused in such a way that would provide a more robust set of observables for location using the multilateration algorithm and location consistency algorithms.
  • the advantage would be the redundancy that results in improved accuracy since the downlink and up link two different frequency bands or in case of the TDD (Time Division Duplexing) improving the system coherency.
  • the location consistency algorithm(s) are extended to isolate the ranges of the DAS antennas from observables generated by the multipath mitigation processing from reference signal(s) (including pilot and/ or synchronization) subearriers and to obtain the location estimates from the multiple DAS emitters (antennas) ranges.
  • the InvisiTrack multi-path mitigation methods and systems for object identification and location finding described i U.S. Patent No. 7,872,583, is applied to the range to signal phase mapping, multipath interference mitigation aid process to generate range based location observables utilizing the LTE downlink, uplink and or both (downlink and uplink), one or more reference signal(s) subcarriers and using multilateration and location consistency to generate a location estimate.
  • E-IJTRA Universal Terrestrial Radio Access
  • 3GPP TS 36.21 1 Release 9 technical Specification it has not been implemented by the wireless operators (carriers).
  • a Downlink locating can be implemented within current, e.g. unmodified, LTE network environment by using the existing physical layer measurements operation(s),
  • the LIE and the eNB are required to make physical layer measurements of the radio characteristics.
  • the measurement definitions are specified in 3GPP TS 36.214. These measurements are performed periodically and are reported to the higher layers and are used for a variety of purposes includin intra- and inter-frequency handover, inter-radio access technology (inter-RAT) handover, timing measurements, and other purposes in support of RRM (Radio Resource Management).
  • inter-RAT inter-radio access technology
  • the RSRP Reference Signal Received Power
  • RSSQ combines signal strength as well as interference level
  • the LTE network provides the UE with eNB neighbor (to serving eNB) lists.
  • the (serving) eNodeB Based on the network knowledge configuration, the (serving) eNodeB provides the UE with neighboring eNB's identifiers, etc. The UE then measures the signal quality of the neighbors it can receive. The UE reports results back to the eNodeB. Note: UE also measures the signal quality of die serving eNB.
  • the RSRP is defined as the linear average over the power contributions (in ⁇ W j) of the resource elements that carry cell-specific reference signals within the considered measurement frequency bandwidth.
  • the measurement bandwidth that is used by the UE to determine .RSRP is left up to the UE implementation with the limitation that corresponding measurement accuracy requirements have to be fulfilled.
  • the cell-specific reference signals that are used in the RSRP measurements can be further processed to determine these reference signals subcarriers phase and amplitude that are in turn applied to the multi-path processor for multipath interference mitigation and generation of range based location observables.
  • other reference signals that are used in the RSRP measurement for example SSS (Secondary Synchronization Signal) might be also used.
  • the location fix can be -estimated using muhilateration and location consistency algorithms.
  • the RF Fingerprinting method(s)/ technology locate accuracy is heavily impacted by multipath dynamics - changes over time, environment (for example weather), people and/ or objects movement, including vertical uncertainty. >100% variability depending upon device Z-height and/ or antenna orientation (see Tsung-Han Lin, et al. Microscopic Examination of an RSSI-Signature-Based Indoor Localization System).
  • the present embodiments can significantly improve the RF Fingerprinting locate accuracy because of the ability (multipath processor) to find and characterize each individual path, including significantly attenuated DLOS. As a result, the RF
  • the locate fix will require position references synchronization in time.
  • these position references may include Access Points, Macro/ Mini/ Pico and Femto cells, as wells as so called Small cells (eNB).
  • eNB Small cells
  • wireless operators do not implement the synchronization accuracy that is needed for an accurate position fix.
  • the standard does not require any time synchronization between eNB(s) for the FDD (Frequency Division Duplexing) networks.
  • FDD Frequency Division Duplexing
  • this time synchronization accuracy is limit is +/- 1.5 microseconds. This is equivalent to 400+ meters locate uncertainty.
  • the LTE FDD networks are also synchronized, but use even larger (than 1,5 microseconds) limits.
  • Wireless LTE operators are using GPS/ GNSS signals to synchronize e-NB(s) in frequency and time.
  • the LTE eNB has to maintain a very accurate carrier frequency: 0.05 ppm for macro/ mini cells and slightly less accurate for other type of cells (0.1 - 0.25 ppm).
  • the GPS/ GNSS signals can also enable a required (for locate) time synchronization accuracy of better than 10 nanoseconds.
  • network operators arid network equipment manufacturers are trying to reduce costs associated with the GPS/ GNSS units in favor of Packet Transport/, e.g. Internet/ Ethernet networking time synchronization by employing NTP (Network Time Protocol) and/ or PTP (Precision Time Protocol), for example IEEE 1588v2 PTP.
  • NTP Network Time Protocol
  • PTP Precision Time Protocol
  • IP network based synchronization has a potential of meeting the minimum frequency and time requirements, but is lacking the GPS/ GNSS precision that is needed for locate fix.
  • [03911 T e approach described herein, is based on the GPS/ GNSS signals and signals generated by the eNB and/ or AJP, or other wireless networks equipment, it also can be based on the IP networking synchronization signals and Protocols and signals generated by the eNB and/ or AP, or other wireless networks equipment. This approach is also applicable to other wireless networks, including WiMax, WiFi and White Space.
  • the eNB signals are received by the Time Observation Unit (TMO) installed at the operator's eNB facility ( Figure 12).
  • TMO Time Observation Unit
  • the TMO also include the External Synchronization Source input.
  • the eNB signals are processed by the TMO and are time stamped using clocks that are synchronized with the External Synchronization Source input,
  • the External Synchronization Source could be from the GPS/ GNSS and ' or Internet/ Ethernet networking, for example PTP or NTP, etc.
  • the time-stamped processed signal for example the LTE frame start (could be other signals, especially in other networks), also includes the eNB (cell) location and/ or cell ID, is sent via the Internet/ Ethernet backhaul to a central TMO Server that creates, maintains and updates a data base of ail eNBs,
  • the UE and/ or eNB( s) involved in the process of ranging and obtaining a location fix will quire the TMO Server and the server will, return the time synchronization offsets between the sN B(s) involved. These time synchronization offsets will be used by the UE and/ or eNB(s) involved in the process of obtaining a location fix to adjust the location fix.
  • the location fix calculations and adjustment can be carried out by the TM O Server when UE and/ or eNB(s) involved in the process of ranging will also supply the obtained ranging information to the TMO Server. The TMO Server will then return an accurate (adjusted) position (locale) fix.
  • the RTT (Round Time Trip) measurements can be used for locating.
  • the drawback is that the RTT ranging is subject to multipara which has drastic impact on the locate accuracy .
  • RTT locating does not require the position references synchronization (in time) in general and in case of LTE the eNB in particular.
  • the multipath mitigation processor, method(s)/techniques and algorithms described in U.S. Patent No. 7,872,583 are capable of determining the channel response for the RTF signal(s), e.g. identify die mdtipath channel that the RTT signal(s) are going through. This allows for correctio of the RTT measurements so that the actual DLOS time will be determined.
  • this embodiment creates a parallel wireless locate infrastructure that uses the wireless network signals to obtain location fix.
  • the InvisiTrack's locate infrastructure will consists of one or more wireless Network Signals Acquisition Units (NSAU) and one or more Locale Server Units (LSU) that collect data from NSAU(s) and analyze it. detenn ing range and locations, and to convert it into a table, e.g. of phone UEs IDs and locations at an instant of time.
  • the LSU interfaces to the wireless network via network's API,
  • the coherent timing can be derived from the GPS clock and/ or other stable clock sources,
  • the NSAU communicates with LSU via LAN (Local Area Network), Metro Area Network (MAN) and/ or Internet.
  • LAN Local Area Network
  • MAN Metro Area Network
  • Internet Internet
  • the NSAU and LSU could be combined/ integrated into a single unit.
  • the transmitters are required to be clock and event synchronized to within tight tolerances. Normall this is accomplished by locking to the 1 PPS signs! of GPS. This will result in timing synchronization in a local area to within 3 nanosecond 1-sigma.
  • This present embodiments provide time offset estimates between the downlink transmitters and tracking of the time offsets in order to provide delay compensation values to the location process so the location process can proceed as if the transmitters were clock and event synchronized. This is accomplished by prior knowledge of the transmit antenna (which is required for any location services) and a receiver with known a priori antenna location. This receiver called the synchronization unit will collect data from all the downlink transmitters and given its knowledge of the locations, calculate the offset timing from a preselected base antenna. These offsets are tracked by the system through the use of a tracking algorithm that compensates for clock drifts the downlink transmitters. Note: The processing to derive pseudo ranges from the received data will utilize the InvisiTraek Multipath mitigation algorithms (described in U.S. Patent No. 7,872,583). Hence the synchronization will not be impacted by multipath.
  • LSU Location Server
  • the synchronization receiver and/ or receiver's antennas will be located based on optimal GDOP for best performance. In large installations multiple synchronization receivers can be utilized to provide an equivalent 3 nsec 1-sigma synchronization offset throughout the network. By utilizing synchronization receivers(s) the requirements for synchronization of the downlink transmitters is eliminated.
  • the synchronization receiver imk can be a standalone nnit communicating with the NSAU and/ or LSU. Alternatively this synchronization receive ca be integrated with th NSAU,
  • Uplink mode - uses wireless network Uplink (UL) signals for the purpose of locating (FIGS. 16 and 17 ⁇
  • Downlink mode - uses wireless network Downlink (DL) signals for the purpose of locating (FIGS. 14 and 15).
  • Two-way mode - uses both: UL and DL signals for locating.
  • multiple antennas are connected to one or more NSAUs. These antennae locations are independent from the wireless network antennas; NSAU(s) antennae locations are selected to minimize the GDOP (Geometric Dilution of Precision).
  • Network' RF signals from the UE/ cell phone devices are collected by NSAU(s) antennae and are processed, by NSAU(s) to produce time stamped samples of the processed network' RF signals during a time interval that is adequate for capturing one or more instances of all signals of interest.
  • NSAU will also receive, process and time stamped samples of Downlink signals to obtain additional information, for example for determining UE/ phone ID, etc.
  • the NSAU will periodically supply data to the LSU. If unscheduled data is needed for one or more UE/ cell phone ID(s) then LSU will request additional data. [0422 ⁇ No changes/ modifications will be needed in wireless network infrastructure and/ or existing UE ceil phone for the UL mode operation.
  • the InvisiTrack enabled UE will be required. Also, the cel l phone F W would have to be modified if phone is used to obtain location fix.
  • BBU Base Band Unit
  • the mode the NSAU will process and time stamp processed RF or baseband (when available) wireless network signals. From captured time stamped samples wireless network signals DL frames starts associated with the network antennas will be determined (obtained) and the difference (offset) between these frame starts will be calculated. This operation can be performed either by the NSAU or by the LSU. Frame starts oifsets for network antennas will be stored on the LSU.
  • offsets of network antennas will be sent from LSU to the UE/ phone device in case when the device will process/ determine its own location fix using InvisiTrack technology. Otherwise, when UE/ cell phone device will periodically send time stamped samples of the processed network' RF signals to the LSU, the LSU will determine the device's location fix and will send the location fix data back to the device.
  • the wireless network RF signals will come from one or more wireless network antennae. To avoid multipath impact on results accuracy the RF signal should be sniffed out from the antenna or the antenna connection to the wireless network equipment.
  • the two-way mode encompasses determination of the location fix from both: UL and DL operations. This allows further improve the locate accuracy.
  • Some Enterprise set ups use one or more BBUs feeding one or more Remote Radio Heads (RRH), with each RRH in turn feeding multiple antennae with the same ID.
  • RRH Remote Radio Heads
  • determining the DL mode frame starts offsets of network antennas might not be required.
  • a configuration configuration whereby antennae that are fed .from multiple BBUs are interleaved in the same zone will require determining the DL mode frame starts offsets of network antennas.
  • location consistency algorithm(s) are extended/ developed to isolate the ranges of the DAS antennas from observable* generated by the multipath mitigation processing from reference signal(s) (including pilot and/ of .synchronization) subcarriers and to obtain the location estimates from the multiple DAS emitters (antennas) ranges.
  • BBUs are capable of supporting up to six sectors
  • Adding a propagation delay element to each antenna would be chosen to be large enough to exceed the delay spread in a particular DAS environment (channel), but smaller than the Cyclic Prefix (CP) length so that the multipath caused by additional delays will not result in ISI (Inter Symbol Interference).
  • CP Cyclic Prefix
  • the addition of a unique delay ID for one or more antenna further reduces the number of antennae that emit the same ID.
  • an autonomous system with no Customer Network [0435] In an embodiment, an autonomous system with no Customer Network
  • the system can operate on a band other than the L IE band.
  • iSM industrial Scientific and Medical
  • White Space bands can be used in places where LTE services are not available.
  • the embodiment can be also integrated with the macro/ mini/ pko/ femto station (s) and/ or UE (cell phone) equipment. Although the integration may require Customer Network Investment, it can reduce cost overhead and can dramatically improve the TCO (Total Cost o f O wnersh i.p).
  • PR.S can be used by the UE for the Downlink Observed Time Difference of Arrival (DL-OTDOA) positioning.
  • DL-OTDOA Downlink Observed Time Difference of Arrival
  • the 3GPP TS 36.305 Stage 2 functional specification of User Equipment (UE) positioning in E-UTRAN specifies transferring timing to the UE,. the timing being r elative to an eNode .8 service of candidate cells (e.g., neighboring cells).
  • the 3 GPP TS 36.305 also specifies Physical cell IDs (PC is) and global ceil IDs (GCls) of candidate cells for measurement purposes.
  • PC Physical cell IDs
  • GCls global ceil IDs
  • this information is delivered from the E- MLC (Enhanced Serving Mobile Location Centre) server. It is to be noted that the TS 36.305 does not specify the abovememioned liming accuracy ,
  • the 3GPP TS 36.305 specifies that the UE shall return to the B- MLC the downlink measurements, which includes Reference Signal Time Difference (RSTD) measurements.
  • RSTD Reference Signal Time Difference
  • the RSTD is the measurement taken between a pair ofeNBs (see TS 36.214 Evolved Universal Terrestrial Radio Access (E-UTRA); Physical Layer measurements; Release 9).
  • the measurement is defined as a relative timing difference between a subframe received from the neighboring cell j and a corresponding subframe of the serving cell i. Positioning Reference Signals are used to take these measurements. The results are reported back to the location server that calculates the position.
  • a hybrid method can be defined to accommodate both the newly introduced PRS and the already existing reference signals.
  • the hybrid method can use/operate with PRS, with other reference signals (e.g., ceil or node-specific reference signals (CRS)), or with both signal types.
  • PRS Physical Uplink Control Signal
  • CRS node-specific reference signals
  • Such a hybrid method provides the ad vantage of allowing network operators) to dynamically choose the mode of operation depending on circumstances or network parameters.
  • the PRS have better hearability than CRS, but may result in up to 7% reduction in the data throughput.
  • CRS signals do not cause any throughput reduction.
  • CRS signals are backward compatible with all previous LTE releases, for example Rel- 8 and lower.
  • the hybrid method provides a network operator the ability to trade-off or balance between hearability, throughput, and compatibility.
  • LTE downlink baseband signals (generated by a cell or wireless node and referred to herein as "nodes") are generally combined into downlink frames.
  • a receiver for detecting and receiving such signals may detect downlink frames from multiple cells or nodes (two or more).
  • Each downlink frame includes multiple CRS or reference signals.
  • these reference signals In a Downlink (DL) frame, these reference signals have predetermined positions in time and frequency, e.g., there are deterministic time offsets between the frame start and each CRS in a given frame.
  • each CRS is modulated with a special code. The modulation and the code are also predetermined. The CRS modulation is the same for all nodes, but the code (seed) is determined by the ID (identification) number of the node.
  • the detector may also demodulate/decode the CRS and then correlate the demodulated/decoded CRS with baseband sub-carriers that are assigned to the CRS.
  • the CRS may also be used as rangin signals by the multipath mitigation processor. Therefore, in addition to finding coarse frame starts the detector's correlation process is also capable of isolating the CRS from other signals (such as payload) in the frame using the code that was used to modulate those signals.
  • a system for tracking and locating one or more wireless network devices in communication with a network comprises a user equipment receiver configured to receive multiple signals from two or more nodes in communication with the network, the multiple signals being modulated with a code determined by an identification of each node of the two or more nodes transmitting the multiple signals, the user equipment receiver including a detector configured to detect and isolate reference signals from the multiple signals based on the identification, and a processor configured to use the reference signals as ranging signals from each node for tracking and locating the one or more wireless network devices.
  • the detector is further configured to estimate a course location of frame starts from each node.
  • the detector Is further configured to estimate tSie course location by correlating the reference signals with known replicas of such reference signals
  • the detector is further configured to isolate the reference signals from any other signals in the frame, and wherein the detector is further configured to isolate the reference signals for each node of the two or more nodes.
  • the processor is at least one multipath mitigation processor, and wherein the multipath mitigation processor is configured to receive the course location and isolated reference signals and estimate a relative time of arrival of the ranging signals from each node.
  • processor is at least one multipath mitigation processor
  • the detector is further configured to estimate a course location of frame starts from each node, wherein the detector is configured to isolate the reference signals from any other signals in the frame, wherein the detector is further configured to isolate the reference signals for each node of the two or more nodes, wherein the detector is configured to pass the course location and isolated reference signals for each node to the multipath mitigation processor, and wherein the multipath mitigation processor is configured to receive the course location and isolated reference signals and estimate a relative time of arrival of the ranging signals from each node.
  • the system further comprises an uplink embodiment where a node receiver is configured to receive device signals from the one or more wireless network devices, the device signals being modulated with a device code determined by a device identification of each wireless network device of the one or more wireless network devices transmitting the device signals, the node receiver including a device detector configured to detect and isolate device reference signals from the device signals based on the device identification, and a second processor is configured to use the device reference signals as ranging signals from each wireless network device for tracking and locating the one or more wireless network devices.
  • a system for tracking and locating one or more wireless network devices in communication with a network comprises a user equipment receiver configured to receive multiple signals from two or more nodes in communication with the network, the multi le signals being modulated with a code determined by an identification of each node of the two or more nodes transmitting the multiple signals, and a processor configured to detect, and. isolate reference signals from the multiple signals based on the identification and. to use the reference signals as ranging signals from each node for tracking and locating the one or more wireless network devices.
  • processor is further configured to estimate the course location by correlating the reference signals with known replicas of the reference signals.
  • the processor is further configured to estimate a relative time of arrival of the ranging signals from each node based on the course location and isolated reference signals.
  • the processor is further configured to isolate the reference signals from any other signals in the frame, and wherein the processor is further configured to isolate the reference signals for each node of the two or more nodes.
  • the processor is further configured to estimate a course location of frame starts from each node by correlating the reference signals with known replicas of the reference signals, wherein the processor is further configured to isolate the reference signals from any other signals in the frame and to isolate the reference signals for each node of the two or more nodes, and wherein the processor is further configured to estimate a relative time of arrival of the ranging signals from each node based on the course location and isolated reference signals.
  • a system for tracking and locating one or more wireless network devices in communication with a network comprises a detector configured to receive multiple signals from two or more nodes in communication with the network, the multiple signals being modulated with a code determined by an identification of each node of the two or more nodes transmitting the multiple signals, and to detect and isolate reference signals from the multiple signals based on the identification, and a processor configured to use the reference signals as ranging signals from each node for tracking and locating the one or more wireless network devices.
  • the detector is further configured to estimate the course location by correlating ihe reference signals with known replicas of such reference signals
  • the detector is further configured to isolate the reference signals from any other signals in the frame, and wherein the detector is further configured to isolate the reference signals for each node of the two or more nodes.
  • the processor is at least one multipath mitigation processor, and wherein the multipath mitigation processor is configured to receive the course location and isolated reference signals and estimate a relative time of arrival of the ranging signals from each node.
  • ihe processor is at least one multipath mitigation processor.
  • the detector is further configured to estimate a course location of frame starts from each node, wherein the detector is configured to isolate the reference signals from any other signals in the frame, wherein the detector is further configured to isolate the reference signals for each node of the two or more nodes, wherein the detector is configured to pass the course location and isolated reference signals for each node to the multipath mitigation processor, and wherein the multipath mitigation processor is configured to receive the course location and isolated reference signals and estimate a relative time of arrival of the ranging signals from each node.
  • a system for tracking and locating one or more wireless devices in communication with a network comprises a node receiver configured to receive device signals from the one or more wireless network devices, the device signals being modulated with a device code determined by a device identification of each wireless network device of the one or more wireless network devices transmitting the device signals, the node receiver including a device detector configured to detect and isolate device reference signals from the device signals based on the device identification, and a processor configured to use the device reference signals as ranging signals from each wireless network device for tracking and locating the one or more wireless network devices.
  • the hybrid method can be transparent to the LTE UE positioning architecture. For instance, the hybrid method can operate in the 3GFP TS 36.305 framework.
  • STD can be measured and, according to the 3 GPP TS 36.305, transferred from a UE to an. E-SMLC.
  • FIG. 18 presents an environment that may include one or more DAS and/ or Femto/ Small ceil antennas.
  • each NSAU is equipped with a single antenna. As depicted, at least three NSAUs are required. However, additional NSAUs can be added to improve hearability because each UE must be "heard" by at least three N SAUs.
  • the NSAU ' (s) can ' be configured as receivers. For example, each NSAU receives but does not transmit information over the air. In operation, each NSAU can listen to the wireless Uplink network signals from UEs.
  • Each of the UEs can be a cell phone, a Tag, and/ or another UE device.
  • the NSAUs can be configured to communicate with a Locate Server Unit (LSU) over an interface, such as a wired service or a LAN.
  • LSU Locate Server Unit
  • the LSU can be configured to communicate with a Locate Server Unit (LSU) over an interface, such as a wired service or a LAN.
  • the LSU can be configured to communicate with a Locate Server Unit (LSU) over an interface, such as a wired service or a LAN.
  • LSU Locate Server Unit
  • the communication can be via a network API, where the LSU can, for example, communicate with an E-SMLC of the LTE network and can use a wired service such as a LAN and/or a WAN,
  • the LSU may also communicate directly with DAS base station(s) and or Femto/ Small cells. This communication can use the same or a modified Network API.
  • the Sounding Reference Signal can be used for locate purposes.
  • other signals may also be employed.
  • the NSAUs can convert the UE Uplink transmission signals to a digital format, for exarapie I/Q samples, and can periodically send a number of the converted signals to the LSU with a time stamp.
  • the DAS base station(s) and or Femto/Small cells can pass to the LSU one or all of the following data:
  • SchedutmgRequesiCoufig information and SRS-UL-Config . information..
  • the information passed to the LSU may not be limited by the abovementioned information. It can include any information needed to correlate each UE device uplink signal, such as a UE SRS, with each UE ID.
  • the LSU functionality can include ranging calculations and obtaining the location fix of a UE. These determinations/ calculations can be based on the information passed from the NSAUs, the DAS bases stations, and or Femto/Smali cells to the LSI),
  • the LSU may also determine timing offsets from the available downlink transmission information passed from the NSAUs to the LSU.
  • the LSU can provide the wireless or LIE network with UE location fix and other calculations and data. Such information, can be communicated, via the Network APT.
  • each NSAU may receive, process, and time stamp samples of Downlink signals. Each NSAU may also periodically send a number of such samples to the LSU, including the time stamp(s).
  • each NSAU may include an input configured for synchronization with external signaS(s).
  • FIG. 19 depicts another embodiment of a UL-TDOA.
  • the environment of this embodiment may include one or more cell towers that can be used in lieu of the DAS base stations and/or Femto/Smali cells. Data from the one or more cell towers can be used to obtain the location fix of a UE.
  • an advantage of this embodiment includes obtaining a location fix with only a single cell tower (eNB).
  • this embodiment can be configured to operate in a similar manner as described under FIG. 18, with the exception that one or more eNBs can replace the DAS base stations and/or the Femto/Smali cells.
  • CID Cell Identification method
  • the UE position may be determined on the cell level.
  • This method is purely network based .
  • the UE for example a handset, is not aware of the fact that it is being tracked. While this is a relatively simple method, it lacks accuracy because the locate uncertainty is equal to the cell diameter.
  • any of the handsets 2000 within the cell diameter 2002 of a serving cell tower 2004 effectively have the same location, even though they are not at the same location.
  • the accuracy of the CID method can be improved when combined with serving sector identification (sector ID) knowledge. For example, as illustrated i» FIG.
  • sector ID 2160 idealities a section 2102 within the cell diameter 2662 that ineiiid.es a mnnber of handsets 2104 thai are known to have a different location than the other handsets 2000 in other sectors of the cell diameter 2002.
  • E-CID Enhanced Ceil ID
  • One enhancement uses timing measurements to calculate how far away the UE is from the eNB (the network node). This distance can be calculated as half the round trip time (RTT), or Timing Advance (TA) in LTE (LTE TA), times the speed of light. If the UE is connected, then RTT or TA may be used for distance estimation. In this case both: the serving cell tower or sector and the UE (upon the serving eNB command) will measure the timing difference between Rx sub-frames and Tx sub-frames. The UE will report its measurements to the eNB (also under the eNB control).
  • LTE Rel-9 adds the TA type 2 measurements that rely on the timing advance estimated from receiving a PRACH preamble during the random access procedure.
  • a PRACH (physical/packet random access channel) preamble specifies the maximum number of preambles to be sen t during one PRACH ramping cycle when no response is received from the UE being tracked.
  • the LTE Type 1 TA measurement is the equivalent to the RTT measurement, as follows:
  • the position of the UE can be calculated by the network.
  • j0491f TSie E-CID locating method is still limited, however, because in one dimension the locate accuracy depends upon the sector width and the distance from the serving cell tower, and in the other dimension the error depends upon the TA (RTT) measurement accuracy.
  • the sector width varies with network topology and is impacted by the propagation phenomena, specifically muitipath. Sector accuracy estimates vary from 200 meters to in excess of 500 meters.
  • the LTE TA measurement resolution is 4 Ts, which corresponds to 39 meters of maximum error.
  • the actual error in the LTE TA measurement is even larger, however, due to calibration inaccuracies and the propagation phenomena (muitipath), and may reach as much as 200 meters.
  • the E-CID method may be further improved with the addition of a feature known as Angle of Arrival (AoA).
  • AoA Angle of Arrival
  • the eNB estimates the direction from which the UE is transmitting using a linear array of equally spaced antenna elemen ts 2200.
  • reference signals are used for the AoA determination.
  • the reference signals may be phase rotated, as shown in FIG. 23 by an amount which depends on the AoA, the carrier frequency, and the element spacing.
  • the AoA will require each e B to be equipped with antenna arrays adaptive antennas. It is also exposed to rmihjpath and. topology variances.
  • a solution to the limitations of other uplink location methods involves the use of AoA capabilities without the need for antenna arrays/adaptive antennas.
  • Such an embodiment may employ TDOA (Time Difference of Arrival) location techniques for AoA determination, which may be based on estimating the difference in the arrival times of the signal from the source at multiple receivers.
  • a particular value of the time difference estimate defines a hyperbola between two receivers in communication with a UE.
  • the TDOA is equivalent to the angle between the baseline of the sensors (receivers antennas) and the incident RF energy from the emitter. If the angle between the baseline and true North is known, then the line of bearing (LOB) and or Ao A can be determined.
  • TDOA Time Difference of Arrival
  • TDOA locate methods have not been used to determine LOB because the TDOA reference points are too close to one another to make the accuracy of such a technique acceptable. Rather, LOB is usually determined using directional antennas and/or beam-forming antennas. The super resolution methods described herein, however, make it possible to use TDOA for LOB determination while dramatically improving accuracy.
  • TDOA e.g. detec t
  • reference signals coming from a UE outside of the serving sectors e.g.
  • TDOA time difference between the non-serving sectors and or antennas.
  • a UE may not be able to detect reference signals coming to the UE from other than serving sectors, e.g. from the non- serving sectors and/or antennas.
  • x is the distance between the two sensors in meters
  • ⁇ is the angle between the baseline of the sensors and the incident RF wave, in degrees ;
  • c is the speed of Sight.
  • TDOA locating embodiment uses Several locate strategies, including: (1) when the TDOA measurements (multilateration) between two or more serving cells are available, e.g., wide separation; (2) when the TDOA measurements are only from two or more sectors at one or more serving cells, e.g., small antenna separations, such LOB/ AoA; (3) a combination of strategies (2) and (3); and (4) a combination of TA
  • the TDOA locating embodiment ma use a line of bearing when the signals from two or more antennas are from the same cell tower. These signals can be detected in the received composite signal. By know ing the tower location and the azimuth of each sector and/or antenna, the line of bearing and/or AoA can be calculated and utilized in the location process.
  • the LOB/AoA accuracy may be impacted by multipath, noise (SN ), etc, However, this impact may be mitigated by advanced signal processing and the multipath mitigation processing techniques described above, which may be based on super resolution technology.
  • Such advanced signal processing includes, but is not limited to, signal correlation correlating, filtering, averaging, synchronous averaging and other methods/ techniques.
  • the serving cell lower 2500 typically consists of multiple sectors, as illustrated in FIG. 25 which shows a three sector (Sector A, Sector B and Sector C) configuration.
  • the three sector deployment illustrated may include one or more antennas 2502 per sector.
  • a single sector, such as sector A, may be in control of the UE (handset) because the handset transmissions will be in Sector A's main lobe (the main lobe's center coincides with the sector azimuth). At the same time the handset transmissions will fall outside Sectors B's and C's main lobes, e.g., into antennas side lobes.
  • the handset signals will still be present in the output signal spectrums of Sectors B and C, but will be significantly attenuated relative to signals from other handset(s) that are located in Sector B's or Sector C's main lobes. Nevertheless, through the use of advanced signal processing, as described above and below, it is possible to obtain sufficient processing gain on ranging signals to make them detectable from the neighboring sectors' side lobes, such as the Sector B and Sector C side lobes.
  • the LTE Uplink S S Sounding Reference Signals
  • the processing gain through reference signal processing methods described herein may be sufficient to allow a calculation of TDOA between the two (or more) sector antennas.
  • the accuracy of this embodiment may be significantly enhanced by the multipath mitigation processing algorithms described above.
  • LOB/AOA intersected with the annulus calculated by the LTE TA timing may provide a UE location to within an error ellipse of approximately 20 meters by 100 meters.
  • Further locale error reduction may be achieved when the UE can be heard by two or more LTE towers, which is highly probable with the processing gains and muhipaih mitigation technology described above, In such a case, the intersection of the TDOA hyperbola and one or more LOB/AoA lines may result in a 30 by 20 meter error ellipse (for a two sector cell tower). If each cell tower supports three or more sectors, then the error ellipse may be further reduced down to 10 - 15 meters. If the UE is heard by three or more eNB's (cell towers), then 5 to 10 meters accuracy may be achieved. In high value areas, such as malls, office parks arid the like, additional small cells or passive listening devices may be used to create the necessary coverage.
  • each sector of the cell tower 2500 may include one or more antennas 2502.
  • signals from each antenna are combined at the sector's receiver input.
  • two or more sector antennas can be viewed as a single antenna with composite directionality pattern, azimuth and elevation.
  • the hypothetical antenna composite directionality and its (main lobe) azimuth and elevation may also be assigned to the sector itself.
  • the received signals (in a digital format) from all sectors of each serving cell tower and neighboring serving cell towers are sent to a locate server unit (LSU) for location determination.
  • LSU locate server unit
  • SRS schedules and TA measurements per each served UE is provided to the LSU by each serving sector from each serving cell tower.
  • the LSU may determine each UE position relative to the serving ceil tower and/or neighboring cell towers.
  • All of the abovementioned information may be sent, through wired networks, for example LAN, WAN, etc., using one or more standardized or proprietary interfaces.
  • the LSU may also interface the wireless network infrastructure using a standardized interface and/or a network carrier's defined interface/ API.
  • the location determination may also be split between the network node and the LSU or performed solely in the network node.
  • the location determination may be performed in the UE or split between the UE and LSU or network node.
  • the UE may communicate over the air using standard networking protocols/interfaces.
  • the location determination can be performed through a combination of the UE, the LSU and/or netw ork nodes, or the LSU functionality can be implemented (embedded) into a SU.PL server, a E-S LC server, and/or a LCS (LoCation Sendees), system that can then foe used in place of the LSU.
  • LCS LiCation Sendees
  • ⁇ OSMJ Embodiments of a Downlink (DL) locate method are reciprocals to the Uplink (UL) locate embodiments described above, in a DL embodiment, a sector may become a transmitter with a transmit pattern, azimuth and elevation that matches the sector's received directionality, azimuth and elevation. Unlike the uplink embodiments, in DL embodiments, the UE typically has a single receive antenna. Thus, for UE there is no sensors baseline that can be used to determine the RF wave incident.
  • the UE can determine the TDOA(s) between different sectors and consequently a hyperbola(s) (multiiateration) between sectors, and because the same cell tower sectors are close to each other the hyperbola becomes interchangeable with the line oi " the RF energy incident or the LOB/AoA, as described above with reference to FIG. 24 While the LOB/AoA accuracy may be impacted by multipaih, noise (SMR), etc, this impact may be mitigated through use of the advanced signal processing and the muitipath mitigation processing, which is based on the super resolution technology, described above.
  • SMR noise
  • UE DL locating can be accomplished in ways that are similar to the UE uplink locating, with the exception of that the RF wave incident angle cannot be determined from the formula above. Instead, the multiiateration technique may be used for determining the LOB/AoA for each serving cell tower.
  • [O508 UE DL locale embodiments also employ reference signals, in the DL case, one approach for such network-based locating may be to employ the LTE Cell-Specific Reference Signals (CRS) as ranging signals. Also, Position Reference Signals (PRS) introduced in LTE Release 9 may be used. Thus, locate may be done using CRS only, PRS only, or both CRS and PRS.
  • CRS Cell-Specific Reference Signals
  • PRS Position Reference Signals
  • a snap-shot of the UE received signal in digital format may be sent to the LSU for processing.
  • the UE may also obtain the TA measurements and provide those to the LSU.
  • TA measurements per each served UE may be provided to the LSU by each serving sector from each serving cell tower (network node).
  • the LSU may determine each UE position relative to the serving cell tower and/or neighboring cell towers.
  • the location determination may be performed in the UE or split between the UE and LSU or network node. In embodiments, all location determinations can be performed in the LSU or the network node or split between the two.
  • the UE will communicate/receive measurements results and other information over the air using standard wireless protocols/ interfaces.
  • the information exchange between the LSU and network «ode(s) may be through wired networks, for example LAN, WAN, etc., using proprietary and/or one or more standardized interfaces.
  • the LSU may interface the wireless network infrastructure using a standardized interface and or network carrier's defined interface/API.
  • the location determination may also be split between the network node and the LSU or performed solely in the network node.
  • antenna port mapping information can also be used to determine location.
  • the 3GPP TS 36.211 LTE standard defines antenna ports for the DL. Separate reference signals (pilot signals) are defined in the LTE standard for each antenna port. Thus, the DL signals also carry the antenna port information.
  • This information is included in the PDSCH (Physical Downlink Shared Channel ⁇ .
  • the PDSCH uses the following antenna ports: 0; 0 and 1 ; 0, 1, 2 and 3); or 5. These logical antenna ports are assigned (mapped) to the physical transmit antennas, as illustrated in FIG. 26 As a result, this antenna port information can be used for the antenna identification (antenna ID).
  • the antenna port mapping information can be used to determine the RF wave incident and the hyperbola(s) (multilateration) between antennas (assuming that the antennas locations are known). Depending upon where the location determination is performed;, the antenna mapping information has to be available to the LSU or UE, or network node. It should be noted that antenna ports are indicated by placing CRS signals in different time slots and different resource elements. Only one CRS signal is transmitted per DL antenna port.
  • receiver ⁇ s In the event of MIMO (Multiple Input Multiple Outputs) deployment in the eNB or network node, receiver ⁇ s) ma be able to determine the time differences of arrivals from a given UE. With knowledge of antennas to the receiver(s) mapping, e.g. MIMO mapping, including antennas locations, it may also be possible to determine the RF wave incident (LOB/AoA) to antennas and the hyperbola(s) (multilateration) for given eNB antennas.
  • LOB/AoA RF wave incident
  • hyperbola(s) multilateration
  • the UE receiver(s) may be able to determine the time differences of arrival(s) from two or more eNB or network node, and MIMO antennas. With knowledge of the eNB antenna locations and antennas mapping, it will be possible to determine the RF wave incident (LOB/AoA) from antennas and the hy erbolae ' s) (mute ' lateratioa) for given eNB antennas. Depending OH where the location determination is performed; the antenna mapping information has to be available to the LSI; or UE, or network node.
  • LOB/AoA RF wave incident
  • hy erbolae ' s mute ' lateratioa
  • SIMO Single Input Multiple Outputs
  • SOMI Single Output Multiple Inputs
  • SISO Single Input Single Output
  • All of these configurations may be defined/determined by the antenna ports mapping and/or MIMO antenna mapping information for locate purposes.
  • the present embodiments relate to methods and systems for RF- based identification, tracking, and locating of objects, including RTLS.
  • the methods and systems employ geographically distributed clusters of receivers and/or transmitters that are precisely synchronized in time, e.g., within 10 ns or better, within each cluster, while the inter-cluster time synchronization can be much less accurate or not required at all. While a precise synchronization time of 10 ns or better is described with respect to one particular embodiment, it is important to note that the predetermined synchronization time required to achieve an accurate location depends on the equipment being utilized.
  • the predetermined time may need to be 10 ns or better, but with other wireless system equipment, a location accuracy of 50 m may be more than sufficient.
  • the predetermined time is based on the desired accuracy location for the wireless system.
  • the relative timing difference between signals coming from neighboring base stations is calculated and the UE position can be estimated in the network with the UE (handset) with or without UE assistance or in the UE (handset) with network assistance (control plane or user plane with SUPL based only) or without the network assistance.
  • the UE measures the relative timing difference between signals coming from a pair of base stations and produces hyperbolic lines of position (LOPs). At least three reference points (base stations not belonging to a straight line) are needed to define two hyperbolas.
  • the location (position fix) of the UE is in the intersection of these two hyperbolas (see FIG. 11).
  • the UE position fix is relative to the base stations' RF emitters' (antennas) locations.
  • LPP LTE Positioning Protocol, Rel-9
  • the DL- OTDOA locating is UE assisted and the E-SMLC (Evolved Serving Mobile Location Centre) is server based.
  • E-SMLC Evolved Serving Mobile Location Centre
  • the U-TDOA technique is similar to the DL-OTDOA, but the roles are reversed.
  • the neighboring Location Management Unit LMU calculates the Relative Time of Arrival of the uplink signal coming from the UE (handset) and the UE position can be estimated in the network without the UE assistance.
  • the U-TJDOA is LMU assisted and the E-SMLC (Evolved Serving Mobile Location Centre) is server based.
  • the network's E-SMLC server produces hyperbolic lines of position (LOPs) and the location (position fix) of the U E (see FIG. 27), The UE position fix is relative to the LMUs antennas locations.
  • the e B's (base station's) time synchronization in case of U-TDOA is not necessary - only (he LM.U(s) will need precision time synchronisation for locating purposes.
  • the LMU is essentially a receiver with computing capabilities.
  • the LMU receiver employs the SDR (Software Defined Radio) technology.
  • the LMU may be a small cell, macro cell or a special purpose small ceil type device that only receives.
  • correlating the location of the SRS for the specific UE, as provisioned by the network will enable identification and location of the UE.
  • Location of the SRS may be done at the network level or within a local sector, such as a DAS for a building, a small cell or combination of small cells and macro cells that serve a specific area. If the location of the SRS for the UE is not known a priori, the solution may be able to correlate the UE's location through the covered area. Doing so will show the location history of where the UE has travelled. In some circumstances, it may be desirable to determine the location of the UE, even if the network does not provide an indication of where the SRS is located for a particular UE.
  • the location of the UE may be correlated with the SRS by determining the location or proximity of the UE to a known point, thereby correlating the UE with the SRS it is transmitting. Such location can be accomplished through other location/proximity solutions, such as Wi-Fi and Bluetooth.
  • the user may also identify their location via a UE application or by walking over to a predetermined location in order to identify their UE to a location solution.
  • FIGS. 11 and 27 only the macro base stations are shown. Also, FIG. 27 depicts the LMUs being co-located with the base stations. These depictions are valid options, but the LIE standards do not specify where the LMUs can be placed, as long as LMUs placement satisfies the multilateration/trilateration requirements. [05201 ⁇ an aspect, a common deployment for Indoor environments is DAS
  • the LMU(s) can be placed indoors and/or within a campus-type environment as well, e.g. the U-TDOA can be used in a DAS and/or small cell environment.
  • the U-TDOA based accurate indoors locating can be achieved with a combination of LMUs positioned indoors and macro cells that are positioned outside, e.g. without the need of deploying DAS and/or small ceils: or have a reduced number of the small cells.
  • the LMUs can b 'deployed with or without DAS and/or small cells being present.
  • the LMUs can be placed in environments where cellular signal amplifiers/boosters are used; with or without DAS and/or small cells being present.
  • the LTE release 1 1 also contemplates the integration of the LMU and the e B into a single unit. This, however, will put additional burden on the time synchronization requirements between small cells if individual small cells eNBs are geographically distributed, which wireless/cellular service providers are not ready to meet, especially indoors and/or in other GPS/G SS denied environments.
  • DAS systems are inherently time synchronized to a much higher degree (precision) than geographically distributed macro/mini/small ce l/LMUs.
  • Using a DL-DTOA solution in a DAS environment will alleviate the time synchronization issue, but in a DAS environment, a single base station serves a large number of distributed antennas, such that multiple antennas are transmitting the same downlink signal with the same cell ID (identification number).
  • identity number the traditional DL-OTDOA approach fails because there are no identifiable neighboring cells (antennas) generating signals with a different ID.
  • One solution is to reduce the number of antennae that emit the same ID, e.g., split a large number of DAS antennas into two or more time synchronized clusters with different IDs. Such an arrangement will increase the system cost (increase the number of base stations) and require the handset UE to support the above-mentioned technology.
  • Employing U-TDOA in a DAS environment will also add cost relative lo adding/ installing LMU units. However, no changes to the UE (handset) will be needed; only the base station software would have to be upgraded to support the U-TDOA functionality. Also, it is possible to integrate multiple LMUs with (into) a DAS system. Therefore, using the U-TDOA method with LMUs has many ad vantages when utilized indoors, in campus environments, and in other GPS/GNSS challenging, geographically limited environments.
  • GPS/GNSS signal(s) quality is very good and macro cells antennas transmissions and/or LMU receivers can be synchronized, using GPS/GNSS to a very high accuracy -- standard deviation 10 os, over a sufficiently large area.
  • time synchronization amongst multiple distributed base station and/or small cells/LMUs is achieved by using an External Synchronization Source that produces the synchronization signal shared by many base stations and/or small cells and/or LMUs.
  • This synchronization signal can be derived from GPS/GNSS, for example the I PPS signal, and/or Internet/Ethernet networking, for example FTP or NIP, etc.
  • GPS/GNSS for example the I PPS signal
  • Internet/Ethernet networking for example FTP or NIP, etc.
  • the latter is a low cost solution, but it cannot provide the time synchronization precision required for accurate location, the GPS/GNSS derived external synchronization signal(s) are more precise - standard deviation down to 20 ns, but require additional hardware and installation requirements, e.g.
  • one embodiment uses a LMU device 2800 having multiple receive antennas 2802 and signal channels 2804.
  • one or more signal channels 2804 can comprising signal processing components such as an RFE (RF front end) 2806, RF down con verter 2808, and/or uplink-locate processor 2810.
  • RFE RF front end
  • RF down con verter 2808 RF down con verter 2808
  • the signal channels 2804 are co-located within the LMU device 2800 and tightly time synchronized (e.g., standard deviation of about 3 lis to about 10 us).
  • antennae 2802 from each LMU signal channel 2804 are geographically distributed, (e.g., similarly to DAS).
  • external time e.g., external time
  • syachranixatjon components e.g., GPS GNSS ⁇ Internet/Ethernet, etc.
  • the Precise time synchronization is more readily achieved inside the device ⁇ e.g., LMU device 2800) than it is by trying to tightly synchronize a number of
  • LMU device 2800 when two or more multichannel LMUs (e.g., LMU device 2800) are deployed, the time synchronization between these LMUs can be relaxed so that a low cost and low complexity approach can be used to synchronize (using an external source signal) a number of distributed multichannel LMUs.
  • latemel Ethernet networking synchronization can be used or a common sensor (device) can be deployed to provide timing synchronization between different multichannel LMUs.
  • the multichannel LMU approach reduces the number of hyperbolic lines of position (LOPs) that can be used in determining the position fix, but the time synchronization improvement overcomes this deficiency (see explanation and example below).
  • LOPs hyperbolic lines of position
  • the UE positioning accuracy is a function of two factors: the geometrical dilution of precision (GDOP), which is due to.
  • GDOP geometrical dilution of precision
  • the GDOP is function of the geographical distribution of transmitting antennas (in case of DL-OTDOA) or receiving antennas (in case of U-TDOA). In ease of the regularly placed antennae, the two dimensional GDOP estimation is equal to 2 v (11 B. LEE,
  • N is the number of emitters (macro cell towers/small cells/ DAS antennas) that are "hearable” by the UE (in case of DL-OTDOA) or the number of LMUs/ LMUs receive channels that can "hear” the UE uplink transmission (in case of U-TDQA).
  • the standard deviation of UE position error can be calculated as follows:
  • LMUs (regularly placed) are detecting the UE uplink transmission and these LMUs are synchronized via the I PPS signal (e.g., standard deviation of 20 ns). in. this case N :::: 8 and there will be seven independent LOPs thai can be used for UE position fix.
  • ranging error standard, deviation, ⁇ 3 ⁇ 4 is 3 meters (about 1 as); then the accuracy of single ranging measurement is:
  • two UE position fixes is generated, each with standard deviation error ti pos of 3.12 meters.
  • Combining these two position fixes by averaging and/or other means/methods will further reduce the UE position fix error.
  • One estimate is that the error reduction is proportional to the square root of the number of the UE position fixes. In the present disclosure, this number is equal two and the final UE position fix error a PQS FmAL is 2.21 meter; obtained as; 3.12 / 2.
  • multichannel LMU e.g., LMU device 2800
  • relaxed synclironizatiori betwee these multichannel LMUs can be used for indoors and other multichannel LMUs
  • the LMUs can be tightly synchronized (e.g., standard deviation of between about 3 ns and about 10 ns).
  • Another embodiment takes advantage of the fact that a number of single channel small cell/LMU and/or small cells with integrated LMU devices electronics (the LMU functionality is embedded into the eNB) can be clustered (e.g., integrated, co-located, etc.) in a rackmount enclosure (FIG. 31, FIG. 32 and FIG. 33) and/or a cabinet, for example a 19 inch rack.
  • Each single channel device antenna can be geographically distributed, like in DAS,
  • the devices within a cluster can be tightly time synchronized (e.g...
  • multiple LMUs can be integrated with (into) the DAS system as illustrated in FIG. 34.
  • the LMU receivers can share the received signal(s) generated by the each DAS antenna, e.g., sharing DAS antennas. The actual distribution of these received signals depends upon the DAS implementation: active DAS vs. passive DAS.
  • the LMU and DAS integration embodiment entails sharing the received signal(s) generated by the each DAS antenna with LMU receiver channel and creating an almanac that matches
  • each DAS antenna coordinates with corresponding LMU/ LMU receiver channel.
  • the clustering approach and/or employing multichannel LMU(s) are preferable ways for LMU and DAS integration.
  • each standalone LMU receive channel can support (be time multiplexed with) two or more antennae, for example serve two or more small cells. This, in turn, can lower the number of LMUs ( in small cells/DAS and/or other U-TDOA locate environments) and reduce the cost of the system (see also FIG. 28).
  • wireless/cellular network E-SMLC server is lacking the functionality required for DL-OTDOA and/or U-TDOA techniques, this functionality can be carried out by a location server that can communicate with the UE and-'or LMU and the wheless/ceilular network infrastructure and/or a location services server (see FIG. 29 and FIG. 3 ⁇ .
  • a location server that can communicate with the UE and-'or LMU and the wheless/ceilular network infrastructure and/or a location services server (see FIG. 29 and FIG. 3 ⁇ .
  • Other configurations can b used.
  • one or more LMU devices can be deployed with WiFi infrastructure, for example, as illustrated in FIG. 35.
  • a listening device could be used to monitor the LMU antenna in the same manner as the WiFi infrastructure.
  • the LMU devices and/or channel antennas servicing the LMUs can be co-located with one or more Wi.Fi/listening devices 3500, such as one or more WiFi access points (APs).
  • the WiFi devices 3500 can be geographically distributed.
  • the WiFi device 3500 can be connected to a power source.
  • An F analog portion 3502 (e.g., circuitry) of one or more LMU devices or channels can be integrated with the LMU antenna such that the RF analog portion 3502 can share the power source with the WiFi device 3S00 (see FIG. 35).
  • the RF analog portion 3502 of the LMU device or channel can be connected via cable to the Uplink-Locate processor circuitry (e.g., Uplink-Locate processor 2810), which can include the baseband signal processing.
  • Uplink-Locate processor 2810 can include the baseband signal processing.
  • the baseband circuitry such an embodiment facilitates improved signal-to-noise ratio (SNR).
  • the RF analog portion 3502 can down-convert the received signal (e.g., down to the baseband) and, because the baseband signal frequencies are several magnitudes smaller than the received signal in antenna, the cable requirements can be relaxed. Such relaxation of cable requirements can translate into cost reduction of the connections and can significantly increase the transmission distance.
  • the ranging signals are not limited to the SRS only and can utilize other reference signals, including MIMO, CRS (Cell-Specific Reference Signal), etc.

Abstract

Systems and methods for determining a location of one or more user equipment (UE) in a wireless system can comprise receiving reference signals via a location management unit having two or more co-located channels, wherein the two or more co-located channels are tightly synchronized with each other and utilizing the received reference signals to calculate a location of at least one UE among the one or more UE. Embodiments include multichannel synchronization with a standard deviation of less than or equal 10 ns. Embodiments can include two LMUs, with each LMU having internal synchronization, or one LMU with tightly synchronized signals.

Description

PARTIALLY SYNCHRONIZED ML'LTILATERATION/ TRILATERATION
ME THOD AND SYSTEM FOR POSITION AL FINDING USING RF
CROSS REFERENCE TO RELATED APPLICATIONS
[0001] This application claims the benefit of U.S. Provisional Patent Application. No. 62/032,371, filed August 1, 2014, entitled PARTIALL Y SYNCHRONIZED
MULTILATERATION/TRILATERATION METHOD AND SYSTEM FOR POSITIONAL FINDING USING RF; and will also be a continuation-in-part of U.S. Patent Application No. .13/566,993, filed August 3, 2012, entitled MULTI-PATH MITIGATION IN
RANGEFINDING AND I RACKING OBJECTS USING REDUCED ATTENUATION RF TECHNOLOGY, which claims benefit under 35 U.S.C. §119(e) of U.S. Provisional Application No. 61/51 ,839, filed August 3, 201 i, entitled MULTI-PATH MITIGATION IN
RANGEFINDING AND TRACKING OBJECTS USING REDUCED ATTENUATION RF TECHNOLOGY; U.S. Provisional Application No. 61/554,945, filed November 2, 201 1, entitled MULTI-PATH MITIGATION IN RANGEFINDING AND TRACKING OBJECTS USING REDUCED ATTENUATION RF TECHNOLOGY; U.S. Provisional Application No.
61/618,472, filed March 30, 2012, entitled MULTI-PATH MITIGATION IN RANGEFINDING AND TRACKING OBJECTS USING REDUCED ATTENUATION RF TECHNOLOGY; and U.S. Provisional Application No. 61/662,270, filed June 20, 2012, entitled MULTI-PATH MITIGATION IN RANGEFINDING AND TRACKING OBJECTS USING REDUCED ATTENUATION RF TECHNOLOGY; which are incorporated herein by reference in its entirety.
[0002] U.S. Patent Application No. 13/566,993 is a continuation-in-part of U.S. Patent
Application No. 13/109,904, filed May 17, 2011 , entitled MULTI-PATH MITIGATION IN
RANGEFINDING AND TRACKING OBJECTS USING REDUCED ATTENUATION RF
TECHNOLOGY, which is a continuation o.fU.S. Patent Application No. 13/008,519, filed
January 1 , 2011, now U.S. Patent No. 7,969,31 1 , issued June 28s 201 1 , entitled. METHODS
AND SYSTEM FOR MULTI-PATH MITIGATION IN TRACKING OBJECTS USING
REDUCED ATTENUATION RF TECHNOLOGY, which is a conti««ation-in-part of U.S.
Patent Application No. 12/502,809, filed on July 14, 2009, now U.S. Patent No. 7,872,583, issued January 8, 2011, entitled METHODS AND SYSTEM FOR REDUCED
ATTENUATION IN TRACKING OBJECTS USING RF TECHNOLOGY, which is a
continuation of U.S. Patent Application No. 1 1/610,595, filed on December 1 , 2006, ROW U.S. 6071 7370V 1 - i - Patent No, 7361 ,048, issued My 14, 2009, entitled METHODS AMD SYSTEM FOR
REDUCED ATTENUATION IN TRACKING OBJECTS USING RF TECHNOLOGY, which claims benefit under 35 U.S.C, § 1.19(e) of U.S. Provisional Patent Application No. 60/597,649 filed OB December 15, 2005, entitled METHOD AND SYSTEM FOR REDUCED
ATTENUATION IN TRACKING OBJECTS USING MULTI-BAND RF TECHNOLOGY, which are incorporated by reference herein in their entirety .
[0003] U.S. Patent Application No. 12/502,809, filed on July 14, 2009, entitled METHODS AND SYSTE FOR REDUCED ATTENUATION IN TRACKING OBJECTS USING RF TECHNOLOGY, also claims benefit under 35 U.S.C. § 119(e) of U. S. Provisional Application No. 61/1 3,270, filed on October 7, 2008, entitled METHODS AND SYSTEM FOR MULTI-PATH MITIGATION IN TRACKING OBJECTS USING REDUCED
ATTENUATION RF TECHNOLOGY, which are incorporated by reference herein in their entirety.
TECHNICAL FIELD
[0004] The present embodiment relates to wireless communications and wireless networks systems and systems for a Radio Frequency (RF)-based identification, tracking and locating of objects, including RTLS (Real Time Locating Service) and LTE based locating services.
BACKGROUND
[0005] RE-based identification and location-finding systems for determination of relative or geographic position of objects are generally used for tracking single objects or groups of objects, as well as for tracking individuals. Conventional location-finding systems have been used for position determination in an open, outdoor environment. RF-based, Global Positioning System (GPS)/Global Navigation Satellite System (GNSS), and assisted GPSs/GNSSs are typically used. However, conventional location-finding systems suffer from certain inaccuracies when locating the objects in closed (i.e., indoor) environments, as well, as outdoors.
[0006] Cellular wireless communication systems provide various methods of locating user equipment (UE) position indoors and in environments that are not well suited for GPS. The most accurate methods are positioning techniques that are based on the
multilateration/trilateration methods. For example, LTE (Long Term Evolution) standard release 9 specifies the DL-OTDOA (Downlink Observed Time Difference of Arrival) and release 1 1 specifies the U-TDOA (Uplink Time Difference of Arrival) techniques, thai are derivatives of the multil erau'on lrilateration methods,
[0007] Since time synchronization errors impact locate accuracy, the fundamental requirement for multiiateration/triiateration based systems is the complete and precise time synchronization of the system to a single common reference time, in cellular networks, the DL- OTDOA and the U-TDOA locating methods also require, in the case of DL-OTDOA, that transmissions from multiple antennas be time synchronized, or in the case of U-TDOA, thai multiple receivers be time synchronized.
[0008] The LTE standards release 9 and release 1 1 do not specify the time
synchronization accuracy for the purpose of locating, leaving this to wireless/cellular service providers. On the other hand, these standards do provide limits for the ranging accuracy. For example, when usin 10 MH* ranging signal bandwidth, the requirement is 50 meters ¾67% reliability for the DL-OTDOA and 100 meters 67% reliability for the U-TDOA.
[0009] The above noted limits are the result of a combination of ranging measurements errors and errors caused by the lack of precision synchronization, e.g. time synchronization errors. From the relevant LTE test specifications (3GPP TS 36.133 version 10.1.0 release 10) and other documents, it is possible to estimate the time synchronization error, assuming that the synchronization error is uniformly distributed. One such estimate amounts to 200 ns (100 ns peak-to-peak). It should be noted that the Voice over LTE (VoLTE) functionality also requires cellular network synchronization down to 150 nanoseconds (75 ns peak-to-peak), assuming that the synchronization error is uniformly distributed. Therefore, going forward, the LTE network's time synchronization accuracy will be assumed to be within Ϊ 50 ns.
[0010] As for distance location accuracy, FCC directive NG 91.1 specifies locate accuracy requirements of 50 meters and 100 meters. However, for the Location Based Services (LBS) market, the indoors location requirements are much more stringent - 3 meters @67% reliability. As such, the ranging and locate error introduced by the time synchronization error of 1 0 ns (the standard deviation of 43 ns) is much larger than the 3 meters ranging error (standard deviation of 1.0 ns).
[001 1 j While cellular network's time synchronization might be adequate to satisfy the mandatory FCC NG E91 1 emergency location requirements, this synchronization accuracy falls short of the needs of LBS or RTLS system users, who require significantly more accurate locating. Thus, there is a need in the art for mitigating the locate error induced by lack of accurate time synchronization for cellular/wireless networks for the purpose of supporting LBS and TLS.
SUMMARY
(00121 The present disclosure relates to methods and systems for Radio Frequency (RF)-based identification, tracking and locating of objects, including Real Time Locating Service (RTLS) systems that substantially obviate one or more of the disadvantages associated with existing systems. The methods and systems can use partially synchronized (in time) receivers and/or transmitters. According to an embodiment, RF-based tracking and locating is
implemented in cellular networks, but could be also implemented in any wireless system and RTLS environments. The proposed system can use software implemented digital signal processing an ! software defined radio technologies (SDR). Digital signal processing (DSP) can be used as well.
[0013] One approach described herein employs clusters of receivers and/or transmitters precisely time synchronized within each cluster, while the inter-cluster time synchronization can be much less accurate or not required at all. The present embodiment can be used in all wireless systems/networks and include simplex, half duplex and full duplex modes of operation. The embodiment described below operates with wireless networks that employ various modulation types, including OFDM modulation and/or its derivatives. Thus, the embodiment described below operates with LTE networks and it is also applicable to other wireless systems/networks.
[0014} As described in one embodiment, RF-based .tracking and locating is
implemented on 3 GPP LTE cellular networks will significantly benefit from ihe precisely synchronized (in time) receivers and or transmitters clusters. The proposed system can use software- and/or hardware-implemented digital signal processing.
[0015] Additional features and advantages of the embodiments will be set forth in the description that follows, and in part will be apparent from the description, or may be learned by practice of the embodiments. The advantages of the embodiments will be realized and attained by the structure particularly pointed out in the written description and claims hereof as well as the appended drawings.
[0016] It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are intended to provide further explanation of the embodiments as claimed. BRIEF DESCRIPTION OF THE DRAWINGS
[001 7| The accompanying drawings, which are included to provide a further understanding of the embodiments and are incorporated in and constitute a part of this specification, illustrate embodiments and together with the description serve to explain the principles of the embodiments. In the drawings:
[0018] FIG. 1 and FIG. 1 A illustrate narrow bandwidth ranging signal frequency components, in accordance with an embodiment:
[001 9} FIG. 2 illustrates exemplary wide bandwidth ranging signal frequency components;
[0020] FIG. 3 A, FIG. 3B and FIG. 3C illustrate block diagrams of master and slave units of an RF mobile tracking and locating system, in accordance with an embodiment;
[00211 FIG . 4 illustrates an embodiment synthesized wideband base band ranging signal;
[0022 J FIG, 5 illustrates elimination of signal precursor by cancellation, in accordance with an embodiment;
[0023J FIG. 6 Illustrates precursor cancellation with fewer carriers, in accordance with an embodiment;
[0024} FIG. 7 illustrates an embodiment of one-way transfer function phase;
[00251 F3CJ- 8 illustrates an embod iment of a location method;
[0026} FIG. 9 illustrates LIE reference signals mapping;
[0027} FIG. 10 illustrates an embodiment of an enhanced Cell ID + RTT locating technique;
[00281 FIG. 1 1 illustrates an embodiment of an OTDOA locating technique;
[00291 FIG. 1.2 illustrates the operation of a Time Observation Unit (TMO) installed at an operator's e B facility, in accordance with an embodiment;
[0030] FIG. 13 illustrates an embodiment of a wireless network locate equipment diagram;
[0031] FIG. 14 illustrates an embodiment of a wireless network locate downlink ecosystem for enterprise applications;
[0032} FIG. 15 illustrates an embodiment of a wireless network locate downlink ecosystem for network wide applications;
[0033} FIG. 16 illustrates an embodiment of a wireless network locate uplink ecosystem for enterprise applications; [00341 FIG. 17 illustrates an embodiment of a wireless network locale uplink
ecosystem for network wide applications;
[0035] FIG. 18 illustrates an embodiment of an UL-TDOA environment that may include one or more DAS and/or femto/small cell antennas;
[0036] FIG. 19 illustrates an embodiment of an UL-TDOA like that of FIG. 1.8 that ma include one or more cell towers that can be used in lieu of DAS base stations and/or femto/small ceils;
[0037] FIG. 20 illustrates an embodiment of cell level locating;
[0038j FIG. 21 illustrates an embodiment of serving ceil and sector ID locating;
[0039] FIG. 22 illustrates an embodiment of E-O'D plus AoA locating;
[0040] FIG. 23 illustrates an embodiment of AoA locating;
[0041 J FIG. 24 illustrates an embodiment of TDOA with wide and close distances between receiving antenna;
[00421 FIG, 25 illustrates an embodiment of a three sector deployment;
[0043 j FIG. 26 illustrates an embodiment of antenna potts mapping;
[0044] FIG. 27 illustrates an embodiment of an LTE Release 11 U-TDOA locating technique;
[0045] FIG. 28 illustrates an embodiment of a multichannel Location Management Unit (LMU) high level block diagram;
[0046] FIG. 29 illustrates an embodiment of a DL-OTDOA technique in
wireless/cellular network with a location Server;
[0047] FIG. 30 illustrates an embodiment of a U-TDOA technique in wireless/cellular network with a location Server;
[0048] FIG. 31 illustrates an embodiment of a depiction of a rackmount enclosure;
(0049 ] FIG. 32 illustrates an embodiment of a high level block diagram of multiple single channel LMUs clustered (integrated) in a rackmount enclosure;
[0050] FIG. 33 illustrates an embodiment of a high level block diagram of multiple small cells with integrated LMU clustered (integrated) in a rackmount enclosure (one-to-one antenna connection mapping); and
[0051 j FIG. 34 illustrates an embodiment of a high level block diagram of LMUs and DAS integration.
J0051] FIG. 35 illusirates an embodiment of a high level block diagram of LMUs and WiFi infrastructure integration. DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS
[00521 Reference will now be made in detail to the preferred embodiment of the present embodiments, examples of which are illustrated in the accompanying drawings.
[0053] The present embodiments relate to a method and system for RF-based identification, tracking and locating of objects, including RTLS. According to an embodiment, the method and system employs a narrow' bandwidth ranging signal. The embodiment operates in VHF band, but can be also used in HF, LF and VLF bands as well as UHF band and higher frequencies. It employs multi-path mitigation processor. Employing multi-path mitigation processor increases the accuracy of tracking and locating implemented by a system.
[0054] The embodiment includes small, highly portable base units that allow users to track, locate and monitor multiple persons and objects. Each unit has its own ID. Each unit broadcasts an RF signal with its ID, and each unit is able to send back a return signal, which can include its ID as well as voice, data and additional information. Each unit processes the returned signals from the other units and, depending on the triangulation or trilateration and/or other methods used, continuously determines their relative and/or actual locations. The preferred embodiment can also be easily integrated with products such as GPS devices, smart phones, two- way radios and PDAs. The resulting product will have all of the functions of the stand-alone devices while leveraging the existing display, sensors (such as altimeters, GPS, accelerometers and compasses) and processing capacity of its host. For example, a GPS device with the device technology describe herein will be able to provide the user's location on a map as well as to ma the locations of the other members of the group.
[0055] The size of the preferred embodiment based on an FPGA implementation is between approximately 2x4x1 inches and 2x2x0.5 inches, or smaller, as integrated circuit technology improves. Depending on the frequency used, the antenna will be either integrated into the device or protrude through the device enclosure. An ASIC (Application Specific Integrated Circuit) based version of the device will be able to incorporate the functions of the FPGA and most of the other electronic components in the unit or Tag. The ASIC -based standalone version of the product will result in the device size of 1x0.5x0.5 inches or smaller. The antenna size will be determined by the frequency used and part of the antenna can be integrated into the enclosure. The ASIC based embodiment is designed to be integrated into products can consist of nothing more than a chipset. There should not be any substantial physical size difference between the Master or Tag units. {00561 The devices can use standard system components (off-the-shelf components) operating at multiple frequency ranges (bands) for processing of multi-path mitigation algorithms. The software for digital signal processing and software-defined radio can be used. The signal processing software combined with minimal hardware, allows assembling the radios that have transmitted and received waveforms defined by the software.
J0057J U.S. Patent No, 7,561 ,048 discloses a narrow-bandwidth ranging signal system, whereby ike narrow -bandwidth ranging signal is designed to fit into a low-bandwidth channel, for example using voice channels that are only several kilohertz wide (though some of low- bandwidth channels may extend into a few tens of kilohertz). This is in contrast to conventional location- finding systems that use channels from hundreds of kilohertz to tens of megahertz w ide.
{005$! The advantage of this narrow-bandwidth ranging signal system is as follows: 1) at lower operating frequencies bands, conventional location-finding systems ranging signal bandwidth exceeds the carrier (operating) frequency value. Thus, such systems cannot be deployed at LF/VLF and other lower frequencies bands, including HF. Unlike conventional location- finding systems, the narrow-bandwidth ranging signal system described in U.S. Patent No. 7,561,048 can be successfully deployed on LF, VLF and other bands because its ranging signal bandwidth is far below the carrier frequency value; 2) at lower end of RF spectrum (some VLF, LF, HF and VHF bands), e.g., up to UHF band, conventional location-finding systems cannot be used because the FCC severely limits the allowable channel bandwidth (12 - 25 kHz), which makes it impossible to use conventional ranging signals. Unlike conventional location- finding systems, the narrow-bandwidth ranging signal system's ranging signal bandwidth is fully compliant with FCC regulations and other international spectrum regulatory bodies; and 3) it is well known (see MRT. the basics, by Ray H. Hashemi, William G. Bradley ... - 2003) that independently of operating frequency/band, a narrow-bandwidth signal has inherently higher SNR (Signal-to-Noise-Ratio) as compared to a wide-bandwidth signal. This increases the operating range of the narrow-bandwidth ranging signal location-finding system independently of the frequency/band it operates, including UHF band .
[0059] Thus, unlike conventional location-finding systems, the narrow-bandwidth ranging signal location- finding system can be deployed on lower end of the RF spectrum - for example VHF and lower frequencies bands, down to LF/VLF bands, where the multipath phenomena is less pronounced. At the same time, the narrow-bandwidth ranging location-finding system can be also deployed on UHF band and beyond, improving the ranging signal SNR and, as a result, increasing the location-finding system operating range. JOO60f To mi imize multipath, e.g., RF energy reflections, it is desirable to operate on VLF/LP bands. However, at these frequencies the efficiency of a portable/mobile antenna is very small (about 0.1% or less because of small antenna length (size) relative to the RF wave length). In addition, at these low frequencies the noise level from natural and manmade sources is much higher than on higher frequencies/bands, for example VHP. Together, these two phenomena may limit the applicability of location-finding system, e.g. its operating range and/or
mobility/portability. Therefore, for certain applications where operating range and/or
mobility/portability are very important a higher RF frequencies/bands may be used, for example HP, VHF, UHF and UWB.
[0061] At VHF and UHF bands, the noise level from natural and manmade sources is significantly lower compared to VLF, LF and HF bands; and at VHF and HF frequencies the multi-path phenomena (e.g., RF energy reflections) is less severe than at UHF and higher frequencies. Also, at VHF, the antenna efficiency is significantly better, than on HF and lower frequencies, and at VHF the RF penetration capabilities are much better than at UHF. Thus, the VHF band provides a good compromise for mobile/portable applications. On the other hand in some special cases, for example GPS where VHF frequencies (or lower frequencies) cannot penetrate the ionosphere (or get deflected/refracted), the UHF can be a good choice. However, in any case (and all cases/applications) the narrow-bandwidth ranging signal system will have advantages over the conventional wide-bandwidth ranging signal location-finding systems.
|Ot 2J The actual applicaiion(s) will determine the exact technical specifications (such as power, emissions, bandwidth and operating frequencies/band). Narrow bandwidth ranging allows the user to either receive licenses or receive exemption from licenses, or use unlicensed bands as set forth in the FCC because narrow band ranging allows for operation on many different bandwidths/frequencies, including the most stringent narrow bandwidibs: 6.25kpi¾, 1 1.25kHz, 12.5kHz, 25kHz and 50kHz set forth in the FCC and comply with the corresponding technical requirements for the appropriate sections. As a result, multiple FCC sections and exemptions within such sections will be applicable. The primary FCC Regulations thai are applicable are: 47 CFR Part 90- Private Land Mobile Radio Services, 47 CFR Part 94 personal Radio Services, 47 CFR Part 15 - Radio Frequency Devices, (By comparison, a wideband signal in this context is from several hundred KHz up to 10-20 MHz.)
[0063] Typically, for Part 90 and Part 94, VHP implementations allow the user to operate the device np to lOOmW under certain exemptions (Low Power Radio Service bein an example). For certain applications the allowable transmitted power at VHF band is between 2 and 5 Watts. For 900 MHz (UHF a»d) it is 1W. On 160 kHz - 190 kHz frequencies (LF band) the allowable transmitted power is I Watt,
[0064] Narrow band ranging can comply with many if not all of the different spectrum allowances and allows for accurate ranging while still complying with the most stringent regulatory requirements. This holds true not just for the FCC, but for other international organizations that regulate the. use of spectrum throughout the world, including Europe, Japan and Korea.
[0065] The following is a list of the common frequencies used, with typical power usage and the distance the tag can communicate with another reader in a real world emironment (see Indoor Propagation and Wavelength Dan Dobkisi, WJ Communications, V 1.4 7/ S O/02):
915 MHz 100 ffiW 150 feet
2.4 GHz 100 roW 100 feet
5.6 Ghz l mW 75 feet
[0066] The proposed system works at VHF frequencies and employs a proprietary method for sending and processing the RF signals. More specifically, it uses DSP techniques and software-defined radio (SDR) to overcome the limitations of the narrow bandwidth requirements at VHF frequencies.
[0067] Operating at lower (VHF) frequencies reduces scatter and provides much better wall penetration. The net result is a roughly ten-fold increase in range over commonly used frequencies. Compare, for example, the measured range of a prototype to that of the RFID technologies listed above:
Figure imgf000011_0001
[ 681 Utilizing narrow band ranging techniques, the range of commonly used frequencies, with typical power usage and the distance the tag communication range will be able to communicate with another reader in a real world environment would increase significantly:
From: To:
915 MHz 100 mW 150 feet 500 feet
2.4 GHz 00 BiW 100 feet 450 feet
5.6 Ghz 100 mW 75 feet 400 feet
[0069] Battery consumption is a function of design, tran smitted power and the duty cycle of the device, e.g., the time interval between two consecutive distance (location) measurements. In many applications the duty cycle is large, l OX to S 000X. In applications with large duly cycle, for example 100X, an FPGA version that franstniis 100 mW of power will have an up time of approximately three weeks. An ASIC based version is expected to increase the up time by 10X. Also, ASICs have inherently lower noise level. Thus, the ASIC -based version may also increase the operating range by about 40%.
[0070] Those skilled in the art will appreciate that the embodiment does not compromise the system long operating range while significantly increases the location-finding accuracy in RF challenging environments (such as, for example, buildings, urban corridors, etc.)
[0071] Typically, tracking and location systems employ Track-Locate-Navigate methods. These methods include Time-Of-Arrival (TO A), Differential-Time-Of-Arrival (DTOA) and combination of TOA and DTOA. Time-Of-Arrival (TO A) as the distance measurement technique is generally described in U.S. Patent No. 5,525,967. A.TOA/DTOA- based system measures the RF ranging signal Direct-Line-Of-Site (DLOS) iirae-of-Hight, e.g., time-delay, which is then converted to a distance range.
(0072] In case of RF reflections (e.g. , multi-path), multiple copies of the RF ranging: signal with various delay times are superimposed onto the DLOS RF ranging signal. A track- locate system that uses a narrow bandwidth ranging signal cannot differentiate between the DLOS signal and reflected signals without multi-path mitigation. As a result, these reflected signals induce an error in the estimated ranging signal DLOS time-of-flight, which, in turn, impacts the range estimating accuracy.
[0073] The embodiment advantageously uses the multi-path mitigation processor to separate the DLOS signal and reflected signals. Thus, the embodiment significantly lowers the error in the estimated ranging signal DLOS time-of-flight. The proposed multi-path mitigation method can be used on all RF bands. It can also be used with wide bandwidth ranging signal location-finding systems. And it can support various modulation/demodulation techniques, including Spread Spectrum techniques, such as DSS (Direct Spread Spectrum) and FH
(Frequency Hopping).
[0074] Additionally, noise reduction methods can be applied in order to further improve the method's accuracy. These noise reduction methods can include, but are not limited to, coherent summing, non-coherent summing, Matched filtering, temporal diversity techniques, etc. The remnants of the multi-path interference error can be further reduced by applying the postprocessing techniques, such as, maximum likelihood estimation (like.g., Viterbi Algorithm), minimal variance estimation (Kalman Filter), etc. [0Θ75| The embodiment cm be used m systems with, simplex, half-duplex and full duplex modes of operation. Full-duplex operation is very demanding in terms of complexity, cost and logistics on the RF transceiver, which limits the system operating range in
portable/mobile device implementations. In half-duplex mode of operation the reader (often referred to as the "master") and the tags (sometimes also referred to as "slaves" or "targets") are controlled by a protocol that only allows the master or the slave to transmit at any given time.
[0076] The alternation of sending and receiving allows a single frequency to be used in distance measurement. Such an arrangement reduces the costs and complexity of the system in comparison with full duplex systems. The simplex mode of operation is conceptually simpler, but requires a more rigorous synchronization of events between master and target unit(s), including the start of the ranging signal sequence.
[0077] In present embodiments the narrow bandwidth ranging signal multi-path mitigation processor does not increase the ranging signal bandwidth. It uses different frequency components, advantageously, to allow propagation of a narrow bandwidth ranging signal.
Further ranging signal processing can be carried out in the frequency domain by way of employing super resolution spectrum estimation algorithms (MUSIC, rootMUSIC, ESPRIT) and/or statistical algorithms like RELAX, or in time-domain by assembling a synthetic ranging signal with a relatively large bandwidth and applying a further processing to this signal. The different frequency component of narrow bandwidth ranging signal can be pseudo randomly selected, it can also be contiguous or spaced apart in frequency, and it can have uniform and/or non-uniform spacing in frequency.
[0078] The embodiment expands multipath mitigation technology. The signal model for the narrowband ranging is a complex exponential (as introduced elsewhere in this document) whose frequency is directly proportional to the delay defined by the range plus similar terms whose delay is defined by the time delay related to the multipath. The model is independent of the actual implementation of the signal structure, e.g., stepped frequency, Linear Frequency Modulation, etc.
[0079J The frequency separation between the direct path and multipath is nominally extremely small and normal frequency domain processing is not sufficient to estimate the direc path, range. For example a stepped frequency ranging signal at a lOO Hz stepping rate over 5 MHz at a. range of 30 meters (100.07 nanoseconds delay) results in a frequency of 0.062875 radians/sec. A multipath reflection with a path length of 35 meters would result in a frequency of 0.073355. The separation is 0.0104792. Frequency resolution of the 50 sample observable has a native frequency resolution of 0.12566 Hz. Consequently it is not possible to use conventional .frequency estimation techniques for the separation of the direct path from the reflected path and accurately estimate the direct path range.
[0080] To overcome this limitation the embodiments use a unique combination of implementations of subspace decomposition high resolution spectral estimation methodologies and multimodal cluster analysis. The subspace decomposition technology relies on breaking the estimated covariance matrix of the observed data into two orthogonal subspaces, the noise subspace and the signal subspace. The theory behind the subspace decomposition methodology is that the projection of the obser able onto the noise subspace consists of only the noise and the projection of the observable onto the signal subspace consists of only the signal.
[0081] The super resolution spectrum estimation algorithms and RELAX algorithm are capable of distinguishing closely placed frequencies (sinusoids) in spectrum in presence of noise. The frequencies do not have to be harmonically related and, unlike the Digital Fourier Transfonn (DFT), the signal model does not introduce any artificial periodicity. For a given bandwidth, these algorithms provide significantly higher resolution than Fourier Transform. Thus, the Direct Line Of Sight (DLOS) can be reliably distinguished from other multi-paths (MP) with high accuracy. Similarly, applying the thresholded method, which will be explained later, to the artificially produced synthetic wider bandwidth ranging signal makes it possible to reliably distinguish DLOS from other paths with high accuracy.
[0082} In accordance with the embodiment, the Digital signal processing (DSP), can be employed by the multi-path mitigation processor to reliably distinguish the DLOS from other MP paths. A variety of super-resolution algorithms/techniques exist in the spectral analysis (spectrum estimation) technology. Examples include subspace based methods: Multiple Signal Characterization (MUSIC) algorithm or root-MUSIC algorithm, Estimation of Signal Parameters via Rotational Invariance Techniques (ESPRIT) algorithm, Pisarenko Harmonic Decomposition (PHD) algorithm, RELAX algorithm, etc.
[0083] The noted super-resolution algorithms work on the premise that the signals impinging on the antennas are not fully correlated. Thus, the performance degrades severely in a highly correlated signal environment as may be encountered in multipath propagation. Multipath mitigation techniques may involve a preprocessing scheme called spatial smoothing. As a result, the multipath mitigation process may become computationally intensive, complicated, i.e., increases the complexity of the system implementation. Multipath mitigation with lower system computational costs and implementation complexity may be achieved hy using the super - resolution Matrix Pencil (MP) algorithm. The MP algorithm is classified as a non-search procedure. Therefore, it is computationally less complicated and eliminates problems encountered in search procedures used in other super-resolution algorithms. Moreover, the MP algorithm is not sensitive to correlated signals and only requires a single channel estimate and can also estimate the delays associated with coherent multipath components.
[0084] In all of the abovementioned super-resolution algorithms the incoming (i.e., received) signal is modeled as a linear combination of complex exponentials and their complex amplitudes of frequencies; In case of a multi-path, the received signal will he as follows:
Figure imgf000015_0001
(0085J where β x e'~'v " is ihe transmitted signal, , is the operating frequency, L is the number of multi-path components, and aK ■■■ |«x | x and rA, are the complex attenuation and propagation delay of the K-th path, respectively. The multi-path components are indexed so that the propagation delays are considered in ascending order. As a result, in this model r0 denotes the propagation delay of the DLOS path. Obviously, the r0 value is of the most interest, as it is the smallest value of all τκ . The phase θκ is normally assumed random from one measurement cycle to another with a uniform probability density function U (0,2/r ). Thus, we assume that a^- = const (i.e., constant value)
[0086] Parameters aK and τκ are random time-variant functions reflecting motions of people and equipment in and around buildings. However, since the rate of their variations is very slow as compared to the measurement time interval, these parameters can be treated as time-invariant random variables within a given measurement cycle.
[0087] All these parameters are frequency-dependent since they are related to radio signal characteristics, such as, transmission and reflection coefficients. However, in the embodiment, the operating frequency changes very little. Thus, the abovementioned parameters can be assumed frequency-independent.
[00881 Equation ( I } can be presented in frequency domain as:
4/) ,, x e
k :: (i ♦ (2) where: A{f ) is complex amplitude of the. received signal, (_2 r x rA- ) are the artificial "frequencies" to be estimated by a super-resolution algorithm and the operating frequency / is the independent variable; o.K is the K-th path amplitude.
[0089] In the equation (2) the super-resolution estimation of 2π x rK ) and subsequently τκ values are based on continuous frequency. In practice, there is a finite number of measurements. Thus, the variable /' will not be a continuous variable, but rather a discrete one. Accordingly, the com lex amplitude A {/ ) can be calculated as follows:
Figure imgf000016_0001
(0090J where A (/,,. ) are discrete complex amplitude estimates (i.e., measurements) at discrete frequencies ,■
[009 if In equation (3) A {/. )can be interpreted as an amplitude and a phase of a sinusoidal signal of frequency f„ after it propagates through the multi-path channel. Note that all spectrum estimation based super-resolution algorithms require complex input data (i.e. complex amplitude).
[0092] In some cases, it is possible to convert real signal data, e.g. Re A (/„. )), into a complex signal (e.g., analytical signal). For example, such a conversion can be accomplished by using Hiibert transformation or other methods. However, in case of short distances the value r0 is very small, which results in very low {ΐ π X s- ) "frequencies".
[0093] These low "frequencies" create problems with Hiibert transform (or other methods) implementations. In addition, if only amplitude values (e.g., Re {,4 (/,. ))} are to be used, then the number of frequencies to be estimated will include not only the \2ft x r A- ) "frequencies", but also theirs combinations. As a rule, increasing the number of unknown frequencies impacts the accuracy of the super-resolution algorithms. Thus, reliable and accurate separation of DLOS path from other multi-path (MP) paths requires complex amplitude estimation.
|0094| The following is a description of a method and the multi-path mitigation processor operation during the task of obtaining complex amplitude A.{fK ) in presence of multi-path. Note that, while the description is focused Oil the ha lf-duplex mode of operation, it can. be easily extended for the full-duplex mode. The simplex, mode of operation is a subset of the half-duplex mode, but would require additional events synchronization.
[0095] In half-duplex mode of operation the reader (often referred to as the "master") and the tags (also referred to as "slaves" or "targets") are controlled by a protocol that only allows the master or the slave to transmit at any given time. In this mode of operation the tags (target devices) serve as Transponders. The tags receive the ranging signal from a reader (master device), store it in the memory and then, after certain time (delay), re-transmit the signal back to the master.
[ 096J An example of rang ing signal is shown in FIG. 1 and FIG. 1A. The exemplary ranging signal employs different frequency components that are contiguous. Other waveforms, including pseudo random, spaced in frequency and/or time or orthogonal, etc. can be also used for as long as the ranging signal bandwidth remains narrow. In FIG.l the time duration T f for every frequency component is long enough to obtain the ranging signal narrow-bandwidth property.
{0097| Another variation of a ranging signal with different frequency components is shown on FIG. 2. It includes multiple frequencies (ft, f>, ΐ £*, Q transmitted over long period of time to make individual frequencies narrow-band. Such signal is more efficient, but it occupies in a wide bandwidth and a wide bandwidth ranging signal impacts the SNR, which, in turn, reduces the operating range. Also, such wide bandwidth ranging signal will violate FCC requirements on the VHF band or lower frequencies bands. However, in certain applications this wide-bandwidth ranging signal allows an easier integration into existing signal and transmission protocols. Also, such a signal decreases the track-locate time.
[0098] These multiple-frequency (fi, f>, ¾, ¾, f„) bursts may be also contiguous and/or pseudo random, spaced in frequency and/or time or orthogonal, etc.
[0099] The narrowband ranging mode will produce the accuracy in the form of instantaneous wide band ranging while increasing the range at wfiich this accuracy can be realized, compared to wide band ranging. This performance is achieved because at a fixed transmit power, the SNR (in the appropriate signal bandwidths) at the receiver of the narrow band ranging signal is greater man the SNR at the receiver of a wideband ranging signal. The SNR gain is on the order of the ratio of the total bandwidth of the wideband ranging signal and the bandwidth of each channel of the narrow band ranging signal. This provides a good trade-off when very rapid ranging is not required, e.g., for stationary and slow-moving targets, such as a person walking or running. [OlOOf Master devices and Tag devices are identical and can operate either in Master or Transponder mode. All devices include data/remote control communication channels. The devices can exchange the information and master device(s) can remotely control tag devices. In this example depicted in FIG. 1 during an operation of a master (i.e., reader) multi-path mitigation processor originates the ranging signal to tag(s) arid, after a certain delay, the.
master/reader receives the repeated ranging signal from the tag(s).
[0101] Thereafter, master's multi-path mitigation, processor compares the received ranging signal with the one that was originally sent from the master and determines the A ( fn ) estimates in form of an amplitude and a phase for every frequency component fn . Note that in the equation (3) A ( , ) is defined for one-w ay ranging signal trip. In the embodiment the. ranging signal makes a round-trip, in other words, it travels both ways: from a master/reader to a target/slave and from the target/slave back to the master/reader. Thus, this round-trip signal complex amplitude, which is received back by the master, cart be calculated as follows:
ji .,, (/·. J - μ if , t a d A «r (/- )■ 2 x A (/„ )) (4)
[0102] There are many techniques available for estimating the complex amplitude and phase values, including, for example, matching filterin A(f„ )| and Z A (/„ ) . According to the embodiment, a complex amplitude determination is based on |^ (/;, J values derived from the master and/or tag receiver SSI (Received Signal Strength indicator) values. The phase values Z ART (/n ) are obtained by comparing the received by a reader/master returned baseband ranging signal phase and the original (i.e., sent by reader/master) base band ranging signal phase. In addition, because master and tag devices have independent clock systems a detailed explanation of devices operation is augmented by analysis of the clock accuracy impac t on the phase estimation error. As the above description shows, the one-way amplitude )| values are directly obtainable from target slave device. However ;. the one-way phase Z A { fn ) values cannot be measured directly.
10103'} In the embodiment, the ranging base band signal is the same as the one depicted in FIG. 1. Howev er , for the sake of simplicity, it is assumed herein that the ranging base band signal consists of only two frequency components each containing multiple periods of cosine or sine waves of different frequency. Fx and F2 . Note that Fx = j dF2 = f2 . The number of periods in a first freq uency component is L and the number of periods in a second frequency component is P. Note that L may or may not be equal to P, because for 1) ~ constant each frequency component can have different number of periods. Also, there is no time gap between each frequency component, and both ja d F2 start from the initial phase equal to zero.
JO 104] Figures A, 3B and 3C depict block diagrams of a master or a slave unit (tag) of an RF mobile tracking and locating system, Fosc refers to the frequency of the device system clock (crystal oscillator 20 in FIG. 3A). All frequencies generated within the device are generated from this system clock crystal oscillator. The following definitions are used: M is a master device (unit); AM is a tag (target) device (unit). The tag device is operating in the transponder mode and is referred to as transponder (AM) unit.
[0105] In the preferred embodiment the device consists of the RF front-end and the RF back-end, base-band and the multi-path mitigation processor. The RF back-end, base-band and the multi-path mitigation processor are implemented in the FPGA 150 (see FIGs. 3B and 3C). The system clock generator 20 (see FIG. 3A) oscillates at; FyS(;: - 20 MHz; or
ω ν-, - 2π χ 20 >: lit. This is an. ideal frequency because in actual devices the system clocks frequencies are not alwa s equal to 20 MHz: j¾r - F^./1* ; F£ - F(m? ■
Note that r M
Figure imgf000019_0001
jOliMJ If should be noted that other than 20 MHz Fosc frequencies can be used without any impact on system performance.
[0107] Both units' (master and tag) electronic makeup is identical and the different modes of operations are software programmable. The base band ranging signal is generated in digital format by the master' FPGA 150, blocks 155 - 180 (see FIG. 2B). It consists of two frequency components each containing multiple periods of cosine or sine waves of different frequency. At the beginning, t = 0 , the FPGA 150 in a master device (FIG, 38) outputs the digital base-band ranging signal to its up-converter 50 via I/Q DACs 120 and 125. The FPGA 150 starts with Fl frequency and after time T start generating F2 frequency for time duration of
[0108 Since crystal oscillator's frequency might differ from 20 MHz the actual frequencies generated by the FPGA will be *]/" and FtyM . Also, time II will be Ί β * and T, will be Τ βΜ . IT is also assumed that 7' s , 7 , „ F are such that Ftfu * Ί βΜ ~ s Ί and
- I S - Κ,γ" * Ί]β* - > where both F \ & F2Tt are integer n umbers. Thai means that the initial phases of Ft arid F3 are equal to zero.
[01091 Since- all frequencies are generated from the system crystal oscillator 20 clocks, the master' base-band I/Q DAC(s) 120 and 125 outputs are as follows:
Fi = r M 20 x 106 x KFi and F2 = γΜ 20 χ 10* χ ΚΛ , where K and are constant coefficients. Similarly, the output frequencies TX LO and HX LO from frequency synthesizer 25 (LO signals for mixers 50 and 85) can be expressed through constant coefficients. These constant coefficients are the same for the master (M) and the transponder (AM) - the difference is in the system crystal oscillator 20 clock frequency of each device.
[0110] The master (M) and the transponder (AM) work in a half-duplex mode.
Master's RF front-end up-converts the base-band ranging signal, generated by the multi-path mitigation processor, using quadrature up-converter (i.e., mixer) 50 and transmits this up- converted signal. After the base-band signal is transmitted the master switches from TX to RX mode using RF Front-end TX/RX Switch 15. The transponder receives and down-converts the received signal back using its RF Front-end mixer 85 (producing First IF) and ADC 140
(producing Second IF).
[ 1 Thereafter, this second IF signal is digitally filtered in the Transponder RF back- end processor using digital filters 190 and further down-converted to the base-band ranging signal using the RF back-end quadrature mixer 200, digital I/Q filters 210 and 230, a digital quadrature oscillator 220 and a summer 270. This base-band ranging signal is stored in the transponder's memory 170 using Ram Data Bus Controller 195 and control logic 180.
|0112| Subsequently, the transponder switches from RX to TX mode using RF front- end switch 15 and after certain delay tRTX begins re-transmitting the stored base-band signal.
Note that the delay is measured in the AM (transponder) system clock. Thus, ¾τ ~ ½x /?AM■ The master receives the transponder transmission and down-converts the received signal back to the base-band signal using its RF back-end quadrature mixer 200, the digital I and Q fi iters 210 and 230, the digital quadrature oscillator 220 (see FIG. 3C).
[0113] Thereafter, the master calculates the phase difference between /' and , in the received (i.e., recovered) base-band signal using multi-path mitigation processor arctan block 250 and phase compare block 255. The amplitude values are derived from the RF back-end RSSS block 240, [01141 For improving the estimation accuracy it is always desirable to improve the SNR of the amplitude estimates from block 240 and phase difference estimates from block 255. In the preferred embodiment the multi-path mitigation processor calculates amplitude and phase difference estimates for many time instances over the ranging signal frequency component duration ( .?' f }. These values, when averaged, improve SNR. The SNR improvement can be in an order thai is proportional to , where N is a number of instances when amplitude and phase difference values were taken (i.e., determined).
[Oil SJ Another approach to the SNR improvement is to determine, amplitude and p ase difference values by applying matching filter techniques over a period, of time. Yet, another approach would be to estimate the phase and the amplitude of the received (i .e., repeated) base band ranging signal frequency components by sampling them and integrating over period T≤ Tf against the original (i.e., sent by the master/reader) base- band ranging signal frequency components in the I/Q form. The integration has the effect of averaging of multiple instances of the amplitude and the phase in the I/Q format. Thereafter, the phase and the amplitude values can be translated from the I/Q format to the i( „^ and zi(/„) format.
[0116} Let's assume that at / 0 under master' multi-path processor control the master base-band processor (both in FPGA 1.50) start the base-band ranging sequence.
¾ 00 = r" x ¾· x C ) f y/ < Ί]βΗ ;
;A ( - rM x ¾c x f
Figure imgf000021_0001
Ί βΜ
where Tf ≥ 2\βΜ .
The phase at master's DAC(s) 1.20 and 125 ontpnts are as follows;
Figure imgf000021_0002
i > Μ + /&Γ
Note that DACs 120 and 125 have internal propagation delay, t sc , that does not depend upon the system clock.
|0l i7j Similarly, the transmitter circuitry components 15, 30, 40 and 50 will introduce additional delay, t l x , that does not depend uport the system clock.
[0118] As a result, the phase of the transmitted RF signal by the master can be calculated as follows;
Figure imgf000022_0001
(01191 The RF signal from the master (M) experiences a phase shift * that is a function of the multi-path phenomena between the master and tag.
[0120] The #>Ml,LT values depend upon. the transmitted frequencies, e.g. I d P . The transponder (AM) receiver' is not able to resolve each path because of limited (i.e., narrow) bandwidth, of the RF portion of the receiver. Thus, after a certain time, for example, 1
microsecond (equivalent to -300 meters of flight), when all reflected signals have arrived at the
formulas apply;
Figure imgf000022_0002
<^¾ = ;.' : xi¾c, xfc;(i;?M) - ¾. (/■- - c-' ) + ^--·4ΐ+^Γ+ί¾ )+ «P& A
[01211 In the AM (transponder) receiver at the first down converter, element 85, an output, e.g. first IF, the phase of the signal is as follows:
Ψη" (n - rH x x ..ϊτ (' ~ ~ ' ))~
Figure imgf000022_0003
^; Γ + ¾ rx (0) - ^.RXJ ( ),i0'fi < / ?/? ! + c + 4- + ^ ;
(f) = f M x m< . x fe, j ?M ) + , it - 7 /r-; - C - - } + ,:ir (t - - ^ ))
^! r + ·ίΧ(0) ~ .RXJ( ),f > r^M + C + + .10 ·
[0122] Note that the propagation delay 4 in the receiver RF section (elements 15 and 60 - 85) does not depend upon the system clock. After passing thiough RF Front-end filters and ampli fiers (elements 95-110 and i 25) the first IF signal is sampled by the RF Back-end ADC 140. It is assimied that ADC 140 is under-sampling the input signal (e.g., first IF). Thus, the ADC also acts like a down-converter producing the second IF. The first IF filters, amplifiers and the ADC add propagation delay time. At the ADC output (second IF):
Figure imgf000023_0001
Hi" / /; M ÷ : >
?M x *½e fc- + *·„ e - ?;?M - - 4 ~ W ~ ^ - >.;L> + .« - · · ···
(0123) In the FPGA 150 the second IF signal (from roe ADC output) is filtered by the RF Back-end digital filters 190 and further down-converted back to base-band ranging signal by the third down-converter (i.e., quadrature mixer 200, digital filters 230 and 210 and digital quadratare oscillator 220), summed in the summer 270 and is stored in the memoiy 170. At the third down-converter output (i.e., quadrature mixer);
Figure imgf000023_0002
Ψ Τ + - <°> ~ ^ ax... (*» ~ ? . α.κ (0) ~ KX (0),
Figure imgf000023_0003
^T" + Tx ({¾ -■ ^¾.Kx,i (0) - Ψ Ι- πκ (0) - >:4 *·,· : <»X
/ > Ίβ 4- C 4- + 4 /£, 4 /** 4 ¾, ?AM + 10-*
101241 Note that propagation delay f = fnR β -A in the FIR section 190 does not depend upon the system clock.
(0125| After RX->TX delay the stored (in memory 170) base-band ranging signal from the master ( ) is retransmitted. Note that RX->TX delay ~ XTX β ΑΜ .
Figure imgf000024_0001
Figure imgf000024_0002
jOI26| By the time the signal from the transponder reaches the master' (M) receiver antenna ti e RF si nal from transponder (AM) experiences another phase shift that is a function of the multi-path. As discussed above, this phase shift happens after a certain time period when all reflected signals have arrived at the master' receiver antenna:
Figure imgf000025_0001
2 1.0'- < / < z; ?- + & c + /;· - / f , + ^ + »fl + + +,:
Figure imgf000025_0002
[0127] In the master receiver the signal from transponder goes through the same down- conversion process as in the transponder receiver. The result is the recovered base-band ranging signal that was originally sent by the master.
For the first frequency component j:
Figure imgf000025_0003
[01281 $w tbe second frequency component 2 :
Figure imgf000026_0001
|0129| Substitutions:
- ? : -f /
where Tn is the propagation delay through roaster (M) and transponder (AM) circuitry.
(° ^ 9 y. («) - (0 ~ where: ^ M.AM (0)is the 1,0 phase shift at time t=0, from master (M) and transponder (AM) frequency mixers, including ADC(s).
Also: K SYN IX i + ;;>, + λ.",..·.
[01301 First frequency component Fl:
Figure imgf000027_0001
[ 1311 First frequency component Fl continued:
Figure imgf000027_0002
0132| Second frequency component F2;
Figure imgf000028_0001
&SYMJX (~ ~ x ~ i- x ~ l.ux ~*mP ~ ''mxfi ~ *ou ~ '.>:>' ~ 1ΆΪ. ~ f it f ~ '-si : ~ *<¾t '
Figure imgf000028_0002
/ 7; ?M÷7;} 2*10··*
j0133| Second frequency component F2, continued:
Figure imgf000028_0003
2xi(r
[0134] Further substituting:
Figure imgf000028_0004
where a is a constant. 0135J Then the final phase equations is: rM x ¾■ x (.£,·.· {/ - r^^^ ))' 2 ^BB j.AM ( ) + a,
2 10* </ < T ^ + l), M^i M x ¾c x (/^ (7: ?^ } ÷ A1; ( - 7\βΜ - ΓΛ w Λ.Α,})*· 2 x ΪΓ + φ^.^ (θ) ·ί· α, 0136| From the equation (5);
„ .
Figure imgf000029_0001
where ί- 2, 3, ; and 2χΔΦ F is equal to 2 χ {fpt-V!J - φ^-7 }.
[0137} For example, the difference 2 x (^ ~ (pphls ) at time instances tl and t2;
2 x Η : - 2 x Λϊ" - 2 ΛΦ .. ,Λ -
Figure imgf000029_0002
2 χ < ί, < + /.. M..Aii iz > + ¾ y...,,, + 2 x i0"ft
{91381 To .find 2χΔΦ;..,;. difference we need to know Tri n ;
¾ . y ··· c ; 4- + 4 . i + 4χ· · ½¾44;.;ί ■·,; = 4 + ^4 + 4-S + 4c 5 ¾m/*AM . where Tls u and j:V/ are propagation delays through the master (M) and transponder
(AM) TX and RX circuitries that are measured by placing devices in the ioop-back mode. Note that the master and the transpooder devices can measure Γ,,¾ ί and Tls automatically; and we also know the ?RTX value.
[0J.39J From the above formulas and ikrx value Tn M ..MS can be determined and consequently, for a given tl3 and the 2 x ΔΦ p_ ,f. value can be found as follows;
2 x Δ .:: i#
Figure imgf000029_0003
) - V&JBCW (fi > " 4 x *
¾,(?/?M) - A; /. - ; 7 ,J - ; ;/; -;- A^J^ --"
2 x 10"* < 5< i;4! + Ί)-, M...,,« ;¾ = /, + Ίβ [A; /, - κ, - fc> - )x r M - k .;; - A, }X r;? Λί '
Figure imgf000030_0001
2 x if) < i, < + τβ Ai.. ;/j :== ?s /;/;"
2 X &Φ f ,. ^ = AI;< >Y t ) - coV (t. ) - γ :M X ¾e x [ ·;;/, - K t. ~ IK, ~ K )x - z;i Μβ» - ΐίβ ΑΜβ^β ~ ( κνκβΜ% <<Ό 2 it)-* < ί, < + /;,_, :<-.. - *, + /;/?"\
Or, assuming that βΜ - β** ~ 1 :
2 ΔΦ,. ;i. = v*_mcm (/2 ) - ^Biascov - M χ¾ χ^/, -KFlt -{KF; -K xii: -!^ (6A) 2 x 10" fi ·<: ti < Ί] -·- 7'β .f.^ ,■■■■■ tx + ;
[01 01 From me equation (6) it can be concluded mat at operating frequency(s) ranging signal(s) complex amplitude values can be found from processing the returned base-band ranging signal.
[0141] The initial phase value 2 x <p ' :JLT can be assumed to be equal zero because the subspace algorithms are not sensitive to a constant phase offset. If necessary, the 2 χ φ^1 value (phase initial value) can be found by determining the TOA (Time Of Arrival) using the narrow-bandwidth ranging signal method as described in U.S. Patent- No.7,561,048, incorporated herein by reference in its entirety. This method estimates the ranging signal round trip delay, which is equal to 2 x κιτβ' and the 2 χ <p Ji:i value can be found from the following equation:
2 x ψν : ι ··· 2 x β* x γ M x ω&κ x fe^ TX K, jx
Or:
2 x φ ' it - 2 x ;¾;Ni. x {f sm J5: + A\ )x (Τηΐ \
[0142 j In the preferred embodiment, the returned base-band ranging signal phase values ,<f( o ( ) are calculated by the multi-path processor's arctan block 250. To improve
SNR, the multi-path mitigation processor phase compare block 255 calculates
2 x ΔΦ ;. fK = (t.m ) - Rtax)V (i„) for many instances n (n.~ 2,3,4 ,,,,,...) using the equation (6A), and then average them out to improve SNR. Note that
2 \^<tn<Tf + TD M_AM-tm = +Tf. [01.431 From the equations 5 and 6 it becomes apparent that the recovered (i.e., received) base-band ranging signal has the same frequency as the original base-band signal thai was sent by the master. Thus, there is no frequency translation despite the fact that the master (M) and the transponder (AM) system clocks can differ. Because the base-band signal consists of several frequency components, each component is consists of multiple periods of a sinusoid, it is also possible to estimate the phase and the amplitude of the received ranging signal by sampling the received base-band signal individual component frequency with the corresponding original (i.e., sent by the master) base-band signal individual frequency component and integrating the resulting signal over period T≤ Ί .
[0144] This operation generates complex amplitude values i ( ; ) of recei ed ranging signal in the I/Q format. Note that each base-band signal individual frequency component that was sent by the master has to be shifted in time b the 7 , is .. w . The integration operation produces effect of averaging out the multiple instances of the amplitude and the phase (e.g., increasing the SNR). Note that the phase and the amplitude values can be translated from
Figure imgf000031_0001
[01.45} This .method of sampling, integrating over period of T≤ Tf and subsequent conversion from the 1 Q format to the U (./,, J and Z A ( „ ) format can he implemented in the phase compare block 255 in FIG. 3C. Thus, depending upon the block's 255 design and implementation, either the method of the preferred embodiment, based on. the equation (5), or an alternative method, described in this section, can be used.
[0146] Although the ranging signal bandwidth is narrow, the frequency difference /„ - /, can be relatively large, for example, in an order of several megahertz. As a result, the receiver's bandwidth has to be kept wide enough to pass all of the . j : fn ranging signal frequencies components. This wide receiver bandwidth impacts the SNR, To reduce the receiver effective bandwidth and improve the SNR, the received ranging signal base-band frequency components can be filtered by the RF back-end processor in FPGA. 150 by the digital narrow bandwidth filters tuned for each individual frequency component of the received baseband ranging signal. However, this large number of digital filters (the number of filters equals to the number of individual frequency components, n) puts additional burden on the FPGA resources, increasing its cost, size and power consumption. [0:1.471 In the preferred embodiment only two .narrow bandwidth digital filters axe used; one filter is always timed for fx frequency component and the other filter can be tuned for all other frequencies components: f2 : f„ . Multiple instances of ranging signal are sent by the master. Each instance consists of only two frequencies: t : f2 ', fi '■ f3 ',./[ '■
Figure imgf000032_0001
'■ f„ ·
Similar strategies are also possible.
[0148] Please note that it is also entirely possible to keep the base-band ranging signal components to only two. (or even one) generating the rest of the frequency components Dyad justing the frequency synthesisers, e.g. changing K syii . It is desirable that LO signals for up- converters and down-converters mixers are generated using the Direct Digital Synthesis (DDS) technology. For high VHP band frequencies this can present an undesired burden on the transceiver/FPGA hardware. However, for lower frequencies this might be a useful approach. Analog .frequency synthesizers can also be used, but may take additional time to settle after frequency is changed. Also, in case of analog synthesizers, two measurements at the same frequency would have to be made in order to cancel a phase offset that might develop after changing the analog synthesizer's frequency.
[0i49 The actual 7'r, ¾ ... ,y that is used in the above equations is measured in both: the master (M) and the transponder (AM) systems clocks, e.g. TfS w and ? sx are counted in the transponder (AM) clocks and T u is counted in the master (M) clock. However, when 2 x Δ is calculated both: Tts m and /RTX are measured (counted) in master ) clock. This introduces an error;
2 xA<¾iM ^ rM x<¾cx(/¾ AM(fi ^ - ?AM)+ ?M ··· /? '·;.} (7)
S 1501 The phase estimation error (7) impacts the accuracy. Therefore, it is necessary to ininimize this error. If0Α =βΑΜ, in other words, all master(s) and transponders (tags) system clocks are synchronized, then the contribution from the tRTX time is eliminated.
[0151] In the preferred embodiment, the master and the transponder units (devices) are capable of synchronizing clocks with any of the devices. For example, a master device can serve as a reference. Clock synchronization is accomplished by using the remote control
communication channel, whereby under FPGA 150 control, the frequency of temperature compensated crystal oscillator TCXO 20 is adjusted. The frequency difference is measured at the output of the summer 270 of the master device while the selected transponder device is transmitting a carrier signal. [0152 j Thereafter, the master sends a command io the transponder to increase/decrease TCXO frequency. This procedure may be repeated several times to achieve greater accuracy by minimizing frequency at the summer 270 output. Please note that in an ideal case the frequency at the summer 270 output should become equal to zero. An alternative method is to measure the frequency difference and make a correction of the estimated phase without adjusting the transponder* TCXO frequency.
[0153J While j * ·-- - ?VM can be considerably reduced there is a phase estimation error when βΜ≠■ 1 , In this case the margin of error depends upon a long term stability of the reference device (usually master ( )) dock generator. In addition, the process of clock synchronization may take considerable amount of time, especially with large number of units in the field. During the synchronization process the track-locate system becomes partially or fully inoperable, which negatively impacts the system readiness and performance. In this case the abovementioned method that does not require the transponder' TCXO frequency adjustment is preferred.
[0154] Commercially available (off the shell) TCXO components have high degree of accuracy and stability. Specifically, TCXO components for the GPS commercial applications are very accurate. With these devices, the phase error impact on locating accuracy can be less than one meter without the need for frequent clock synchronization.
[0155] After narrow bandwidth ranging signal multi-path mitigation processor obtains the returned narrow bandwidth ranging signal complex amplitude A RT (jls ) , the further processing (i.e., execution of super-resolution algorithms), is implemented in the software-based component, which is a part of the multi-path mitigation processor. This software component can be implemented in the master (reader) host computer CPU and/or the microprocessor that is embedded in the FPO A 1 50 (not shown). In the preferred embodiment the multi-path mitigation aigorithm(.s) software component is executed by the master host computer CPU.
[0156] The super-resolution a!gorithrn(s) produce estimation of (2 x rK )
"frequencies"., e.g. τκ values. At the final step the multi-path mitigation processor selects t with the smallest value (i.e., the DLOS delay time),
[0157| In certain cases here the ranging signal narrow bandwidth requirements are somewhat relaxed, the DLOS path can be separated from MP paths by employing a continuous (in time) chirp. In the preferred embodiment this continuous chirp is Linear Frequency
Modulation (LFM). However, other chirp waveforms can be also used. [01.581 Let's assume thai under muiii-path. mitigation processor control a chirp with
B
bandwidth ot B and duration of I is transmitted, lhat gives a chirp raie oi β = ·2Λ?— radians per
1
second. Multiple chirps are transmitted and received back. Note that chirps signals are generated digitally with each chirp started at the same phase.
[0159] In the multi-path processor each recei ved single chirp is aligned so thai the returned chirp is from the middle of the area of interest
[0160) The chirp waveform equation is:
- ( f ) - exp S ( t + fit 11 , where <¾ is the initial frequency for 0 < t < T .
For a single delay round-trip τ , e.g. no multi-path, the relumed signal (dip) is sii - r) .
0161.1 The multi-path mi ligation processor then "deramps" the ,v(/- r) by performing complex, conjugate mix with the originally transmitted chirp. The resulting signal is a complex sinusoid:
fr {/) = exp(~ r)exp{ ~2ίβτί )&κρ{ ϊ.βτ2 ) , (8) where εχ (~ί% } is the amplitude and 2βτ is the frequency and 0≤t≤l Note thai the last terra is a phase and it is negligible.
[01621 h* case of multi-path, the composi te decamped signal consists of multiple complex sinusoids :
Figure imgf000034_0001
where L is the number of ranging signal paths, including the DLOS path and 0 <i < l
j0163| Multiple chirps are transmitted and processed. Each chirp is individually ireaied/processed as described above. Thereafter, the multi-path mitigation processor assembles result of individual chirps processing:
J,i/ H ¾)>(/ - np ) I x I ¾exp(- iH¾T,)exp(~ ϊ2βτ > j (10) L " J L*3 J where is the number of chirps, lit) ~ < ^ "' 1
\(kr> T '"" ? J I , p - T + ; *" is the dead, time zo e between two consecutive chirps; 20rk are artificial delay "frequencies". Again, the most interesting is the lowest "frequency", which corresponds to the DLOS path delay. {0364J In the equation (10) can be thought of as N samples of a sum of complex sinusoids at times:
0 < ta≤ T tl = ta + p;t2 = ta + 2p = ta ÷ (N™ l)p; e 0 : w ~ l;
Thus, the number of samples can be a multiple of N, e.g. <xN;a 1,2,
[0165] From the equation (10) the multi-path mitigation processor produces a^ complex amplitude samples in time domain that are used in further processing (i.e., execution of super-resolution algorithms). This further processing is implemented in the software component, which is a part of the multi-path mitigation processor. This software component can be executed by the master (reader) host computer CPU and/or by the
microprocessor that is embedded in the FPGA 150 (not shown), or both. In the preferred embodiment the multi-path mitigation algorithm(s) software is executed by the master host computer CPU.
[0166] The super-resolution algorithm(s) produce estimation of 2βτΙ; "frequencies", e.g. TK values. At the final step the multi-path mitigation processor selects τ with the smallest value, i.e. the DLOS delay time,
[0167$ An explanation will be given of a special processing method, called the
"threshold technique," which can serve as an alternative to the super-resolution algorithms, in other words, it is used to enhance reliability and accuracy in distinguishing DLOS path from other MP paths using the artificially generated synthetic wider bandwidth ranging signal.
[0168] The frequency domain base-band ranging signal shown in FIG. 1 and FIG, 1 A can be
Figure imgf000035_0001
V i t
It is readily verified that " ' is periodic with period V , and for an integer k, that
* (A - ^ ! s which is the peak value of the signal. Where n=N in FIG- 1 and FIG, 1 A.
^ 1 f
0169J FIG, 4 shows two periods of ' ·' for the case where ~ 11 and ■> ~ 250 kHz. The signal appears as a sequence of pulses of height H-T = 3 separated by ~ 4 microseconds. Between the pulses is a sinusoidal waveibon with varying amplitude and 2N zeros. The wide bandwidth of the signal can be attributed to the narrowness of the tall pulses, it
A' \ f
can be also seen that the bandwidth extends from zero frequency to ' ::: 2.75 MHz. [01.701 The basic idea of the thresholded method that is used hi the preferred embodiment is to enhance the artificially generated synthetic wider bandwidth ranging reliability and accuracy in distinguishing DLOS path from other MP paths. The threshold method detects when the start of the leading edge of a wideband pulse arrives at a receiver. Because of filtering in the transmitter and receiver, the leading edge does not rise instantaneously, but rises out of the noise with smoothly increasing slope. The TOA of the leading edge is measured by detecting when the leading edge crosses a predetermined threshold T.
[0171] A small threshold is desirable because it gets crossed sooner and the error delay τ between the true start of the pulse and the threshold crossing is small. Thus, any pulse replica arriving due to multi-path has no effect if the start of the replic a having a delay greater than r . However, the presence of noise places a limit on how small the threshold T can be. One way to decrease the delay τ is to use the derivative of the received pulse instead of the pulse itself, because the derivative rises faster. The second derivative has an even faster rise. Higher order derivatives might be used, but in practice they can raise the noise level to an unacceptable value, so the thresholded second derivative is used.
0172| Although the 2.75 Mffcs wide signal depicted in FIG. 4 has a fairly wide bandwidth, it is not suitable for measuring range by the abovementioned method. That method requires transmitted pulses each having a zero-signal precursor. However, it is possible to achieve that goal by modifying the signal so that the sinusoidal waveform between the pulses is essentially cancelled out. In the preferred embodiment it is done by constructing a waveform which closely approximates the signal on a chosen interval between the tall pulses, and then subtracting it from the original signal.
[0173] The technique can be illustrated by applying it to the signal in FIG. 1. The two black dots shown on the waveform are the endpoints of an interval I centered between the first two pulses. The left and right endpoints of the interval I, which have been experimentally determined to provide the best results, are respectively at:
1 - 1 1 1
t. = = ≡ 191.3 usee
(2N + 1 / 23 x 250, 000
( 12)
1 1 i . i
/ . t, = 3, 808. / osee
" Af 250, 000 23 x 250, 000
[0174] An attempt to generate a function g ( t) which essentially cancels out the signal
,s (i ) on this interval, but does not cause much harm outside the interval, is performed. Since the expression ( 11) indicates that s{ t) is the sinusoid siu jr (2N + \ ) fi modulated by 1/sin xAff , first a function ( t) which closely approximates Ι/ήη π&β on the interval I. is found, aad i a form g (f ) as (he product;
g(t) h(t)sm x(2N + l)&fi (13) h (t) is■ generated by the following sum;
Figure imgf000037_0001
where
Figure imgf000037_0002
and the coefficients are chosen to minimize the least-square, error
Figure imgf000037_0003
over the interval 3.
[01751 The solution is readily obtaisied by taking partial derivatives of J with respeci to the <¾ and setting them equal io zero. The result is the linear system of M + 1 equations
« Λ,, j - 0J,2,..., (17) {0176 that can b solved for the ak , where
[0
Figure imgf000037_0004
{ M ^ (19)
(0178J Using the definition of the .functions≠k (>) given by (12)
Figure imgf000037_0005
[0179J T.lieg( f) is subtracted from .?{?) to get a function r (t) , which should essentially cancel s {t) on the interval 1. As indicated in the Appendix, an appropriate choice for the upper limit M for the summation in the equation (20) is M™ 2N - J , Using this value and the results from the Appendix, (21 )
¾ ÷ y. ; cos 2π%£φ 4- c sin.2π { N + ) &ft where
Figure imgf000038_0001
(0180J From the equation (17) it is seen that a lota! of 2N-†-3 frequencies (rncludin zero- frequency DC term) are required to obtain the desired signal r {7 ) , FIG. 5 shows the resulting signal r(t) for the original signal s(t) shown in FIG. 1, where N - 11. In this case the construction of r(i) requires 25 carriers (including the DC terra ¾ ).
jOJ8l| The important characteristics of r(t) as constructed above arc as follows:
[ΘΊ82] 1. The lowest frequency is zero Hz and the highest frequency is (2Λ- - 1 }Δ Hz, as seen from (14). Thus, the total bandwidth is (2N + \)&f Hz.
{0183J 2. Aii carriers are cosine functions (including DC) spaced Af apart, except for one carrier, which is a sine function located at frequency (Λ? )4/ ,
[0184] 3. Although the original signal s(t ) has period l/Af , r(t) has period 2/4 ' - The first half of each period of r ( t ) , which is a full period of s ( t ) , contains a cancelled portion of the signal, and the second half-period of r (7) is a large oscillatory segment. Thus, cancellation of the precursor occurs only in every other period of s (t ) .
[01851 This occurs because the cancelin function g (/) actually strengthens s (t) in every other period of s (?) . The reason is thai g (t ) reverses its polarity at every peak of s ( t) , whereas s (t) does not. A method of making every period of .v ( ) contain a cancelled portion to increase processing gain by 3 dB is described below,
[0186] 4. The length of the cancelled portion of s(() is about 80-90% of .1 / f .
Therefore, Af needs to be small enough to make this length long enough to eliminate any residual signal from previous non-zero portions of ?· (/ } due to multi-path. |0187f 5. Immediately following each zero portion of r(t) is the first cycle of an oscillatory portion. In the preferred embodiment, in the TOA measurement method as described above, the first half of this cycle is used for measuring TOA, specifically the beginning of its rise. It is interesting to note that the peak value of this first half-cycle (which will be called the main peak) is somewhat larger than the corresponding peak of s ( t) located at approximately the same point in time. The width of the first half-cycle is roughly inversely proportional to Λ'Δ * ,
[01881 6. A large amount of processing gain can be achieved by;
[01891 (a) Using the repetitions of the signal r(/ } , because r (i) is periodic with period 2/Af . Also, an additional 3 dB of processing gain is possible by a method to be described later.
[01 0J (b) Narrowband filtering. Because each of the 2JV + 3 carriers is a narrowband signal, the occupied bandwidth of the signal is much smaller than that of a wideband signal spread out across the entire allocated band of frequencies.
[0191] For the signal r ( t) shown in Figure 5, where N = 1 1 and Δ/' = 250 kHz, the length of the cancelled portion of s (t ) is about 3.7 microseconds or 1 , 1 10 meters. This is more than enough to eliminate any residual signal from previous non-zero portions of r (t) due to the multi-path. The main peak has value of approximately 35, and the largest magnitude in the precursor (i.e., cancellation) region is about 0.02, which is 65 dB below the main peak. This is desirable for getting good performance using the TOA measurement thresholded technique as described above.
[01 21 Use of fewer carriers is depicted in FIG, 6, which illustrates a signal that is generated using Af = 850 kHz, N ** 3, and M « 2JV + 1 = 7, for a total of only 2N+ 3 = 9 carriers. In this case, the period of the signal is only 2 Af ~ 2,35 microseconds as compared, to the signal in FIG. 5, where the period is 8 microseconds. Since this example has more periods per unit time, one might expect that more processing gain could be achieved.
01 3J However, since fewer carriers are used, the amplitude of the main peak is about 1/3 as large as before, which tends to cancel the expected extra processing gain. Also, the length of the zero-signal precursor segments is shorter, about 0.8 microseconds or 240 meters. This should still be enough to eliminate any residual signal from previous non-zero portions of r ( t) due to the multi-path. Note that the total bandwidth of (2N ' + 1 } Δ/ ~ 5.95 MHz is about the same as before, and that the width of the half-cycle of the main peak is also roughly the same. Since fewer carriers are used, there should be some extra processing gain when each carrier is narrowband filtered at the receiver. Moreover, the largest magnitude in the precursor (i.e., cancellation) region is now about 75 dB below the main peak, a 10 dB improvement from the previous example,
\n94 Transmission at RF Frequencies: τφ to this point r (t) has been described as a base-band signal for purposes of simplicity. However, it can be translated up to RF, transmitted, received, and then reconstituted as a base-band signal at the receiver. To illustrate, consider what happens to one of the frequency components <&k in the base-band signal r (/) traveling via one of the multi-path propagation paths having index j (radian/sec frequencies are used for notational simplicity):
bk cos (okt (at baseband in transmitter)
l cos(<».+ i¾ ) (translated by frequency ω up to RF)
r .- > i (23) a s- cos ( ω 4 mk } / - -r .. ) ÷ ι (at receiver antenna)
( ? - t , ) φ. 4 # (translated by frequenc - a> to baseband)
Figure imgf000040_0001
[0195} It is assumed here that the transmitter and receiver are frequency synchronized. The parameter bk is the km coefficient in expression (21) for r(f) . The parameters r and φ are respectively the path delay and phase shift (due to dielectric properties of a reflector) of the h propagation path. The parameter Θ is the phase shift occurring in the down-conversion to baseband in the receiver. A similar sequence of functions can be presented for the sine component of the equation ( 1 ).
[01961 It is important to note that as long as the zero-signal precursors in r{t) have length sufficiently larger than the largest significant propagation delay, the final base-band signal i the equation (20) will still have zero-signai precursors. Of course, when all frequency components (index k) over all paths (index/) are combined, the base-band signal at the receiver will be a distorted version of r(t) , including ail phase shifts.
[01 7J Sequential Carrier Transmissions and Signal Reconstruction are illustrated in FIG. 1 and FIG. 1A. It is assumed that the transmitter and the receiver are time and frequency synchronized, the 2N + 3 transmitted carriers need not be transmitted simultaneously. As an example, consider the transmission, of the signal whose base-band representation is that of FIG. l and MG. 6. [01981 Ifl FIG. 6, N ::: 3, and suppose each of the 9 frequency components for 1 millisecond are sequentially transmitted. The start and the end times for each frequency transmission are known at the receiver, so it can sequentially start and end its reception of each frequency component at those respective times. Since the signal propagation time is very short compared to 1 millisecond (it will normally be less than several microseconds in the intended applicat ion), a small portion of each recei ved frequency component should be ignored, and the receiver can easily blank it out.
[0199] The entire process of receiving 9 frequency components can be repeated in 9- millisecond blocks of additional reception to increase the processing gain. In one second of total reception time there would be about 1 11 such 9-millisecond blocks available for processing gain. Additionally, within each block there w ould be additional processing gain available from
0.009/(2/4/ )≡ 383 main peaks.
[0200] It is worth noting that in general the signal reconstruction can be made very economical, and will inherently permit all possible processing gain . For each of the 2iV + 3 received frequencies:
1. Measure the phase and amplitude of each 1 -millisecond reception of that frequency to form a sequence of stored vectors (phasors) corresponding to that frequency.
2 , Average the stored vectors for that frequency.
3. Finally, use the 2Ν·* 3 vector a verages for the 2N + 3 frequencies to reconstruct 1 period of base-band signal having duration 2/Δ/' , and use the reconstruction to estimate signal TOA.
[0201] This method is not restricted to 1 -millisecond transmissions, and the length of the transmissions may be increased or decreased. However, the total time for all transmissions should be short enough to freeze any motion of the receiver or transmitter.
(02021 Obtaining Cancellation on Alternate Half-Cycles of r(t): by simply reversing the polarity of the canceling function g ( ) , cancellation between the peaks of ,s (/ } is possible where r(t) was formerly oscillatory. However, to obtain cancellation between all peaks of
.v{ r) , the function g {( ) and its polarity reversed version must be applied at the receiver, and this involves coefficient weighting at the receiver.
[0203] Coefficient Weighting at the Receiver: if desired, the coefficients bk in the equation (21) are used for construction of r {t ) at the transmitter and may be introduced at the receiver instead. This is easily seen by considering the sequence of signals in the equation (20) in which the final signal is the same if % is introduced at the last step instead of at. the beginnin Ignoring noise, the values are as follows:
cos<¾./ (at baseband in transmitter)
eosf ω-ϊ )/ (translated by frequency o> up to RF)
Figure imgf000042_0001
a; cos (translated by frequency - ω to baseband)
a,h, cosi e¾ (i - r,. ) + + 0 j (weighted by coefficient bk at baseband)
02041 The transmitter can then transmit all frequencies with the same amplitude, which simplifies its design. It should be noted, that this method also weights the noise at each frequency, the effect of which should be considered . It should also be noted that coefficient weighting should be done at the receiver in order to effect the polarity reversal of g (?) to get twice as many useable main peaks,
[02051 Scaling of Δ/'to Center Frequencies m Channels; to meet the FCC requirements at the VHF or lower frequencies a channelized transmission with constant channel spacing will be required. In a channelized transmission band with constant channel spacing that is small compared to the total allocated band, which is the case for the VHF and lower frequencies band(s), small adjustments to Λ/' , if necessary, permit all transmitted frequencies to be at channel centers without materially changing performance from original design values. In the two examples of base-band signals previously presented, all frequency components are multiples of Af/2 , so if the channel spacing divides Δ/ /2 , the lowest RF transmitted frequency can be centered in one channel and all other frequencies fall at the center of channels.
0206'} Irs some Radio Frequency ( F)-based identification, tracking and locating systems in addition to performing the distance measurement function, both: the Master Unit and the Tag Unit also perform voice, data and control communication functions. Similarly, in the preferred embodiment both the Master Unit and the Tag perform voice, data and control communication functions in addition to the distance measurement function.
[0207] According to the preferred embodiment, the ranging signal(s) are subject to the extensive sophisticated signal processing techniques, including the multi-path mitigation.
However, these techniques may not lend themselves to the voice, data and control signals. As a result, the operating range of the proposed system (as well as other existing systems) may be limited not by its ability to measure distance reliably and accurately, but by being out of range during voice and/or data and/or control communications. [02081 ΪΛ other Radio Frequency (RF)-based identification, tracking and locating systems the distance measurement functionality is separated from the voice, data and control communication functionality. In these systems separate RF Transceivers are used to perform voice, data and control communication functions. The drawback of this approach is system increased cost, complexity, size, etc,
[02091 To avoid abovementioned drawbacks, in. the preferred embodiment, a narrow bandwidth ranging signal or base-band narrow bandwidth ranging signal several individual frequency components are modulated with the identical data/control signals and in case of voice with digitized voice packets data. At the receiver the indmdual frequency components that have the highest signal strength are demodulated and the obtained information reliability may be further enhanced by performing ''voting" or other signal processing techniques that utilize the information red undancy .
[02101 This method allows to avoid the "null" phenomena, wherein the incoming RF signals from multiple paths are destmctively combining with the DLOS path and each other, thus significantly reducing the received signal strength and associated with it SNR, Moreover, such method allows to find a set of frequencies at which the incoming signals from multiple paths are constructively combining with DLOS path am! each other, thus increasing the received signal strength and associated with it SNR.
[0211] As mentioned earlier, spectrum estimation-based super-resolution algorithms generally use the same model: a linear combination of complex exponentials and their complex amplitudes of frequencies. This complex amplitude is given by equation 3 above.
[0212] All spectrum estimation-based super-resolution algorithms require a priori knowledge of number of complex exponentials, i.e., the number of multipath paths. This number of complex exponentials is called the model size and is detennined by the number of multi-path components L as shown in equations 1 - 3. However, when estimating path delay, which is the case for RF track-locate applications, this information is not available. This adds another dimension, i.e., the model size estimation,, to the spectrum estimation process via super- resolution algorithms.
[02J3J It has been shown (Kei Sakaguchi et al., Influence of the Model Order
Estimation Error in the ESPRIT Based High Resolution Techniques) that in case of model size underestimation the accuracy of frequency estimation is impacted and when the model size is overestimated the algorithm generates spurious, e.g., non-existent, frequencies. Existing methods of model size estimation such as AIC (Akaikes Information Criterion), MDL (Minimum Description Length), etc. have a high sensitivity to correlation between signals (complex exponentials). But in the case of RP mullipath, this is always the case. Even, for example, after Forward-Backward smoothing algorithms are applied, there will always be a residual amount of correlation.
[02 41 1» the Sakaguchi paper, it is suggested to use an overestimated mode! and differentiating actual frequencies (signals) from spurious frequencies '(signals) by estimating these signals power (amplitude) and then rejecting the signals with very low power. Although this method is an improvement over existing methods, it is not guaranteed. The inventors implemented the Kei Sakaguchi et al. method and ran simulations for more complex cases with a larger model size. It was observed that, in some cases, a spurious signal may have amphtude that is very close to actual signals amplitude.
[02151 All spectrum estimation-based. super«resolution algorithms work by splitting the incoming signal complex amplitude data into two sub-spaces', the noise sub-space and signals sub-space. If these sub-spaces are properly defined (separated), then the model size is equal to the signal sub-space size (dimension).
[0216] In one embodiment, the model size estimation is accomplished using an "F" statistic. For example, for ESPRIT algorithm, the singular value decomposition of the estimate of the covariance matrix (with forward/backward correlation smoothing) is ordered in ascending order. Thereafter, a division is made whereby the (n÷1.) eigenvalue is divided by the n-th eigenvalue. This ratio is an "F" random variable. The worst case is an "F" random variable of (1 ,1) degree of freedom. The 95% confidence interval for a "F" random variable with (1,1) degrees of freedom is 161. Setting that value as a threshold determines the model size. Note also that for the noise subspace, the eigenvalues represent an estimate of the noise power.
[0217 j This met hod of applying " F>! statistics to the ratio of the eigenvalues is a more accurate method of estimating the model size. It should be noted that other degrees of freedom in "F" statistics can be also used for threshold calculation and consequently model size estimation.
[0218] Nevertheless, in some cases, two or more very closely spaced (in time) signals can degenerate into one signal because of real-world measurement imperfections. As a result, the above mentioned method will underestimate the number of signals, i.e., the model size. Since model size underestimation reduces the frequency estimation accuracy, it is prudent to increase the model size by adding a certain number. This number can be determined experimentally and or' from simulations. However, when signals are not closely spaced, the model size will be overestimated. [0239f In such cases spurious, i.e., non-exislent, frequencies may appear. As noted earlier, using signal amplitude for spurious signals detection does not always work because in some cases a spurious signal(s) was observed to have amplitude that is very close to actual signal(s) amplitude. Therefore, in addition to the amplitude discrimination, filters can be implemented to improve spurious frequencies elimination probability.
[0220] The frequencies that are estimated by super-resolution algorithms are artificial frequencies (equation 2). In fact, these frequencies are individual paths delays of the multipath environment. As a result, there should be no negative frequencies and all negative frequencies that are produced by a super-resolution algorithm are spurious frequencies to be rejected.
{022 ! J Furthermore, a DLOS distance range can be estimated from the complex
2( f )
amplitude / " ' values obtained during measurements using methods that are different from super-resolution methods. While these methods have lower accuracy, this approach establishes range thai is used to discriminate delays, i.e., frequencies. For example, he ratio of
A [ZJ (2 ^A )]
{0222J in ^ intervals where the signal amplitude ^ ^ x ^ is close to maximum, i.e., avoiding nulls, provides a DLOS delay range. Although actual DLOS delay can be up to two times larger or smaller, this defines a range that helps to reject spurious results.
[0223] In the embodiment, the ranging signal makes a round-trip. In other words, it travels both ways: from a master/reader to a target/slave and from the target/slave back to the master/reader:
[0224} Master transmits a tone: x e " , where ω is an operating frequency in the operating band and a is the tone signal amplitude.
[0225] At the target's receiver, the received signal (one-way) is as follows:
S, {() ~a x YX„ " xe*
(25)
[0226 j Where: N is number of signal paths in the multipath environment;. K0 and 'ΰ are amplitude and time-of-flig'ht of the DLOS signal; , ^<> > t! , ari ^**0 can be positive or negative.
(0 :::- <* * e~J"* A x e "!ί){β! (26) [0227| Where:
Figure imgf000046_0001
is one way multipath RF channel transfer function in the frequency domain; and Α (ω )≥ 0
[0228 Target retransmits the received signal:
Figure imgf000046_0002
[0229} At the master receiver, the round-trip signal is
Figure imgf000046_0003
[0231}
[02321 ,i'"' {m}x e
[0233 j On the other hand from equations (26) and (28):
Figure imgf000046_0004
0234] Where: " r a ·-' is ro iidtrip multipath RF channel transfer function in the frequency domain.
[02351 From equation 29, the round trt multipath channel h as a l arger number o f paths than one-way channel multipath because the
Figure imgf000046_0005
expression in addition to the
N paths delays, includes combinations of these paths delays, for example:
r0 + T2 r, i r, τχ + r3 .., etc.
[0236] These combinations dramatically increase the number of signals (complex exponentials). Hence the probability of very closel spaced (in time) signals will also increase and may lead to significant model size underestimation. Tims, it is desirable to obtain one-way multipath RF channel transfer function.
[0237] In preferred embodiment, the one-way amplitude values are directlv obtainable from target/slave device. However, the owe-way phase values '"" " / « ·' 'cannot be measured directly. It is possible to determine the phase of the one- way from the roundtrip phase measurements observation: ∑ K - * e
» ~o and » ** '
[0239| However, for each value of ω , there are two values of phase 1 * ·' such that
[0240J "■" = «*·
[0241} A detailed description of resolving this ambiguity is shows below. If the ranging signal different frequency components are close to each other, then for most part the one-way phase can be found by dividing the roimdtrip phase by two. Exceptions will include the areas that are close to the "null", where the phase can undergo a significant change even with small frequency step. Note: the "null" phenomena is where the incoming RF signals from multiple paths are destructively combining with the DLOS path and each other, thus significantly reducing the received signal strength and associated with it SNR.
[0242} Let h ?) be the one-way impulse response of a communications channel. The corresponding transfer function in the frequency domain is
Figure imgf000047_0001
[0243} (30) 0244| where Α {ω) > 0 is the magnitude and a{&) is the phase of the transfer function. If the one-way impulse response is retransmitted back through the same channel as it is being received, the resulting two-way transfer function is
[02451 <?W^(^ ) - Η* {ω) ~ Λ* {φ)^ (3 | )
[0246} where B ( a ≥Q . Suppose the two-way transfer function 0(ω) is known for all ω in some open frequency interval {<% , ω, ) . Is it possible to determine the one-way transfer function Η (ω) defined on (ω,, βχ, ) that produced (?{<»)?
[0247} Since the magnitude of the two-way transfer function is the square of the one- wav magnitude, it is clear that
[02481 ^fflW'*M (32) [0249} However, in trying to recover the phase of the one-way transfer function from observation of G ( ω) , the siiitation is more subtle. For each value of ω , there are two values of phase ω) such that ef"iM ~ £:βί* (33) [02511 A large number of different solutions .might be generated by independently choosing one of two possible phase values for each different frequency ω .
[0252] The following theorems, which assume that any one-way transfer function is continuous at all frequencies, help resolve this situation.
[0253 j Theorem /: Let. / be an open interval of frequencies ^containing no zeros of the two-way transfer function ( ( ω) ~ S(<»)r!i ">! . Let </(« ) ·~ ^jB ( <»')<r? ( > be a continuous function on where β{ω) - 2γ{ω) . Then J {&) and -<1{ω) are the one-way transfer functions which produce G(&>) on /, and there are no others.
[0254] Proof: One of the solutions for the one-way transfer function is the function
H { 6>) - -y' if <^ϊ)ί>·ί'< 'S!', , continuous on / since it is difterentiable on /, and where β{ ω) - 2a (ω) . Since G (<¾>≠<) on A H c&) and ./(«*) are nonzero on /. Then,
Figure imgf000048_0001
[0256} Since H { ω) and J ( ω) are continuous and nonzero on /, their ratio is continuous on Λ hence the right Side of (34) is continuous on /. The conditions
β{ω) - 2<χ(ω) - 2γ{ω) imply that For each ω ζ-Ι , α(ω)~γ(ω) is either 0 or π . However, a Ua) - γ {ω) cannot switch between these two values without causing a discontinuity on the right side of (34), Thus, either a { &>) - γ (ω) 0 for all ω e / , or a (ω) - γ {ω) ~ π for all (¾€/ . In the first case, we get J fo.>) ~ Η(ω) , and in the second we get J{ } ~ ~H (&} ,
[02S7J This theorem proves that to get a one-way solution on any open interval I containin no zeros of the transfer function G(m)■■■■■■■ 5(»)e"'w- we form the function choosing the values of / &) satisfying β(ω) - 2γ (ω) in such a way as
Figure imgf000048_0002
to make ./ (ω) continuous. Since it is known that there is a solution having this property, namely H (<¾>) , it is al ways possible to do this.
{0258J An alternate procedure for finding a one-way solution is based on the following theorem: [0259} Theorem 2: Let H (co) ~ .4 he a one-way transfer function and let ./ be an open interval of frequencies ^containing no ¾ros of Η{ω), Then the phase function «(&>) of H ( ή must be continuous on /.
[026 } Proof; Let be a frequency in the interval /. in FIG.7, the complex value
Η (ω9 } has been plotted as a point in the complex plane, and by hypothesis, H (ω(ί )≠ 0. Let ε > 0 be an arbitrarily small real number, and consider the two angles of measure ε shown in the FIG.7, as well as the circle centered at H(<»0) and tangent to the two rays OA and OB. By assumption, H(i») is continuous for all ω . Thus, if ω is sufficiently close to o0 , the complex value Η{<ω) will Ue in the circle, and it is -seen, that |#((y)-«(fi¼)j<£. Since s>0 was chosen arbitrarily, we conclude that «{<¾>)-·>«("<¾} as (¾>→■(¾, so that the phase function α(ω) is continuous at *¾ .
[0261} Theorem 3: Let / be an open interval of frequencies <» containing no zeros of th two-way transfer function G ( co )- ( ω ) e'^r) , Let J [ω)~ -^B (m)e!;{m be a function on / where and γ{&>) is continuous o ./. Then J{co) and ~-J{m) are the one-way
Figure imgf000049_0001
transfer functions which produce G(m) on /, and there are no others.
[0262} Proof: The proof is similar to the proof of Theorem 1. We know that one of the solutions for the one-way transfer function is the function H {(») - {(:o)eja'"n , where β{ω)~-2σ.{ω). Since ( ω}≠0 on/, Η{&) and J (co) are nonzero on /. Then,
Figure imgf000049_0002
[02641 By hypothesis y &) is continuous on / and by Theorem 2 a(m) is also continuous on /. Thus, α(ω - χ(ω) is continuous on i. The conditions
β(ω) = 2 (ω) = 2γ ω) imply that for each <¾€/, a{m) -- γ{ω) is either 0 or π , However, {ω) - y{<®) cannot switch between these two values without becoming discontinuous on /. Thus, either α(ω)-γ(ω ΰ for all eve I , ox (ω)- '(ω)~χ for all <»€./. in the .first case, we get /(<¾:>) - H ( ω) , and in the second { ω) = -H(co) . [02651 Theorem 3 tells us that to get a one-way solution on any open interval containing no xeros of the transfer function (?(.») - B (t»)e i9i, we simply form the function
J { m) -
Figure imgf000050_0001
values of γ{ω) satisfying β(ω) ~ 2γ (ω) in such a way as to make the phase function γ{ω) continuous. Since it is known that there is a solution having this property, namely H(o) , it is always possible to do this.
[0266} Although the above theorems show how to reconstruct the two one-way transfer functions which generate the two-way function G{m), they are useful only on a frequency interval / containing no zeros of G(e ) . in general, Ο (ω) will be observed on a frequency interval (ί»,, ω? ) which may contain zeros. The following is a method that might get around this problem, assuming that there are only a finite number of zeros of G ( ω) in ( ωι , ω2 ) , and that a one-way transfer function has derivati es of all orders on (<», , co2 ) , not ail of which are zero at any given frequency & :
{02671 Let " (ω) be a one-way function that generates ^ l^) on the interval (6^ , m^ t and assume that has at least one zero on * -■ . The zeros or ' will separate ( ωι , ω2 ) -nto ^ finite number of abutting open frequency intervals JL , J, , ... , J„ . On each such interval the solution ^ (ω) or Η (ω) wuj De foun(j using either Theorem 1 or Theorem 3. We need to "stitch together" these solutions so that the stitched solution is either ^ (ω^ or ^{ω) across ail of 5 ' ~ ·· ' . in order to do this, we need to know how to pair the solutions in two adjacent subintervals so that wre aren't switching from ^ ^ ' to ^ or from ^ (ω)†0
^ in moving from one subinterval to the next.
{02681 We. illustrate the stitching procedure starting with the first two adjacent open sub intervals J:. and < , . These subintervais will abut at a frequency <». which is a zero of G(a>)
(of course, &{ is not contained in either subinterval). By our above assumption about the properties of a one-way transfer function, there must be a minimum positive integer n such that Hi,: '' (&>, )≠ , where the superscript («) denotes the n1' derivative. Then the limit of the »w derivative of our one-way solution in J, as ω→ o from the left will be either H{r) coi ) or ~ (" * (<»« ) according to whether our solution in Jt is (ω) or ~Η { ω) . Similarly, the limit of the ft* derivative of our one-way solution in as ίύ→ω{ from the right will be either
H("' (<Wj ) or -//'·"' (es/j ) according to whether our solution in z is H(«>} or -H (<2>) . Since H : (fi>, )≠ 0 , the two limits will be equal if and only if the solutions in Jx and J, are both {m) or both -H(«>) . If the left and right hand limits are unequal we invert the solution in subinterval J . Otherwise, we don't.
026"9| After nwerting the solution in subinterva! (if necessary), we perform an identical procedure for subintervals J2 and J- , inverting the solution in subinterval J, (if necessary). Con tinning in this fashion, we ev entually build up a complete solution on the interval (β¾,ή¾ ) .
[0270] It would be desirable that high-order derivatives of H(o)not be required in the above reconstruction procedure, since they are difficult to compute accurately in the presence of noise. This problem is unlikely to occur, because at any zero of G (t») it seems very likely that the first derivative of (<y) will be nonzero, and if not, very likely that the second derivative will be nonzero.
[0271 J In a practical scheme, the two-way transfer function Ο(ω) will be measured at discrete frequencies, which most be close enough together to enable reasonably accurate computation of derivatives near the zeros of G {<o) .
[0272] For RF-based distance measurements it is necessary to resolve an unknown number of closely spaced, overlapping, and noisy echoes of a ranging signal with a priori known shape. Assuming that ranging signal is a narrow-band, in frequency domain this RF phenomena can be described (modeled) as a sum of a number of sine waves, each per multipath component, and each with the complex attenuation and propagation delay of the path.
[0273] Taking the Fourier transform of the above mentioned sum will express this multipath model in the time domain. Exchanging the role of time and frequency variables in this time domain expression, this multipath model will become harmonic signals spectrum in which the propagation delay of the path is transformed to a harmonic signal.
[0274] The super (high) resolution spectral estimation methods are designed to distinguish closely-placed frequencies in the spectrum and used for estimating the individual frequencies of multiple harmonic signals, e.g., paths delays. As a result, path delays can be accurately estimate .
[0275] The super resolution spectral estimation makes use of the eigen-structure of the covariance matrix of the baseband ranging signal samples and covariance matrix intrinsic properties to provide a solution to an underlying estimation of individual frequencies, e.g. paths delays. One of the eigen-structure properties is thai the eigenvalues can. be combined and consequently divided into orthogonal noise and signal eigenvectors, aka subspaces. Another eigen-structure property is the rotation-invariant signal subspaces property.
[0276] The subspace decomposition technology (MUSIC, rootMUSIC, ESPRIT, etc.) relies on breaking the estimated covariance matrix of the observed data into two orthogonal subspaces, the noise subspace and the signal subspace. The theory behind the subspace decomposition methodology is that the projection of the observable onto the noise subspace consists of only the noise and the projection of the observable onto the signal subspace consists of only the signal
[0277} The spectral estimation methods assume that signals are narrow-band, and the number of harmonic signals is also known, i.e., the size of the signal subspace needs to be known. The size of the signal subspace is called as the model size. In general, it cannot be known in any detail and can change rapidly - particularly indoors - as the environment changes. One of the most difficult and subtle issues when applying any subspace decomposition algorithm is the dimension of the signal subspace that can be taken as the number of frequency components present, and which is the number multipath reflections plus the direct path. Because of real-w orld measurement imperfections there always will be an error in the model size estimation, which in turn will result in loss of accuracy of frequencies estimation, i.e., distances.
[0278] To improve Che. distance measurement accuracy, one embodiment includes six features that advance the state of the art in the methodology of subspace decomposition high resolution estimation. Included is combining two or more algorithms estimating individual frequencies by using different eigen-structure properties that further reduces the delay path determination ambiguity.
[0279J Root Music finds the individual frequencies, that when the observable is projected onto the noise subspace, minimizes the energy of the projection. The Esprit algorithm determines the individual frequencies from the rotation operator. And in many respects this operation is the conjugate of Music in that it finds the frequencies thai, when the observable is projected onto the signal subspace, maximizes the energ of the projection. [02801 The model size is the key to both of these algorithms, and in practice, in a complex signal environment such as seen in indoor ranging - the model size which provides the best performance for Music nd Esprit are in general not equal, for reasons that will be discussed below,
[0 811 for Music it is preferable to err on the side of identifying a basis element of the decomposition as a "signal eigen value" (Type I Error). This will minimize the amount of signal energy that is projected on the noise subspace and improve the accuracy. For Esprit - the opposite is true - it is preferable to err on the side of identifying a basis element of the decomposition as a "noise eigenvalue." This is again a Type I Error. This will minimize the impact of noise on the energy projected onto the signal subspace. Therefore, the model size for Music will, in general, be somewhat larger than that for Esprit.
[0282} Secondly, in. a complex signal environment, there arise occasions where, with the strong reflections and the potential that the direct path is in fact much weaker than some of the muitipath reflections, the model size is difficult to estimate with sufficient statistical reliability. This issue is addressed by estimating a "base" mode! size for both Music and Esprit and the processing the observable data using Music and Esprit in a window of model sizes defined by the base model size for each. This results in multiple measurements for each measurement.
[02831 The first feature of the embodiment is the use of the F-statistic to estimate the model size (see above) . The second feature is the use of different Type I Error probabilities in the F-statistic for Music and Esprit. This implements the Type I Error differences between Music and Esprit as discussed above. The third feature is the use of a base model size and a window in order to maximize the probability of detecting the direct path.
[02841 Because of the potentially rapidly changing physical and electronic
environment, not every measurement will provide robust answers. This is addressed by using cluster analysis on multiple measurements to provide a robust range estimate. The fourth feature of the embodiment is the use of multiple measurements.
[0285] Because there are multiple signals present, the probability distribution of the multiple answers resulting from multiple measurements, each using multiple model sizes from both a Music and Esprit implementation, will be multimodal. Conventional cluster analysis will not be sufficient for this application. The fifth feature is the development of multimodal cluster analysis to estimate the direct range and equivalent range of the reflected muitipath components. The sixth feature is the analysis of the statistics of the range estimates provided by the cluster analysis (range and standard deviation and combing those estimates that are statistically
identical. This results in a more accurate range estimate.
ji)286J The aboveraentidned methods can be also used in wide bandwidth ranging signal location-finding systems.
|0287| For the derivation of r(t) in the thresholded method, starting with expression (20), we obtain
g(f)~\ + si &ft js /r(2iv"-
M
- αϋ sin π 2 N 4- 1 ) Aft + ^ . sin π(2Ν -i- l} f1 si kxbjl ~ 0 sin r(2iV 4·1)Δ^
(A!)
+ ^ cos n ( 2 N -f\~k } Δ// -^ · ai cos ( 2.N + Ai
Figure imgf000054_0001
-÷- 2 f <¾ cos 2?r ( N + ■ j) Δ/ί - J I ¾ cos 2 Λ· ( Af + { ~† ) Δ/
|0288| where the trigonometric identity sin x sin ! - f cos (x - y) - ·§· cos( + y ) is used.
[02891 Except for <¾, the coefficients ¾ are zero for even /t. The reason for this is that on the interval L the function l sin that we are trying to approximate by r(/)is even about the center 'of i, but the basis' functions $iakn&fi for even k, k≠G, are odd about the center of Λ, hence are orthogonal to l s // on I. Thus, we can make the substitution /f -2« + ! and let M be an odd positive integer, hi fact, we will let M - 2N+ 1. This choice has been experimentally determined to provide a good amount of cancellation of the oscillations in the interval I.
Figure imgf000054_0002
[0290] Now we make the substitution k = N- n in the first summation and k = N+ n + 1 in the second summation to obtain g (( ) -
Figure imgf000055_0001
[0291 J Subtracting ^ ' from results in r(t) = s
Figure imgf000055_0002
" ∑ i¾*'*- cos Znkisft. - ¾ sin 2J? (N -ί- -)Δ/'
[0292} Now Set.
¾ -2~¼,{V..J: for* ...1.2,...:.V
(A5) >ξ s iff for' - N - i, <V - 2, ... , 2<¥ + ί
[0293} Then (A4) can be written as
Figure imgf000055_0003
0294J The present embodiments relate to a positioning/locating method m wireless communication and other wireless networks that substantially obviate one or more of the disadvantages of the related art. The present embodiments advantageoosl improve the accuracy of tracking and locating functionality in multiple types of wireless network by utilizing rauiii- path mitigation processes, techniques and algorithms, described in U.S. Patent No.7,872,583, These wireless networks include Wireless Personal Area Networks (WPGAN) such as ZigBee and Blue Tooth, wireless local area network (WLAN) such as WiFi and UWB, Wireless Metropolitan Area Networks, (WMAN) typically consisting of multiple WLANs, W Max being the primary example, wireless Wide Area Networks ( W AN) such as White Space TV Bands, and Mobile Devices Networks (MDN) that are typically used to transmit voice and data. MDNs are typically based on Glojbal System, for Mobile Coinmunkations (GSM) and Personal
Communications Service (PCS) standards. A more recent MDN is based on the Long Term Evolution (LTE) standard. These wireless networks are typically comprised of a combination of devices, including base stations, desktop, tablet and laptop computers, handsets, smartphones, actuators, dedicated tags, sensors as well as other communication and data devices (generally, all these devices are referred to as "wireless network, devices").
029SJ Existing location and positioning information solutions use multiple
technologies and networks, including GPS, AGPS, Cell Phone Tower Triangulation, and Wi-Fi. Some of the methods used to derive this location information include RF Fingerprinting, RSSI, and TDOA. Although acceptable for the current E 1 1 requirements, existing location and ranging methods do not have the reliability and accuracy required, to support the upcoming E 11 requirements as well as LBS and/or RTLS applications requirements, especially indoors and urban environments,
[02 61 T e methods described in U.S. Patent No. 7,872,583 significantly improve the ability to accurately locate and track targeted devices within a single wireless network or a combination of multiple wireless networks. The embodiment is a significant improvement to the existing implementation of tracking and location methods used by wireless networks that use Enhanced Cell- ID and OTDOA (Observed Time Difference of Arrival), including DL-OTDOA (Downlink OTDOA), U-TDOA, UL-TDOA and others
[0297] Cell ID location technique allows estimating the position of the user (UE- User Equipment) with the accuracy of the particular sector coverage area. Thus, the attainable accuracy depends on the cell (base station) sectoring scheme and antenna beam-width. In order to improve accuracy the Enhanced Cell ID technique adds RTT (Round Trip Time)
measurements from the eNB. Note: Here, the RTT constitutes the difference between
transmission of a downlink DPCH - Dedicated Physical Channel, (DPDCH)/DPCCH: Dedicated Physical Data Channel/Dedicated Physical Control Channel) frame and the beginning of a corresponding uplink physical frame. In this instance the abovementioned frame(s) act as a ranging signal. Based on the information of how long this signal propagates from eNB to the UE, the distance from eNB can be calculated (see Figure 10). [02981 ΪΛ the Observed Time Difference of Arrival (OTDOA) technique the time of arrival, of the signal earning from neighboring base stations (eNB) is calculated The UE position can be estimated in the handset (UE-based method) or in the network (NT-based, UE-assisted method) once the signals from three base stations are received. The measured signal is the CPICH (Common Pilot Channel). The propagation time of signals is correlated with a locally generated replica. The peak of correlation indicates the observed time of propagation of the measured signal. Time difference of arrival values between two base stations determines a hyperbola. At least three reference points are needed to define two hyperbolas. The location of the UE is in the intersection of these two hyperbolas (see Figure i i).
[0299] Idle Period Downlink (IPDL) is further OTDOA enhancement. The OTDOA- IPDL technique is based on the same measurements as the regular OTDOA Time measurements are taken during idle periods, in which serving eNB ceases its transmissions and allows the UE within the coverage of this cell to hear pilots coming from distant eNB(s). Serving eNB provides idle periods in continuous or burst mode. In the continuous mode, one idle period is inserted in every downlink physical frame (10 ms). In the burst mode, idle periods occur in a pseudorandom way. Further improvement is obtained via Time Aligned IPDL (TA-IPDL). Time alignment creates a common idle period, during which, each base station will either cease its transmission or transmit the common pilot. The pilot signal measurements will occur in idle period. There are several other techniques that may further enhance the DL OTDOA-IPDL method, for example Cumulative Virtual Blanking, UTDOA (Uplink TDOA), etc. All these techniques improve the ability to hear other (non-serving) eNB(s).
[0300] One significant drawback of the OTDOA based techniques is that the base stations timing relationships must be known, or measured (synchronized), for this method to be viable. For trasynchronixed UMTS networks the 3GPP standard offers suggestion of how this timing may be recovered. However, networks operators are not implementing such solution. As a result, an alternative that uses the RTT measurements in lieu of the CPICH signal measurements was proposed (see U.S. Patent Publication No. 20080285505, John Carlson ei at, SYSTEM AND METHOD FOR NETWORK TIMING RECOVERY IN COMMUNICATIONS NETWORKS).
[0301] All abovementioned methods/techniques are based on the terrestrial signals time of arrival and/or time difference of arrival measurements (RTT, CPICH, etc.). An issue with such measurements is that these are severely impacted by the multi-path. This, in turn, significantly degrades the abovementkmed methods/techniques locate/track accuracy (see Jak b Marek Borkowski: Performance of Cell ID+RTT Hybrid Positioning Method for UMTS).
{0302] One Multi-path mitigation, technique uses detections/measurements from excess number of e B(s) or Radio Base Stations (RBS). The minimum is three, but for multipath mitigation the number of RBS's required is at least six to eight (see METHOD AND
ARRANGEMENT FOR DL-OTDOA (DOWNLINK OBSERVED TIME DIFFERENCE OF ARRIVAL) POSITIONING IN A LTE (LONG TERM EVOLUTION) WIRELESS
COMMUNICATIONS SYSTEM, WO/2010/104436). However, {he probability of .an UE hearing from this large number of eNB(s) is much lower than from three eNB(s). This is because with large number of RBS (eNBs) there will be several ones that are far away from the UE and the received signal from these RBS (es) may fall below the UE receiver sensitivity level or the received signal will have low SNR.
[03031 I case of RF reflections (e.g., multi-path), multiple copies of the RF signal with various delay times are superimposed onto the DLOS (Direct Liae of Site) signal Because CPICH, uplink DPCCH DPDCH and other signals that are used in various CELL ID and OTDOA methods/techniques, including the RTT measurements, are of a limited bandwidth the DLOS signal and reflected signals cannot be differentiated without proper multi-path processing/mitigation; and without this multi-path processing these reflected signals will induce an error in the estimated time difference of arrival (TDOA) and time of arrival (TOA) measurements, including RTT measurements.
[0304] For example, 3 G TS 25.515 v.3.0.0 (199-10) standards define the RTT as ".... the difference between transmission of a downlink DPCH frame (signal) and the reception of the beginning (first significant path) of the corresponding uplink DPCCH/DPDCH frame (signal) from UE". The standard does not define what constitutes this "first significant path". The standard goes on noting that "The definition of the first significant path needs further elaboration". For example, in heavy multipath environment it is a common occurrence whereby the DLOS signal, which is the first significant path, is severely attenuated (10 dB - 20 dB) relatively to one or more reflected signal(s). If the "first significant path" is detennined by measuring the signal strength, it may be one of the reflected signal(s) and not the DLOS signal. This will result in erroneous TOA/DTOA/RTT measurement s) and loss of locating accuracy.
[0305] In prior wireless networks generations the locating accuracy was also impacted by the low sampling rate of frames (signals) that are used by the locate meihods - RTT, CPCIH and other signals. The current third and following wireless network generations have much higher 'sampling rate. As a result, m these networks the locating accuracy real impact is from the terrestrial RF propagation phenomena (mulii ath).
[0306] The embodiment can be used in all wireless networks that employ reference and/or pilot signals, and/ or synchronization signals, including simplex, half-duplex and full duplex modes of operation. For e ample, the embodiment operates with wireless networks that employ OFDM modulation and/or its derivatives. Thus, the embodiment operates with LTE networks.
[0307] It is also applicable to other wireless networks, including WiMax, WiFi, and White Space. Other wireless networks that do not use reference and/or pilot or synchronization signals may employ one or more of the following types of alternate modulation embodiments as described in U.S. Patent No, 7,872,583: 1 ) where a portion of frame is dedicated to the ranging signal/ranging signal elements as described in U.S. Patent No. 7,872,583; 2) where the ranging signal elements (U.S. Patent No. 7,872,583) are embedded into transmit/receive signals frame(s); and 3) where the ranging signal elements (described in U.S. Patent No. 7,872,583) are embedded with the data.
[0308] These alternate embodiments employ multi-path mitigation processor and multi- path mitigation techniques/algorithms described in U.S. Patent No. 7,872,583 and can be used in all modes of operation: simplex, half-duplex and full duplex.
[030^1 It is also likely that multiple wireless networks will, at the same time, utilize the preferred and/or alternate embodiments. By way of example, a smart phone can have Blue Tooth, WiFi, GSM and LTE functionality with the capability of operating on multiple networks at the same time. Depending on application demands and/or network availability, different wireless networks can be utilized to provide positioning/locating information.
[0310] The proposed embodiment method and system leverages the wireless network reference/pilot and/ or synchronization signals. Furthermore, the reference/pilot signal/ synchronization signals measurements might be combined with RTT (Round Trip Time) measurements or system timing, According to an embodiment, RF -based tracking and locating is implemented on 3 GPP LTE cellular networks, but could be also implemented on other wireless networks, for example WiMax, Wi-Fi, LTE, sensors networks, etc. that employ a variety of signaling techniques. Both the exemplary and mentioned above alternative embodiments employ multi-path mitigation method/techniques and algorithms that are described in U.S. Patent No. 7,872,583. The proposed system can use software implemented digital signal processing. [033 J 1 The system of the embodiment leverages User Equipment (UE),. e.g. eel! phone or smart phone, haidware/sofiware as well as Base Station (Node B)/enhaiiced Base Station (eNB) hardware/software. A base station generally consists of transmitters and receivers in a cabin or cabinet connected to antennas by feeders. These base stations include, Micro Cell, Pico Cell, Macro Cell, Umbrella Cell, Cell Phone towers, Routers and Femtocells. As a result, there will be little or no incremental cost to the UE device and overall system. At the same time the locate accuracy will be significantly improved.
[0312| The improved accuracy comes from the multipara mitigation that is provided by the present embodiments md U.S. Patent No. 7,872,583. The embodiments use multi-path mitigation algorithms, network reference/ pilot and/ or synchronization signals and network node (eNB). These might be supplemented with RTT (Round Time Trip) measurements. The multi- path mitigation algorithms are implemented in UE and/or base station (eNB), or both: UE and eNB
[03J 3 j The embodiments advantageously use the multi-path mitigation
processor/algorithms (see U.S. Patent No. 7.872,583) that allow separating the DLOS signal and reflected signals, even when DLOS signal is significantly attenuated (10 dB - 20 dB lower) relatively to one or more reflected signals. Thus, the embodiments significantly lower the error in the estimated ranging signal DLOS time-of-fhght and consequently TOA, RTT and DTOA measurements. The proposed multi-path mitigation and DLOS differentiating (recognizing) method can be used on all RF bands and wireless systems/networks. And it can support various modulation/demodulation techniques, including Spread Spectrum techniques, such as DSS (Direct Spread Spectrum) and FH (Frequency Hopping).
[0314] Additionally, noise reduction methods can be applied in order to further improve the method's accuracy. These noise reduction methods can include, but are not limited to, coherent summing, non-coherent summing, Matched filtering, temporal diversity techniques, etc. The remnants of the multi-path interference error can be further reduced by applying the postprocessing techniques, such as, maximum likelihood estimation (e.g., Viterbi Algorithm), minimal variance estimation (Kalman Filter), etc..
[0315] In present embodiments the multi-path mitigation processor and multi-path mitigation techniques/algorithms do not change the RTT, CPCIH and other signals and/or frames. The present embodiments leverage wireless network reference, pilot and; or
synchronization signals that are used to obtain the channel response/estimation. The invention uses the channel estimation statistics that is generated by UE and/or eNB (see Iwamatsu et aL APPARATUS FOR ESTIMATING PROPAGATION PATH CHARACTERISTICS, US
2003/00 156; US 716745 .82).
[0316] LTE networks use specific (non-data) reference/ pilot and/ or synchronization s signals (known signals) that are transmitted in every downlink and uplink subframe, and might span entire cell bandwidth. For simplicity from now on we will refer to reference/ pilot and synchronization signals as reference signals. An example of the LTE reference signals is in FIG. 9 (these signals are interspersed among LTE resource elements). From FIG. 2, reference signals (symbols) are transmitted every sixth subcarrier. Further, reference signals (symbols) are staggered in both time and frequency. In total, reference signals are covering every third subcarrier.
[0317] These reference signals are used in the initial cell search by the UE, downlink signal strength measurements, scheduling and handover, etc. Included in the reference signals are UE-specific reference signals for channel estimation (response determination) for coherent demodulation. In addition to the UE-specific reference signals, other reference signals may be also used for channel estimation purposes, (see Chen et al., US patent publication No.
2010/0091826 A.1 ).
10318| LTE employs the OFDM (Orthogonal Frequency Division Multiplexing) modulation (technique), in. LTE the 1ST (inter Symbol Interference) caused by multipath is handled by inserting Cyclic prefix (CP) at the beginning of each OFDM symbol The CP provides enough delay so that delayed reflected signals of the previous OFDM symbol will die out before reaching the next OFDM symbol,
[03191 An OFDM symbol consists of multiple very tightly spaced subcarriers. Inside the OFDM symbol time-staggered copies of the current symbol (caused by multipath.) result in Inter Carrier Interference (ICI). In LTE the ICI is handled (mitigated) by determining the multipath channel response and correcting the channel response in the receiver.
[0320] In LTE the multipath channel response (estimation) is computed in the receiver from subcarriers bearing the reference symbols. Interpolation is used to estimate the channel response on the remaining subcarriers. The channel response is calculated (estimated) in form of channel amplitude and phase. Once the channel response is determined (by periodic transmission of known reference signals), the channel distortion caused by multipath is mitigated by applying an amplitude and phase shift on a subcarrier-by-subcarrier basis (see Jim Zyren, Overview of the 3 GPP Long Term Evolution Physical Layer, white paper). [03211 LTE multipath mitigation is designed Co remove the ISI (by inserting a Cyclic Prefix) and ICl, but not to separate the DLOS signal from reflected signals. For example, lime- staggered copies of the current symbol make each modulated subcarrier signals spread in time, thus causing ICI. Correcting multipath channel response using the abovementioned LTE technique will shrink modulated subcarrier signals in time, but this type of correction does not guarantee that the resulting modulated subcarrier signals (inside the OFDM symbol) are DLOS signals. If DLOS modulated subcarrier signals are significantly attenuated relati vely to delayed reflected sigaal(s), the resulting output signal will be the delayed reflected signal(s) and the DLOS .signal will be lost.
[0322] In LTE compliant receiver, further signal processing includes DFT (Digital Fourier Transformation). It is well known that DFT technique(s) can resolve (remove) only copies of signal(s) that are delayed for times that are longer than or equal to the time that is inversely proportional to the signal and/or channel bandwidth. This method accuracy may be adequate for an efficient data transfer, but not accurate enough for precise distance measurement in a heavy multipath environment. For example, to achieve thirty meters accuracy, the signal and receiver channel bandwidths should be larger than or equal to ten megahertz (1/10 MHz = 100 ns.). For better accuracy the signal and receiver channel bandwidths should be wider - one hundred megahertz for three meters.
[03231 However, CPJCTi uplink DPCCH DPDCH and other signals that are used in various CELL ID and OTDOA methods techniques, including the RTT measurements, as well as the LTE received signal subcarriers have bandwidths that are significantly lower than ten megahertz. As a result, the currently employed (in LTE) method/technique will produce locating errors in the range of 100 meters.
[0324] To overcome the abovementioned limitations the embodiments use a unique combination of implementations of subspace decomposition high resolution spectral estimation methodologies and multimodal cluster analysis. This analysis and related multi-path mitigation method/techniques and algorithms, described in U.S. Patent No. 7,872,583, allow a reliable and accurate separation of DLOS path from other reflected signals paths.
[0325] Compared to nvethods/iechiisques used in the LTE, in a heavy multipath environment this method/techniques and algorithms (U.S. Patent No. 7,872,583) deliver 2 OX to 50X accuracy improvement in the distance measurement via reliable and accurate separation of DLOS path from oilier multi-path (MP) paths. [03261 Methods/techniques and algorithms described hi U.S. Patent No. 7,872,583 require ranging signal complex amplitude estimation. Accordingly, the LTE reference signals used for channel estimation (response determination) as well as other reference signals
(including pilot and/ or synchronization signals, can be also construed as a ranging signal in methods/techniques and algorithms described in U.S. Patent No. 7,872,583. In this case the ranging signal complex amplitude is the channel response that is calculated (estimated) by the LTE receiver in form of amplitude and phase. Is other words, the channel response statistics that is calculated (estimated) by the LTE receiver can provide complex amplitude Information that is required by the method/techniques and algorithms described in U.S. Patent No. 7,872,583.
[0327] In ideal open space RF propagation environment with no multipath the phase change of the received signal (ranging signal), e.g. channel response phase, will be directly proportional to the signal's frequency (a straigh line); and the RF signal time-of-flight
(propagation delay) in such environment can be directly computed from the phase vs. frequency dependency by computing fast derivative of the phase vs. frequency dependency. The result will be the propagation delay constant.
[0328] In this ideal environment the absolute phase value at initial (or any) frequency is not important because the derivative is not affected by the phase absolute values.
[0329] In a heavy multipath environment the received signal phase change vs.
frequency is a complicated curve (not a straight line); and the first derivative does not provide information that could be used for accurate separation of DLOS path from other reflected signals paths. This is the reason for employing multipath mitigation processor and method(s)/techniques and algorithms described in U.S. Patent No. 7,872,583.
[0330] If the phase and frequency synchronization (phase coherency) achieved in a given wireless network/system is very good, then multipath mitigation processor and method(s)/techniques and algorithms described in U.S. Patent No. 7,872,583 will accurately separate DLOS path from odier reflected signals paths and determine this DLOS path length (time-of-ilighi).
[03311 In this phase coherent network/system no additional measurements are required.
In other words, one way ranging (simplex ranging) can be realized.
[03321 However, if the degree of synchronization (phase coherency) achieved in a given wireless network/system is not accurate enough, then in a heavy multipath environment the received signal phase and amplitude change vs. frequency might be very similar for measurements conducted at two or more different locations (distances). This phenomenon might lead to an ambiguity in received signal DLOS distance (time-of-flight) determination.
[0333] To resolve this ambiguity it is necessary to know the actual (absolute) phase value for at least one frequency.
[03341 However, the amplitude and phase vs. frequency dependency that is computed by the LIE receiver does not include an actual phase value because all amplitude and phase valises are computed from the downlink uplink reference signals, e.g. relative to each other. Thus, the amplitude and phase of the channel response that is calculated (estimated) by the LTE receiver needs actual phase value at least at one frequency (subcarrier frequency).
[0335] In LTE this actual phase value can be determined from one or more RTT measurement(s), TOA measurements; or
[0336] from time-stamping of one or more received reference signals, provided that 1.) these time stamps of transmitting these signals by eNB are also known at the receiver (or vice versa), 2) the receiver and eNB clocks are well synchronized in time, and/ or 3) by using multilateration techniques,
[0337] All of the above methods provide the time-of-flight values of one or more reference signals. From the time-of-flight values and frequencies of these reference signals actual phase values at one or more frequencies can be calculated.
[0338] The present embodiments achieve a highly accurate DLOS distance
determination locating in a heavy multipath environment by combining multi-path mitigation processor, method(s)/techniques and algorithms described in U.S. Patent o. 7,872,583 with; ί .) the amplitude and phase vs. frequency dependency that is computed by the LTE UE and/ or eNB receiver or 2) a combination of the amplitude and phase vs. frequency dependency that is computed by the LTE UE and/ or eNB receiver and actual phase value(s) for one or more frequencies obtained via RTT and/or TOA; and/or time-stamping measurements .
[0339] In these cases the actual phase value(s) is affected by the multipath. However, this does not impact the performance of methods/ techniques and algorithms described in U.S. Patent No. 7,872,583.
[0340] In LTE RTT/TOA/TDOA/OTDOA, including DL-OTDOA, U-TDOA, UL- TDOA, etc., measurements can be carried out with the resolution of 5 meters. RTT
measurements are carried during dedicated connections. Thus, multiple simultaneous measurements are possible when UE is in handover state and times when UE periodically collects and reports measurements back to the UE, in which the DPCH frames are exchanged between the UE and different networks (base stations). Similar to RTT, TOA measurements provide the signal's time-of-ffight (propagation delay), but TOA measurements cannot be made simultaneously (Jakub Marek Borkowskt: Performance of Cell ID+R1T Hybrid Positioning Method for UMTS).
[03411 I» order to locate UE on plane DLOS distances have to be determined at least from/to three eN B(s), To locate UE in tru^ee-dimensional space minimum four DLOS distances from/to four eNB(s) would have to be determined (assuming that at least one eNB is not on the same plane).
[0342f An example of UE positioning method is shown in FIG. 1.
[0343] In case of very good synchronization RTT measurements are not required.
|0344| If the degree of synchronization is not accurate enough, then methods like OTD0A, Cell ID + RTT and others, for example AOA (Angle-of-Arrival) and its combinations with other methods, can be used, for the UE locating,
[0345} The Cell ID ÷ RTT track-locate method accuracy is impacted by the multipath (RTT measurements) and the eNB (base station) antenna beamwidth. Base stations antennas beamwidths are between 33 and 65 degrees. These wide beamwidths results in locating error of 50 - 150 meters in urban areas (Jakub Marek Borkowski: Performance of Cell ID+RTT Hybrid Positioning Method for UMTS). Considering that in a heavy multipath environment the current LTE RTT distance measurement average error is approximately 100 meters, the overall expected average locate error of the currently employed by LTE Ceil ID ÷ RTT method is approximately 150 meters.
[0346] One of the embodiments is the UE locating based on the AOA method, whereby one or more reference signals from the UE is used for the UE locate purposes, f involves an AOA determination device location for determining the DLOS AOA. The device can be collocated with the base station and/or installed at another one or more locations independent from the base station location. The coordinates of these locations are presumably known. No changes are required on the UE side.
[0347] This device includes a small antenna arm and is based on a variation of the same multipath mitigation processor, method(s)/teehniqites and algorithms described in U.S. Patent No. 7,872,583. This one possible embodiment has the advantage of precise determination (very narrow beamwidth) of the AO of the DLOS RF energy from, an UE unit,
[0348} In one other option this added device is receive only device. As a result, its size/weight and cost are ver low. [03491 The combination of embodiments in which accurate DLOS distance
measurements are obtained and embodiments in which an accurate DLOS AOA determination can be made will greatly improve the Cell ID + RTT track- locate method precision - 10X or greater. Another advantage of this approach is that the UE location can be determined at any moment with a single tower, (does not require placing UE in soft handover mode). Because an accurate location fix can be obtained with a single tower there is no need to synchronize multiple ceil towers. Another option of determining the DLOS AOA is to use the existing eNB antenna array and the eNB equipment. This option may further lower the cost of implementation of the improved Ceil ID + RTT method. However, because eNB antennas are not designed for the locating applications, the positioning accuracy may be degraded. Also, network operators may be unwilling to implement required changes in base station (software/hardware).
[03501 I» the LTE (Evolved Universal Terrestrial Radio Access (E-UT A); Physical channels and modulation; 3 GPP TS 36,2 i 1 Release 9 technical Specification) Positioning Reference Signals (P SX were added. These signals are to be used by the UE for the DL-OTDA (Downlink OTDOA), positioning. Also, this release 9 requires eNB(s) to be synchronized. Thus, clearing the last obstacle for OTDOA methods (see paragraph 274 above). The PRS improves UE hear-ability at UE of multiple eNBs. Note: the Release 9 did not specify the eN B
synchronization accuracy (some proposals: 100 ns.).
[0351] The U-TDOA/ UL-TDOA are in a study phase; to be standardized in 20 i 1 .
[0352] The DL-OTDOA method (in Release 9) is detailed in the US patent US
2011/0124347 Al (Method and Apparatus for UE positioning in LTE networks, Chen, at al.). The Release 9 DL-OTDOA suffers from the multipath. Some of the multipath mitigation can be achieved via increased PRS signal bandwidth. However, the hade-off is increased scheduling comple ity and longer times between UE positions fixes. Moreover, for networks with limited operating bandwidth, for example 10 MHz, the best possible accuracy is 100 meters, see Chen, Table 1.
[0353J The above numbers are the best possible case. Other cases, especially when the DLOS signal strength is signi ficantly lower ( 10 - 20 dB) compared to the reflected signai(s) strength, resul t in significantly larger (2X - 4X) of the abovementioned locate/ ranging errors.
[0354] Embodiments described herein allow for up t,q 50X ranging/locate accuracy improvement for a given signal bandwidth over the performance achieved by the Release 9 DL- OTDOA method and the UL-PRS method of Chen e al. described in the Background section. Thus, applying embodiments of the methods described herein to the Release 9 PRS processing reduces the locale error down to 3 meters or better in 95% of all possible cases . In addition, this accuracy gain will reduce the scheduling complexity and the time between HE position fixes.
[0355] With the embodiments described herein further improvements for the OTDOA method are possible. For example, the ranging to the serving cell can be determined from other serving cells' signals, thus improvin the neighboring cells hearability and reducing the.
scheduling complexity, including the time between UE positions fixes.
[0356} Embodiments also enable the acc uracy of the U-TDOA method and UL-TDOA from. Chen et al (described in the Background) to be improved up to 50 times. Applying
embodiments to the Chen's UL-TDOA variant,, reduces the locate error down to 3 meters or better in 95% of all possible cases. Moreover, this accuracy gain further reduces the scheduling complexity and the time between UE positions fixes.
[03571 Again, with the present embodiments, Chen's UL-TDOA method accuracy can be improved up to 5 X, Thus, applying the present embodiments io the Chen's U-TDO variant, will reduce the locate error down to 3 meters or belter in 95% of all possible cases. Moreover, this accuracy gain will further reduce the scheduling complexity and the time between UE position fixes.
[0358J The above-mentioned DL-TDQA and U-TDOA/UL-TDOA methods rely on oneway measurements (ranging). Present embodiments and practically all other ranging
technologies require that the PRS and/or other signals used in the proces of one-way ranging would be frequency and phase coherent. The OFDM based systems, like LTE. ate frequency coherent. However, the UE units and e B(s) are not phase or time synchronized by a common source - like UTC, to a couple nanoseconds, e.g. there exists a random phase adder.
[0359} To avoid the phase coherency impact on the ranging accuracy, the embodiment of the multipath processor calculates the differential phase between the ranging signal(s), e.g. reference signals, individual components (subcarriers). This eliminates the random phase term adder.
[0360] As identified above in the discussion of Chen et al., applying the embodiments described herein result in significant accuracy improvement in indoor environments compared to the performance achieved by Chen et al. For example, according to Chen, at al. the DL-OTDOA and/ or U-TDOA/UL-TDOA are mostly for outdoor environments, indoors (buildings, campuses, etc.) the DL-OTDOA and U-TDOA technologies may not perform well. Several reasons are noted (see Chen, #161 - 164), including the Distributed Antenna Systems (DAS) that are commonly employed indoors, whereby each antenna does not have a unique ID. } [036 J 1 The embodiment described below operates with wireless networks that employ OFDM modulation and/or its derivatives; and reference/ pilot/ and or synchronization signals. Thus, the embodiment described below operates with LTE networks and it is also applicable to other wireless systems and other wireless networks, including other types of modulation, with or without reference/ pilot/ and/or synchronization signals.
[03621 The approach described herein is also applicable to other wireless networks, including WiMax, WiFi, and White Space. Other wireless networks that do not use reference / pilot and/ or synchronization signals may employ one or more of the following types of alternate modulation embodiments as described in U.S. Patent No. 7,872,583: 1 ) where a portion of frame is dedicated to the ranging signal/ranging signal elements; 2) where the ranging signal elements are embedded into transmit receive signals frame(s); and 3) where the ranging signal elements are embedded with the data.
[0363] Embodiments of the muitipath mitigation range estimation algorithm described herein (also described in U.S. Patent Nos. 7,969,33 ! and 8,305,215) works by providing estimates of the ranges in the ensemble made up of the direct path (DLOS) of a signal plus the mul tipa th reflections.
1036 | The LTE DAS system produces multiple copies of the same signal seen at various time offsets to a mobile receiver (UE . The delays are used to uniquely determine geometric relationships between the antennas and the mobile receiver. The signal seen by the receiver resembles that seen in a muitipath environment - except the major "muitipath" components result from the sum of the offset signals from the multiple DAS antennas.
[0365] The signal ensemble seen by the receiver is identical to the type of signal ensemble embodiments are designed to exploit - except that in this case the major muitipath components are not traditional muitipath. The present muitipath mitigation processor
(algorithms) is capable of determining the attenuation and propagation delay of the DLOS and each path, e.g. reflection, (see equations 1 - 3 and associated descriptions). While muitipath can be present because of the dispersive RF channel (environment), the major muitipath components in this signal ensemble are associated w ith transmissions from multiple antennas. Embodiments of the present muitipath algorithm can estimate these muitipath components, isolate the ranges of the DAS antennas to the receiver, and provide range data to the location processor (implemented in software). Depending on the antenna placing geometry, this solution can provide both X, Y and X, Y, Z location coordinates. [03661 As a result present embodiments do not require any hardware and/or new network signals) additions. Moreover, the positioning accuracy can. be significantly improved by 1) mitigating the multipath and 2) in case of active DAS the lower bound of positioning error can be drastically reduced, such as reducing from approximately 50 meters to approximately 3 meters.
[03671 It is assumed that the position (location) of each antenna of a DAS is known. The signal propagation delay of each antenna (or relati ve to other antenna) also has to be determined (known),
[03681 For active DAS systems the signal propagation delay may be determined automatically, using the loopback techniques, whereby the known signal is sent round trip and this round trip time is measured. This loopback technique also eliminates the signal propagation delay changes (drift) with temperature, time, etc.
03691 Using multiple macro ceils and associated antennas, Pico cells and micro cells further enhance the resolution by providing additional reference points.
The embodiment described above of individual range estimates in a signal ensemble of multiple copies from multiple antenna can be further enhanced by changes to the signal transmit structure in the following two ways. The first is to time multiplex the transmissions from each antenna. The second approach is to frequency multiplex for each of the antennas. Using both enhancements, time and frequency multiplexing simultaneously, further improve the ranging and location accuracy of the system. Another approach is to add a propagation delay to each antenna. The delay values would be chosen to be large enough to exceed the delay spread in a particular DAS environment (channel), but smaller than the Cyclic Prefix (CP) length so that the multipath caused by additional delays will not result in ISI (Inter Symbol Interference).
[0370] The addition of a unique ID or unique identifier for each antenna increases the efficiency of the resulting solution. For example, it eliminates the need for the processor to estimate all the ranges from the signals from each of the antennas
[0371] In one embodiment utilizing the LTE downlink, one or more reference signal(s) subcarriers, including pilot and or synchronization signal(s) subcarriers, are used to determine subcarriers phase and amplitude that are in turn applied to the multi-path processor for multipath interference mitigation and generation of range based location observables and locate estimate using multilateration and location consistency algorithms to edit out wild points.
[0372] Another embodiment takes advantage of the fact that the LTE uplink signaling also includes reference signals, mobile devic to base, which also contains reference subcarriers. In fact there is more than one mode in which contain these subearriers from a full sounding mode used by the network to assign a frequency band to the uplink device to a mode where are reference subearriers are used to generate a channel impulse responses to aid in demodulation of the uplink signal, etc. Also, similarly to the DL PRS added in rel.9 additional UL reference signals might be added in the upcoming and future standard releases. In this embodiment, the uplink signal is processed by multiple base units (eNB) using the same range to phase, multipath mitigation processing to generate range related observables. In this embodiment, location consistency algorithms are used as established by the multilateration algorithm to edit wild point observables and generate a location estimate.
[0373] Yet another embodiment, relevant one or more reference (including pilot and/ or synchronization) subearriers of both the LTE d ownlink and LTE uplink are collected, the range to phase mapping is applied, multipath mitigation is applies and the range associated observable is estimated. These data would then be fused in such a way that would provide a more robust set of observables for location using the multilateration algorithm and location consistency algorithms. The advantage would be the redundancy that results in improved accuracy since the downlink and up link two different frequency bands or in case of the TDD (Time Division Duplexing) improving the system coherency.
[0374] In a DAS (Distributed Antenna System) environment where multiple antennas transmitting the same downlink signal from a microcell the location consistency algorithm(s) are extended to isolate the ranges of the DAS antennas from observables generated by the multipath mitigation processing from reference signal(s) (including pilot and/ or synchronization) subearriers and to obtain the location estimates from the multiple DAS emitters (antennas) ranges.
[0375] In a DAS system (environment) obtaining accurate location estimate is possible only if the signals paths from individual antennas can be resolved with a lisgh accuracy, whereby the path error is only a fraction of the distance between antennas {accuracy of 1 meters or better). Because all existing techniques/ methods cannot provide such accuracy in a heavy multipath environment (signals from multiple DAS antennas will appear as induced heavy multipath) the existing techniques/ methods cannot take advantage of the abovementioned extension of the location consistency algorithm(s) and this locate method/ technique in the DAS environment.
[0376] The InvisiTrack multi-path mitigation methods and systems for object identification and location finding, described i U.S. Patent No. 7,872,583, is applied to the range to signal phase mapping, multipath interference mitigation aid process to generate range based location observables utilizing the LTE downlink, uplink and or both (downlink and uplink), one or more reference signal(s) subcarriers and using multilateration and location consistency to generate a location estimate.
[03??1 In all above embodiments trilateration positioning algorithms can be also employed.
[0378} The DL-OTDOA locating was specified in the LTE release 9: Evolved
Universal Terrestrial Radio Access (E-IJTRA); Physical channels and modulation; 3GPP TS 36.21 1 Release 9 technical Specification. However, it has not been implemented by the wireless operators (carriers). In the meantime a Downlink locating can be implemented within current, e.g. unmodified, LTE network environment by using the existing physical layer measurements operation(s),
[03791 In LTE the LIE and the eNB are required to make physical layer measurements of the radio characteristics. The measurement definitions are specified in 3GPP TS 36.214. These measurements are performed periodically and are reported to the higher layers and are used for a variety of purposes includin intra- and inter-frequency handover, inter-radio access technology (inter-RAT) handover, timing measurements, and other purposes in support of RRM (Radio Resource Management).
[0380 For example, the RSRP (Reference Signal Received Power) is the average of the power of all resource elements which carry cell-specific reference signals over the entire bandwidth,
[038 1 Another example is the RSRQ (Reference Signal Received Quality)
measurement, that provides additional information (RSRQ combines signal strength as well as interference level).
|0382| The LTE network provides the UE with eNB neighbor (to serving eNB) lists.
Based on the network knowledge configuration, the (serving) eNodeB provides the UE with neighboring eNB's identifiers, etc. The UE then measures the signal quality of the neighbors it can receive. The UE reports results back to the eNodeB. Note: UE also measures the signal quality of die serving eNB.
[03831 According to the specification, the RSRP is defined as the linear average over the power contributions (in { W j) of the resource elements that carry cell-specific reference signals within the considered measurement frequency bandwidth. The measurement bandwidth that is used by the UE to determine .RSRP is left up to the UE implementation with the limitation that corresponding measurement accuracy requirements have to be fulfilled.
[0384] Considering the measurement bandwidth accuracy requirements this bandwidth is fairly large and the cell-specific reference signals that are used in the RSRP measurements can be further processed to determine these reference signals subcarriers phase and amplitude that are in turn applied to the multi-path processor for multipath interference mitigation and generation of range based location observables. In addition, other reference signals that are used in the RSRP measurement, for example SSS (Secondary Synchronization Signal) might be also used.
[0385] Thereafter, based on range observ ables from three or more cells the location fix can be -estimated using muhilateration and location consistency algorithms.
[03861 As was mentioned previously while there ate several causes of the RF
fingerprinting database instability one of the major ones is the multipath (the RF signature is very sensitive to multipath). As a result, the RF Fingerprinting method(s)/ technology locate accuracy is heavily impacted by multipath dynamics - changes over time, environment (for example weather), people and/ or objects movement, including vertical uncertainty. >100% variability depending upon device Z-height and/ or antenna orientation (see Tsung-Han Lin, et al. Microscopic Examination of an RSSI-Signature-Based Indoor Localization System).
[0387] The present embodiments can significantly improve the RF Fingerprinting locate accuracy because of the ability (multipath processor) to find and characterize each individual path, including significantly attenuated DLOS. As a result, the RF
Figure imgf000072_0001
decision on the location fix can be supplemented with the real-time multipath distribution information
[0388] As was mentioned above, the locate fix will require position references synchronization in time. In wireless networks these position references may include Access Points, Macro/ Mini/ Pico and Femto cells, as wells as so called Small cells (eNB). However, wireless operators do not implement the synchronization accuracy that is needed for an accurate position fix. For example, in case of LTE the standard does not require any time synchronization between eNB(s) for the FDD (Frequency Division Duplexing) networks. For LTE TDD (Time Division. Duplexing) this time synchronization accuracy is limit is +/- 1.5 microseconds. This is equivalent to 400+ meters locate uncertainty. Although not required, the LTE FDD networks are also synchronized, but use even larger (than 1,5 microseconds) limits. [03891 Wireless LTE operators are using GPS/ GNSS signals to synchronize e-NB(s) in frequency and time. Note: The LTE eNB has to maintain a very accurate carrier frequency: 0.05 ppm for macro/ mini cells and slightly less accurate for other type of cells (0.1 - 0.25 ppm). The GPS/ GNSS signals can also enable a required (for locate) time synchronization accuracy of better than 10 nanoseconds. However, network operators arid network equipment manufacturers are trying to reduce costs associated with the GPS/ GNSS units in favor of Packet Transport/, e.g. Internet/ Ethernet networking time synchronization by employing NTP (Network Time Protocol) and/ or PTP (Precision Time Protocol), for example IEEE 1588v2 PTP.
[03901 The IP network based synchronization has a potential of meeting the minimum frequency and time requirements, but is lacking the GPS/ GNSS precision that is needed for locate fix.
[03911 T e approach described herein, is based on the GPS/ GNSS signals and signals generated by the eNB and/ or AJP, or other wireless networks equipment, it also can be based on the IP networking synchronization signals and Protocols and signals generated by the eNB and/ or AP, or other wireless networks equipment. This approach is also applicable to other wireless networks, including WiMax, WiFi and White Space.
{0392J The eNB signals are received by the Time Observation Unit (TMO) installed at the operator's eNB facility (Figure 12). The TMO also include the External Synchronization Source input.
[0393| The eNB signals are processed by the TMO and are time stamped using clocks that are synchronized with the External Synchronization Source input,
[0394] The External Synchronization Source could be from the GPS/ GNSS and' or Internet/ Ethernet networking, for example PTP or NTP, etc.
[i)395j The time-stamped processed signal, for example the LTE frame start (could be other signals, especially in other networks), also includes the eNB (cell) location and/ or cell ID, is sent via the Internet/ Ethernet backhaul to a central TMO Server that creates, maintains and updates a data base of ail eNBs,
0396J The UE and/ or eNB( s) involved in the process of ranging and obtaining a location fix will quire the TMO Server and the server will, return the time synchronization offsets between the sN B(s) involved. These time synchronization offsets will be used by the UE and/ or eNB(s) involved in the process of obtaining a location fix to adjust the location fix.
j0397} Alternatively, the location fix calculations and adjustment can be carried out by the TM O Server when UE and/ or eNB(s) involved in the process of ranging will also supply the obtained ranging information to the TMO Server. The TMO Server will then return an accurate (adjusted) position (locale) fix.
[0398J If more t han one cell eNB equipment is co-located together a single TMO can process and time stamp signals from, ail e B(s).
[03991 The RTT (Round Time Trip) measurements (ranging) can be used for locating. The drawback is that the RTT ranging is subject to multipara which has drastic impact on the locate accuracy .
[0400] On the other hand, RTT locating does not require the position references synchronization (in time) in general and in case of LTE the eNB in particular.
[0401] At the same time, when operating with Pilot Reference and/ or other signals of the wireless network the multipath mitigation processor, method(s)/techniques and algorithms described in U.S. Patent No. 7,872,583 are capable of determining the channel response for the RTF signal(s), e.g. identify die mdtipath channel that the RTT signal(s) are going through. This allows for correctio of the RTT measurements so that the actual DLOS time will be determined.
[0402] With DLOS time known it will be possible to obtain the location fix using trilateration and/ or similar locating methods without the need of eNB or position references synchronization in time.
[0403] Even with TMO and TMO Server in place the InvisiTrack's technology integration will require changes in the macro/ mini/ pico and small cells and/ or UE (cell phone). Although these changes are limited only to SW/ FW (software/ firmware) it takes a lot of effort to revamp the existing infrastructure. Also, in some cases network operators and/ or UE/ cell phone manufacturers/ suppliers resisting equipment modifications. Note: UE is wireless network User Equipment.
[0404] This SW/ FW change can be completely avoided if the TMO and TMO Server functionality is expanded to support the InvisiTrack locate technology. In other words, another embodiment described below operates with wireless networks signals, but do not require any modifications of the w ireless network equ ipment/ infrastructure. Thus, the embodiment described below operates with LTE networks and it is also applicable to other wireless systems/networks, including Wi-Fi.
[0405] In essence this embodiment creates a parallel wireless locate infrastructure that uses the wireless network signals to obtain location fix.
[0406] Similarly to TMO and TMO Server, the InvisiTrack's locate infrastructure will consists of one or more wireless Network Signals Acquisition Units (NSAU) and one or more Locale Server Units (LSU) that collect data from NSAU(s) and analyze it. detenn ing range and locations, and to convert it into a table, e.g. of phone UEs IDs and locations at an instant of time. The LSU interfaces to the wireless network via network's API,
[0407] Multiple of these units could be deployed in various locations in a large infrastructure. If NSAU(s) have coherent timing - the. -results for all can be used which will give better accuracy.
[O408| The coherent timing can be derived from the GPS clock and/ or other stable clock sources,
[0409! The NSAU communicates with LSU via LAN (Local Area Network), Metro Area Network (MAN) and/ or Internet.
[0410] In some installation/ instances the NSAU and LSU could be combined/ integrated into a single unit.
[0411] In order to support location services using LTE or other wireless networks, the transmitters are required to be clock and event synchronized to within tight tolerances. Normall this is accomplished by locking to the 1 PPS signs! of GPS. This will result in timing synchronization in a local area to within 3 nanosecond 1-sigma.
[0412] However, there are many instances when this type of synchronization is not practical. This present embodiments provide time offset estimates between the downlink transmitters and tracking of the time offsets in order to provide delay compensation values to the location process so the location process can proceed as if the transmitters were clock and event synchronized. This is accomplished by prior knowledge of the transmit antenna (which is required for any location services) and a receiver with known a priori antenna location. This receiver called the synchronization unit will collect data from all the downlink transmitters and given its knowledge of the locations, calculate the offset timing from a preselected base antenna. These offsets are tracked by the system through the use of a tracking algorithm that compensates for clock drifts the downlink transmitters. Note: The processing to derive pseudo ranges from the received data will utilize the InvisiTraek Multipath mitigation algorithms (described in U.S. Patent No. 7,872,583). Hence the synchronization will not be impacted by multipath.
[0413] These offset data are used by the location processor (Location Server, LSU) to properly align the data from each downlink transmitter so that it appears to have been generated by synchronized transmitters. The time accuracy is comparable with the best 1-PPS tracking and will support 3 meter location accuracy (1-sigma). [04141 The synchronization receiver and/ or receiver's antennas will be located based on optimal GDOP for best performance. In large installations multiple synchronization receivers can be utilized to provide an equivalent 3 nsec 1-sigma synchronization offset throughout the network. By utilizing synchronization receivers(s) the requirements for synchronization of the downlink transmitters is eliminated.
[0415J The synchronization receiver imk can be a standalone nnit communicating with the NSAU and/ or LSU. Alternatively this synchronization receive ca be integrated with th NSAU,
[0416J The exemplary wireless network locate equipment diagram is depicted in FIG.
13.
{0417J The embodiment of a completely autonomous system, no Customer Network- Investment, which utilizes LT.E signals operates in the following modes:
1 , Uplink mode - uses wireless network Uplink (UL) signals for the purpose of locating (FIGS. 16 and 17}
2, Downlink mode - uses wireless network Downlink (DL) signals for the purpose of locating (FIGS. 14 and 15).
3, Two-way mode - uses both: UL and DL signals for locating.
In the Uplink mode multiple antennas are connected to one or more NSAUs. These antennae locations are independent from the wireless network antennas; NSAU(s) antennae locations are selected to minimize the GDOP (Geometric Dilution of Precision).
[0418j Network' RF signals from the UE/ cell phone devices are collected by NSAU(s) antennae and are processed, by NSAU(s) to produce time stamped samples of the processed network' RF signals during a time interval that is adequate for capturing one or more instances of all signals of interest.
{0419J Optionally, NSAU will also receive, process and time stamped samples of Downlink signals to obtain additional information, for example for determining UE/ phone ID, etc.
[0420] From captured time stamped samples the UE/ ceil phone devices identification numbers (ID) together with time stamped wireless network signals of interest that associated with each UE/ cell phone ID(s) will be determined (obtained). This operation can be performed either by the NSAU or by the LSU.
[0421J The NSAU will periodically supply data to the LSU. If unscheduled data is needed for one or more UE/ cell phone ID(s) then LSU will request additional data. [0422} No changes/ modifications will be needed in wireless network infrastructure and/ or existing UE ceil phone for the UL mode operation.
{0423| in the Downlink (DL) mode the InvisiTrack enabled UE will be required. Also, the cel l phone F W would have to be modified if phone is used to obtain location fix.
[04241 Ϊ» some instances operators can make baseband signals available from BBU(s) (Base Band Units). In such cases SAU(s) will also be capable process these available base band wireless network signals instead of RF wireless network signals.
[0425] In the DL mode there is no need to associate the UE/ cell phone ID with one or more wireless network signals because these signals will be processed in the UE/ cell phone or UE/ cell phone will periodically produce time stamped samples of the processed network' RF signals and send these to the LSU; and the LSU will send result(s) back to the UE/ cell phone.
[04261 bJ the mode the NSAU will process and time stamp processed RF or baseband (when available) wireless network signals. From captured time stamped samples wireless network signals DL frames starts associated with the network antennas will be determined (obtained) and the difference (offset) between these frame starts will be calculated. This operation can be performed either by the NSAU or by the LSU. Frame starts oifsets for network antennas will be stored on the LSU.
[0427] In the DL mode frame starts offsets of network antennas will be sent from LSU to the UE/ phone device in case when the device will process/ determine its own location fix using InvisiTrack technology. Otherwise, when UE/ cell phone device will periodically send time stamped samples of the processed network' RF signals to the LSU, the LSU will determine the device's location fix and will send the location fix data back to the device.
[0428] In DL mode the wireless network RF signals will come from one or more wireless network antennae. To avoid multipath impact on results accuracy the RF signal should be sniffed out from the antenna or the antenna connection to the wireless network equipment.
[0429] The two-way mode encompasses determination of the location fix from both: UL and DL operations. This allows further improve the locate accuracy.
[0430] Some Enterprise set ups use one or more BBUs feeding one or more Remote Radio Heads (RRH), with each RRH in turn feeding multiple antennae with the same ID. In such environments, depending on wireless network configuration, determining the DL mode frame starts offsets of network antennas might not be required. This includes a single BBU set up as well as multiple BBUs, whereby antennae of each BBU are assigned to a certain zone and adjacent zones coverage's are overlapping. [043 If On the other hand a configuration, configuration whereby antennae that are fed .from multiple BBUs are interleaved in the same zone will require determining the DL mode frame starts offsets of network antennas.
[0432] In DL mode of operation in DAS environment multiple antennae may share the same ID.
[0433} In the present embodiments, location consistency algorithm(s) are extended/ developed to isolate the ranges of the DAS antennas from observable* generated by the multipath mitigation processing from reference signal(s) (including pilot and/ of .synchronization) subcarriers and to obtain the location estimates from the multiple DAS emitters (antennas) ranges.
[0434] However, these consistency algorithms have limits of number of antennae that emit the same ID. It is possible to reduce the number of antennae that emit the same ID by
1. For a given coverage zone interleave Antennas that are fed from different sectors of sectorized BBU (BBUs are capable of supporting up to six sectors)
2. For a given coverage zone interleave Antennas that are fed from different sectors of sectorized BBU as well as Antennas that are fed from different BBUs
3. Adding a propagation delay element to each antenna. The delay values would be chosen to be large enough to exceed the delay spread in a particular DAS environment (channel), but smaller than the Cyclic Prefix (CP) length so that the multipath caused by additional delays will not result in ISI (Inter Symbol Interference). The addition of a unique delay ID for one or more antenna further reduces the number of antennae that emit the same ID.
[0435] In an embodiment, an autonomous system with no Customer Network
Investment can be offered . In such embodiment, the system can operate on a band other than the L IE band. For example, iSM (industrial Scientific and Medical) bands and/ or White Space bands can be used in places where LTE services are not available.
[0436] The embodiment can be also integrated with the macro/ mini/ pko/ femto station (s) and/ or UE (cell phone) equipment. Although the integration may require Customer Network Investment, it can reduce cost overhead and can dramatically improve the TCO (Total Cost o f O wnersh i.p).
[04371 As mentioned herein above, PR.S can be used by the UE for the Downlink Observed Time Difference of Arrival (DL-OTDOA) positioning. Regarding the synchronization of neighborin base stations (eNBs), the 3GPP TS 36.305 (Stage 2 functional specification of User Equipment (UE) positioning in E-UTRAN) specifies transferring timing to the UE,. the timing being r elative to an eNode .8 service of candidate cells (e.g., neighboring cells). The 3 GPP TS 36.305 also specifies Physical cell IDs (PC is) and global ceil IDs (GCls) of candidate cells for measurement purposes.
i0438 According to the GPP TS 36.305, this information is delivered from the E- MLC (Enhanced Serving Mobile Location Centre) server. It is to be noted that the TS 36.305 does not specify the abovememioned liming accuracy ,
439| Additionally, the 3GPP TS 36.305 specifies that the UE shall return to the B- MLC the downlink measurements, which includes Reference Signal Time Difference (RSTD) measurements.
[0440] The RSTD is the measurement taken between a pair ofeNBs (see TS 36.214 Evolved Universal Terrestrial Radio Access (E-UTRA); Physical Layer measurements; Release 9). The measurement is defined as a relative timing difference between a subframe received from the neighboring cell j and a corresponding subframe of the serving cell i. Positioning Reference Signals are used to take these measurements. The results are reported back to the location server that calculates the position.
[0441] In an embodiment, a hybrid method can be defined to accommodate both the newly introduced PRS and the already existing reference signals. In other words, the hybrid method can use/operate with PRS, with other reference signals (e.g., ceil or node-specific reference signals (CRS)), or with both signal types.
[0442} Such a hybrid method provides the ad vantage of allowing network operators) to dynamically choose the mode of operation depending on circumstances or network parameters. For example, the PRS have better hearability than CRS, but may result in up to 7% reduction in the data throughput. On the other hand, CRS signals do not cause any throughput reduction. In addition, CRS signals are backward compatible with all previous LTE releases, for example Rel- 8 and lower. As such, the hybrid method provides a network operator the ability to trade-off or balance between hearability, throughput, and compatibility.
[0443] In Long Term Evolution (LTE) implementations, LTE downlink baseband signals (generated by a cell or wireless node and referred to herein as "nodes") are generally combined into downlink frames. A receiver for detecting and receiving such signals may detect downlink frames from multiple cells or nodes (two or more). Each downlink frame includes multiple CRS or reference signals. In a Downlink (DL) frame, these reference signals have predetermined positions in time and frequency, e.g., there are deterministic time offsets between the frame start and each CRS in a given frame. [04441 ΪΛ addition, each CRS is modulated with a special code. The modulation and the code are also predetermined. The CRS modulation is the same for all nodes, but the code (seed) is determined by the ID (identification) number of the node.
[044$! As a result, by knowing the node ID(s), it is possible to estimate a course location of a frame start time for each frame from each node (cell), in the spectrum of the reference signals. To do so, it is first necessary to determine the frame start times or frame starts for all DL signals from different nodes. For example, in an embodiment, by correlating the received DL baseband signal with known replicas of code modulated CRS (generated internally by a detector and/or a multipath mitigation processor) it is possible to find all CRS sequences or other reference signals from various nodes, and w ith this information find coarse location frame starts of all observable nodes. In an embodiment, the detector may also demodulate/decode the CRS and then correlate the demodulated/decoded CRS with baseband sub-carriers that are assigned to the CRS.
[0446] At the same time, in an embodiment, the CRS may also be used as rangin signals by the multipath mitigation processor. Therefore, in addition to finding coarse frame starts the detector's correlation process is also capable of isolating the CRS from other signals (such as payload) in the frame using the code that was used to modulate those signals.
Thereafter, these isolated CRS, and associated frames starts, are transferred to a multipath mitigation processor for ranging.
[0447J A similar approach can be used in the Uplink mode, whereby timing offsets between different node receivers can be determined.
[0448] In a downlink embodiment, a system for tracking and locating one or more wireless network devices in communication with a network comprises a user equipment receiver configured to receive multiple signals from two or more nodes in communication with the network, the multiple signals being modulated with a code determined by an identification of each node of the two or more nodes transmitting the multiple signals, the user equipment receiver including a detector configured to detect and isolate reference signals from the multiple signals based on the identification, and a processor configured to use the reference signals as ranging signals from each node for tracking and locating the one or more wireless network devices.
[0449] In the embodiment, wherein the mdtiple signals from each node of the two or more nodes are combined into a frame that includes the reference signals, and wherein the detector is further configured to estimate a course location of frame starts from each node. [04501 In the embodiment, wherein the detector Is further configured to estimate tSie course location by correlating the reference signals with known replicas of such reference signals,
[04511 to e embodiment. v\ herein the detector is further configured to isolate the reference signals from any other signals in the frame, and wherein the detector is further configured to isolate the reference signals for each node of the two or more nodes.
[0452] In the embodiment, wherein the processor is at least one multipath mitigation processor, and wherein the multipath mitigation processor is configured to receive the course location and isolated reference signals and estimate a relative time of arrival of the ranging signals from each node.
04531 In the embodiment, wherein the processor is at least one multipath mitigation processor,
[0454j In the embodiment, wherein the multiple signals from each node of the two or more nodes are in a frame, wherein the detector is further configured to estimate a course location of frame starts from each node, wherein the detector is configured to isolate the reference signals from any other signals in the frame, wherein the detector is further configured to isolate the reference signals for each node of the two or more nodes, wherein the detector is configured to pass the course location and isolated reference signals for each node to the multipath mitigation processor, and wherein the multipath mitigation processor is configured to receive the course location and isolated reference signals and estimate a relative time of arrival of the ranging signals from each node.
[0455] In the embodiment, the system further comprises an uplink embodiment where a node receiver is configured to receive device signals from the one or more wireless network devices, the device signals being modulated with a device code determined by a device identification of each wireless network device of the one or more wireless network devices transmitting the device signals, the node receiver including a device detector configured to detect and isolate device reference signals from the device signals based on the device identification, and a second processor is configured to use the device reference signals as ranging signals from each wireless network device for tracking and locating the one or more wireless network devices.
[0456] In an embodiment, a system for tracking and locating one or more wireless network devices in communication with a network, comprises a user equipment receiver configured to receive multiple signals from two or more nodes in communication with the network, the multi le signals being modulated with a code determined by an identification of each node of the two or more nodes transmitting the multiple signals, and a processor configured to detect, and. isolate reference signals from the multiple signals based on the identification and. to use the reference signals as ranging signals from each node for tracking and locating the one or more wireless network devices.
[0457] In the embodiment, wherein the multiple signals from each node of the two or more nodes are combined into a frame that includes the reference signals, and wherein the processor is further configured to estimate a course location of frame starts from each node.
[0458] In the embodiment, wherein the processor is further configured to estimate the course location by correlating the reference signals with known replicas of the reference signals.
[0459] In the embodiment, wherein the processor is further configured to estimate a relative time of arrival of the ranging signals from each node based on the course location and isolated reference signals.
[0460] In the embodiment, wherein the processor is further configured to isolate the reference signals from any other signals in the frame, and wherein the processor is further configured to isolate the reference signals for each node of the two or more nodes.
[0461] In the embodiment, wherein the multiple signals from each node of the two or more nodes are in a frame, wherein the processor is further configured to estimate a course location of frame starts from each node by correlating the reference signals with known replicas of the reference signals, wherein the processor is further configured to isolate the reference signals from any other signals in the frame and to isolate the reference signals for each node of the two or more nodes, and wherein the processor is further configured to estimate a relative time of arrival of the ranging signals from each node based on the course location and isolated reference signals.
[0462] In an embodiment, a system for tracking and locating one or more wireless network devices in communication with a network, comprises a detector configured to receive multiple signals from two or more nodes in communication with the network, the multiple signals being modulated with a code determined by an identification of each node of the two or more nodes transmitting the multiple signals, and to detect and isolate reference signals from the multiple signals based on the identification, and a processor configured to use the reference signals as ranging signals from each node for tracking and locating the one or more wireless network devices. [04631 ΪΛ the embodiment, wherein ihe multiple signals from each node of the two or more nodes are combined into a frame that includes the reference signals, and wherein the detector is further configured to estimate a course location of frame starts from each node.
[0464] In the embodiment, wherein the detector is further configured to estimate the course location by correlating ihe reference signals with known replicas of such reference signals,
[0465} in the embodiment, wherein the detector is further configured to isolate the reference signals from any other signals in the frame, and wherein the detector is further configured to isolate the reference signals for each node of the two or more nodes.
[0466] In the embodiment, wherein the processor is at least one multipath mitigation processor, and wherein the multipath mitigation processor is configured to receive the course location and isolated reference signals and estimate a relative time of arrival of the ranging signals from each node.
[0467} In the embodiment, wherein ihe processor is at least one multipath mitigation processor.
[0468] In the embodiment, wherein the multiple signals from each node of the two or more nodes are in a frame, wherein the detector is further configured to estimate a course location of frame starts from each node, wherein the detector is configured to isolate the reference signals from any other signals in the frame, wherein the detector is further configured to isolate the reference signals for each node of the two or more nodes, wherein the detector is configured to pass the course location and isolated reference signals for each node to the multipath mitigation processor, and wherein the multipath mitigation processor is configured to receive the course location and isolated reference signals and estimate a relative time of arrival of the ranging signals from each node.
[0469] In an embodiment, a system for tracking and locating one or more wireless devices in communication with a network, comprises a node receiver configured to receive device signals from the one or more wireless network devices, the device signals being modulated with a device code determined by a device identification of each wireless network device of the one or more wireless network devices transmitting the device signals, the node receiver including a device detector configured to detect and isolate device reference signals from the device signals based on the device identification, anda processor configured to use the device reference signals as ranging signals from each wireless network device for tracking and locating the one or more wireless network devices. [04701 Furthermore, the hybrid method can be transparent to the LTE UE positioning architecture. For instance, the hybrid method can operate in the 3GFP TS 36.305 framework.
(04711 in art embodiment, STD can be measured and, according to the 3 GPP TS 36.305, transferred from a UE to an. E-SMLC.
[04721 The UL-TDOA (l!-TDOA) is currently in a study phase and is expected to be standardized in the upcoming release 11.
[0473} 'Embodiments of the 'UL-TDOA (Uplink) are described herein above and are also shown in FIGS. 16 and 17. FIG. 18 and 19, described herein below, provide examples of alternative embodiments of the UL-TDOA.
[0474] FIG. 18 presents an environment that may include one or more DAS and/ or Femto/ Small ceil antennas. In this example embodiment, each NSAU is equipped with a single antenna. As depicted, at least three NSAUs are required. However, additional NSAUs can be added to improve hearability because each UE must be "heard" by at least three N SAUs.
[0475} Furthermore, the NSAU'(s) can 'be configured as receivers. For example, each NSAU receives but does not transmit information over the air. In operation, each NSAU can listen to the wireless Uplink network signals from UEs. Each of the UEs can be a cell phone, a Tag, and/ or another UE device.
[04761 Moreover, the NSAUs can be configured to communicate with a Locate Server Unit (LSU) over an interface, such as a wired service or a LAN. In turn, the LSU can
communicate with a wireless or an LTE network. The communication can be via a network API, where the LSU can, for example, communicate with an E-SMLC of the LTE network and can use a wired service such as a LAN and/or a WAN,
[0477] Optionally, the LSU may also communicate directly with DAS base station(s) and or Femto/ Small cells. This communication can use the same or a modified Network API.
[0478] In this embodiment, the Sounding Reference Signal (SRS) can be used for locate purposes. However, other signals may also be employed.
[0479J The NSAUs can convert the UE Uplink transmission signals to a digital format, for exarapie I/Q samples, and can periodically send a number of the converted signals to the LSU with a time stamp.
[0480} The DAS base station(s) and or Femto/Small cells can pass to the LSU one or all of the following data:
1) the SRS, the I/Q samples, and the lime stamp;
2) a list of served UE IDs; and 3) SRS schedule per UE with a UE ID, the schedule including S S
SchedutmgRequesiCoufig information and SRS-UL-Config . information..
[0481] The information passed to the LSU may not be limited by the abovementioned information. It can include any information needed to correlate each UE device uplink signal, such as a UE SRS, with each UE ID.
[0482} The LSU functionality can include ranging calculations and obtaining the location fix of a UE. These determinations/ calculations can be based on the information passed from the NSAUs, the DAS bases stations, and or Femto/Smali cells to the LSI),
[ 483J The LSU may also determine timing offsets from the available downlink transmission information passed from the NSAUs to the LSU.
[0484} In turn, the LSU can provide the wireless or LIE network with UE location fix and other calculations and data. Such information, can be communicated, via the Network APT.
[0485} For synchronisation purposes, each NSAU may receive, process, and time stamp samples of Downlink signals. Each NSAU may also periodically send a number of such samples to the LSU, including the time stamp(s).
[0486} Additionally, each NSAU may include an input configured for synchronization with external signaS(s).
[0487} FIG. 19 depicts another embodiment of a UL-TDOA. In addition to the components depicted under FIG. 18, the environment of this embodiment may include one or more cell towers that can be used in lieu of the DAS base stations and/or Femto/Smali cells. Data from the one or more cell towers can be used to obtain the location fix of a UE.
[0488] As such, an advantage of this embodiment includes obtaining a location fix with only a single cell tower (eNB). In addition, this embodiment can be configured to operate in a similar manner as described under FIG. 18, with the exception that one or more eNBs can replace the DAS base stations and/or the Femto/Smali cells.
[0489} One method of uplink locating of UE is the Cell Identification method (CID). In the basic CID method the UE position may be determined on the cell level. This method is purely network based . As a result, the UE , for example a handset, is not aware of the fact that it is being tracked. While this is a relatively simple method, it lacks accuracy because the locate uncertainty is equal to the cell diameter. For example, as illustrated i FIG. 20, any of the handsets 2000 within the cell diameter 2002 of a serving cell tower 2004 effectively have the same location, even though they are not at the same location. The accuracy of the CID method can be improved when combined with serving sector identification (sector ID) knowledge. For example, as illustrated i» FIG. 21 , sector ID 2160 idealities a section 2102 within the cell diameter 2662 that ineiiid.es a mnnber of handsets 2104 thai are known to have a different location than the other handsets 2000 in other sectors of the cell diameter 2002.
[0490] Further enhancement to the CID method may be possible through the Enhanced Ceil ID (E-CID) method, which provides further refinements to the basic CID method described above. One enhancement uses timing measurements to calculate how far away the UE is from the eNB (the network node). This distance can be calculated as half the round trip time (RTT), or Timing Advance (TA) in LTE (LTE TA), times the speed of light. If the UE is connected, then RTT or TA may be used for distance estimation. In this case both: the serving cell tower or sector and the UE (upon the serving eNB command) will measure the timing difference between Rx sub-frames and Tx sub-frames. The UE will report its measurements to the eNB (also under the eNB control). It should be noted that LTE Rel-9 adds the TA type 2 measurements that rely on the timing advance estimated from receiving a PRACH preamble during the random access procedure. A PRACH (physical/packet random access channel) preamble specifies the maximum number of preambles to be sen t during one PRACH ramping cycle when no response is received from the UE being tracked. The LTE Type 1 TA measurement is the equivalent to the RTT measurement, as follows:
RTT - TA iiype 1) - eNBiRx - 1.x. ! + UEiRx - Tx)
With knowledge of the eNB's coordinates and the height of the serving cell tower antenna, the position of the UE can be calculated by the network.
j0491f TSie E-CID locating method is still limited, however, because in one dimension the locate accuracy depends upon the sector width and the distance from the serving cell tower, and in the other dimension the error depends upon the TA (RTT) measurement accuracy. The sector width varies with network topology and is impacted by the propagation phenomena, specifically muitipath. Sector accuracy estimates vary from 200 meters to in excess of 500 meters. The LTE TA measurement resolution is 4 Ts, which corresponds to 39 meters of maximum error. The actual error in the LTE TA measurement is even larger, however, due to calibration inaccuracies and the propagation phenomena (muitipath), and may reach as much as 200 meters.
f©492J As illustrated in FIG, 22 the E-CID method may be further improved with the addition of a feature known as Angle of Arrival (AoA). The eNB estimates the direction from which the UE is transmitting using a linear array of equally spaced antenna elemen ts 2200. Typically, reference signals are used for the AoA determination. When reference signals are received from the UE at two adjacent antenna elements 2200, the reference signals may be phase rotated, as shown in FIG. 23 by an amount which depends on the AoA, the carrier frequency, and the element spacing. The AoA will require each e B to be equipped with antenna arrays adaptive antennas. It is also exposed to rmihjpath and. topology variances. Nevertheless, sophisticated antenna arrays can significantly reduce the width 2202 of the sector 2100, which may lead to better locate accuracy. Moreover, if two or more serving cell towers 2300 (eNB's base stations equipped with directional antenna arrays) can be used to make the handset AoA determination, as illustrated in FIG. 23 then the accuracy may be considerably improved. In such a case, the accuracy is still subject to the multipath/propagation phenomena.
[0493] Deploying antenna arrays/adaptive antennas network-wide over multiple LTE bands requires a monumental effort in terms of capital, time, maintenance, etc. As a result, the antenna arrays/adaptive antennas have not been deployed for the purpose of UE locating. Other approaches, such as signal strength based methods, do not produce significant accuracy improvement. One such signal strength approach is fingerprinting, which requires creating and continuously updating an enormous, continuously changing (in time) fingerprint database, e.g. large capital and reoccurring expenses without significant accuracy improvement. Moreover, fingerprinting is UE based technology, whereby the UE position cannot be determined without UE assistance on the UE application level.
[0494] A solution to the limitations of other uplink location methods involves the use of AoA capabilities without the need for antenna arrays/adaptive antennas. Such an embodiment may employ TDOA (Time Difference of Arrival) location techniques for AoA determination, which may be based on estimating the difference in the arrival times of the signal from the source at multiple receivers. A particular value of the time difference estimate defines a hyperbola between two receivers in communication with a UE. When the distance between the receiving antennas is small relative to the distance of the emitter (the handset) being located, then the TDOA is equivalent to the angle between the baseline of the sensors (receivers antennas) and the incident RF energy from the emitter. If the angle between the baseline and true North is known, then the line of bearing (LOB) and or Ao A can be determined.
[0495J While general locate methods that use either TDOA or LOB (also known as AoA) are known, TDOA locate methods have not been used to determine LOB because the TDOA reference points are too close to one another to make the accuracy of such a technique acceptable. Rather, LOB is usually determined using directional antennas and/or beam-forming antennas. The super resolution methods described herein, however, make it possible to use TDOA for LOB determination while dramatically improving accuracy. In addition, without the reference signal processing techniques described herein, it may not be possible to "hear", e.g. detec t, reference signals coming from a UE outside of the serving sectors, e.g. by the non-serving sectors and or antennas; Without the resolution and processing capabilities described herein, it may not be possible to employ TDOA for LOB determination because at least two points of reference are needed, e.g. two or more sectors and/or antennas). Similarly, a UE may not be able to detect reference signals coming to the UE from other than serving sectors, e.g. from the non- serving sectors and/or antennas.
[0496J For example, in FiG. 24 two antenna separation scenarios are illustrated: wide separation and close (small) .separation in both scenarios the hyperbola 2400 and the incident line 2402 are crossi n g at the handset 2000 location, but in the case of where the antenna 2404 separation is wide, this happens at a steeper angle, which in turn substantially reduces the locat error. At the same time, in case of the antennas 2404 being close to each other the hyperbola 2400 becomes interchangeable with the line 2402 of the RF energy incident or the LOB/ AoA.
[0497] The formula set forth belowr can be used to determine the incident RF energy from the emitter, where the time difference in arrival time of RF energy between two antennas (sensors) is given by: sin 0
At =
c
where:
At is the time difference in seconds;
x is the distance between the two sensors in meters;
Θ is the angle between the baseline of the sensors and the incident RF wave, in degrees ; and
c is the speed of Sight.
[04981 Several locate strategies are available through, use of the TDOA locating embodiment, including: (1) when the TDOA measurements (multilateration) between two or more serving cells are available, e.g., wide separation; (2) when the TDOA measurements are only from two or more sectors at one or more serving cells, e.g., small antenna separations, such LOB/ AoA; (3) a combination of strategies (2) and (3); and (4) a combination of TA
measurements and strategies (.1 )-{3), e.g.. improved E-CID.
[i)499| As further explained below, b the case of closely positioned antennas, the TDOA locating embodiment ma use a line of bearing when the signals from two or more antennas are from the same cell tower. These signals can be detected in the received composite signal. By know ing the tower location and the azimuth of each sector and/or antenna, the line of bearing and/or AoA can be calculated and utilized in the location process. The LOB/AoA accuracy may be impacted by multipath, noise (SN ), etc, However, this impact may be mitigated by advanced signal processing and the multipath mitigation processing techniques described above, which may be based on super resolution technology. Such advanced signal processing includes, but is not limited to, signal correlation correlating, filtering, averaging, synchronous averaging and other methods/ techniques.
[0 001 The serving cell lower 2500 typically consists of multiple sectors, as illustrated in FIG. 25 which shows a three sector (Sector A, Sector B and Sector C) configuration. The three sector deployment illustrated may include one or more antennas 2502 per sector. A single sector, such as sector A, may be in control of the UE (handset) because the handset transmissions will be in Sector A's main lobe (the main lobe's center coincides with the sector azimuth). At the same time the handset transmissions will fall outside Sectors B's and C's main lobes, e.g., into antennas side lobes. Thus, the handset signals will still be present in the output signal spectrums of Sectors B and C, but will be significantly attenuated relative to signals from other handset(s) that are located in Sector B's or Sector C's main lobes. Nevertheless, through the use of advanced signal processing, as described above and below, it is possible to obtain sufficient processing gain on ranging signals to make them detectable from the neighboring sectors' side lobes, such as the Sector B and Sector C side lobes. For network-based locating purposes, the LTE Uplink S S (Sounding Reference Signals) may be employed as ranging signals.
[0501] In other words, while the UE uplink reference signal might be in the side lobe of the neighboring sector(s) antennas, the processing gain through reference signal processing methods described herein may be sufficient to allow a calculation of TDOA between the two (or more) sector antennas. The accuracy of this embodiment may be significantly enhanced by the multipath mitigation processing algorithms described above. Thus, LOB/AOA intersected with the annulus calculated by the LTE TA timing may provide a UE location to within an error ellipse of approximately 20 meters by 100 meters. [0502} Further locale error reduction may be achieved when the UE can be heard by two or more LTE towers, which is highly probable with the processing gains and muhipaih mitigation technology described above, In such a case, the intersection of the TDOA hyperbola and one or more LOB/AoA lines may result in a 30 by 20 meter error ellipse (for a two sector cell tower). If each cell tower supports three or more sectors, then the error ellipse may be further reduced down to 10 - 15 meters. If the UE is heard by three or more eNB's (cell towers), then 5 to 10 meters accuracy may be achieved. In high value areas, such as malls, office parks arid the like, additional small cells or passive listening devices may be used to create the necessary coverage.
{0503} As was mentioned, above each sector of the cell tower 2500 may include one or more antennas 2502. In a typical installation, for a given sector, signals from each antenna are combined at the sector's receiver input. As a result, for locate purposes, two or more sector antennas can be viewed as a single antenna with composite directionality pattern, azimuth and elevation. The hypothetical antenna composite directionality and its (main lobe) azimuth and elevation may also be assigned to the sector itself.
[0504] In an embodiment, the received signals (in a digital format) from all sectors of each serving cell tower and neighboring serving cell towers are sent to a locate server unit (LSU) for location determination. Also, SRS schedules and TA measurements per each served UE is provided to the LSU by each serving sector from each serving cell tower. Assuming that each serving cell tower and each neighboring cell tower location coordinates, the number of sectors per tower with each hypothetical (composite) sector antenna azimuth and elevation, and each sector position at the cell tower are known, the LSU may determine each UE position relative to the serving ceil tower and/or neighboring cell towers. All of the abovementioned information may be sent, through wired networks, for example LAN, WAN, etc., using one or more standardized or proprietary interfaces. The LSU may also interface the wireless network infrastructure using a standardized interface and/or a network carrier's defined interface/ API. The location determination may also be split between the network node and the LSU or performed solely in the network node.
[0505] In an embodiment, the location determination may be performed in the UE or split between the UE and LSU or network node. In such cases, the UE may communicate over the air using standard networking protocols/interfaces. In addition, the location determination can be performed through a combination of the UE, the LSU and/or netw ork nodes, or the LSU functionality can be implemented (embedded) into a SU.PL server, a E-S LC server, and/or a LCS (LoCation Sendees), system that can then foe used in place of the LSU.
{OSMJ Embodiments of a Downlink (DL) locate method are reciprocals to the Uplink (UL) locate embodiments described above, in a DL embodiment, a sector may become a transmitter with a transmit pattern, azimuth and elevation that matches the sector's received directionality, azimuth and elevation. Unlike the uplink embodiments, in DL embodiments, the UE typically has a single receive antenna. Thus, for UE there is no sensors baseline that can be used to determine the RF wave incident. However, the UE can determine the TDOA(s) between different sectors and consequently a hyperbola(s) (multiiateration) between sectors, and because the same cell tower sectors are close to each other the hyperbola becomes interchangeable with the line oi" the RF energy incident or the LOB/AoA, as described above with reference to FIG. 24 While the LOB/AoA accuracy may be impacted by multipaih, noise (SMR), etc, this impact may be mitigated through use of the advanced signal processing and the muitipath mitigation processing, which is based on the super resolution technology, described above.
105071 As noted, UE DL locating can be accomplished in ways that are similar to the UE uplink locating, with the exception of that the RF wave incident angle cannot be determined from the formula above. Instead, the multiiateration technique may be used for determining the LOB/AoA for each serving cell tower.
[O508 UE DL locale embodiments also employ reference signals, in the DL case, one approach for such network-based locating may be to employ the LTE Cell-Specific Reference Signals (CRS) as ranging signals. Also, Position Reference Signals (PRS) introduced in LTE Release 9 may be used. Thus, locate may be done using CRS only, PRS only, or both CRS and PRS.
[0509j As with UE uplink locate embodiments, for UE downlink locate embodiments, a snap-shot of the UE received signal in digital format may be sent to the LSU for processing. The UE may also obtain the TA measurements and provide those to the LSU. Optionally , TA measurements per each served UE may be provided to the LSU by each serving sector from each serving cell tower (network node). As previously noted, assuming that each serving cell tower and each neighboring cell tower location coordinates, the number of sectors per tower with each sector transmit pattern azimuth and elevation, and each sector position at the tower are known, the LSU may determine each UE position relative to the serving cell tower and/or neighboring cell towers. In embodiments, the location determination may be performed in the UE or split between the UE and LSU or network node. In embodiments, all location determinations can be performed in the LSU or the network node or split between the two.
[0510] The UE will communicate/receive measurements results and other information over the air using standard wireless protocols/ interfaces. The information exchange between the LSU and network «ode(s) may be through wired networks, for example LAN, WAN, etc., using proprietary and/or one or more standardized interfaces. The LSU may interface the wireless network infrastructure using a standardized interface and or network carrier's defined interface/API. The location determination may also be split between the network node and the LSU or performed solely in the network node.
[0511] For the UE DL location embodiments described above, antenna port mapping information can also be used to determine location. The 3GPP TS 36.211 LTE standard defines antenna ports for the DL. Separate reference signals (pilot signals) are defined in the LTE standard for each antenna port. Thus, the DL signals also carry the antenna port information. This information is included in the PDSCH (Physical Downlink Shared Channel}. The PDSCH uses the following antenna ports: 0; 0 and 1 ; 0, 1, 2 and 3); or 5. These logical antenna ports are assigned (mapped) to the physical transmit antennas, as illustrated in FIG. 26 As a result, this antenna port information can be used for the antenna identification (antenna ID).
[0512] For example, the antenna port mapping information can be used to determine the RF wave incident and the hyperbola(s) (multilateration) between antennas (assuming that the antennas locations are known). Depending upon where the location determination is performed;, the antenna mapping information has to be available to the LSU or UE, or network node. It should be noted that antenna ports are indicated by placing CRS signals in different time slots and different resource elements. Only one CRS signal is transmitted per DL antenna port.
[0513] In the event of MIMO (Multiple Input Multiple Outputs) deployment in the eNB or network node, receiver{s) ma be able to determine the time differences of arrivals from a given UE. With knowledge of antennas to the receiver(s) mapping, e.g. MIMO mapping, including antennas locations, it may also be possible to determine the RF wave incident (LOB/AoA) to antennas and the hyperbola(s) (multilateration) for given eNB antennas.
Likewise, at the UE, the UE receiver(s) may be able to determine the time differences of arrival(s) from two or more eNB or network node, and MIMO antennas. With knowledge of the eNB antenna locations and antennas mapping, it will be possible to determine the RF wave incident (LOB/AoA) from antennas and the hy erbolae's) (mute'lateratioa) for given eNB antennas. Depending OH where the location determination is performed; the antenna mapping information has to be available to the LSI; or UE, or network node.
[0514] There are other configurations that are subsets of MIMO, such as Single Input Multiple Outputs (SIMO), Single Output Multiple Inputs (SOMI), Single Input Single Output (SISO), etc. All of these configurations may be defined/determined by the antenna ports mapping and/or MIMO antenna mapping information for locate purposes.
[0515} In an aspect, the present embodiments relate to methods and systems for RF- based identification, tracking, and locating of objects, including RTLS. According to one embodiment, the methods and systems employ geographically distributed clusters of receivers and/or transmitters that are precisely synchronized in time, e.g., within 10 ns or better, within each cluster, while the inter-cluster time synchronization can be much less accurate or not required at all. While a precise synchronization time of 10 ns or better is described with respect to one particular embodiment, it is important to note that the predetermined synchronization time required to achieve an accurate location depends on the equipment being utilized. For example, for some wireless system equipment, where an accuracy of 3 m is required for an accurate location determination, the predetermined time may need to be 10 ns or better, but with other wireless system equipment, a location accuracy of 50 m may be more than sufficient. Hence, the predetermined time is based on the desired accuracy location for the wireless system. The disclosed methods and systems are a significant improvement to the existing implementation of tracking and location DL-OTDOA and U-TDOA techniques, which rely on geographically- distributed standalone (individual) transmitters and/or receivers.
[0516] For example, in the DL-OTDOA technique, the relative timing difference between signals coming from neighboring base stations (eNB) is calculated and the UE position can be estimated in the network with the UE (handset) with or without UE assistance or in the UE (handset) with network assistance (control plane or user plane with SUPL based only) or without the network assistance. In DL-OTDOA, once the signals from three or more base stations are received, the UE measures the relative timing difference between signals coming from a pair of base stations and produces hyperbolic lines of position (LOPs). At least three reference points (base stations not belonging to a straight line) are needed to define two hyperbolas. The location (position fix) of the UE is in the intersection of these two hyperbolas (see FIG. 11). The UE position fix is relative to the base stations' RF emitters' (antennas) locations. As an example, when using the LPP (LTE Positioning Protocol, Rel-9) the DL- OTDOA locating is UE assisted and the E-SMLC (Evolved Serving Mobile Location Centre) is server based.
[0517] The U-TDOA technique is similar to the DL-OTDOA, but the roles are reversed. Here, the neighboring Location Management Unit (LMU) calculates the Relative Time of Arrival of the uplink signal coming from the UE (handset) and the UE position can be estimated in the network without the UE assistance. Thus, the U-TJDOA is LMU assisted and the E-SMLC (Evolved Serving Mobile Location Centre) is server based. Once the Relati ve Time of Arrival values from three or more LMUs are available, the network's E-SMLC server produces hyperbolic lines of position (LOPs) and the location (position fix) of the U E (see FIG. 27), The UE position fix is relative to the LMUs antennas locations. In an aspect, unlike the DL-OTDOA, the e B's (base station's) time synchronization in case of U-TDOA is not necessary - only (he LM.U(s) will need precision time synchronisation for locating purposes. As an example, the LMU is essentially a receiver with computing capabilities. As a further example, the LMU receiver employs the SDR (Software Defined Radio) technology. In a further example, the LMU may be a small cell, macro cell or a special purpose small ceil type device that only receives.
[0518] .Regardless of the implementation, correlating the location of the SRS for the specific UE, as provisioned by the network, will enable identification and location of the UE. Location of the SRS ma be done at the network level or within a local sector, such as a DAS for a building, a small cell or combination of small cells and macro cells that serve a specific area. If the location of the SRS for the UE is not known a priori, the solution may be able to correlate the UE's location through the covered area. Doing so will show the location history of where the UE has travelled. In some circumstances, it may be desirable to determine the location of the UE, even if the network does not provide an indication of where the SRS is located for a particular UE. The location of the UE may be correlated with the SRS by determining the location or proximity of the UE to a known point, thereby correlating the UE with the SRS it is transmitting. Such location can be accomplished through other location/proximity solutions, such as Wi-Fi and Bluetooth. The user may also identify their location via a UE application or by walking over to a predetermined location in order to identify their UE to a location solution.
[0519] In FIGS. 11 and 27 only the macro base stations are shown. Also, FIG. 27 depicts the LMUs being co-located with the base stations. These depictions are valid options, but the LIE standards do not specify where the LMUs can be placed, as long as LMUs placement satisfies the multilateration/trilateration requirements. [05201 ΪΛ an aspect, a common deployment for Indoor environments is DAS
(Distributed Antenna System) and/or 'small cells, which are inexpensive base stations highly integrated with the F. The LMU(s) can be placed indoors and/or within a campus-type environment as well, e.g. the U-TDOA can be used in a DAS and/or small cell environment. In another aspect, the U-TDOA based accurate indoors locating can be achieved with a combination of LMUs positioned indoors and macro cells that are positioned outside, e.g. without the need of deploying DAS and/or small ceils: or have a reduced number of the small cells. Thus, the LMUs can b 'deployed with or without DAS and/or small cells being present. In a further aspect, the LMUs can be placed in environments where cellular signal amplifiers/boosters are used; with or without DAS and/or small cells being present.
[0521] The LTE release 1 1 also contemplates the integration of the LMU and the e B into a single unit. This, however, will put additional burden on the time synchronization requirements between small cells if individual small cells eNBs are geographically distributed, which wireless/cellular service providers are not ready to meet, especially indoors and/or in other GPS/G SS denied environments.
[0522] DAS systems are inherently time synchronized to a much higher degree (precision) than geographically distributed macro/mini/small ce l/LMUs. Using a DL-DTOA solution in a DAS environment will alleviate the time synchronization issue, but in a DAS environment, a single base station serves a large number of distributed antennas, such that multiple antennas are transmitting the same downlink signal with the same cell ID (identification number). As a result, the traditional DL-OTDOA approach fails because there are no identifiable neighboring cells (antennas) generating signals with a different ID. Nevertheless, it is possible to use the DL-OTDOA technique when employing a multi-path mitigation processor and multi-path mitigation techniques/algorithms, as described in U.S. Patent No. 7,872,583, and extending the use of location consistency algorithm(s), as described in U.S. Nonpro visional Application No. 13/566,993, filed August 3, 2012, entitled MULTI-PATH MITIGATION IN RANGEFINDING AND TRACKING OBJECTS USING REDUCED ATTENUATION RF TECHNOLOGY; which are incorporated herein by reference in their entirety. However, these consistency algorithms have limits of the number of antennae that emit signal(s) with the same ID. One solution is to reduce the number of antennae that emit the same ID, e.g., split a large number of DAS antennas into two or more time synchronized clusters with different IDs. Such an arrangement will increase the system cost (increase the number of base stations) and require the handset UE to support the above-mentioned technology. [05231 Employing U-TDOA in a DAS environment will also add cost relative lo adding/ installing LMU units. However, no changes to the UE (handset) will be needed; only the base station software would have to be upgraded to support the U-TDOA functionality. Also, it is possible to integrate multiple LMUs with (into) a DAS system. Therefore, using the U-TDOA method with LMUs has many ad vantages when utilized indoors, in campus environments, and in other GPS/GNSS challenging, geographically limited environments.
[0524} Precise time synchronization amongst geographically distributed multiple base stations and/or small cells and/or LMUs in indoors and other GPS/GNSS denied environments is more complex than time synchronizing macro cells and/or the LMU equipment used in the macro cell outdoor, e.g., GPS/GNSS friendly environment. This is because the macro cells in the outdoor environment have antennas, thai are elevated and in the open. As a result, the
GPS/GNSS signal(s) quality is very good and macro cells antennas transmissions and/or LMU receivers can be synchronized, using GPS/GNSS to a very high accuracy -- standard deviation 10 os, over a sufficiently large area.
[0525} In an aspect, for indoor and other GPS/GNSS denied environments, time synchronization amongst multiple distributed base station and/or small cells/LMUs is achieved by using an External Synchronization Source that produces the synchronization signal shared by many base stations and/or small cells and/or LMUs. This synchronization signal can be derived from GPS/GNSS, for example the I PPS signal, and/or Internet/Ethernet networking, for example FTP or NIP, etc. The latter is a low cost solution, but it cannot provide the time synchronization precision required for accurate location, the GPS/GNSS derived external synchronization signal(s) are more precise - standard deviation down to 20 ns, but require additional hardware and installation requirements, e.g. wiring up these signals, is more complex/ expensive. Also, changes to base station and/or small cells hardware/ low level firmware might be needed to accommodate the external synchronization signal higher level of precision. Beside the 20 ns standard deviation is not accurate enough to satisfy the 3 meters requirements, e.g. standard deviation -of about 10 ns.
[0526} In order to overcome the above mentioned limitations, as illustrated by the multichannel LMU high level block diagram of FIG. 28, one embodiment uses a LMU device 2800 having multiple receive antennas 2802 and signal channels 2804. As an example, one or more signal channels 2804 can comprising signal processing components such as an RFE (RF front end) 2806, RF down con verter 2808, and/or uplink-locate processor 2810. Other
components and configurations can be used. In an aspect, the signal channels 2804 are co-located within the LMU device 2800 and tightly time synchronized (e.g., standard deviation of about 3 lis to about 10 us). In another example, antennae 2802 from each LMU signal channel 2804 are geographically distributed, (e.g., similarly to DAS). As a further example, external time
syachranixatjon components (e.g., GPS GNSS^ Internet/Ethernet, etc.) can be in communication with the LMU device 2800, The Precise time synchronization is more readily achieved inside the device {e.g., LMU device 2800) than it is by trying to tightly synchronize a number of
geographicall distributed devices.
0527| As an example, when two or more multichannel LMUs (e.g., LMU device 2800) are deployed, the time synchronization between these LMUs can be relaxed so that a low cost and low complexity approach can be used to synchronize (using an external source signal) a number of distributed multichannel LMUs. For example, latemel Ethernet networking synchronization can be used or a common sensor (device) can be deployed to provide timing synchronization between different multichannel LMUs.
[0528} On the other hand, the multichannel LMU approach reduces the number of hyperbolic lines of position (LOPs) that can be used in determining the position fix, but the time synchronization improvement overcomes this deficiency (see explanation and example below).
[0529] When using multilateration/trilateration methods, the UE positioning accuracy is a function of two factors: the geometrical dilution of precision (GDOP), which is due to.
geometrical arrangement of macro cell towers/smal! cel!s/L Us, and the accuracy of single ranging <rR pseudo measurement (See Giinter Seeber, Satellite Geodesy, 2003):
ffpos ~ (OOP x pSm!{}0
[0530} The GDOP is function of the geographical distribution of transmitting antennas (in case of DL-OTDOA) or receiving antennas (in case of U-TDOA). In ease of the regularly placed antennae, the two dimensional GDOP estimation is equal to 2 v (11 B. LEE,
ACCURACY LIMITATIONS OF HYPERBOLIC MULTILATERATION SYSTEMS, 1973); where in case of cellular networks N is the number of emitters (macro cell towers/small cells/ DAS antennas) that are "hearable" by the UE (in case of DL-OTDOA) or the number of LMUs/ LMUs receive channels that can "hear" the UE uplink transmission (in case of U-TDQA).
Therefore, the standard deviation of UE position error can be calculated as follows:
_ 2
OPOS — X °R_pseudo
[0531] Assume that eight geographically distributed (indoors) single receive channel
LMUs (regularly placed) are detecting the UE uplink transmission and these LMUs are synchronized via the I PPS signal (e.g., standard deviation of 20 ns). in. this case N::::8 and there will be seven independent LOPs thai can be used for UE position fix. Let's further assume that ranging error standard, deviation, <¾ is 3 meters (about 1 as); then the accuracy of single ranging measurement is:
^R pseudo = VO ) + Ojywc) = Vl02 + 202 = 22.4 ns (6.7 meters); where &SYNC s the external time synchronization signal standard deviation (20 ns).
In t his case (N::: 8) the single ranging measurement and the standard deviation of UE position error σΡ05 is equal to 4.74 meters.
[0532 j As an. example, if two, four receive channel, LMUs (e.g., multichannel LMU device 28ΘΟ) with regularly placed distributed antennae, are detecting, the UE uplink
transmission, then each LMU will produce a set of three tightly time synchronized LOPs (e.g., standard deviation of about 3 ns); and for three independent LOPs die =4. in this case, two UE position fixes is generated, each with standard deviation error tipos of 3.12 meters. Combining these two position fixes by averaging and/or other means/methods will further reduce the UE position fix error. One estimate is that the error reduction is proportional to the square root of the number of the UE position fixes. In the present disclosure, this number is equal two and the final UE position fix error aPQS FmAL is 2.21 meter; obtained as; 3.12 / 2.
|0533| In an aspect, several multichannel LMU (e.g., LMU device 2800) with relaxed synclironizatiori betwee these multichannel LMUs can be used for indoors and other
GPS/G SS denied environments. As an example, within the multichannel LMU device, the LMUs can be tightly synchronized (e.g., standard deviation of between about 3 ns and about 10 ns). Another embodiment takes advantage of the fact that a number of single channel small cell/LMU and/or small cells with integrated LMU devices electronics (the LMU functionality is embedded into the eNB) can be clustered (e.g., integrated, co-located, etc.) in a rackmount enclosure (FIG. 31, FIG. 32 and FIG. 33) and/or a cabinet, for example a 19 inch rack. Each single channel device antenna can be geographically distributed, like in DAS, The devices within a cluster can be tightly time synchronized (e.g... standard deviation of less than or equal to 1 ns). Multiple rackmount enclosures can be synchronized per communication requirements, for example VoLTE, whereby a low cost and low complexity approach can be used. Precise (tight) time synchronisation between a number of devices clustered (integrated) inside the rackmount enclosure cabinet is more readily achieved and. less costly than in the case of tightly time synchronizing a number of geographically distributed devices.
[0534| in another aspect, multiple LMUs can be integrated with (into) the DAS system as illustrated in FIG. 34. As an example, the LMU receivers can share the received signal(s) generated by the each DAS antenna, e.g., sharing DAS antennas. The actual distribution of these received signals depends upon the DAS implementation: active DAS vs. passive DAS. However, the LMU and DAS integration embodiment entails sharing the received signal(s) generated by the each DAS antenna with LMU receiver channel and creating an almanac that matches
(correlates) each DAS antenna coordinates with corresponding LMU/ LMU receiver channel. Again, the clustering approach and/or employing multichannel LMU(s) are preferable ways for LMU and DAS integration.
[0535] Also, in a similar fashion, it is possible to share the received signal(s) generated by the each small cell antenna with the LMU receiver channel. Here, the small cell's time synchronization can be relaxed, e.g. does not need to meet the locate requirements, while the LMU/LMU channels will require precision time synchronization. The clustering approach a»d or employing multichannel LMU(s) are a preferable way for LMU(s) for such option,
βδ36| integration of the LMU and the eNB into a single unit has a cost advantage over a combination of standalone eNB and LMU devices. However, unlike the integrated LMU and the eNB receiver, a standalone LMU receive channel does not have to process the data payload from UE. Furthermore, because the UE uplink ranging signals (SRS, sounding reference signal, in case of LTE) are repeatable and time synchronized (to the serving cell), each standalone LMU receive channel can support (be time multiplexed with) two or more antennae, for example serve two or more small cells. This, in turn, can lower the number of LMUs ( in small cells/DAS and/or other U-TDOA locate environments) and reduce the cost of the system (see also FIG. 28).
[0537] If wireless/cellular network E-SMLC server is lacking the functionality required for DL-OTDOA and/or U-TDOA techniques, this functionality can be carried out by a location server that can communicate with the UE and-'or LMU and the wheless/ceilular network infrastructure and/or a location services server (see FIG. 29 and FIG. 3Θ . Other configurations can b used.
{0538J In another aspect, one or more LMU devices (e.g., LMU 2802) can be deployed with WiFi infrastructure, for example, as illustrated in FIG. 35. Alternatively, a listening device could be used to monitor the LMU antenna in the same manner as the WiFi infrastructure. As such, the LMU devices and/or channel antennas servicing the LMUs can be co-located with one or more Wi.Fi/listening devices 3500, such as one or more WiFi access points (APs). As an example, the WiFi devices 3500 can be geographically distributed.
[0539J In one embodiment the WiFi device 3500 can be connected to a power source. An F analog portion 3502 (e.g., circuitry) of one or more LMU devices or channels can be integrated with the LMU antenna such that the RF analog portion 3502 can share the power source with the WiFi device 3S00 (see FIG. 35). As an example, the RF analog portion 3502 of the LMU device or channel can be connected via cable to the Uplink-Locate processor circuitry (e.g., Uplink-Locate processor 2810), which can include the baseband signal processing. As a further example, because there can be signal amplification between the antenna and the interconnecting cable between the RF analog portion 3S02 am! the baseband circuitry,, such an embodiment facilitates improved signal-to-noise ratio (SNR). Moreover, the RF analog portion 3502 can down-convert the received signal (e.g., down to the baseband) and, because the baseband signal frequencies are several magnitudes smaller than the received signal in antenna, the cable requirements can be relaxed. Such relaxation of cable requirements can translate into cost reduction of the connections and can significantly increase the transmission distance.
[0540] It is understood that the ranging signals are not limited to the SRS only and can utilize other reference signals, including MIMO, CRS (Cell-Specific Reference Signal), etc.
[0541] Having thus described the different embodiments of a system and methods, it should be apparent to those skilled in the art that certain advantages of the described method and apparatus have been achieved. In particular, it should be appreciated by those skilled in the art that a system for tracking and locating objects can be assembled using FGPA or ASIC and standard signal processing software/hardware combination at a very small incremental cost. Such a system is useful in a variety of applications, e.g. locating people in indoor or in outdoor environments, harsh and hostile environments etc.
[0542] It should also be appreciated that various modifications, adaptations, and alternative embodiments thereof may be made within the scope and spirit of the present invention.

Claims

What is Claimed:
1. A method for determin ing a location of one or more user equipment (UE) in a wireless system, the method comprising:
receiving reference signals via two or more co-located channels:
synchronizing timings of the two or more co-located channels within a standard deviation of less than or equal to a predetermined time based on a desired accuracy of a location for the wireless system; and
utilizi ng the received reference signals to calculate the location of at least one UE among the one or more OB.
2. The method of claim 1 , further comprising:
employing a muitipath mitigation processor configured to receive and process the received reference signals, wherein the muitipath mitigation processor utilizes a high-resolution spectrum estimation analysis to reduce spatial ambiguity associated with the received reference signals, the high-resolution spectrum estimation including estimating a model size for a number of frequency components of the received reference signals and calculating the location of the at least one UE based on a distribution of a plurality of artificial frequencies of the frequency components.
3. The method recited in claim 2, wherein the high-resolution spectrum estimation analysis employs one or more high-resolution spectrum estimation algorithms
4. The method as recited in claim 3, wherein the one or more high-resolution spectrum estimation algorithms include a Matrix Pencil algorithm.
5. The method of claim 1 , wherein each of the two or more co- located channels comprise a location management unit card or a small cell.
6. The method of claim 1, wherein the predetermined time is between about 3 ns and about 10 ns.
7. The method of claim 1 , wherein the received reference signals are uplink reference signals, downlink reference signals, distributed antennae system references signals, or a
combination thereof.
8. The method of claim 1, wherein the wireless system includes one or more nodes and each of the one or more nodes includes at least one sector, wherein a sector of each node is configured to communicate with a locate server unit (LSU), and wherein the step of utilizing is performed by the LSU or a combination of the one or more nodes and the LSU.
9. The method of claim 1, wherein the wireless system includes one or more nodes and each of the one or more nodes includes at least one sector, wherein a sector of each node is configured to communicate with a locate server unit (LSU), and wherein the step of utilizing is performed by the one or more UE, the LSU, the one or more nodes, or a combination thereof.
10. The method of claim 9, wherein the one or more UE are configured to communicate with the LSU
11. The method of claim 10, wherein the one or more UE, the LSU, the one or more nodes, or a combination thereof are configured to support multipath mitigation and reference signals processing to calculate the location of each UE.
.12. The method of claim 1 , wherein the wireless system is configured to include functionality of a LSU in a network SUPL server, a E-SMLC server, a LCS (Location Services) system, or a combination thereof.
13, The method of claim I , wherein the wireless system includes a LSU and one or more nodes, and wherein the LSU is configured to interface the one or mor nodes and network infrastructure of the wireless system.
14. The method of claim 1 , wherein the step of utilizing includes utilizing one or more line of position (LOP).
15. The method of claim 1 , wherein the reference signals are received from geographically distributed antennae.
1 . A method for determining a location of one or more user equipment (UE) in a wireless system, the method comprising:
receiving reference signals via a first location management unit having two or more co- loeated channels;
synchronizing timings of the two or more co-located channels within a first standard deviation of less than or equal to a first predetermined time based on a desired accuracy of a location for the wireless system;
receiving reference signals via a second location management unit having two or more co-located channels;
synchronizing timings of the co-located channels of the second location management unit within a second standard deviation of less than or equal to a second predetermined time based on the desired accuracy of the location for the wireless system; and utilizing the received reference signals from the first location management unit and the received reference signals from the second location management unit to calculate the location of at least one UE among the one or more UE.
17. The method as recited in claim 16, further comprising:
employing a multipath mitigation processor configured to receive and process the reference signals from the first location management unit and the second location management unit, wherein the multipath mitigation processor utilizes a high-resolution spectnim estimation analysis to reduce spatial ambiguity associated with the received reference signals of the first location management unit and the second location management unit, the high-resolution spectnim estimation including estimating a model size for a number of frequency components of the received reference signals of the first location management unit and the second location management unit and calculating the location of the at least one UE based on a distribution of a plurality of artificial frequencies of the frequency components.
18. The method as recited in claim 17, wherein the high-resolution spectrum estimation analysis employs one or more high-resolution spectrum estimation algorithms
19. The■ method as red led in claim i 8, wherein the one or more high-resolution spectnim estimation, algorithms include a Matrix Pencil algorithm.
20. The method of claim 16, wherein the first predetermined time and the second
predetermined time are between about 3 ns and about 10 n«
21. The method of claim ! 6, wherein the second predetermined time is between about 3 ns and about J O n .
22. The method of claim 16, wherein the received reference signals are uplink reference signals, downlink reference signals, distributed antennae system references signals, or a combination thereof.
23. The method of claim 16, wherein the step of utilizing includes utilizing one or more line of position (LOP).
24, The method of claim 16, wherein the first location management twit receives reference signals front geographically distributed antennae.
25. The method of claim 16, wherein the second location management unit recei ves reference signals from geographically distributed antennae.
26. The method of claim 16, wherein the first predetermined time and the second
predetermined time are greater than about 10 ns.
27. A method for determining a location of one or more user equipment (UE) in a wireless system, the method comprising:
receiving reference signals via a location management unit having two or more co- located channels, wherein the two or more co-located channels are tightly time synchronized with each other; and
utilizing the received reference signals to calculate a loca tion of at least one UE among the one or more UE.
28. The method of claim 27, wherein each of the two or more co-located channels comprise a location management unit card or a small ceil. 2s.),. The method of claim 27, wherein each of the two or more co-located chamois are integrated with the same rack mount system.
30. The method of claim 27, wherein the co-iocated channels are synchronized within a standard deviation of a predetermined time based on a desired accuracy of the iocatioii for the wireiess system
31 , The method of claim 30, wherein the predetermined time is between about 3 ns and about
10 as.
32. The method of claim 27, wherein the step of utilizing includes utilizing one or more line of position (LOP).
33. The method of claim 27, wherein the reference signals are received from geographically distributed antennae.
34. The method of claim 27, wherein the reference signals are received from a shared group of antennae in communication with the two or more co-located channels.
35. The method of claim 27, wherein the location management unit or an antenna servicing the location management unit is co-located with a WiFi device.
36. The method of claim 5, wherein the l cation management unit shares a power source with the WiFi device.
37. The method of claim 35, wherein the antenna servicing the location management unit shares a power source with the WiFi device.
PCT/US2015/043321 2005-12-15 2015-07-31 Partially synchronized multilateration/trilateration method and system for positional finding using rf WO2016019354A1 (en)

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US15/501,169 US10281557B2 (en) 2005-12-15 2015-07-31 Partially synchronized multilateration/trilateration method and system for positional finding using RF
KR1020177005271A KR102166578B1 (en) 2014-08-01 2015-07-31 Partially synchronized multilateration/trilateration method and system for positional finding using rf
CN201580053072.5A CN106922219B (en) 2014-08-01 2015-07-31 Partially synchronized multilateration/trilateration method and system for location finding using RF
EP15827815.0A EP3175668A4 (en) 2014-08-01 2015-07-31 Partially synchronized multilateration/trilateration method and system for positional finding using rf
JP2017505477A JP6557849B2 (en) 2014-08-01 2015-07-31 Partially synchronized multi-side / tri-side survey method and system for position finding using RF
US14/923,299 US9913244B2 (en) 2005-12-15 2015-10-26 Partially synchronized multilateration or trilateration method and system for positional finding using RF
US15/442,277 US10117218B2 (en) 2005-12-15 2017-02-24 Partially synchronized multilateration or trilateration method and system for positional finding using RF
US15/595,702 US11835639B2 (en) 2011-08-03 2017-05-15 Partially synchronized multilateration or trilateration method and system for positional finding using RF
US15/900,654 US11131744B2 (en) 2005-12-15 2018-02-20 Partially synchronized multilateration or trilateration method and system for positional finding using RF
US16/164,724 US10863313B2 (en) 2014-08-01 2018-10-18 Network architecture and methods for location services
US16/367,014 US10845453B2 (en) 2012-08-03 2019-03-27 Network architecture and methods for location services
US16/389,827 US11474188B2 (en) 2005-12-15 2019-04-19 Partially synchronized multilateration or trilateration method and system for positional finding using RF
US16/398,121 US10873830B2 (en) 2014-08-01 2019-04-29 Network architecture and methods for location services
US16/734,205 US11125850B2 (en) 2011-08-03 2020-01-03 Systems and methods for determining a timing offset of emitter antennas in a wireless network
US17/090,397 US11395105B2 (en) 2014-08-01 2020-11-05 Network architecture and methods for location services
US17/090,247 US11375341B2 (en) 2014-08-01 2020-11-05 Network architecture and methods for location services
US17/090,486 US11388554B2 (en) 2014-08-01 2020-11-05 Network architecture and methods for location services
US17/837,944 US11917493B2 (en) 2014-08-01 2022-06-10 Network architecture and methods for location services

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US15/595,702 Division US11835639B2 (en) 2011-08-03 2017-05-15 Partially synchronized multilateration or trilateration method and system for positional finding using RF
US15/900,654 Continuation US11131744B2 (en) 2005-12-15 2018-02-20 Partially synchronized multilateration or trilateration method and system for positional finding using RF
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EP3175668A1 (en) 2017-06-07
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