WO2015145713A1 - Rectifier circuit - Google Patents
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- WO2015145713A1 WO2015145713A1 PCT/JP2014/059073 JP2014059073W WO2015145713A1 WO 2015145713 A1 WO2015145713 A1 WO 2015145713A1 JP 2014059073 W JP2014059073 W JP 2014059073W WO 2015145713 A1 WO2015145713 A1 WO 2015145713A1
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
- H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
- H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M7/23—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only arranged for operation in parallel
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/12—Arrangements for reducing harmonics from ac input or output
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
- H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
- H02M7/06—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
- H02M7/08—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode arranged for operation in parallel
Definitions
- the present invention relates to a rectifier circuit that converts alternating current into direct current, and more particularly to a rectifier circuit that is suitable for suppressing harmonic components contained in an alternating current input current waveform.
- each three-phase bridge rectifier circuit is connected by an interphase reactor, and the interphase reactor is configured to supply a rectified current to a load via a smoothing reactor.
- the interphase reactor is provided with a plurality of taps symmetrically on the left and right, centering on the midpoint thereof, and connected via the respective taps, or a secondary winding is provided on the interphase reactor. It is reported that the midpoint of the secondary winding and the midpoint of the interphase reactor are combined, and a plurality of taps are provided symmetrically about the midpoint of the secondary winding and connected via the taps. Has been. That is, one is selected from those connected to a plurality of positions on the interphase reactor at a predetermined time, and the coupling current is taken out. Such a technique is described in Japanese Patent Laid-Open No. 59-213282.
- an object of the present invention is to provide a rectifier circuit that improves the suppression effect of harmonic components without increasing the number of switching elements and eliminates the need for an interphase reactor.
- the present invention provides a conversion transformer that outputs two sets of alternating currents having different phases, two sets of rectifier circuits that convert the alternating current into direct current, and the two sets of rectifier circuits.
- a coupling circuit that couples the outputs of the first and second outputs in parallel, the coupling circuit includes a first switching element, a second switching element, and one DC reactor, and is connected to one of the two sets of rectification circuits.
- the first input connection point, the second input connection point connected to the other of the two sets of rectifier circuits, and the output connection point are formed, and the first input connection point and the first input connection point are formed.
- each switching element One end of each switching element is connected, one end of the second switching element is connected to one end of the second input connection point, and the other end of each switching element is connected to one end of the DC reactor.
- the first switching element is switched at a cycle shorter than the cycle of the pulsating current output from the first rectifier circuit, and the second switching element is the first switching element. The switching is performed at a cycle shorter than the cycle of the pulsating current output from the rectifier circuit 2.
- a transformer for conversion that outputs two sets of three-phase alternating currents having different phases, two sets of bridge circuits that respectively convert the three-phase alternating currents into direct current, and a coupling that couples the two sets of direct current outputs in parallel
- the coupling circuit has two switching elements and one DC reactor, and one end of each switching element is connected to each of the DC output ends of the two sets of bridge circuits, and the other end Are connected to one end of the DC reactor.
- a parallel resonance circuit is connected to the other end of the DC reactor.
- the effect of suppressing harmonic components can be improved without increasing the number of switching elements, and an interphase reactor can be made unnecessary.
- FIG. 1 is a main circuit configuration diagram according to Embodiment 1.
- FIG. FIG. 3 is a diagram illustrating each DC current ratio-phase angle diagram according to the first embodiment.
- FIG. 3 is a voltage-phase angle diagram of each DC voltage according to the first embodiment.
- FIG. 6 is a primary side converted current-phase angle diagram of each secondary side current according to the first embodiment.
- 1 is a primary current-phase angle diagram according to Embodiment 1.
- FIG. FIG. 6 is a main circuit configuration diagram according to the second embodiment.
- FIG. 6 is a diagram illustrating each DC current ratio-phase angle diagram according to the second embodiment.
- FIG. 6 is a -phase angle diagram of each DC voltage according to the second embodiment.
- FIG. 6 is a primary current converted value-phase angle diagram of each secondary current according to the second embodiment.
- FIG. 10 is a primary side current-phase angle diagram according to the second embodiment.
- FIG. 6 is a main circuit configuration diagram according to the third embodiment.
- FIG. 10 is a -phase angle diagram of each DC voltage according to the third embodiment.
- FIG. 12 is a primary side current converted value-phase angle diagram of each secondary side current according to the third embodiment.
- FIG. 10 is a primary current-phase angle diagram according to the third embodiment.
- FIG. 6 is a main circuit configuration diagram according to the fourth embodiment. The voltage-phase angle diagram of each voltage which concerns on an Example. Switching element operation-phase angle diagram according to the embodiment.
- FIG. 1 is a main circuit configuration diagram of the rectifier circuit of this embodiment
- FIG. 2 is a DC current ratio-phase angle diagram of the rectifier circuit
- FIG. 3 is a voltage-phase angle diagram of each DC voltage in the rectifier circuit
- FIG. 4 is a primary current converted value—phase angle diagram of each secondary current of the conversion transformer u phase in the rectifier circuit
- FIG. 5 is a primary phase current—phase angle diagram of the u phase.
- the rectifier circuit includes a primary winding 5 (Y connection, number of turns N0) to which a three-phase AC voltage is input from the outside, and a three-phase circuit having the same phase as the primary three-phase AC voltage.
- Secondary winding 6 (Y connection, number of turns N1) for outputting AC voltage
- secondary winding 7 ( ⁇ connection for outputting three-phase AC voltage that is 30 ° out of phase with the primary three-phase AC voltage) ,
- the number of turns N2 The number of turns N2).
- the primary winding 5 (Y connection, number of turns N0) is a Y connection of the windings u0, v0, w0, and the currents flowing through the windings u0, v0, w0 are defined as Iu0, Iv0, Iw0. .
- the primary winding 5 (Y connection, number of turns N0) has system input force terminals u, v, and w, and a system voltage is applied to the system input force terminals u, v, and w. For example, a system voltage of 50 Hz or 60 Hz is applied.
- the secondary winding 6 (Y connection, number of turns N1) is Y connection of the windings u1, v1, and w1, and the currents flowing through the respective windings u1, v1, and w1 are defined as Iu1, Iv1, and Iw1.
- the secondary winding 7 ( ⁇ connection, number of turns N2) is obtained by ⁇ connection of the windings u2, v2, and w2, and the currents flowing in the respective windings u2, v2, and w2 are defined as Iu2, Iv2, and Iw2. .
- the bridge circuit 21 includes a diode DI11 and a diode DI12 connected in series, a diode DI13 and a diode DI14 connected in series, and a diode DI15 and a diode DI16 connected in series. It is.
- a connection point between the diode DI11 and the diode DI12, a connection point between the diode DI13 and the diode DI14, and a connection point between the diode DI15 and the diode DI16 are connected to the windings u1, v1, and w1, respectively.
- One of the parallel connection points is connected to the input connection point a of the coupling circuit 10, and the other of the parallel connection points is connected to the output connection point d to the load 60.
- a capacitor C1 is connected between the parallel connection points.
- the voltage between the parallel connection points is defined as Va.
- the current to the input connection point a of the coupling circuit 10 is defined as Ia.
- the bridge circuit 22 includes a diode DI21 and a diode DI22 connected in series, a diode DI23 and a diode DI24 connected in series, and a diode DI25 and a diode DI26 connected in series. It is.
- the connection point of the diode DI21 and the diode DI22, the connection point of the diode DI23 and the diode DI24, the connection point of the diode DI25 and the diode DI26 are respectively the connection point of the windings w2 and u2, the connection point of the windings u2 and v2, and the winding point. Connected to the connection point of v2 and w2.
- One of the parallel connection points is connected to the input connection point b of the coupling circuit 10, and the other of the parallel connection points is connected to the output connection point d to the load 60.
- a capacitor C2 is connected between the parallel connection points.
- the voltage between the parallel connection points is defined as Vb.
- the current to the input connection point b of the coupling circuit 10 is defined as Ia.
- the direct current outputs are coupled in parallel by the coupling circuit 10.
- This coupling circuit 10 has two switching elements SW1 and SW2 and one DC reactor L0, and one ends of the switching elements SW1 and SW2 are connected to the DC output ends of the bridge circuits 21 and 22 and connection points a and b, respectively. The other end is connected to one end of the DC reactor.
- a load 60 is connected to the other end of the DC reactor at a connection point c.
- the switching element for example, a well-known element such as an IGBT and a reverse blocking type element can be used. When using a switching element that is not a reverse blocking type, a diode may be added in series.
- a sine wave is considered as a three-phase AC voltage waveform input from the outside.
- a system voltage for example, a system voltage of 50 Hz or 60 Hz
- the system input force terminals u, v, w are externally applied.
- Vu, Vv, and Vw are defined as Vu, Vv, and Vw.
- the horizontal axis (phase angle) in FIG. 17 is shown as the phase angle of the system voltage applied to the system input force terminals u, v, and w. That is, for example, one cycle of the system voltage given at 50 Hz or 60 Hz is indicated as 360 ° (deg).
- the reference for the phase angle is the same in the description and drawings described later.
- the u-phase voltage Vu is used as a reference, and the v-phase voltage Vv is 120 ° and the w-phase voltage Vw is 240 °, which are out of phase.
- the output voltages Va and Vb from the bridge circuits 21 and 22 are as shown in FIG. That is, it has the same waveform shape including an AC fluctuation component whose basic frequency is 6 times the input AC frequency, and becomes a voltage waveform whose phase is shifted by 30 ° from each other (corresponding to 60 ° of the system voltage is one cycle. ).
- the output voltages Va and Vb are interchanged with each other every 30 °.
- the switching elements SW1 and SW2 are ON / OFF controlled.
- An ON / OFF period corresponding to 1.5 ° of the system voltage is set as an ON / OFF cycle, and a predetermined period is turned ON, and the remaining period is turned OFF.
- SW2 is always turned on, and SW1 is turned on / off, thereby passing through the connection point a and flowing through the connection point c, and passing through the connection point b.
- the ratio of the current Ib flowing through the connection point c can be changed.
- Vc ⁇ Ic Va ⁇ Ia + Vb ⁇ Ib Formula (1)
- Vc Va ⁇ Ia / Ic + Vb ⁇ Ib / Ic (2)
- the ratio of the currents Ia and Ib can be changed by ON / OFF control of the switching element SW1 or SW2, and the value of Vc can be arbitrarily controlled in the intermediate range between Va and Vb.
- the current ratio Ia / Ic is 0.5 + ⁇ at a phase angle of 0 °, decreases monotonously to 0.5- ⁇ at a phase angle of 30 °, and increases monotonically from here.
- Ib / Ic is 0.5- ⁇ at a phase angle of 0 °, and is slightly increased to 0.5 + ⁇ at a phase angle of 30 °, so that the operation of 0.5 + ⁇ is repeated at an angle of 60 °.
- the switching elements SW1 and SW2 are ON / OFF controlled so as to repeat the operation of monotonically decreasing and reaching 0.5- ⁇ at a phase angle of 60 °. In this way, if Ia / Ic and Ib / Ic are controlled, Vc can be set to a constant value as shown in FIG. 3, and it is not necessary to install a DC filter before the load.
- FIG. 4 shows a primary-side converted current-phase angle diagram of each secondary-side current of the conversion transformer u-phase when the above control is performed.
- the u-phase primary side current is the sum of the primary side current converted values of these secondary side currents. This is shown in FIG. Even in the current waveform shown in FIG. 5, it can be said that the harmonic component is sufficiently close to a sine wave depending on the application and has a small harmonic component.
- Example 2 will be described with reference to FIGS. In the figure, the same reference numerals are basically equivalent to those in the first embodiment. In the following embodiments, the description will focus on the differences from the embodiments described so far, and the portions that are not described are the same as long as they are not technically different from the embodiments described so far. Is.
- FIG. 6 is a main circuit configuration diagram of the rectifier circuit of this embodiment
- FIG. 7 is a DC current ratio-phase angle diagram of the rectifier circuit
- FIG. 8 is a voltage-phase angle diagram of each DC voltage in the rectifier circuit
- FIG. 9 is a primary current converted value—phase angle diagram of each secondary current of the conversion transformer u phase in the rectifier circuit
- FIG. 5 is a primary phase current—phase angle diagram of the u phase.
- connection positions of the switching current smoothing capacitors C1 and C2 in the first embodiment shown in FIG. 1 are between the connection point a and the connection point c, and between the connection point b and the connection point c, respectively.
- the parallel resonance circuit 40 is inserted between the connection point c and the load 60.
- the inductance of the reactor L41 and the electrostatic capacity of the capacitor C41 constituting the parallel resonance circuit 40 are selected so as to resonate at a frequency 12 times the frequency of the primary side three-phase alternating current.
- a sine wave having a frequency 12 times the frequency of the primary three-phase alternating current is generated in the DC voltage as the voltage waveform of Vc.
- the superimposition can be output.
- the superimposed sine wave is removed by the parallel resonance circuit 40, and only a DC voltage is applied to the load 60. If the load 60 is a pure resistance load, the current Ic has a constant value.
- FIG. 9 is a primary side current converted value-phase angle diagram of each secondary side current of the conversion transformer u phase when the above control is performed.
- the primary current of the u phase is the sum of the primary current conversion values of these secondary currents. This is shown in FIG. In the current waveform shown in FIG. 10, further suppression of harmonic components is realized as compared with that of the first embodiment.
- Example 3 will be described with reference to FIGS. 11 is a main circuit configuration diagram of the rectifier circuit of the present embodiment
- FIG. 12 is a DC current ratio-phase angle diagram of the rectifier circuit
- FIG. 13 is a voltage-phase angle diagram of each DC voltage in the rectifier circuit
- FIG. 14 is a primary current converted value—phase angle diagram of each secondary current of the transformer u phase for conversion in the rectifier circuit
- FIG. 15 is a primary current-phase angle diagram of the u phase.
- the second embodiment configures the parallel resonant circuit 40 to have a primary three-phase frequency. It is configured to resonate at a frequency 12 times and 24 times the frequency of the alternating current.
- the inductance of the reactor L41 and the capacitance of the capacitor C41 constituting the parallel resonant circuit 40 are selected so as to resonate at a frequency 12 times the frequency of the primary side three-phase alternating current, and the inductance and the capacitor of the reactor L42.
- the capacitance of C42 is selected so as to resonate at a frequency 24 times the frequency of the primary side three-phase alternating current.
- a sine wave having a frequency 12 times the frequency of the primary side three-phase alternating current is added to the DC voltage as shown in FIG.
- a signal on which a sine wave having a frequency 24 times is superimposed can be output.
- the superimposed sine wave is removed by the parallel resonance circuit 40, and only a DC voltage is applied to the load 60. If the load 60 is a pure resistance load, the current Ic has a constant value.
- FIG. 14 shows a primary current converted value-phase angle diagram of each secondary current of the conversion transformer u-phase when the above control is performed.
- the primary current of the u phase is the sum of the primary current conversion values of these secondary currents. This is shown in FIG.
- the current waveform shown in FIG. 15 realizes further suppression of higher harmonic components than that of the second embodiment.
- FIG. 16 is a main circuit configuration diagram of the rectifier circuit of this embodiment.
- the diodes DI11, DI13, DI15, DI21, DI23, and DI25 of the bridge circuit in the first embodiment shown in FIG. 1 are replaced with GTO, GT11, GT13, GT15, GT21, GT23, and GT25, respectively. It has become.
- the basic circuit operation is the same as in the first embodiment.
- by replacing the GTO there is an effect that the current can be surely interrupted by turning off the GTO at the time of abnormality.
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Abstract
Provided is a rectifier circuit in which the effect of suppressing harmonic components is improved without increasing the number of switching elements and no interphase reactor is required.
The rectifier circuit has: a conversion transformer that outputs two sets of three-phase alternating currents having different phases; two sets of bridge circuits respectively converting the three-phase alternating currents to direct currents; and a coupling circuit coupling the two sets of direct current outputs in parallel. The coupling circuit has two switching elements and a DC reactor, wherein one ends of the switching elements are respectively connected to DC output terminals of the two sets of bridge circuits and the other ends are both connected to one end of the DC reactor.
Description
本発明は、交流を直流に変換する整流回路に係り、特に交流入力電流波形に含まれる高調波成分を抑制するのに好適な整流回路に関する。
The present invention relates to a rectifier circuit that converts alternating current into direct current, and more particularly to a rectifier circuit that is suitable for suppressing harmonic components contained in an alternating current input current waveform.
The present invention relates to a rectifier circuit that converts alternating current into direct current, and more particularly to a rectifier circuit that is suitable for suppressing harmonic components contained in an alternating current input current waveform.
三相交流を直流に変換する整流回路として、位相が異なる2組の交流を出力する変換用変圧器を利用した整流回路が知られている。この整流回路では、例えば、ブリッジ整流器を2組用意し、一方のブリツジ整流回路を変換用変圧器の一方の相の出力に接続し、他方のブリツジ整流回路を変換用変圧器の他方の相の出力に接続し、さらに、各々の3相ブリッジ整流回路を相間リアクトルで接続し、この相間リアクトルについて平滑用のリアクトルを介して負荷に整流した電流を供給するように構成されるものである。
As a rectifier circuit that converts three-phase alternating current into direct current, a rectifier circuit that uses a conversion transformer that outputs two sets of alternating currents with different phases is known. In this rectifier circuit, for example, two sets of bridge rectifiers are prepared, one bridge rectifier circuit is connected to the output of one phase of the conversion transformer, and the other bridge rectifier circuit is connected to the other phase of the conversion transformer. Connected to the output, each three-phase bridge rectifier circuit is connected by an interphase reactor, and the interphase reactor is configured to supply a rectified current to a load via a smoothing reactor.
このように構成された整流回路において、交流入力電流波形に含まれる高調波成分を抑制する回路構成手法が報告されている。例えば、相間リアクトルに、その中点を中心にして、左右に対称的に複数個のタップを設け、その各タップを介して接続する、或いは、相間リアクトルに2次巻線を設けて、この2次巻線の中点と相間リアクトルの中点とを結合し、2次巻線の中点を中心にして左右に対称的に複数個のタップを設け、そのタップを介して接続するものが報告されている。すなわち、所定時刻において当該相間リアクトル上の複数の位置に接続されたもののうちから1つを選んで結合電流を取出すものである。このような技術は、特開昭59-213282号公報に記載されている。
In the rectifier circuit configured as described above, a circuit configuration technique for suppressing harmonic components included in the AC input current waveform has been reported. For example, the interphase reactor is provided with a plurality of taps symmetrically on the left and right, centering on the midpoint thereof, and connected via the respective taps, or a secondary winding is provided on the interphase reactor. It is reported that the midpoint of the secondary winding and the midpoint of the interphase reactor are combined, and a plurality of taps are provided symmetrically about the midpoint of the secondary winding and connected via the taps. Has been. That is, one is selected from those connected to a plurality of positions on the interphase reactor at a predetermined time, and the coupling current is taken out. Such a technique is described in Japanese Patent Laid-Open No. 59-213282.
In the rectifier circuit configured as described above, a circuit configuration technique for suppressing harmonic components included in the AC input current waveform has been reported. For example, the interphase reactor is provided with a plurality of taps symmetrically on the left and right, centering on the midpoint thereof, and connected via the respective taps, or a secondary winding is provided on the interphase reactor. It is reported that the midpoint of the secondary winding and the midpoint of the interphase reactor are combined, and a plurality of taps are provided symmetrically about the midpoint of the secondary winding and connected via the taps. Has been. That is, one is selected from those connected to a plurality of positions on the interphase reactor at a predetermined time, and the coupling current is taken out. Such a technique is described in Japanese Patent Laid-Open No. 59-213282.
上記の従来技術では、交流入力電流波形に含まれる高調波成分の抑制効果を向上するためには、相間リアクトルのタップの数を多くする必要があるので、それに伴い、整流回路の素子数を増やす必要があった。また、相間リアクトルの両端には入力交流の周波数よりも高い周波数を基本周波数とする交流電圧が加わることになるが、この交流電圧により生じる交流電流を抑制するためには相間リアクトルのインダクタンスを大きくとる必要があった。
In the above prior art, in order to improve the suppression effect of the harmonic component included in the AC input current waveform, it is necessary to increase the number of taps of the interphase reactor, and accordingly, the number of elements of the rectifier circuit is increased. There was a need. In addition, an AC voltage having a fundamental frequency higher than the input AC frequency is applied to both ends of the interphase reactor. In order to suppress the AC current generated by the AC voltage, the inductance of the interphase reactor is increased. There was a need.
そこで本発明の目的は、スイッチング素子の数を増やすこと無く高調波成分の抑制効果を向上するとともに、相間リアクトルを不要とした整流回路を提供することにある。
Accordingly, an object of the present invention is to provide a rectifier circuit that improves the suppression effect of harmonic components without increasing the number of switching elements and eliminates the need for an interphase reactor.
Accordingly, an object of the present invention is to provide a rectifier circuit that improves the suppression effect of harmonic components without increasing the number of switching elements and eliminates the need for an interphase reactor.
上記の課題を解決するために、本発明は、位相が異なる2組の交流を出力する変換用変圧器と、前記交流を各々直流に変換する2組の整流回路と、前記2組の整流回路の出力を並列結合する結合回路とを有する整流回路において、前記結合回路が第1のスイッチング素子と第2のスイッチング素子と1つの直流リアクトルとを有するとともに、前記2組の整流回路の一方に接続される第1の入力接続点と、前記2組の整流回路の他方に接続される第2の入力接続点と、出力接続点が形成されており、前記第1の入力接続点と前記第1のスイッチング素子の一端が接続され、前記第2の入力接続点と前記第2のスイッチング素子の一端が接続され、夫々のスイッチング素子の他端がともに前記直流リアクトルの一端に接続され、前記直流リアクトルの他端が前記出力接続点に接続され、前記第1のスイッチング素子は前記第1の整流回路の出力する脈流の周期よりも短い周期でスイッチングし、前記第2のスイッチング素子は前記第2の整流回路の出力する脈流の周期よりも短い周期でスイッチングするように構成した。
In order to solve the above-described problems, the present invention provides a conversion transformer that outputs two sets of alternating currents having different phases, two sets of rectifier circuits that convert the alternating current into direct current, and the two sets of rectifier circuits. And a coupling circuit that couples the outputs of the first and second outputs in parallel, the coupling circuit includes a first switching element, a second switching element, and one DC reactor, and is connected to one of the two sets of rectification circuits. The first input connection point, the second input connection point connected to the other of the two sets of rectifier circuits, and the output connection point are formed, and the first input connection point and the first input connection point are formed. One end of each switching element is connected, one end of the second switching element is connected to one end of the second input connection point, and the other end of each switching element is connected to one end of the DC reactor. And the first switching element is switched at a cycle shorter than the cycle of the pulsating current output from the first rectifier circuit, and the second switching element is the first switching element. The switching is performed at a cycle shorter than the cycle of the pulsating current output from the rectifier circuit 2.
具体的には、位相が異なる2組の三相交流を出力する変換用変圧器と、前記三相交流を各々直流に変換する2組のブリッジ回路と、2組の直流出力を並列結合する結合回路とを有しており、前記結合回路が、2つのスイッチング素子と1つの直流リアクトルとを有し、各スイッチング素子の一端が前記2組のブリッジ回路の直流出力端に各々接続され、他端が共に直流リアクトルの一端に接続される様に構成した。また、前記直流リアクトルの他端に並列共振回路が接続されるように構成した。
Specifically, a transformer for conversion that outputs two sets of three-phase alternating currents having different phases, two sets of bridge circuits that respectively convert the three-phase alternating currents into direct current, and a coupling that couples the two sets of direct current outputs in parallel The coupling circuit has two switching elements and one DC reactor, and one end of each switching element is connected to each of the DC output ends of the two sets of bridge circuits, and the other end Are connected to one end of the DC reactor. In addition, a parallel resonance circuit is connected to the other end of the DC reactor.
Specifically, a transformer for conversion that outputs two sets of three-phase alternating currents having different phases, two sets of bridge circuits that respectively convert the three-phase alternating currents into direct current, and a coupling that couples the two sets of direct current outputs in parallel The coupling circuit has two switching elements and one DC reactor, and one end of each switching element is connected to each of the DC output ends of the two sets of bridge circuits, and the other end Are connected to one end of the DC reactor. In addition, a parallel resonance circuit is connected to the other end of the DC reactor.
本発明によれば、整流回路において、スイッチング素子の数を増やすこと無く高調波成分の抑制効果を向上するとともに、相間リアクトルを不要とすることができる。
According to the present invention, in the rectifier circuit, the effect of suppressing harmonic components can be improved without increasing the number of switching elements, and an interphase reactor can be made unnecessary.
According to the present invention, in the rectifier circuit, the effect of suppressing harmonic components can be improved without increasing the number of switching elements, and an interphase reactor can be made unnecessary.
以下、実施例を図面を用いて説明する。なお、本発明の本質を明らかにするため、以下の実施例では回路における各スイッチング素子、ダイオードにおける電圧低下や配線の抵抗、インダクタンス、寄生容量、変圧器の励磁インダクタンス等が無視できる理想的な状態であるものとして説明する。
Hereinafter, examples will be described with reference to the drawings. In order to clarify the essence of the present invention, in the following embodiments, each switching element in the circuit, the voltage drop in the diode, the resistance of the wiring, the inductance, the parasitic capacitance, the exciting inductance of the transformer, etc. can be ignored. It is assumed that
実施例1について図1ないし図5、図17ないし図18を用いて説明する。図1は本実施例の整流回路の主回路構成図、図2はその整流回路の各直流電流比率-位相角線図、図3は当該整流回路における各直流電圧の電圧-位相角線図、図4は当該整流回路における変換用変圧器u相の各二次側電流の一次側電流換算値-位相角線図、図5はu相の一次側電流-位相角線図である。
Example 1 will be described with reference to FIGS. 1 to 5 and FIGS. 17 to 18. FIG. 1 is a main circuit configuration diagram of the rectifier circuit of this embodiment, FIG. 2 is a DC current ratio-phase angle diagram of the rectifier circuit, FIG. 3 is a voltage-phase angle diagram of each DC voltage in the rectifier circuit, FIG. 4 is a primary current converted value—phase angle diagram of each secondary current of the conversion transformer u phase in the rectifier circuit, and FIG. 5 is a primary phase current—phase angle diagram of the u phase.
本実施例の整流回路は、図1において、外部から三相交流電圧が入力される1次側巻線5(Y結線、巻数N0)と、1次側三相交流電圧と同じ位相の三相交流電圧を出力する2次側巻線6(Y結線、巻数N1)と、1次側三相交流電圧と位相が30°ずれた三相交流電圧を出力する2次側巻線7(Δ結線、巻数N2)とを有する変換用変圧器30を有している。1次側巻線5(Y結線、巻数N0)は、巻線u0、v0、w0をY結線したもので、各々の巻線u0、v0、w0に流れる電流をIu0、Iv0、Iw0と定義する。1次側巻線5(Y結線、巻数N0)は、系統入流力端子u、v、wを有し、この系統入流力端子u、v、wには、系統電圧が印加される。例えば、50Hzあるいは60Hzの系統電圧が印加される。2次側巻線6(Y結線、巻数N1)は、巻線u1、v1、w1をY結線したもので、各々の巻線u1、v1、w1に流れる電流をIu1、Iv1、Iw1と定義する。2次側巻線7(Δ結線、巻数N2)は、巻線u2、v2、w2をΔ結線したもので、各々の巻線u2、v2、w2に流れる電流をIu2、Iv2、Iw2と定義する。
In FIG. 1, the rectifier circuit according to this embodiment includes a primary winding 5 (Y connection, number of turns N0) to which a three-phase AC voltage is input from the outside, and a three-phase circuit having the same phase as the primary three-phase AC voltage. Secondary winding 6 (Y connection, number of turns N1) for outputting AC voltage, and secondary winding 7 (Δ connection for outputting three-phase AC voltage that is 30 ° out of phase with the primary three-phase AC voltage) , The number of turns N2). The primary winding 5 (Y connection, number of turns N0) is a Y connection of the windings u0, v0, w0, and the currents flowing through the windings u0, v0, w0 are defined as Iu0, Iv0, Iw0. . The primary winding 5 (Y connection, number of turns N0) has system input force terminals u, v, and w, and a system voltage is applied to the system input force terminals u, v, and w. For example, a system voltage of 50 Hz or 60 Hz is applied. The secondary winding 6 (Y connection, number of turns N1) is Y connection of the windings u1, v1, and w1, and the currents flowing through the respective windings u1, v1, and w1 are defined as Iu1, Iv1, and Iw1. . The secondary winding 7 (Δ connection, number of turns N2) is obtained by Δ connection of the windings u2, v2, and w2, and the currents flowing in the respective windings u2, v2, and w2 are defined as Iu2, Iv2, and Iw2. .
ブリッジ回路21は、ダイオードDI11とダイオードDI12を直列に接続したものと、ダイオードDI13とダイオードDI14を直列に接続したものと、ダイオードDI15とダイオードDI16を直列に接続したものとを、並列に接続したものである。ダイオードDI11とダイオードDI12の接続点、ダイオードDI13とダイオードDI14の接続点、ダイオードDI15とダイオードDI16の接続点は、各々、巻線u1、v1、w1に接続される。並列接続点の一方は、結合回路10の入力接続点aに接続され、並列接続点の他方は、負荷60への出力接続点dに接続される。並列接続点間にはコンデンサC1が接続される。並列接続点間の電圧はVaと定義される。結合回路10の入力接続点aへの電流はIaと定義される。
The bridge circuit 21 includes a diode DI11 and a diode DI12 connected in series, a diode DI13 and a diode DI14 connected in series, and a diode DI15 and a diode DI16 connected in series. It is. A connection point between the diode DI11 and the diode DI12, a connection point between the diode DI13 and the diode DI14, and a connection point between the diode DI15 and the diode DI16 are connected to the windings u1, v1, and w1, respectively. One of the parallel connection points is connected to the input connection point a of the coupling circuit 10, and the other of the parallel connection points is connected to the output connection point d to the load 60. A capacitor C1 is connected between the parallel connection points. The voltage between the parallel connection points is defined as Va. The current to the input connection point a of the coupling circuit 10 is defined as Ia.
ブリッジ回路22は、ダイオードDI21とダイオードDI22を直列に接続したものと、ダイオードDI23とダイオードDI24を直列に接続したものと、ダイオードDI25とダイオードDI26を直列に接続したものとを、並列に接続したものである。ダイオードDI21とダイオードDI22の接続点、ダイオードDI23とダイオードDI24の接続点、ダイオードDI25とダイオードDI26の接続点は、各々、巻線w2とu2の接続点、巻線u2とv2の接続点、巻線v2とw2の接続点に接続される。並列接続点の一方は、結合回路10の入力接続点bに接続され、並列接続点の他方は、負荷60への出力接続点dに接続される。並列接続点間にはコンデンサC2が接続される。並列接続点間の電圧はVbと定義される。結合回路10の入力接続点bへの電流はIaと定義される。
The bridge circuit 22 includes a diode DI21 and a diode DI22 connected in series, a diode DI23 and a diode DI24 connected in series, and a diode DI25 and a diode DI26 connected in series. It is. The connection point of the diode DI21 and the diode DI22, the connection point of the diode DI23 and the diode DI24, the connection point of the diode DI25 and the diode DI26 are respectively the connection point of the windings w2 and u2, the connection point of the windings u2 and v2, and the winding point. Connected to the connection point of v2 and w2. One of the parallel connection points is connected to the input connection point b of the coupling circuit 10, and the other of the parallel connection points is connected to the output connection point d to the load 60. A capacitor C2 is connected between the parallel connection points. The voltage between the parallel connection points is defined as Vb. The current to the input connection point b of the coupling circuit 10 is defined as Ia.
各々の三相交流を直流に変換するブリッジ回路21、22は、結合回路10により各々の直流出力が並列結合される。この結合回路10は、2つのスイッチング素子SW1、SW2と1つの直流リアクトルL0とを有し、スイッチング素子SW1、SW2の一端が前記ブリッジ回路21、22の直流出力端と接続点a,bで各々接続され、他端が共に直流リアクトルの一端に接続されている。そして前記直流リアクトルの他端に接続点cで負荷60が接続されている。なお、スイッチング素子は、例えば、IGBT等の周知の素子であって逆阻止型のものが使用できる。逆阻止型でないスイッチング素子を使用する場合は各々直列にダイオードを追加すれば良い。
In the bridge circuits 21 and 22 for converting each three-phase alternating current into direct current, the direct current outputs are coupled in parallel by the coupling circuit 10. This coupling circuit 10 has two switching elements SW1 and SW2 and one DC reactor L0, and one ends of the switching elements SW1 and SW2 are connected to the DC output ends of the bridge circuits 21 and 22 and connection points a and b, respectively. The other end is connected to one end of the DC reactor. A load 60 is connected to the other end of the DC reactor at a connection point c. As the switching element, for example, a well-known element such as an IGBT and a reverse blocking type element can be used. When using a switching element that is not a reverse blocking type, a diode may be added in series.
次に本実施例の整流回路の動作を説明する。一連の説明において、外部から入力される三相交流電圧波形として正弦波を考える。図17において、図1に示すように系統入流力端子u、v、wに系統電圧(例えば、50Hzあるいは60Hzの系統電圧)が印加されるところ、外部からから系統入流力端子u、v、wに与えられる電圧がVu、Vv、Vwと定義されて示される。ここで、図17において図の横軸(位相角)は、系統入流力端子u、v、wに与えられる系統電圧の位相角として示される。すなわち、例えば、50Hzあるいは60Hzで与えられる系統電圧の1周期を360°(deg)として示される。この位相角の基準は後述する説明・図面でも同様である。
Next, the operation of the rectifier circuit of this embodiment will be described. In a series of descriptions, a sine wave is considered as a three-phase AC voltage waveform input from the outside. In FIG. 17, when a system voltage (for example, a system voltage of 50 Hz or 60 Hz) is applied to the system input force terminals u, v, w as shown in FIG. 1, the system input force terminals u, v, w are externally applied. Are defined as Vu, Vv, and Vw. Here, the horizontal axis (phase angle) in FIG. 17 is shown as the phase angle of the system voltage applied to the system input force terminals u, v, and w. That is, for example, one cycle of the system voltage given at 50 Hz or 60 Hz is indicated as 360 ° (deg). The reference for the phase angle is the same in the description and drawings described later.
図17に示すとおりu相電圧Vuを基準とし、これに対してv相電圧Vvは120°、w相電圧Vwは240°、夫々位相がずれているものとする。変換用変圧器30の巻数比を、N1/N0=1、N2/N0=√3として構成すれば、ブリッジ回路21、22からの出力電圧Va、Vbは夫々図17に示したものとなる。即ち、基本周波数が入力交流周波数の6倍となる交流変動分を含んだ同一の波形形状を有し、互いに位相が30°ずれた電圧波形となる(系統電圧の60°相当が1周期となる)。そして、出力電圧Va、Vbは互いに30°毎に電圧の大小が入れ替わるものとなっている。
As shown in FIG. 17, the u-phase voltage Vu is used as a reference, and the v-phase voltage Vv is 120 ° and the w-phase voltage Vw is 240 °, which are out of phase. If the turns ratio of the conversion transformer 30 is configured as N1 / N0 = 1 and N2 / N0 = √3, the output voltages Va and Vb from the bridge circuits 21 and 22 are as shown in FIG. That is, it has the same waveform shape including an AC fluctuation component whose basic frequency is 6 times the input AC frequency, and becomes a voltage waveform whose phase is shifted by 30 ° from each other (corresponding to 60 ° of the system voltage is one cycle. ). The output voltages Va and Vb are interchanged with each other every 30 °.
ここで、図18に示した様に、スイッチング素子SW1、SW2をON/OFF制御する。系統電圧の1.5°相当をON/OFF周期としてその中の所定の期間をオンとして残りの期間をオフとする。出力電圧Va≧Vbとなる位相角の範囲においてSW2を常時ONとし、SW1をON/OFF制御することにより接続点aを通過して接続点cに流れる電流Iaと、接続点bを通過して接続点cに流れる電流Ibの比を変えることができる。ここで、電流Ia、Ibは一回のON/OFF動作における平均電流の意味で使用しており、接続点cを流れる電流Icとの関係式は以下となる。
(Ia+Ib=Ic)または(Ia/Ic+Ib/Ic=1)
同様にVa≦Vbとなる位相角の範囲においてはSW1を常時ONとし、SW2をON/OFF制御することにより電流IaとIbの比率を変えることができる。また、結合回路10はスイッチング素子とリアクトルのみで構成されていて、理想的にはエネルギー損失が生じないことから、一回のON/OFF周期が十分短く、当該時間における電圧の変化が無視できる場合、電圧と電流の関係式として次式(1)または(2)が成り立つ。
Vc・Ic=Va・Ia+Vb・Ib … 式(1)
または
Vc=Va・Ia/Ic+Vb・Ib/Ic … 式(2)
Here, as shown in FIG. 18, the switching elements SW1 and SW2 are ON / OFF controlled. An ON / OFF period corresponding to 1.5 ° of the system voltage is set as an ON / OFF cycle, and a predetermined period is turned ON, and the remaining period is turned OFF. In the phase angle range where the output voltage Va ≧ Vb, SW2 is always turned on, and SW1 is turned on / off, thereby passing through the connection point a and flowing through the connection point c, and passing through the connection point b. The ratio of the current Ib flowing through the connection point c can be changed. Here, the currents Ia and Ib are used to mean the average current in one ON / OFF operation, and the relational expression with the current Ic flowing through the connection point c is as follows.
(Ia + Ib = Ic) or (Ia / Ic + Ib / Ic = 1)
Similarly, in the phase angle range where Va ≦ Vb, the ratio of the currents Ia and Ib can be changed by always turning SW1 ON and controlling SW2 ON / OFF. In addition, since thecoupling circuit 10 is composed only of a switching element and a reactor, and ideally no energy loss occurs, a single ON / OFF cycle is sufficiently short, and a change in voltage at the time can be ignored. As a relational expression between voltage and current, the following expression (1) or (2) is established.
Vc · Ic = Va · Ia + Vb · Ib Formula (1)
Or Vc = Va · Ia / Ic + Vb · Ib / Ic (2)
(Ia+Ib=Ic)または(Ia/Ic+Ib/Ic=1)
同様にVa≦Vbとなる位相角の範囲においてはSW1を常時ONとし、SW2をON/OFF制御することにより電流IaとIbの比率を変えることができる。また、結合回路10はスイッチング素子とリアクトルのみで構成されていて、理想的にはエネルギー損失が生じないことから、一回のON/OFF周期が十分短く、当該時間における電圧の変化が無視できる場合、電圧と電流の関係式として次式(1)または(2)が成り立つ。
Vc・Ic=Va・Ia+Vb・Ib … 式(1)
または
Vc=Va・Ia/Ic+Vb・Ib/Ic … 式(2)
Here, as shown in FIG. 18, the switching elements SW1 and SW2 are ON / OFF controlled. An ON / OFF period corresponding to 1.5 ° of the system voltage is set as an ON / OFF cycle, and a predetermined period is turned ON, and the remaining period is turned OFF. In the phase angle range where the output voltage Va ≧ Vb, SW2 is always turned on, and SW1 is turned on / off, thereby passing through the connection point a and flowing through the connection point c, and passing through the connection point b. The ratio of the current Ib flowing through the connection point c can be changed. Here, the currents Ia and Ib are used to mean the average current in one ON / OFF operation, and the relational expression with the current Ic flowing through the connection point c is as follows.
(Ia + Ib = Ic) or (Ia / Ic + Ib / Ic = 1)
Similarly, in the phase angle range where Va ≦ Vb, the ratio of the currents Ia and Ib can be changed by always turning SW1 ON and controlling SW2 ON / OFF. In addition, since the
Vc · Ic = Va · Ia + Vb · Ib Formula (1)
Or Vc = Va · Ia / Ic + Vb · Ib / Ic (2)
従って、スイッチング素子SW1あるいはSW2をON/OFF制御することにより電流IaとIbの比率を変え、VaとVbの中間範囲においてVcの値を任意に制御することができる。ここで図2に示した如く、電流比Ia/Icが、位相角0°で0.5+αであり、単調減少して位相角30°で0.5-αとなり、ここから単調増加して位相角60°で0.5+αとなる動作を繰り返すように、また、Ib/Icが、位相角0°で0.5-αであり、単調増加少して位相角30°で0.5+αとなり、さらに、単調減少して位相角60°で0.5-αとなる動作を繰り返すように、スイッチング素子SW1、SW2をON/OFF制御する。このように、Ia/Ic、Ib/Icを制御すれば、図3に示した如くVcを一定値とすることができ、負荷の手前に直流フィルタを設置することが不要となる。
Therefore, the ratio of the currents Ia and Ib can be changed by ON / OFF control of the switching element SW1 or SW2, and the value of Vc can be arbitrarily controlled in the intermediate range between Va and Vb. Here, as shown in FIG. 2, the current ratio Ia / Ic is 0.5 + α at a phase angle of 0 °, decreases monotonously to 0.5-α at a phase angle of 30 °, and increases monotonically from here. In addition, Ib / Ic is 0.5-α at a phase angle of 0 °, and is slightly increased to 0.5 + α at a phase angle of 30 °, so that the operation of 0.5 + α is repeated at an angle of 60 °. The switching elements SW1 and SW2 are ON / OFF controlled so as to repeat the operation of monotonically decreasing and reaching 0.5-α at a phase angle of 60 °. In this way, if Ia / Ic and Ib / Ic are controlled, Vc can be set to a constant value as shown in FIG. 3, and it is not necessary to install a DC filter before the load.
図4は上記制御を実施したときの変換用変圧器u相の各二次側電流の一次側電流換算値-位相角線図を示したものである。u相の一次側電流はこれら各二次側電流の一次側電流換算値の和となる。これを図5に示す。図5に示された電流波形でも用途によっては十分に正弦波に近い、高調波成分が小さいものと言える。
FIG. 4 shows a primary-side converted current-phase angle diagram of each secondary-side current of the conversion transformer u-phase when the above control is performed. The u-phase primary side current is the sum of the primary side current converted values of these secondary side currents. This is shown in FIG. Even in the current waveform shown in FIG. 5, it can be said that the harmonic component is sufficiently close to a sine wave depending on the application and has a small harmonic component.
FIG. 4 shows a primary-side converted current-phase angle diagram of each secondary-side current of the conversion transformer u-phase when the above control is performed. The u-phase primary side current is the sum of the primary side current converted values of these secondary side currents. This is shown in FIG. Even in the current waveform shown in FIG. 5, it can be said that the harmonic component is sufficiently close to a sine wave depending on the application and has a small harmonic component.
実施例2について図6ないし図10を用いて説明する。図において符号の番号が同じものは実施例1と基本的に同等なものである。以下の実施例では、それまでに説明した実施例と異なる部分を中心に説明するのであって、説明が省略された部分はそれまでに説明された実施例と技術的に異なっていない限りにおいて同じものである。
Example 2 will be described with reference to FIGS. In the figure, the same reference numerals are basically equivalent to those in the first embodiment. In the following embodiments, the description will focus on the differences from the embodiments described so far, and the portions that are not described are the same as long as they are not technically different from the embodiments described so far. Is.
図6は本実施例の整流回路の主回路構成図、図7はその整流回路の各直流電流比率-位相角線図、図8は当該整流回路における各直流電圧の電圧-位相角線図、図9は当該整流回路における変換用変圧器u相の各二次側電流の一次側電流換算値-位相角線図、図5はu相の一次側電流-位相角線図である。
6 is a main circuit configuration diagram of the rectifier circuit of this embodiment, FIG. 7 is a DC current ratio-phase angle diagram of the rectifier circuit, FIG. 8 is a voltage-phase angle diagram of each DC voltage in the rectifier circuit, FIG. 9 is a primary current converted value—phase angle diagram of each secondary current of the conversion transformer u phase in the rectifier circuit, and FIG. 5 is a primary phase current—phase angle diagram of the u phase.
本実施例では、図1に示した実施例1におけるスイッチング電流平滑用コンデンサC1、C2の接続位置が、夫々接続点aと接続点cとの間、接続点bと接続点cとの間に変更されるとともに、接続点cと負荷60の間に並列共振回路40が挿入されたものとなっている。並列共振回路40を構成するリアクトルL41のインダクタンスとコンデンサC41の静電容量は、1次側三相交流の周波数の12倍の周波数で共振する様に選定されている。
In the present embodiment, the connection positions of the switching current smoothing capacitors C1 and C2 in the first embodiment shown in FIG. 1 are between the connection point a and the connection point c, and between the connection point b and the connection point c, respectively. In addition to the change, the parallel resonance circuit 40 is inserted between the connection point c and the load 60. The inductance of the reactor L41 and the electrostatic capacity of the capacitor C41 constituting the parallel resonance circuit 40 are selected so as to resonate at a frequency 12 times the frequency of the primary side three-phase alternating current.
ここで、図7に示した通りに電流比率を制御すれば、図8に示した様にVcの電圧波形として、直流電圧に1次側三相交流の周波数の12倍の周波数の正弦波が重畳されたものを出力することができる。この重畳された正弦波は並列共振回路40により除去され、負荷60には直流電圧のみが印加される。負荷60が純抵抗負荷であれば電流Icは一定値となる。
Here, if the current ratio is controlled as shown in FIG. 7, as shown in FIG. 8, a sine wave having a frequency 12 times the frequency of the primary three-phase alternating current is generated in the DC voltage as the voltage waveform of Vc. The superimposition can be output. The superimposed sine wave is removed by the parallel resonance circuit 40, and only a DC voltage is applied to the load 60. If the load 60 is a pure resistance load, the current Ic has a constant value.
図9は上記制御を実施し0たときの変換用変圧器u相の各二次側電流の一次側電流換算値-位相角線図を示したものである。u相の一次側電流はこれら各二次側電流の一次側電流換算値の和となる。これを図10に示す。図10に示された電流波形は実施例1のものと比べて更なる高調波成分の抑制が実現されている。
FIG. 9 is a primary side current converted value-phase angle diagram of each secondary side current of the conversion transformer u phase when the above control is performed. The primary current of the u phase is the sum of the primary current conversion values of these secondary currents. This is shown in FIG. In the current waveform shown in FIG. 10, further suppression of harmonic components is realized as compared with that of the first embodiment.
また、コンデンサC1、C2を図示の如く配置することにより、実施例1と比べて耐電圧の小さな素子を選定することが可能となっている。
Further, by disposing the capacitors C1 and C2 as shown in the drawing, it is possible to select an element having a smaller withstand voltage compared to the first embodiment.
Further, by disposing the capacitors C1 and C2 as shown in the drawing, it is possible to select an element having a smaller withstand voltage compared to the first embodiment.
実施例3について図11ないし図15を用いて説明する。図11は本実施例の整流回路の主回路構成図、図12はその整流回路の各直流電流比率-位相角線図、図13は当該整流回路における各直流電圧の電圧-位相角線図、図14は当該整流回路における変換用変圧器u相の各二次側電流の一次側電流換算値-位相角線図、図15はu相の一次側電流-位相角線図である。
Example 3 will be described with reference to FIGS. 11 is a main circuit configuration diagram of the rectifier circuit of the present embodiment, FIG. 12 is a DC current ratio-phase angle diagram of the rectifier circuit, FIG. 13 is a voltage-phase angle diagram of each DC voltage in the rectifier circuit, FIG. 14 is a primary current converted value—phase angle diagram of each secondary current of the transformer u phase for conversion in the rectifier circuit, and FIG. 15 is a primary current-phase angle diagram of the u phase.
実施例2が並列共振回路40を1次側三相交流の周波数の12倍の周波数で共振する様に構成していたのに対し、本実施例は、並列共振回路40を1次側三相交流の周波数の12倍の周波数及び24倍の周波数で共振する様に構成している。即ち、並列共振回路40を構成するリアクトルL41のインダクタンスとコンデンサC41の静電容量を、1次側三相交流の周波数の12倍の周波数で共振する様に選定するとともに、リアクトルL42のインダクタンスとコンデンサC42の静電容量を1次側三相交流の周波数の24倍の周波数で共振する様に選定している。
In contrast to the second embodiment in which the parallel resonant circuit 40 is configured to resonate at a frequency 12 times the frequency of the primary three-phase alternating current, the second embodiment configures the parallel resonant circuit 40 to have a primary three-phase frequency. It is configured to resonate at a frequency 12 times and 24 times the frequency of the alternating current. In other words, the inductance of the reactor L41 and the capacitance of the capacitor C41 constituting the parallel resonant circuit 40 are selected so as to resonate at a frequency 12 times the frequency of the primary side three-phase alternating current, and the inductance and the capacitor of the reactor L42. The capacitance of C42 is selected so as to resonate at a frequency 24 times the frequency of the primary side three-phase alternating current.
ここで、図12に示した通りに電流比率を制御すれば、図13に示した様にVcの電圧波形として、直流電圧に1次側三相交流の周波数の12倍の周波数の正弦波と24倍の周波数の正弦波が重畳されたものを出力することができる。この重畳された正弦波は並列共振回路40により除去され、負荷60には直流電圧のみが印加される。負荷60が純抵抗負荷であれば電流Icは一定値となる。
If the current ratio is controlled as shown in FIG. 12, a sine wave having a frequency 12 times the frequency of the primary side three-phase alternating current is added to the DC voltage as shown in FIG. A signal on which a sine wave having a frequency 24 times is superimposed can be output. The superimposed sine wave is removed by the parallel resonance circuit 40, and only a DC voltage is applied to the load 60. If the load 60 is a pure resistance load, the current Ic has a constant value.
図14は上記制御を実施したときの変換用変圧器u相の各二次側電流の一次側電流換算値-位相角線図を示したものである。u相の一次側電流はこれら各二次側電流の一次側電流換算値の和となる。これを図15に示す。図15に示された電流波形は実施例2のものと比べて更なる高調波成分の抑制が実現されている。
FIG. 14 shows a primary current converted value-phase angle diagram of each secondary current of the conversion transformer u-phase when the above control is performed. The primary current of the u phase is the sum of the primary current conversion values of these secondary currents. This is shown in FIG. The current waveform shown in FIG. 15 realizes further suppression of higher harmonic components than that of the second embodiment.
FIG. 14 shows a primary current converted value-phase angle diagram of each secondary current of the conversion transformer u-phase when the above control is performed. The primary current of the u phase is the sum of the primary current conversion values of these secondary currents. This is shown in FIG. The current waveform shown in FIG. 15 realizes further suppression of higher harmonic components than that of the second embodiment.
実施例4について図16を用いて説明する。図16は本実施例の整流回路の主回路構成図である。本実施例では、図1に示した実施例1におけるブリッジ回路のダイオードDI11,DI13、DI15、DI21,DI23、DI25がそれぞれGTO、GT11,GT13、GT15、GT21,GT23、GT25に置き換えられたものとなっている。基本的な回路動作は実施例1と同一である。一方でGTOに置き換えたことにより、異常時において当該GTOをOFFすることで確実に電流を遮断できる等の効果がある。
Example 4 will be described with reference to FIG. FIG. 16 is a main circuit configuration diagram of the rectifier circuit of this embodiment. In the present embodiment, the diodes DI11, DI13, DI15, DI21, DI23, and DI25 of the bridge circuit in the first embodiment shown in FIG. 1 are replaced with GTO, GT11, GT13, GT15, GT21, GT23, and GT25, respectively. It has become. The basic circuit operation is the same as in the first embodiment. On the other hand, by replacing the GTO, there is an effect that the current can be surely interrupted by turning off the GTO at the time of abnormality.
Example 4 will be described with reference to FIG. FIG. 16 is a main circuit configuration diagram of the rectifier circuit of this embodiment. In the present embodiment, the diodes DI11, DI13, DI15, DI21, DI23, and DI25 of the bridge circuit in the first embodiment shown in FIG. 1 are replaced with GTO, GT11, GT13, GT15, GT21, GT23, and GT25, respectively. It has become. The basic circuit operation is the same as in the first embodiment. On the other hand, by replacing the GTO, there is an effect that the current can be surely interrupted by turning off the GTO at the time of abnormality.
10…結合回路、21、22…ブリッジ回路、30…変換用変圧器、40…並列共振回路、SW1,SW2…主スイッチング素子、L0…直流リアクトル、L41,L42…リアクトル、C41,C42…コンデンサ、D11,D12,D13,D14,D15,D16…ダイオード、D21,D22,D23,D24,D25,D26…ダイオード、GT11,GT13,GT15…副スイッチング素子、GT21,GT23,GT25…副スイッチング素子。
DESCRIPTION OF SYMBOLS 10 ... Coupling circuit, 21, 22 ... Bridge circuit, 30 ... Conversion transformer, 40 ... Parallel resonance circuit, SW1, SW2 ... Main switching element, L0 ... DC reactor, L41, L42 ... Reactor, C41, C42 ... Capacitor, D11, D12, D13, D14, D15, D16 ... diode, D21, D22, D23, D24, D25, D26 ... diode, GT11, GT13, GT15 ... sub-switching element, GT21, GT23, GT25 ... sub-switching element.
Claims (7)
- 位相が異なる2組の交流を出力する変換用変圧器と、前記交流を各々直流に変換する2組の整流回路と、前記2組の整流回路の出力を並列結合する結合回路とを有する整流回路において、前記結合回路が第1のスイッチング素子と第2のスイッチング素子と1つの直流リアクトルとを有するとともに、前記2組の整流回路の一方に接続される第1の入力接続点と、前記2組の整流回路の他方に接続される第2の入力接続点と、出力接続点が形成されており、前記第1の入力接続点と前記第1のスイッチング素子の一端が接続され、前記第2の入力接続点と前記第2のスイッチング素子の一端が接続され、夫々のスイッチング素子の他端がともに前記直流リアクトルの一端に接続され、前記直流リアクトルの他端が前記出力接続点に接続され、前記第1のスイッチング素子は前記第1の整流回路の出力する脈流の周期よりも短い周期でスイッチングし、前記第2のスイッチング素子は前記第2の整流回路の出力する脈流の周期よりも短い周期でスイッチングすることを特徴とする整流回路。
A rectifier circuit comprising: a transformer for conversion that outputs two sets of alternating currents having different phases; two sets of rectifier circuits that each convert the alternating current into direct current; and a coupling circuit that couples the outputs of the two sets of rectifier circuits in parallel. And the coupling circuit includes a first switching element, a second switching element, and one DC reactor, and a first input connection point connected to one of the two sets of rectifier circuits, and the two sets A second input connection point connected to the other of the rectifier circuit and an output connection point are formed, and the first input connection point and one end of the first switching element are connected to each other. An input connection point and one end of the second switching element are connected, the other end of each switching element is connected to one end of the DC reactor, and the other end of the DC reactor is connected to the output connection point, The first switching element is switched with a cycle shorter than the cycle of the pulsating current output from the first rectifier circuit, and the second switching element is switched over with the cycle of the pulsating flow output from the second rectifier circuit. A rectifier circuit that switches at a short cycle.
- 請求項1において、前記第1のスイッチング素子と前記第2のスイッチング素子は、一方がオン動作のときに他方がオン/オフ動作となるように動作することを特徴とする整流回路。
2. The rectifier circuit according to claim 1, wherein the first switching element and the second switching element operate so that one of the first switching element and the second switching element is turned on / off when the other is turned on.
- 請求項1または2において、前記出力電の電圧を一定に保つように、前記第1の入力接続点に流れる電流と、前記第2の入力接続点に流れる電流を制御することを特徴とする整流回路。
3. The rectification according to claim 1, wherein a current flowing through the first input connection point and a current flowing through the second input connection point are controlled so as to keep the voltage of the output electricity constant. circuit.
- 請求項1、2または3において、前記変換用変圧器は、二次側の一方がY結線であり、他方がΔ結線であることを特徴とする整流回路。
4. The rectifier circuit according to claim 1, wherein one of the secondary transformers is Y-connected and the other is Δ-connected.
- 請求項1、2、3または4において、前記整流回路は、ブリッジ回路として構成されることを特徴とする整流回路。
5. The rectifier circuit according to claim 1, wherein the rectifier circuit is configured as a bridge circuit.
- 請求項1、2、3、4または5に記載の整流回路であって、前記出力接続点に並列共振回路が接続されていることを特徴とする整流回路。
6. The rectifier circuit according to claim 1, wherein a parallel resonant circuit is connected to the output connection point.
- 請求項6に記載の整流回路であって、前記出力接続点と前記並列共振回路の間にさらに並列共振回路を設け、これらの並列共振回路は互いに異なる周波数で共振するように構成されていることを特徴とする整流回路。 7. The rectifier circuit according to claim 6, wherein a parallel resonant circuit is further provided between the output connection point and the parallel resonant circuit, and these parallel resonant circuits are configured to resonate at mutually different frequencies. A rectifier circuit characterized by.
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JP2017153353A (en) * | 2016-02-22 | 2017-08-31 | ドクター エンジニール ハー ツェー エフ ポルシェ アクチエンゲゼルシャフトDr. Ing. h.c. F. Porsche Aktiengesellschaft | Method and apparatus for controlling charge station |
WO2018119528A1 (en) * | 2016-12-30 | 2018-07-05 | Universidad De Santiago De Chile | System for transmitting electrical energy using direct current |
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JP2017153353A (en) * | 2016-02-22 | 2017-08-31 | ドクター エンジニール ハー ツェー エフ ポルシェ アクチエンゲゼルシャフトDr. Ing. h.c. F. Porsche Aktiengesellschaft | Method and apparatus for controlling charge station |
WO2018119528A1 (en) * | 2016-12-30 | 2018-07-05 | Universidad De Santiago De Chile | System for transmitting electrical energy using direct current |
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