WO2015145713A1 - Rectifier circuit - Google Patents

Rectifier circuit Download PDF

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Publication number
WO2015145713A1
WO2015145713A1 PCT/JP2014/059073 JP2014059073W WO2015145713A1 WO 2015145713 A1 WO2015145713 A1 WO 2015145713A1 JP 2014059073 W JP2014059073 W JP 2014059073W WO 2015145713 A1 WO2015145713 A1 WO 2015145713A1
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Prior art keywords
rectifier circuit
connection point
switching element
current
circuit
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PCT/JP2014/059073
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French (fr)
Japanese (ja)
Inventor
市村　智
加藤　修治
井上　重徳
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株式会社日立製作所
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Priority to PCT/JP2014/059073 priority Critical patent/WO2015145713A1/en
Publication of WO2015145713A1 publication Critical patent/WO2015145713A1/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/23—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only arranged for operation in parallel
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/06—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
    • H02M7/08—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode arranged for operation in parallel

Definitions

  • the present invention relates to a rectifier circuit that converts alternating current into direct current, and more particularly to a rectifier circuit that is suitable for suppressing harmonic components contained in an alternating current input current waveform.
  • each three-phase bridge rectifier circuit is connected by an interphase reactor, and the interphase reactor is configured to supply a rectified current to a load via a smoothing reactor.
  • the interphase reactor is provided with a plurality of taps symmetrically on the left and right, centering on the midpoint thereof, and connected via the respective taps, or a secondary winding is provided on the interphase reactor. It is reported that the midpoint of the secondary winding and the midpoint of the interphase reactor are combined, and a plurality of taps are provided symmetrically about the midpoint of the secondary winding and connected via the taps. Has been. That is, one is selected from those connected to a plurality of positions on the interphase reactor at a predetermined time, and the coupling current is taken out. Such a technique is described in Japanese Patent Laid-Open No. 59-213282.
  • an object of the present invention is to provide a rectifier circuit that improves the suppression effect of harmonic components without increasing the number of switching elements and eliminates the need for an interphase reactor.
  • the present invention provides a conversion transformer that outputs two sets of alternating currents having different phases, two sets of rectifier circuits that convert the alternating current into direct current, and the two sets of rectifier circuits.
  • a coupling circuit that couples the outputs of the first and second outputs in parallel, the coupling circuit includes a first switching element, a second switching element, and one DC reactor, and is connected to one of the two sets of rectification circuits.
  • the first input connection point, the second input connection point connected to the other of the two sets of rectifier circuits, and the output connection point are formed, and the first input connection point and the first input connection point are formed.
  • each switching element One end of each switching element is connected, one end of the second switching element is connected to one end of the second input connection point, and the other end of each switching element is connected to one end of the DC reactor.
  • the first switching element is switched at a cycle shorter than the cycle of the pulsating current output from the first rectifier circuit, and the second switching element is the first switching element. The switching is performed at a cycle shorter than the cycle of the pulsating current output from the rectifier circuit 2.
  • a transformer for conversion that outputs two sets of three-phase alternating currents having different phases, two sets of bridge circuits that respectively convert the three-phase alternating currents into direct current, and a coupling that couples the two sets of direct current outputs in parallel
  • the coupling circuit has two switching elements and one DC reactor, and one end of each switching element is connected to each of the DC output ends of the two sets of bridge circuits, and the other end Are connected to one end of the DC reactor.
  • a parallel resonance circuit is connected to the other end of the DC reactor.
  • the effect of suppressing harmonic components can be improved without increasing the number of switching elements, and an interphase reactor can be made unnecessary.
  • FIG. 1 is a main circuit configuration diagram according to Embodiment 1.
  • FIG. FIG. 3 is a diagram illustrating each DC current ratio-phase angle diagram according to the first embodiment.
  • FIG. 3 is a voltage-phase angle diagram of each DC voltage according to the first embodiment.
  • FIG. 6 is a primary side converted current-phase angle diagram of each secondary side current according to the first embodiment.
  • 1 is a primary current-phase angle diagram according to Embodiment 1.
  • FIG. FIG. 6 is a main circuit configuration diagram according to the second embodiment.
  • FIG. 6 is a diagram illustrating each DC current ratio-phase angle diagram according to the second embodiment.
  • FIG. 6 is a -phase angle diagram of each DC voltage according to the second embodiment.
  • FIG. 6 is a primary current converted value-phase angle diagram of each secondary current according to the second embodiment.
  • FIG. 10 is a primary side current-phase angle diagram according to the second embodiment.
  • FIG. 6 is a main circuit configuration diagram according to the third embodiment.
  • FIG. 10 is a -phase angle diagram of each DC voltage according to the third embodiment.
  • FIG. 12 is a primary side current converted value-phase angle diagram of each secondary side current according to the third embodiment.
  • FIG. 10 is a primary current-phase angle diagram according to the third embodiment.
  • FIG. 6 is a main circuit configuration diagram according to the fourth embodiment. The voltage-phase angle diagram of each voltage which concerns on an Example. Switching element operation-phase angle diagram according to the embodiment.
  • FIG. 1 is a main circuit configuration diagram of the rectifier circuit of this embodiment
  • FIG. 2 is a DC current ratio-phase angle diagram of the rectifier circuit
  • FIG. 3 is a voltage-phase angle diagram of each DC voltage in the rectifier circuit
  • FIG. 4 is a primary current converted value—phase angle diagram of each secondary current of the conversion transformer u phase in the rectifier circuit
  • FIG. 5 is a primary phase current—phase angle diagram of the u phase.
  • the rectifier circuit includes a primary winding 5 (Y connection, number of turns N0) to which a three-phase AC voltage is input from the outside, and a three-phase circuit having the same phase as the primary three-phase AC voltage.
  • Secondary winding 6 (Y connection, number of turns N1) for outputting AC voltage
  • secondary winding 7 ( ⁇ connection for outputting three-phase AC voltage that is 30 ° out of phase with the primary three-phase AC voltage) ,
  • the number of turns N2 The number of turns N2).
  • the primary winding 5 (Y connection, number of turns N0) is a Y connection of the windings u0, v0, w0, and the currents flowing through the windings u0, v0, w0 are defined as Iu0, Iv0, Iw0. .
  • the primary winding 5 (Y connection, number of turns N0) has system input force terminals u, v, and w, and a system voltage is applied to the system input force terminals u, v, and w. For example, a system voltage of 50 Hz or 60 Hz is applied.
  • the secondary winding 6 (Y connection, number of turns N1) is Y connection of the windings u1, v1, and w1, and the currents flowing through the respective windings u1, v1, and w1 are defined as Iu1, Iv1, and Iw1.
  • the secondary winding 7 ( ⁇ connection, number of turns N2) is obtained by ⁇ connection of the windings u2, v2, and w2, and the currents flowing in the respective windings u2, v2, and w2 are defined as Iu2, Iv2, and Iw2. .
  • the bridge circuit 21 includes a diode DI11 and a diode DI12 connected in series, a diode DI13 and a diode DI14 connected in series, and a diode DI15 and a diode DI16 connected in series. It is.
  • a connection point between the diode DI11 and the diode DI12, a connection point between the diode DI13 and the diode DI14, and a connection point between the diode DI15 and the diode DI16 are connected to the windings u1, v1, and w1, respectively.
  • One of the parallel connection points is connected to the input connection point a of the coupling circuit 10, and the other of the parallel connection points is connected to the output connection point d to the load 60.
  • a capacitor C1 is connected between the parallel connection points.
  • the voltage between the parallel connection points is defined as Va.
  • the current to the input connection point a of the coupling circuit 10 is defined as Ia.
  • the bridge circuit 22 includes a diode DI21 and a diode DI22 connected in series, a diode DI23 and a diode DI24 connected in series, and a diode DI25 and a diode DI26 connected in series. It is.
  • the connection point of the diode DI21 and the diode DI22, the connection point of the diode DI23 and the diode DI24, the connection point of the diode DI25 and the diode DI26 are respectively the connection point of the windings w2 and u2, the connection point of the windings u2 and v2, and the winding point. Connected to the connection point of v2 and w2.
  • One of the parallel connection points is connected to the input connection point b of the coupling circuit 10, and the other of the parallel connection points is connected to the output connection point d to the load 60.
  • a capacitor C2 is connected between the parallel connection points.
  • the voltage between the parallel connection points is defined as Vb.
  • the current to the input connection point b of the coupling circuit 10 is defined as Ia.
  • the direct current outputs are coupled in parallel by the coupling circuit 10.
  • This coupling circuit 10 has two switching elements SW1 and SW2 and one DC reactor L0, and one ends of the switching elements SW1 and SW2 are connected to the DC output ends of the bridge circuits 21 and 22 and connection points a and b, respectively. The other end is connected to one end of the DC reactor.
  • a load 60 is connected to the other end of the DC reactor at a connection point c.
  • the switching element for example, a well-known element such as an IGBT and a reverse blocking type element can be used. When using a switching element that is not a reverse blocking type, a diode may be added in series.
  • a sine wave is considered as a three-phase AC voltage waveform input from the outside.
  • a system voltage for example, a system voltage of 50 Hz or 60 Hz
  • the system input force terminals u, v, w are externally applied.
  • Vu, Vv, and Vw are defined as Vu, Vv, and Vw.
  • the horizontal axis (phase angle) in FIG. 17 is shown as the phase angle of the system voltage applied to the system input force terminals u, v, and w. That is, for example, one cycle of the system voltage given at 50 Hz or 60 Hz is indicated as 360 ° (deg).
  • the reference for the phase angle is the same in the description and drawings described later.
  • the u-phase voltage Vu is used as a reference, and the v-phase voltage Vv is 120 ° and the w-phase voltage Vw is 240 °, which are out of phase.
  • the output voltages Va and Vb from the bridge circuits 21 and 22 are as shown in FIG. That is, it has the same waveform shape including an AC fluctuation component whose basic frequency is 6 times the input AC frequency, and becomes a voltage waveform whose phase is shifted by 30 ° from each other (corresponding to 60 ° of the system voltage is one cycle. ).
  • the output voltages Va and Vb are interchanged with each other every 30 °.
  • the switching elements SW1 and SW2 are ON / OFF controlled.
  • An ON / OFF period corresponding to 1.5 ° of the system voltage is set as an ON / OFF cycle, and a predetermined period is turned ON, and the remaining period is turned OFF.
  • SW2 is always turned on, and SW1 is turned on / off, thereby passing through the connection point a and flowing through the connection point c, and passing through the connection point b.
  • the ratio of the current Ib flowing through the connection point c can be changed.
  • Vc ⁇ Ic Va ⁇ Ia + Vb ⁇ Ib Formula (1)
  • Vc Va ⁇ Ia / Ic + Vb ⁇ Ib / Ic (2)
  • the ratio of the currents Ia and Ib can be changed by ON / OFF control of the switching element SW1 or SW2, and the value of Vc can be arbitrarily controlled in the intermediate range between Va and Vb.
  • the current ratio Ia / Ic is 0.5 + ⁇ at a phase angle of 0 °, decreases monotonously to 0.5- ⁇ at a phase angle of 30 °, and increases monotonically from here.
  • Ib / Ic is 0.5- ⁇ at a phase angle of 0 °, and is slightly increased to 0.5 + ⁇ at a phase angle of 30 °, so that the operation of 0.5 + ⁇ is repeated at an angle of 60 °.
  • the switching elements SW1 and SW2 are ON / OFF controlled so as to repeat the operation of monotonically decreasing and reaching 0.5- ⁇ at a phase angle of 60 °. In this way, if Ia / Ic and Ib / Ic are controlled, Vc can be set to a constant value as shown in FIG. 3, and it is not necessary to install a DC filter before the load.
  • FIG. 4 shows a primary-side converted current-phase angle diagram of each secondary-side current of the conversion transformer u-phase when the above control is performed.
  • the u-phase primary side current is the sum of the primary side current converted values of these secondary side currents. This is shown in FIG. Even in the current waveform shown in FIG. 5, it can be said that the harmonic component is sufficiently close to a sine wave depending on the application and has a small harmonic component.
  • Example 2 will be described with reference to FIGS. In the figure, the same reference numerals are basically equivalent to those in the first embodiment. In the following embodiments, the description will focus on the differences from the embodiments described so far, and the portions that are not described are the same as long as they are not technically different from the embodiments described so far. Is.
  • FIG. 6 is a main circuit configuration diagram of the rectifier circuit of this embodiment
  • FIG. 7 is a DC current ratio-phase angle diagram of the rectifier circuit
  • FIG. 8 is a voltage-phase angle diagram of each DC voltage in the rectifier circuit
  • FIG. 9 is a primary current converted value—phase angle diagram of each secondary current of the conversion transformer u phase in the rectifier circuit
  • FIG. 5 is a primary phase current—phase angle diagram of the u phase.
  • connection positions of the switching current smoothing capacitors C1 and C2 in the first embodiment shown in FIG. 1 are between the connection point a and the connection point c, and between the connection point b and the connection point c, respectively.
  • the parallel resonance circuit 40 is inserted between the connection point c and the load 60.
  • the inductance of the reactor L41 and the electrostatic capacity of the capacitor C41 constituting the parallel resonance circuit 40 are selected so as to resonate at a frequency 12 times the frequency of the primary side three-phase alternating current.
  • a sine wave having a frequency 12 times the frequency of the primary three-phase alternating current is generated in the DC voltage as the voltage waveform of Vc.
  • the superimposition can be output.
  • the superimposed sine wave is removed by the parallel resonance circuit 40, and only a DC voltage is applied to the load 60. If the load 60 is a pure resistance load, the current Ic has a constant value.
  • FIG. 9 is a primary side current converted value-phase angle diagram of each secondary side current of the conversion transformer u phase when the above control is performed.
  • the primary current of the u phase is the sum of the primary current conversion values of these secondary currents. This is shown in FIG. In the current waveform shown in FIG. 10, further suppression of harmonic components is realized as compared with that of the first embodiment.
  • Example 3 will be described with reference to FIGS. 11 is a main circuit configuration diagram of the rectifier circuit of the present embodiment
  • FIG. 12 is a DC current ratio-phase angle diagram of the rectifier circuit
  • FIG. 13 is a voltage-phase angle diagram of each DC voltage in the rectifier circuit
  • FIG. 14 is a primary current converted value—phase angle diagram of each secondary current of the transformer u phase for conversion in the rectifier circuit
  • FIG. 15 is a primary current-phase angle diagram of the u phase.
  • the second embodiment configures the parallel resonant circuit 40 to have a primary three-phase frequency. It is configured to resonate at a frequency 12 times and 24 times the frequency of the alternating current.
  • the inductance of the reactor L41 and the capacitance of the capacitor C41 constituting the parallel resonant circuit 40 are selected so as to resonate at a frequency 12 times the frequency of the primary side three-phase alternating current, and the inductance and the capacitor of the reactor L42.
  • the capacitance of C42 is selected so as to resonate at a frequency 24 times the frequency of the primary side three-phase alternating current.
  • a sine wave having a frequency 12 times the frequency of the primary side three-phase alternating current is added to the DC voltage as shown in FIG.
  • a signal on which a sine wave having a frequency 24 times is superimposed can be output.
  • the superimposed sine wave is removed by the parallel resonance circuit 40, and only a DC voltage is applied to the load 60. If the load 60 is a pure resistance load, the current Ic has a constant value.
  • FIG. 14 shows a primary current converted value-phase angle diagram of each secondary current of the conversion transformer u-phase when the above control is performed.
  • the primary current of the u phase is the sum of the primary current conversion values of these secondary currents. This is shown in FIG.
  • the current waveform shown in FIG. 15 realizes further suppression of higher harmonic components than that of the second embodiment.
  • FIG. 16 is a main circuit configuration diagram of the rectifier circuit of this embodiment.
  • the diodes DI11, DI13, DI15, DI21, DI23, and DI25 of the bridge circuit in the first embodiment shown in FIG. 1 are replaced with GTO, GT11, GT13, GT15, GT21, GT23, and GT25, respectively. It has become.
  • the basic circuit operation is the same as in the first embodiment.
  • by replacing the GTO there is an effect that the current can be surely interrupted by turning off the GTO at the time of abnormality.

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Abstract

Provided is a rectifier circuit in which the effect of suppressing harmonic components is improved without increasing the number of switching elements and no interphase reactor is required. The rectifier circuit has: a conversion transformer that outputs two sets of three-phase alternating currents having different phases; two sets of bridge circuits respectively converting the three-phase alternating currents to direct currents; and a coupling circuit coupling the two sets of direct current outputs in parallel. The coupling circuit has two switching elements and a DC reactor, wherein one ends of the switching elements are respectively connected to DC output terminals of the two sets of bridge circuits and the other ends are both connected to one end of the DC reactor.

Description

整流回路Rectifier circuit
　本発明は、交流を直流に変換する整流回路に係り、特に交流入力電流波形に含まれる高調波成分を抑制するのに好適な整流回路に関する。  
The present invention relates to a rectifier circuit that converts alternating current into direct current, and more particularly to a rectifier circuit that is suitable for suppressing harmonic components contained in an alternating current input current waveform.
　三相交流を直流に変換する整流回路として、位相が異なる２組の交流を出力する変換用変圧器を利用した整流回路が知られている。この整流回路では、例えば、ブリッジ整流器を２組用意し、一方のブリツジ整流回路を変換用変圧器の一方の相の出力に接続し、他方のブリツジ整流回路を変換用変圧器の他方の相の出力に接続し、さらに、各々の３相ブリッジ整流回路を相間リアクトルで接続し、この相間リアクトルについて平滑用のリアクトルを介して負荷に整流した電流を供給するように構成されるものである。 As a rectifier circuit that converts three-phase alternating current into direct current, a rectifier circuit that uses a conversion transformer that outputs two sets of alternating currents with different phases is known. In this rectifier circuit, for example, two sets of bridge rectifiers are prepared, one bridge rectifier circuit is connected to the output of one phase of the conversion transformer, and the other bridge rectifier circuit is connected to the other phase of the conversion transformer. Connected to the output, each three-phase bridge rectifier circuit is connected by an interphase reactor, and the interphase reactor is configured to supply a rectified current to a load via a smoothing reactor.
　このように構成された整流回路において、交流入力電流波形に含まれる高調波成分を抑制する回路構成手法が報告されている。例えば、相間リアクトルに、その中点を中心にして、左右に対称的に複数個のタップを設け、その各タップを介して接続する、或いは、相間リアクトルに２次巻線を設けて、この２次巻線の中点と相間リアクトルの中点とを結合し、２次巻線の中点を中心にして左右に対称的に複数個のタップを設け、そのタップを介して接続するものが報告されている。すなわち、所定時刻において当該相間リアクトル上の複数の位置に接続されたもののうちから１つを選んで結合電流を取出すものである。このような技術は、特開昭５９－２１３２８２号公報に記載されている。
In the rectifier circuit configured as described above, a circuit configuration technique for suppressing harmonic components included in the AC input current waveform has been reported. For example, the interphase reactor is provided with a plurality of taps symmetrically on the left and right, centering on the midpoint thereof, and connected via the respective taps, or a secondary winding is provided on the interphase reactor. It is reported that the midpoint of the secondary winding and the midpoint of the interphase reactor are combined, and a plurality of taps are provided symmetrically about the midpoint of the secondary winding and connected via the taps. Has been. That is, one is selected from those connected to a plurality of positions on the interphase reactor at a predetermined time, and the coupling current is taken out. Such a technique is described in Japanese Patent Laid-Open No. 59-213282.
特開昭５９－２１３２８２号公報JP 59-213282 A
　上記の従来技術では、交流入力電流波形に含まれる高調波成分の抑制効果を向上するためには、相間リアクトルのタップの数を多くする必要があるので、それに伴い、整流回路の素子数を増やす必要があった。また、相間リアクトルの両端には入力交流の周波数よりも高い周波数を基本周波数とする交流電圧が加わることになるが、この交流電圧により生じる交流電流を抑制するためには相間リアクトルのインダクタンスを大きくとる必要があった。 In the above prior art, in order to improve the suppression effect of the harmonic component included in the AC input current waveform, it is necessary to increase the number of taps of the interphase reactor, and accordingly, the number of elements of the rectifier circuit is increased. There was a need. In addition, an AC voltage having a fundamental frequency higher than the input AC frequency is applied to both ends of the interphase reactor. In order to suppress the AC current generated by the AC voltage, the inductance of the interphase reactor is increased. There was a need.
　そこで本発明の目的は、スイッチング素子の数を増やすこと無く高調波成分の抑制効果を向上するとともに、相間リアクトルを不要とした整流回路を提供することにある。
Accordingly, an object of the present invention is to provide a rectifier circuit that improves the suppression effect of harmonic components without increasing the number of switching elements and eliminates the need for an interphase reactor.
　上記の課題を解決するために、本発明は、位相が異なる２組の交流を出力する変換用変圧器と、前記交流を各々直流に変換する２組の整流回路と、前記２組の整流回路の出力を並列結合する結合回路とを有する整流回路において、前記結合回路が第１のスイッチング素子と第２のスイッチング素子と１つの直流リアクトルとを有するとともに、前記２組の整流回路の一方に接続される第１の入力接続点と、前記２組の整流回路の他方に接続される第２の入力接続点と、出力接続点が形成されており、前記第１の入力接続点と前記第１のスイッチング素子の一端が接続され、前記第２の入力接続点と前記第２のスイッチング素子の一端が接続され、夫々のスイッチング素子の他端がともに前記直流リアクトルの一端に接続され、前記直流リアクトルの他端が前記出力接続点に接続され、前記第１のスイッチング素子は前記第１の整流回路の出力する脈流の周期よりも短い周期でスイッチングし、前記第２のスイッチング素子は前記第２の整流回路の出力する脈流の周期よりも短い周期でスイッチングするように構成した。 In order to solve the above-described problems, the present invention provides a conversion transformer that outputs two sets of alternating currents having different phases, two sets of rectifier circuits that convert the alternating current into direct current, and the two sets of rectifier circuits. And a coupling circuit that couples the outputs of the first and second outputs in parallel, the coupling circuit includes a first switching element, a second switching element, and one DC reactor, and is connected to one of the two sets of rectification circuits. The first input connection point, the second input connection point connected to the other of the two sets of rectifier circuits, and the output connection point are formed, and the first input connection point and the first input connection point are formed. One end of each switching element is connected, one end of the second switching element is connected to one end of the second input connection point, and the other end of each switching element is connected to one end of the DC reactor. And the first switching element is switched at a cycle shorter than the cycle of the pulsating current output from the first rectifier circuit, and the second switching element is the first switching element. The switching is performed at a cycle shorter than the cycle of the pulsating current output from the rectifier circuit 2.
　具体的には、位相が異なる２組の三相交流を出力する変換用変圧器と、前記三相交流を各々直流に変換する２組のブリッジ回路と、２組の直流出力を並列結合する結合回路とを有しており、前記結合回路が、２つのスイッチング素子と１つの直流リアクトルとを有し、各スイッチング素子の一端が前記２組のブリッジ回路の直流出力端に各々接続され、他端が共に直流リアクトルの一端に接続される様に構成した。また、前記直流リアクトルの他端に並列共振回路が接続されるように構成した。
Specifically, a transformer for conversion that outputs two sets of three-phase alternating currents having different phases, two sets of bridge circuits that respectively convert the three-phase alternating currents into direct current, and a coupling that couples the two sets of direct current outputs in parallel The coupling circuit has two switching elements and one DC reactor, and one end of each switching element is connected to each of the DC output ends of the two sets of bridge circuits, and the other end Are connected to one end of the DC reactor. In addition, a parallel resonance circuit is connected to the other end of the DC reactor.
　本発明によれば、整流回路において、スイッチング素子の数を増やすこと無く高調波成分の抑制効果を向上するとともに、相間リアクトルを不要とすることができる。
According to the present invention, in the rectifier circuit, the effect of suppressing harmonic components can be improved without increasing the number of switching elements, and an interphase reactor can be made unnecessary.
実施例１に係る主回路構成図。1 is a main circuit configuration diagram according to Embodiment 1. FIG. 実施例１に係る各直流電流比率－位相角線図。FIG. 3 is a diagram illustrating each DC current ratio-phase angle diagram according to the first embodiment. 実施例１に係る各直流電圧の電圧－位相角線図。FIG. 3 is a voltage-phase angle diagram of each DC voltage according to the first embodiment. 実施例１に係る各二次側電流の一次側電流換算値－位相角線図。FIG. 6 is a primary side converted current-phase angle diagram of each secondary side current according to the first embodiment. 実施例１に係る一次側電流－位相角線図。1 is a primary current-phase angle diagram according to Embodiment 1. FIG. 実施例２に係る主回路構成図。FIG. 6 is a main circuit configuration diagram according to the second embodiment. 実施例２に係る各直流電流比率－位相角線図。FIG. 6 is a diagram illustrating each DC current ratio-phase angle diagram according to the second embodiment. 実施例２に係る各直流電圧の－位相角線図。FIG. 6 is a -phase angle diagram of each DC voltage according to the second embodiment. 実施例２に係る各二次側電流の一次側電流換算値－位相角線図。FIG. 6 is a primary current converted value-phase angle diagram of each secondary current according to the second embodiment. 実施例２に係る一次側電流－位相角線図。FIG. 10 is a primary side current-phase angle diagram according to the second embodiment. 実施例３に係る主回路構成図。FIG. 6 is a main circuit configuration diagram according to the third embodiment. 実施例３に係る各直流電流比率－位相角線図。Each DC current ratio-phase angle diagram according to Embodiment 3. FIG. 実施例３に係る各直流電圧の－位相角線図。FIG. 10 is a -phase angle diagram of each DC voltage according to the third embodiment. 実施例３に係る各二次側電流の一次側電流換算値－位相角線図。FIG. 12 is a primary side current converted value-phase angle diagram of each secondary side current according to the third embodiment. 実施例３に係る一次側電流－位相角線図。FIG. 10 is a primary current-phase angle diagram according to the third embodiment. 実施例４に係る主回路構成図。FIG. 6 is a main circuit configuration diagram according to the fourth embodiment. 実施例に係る各電圧の電圧－位相角線図。The voltage-phase angle diagram of each voltage which concerns on an Example. 実施例に係るスイッチング素子動作－位相角線図。Switching element operation-phase angle diagram according to the embodiment.
　以下、実施例を図面を用いて説明する。なお、本発明の本質を明らかにするため、以下の実施例では回路における各スイッチング素子、ダイオードにおける電圧低下や配線の抵抗、インダクタンス、寄生容量、変圧器の励磁インダクタンス等が無視できる理想的な状態であるものとして説明する。 Hereinafter, examples will be described with reference to the drawings. In order to clarify the essence of the present invention, in the following embodiments, each switching element in the circuit, the voltage drop in the diode, the resistance of the wiring, the inductance, the parasitic capacitance, the exciting inductance of the transformer, etc. can be ignored. It is assumed that
　実施例１について図１ないし図５、図１７ないし図１８を用いて説明する。図１は本実施例の整流回路の主回路構成図、図２はその整流回路の各直流電流比率－位相角線図、図３は当該整流回路における各直流電圧の電圧－位相角線図、図４は当該整流回路における変換用変圧器u相の各二次側電流の一次側電流換算値－位相角線図、図５はu相の一次側電流－位相角線図である。 Example 1 will be described with reference to FIGS. 1 to 5 and FIGS. 17 to 18. FIG. 1 is a main circuit configuration diagram of the rectifier circuit of this embodiment, FIG. 2 is a DC current ratio-phase angle diagram of the rectifier circuit, FIG. 3 is a voltage-phase angle diagram of each DC voltage in the rectifier circuit, FIG. 4 is a primary current converted value—phase angle diagram of each secondary current of the conversion transformer u phase in the rectifier circuit, and FIG. 5 is a primary phase current—phase angle diagram of the u phase.
　本実施例の整流回路は、図１において、外部から三相交流電圧が入力される１次側巻線５(Ｙ結線、巻数Ｎ０)と、１次側三相交流電圧と同じ位相の三相交流電圧を出力する２次側巻線６(Ｙ結線、巻数Ｎ１)と、１次側三相交流電圧と位相が３０°ずれた三相交流電圧を出力する２次側巻線７(Δ結線、巻数Ｎ２)とを有する変換用変圧器３０を有している。１次側巻線５(Ｙ結線、巻数Ｎ０)は、巻線ｕ０、ｖ０、ｗ０をＹ結線したもので、各々の巻線ｕ０、ｖ０、ｗ０に流れる電流をＩｕ０、Ｉｖ０、Ｉｗ０と定義する。１次側巻線５(Ｙ結線、巻数Ｎ０)は、系統入流力端子ｕ、ｖ、ｗを有し、この系統入流力端子ｕ、ｖ、ｗには、系統電圧が印加される。例えば、５０Ｈｚあるいは６０Ｈｚの系統電圧が印加される。２次側巻線６(Ｙ結線、巻数Ｎ１)は、巻線ｕ１、ｖ１、ｗ１をＹ結線したもので、各々の巻線ｕ１、ｖ１、ｗ１に流れる電流をＩｕ１、Ｉｖ１、Ｉｗ１と定義する。２次側巻線７(Δ結線、巻数Ｎ２)は、巻線ｕ２、ｖ２、ｗ２をΔ結線したもので、各々の巻線ｕ２、ｖ２、ｗ２に流れる電流をＩｕ２、Ｉｖ２、Ｉｗ２と定義する。 In FIG. 1, the rectifier circuit according to this embodiment includes a primary winding 5 (Y connection, number of turns N0) to which a three-phase AC voltage is input from the outside, and a three-phase circuit having the same phase as the primary three-phase AC voltage. Secondary winding 6 (Y connection, number of turns N1) for outputting AC voltage, and secondary winding 7 (Δ connection for outputting three-phase AC voltage that is 30 ° out of phase with the primary three-phase AC voltage) , The number of turns N2). The primary winding 5 (Y connection, number of turns N0) is a Y connection of the windings u0, v0, w0, and the currents flowing through the windings u0, v0, w0 are defined as Iu0, Iv0, Iw0. . The primary winding 5 (Y connection, number of turns N0) has system input force terminals u, v, and w, and a system voltage is applied to the system input force terminals u, v, and w. For example, a system voltage of 50 Hz or 60 Hz is applied. The secondary winding 6 (Y connection, number of turns N1) is Y connection of the windings u1, v1, and w1, and the currents flowing through the respective windings u1, v1, and w1 are defined as Iu1, Iv1, and Iw1. . The secondary winding 7 (Δ connection, number of turns N2) is obtained by Δ connection of the windings u2, v2, and w2, and the currents flowing in the respective windings u2, v2, and w2 are defined as Iu2, Iv2, and Iw2. .
　ブリッジ回路２１は、ダイオードＤＩ１１とダイオードＤＩ１２を直列に接続したものと、ダイオードＤＩ１３とダイオードＤＩ１４を直列に接続したものと、ダイオードＤＩ１５とダイオードＤＩ１６を直列に接続したものとを、並列に接続したものである。ダイオードＤＩ１１とダイオードＤＩ１２の接続点、ダイオードＤＩ１３とダイオードＤＩ１４の接続点、ダイオードＤＩ１５とダイオードＤＩ１６の接続点は、各々、巻線ｕ１、ｖ１、ｗ１に接続される。並列接続点の一方は、結合回路１０の入力接続点ａに接続され、並列接続点の他方は、負荷６０への出力接続点ｄに接続される。並列接続点間にはコンデンサＣ１が接続される。並列接続点間の電圧はＶａと定義される。結合回路１０の入力接続点ａへの電流はＩａと定義される。 The bridge circuit 21 includes a diode DI11 and a diode DI12 connected in series, a diode DI13 and a diode DI14 connected in series, and a diode DI15 and a diode DI16 connected in series. It is. A connection point between the diode DI11 and the diode DI12, a connection point between the diode DI13 and the diode DI14, and a connection point between the diode DI15 and the diode DI16 are connected to the windings u1, v1, and w1, respectively. One of the parallel connection points is connected to the input connection point a of the coupling circuit 10, and the other of the parallel connection points is connected to the output connection point d to the load 60. A capacitor C1 is connected between the parallel connection points. The voltage between the parallel connection points is defined as Va. The current to the input connection point a of the coupling circuit 10 is defined as Ia.
　ブリッジ回路２２は、ダイオードＤＩ２１とダイオードＤＩ２２を直列に接続したものと、ダイオードＤＩ２３とダイオードＤＩ２４を直列に接続したものと、ダイオードＤＩ２５とダイオードＤＩ２６を直列に接続したものとを、並列に接続したものである。ダイオードＤＩ２１とダイオードＤＩ２２の接続点、ダイオードＤＩ２３とダイオードＤＩ２４の接続点、ダイオードＤＩ２５とダイオードＤＩ２６の接続点は、各々、巻線ｗ２とｕ２の接続点、巻線ｕ２とｖ２の接続点、巻線ｖ２とｗ２の接続点に接続される。並列接続点の一方は、結合回路１０の入力接続点ｂに接続され、並列接続点の他方は、負荷６０への出力接続点ｄに接続される。並列接続点間にはコンデンサＣ２が接続される。並列接続点間の電圧はＶｂと定義される。結合回路１０の入力接続点ｂへの電流はＩａと定義される。 The bridge circuit 22 includes a diode DI21 and a diode DI22 connected in series, a diode DI23 and a diode DI24 connected in series, and a diode DI25 and a diode DI26 connected in series. It is. The connection point of the diode DI21 and the diode DI22, the connection point of the diode DI23 and the diode DI24, the connection point of the diode DI25 and the diode DI26 are respectively the connection point of the windings w2 and u2, the connection point of the windings u2 and v2, and the winding point. Connected to the connection point of v2 and w2. One of the parallel connection points is connected to the input connection point b of the coupling circuit 10, and the other of the parallel connection points is connected to the output connection point d to the load 60. A capacitor C2 is connected between the parallel connection points. The voltage between the parallel connection points is defined as Vb. The current to the input connection point b of the coupling circuit 10 is defined as Ia.
　各々の三相交流を直流に変換するブリッジ回路２１、２２は、結合回路１０により各々の直流出力が並列結合される。この結合回路１０は、２つのスイッチング素子ＳＷ１、ＳＷ２と１つの直流リアクトルＬ０とを有し、スイッチング素子ＳＷ１、ＳＷ２の一端が前記ブリッジ回路２１、２２の直流出力端と接続点ａ，ｂで各々接続され、他端が共に直流リアクトルの一端に接続されている。そして前記直流リアクトルの他端に接続点ｃで負荷６０が接続されている。なお、スイッチング素子は、例えば、ＩＧＢＴ等の周知の素子であって逆阻止型のものが使用できる。逆阻止型でないスイッチング素子を使用する場合は各々直列にダイオードを追加すれば良い。 In the bridge circuits 21 and 22 for converting each three-phase alternating current into direct current, the direct current outputs are coupled in parallel by the coupling circuit 10. This coupling circuit 10 has two switching elements SW1 and SW2 and one DC reactor L0, and one ends of the switching elements SW1 and SW2 are connected to the DC output ends of the bridge circuits 21 and 22 and connection points a and b, respectively. The other end is connected to one end of the DC reactor. A load 60 is connected to the other end of the DC reactor at a connection point c. As the switching element, for example, a well-known element such as an IGBT and a reverse blocking type element can be used. When using a switching element that is not a reverse blocking type, a diode may be added in series.
　次に本実施例の整流回路の動作を説明する。一連の説明において、外部から入力される三相交流電圧波形として正弦波を考える。図１７において、図１に示すように系統入流力端子ｕ、ｖ、ｗに系統電圧（例えば、５０Ｈｚあるいは６０Ｈｚの系統電圧）が印加されるところ、外部からから系統入流力端子ｕ、ｖ、ｗに与えられる電圧がＶｕ、Ｖｖ、Ｖｗと定義されて示される。ここで、図１７において図の横軸（位相角）は、系統入流力端子ｕ、ｖ、ｗに与えられる系統電圧の位相角として示される。すなわち、例えば、５０Ｈｚあるいは６０Ｈｚで与えられる系統電圧の１周期を３６０°（ｄｅｇ）として示される。この位相角の基準は後述する説明・図面でも同様である。 Next, the operation of the rectifier circuit of this embodiment will be described. In a series of descriptions, a sine wave is considered as a three-phase AC voltage waveform input from the outside. In FIG. 17, when a system voltage (for example, a system voltage of 50 Hz or 60 Hz) is applied to the system input force terminals u, v, w as shown in FIG. 1, the system input force terminals u, v, w are externally applied. Are defined as Vu, Vv, and Vw. Here, the horizontal axis (phase angle) in FIG. 17 is shown as the phase angle of the system voltage applied to the system input force terminals u, v, and w. That is, for example, one cycle of the system voltage given at 50 Hz or 60 Hz is indicated as 360 ° (deg). The reference for the phase angle is the same in the description and drawings described later.
　図１７に示すとおりｕ相電圧Ｖｕを基準とし、これに対してｖ相電圧Ｖｖは１２０°、ｗ相電圧Ｖｗは２４０°、夫々位相がずれているものとする。変換用変圧器３０の巻数比を、Ｎ１/Ｎ０＝１、Ｎ２/Ｎ０＝√３として構成すれば、ブリッジ回路２１、２２からの出力電圧Ｖａ、Ｖｂは夫々図１７に示したものとなる。即ち、基本周波数が入力交流周波数の６倍となる交流変動分を含んだ同一の波形形状を有し、互いに位相が３０°ずれた電圧波形となる（系統電圧の６０°相当が１周期となる）。そして、出力電圧Ｖａ、Ｖｂは互いに３０°毎に電圧の大小が入れ替わるものとなっている。 As shown in FIG. 17, the u-phase voltage Vu is used as a reference, and the v-phase voltage Vv is 120 ° and the w-phase voltage Vw is 240 °, which are out of phase. If the turns ratio of the conversion transformer 30 is configured as N1 / N0 = 1 and N2 / N0 = √3, the output voltages Va and Vb from the bridge circuits 21 and 22 are as shown in FIG. That is, it has the same waveform shape including an AC fluctuation component whose basic frequency is 6 times the input AC frequency, and becomes a voltage waveform whose phase is shifted by 30 ° from each other (corresponding to 60 ° of the system voltage is one cycle. ). The output voltages Va and Vb are interchanged with each other every 30 °.
　ここで、図１８に示した様に、スイッチング素子ＳＷ１、ＳＷ２をＯＮ/ＯＦＦ制御する。系統電圧の１．５°相当をＯＮ/ＯＦＦ周期としてその中の所定の期間をオンとして残りの期間をオフとする。出力電圧Ｖａ≧Ｖｂとなる位相角の範囲においてＳＷ２を常時ＯＮとし、ＳＷ１をＯＮ/ＯＦＦ制御することにより接続点ａを通過して接続点ｃに流れる電流Ｉａと、接続点ｂを通過して接続点ｃに流れる電流Ｉｂの比を変えることができる。ここで、電流Ｉａ、Ｉｂは一回のＯＮ/ＯＦＦ動作における平均電流の意味で使用しており、接続点ｃを流れる電流Ｉｃとの関係式は以下となる。
（Ｉａ＋Ｉｂ＝Ｉｃ）または（Ｉａ/Ｉｃ＋Ｉｂ/Ｉｃ＝１）
　同様にＶａ≦Ｖｂとなる位相角の範囲においてはＳＷ１を常時ＯＮとし、ＳＷ２をＯＮ/ＯＦＦ制御することにより電流ＩａとＩｂの比率を変えることができる。また、結合回路１０はスイッチング素子とリアクトルのみで構成されていて、理想的にはエネルギー損失が生じないことから、一回のＯＮ/ＯＦＦ周期が十分短く、当該時間における電圧の変化が無視できる場合、電圧と電流の関係式として次式（１）または（２）が成り立つ。

Ｖｃ・Ｉｃ＝Ｖａ・Ｉａ＋Ｖｂ・Ｉｂ　…　式（１）
または
Ｖｃ＝Ｖａ・Ｉａ/Ｉｃ＋Ｖｂ・Ｉｂ/Ｉｃ　…　式（２）
Here, as shown in FIG. 18, the switching elements SW1 and SW2 are ON / OFF controlled. An ON / OFF period corresponding to 1.5 ° of the system voltage is set as an ON / OFF cycle, and a predetermined period is turned ON, and the remaining period is turned OFF. In the phase angle range where the output voltage Va ≧ Vb, SW2 is always turned on, and SW1 is turned on / off, thereby passing through the connection point a and flowing through the connection point c, and passing through the connection point b. The ratio of the current Ib flowing through the connection point c can be changed. Here, the currents Ia and Ib are used to mean the average current in one ON / OFF operation, and the relational expression with the current Ic flowing through the connection point c is as follows.
(Ia + Ib = Ic) or (Ia / Ic + Ib / Ic = 1)
Similarly, in the phase angle range where Va ≦ Vb, the ratio of the currents Ia and Ib can be changed by always turning SW1 ON and controlling SW2 ON / OFF. In addition, since the coupling circuit 10 is composed only of a switching element and a reactor, and ideally no energy loss occurs, a single ON / OFF cycle is sufficiently short, and a change in voltage at the time can be ignored. As a relational expression between voltage and current, the following expression (1) or (2) is established.

Vc · Ic = Va · Ia + Vb · Ib Formula (1)
Or Vc = Va · Ia / Ic + Vb · Ib / Ic (2)
　従って、スイッチング素子ＳＷ１あるいはＳＷ２をＯＮ/ＯＦＦ制御することにより電流ＩａとＩｂの比率を変え、ＶａとＶｂの中間範囲においてＶｃの値を任意に制御することができる。ここで図２に示した如く、電流比Ｉａ/Ｉｃが、位相角０°で０．５＋αであり、単調減少して位相角３０°で０．５－αとなり、ここから単調増加して位相角６０°で０．５＋αとなる動作を繰り返すように、また、Ｉｂ/Ｉｃが、位相角０°で０．５－αであり、単調増加少して位相角３０°で０．５＋αとなり、さらに、単調減少して位相角６０°で０．５－αとなる動作を繰り返すように、スイッチング素子ＳＷ１、ＳＷ２をＯＮ/ＯＦＦ制御する。このように、Ｉａ/Ｉｃ、Ｉｂ/Ｉｃを制御すれば、図３に示した如くＶｃを一定値とすることができ、負荷の手前に直流フィルタを設置することが不要となる。 Therefore, the ratio of the currents Ia and Ib can be changed by ON / OFF control of the switching element SW1 or SW2, and the value of Vc can be arbitrarily controlled in the intermediate range between Va and Vb. Here, as shown in FIG. 2, the current ratio Ia / Ic is 0.5 + α at a phase angle of 0 °, decreases monotonously to 0.5-α at a phase angle of 30 °, and increases monotonically from here. In addition, Ib / Ic is 0.5-α at a phase angle of 0 °, and is slightly increased to 0.5 + α at a phase angle of 30 °, so that the operation of 0.5 + α is repeated at an angle of 60 °. The switching elements SW1 and SW2 are ON / OFF controlled so as to repeat the operation of monotonically decreasing and reaching 0.5-α at a phase angle of 60 °. In this way, if Ia / Ic and Ib / Ic are controlled, Vc can be set to a constant value as shown in FIG. 3, and it is not necessary to install a DC filter before the load.
　図４は上記制御を実施したときの変換用変圧器ｕ相の各二次側電流の一次側電流換算値－位相角線図を示したものである。ｕ相の一次側電流はこれら各二次側電流の一次側電流換算値の和となる。これを図５に示す。図５に示された電流波形でも用途によっては十分に正弦波に近い、高調波成分が小さいものと言える。
FIG. 4 shows a primary-side converted current-phase angle diagram of each secondary-side current of the conversion transformer u-phase when the above control is performed. The u-phase primary side current is the sum of the primary side current converted values of these secondary side currents. This is shown in FIG. Even in the current waveform shown in FIG. 5, it can be said that the harmonic component is sufficiently close to a sine wave depending on the application and has a small harmonic component.
　実施例２について図６ないし図１０を用いて説明する。図において符号の番号が同じものは実施例１と基本的に同等なものである。以下の実施例では、それまでに説明した実施例と異なる部分を中心に説明するのであって、説明が省略された部分はそれまでに説明された実施例と技術的に異なっていない限りにおいて同じものである。 Example 2 will be described with reference to FIGS. In the figure, the same reference numerals are basically equivalent to those in the first embodiment. In the following embodiments, the description will focus on the differences from the embodiments described so far, and the portions that are not described are the same as long as they are not technically different from the embodiments described so far. Is.
　図６は本実施例の整流回路の主回路構成図、図７はその整流回路の各直流電流比率－位相角線図、図８は当該整流回路における各直流電圧の電圧－位相角線図、図９は当該整流回路における変換用変圧器u相の各二次側電流の一次側電流換算値－位相角線図、図５はu相の一次側電流－位相角線図である。 6 is a main circuit configuration diagram of the rectifier circuit of this embodiment, FIG. 7 is a DC current ratio-phase angle diagram of the rectifier circuit, FIG. 8 is a voltage-phase angle diagram of each DC voltage in the rectifier circuit, FIG. 9 is a primary current converted value—phase angle diagram of each secondary current of the conversion transformer u phase in the rectifier circuit, and FIG. 5 is a primary phase current—phase angle diagram of the u phase.
　本実施例では、図１に示した実施例１におけるスイッチング電流平滑用コンデンサＣ１、Ｃ２の接続位置が、夫々接続点ａと接続点ｃとの間、接続点ｂと接続点ｃとの間に変更されるとともに、接続点ｃと負荷６０の間に並列共振回路４０が挿入されたものとなっている。並列共振回路４０を構成するリアクトルＬ４１のインダクタンスとコンデンサＣ４１の静電容量は、１次側三相交流の周波数の１２倍の周波数で共振する様に選定されている。 In the present embodiment, the connection positions of the switching current smoothing capacitors C1 and C2 in the first embodiment shown in FIG. 1 are between the connection point a and the connection point c, and between the connection point b and the connection point c, respectively. In addition to the change, the parallel resonance circuit 40 is inserted between the connection point c and the load 60. The inductance of the reactor L41 and the electrostatic capacity of the capacitor C41 constituting the parallel resonance circuit 40 are selected so as to resonate at a frequency 12 times the frequency of the primary side three-phase alternating current.
　ここで、図７に示した通りに電流比率を制御すれば、図８に示した様にＶｃの電圧波形として、直流電圧に１次側三相交流の周波数の１２倍の周波数の正弦波が重畳されたものを出力することができる。この重畳された正弦波は並列共振回路４０により除去され、負荷６０には直流電圧のみが印加される。負荷６０が純抵抗負荷であれば電流Ｉｃは一定値となる。 Here, if the current ratio is controlled as shown in FIG. 7, as shown in FIG. 8, a sine wave having a frequency 12 times the frequency of the primary three-phase alternating current is generated in the DC voltage as the voltage waveform of Vc. The superimposition can be output. The superimposed sine wave is removed by the parallel resonance circuit 40, and only a DC voltage is applied to the load 60. If the load 60 is a pure resistance load, the current Ic has a constant value.
　図９は上記制御を実施し0たときの変換用変圧器u相の各二次側電流の一次側電流換算値－位相角線図を示したものである。u相の一次側電流はこれら各二次側電流の一次側電流換算値の和となる。これを図１０に示す。図１０に示された電流波形は実施例１のものと比べて更なる高調波成分の抑制が実現されている。 FIG. 9 is a primary side current converted value-phase angle diagram of each secondary side current of the conversion transformer u phase when the above control is performed. The primary current of the u phase is the sum of the primary current conversion values of these secondary currents. This is shown in FIG. In the current waveform shown in FIG. 10, further suppression of harmonic components is realized as compared with that of the first embodiment.
　また、コンデンサＣ１、Ｃ２を図示の如く配置することにより、実施例１と比べて耐電圧の小さな素子を選定することが可能となっている。
Further, by disposing the capacitors C1 and C2 as shown in the drawing, it is possible to select an element having a smaller withstand voltage compared to the first embodiment.
　実施例３について図１１ないし図１５を用いて説明する。図１１は本実施例の整流回路の主回路構成図、図１２はその整流回路の各直流電流比率－位相角線図、図１３は当該整流回路における各直流電圧の電圧－位相角線図、図１４は当該整流回路における変換用変圧器u相の各二次側電流の一次側電流換算値－位相角線図、図１５はu相の一次側電流－位相角線図である。 Example 3 will be described with reference to FIGS. 11 is a main circuit configuration diagram of the rectifier circuit of the present embodiment, FIG. 12 is a DC current ratio-phase angle diagram of the rectifier circuit, FIG. 13 is a voltage-phase angle diagram of each DC voltage in the rectifier circuit, FIG. 14 is a primary current converted value—phase angle diagram of each secondary current of the transformer u phase for conversion in the rectifier circuit, and FIG. 15 is a primary current-phase angle diagram of the u phase.
　実施例２が並列共振回路４０を１次側三相交流の周波数の１２倍の周波数で共振する様に構成していたのに対し、本実施例は、並列共振回路４０を１次側三相交流の周波数の１２倍の周波数及び２４倍の周波数で共振する様に構成している。即ち、並列共振回路４０を構成するリアクトルＬ４１のインダクタンスとコンデンサＣ４１の静電容量を、１次側三相交流の周波数の１２倍の周波数で共振する様に選定するとともに、リアクトルＬ４２のインダクタンスとコンデンサＣ４２の静電容量を１次側三相交流の周波数の２４倍の周波数で共振する様に選定している。 In contrast to the second embodiment in which the parallel resonant circuit 40 is configured to resonate at a frequency 12 times the frequency of the primary three-phase alternating current, the second embodiment configures the parallel resonant circuit 40 to have a primary three-phase frequency. It is configured to resonate at a frequency 12 times and 24 times the frequency of the alternating current. In other words, the inductance of the reactor L41 and the capacitance of the capacitor C41 constituting the parallel resonant circuit 40 are selected so as to resonate at a frequency 12 times the frequency of the primary side three-phase alternating current, and the inductance and the capacitor of the reactor L42. The capacitance of C42 is selected so as to resonate at a frequency 24 times the frequency of the primary side three-phase alternating current.
　ここで、図１２に示した通りに電流比率を制御すれば、図１３に示した様にＶｃの電圧波形として、直流電圧に１次側三相交流の周波数の１２倍の周波数の正弦波と２４倍の周波数の正弦波が重畳されたものを出力することができる。この重畳された正弦波は並列共振回路４０により除去され、負荷６０には直流電圧のみが印加される。負荷６０が純抵抗負荷であれば電流Ｉｃは一定値となる。 If the current ratio is controlled as shown in FIG. 12, a sine wave having a frequency 12 times the frequency of the primary side three-phase alternating current is added to the DC voltage as shown in FIG. A signal on which a sine wave having a frequency 24 times is superimposed can be output. The superimposed sine wave is removed by the parallel resonance circuit 40, and only a DC voltage is applied to the load 60. If the load 60 is a pure resistance load, the current Ic has a constant value.
　図１４は上記制御を実施したときの変換用変圧器u相の各二次側電流の一次側電流換算値－位相角線図を示したものである。u相の一次側電流はこれら各二次側電流の一次側電流換算値の和となる。これを図１５に示す。図１５に示された電流波形は実施例２のものと比べて更なる高調波成分の抑制が実現されている。
FIG. 14 shows a primary current converted value-phase angle diagram of each secondary current of the conversion transformer u-phase when the above control is performed. The primary current of the u phase is the sum of the primary current conversion values of these secondary currents. This is shown in FIG. The current waveform shown in FIG. 15 realizes further suppression of higher harmonic components than that of the second embodiment.
　実施例４について図１６を用いて説明する。図１６は本実施例の整流回路の主回路構成図である。本実施例では、図１に示した実施例１におけるブリッジ回路のダイオードＤＩ１１，ＤＩ１３、ＤＩ１５、ＤＩ２１，ＤＩ２３、ＤＩ２５がそれぞれＧＴＯ、ＧＴ１１，ＧＴ１３、ＧＴ１５、ＧＴ２１，ＧＴ２３、ＧＴ２５に置き換えられたものとなっている。基本的な回路動作は実施例１と同一である。一方でＧＴＯに置き換えたことにより、異常時において当該ＧＴＯをＯＦＦすることで確実に電流を遮断できる等の効果がある。
Example 4 will be described with reference to FIG. FIG. 16 is a main circuit configuration diagram of the rectifier circuit of this embodiment. In the present embodiment, the diodes DI11, DI13, DI15, DI21, DI23, and DI25 of the bridge circuit in the first embodiment shown in FIG. 1 are replaced with GTO, GT11, GT13, GT15, GT21, GT23, and GT25, respectively. It has become. The basic circuit operation is the same as in the first embodiment. On the other hand, by replacing the GTO, there is an effect that the current can be surely interrupted by turning off the GTO at the time of abnormality.
１０…結合回路、２１、２２…ブリッジ回路、３０…変換用変圧器、４０…並列共振回路、ＳＷ１，ＳＷ２…主スイッチング素子、Ｌ０…直流リアクトル、Ｌ４１，Ｌ４２…リアクトル、Ｃ４１，Ｃ４２…コンデンサ、Ｄ１１，Ｄ１２，Ｄ１３，Ｄ１４，Ｄ１５，Ｄ１６…ダイオード、Ｄ２１，Ｄ２２，Ｄ２３，Ｄ２４，Ｄ２５，Ｄ２６…ダイオード、ＧＴ１１，ＧＴ１３，ＧＴ１５…副スイッチング素子、ＧＴ２１，ＧＴ２３，ＧＴ２５…副スイッチング素子。 DESCRIPTION OF SYMBOLS 10 ... Coupling circuit, 21, 22 ... Bridge circuit, 30 ... Conversion transformer, 40 ... Parallel resonance circuit, SW1, SW2 ... Main switching element, L0 ... DC reactor, L41, L42 ... Reactor, C41, C42 ... Capacitor, D11, D12, D13, D14, D15, D16 ... diode, D21, D22, D23, D24, D25, D26 ... diode, GT11, GT13, GT15 ... sub-switching element, GT21, GT23, GT25 ... sub-switching element.

Claims (7)

  1. 　位相が異なる２組の交流を出力する変換用変圧器と、前記交流を各々直流に変換する２組の整流回路と、前記２組の整流回路の出力を並列結合する結合回路とを有する整流回路において、前記結合回路が第１のスイッチング素子と第２のスイッチング素子と１つの直流リアクトルとを有するとともに、前記２組の整流回路の一方に接続される第１の入力接続点と、前記２組の整流回路の他方に接続される第２の入力接続点と、出力接続点が形成されており、前記第１の入力接続点と前記第１のスイッチング素子の一端が接続され、前記第２の入力接続点と前記第２のスイッチング素子の一端が接続され、夫々のスイッチング素子の他端がともに前記直流リアクトルの一端に接続され、前記直流リアクトルの他端が前記出力接続点に接続され、前記第１のスイッチング素子は前記第１の整流回路の出力する脈流の周期よりも短い周期でスイッチングし、前記第２のスイッチング素子は前記第２の整流回路の出力する脈流の周期よりも短い周期でスイッチングすることを特徴とする整流回路。
    A rectifier circuit comprising: a transformer for conversion that outputs two sets of alternating currents having different phases; two sets of rectifier circuits that each convert the alternating current into direct current; and a coupling circuit that couples the outputs of the two sets of rectifier circuits in parallel. And the coupling circuit includes a first switching element, a second switching element, and one DC reactor, and a first input connection point connected to one of the two sets of rectifier circuits, and the two sets A second input connection point connected to the other of the rectifier circuit and an output connection point are formed, and the first input connection point and one end of the first switching element are connected to each other. An input connection point and one end of the second switching element are connected, the other end of each switching element is connected to one end of the DC reactor, and the other end of the DC reactor is connected to the output connection point, The first switching element is switched with a cycle shorter than the cycle of the pulsating current output from the first rectifier circuit, and the second switching element is switched over with the cycle of the pulsating flow output from the second rectifier circuit. A rectifier circuit that switches at a short cycle.
  2. 請求項１において、前記第１のスイッチング素子と前記第２のスイッチング素子は、一方がオン動作のときに他方がオン／オフ動作となるように動作することを特徴とする整流回路。
    2. The rectifier circuit according to claim 1, wherein the first switching element and the second switching element operate so that one of the first switching element and the second switching element is turned on / off when the other is turned on.
  3. 請求項１または２において、前記出力電の電圧を一定に保つように、前記第1の入力接続点に流れる電流と、前記第２の入力接続点に流れる電流を制御することを特徴とする整流回路。
    3. The rectification according to claim 1, wherein a current flowing through the first input connection point and a current flowing through the second input connection point are controlled so as to keep the voltage of the output electricity constant. circuit.
  4. 請求項１、２または３において、前記変換用変圧器は、二次側の一方がＹ結線であり、他方がΔ結線であることを特徴とする整流回路。
    4. The rectifier circuit according to claim 1, wherein one of the secondary transformers is Y-connected and the other is Δ-connected.
  5. 請求項１、２、３または４において、前記整流回路は、ブリッジ回路として構成されることを特徴とする整流回路。
    5. The rectifier circuit according to claim 1, wherein the rectifier circuit is configured as a bridge circuit.
  6. 　請求項１、２、３、４または５に記載の整流回路であって、前記出力接続点に並列共振回路が接続されていることを特徴とする整流回路。
    6. The rectifier circuit according to claim 1, wherein a parallel resonant circuit is connected to the output connection point.
  7. 　請求項６に記載の整流回路であって、前記出力接続点と前記並列共振回路の間にさらに並列共振回路を設け、これらの並列共振回路は互いに異なる周波数で共振するように構成されていることを特徴とする整流回路。 7. The rectifier circuit according to claim 6, wherein a parallel resonant circuit is further provided between the output connection point and the parallel resonant circuit, and these parallel resonant circuits are configured to resonate at mutually different frequencies. A rectifier circuit characterized by.
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Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2017153353A (en) * 2016-02-22 2017-08-31 ドクター エンジニール ハー ツェー エフ ポルシェ アクチエンゲゼルシャフトＤｒ． Ｉｎｇ． ｈ．ｃ． Ｆ． Ｐｏｒｓｃｈｅ Ａｋｔｉｅｎｇｅｓｅｌｌｓｃｈａｆｔ Method and apparatus for controlling charge station
WO2018119528A1 (en) * 2016-12-30 2018-07-05 Universidad De Santiago De Chile System for transmitting electrical energy using direct current

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JPS57115580U (en) * 1981-01-08 1982-07-17
JPH01501834A (en) * 1986-12-30 1989-06-22 サンドストランド・コーポレーション Adjusted AC/DC converter
JPH03124783U (en) * 1990-03-30 1991-12-17
JP2008178180A (en) * 2007-01-17 2008-07-31 Fuji Electric Systems Co Ltd Rectifier circuit

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPS57115580U (en) * 1981-01-08 1982-07-17
JPH01501834A (en) * 1986-12-30 1989-06-22 サンドストランド・コーポレーション Adjusted AC/DC converter
JPH03124783U (en) * 1990-03-30 1991-12-17
JP2008178180A (en) * 2007-01-17 2008-07-31 Fuji Electric Systems Co Ltd Rectifier circuit

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2017153353A (en) * 2016-02-22 2017-08-31 ドクター エンジニール ハー ツェー エフ ポルシェ アクチエンゲゼルシャフトＤｒ． Ｉｎｇ． ｈ．ｃ． Ｆ． Ｐｏｒｓｃｈｅ Ａｋｔｉｅｎｇｅｓｅｌｌｓｃｈａｆｔ Method and apparatus for controlling charge station
WO2018119528A1 (en) * 2016-12-30 2018-07-05 Universidad De Santiago De Chile System for transmitting electrical energy using direct current

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