WO2015052894A1 - Carrier frequency deviation estimation device and carrier frequency deviation estimation method - Google Patents
Carrier frequency deviation estimation device and carrier frequency deviation estimation method Download PDFInfo
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- WO2015052894A1 WO2015052894A1 PCT/JP2014/005001 JP2014005001W WO2015052894A1 WO 2015052894 A1 WO2015052894 A1 WO 2015052894A1 JP 2014005001 W JP2014005001 W JP 2014005001W WO 2015052894 A1 WO2015052894 A1 WO 2015052894A1
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
- H04B10/60—Receivers
- H04B10/61—Coherent receivers
- H04B10/616—Details of the electronic signal processing in coherent optical receivers
- H04B10/6164—Estimation or correction of the frequency offset between the received optical signal and the optical local oscillator
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/0014—Carrier regulation
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
- H04L27/22—Demodulator circuits; Receiver circuits
- H04L27/227—Demodulator circuits; Receiver circuits using coherent demodulation
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
- H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
- H04L27/38—Demodulator circuits; Receiver circuits
- H04L27/3818—Demodulator circuits; Receiver circuits using coherent demodulation, i.e. using one or more nominally phase synchronous carriers
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/0014—Carrier regulation
- H04L2027/0024—Carrier regulation at the receiver end
- H04L2027/0026—Correction of carrier offset
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/0014—Carrier regulation
- H04L2027/0044—Control loops for carrier regulation
- H04L2027/0046—Open loops
- H04L2027/0048—Frequency multiplication
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/0014—Carrier regulation
- H04L2027/0044—Control loops for carrier regulation
- H04L2027/0063—Elements of loops
- H04L2027/0067—Phase error detectors
Definitions
- the present invention relates to a carrier frequency deviation estimation apparatus and a carrier frequency deviation estimation method, and more particularly to a carrier frequency deviation estimation apparatus and a carrier frequency deviation estimation method used in an optical communication system.
- the conventionally used light intensity modulation method performs data modulation on the light intensity of the transmission laser beam
- the optical phase modulation method performs data modulation on the phase of the transmission laser beam.
- Known optical phase modulation methods include QPSK (Quadrature Phase Shift Keying) method and 16QAM (Quadrature Amplitude Modulation) method.
- the polarization multiplexing / demultiplexing technique in an optical transmitter, two independent single-polarized optical signals whose carrier waves are arranged in the same frequency band and whose polarization states are orthogonal to each other are polarization-multiplexed. Then, in the optical receiver, the above-described two independent single-polarized optical signals are separated from the received optical signal. With such a configuration, according to the polarization multiplexing / demultiplexing technique, a double transmission rate can be realized.
- FIG. 5 is a block diagram showing a configuration of the optical receiver 500 using the related digital coherent method described above.
- the local oscillation light generation unit 501 transmits local oscillation light having the same frequency band as the received optical signal.
- the received optical signal and the local oscillation light are input to the 90 degree hybrid 510.
- Eight optical signals output from the 90-degree hybrid 510 are converted into electrical signals by photoelectric conversion units 521 to 524, and further converted from analog signals to digital signals by AD converters (Analog-to-Digital Converters: ADCs) 531 to 534. Is converted to The four digital signals generated in this way correspond to the next signal component of the received optical signal.
- AD converters Analog-to-Digital Converters: ADCs
- the digital signals generated by the AD converters 531 to 534 are demodulated by the digital signal processing unit 540 and then decoded into bit strings by the symbol identification units 551 and 552.
- FIG. 6 is a block diagram showing the configuration of the digital signal processing unit 540.
- An X polarization signal is generated as a complex number from the digital signal input from the AD converters 531 and 532 to the X polarization signal generation unit 611.
- a Y polarization signal is generated as a complex number from a digital signal input from the AD converters 533 and 534 to the Y polarization signal generation unit 612.
- the frequency deviation coarse compensation units 621 and 622 compensate the deviation between the center frequency of the received optical signal and the transmission frequency of the local oscillation light (optical carrier frequency deviation) with rough accuracy. This is because when the optical carrier frequency deviation becomes large due to the type of phase modulation method of the received optical signal and the optical SN (signal-to-noise) ratio, the polarization separation unit 640 disposed in the subsequent stage is normally operated. This is because it may not work. Another reason is that when a matched filter is used in the waveform distortion compensators 631 and 632 arranged in the subsequent stage, signal quality may deteriorate if there is a deviation between the received optical signal and the center frequency of the matched filter. is there. If the above-described problem does not occur, the frequency deviation rough compensation units 621 and 622 can be omitted.
- FIG. 7 is a block diagram illustrating a configuration example of the frequency deviation rough compensation unit 620 (621, 622).
- the frequency deviation rough estimation unit 623 estimates the frequency deviation using one of the two branched input signals, and the phase compensation amount calculation unit 624 calculates the phase compensation amount.
- the other signal waits in the delay unit 625 until the phase compensation amount is calculated.
- the phase compensation amount is calculated as the sum of products of the frequency deviation estimated value and the unit sample time (the reciprocal of the sampling rate in the AD converters 531 to 534).
- the input signal that has been waiting in the delay device 625 is phase rotated clockwise by the calculated phase compensation amount to compensate for the frequency deviation.
- the configuration of the frequency deviation rough compensation unit 620 is not limited to that shown in FIG. 7, and a configuration in which the frequency deviation is compensated by shifting the optical spectrum in the frequency direction in the frequency domain as shown in FIG. 8 can also be used.
- the input signal is subjected to fast Fourier transform (FFT) to a frequency domain signal in the FFT unit 626.
- the data shift unit 627 frequency-shifts in the frequency direction opposite to the estimated frequency deviation value, and then the signal is converted into a time-domain signal by inverse fast Fourier transform (IFFT) processing in the IFFT unit 628.
- IFFT inverse fast Fourier transform
- computation distortion may occur at both ends of the processing units in the FFT and IFFT due to the assumption that the FFT and IFFT assume repeated signals. In such a case, processing such as overlap may be performed separately. .
- the oscillation frequency of the local oscillation light is controlled in the direction opposite to the frequency deviation estimated value. It is also possible to compensate for the frequency deviation.
- FIG. 9 is a block diagram showing a configuration of a related frequency deviation rough estimation unit 623 described in Patent Document 2.
- the related frequency deviation rough estimation unit 623 shown in FIG. 9 calculates a product of two consecutive samples for each of the real part and the imaginary part of the input signal, calculates the difference between them, and then moves.
- a low-pass filter that performs processing such as averaging is transmitted. Since it is clear by simulation that the output value of the low-pass filter and the frequency deviation are proportional to each other within a predetermined frequency deviation range, the frequency deviation is estimated from the output value of the low-pass filter. It is possible.
- the waveform distortion compensators 631 and 632 illustrated in FIG. 6 transmit chromatic dispersion compensation, waveform shaping using a matched filter, nonlinear waveform distortion compensation, and the like to the signals input from the frequency deviation coarse compensation units 621 and 622. Various compensation processes are performed to improve quality.
- the polarization separation unit 640 separates the received optical signal into two digital signals corresponding to two independent optical signals that are polarization-multiplexed in the optical transmitter.
- CMA Continuous Modulus Algorithm
- DD-LMS Decision Decided Last Mean Square
- the signal output from the polarization separation unit 640 is converted into a signal that has been oversampled by a factor of 1 in a state where the sample timing is optimized by the resampling units 651 and 652, respectively.
- the resampling units 651 and 652 can be arranged at other positions such as immediately before the polarization separation unit 640, but oversampling of signals input to the frequency deviation compensation units 661 and 662 is 1 time. There is a need.
- the frequency deviation compensation units 661 and 662 completely compensate for the optical carrier frequency deviation that the frequency deviation coarse compensation units 621 and 622 could not compensate.
- the phase deviation compensation units 671 and 672 compensate for the optical phase deviation.
- FIG. 10 shows the configuration of the frequency deviation compensation unit 660 (661, 662).
- the configuration of the frequency deviation compensator 660 is obtained by replacing the frequency deviation coarse estimator 623 included in the frequency deviation coarse compensator 620 shown in FIG. 7 with a frequency deviation estimator 663, and the other configurations are the same. That is, the frequency deviation compensation unit 660 includes a delay unit 625, a frequency deviation estimation unit 663, and a phase compensation amount calculation unit 624.
- FIG. 11 shows an example of the frequency deviation estimation unit 663.
- the configuration of the frequency deviation estimator 663 shown in FIG. 11 is a configuration that employs an algorithm called an M-th power algorithm (M-th Power Algorithm) or a Viterbi algorithm.
- M-th Power Algorithm M-th Power Algorithm
- Viterbi algorithm a Viterbi algorithm
- JP 2012-248944 A (paragraphs “0022” to “0052”) JP 2009-038801 A (paragraphs “0002” to “0015”, FIG. 1)
- the related frequency deviation estimation unit described above has a problem that it is difficult to accurately estimate the frequency deviation over a wide range because the range of the frequency deviation that can be estimated is limited. Furthermore, since it is necessary to use a large number of arithmetic units, there is a problem that when mounted on an integrated circuit, the circuit scale increases and power consumption increases.
- the related carrier frequency deviation estimation apparatus has a problem that it is difficult to accurately estimate the frequency deviation over a wide range without causing an increase in power consumption.
- An object of the present invention is to solve the above-described problem that it is difficult to accurately estimate a frequency deviation over a wide range without causing an increase in power consumption. It is to provide an estimation method.
- the carrier frequency deviation estimating apparatus includes a resampling unit that converts an input signal into an oversampling signal that is oversampled at a multiple of the symbol rate of the input signal, and two temporally continuous signals included in the oversampling signal.
- a time variation vector calculating means for calculating a time variation vector between the sample signals, a filter means for calculating an average value of the time variation vectors, a frequency deviation estimating means for calculating a carrier frequency deviation estimated value based on the average value, and Have
- the carrier frequency deviation compensation device of the present invention has a carrier frequency deviation estimation device for calculating a carrier frequency deviation estimation value, and a frequency deviation compensation means for compensating the carrier frequency deviation of the input signal based on the carrier frequency deviation estimation value
- the carrier frequency deviation estimating apparatus includes a resampling unit that converts an input signal into an oversampling signal that is oversampled at a multiple of a symbol rate of the input signal, and two temporally continuous sample signals included in the oversampling signal.
- a time variation vector calculating unit that calculates a time variation vector between the filter unit, a filter unit that calculates an average value of the time variation vector, and a frequency deviation estimation unit that calculates a carrier frequency deviation estimated value based on the average value.
- an input signal is converted into an oversampling signal obtained by oversampling at a multiple of the symbol rate of the input signal, and two temporally continuous sample signals included in the oversampling signal are converted.
- an average value of the time variation vectors is calculated by filtering, and a carrier frequency deviation estimated value is calculated based on the average value.
- the frequency deviation can be accurately estimated over a wide range without causing an increase in power consumption.
- FIG. 1 is a block diagram showing a configuration of a carrier frequency deviation estimation apparatus 100 according to the first embodiment of the present invention.
- the carrier frequency deviation estimation apparatus 100 includes resampling means 110, time variation vector calculation means 120, filter means 130, and frequency deviation estimation means 140.
- the resampling means 110 converts the input signal into an oversampling signal that is oversampled at a multiple of the symbol rate of the input signal.
- the time variation vector calculation means 120 calculates a time variation vector between two temporally continuous sample signals included in the oversampling signal.
- the filter means 130 calculates the average value of the time variation vectors.
- the frequency deviation estimating means 140 calculates a carrier frequency deviation estimated value based on this average value.
- the time variation vector calculation means 120 can be configured to include a delay device, a complex conjugate device, and an integrator.
- the delay unit delays the sample signal that precedes in time among the two sample signals output by the resampling means 110 by one sample time.
- the complex conjugate unit calculates a complex conjugate of the sample signal output from the delay unit and outputs a complex conjugate sample signal. Then, the accumulator calculates the product of the subsequent sample signal of the two sample signals and the complex conjugate sample signal.
- the frequency deviation estimating means 140 can be configured to include a declination angle calculating unit 141 and a frequency deviation calculating unit 142.
- the deflection angle calculation unit 141 calculates the deflection angle from the average value of the time variation vectors calculated by the filter unit 130.
- the frequency deviation calculation unit 142 calculates a carrier frequency deviation estimated value based on the relationship between the deviation angle and the carrier frequency deviation.
- the carrier frequency deviation estimating apparatus 100 converts the input signal into a signal having a predetermined sampling rate in the resampling means 110. Thereafter, the time variation vector calculation means 120 calculates the product of the complex conjugate of the sample at a certain time and the immediately preceding sample. The value of this product means a vector representing the time variation of the optical signal per one sample time (reciprocal of the sampling rate), that is, a time variation vector.
- the filter means 130 can be a low-pass filter that performs moving average processing or the like. Then, after the time variation vector described above passes through the filter means 130, the deflection angle calculator 141 calculates the deflection angle. Since this declination has a one-to-one correspondence with the frequency deviation as will be described later, the frequency deviation can be estimated.
- FIG. 2 is a diagram illustrating a result of calculating the relationship between the frequency deviation and the declination of an optical signal by a numerical simulation using the carrier frequency deviation estimation apparatus 100 according to the present embodiment.
- a 128 Gbps polarization multiplexed QPSK optical signal was used as the optical signal.
- FIG. 2 shows the results when oversampling after resampling is set to 1 ⁇ , 2 ⁇ , and 4 ⁇ , respectively.
- the oversampling is performed at 2 times and 4 times, the frequency deviation and the declination are proportional to each other in a wide frequency range.
- FIG. 3 shows the result of calculating the relationship between the frequency deviation and the deflection angle of an optical signal by a numerical simulation for a 256 Gbps polarization multiplexed 16QAM optical signal using the carrier frequency deviation estimation apparatus 100 according to the present embodiment. . Similar to the results shown in FIG. 2, when the oversampling is performed twice or four times, it can be seen that the frequency deviation and the declination are proportional to each other in a wide frequency range. On the other hand, it can be seen that there is no proportional relationship when oversampling is performed by a factor of 1.
- the carrier frequency deviation estimating apparatus 100 As described above, in the carrier frequency deviation estimating apparatus 100 according to the present embodiment, the phase rotation amount caused by the frequency deviation slightly present in the time variation vector is extracted by the low-pass filter as the filter means 130.
- the sampling rate is low, in other words, when oversampling is as small as 1 time, information on the amount of phase rotation included in the time variation vector is extremely small. Therefore, the relationship between the frequency deviation and the declination does not have a one-to-one correspondence, and as a result, it is difficult to normally estimate the frequency deviation.
- the carrier frequency deviation estimation apparatus 100 according to the present embodiment is configured to perform oversampling by multiple times in the resampling means 110, so that the frequency deviation and the deflection angle are in a proportional relationship. Therefore, it is possible to estimate the frequency deviation over a wide range.
- the circuit scale and power consumption do not increase.
- the carrier frequency deviation estimation method according to this embodiment, first, an input signal is converted into an oversampling signal that is oversampled at a multiple of the symbol rate of the input signal. Subsequently, a temporal variation vector between two temporally continuous sample signals included in the oversampling signal is calculated. Then, an average value of time variation vectors is calculated by filtering, and a carrier frequency deviation estimated value is calculated based on the average value.
- the carrier frequency deviation estimation method since the frequency deviation and the declination are in a proportional relationship, it is possible to compensate for the frequency deviation over a wide range.
- the frequency deviation can be accurately estimated over a wide range without causing an increase in power consumption.
- the carrier frequency deviation compensating apparatus of this embodiment includes a carrier frequency deviation estimating apparatus 100 that calculates a carrier frequency deviation estimated value, and a frequency deviation compensating unit that compensates the carrier frequency deviation of the input signal based on the carrier frequency deviation estimated value.
- the configuration of the carrier frequency deviation estimation apparatus 100 is as described above with reference to FIG.
- the configuration of the carrier frequency deviation compensator according to the present embodiment may be a configuration in which the frequency deviation rough estimator 623 is replaced with the carrier frequency deviation estimator 100 in the frequency deviation coarse compensator 620 shown in FIG. it can. That is, the frequency deviation compensation means included in the carrier frequency deviation compensation device of the present embodiment can be configured to include a phase compensation amount calculation unit and a phase compensation unit.
- the phase compensation amount calculation unit calculates the phase compensation amount for the input signal based on the carrier frequency deviation estimated value calculated by the carrier frequency deviation estimation device 100 and the unit sampling time.
- the phase compensator compensates for the carrier frequency deviation by rotating the phase of the input signal by the phase compensation amount.
- the carrier frequency deviation estimation apparatus 100 uses one of the two branched input signals among the input signals input to the carrier frequency deviation compensation apparatus of the present embodiment.
- the frequency deviation is estimated, and the phase compensation amount calculation unit (624) calculates the phase compensation amount.
- the other signal waits in the delay unit (625) included in the phase compensation unit until the phase compensation amount is calculated.
- the phase compensation amount is calculated as the sum of products of the frequency deviation estimated value and the unit sample time.
- the input signal waiting in the delay unit (625) is phase-shifted clockwise by the calculated phase compensation amount to compensate for the frequency deviation.
- the configuration of the carrier frequency deviation compensator of this embodiment can be configured such that the frequency deviation rough compensator 620 shown in FIG. . That is, a configuration in which the frequency deviation is compensated by shifting the optical spectrum in the frequency direction in the frequency domain can also be used.
- the frequency deviation compensation means provided in the carrier frequency deviation compensation device of the present embodiment can be configured to include an FFT unit (626), a data shift unit (627), and an IFFT unit (628).
- the FFT unit (626) performs a fast Fourier transform process on the input signal of the carrier frequency deviation compensator.
- the data shift unit (627) shifts the frequency of the fast Fourier transform processed input signal output from the FFT unit (626) by the carrier frequency deviation estimation value calculated by the carrier frequency deviation estimation apparatus 100.
- the IFFT unit (628) performs an inverse Fourier transform process on the output signal of the data shift unit (627).
- the carrier frequency deviation compensating apparatus of the present embodiment since a wide range of carrier frequency deviation estimated values can be obtained by the carrier frequency deviation estimating apparatus 100, the frequency deviation can be compensated over a wide range.
- FIG. 4 is a block diagram showing a configuration of a carrier frequency deviation estimating apparatus 200 and a carrier frequency deviation compensating apparatus 300 using the same according to the second embodiment of the present invention.
- the carrier frequency deviation estimating apparatus 200 includes a first carrier frequency deviation estimating apparatus 211, a second carrier frequency deviation estimating apparatus 212, and a frequency deviation averaging means 220.
- the configurations of the first carrier frequency deviation estimation device 211 and the second carrier frequency deviation estimation device 212 are the same as the configurations of the carrier frequency deviation estimation device 100 according to the first embodiment.
- the first carrier frequency deviation estimation device 211 and the second carrier frequency deviation estimation device 212 are respectively a first polarization input signal (X polarization input signal) and a second polarization input signal (Y polarization input signal). ) As an input signal.
- the first polarization input signal and the second polarization input signal detect a polarization multiplexed optical signal obtained by polarization-multiplexing two single polarization optical signals which are arranged in the same frequency band and orthogonal to each other. Can be obtained.
- the first carrier frequency deviation estimation device 211 and the second carrier frequency deviation estimation device 212 calculate a first carrier frequency deviation estimation value and a second carrier frequency deviation estimation value, which are carrier frequency deviation estimation values, respectively. Output to the frequency deviation averaging means 220. Then, the frequency deviation averaging means 220 calculates a frequency deviation average value that is an average of the first carrier frequency deviation estimated value and the second carrier frequency deviation estimated value.
- the carrier frequency deviation estimation method according to this embodiment, the first polarization input signal and the second polarization input signal are used as input signals.
- the first polarization input signal and the second polarization input signal are detected from a polarization multiplexed optical signal obtained by polarization multiplexing two single polarization optical signals that are arranged in the same frequency band and orthogonal to each other. It is obtained by doing.
- a first carrier frequency deviation estimated value that is a carrier frequency deviation estimated value is calculated for the first polarization input signal, and a second carrier frequency deviation estimated value is calculated for the second polarization input signal.
- the estimated carrier frequency deviation is calculated.
- a frequency deviation average value that is an average of the first carrier frequency deviation estimated value and the second carrier frequency deviation estimated value is calculated.
- the carrier frequency deviation estimation value is obtained using each of the first polarization input signal and the second polarization input signal.
- the averaged frequency deviation average value is calculated.
- the carrier frequency deviation compensating apparatus 300 includes the above-described carrier frequency deviation estimating apparatus 200 and frequency deviation compensating means.
- the frequency deviation compensation means is configured to output a first polarization input signal (X polarization input signal) and a second polarization input signal (Y polarization input signal) based on the average frequency deviation calculated by the carrier frequency deviation estimation apparatus 200. ) To compensate for the carrier frequency deviation.
- the frequency deviation compensation means includes a phase compensation amount calculation unit for calculating a phase compensation amount for the input signal based on the average frequency deviation value and a unit sampling time, and a carrier frequency deviation by rotating the phase of the input signal by the phase compensation amount.
- a phase compensation unit for compensation The configuration of the frequency deviation compensating means for the X polarization input signal and the Y polarization input signal is the same.
- the carrier frequency deviation estimation apparatus 200 uses the one of the two branched X-polarization input signals and Y-polarization input signals of the X-polarization input signal and the Y-polarization input signal. The average value is calculated.
- the phase compensation amount calculation unit 311 (312) calculates the phase compensation amount based on the average frequency deviation value.
- the other signal of the X polarization input signal and the Y polarization input signal waits in the delay unit 321 (322) included in the phase compensation unit until the phase compensation amount is calculated.
- the phase compensation amount is calculated as the sum of products of the frequency deviation average value and the unit sample time.
- the X polarization input signal (Y polarization input signal) that has been waiting in the delay device 321 (322) is phase rotated clockwise by the calculated phase compensation amount, so that the carrier frequency deviation is compensated.
- the accuracy of estimating the carrier frequency deviation by the carrier frequency deviation estimating apparatus 200 can be improved, so that the frequency deviation can be compensated with high accuracy. Become. Thereby, the transmission characteristic of an optical communication system can be improved.
- Frequency deviation estimation device 100, 200 Carrier frequency deviation estimation device 110 Re-sampling means 120 Time variation vector calculation means 130 Filter means 140 Frequency deviation estimation means 141 Deflection angle calculation unit 142 Frequency deviation calculation unit 211 First carrier frequency deviation estimation device 212 Second carrier wave Frequency deviation estimator 220 Frequency deviation averaging means 300 Carrier frequency deviation compensator 311, 312 Phase compensation amount calculator 321, 322 Delay 500 Optical receiver using related digital coherent method 501 Local oscillation light generator 510 90 degree hybrid 521 to 524 Photoelectric conversion unit 531 to 534 AD converter (ADC) 540 Digital signal processing unit 551, 552 Symbol identification unit 611 X-polarization signal generation unit 612 Y-polarization signal generation unit 620, 621, 622 Frequency deviation rough compensation unit 623 Frequency deviation rough estimation unit 624 Phase compensation amount calculation unit 625 delay unit 626 FFT unit 627 Data shift unit 628 IFFT unit 631, 632 Waveform distortion compensation unit 640 Polarization separation unit 651, 652 Resamp
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Abstract
In related carrier frequency deviation estimation devices, it is difficult to accurately estimate a frequency deviation over a wide range without causing an increase in power consumption. Therefore, this carrier frequency deviation estimation device comprises: a resampling means for converting an input signal into an oversampling signal obtained by oversampling at a plurality of times the symbol rate of the input signal; a time-varying vector calculation means for calculating a time-varying vector between two temporally continuous sample signals included in the oversampling signal; a filter means for calculating an average value of the time-varying vectors; and a frequency deviation estimation means for calculating a carrier frequency deviation estimation value on the basis of the average value.
Description
本発明は、搬送波周波数偏差推定装置および搬送波周波数偏差推定方法に関し、特に、光通信システムに用いられる搬送波周波数偏差推定装置および搬送波周波数偏差推定方法に関する。
The present invention relates to a carrier frequency deviation estimation apparatus and a carrier frequency deviation estimation method, and more particularly to a carrier frequency deviation estimation apparatus and a carrier frequency deviation estimation method used in an optical communication system.
インターネットの普及により、基幹系通信システムのトラフィック量が急激に増大していることから、100Gbpsを越える超高速の光通信システムの実用化が期待されている。このような超高速の光通信システムを実現する技術として、光位相変調方式と偏波多重分離技術を組み合わせたディジタルコヒーレント方式が注目されている(例えば、特許文献1参照)。
Because of the widespread use of the Internet, the amount of traffic in the backbone communication system has increased rapidly. Therefore, the practical application of ultra-high-speed optical communication systems exceeding 100 Gbps is expected. As a technique for realizing such an ultra-high-speed optical communication system, a digital coherent system combining an optical phase modulation system and a polarization multiplexing / demultiplexing technique has attracted attention (for example, see Patent Document 1).
従来から用いられている光強度変調方式は送信レーザ光の光強度に対してデータ変調を行うのに対して、光位相変調方式は送信レーザ光の位相に対してデータ変調を行う。光位相変調方式としては、QPSK(4位相偏移変調:Quadruple Phase Shift Keying)方式や16QAM(直交振幅変調:Quadrature Amplitude Modulation)方式などが知られている。
The conventionally used light intensity modulation method performs data modulation on the light intensity of the transmission laser beam, whereas the optical phase modulation method performs data modulation on the phase of the transmission laser beam. Known optical phase modulation methods include QPSK (Quadrature Phase Shift Keying) method and 16QAM (Quadrature Amplitude Modulation) method.
また、偏波多重分離技術では、光送信器において、搬送波が同一の周波数帯に配備され、かつ偏光状態が互いに直交する2個の独立した単一偏光の光信号を偏光多重する。そして、光受信器において、受信光信号から前述の2個の独立した単一偏光の光信号を分離する。このような構成とすることにより偏波多重分離技術によれば、2倍の伝送速度を実現することができる。
In the polarization multiplexing / demultiplexing technique, in an optical transmitter, two independent single-polarized optical signals whose carrier waves are arranged in the same frequency band and whose polarization states are orthogonal to each other are polarization-multiplexed. Then, in the optical receiver, the above-described two independent single-polarized optical signals are separated from the received optical signal. With such a configuration, according to the polarization multiplexing / demultiplexing technique, a double transmission rate can be realized.
図5は、上述した関連するディジタルコヒーレント方式を用いた光受信器500の構成を示すブロック図である。局所発振光生成部501は受信光信号と同一周波数帯の局所発振光を送出する。受信光信号と局所発振光が90度ハイブリッド510に入力される。90度ハイブリッド510から出力される8個の光信号は、光電変換部521~524によって電気信号に変換され、さらにADコンバータ(Analog-to-Digital Converter:ADC)531~534によりアナログ信号からディジタル信号に変換される。このようにして生成された4個のディジタル信号は、受信光信号のうち次の信号成分に相当する。すなわち、90度ハイブリッド510の偏光軸に平行な信号成分(X偏波信号)の実数部と虚数部、および90度ハイブリッド510の偏光軸に直交する信号成分(Y偏波信号)の実数部と虚数部に相当する信号である。ADコンバータ531~534によって生成されたディジタル信号は、ディジタル信号処理部540により復調処理が施された後、シンボル識別部551、552によりビット列に復号される。
FIG. 5 is a block diagram showing a configuration of the optical receiver 500 using the related digital coherent method described above. The local oscillation light generation unit 501 transmits local oscillation light having the same frequency band as the received optical signal. The received optical signal and the local oscillation light are input to the 90 degree hybrid 510. Eight optical signals output from the 90-degree hybrid 510 are converted into electrical signals by photoelectric conversion units 521 to 524, and further converted from analog signals to digital signals by AD converters (Analog-to-Digital Converters: ADCs) 531 to 534. Is converted to The four digital signals generated in this way correspond to the next signal component of the received optical signal. That is, the real part and imaginary part of the signal component (X polarization signal) parallel to the polarization axis of the 90-degree hybrid 510, and the real part of the signal component (Y polarization signal) orthogonal to the polarization axis of the 90-degree hybrid 510, It is a signal corresponding to the imaginary part. The digital signals generated by the AD converters 531 to 534 are demodulated by the digital signal processing unit 540 and then decoded into bit strings by the symbol identification units 551 and 552.
次に、関連するディジタルコヒーレント方式を用いた光受信器500におけるディジタル信号処理の動作について詳細に説明する。
Next, the operation of digital signal processing in the optical receiver 500 using the related digital coherent method will be described in detail.
図6は、ディジタル信号処理部540の構成を示すブロック図である。ADコンバータ531、532からX偏波信号生成部611に入力されたディジタル信号からX偏波信号が複素数として生成される。同様に、ADコンバータ533、534からY偏波信号生成部612に入力されたディジタル信号からY偏波信号が複素数として生成される。
FIG. 6 is a block diagram showing the configuration of the digital signal processing unit 540. An X polarization signal is generated as a complex number from the digital signal input from the AD converters 531 and 532 to the X polarization signal generation unit 611. Similarly, a Y polarization signal is generated as a complex number from a digital signal input from the AD converters 533 and 534 to the Y polarization signal generation unit 612.
周波数偏差粗補償部621、622は、受信光信号の中心周波数と局所発振光の発信周波数の偏差(光搬送波周波数偏差)を粗い精度で補償する。これは受信光信号の位相変調方式の種類や光SN(signal-to-noise)比に起因して光搬送波周波数偏差が大きくなる場合には、後段に配置される偏波分離部640が正常に動作しない可能性があるからである。また別の理由として、後段に配置される波形歪み補償部631、632においてマッチドフィルタを用いる場合、受信光信号とマッチドフィルタの中心周波数に偏差が存在すると信号品質が劣化する可能性があるからである。なお、上述した問題が発生しない場合には、周波数偏差粗補償部621、622を省略することができる。
The frequency deviation coarse compensation units 621 and 622 compensate the deviation between the center frequency of the received optical signal and the transmission frequency of the local oscillation light (optical carrier frequency deviation) with rough accuracy. This is because when the optical carrier frequency deviation becomes large due to the type of phase modulation method of the received optical signal and the optical SN (signal-to-noise) ratio, the polarization separation unit 640 disposed in the subsequent stage is normally operated. This is because it may not work. Another reason is that when a matched filter is used in the waveform distortion compensators 631 and 632 arranged in the subsequent stage, signal quality may deteriorate if there is a deviation between the received optical signal and the center frequency of the matched filter. is there. If the above-described problem does not occur, the frequency deviation rough compensation units 621 and 622 can be omitted.
図7は、周波数偏差粗補償部620(621、622)の構成例を示すブロック図である。2分岐された入力信号の一方を用いて周波数偏差粗推定部623が周波数偏差を推定し、位相補償量算出部624が位相補償量を算出する。他方の信号は位相補償量が算出されるまで遅延器625において待機される。位相補償量は周波数偏差推定値と単位サンプル時間(ADコンバータ531~534におけるサンプリングレートの逆数)の積の総和として算出される。遅延器625において待機していた入力信号は、算出された位相補償量の分だけ時計方向に位相回転されることにより周波数偏差が補償される。
FIG. 7 is a block diagram illustrating a configuration example of the frequency deviation rough compensation unit 620 (621, 622). The frequency deviation rough estimation unit 623 estimates the frequency deviation using one of the two branched input signals, and the phase compensation amount calculation unit 624 calculates the phase compensation amount. The other signal waits in the delay unit 625 until the phase compensation amount is calculated. The phase compensation amount is calculated as the sum of products of the frequency deviation estimated value and the unit sample time (the reciprocal of the sampling rate in the AD converters 531 to 534). The input signal that has been waiting in the delay device 625 is phase rotated clockwise by the calculated phase compensation amount to compensate for the frequency deviation.
周波数偏差粗補償部620の構成は図7に示したものに限らず、図8に示すように、周波数領域において光スペクトルを周波数方向にシフトすることにより周波数偏差を補償する構成を用いることもできる。図8に示した構成では、入力信号はFFT部626において周波数領域の信号に高速フーリエ変換(Fast Fourier Transform:FFT)される。その後、データシフト部627において、周波数偏差推定値と逆の周波数方向に周波数シフトされた後、IFFT部628における逆高速フーリエ変換(Inverse Fast Fourier Transform:IFFT)処理により時間領域の信号に変換される。なお、FFTおよびIFFTにおける処理単位の両端において、FFTおよびIFFTが繰り返し信号を仮定することによる演算歪みが生じる可能性があるが、その場合にはオーバーラップ等の処理を別途行うこととすればよい。
The configuration of the frequency deviation rough compensation unit 620 is not limited to that shown in FIG. 7, and a configuration in which the frequency deviation is compensated by shifting the optical spectrum in the frequency direction in the frequency domain as shown in FIG. 8 can also be used. . In the configuration illustrated in FIG. 8, the input signal is subjected to fast Fourier transform (FFT) to a frequency domain signal in the FFT unit 626. After that, the data shift unit 627 frequency-shifts in the frequency direction opposite to the estimated frequency deviation value, and then the signal is converted into a time-domain signal by inverse fast Fourier transform (IFFT) processing in the IFFT unit 628. . Note that computation distortion may occur at both ends of the processing units in the FFT and IFFT due to the assumption that the FFT and IFFT assume repeated signals. In such a case, processing such as overlap may be performed separately. .
また、特許文献2に記載されているように、発振周波数を制御することが可能な局所発振光生成部501を用いる場合には、局所発振光の発振周波数を周波数偏差推定値と逆方向に制御することによって周波数偏差を補償することも可能である。
Further, as described in Patent Document 2, when using the local oscillation light generation unit 501 capable of controlling the oscillation frequency, the oscillation frequency of the local oscillation light is controlled in the direction opposite to the frequency deviation estimated value. It is also possible to compensate for the frequency deviation.
周波数偏差を粗く推定する周波数偏差粗推定部623の構成の一例が特許文献2に記載されている。図9は、特許文献2に記載されている関連する周波数偏差粗推定部623の構成を示すブロック図である。図9に示した関連する周波数偏差粗推定部623においては、入力信号の実数部および虚数部のそれぞれに対して時間的に連続した2サンプルの積を求め、それらの差分を算出したうえで移動平均等の処理を行う低域通過フィルタを透過させる。所定の周波数偏差の範囲内であれば、低域通過フィルタの出力値と周波数偏差が比例関係になることがシミュレーションによって明らかになっているので、低域通過フィルタの出力値から周波数偏差を推定することが可能である。
An example of the configuration of the frequency deviation rough estimation unit 623 that roughly estimates the frequency deviation is described in Patent Document 2. FIG. 9 is a block diagram showing a configuration of a related frequency deviation rough estimation unit 623 described in Patent Document 2. The related frequency deviation rough estimation unit 623 shown in FIG. 9 calculates a product of two consecutive samples for each of the real part and the imaginary part of the input signal, calculates the difference between them, and then moves. A low-pass filter that performs processing such as averaging is transmitted. Since it is clear by simulation that the output value of the low-pass filter and the frequency deviation are proportional to each other within a predetermined frequency deviation range, the frequency deviation is estimated from the output value of the low-pass filter. It is possible.
図6に示した波形歪み補償部631、632は、周波数偏差粗補償部621、622から入力された信号に対して、波長分散補償、マッチドフィルタを用いた波形整形、非線形波形歪み補償等の伝送品質を向上させるための各種の補償処理を行う。
The waveform distortion compensators 631 and 632 illustrated in FIG. 6 transmit chromatic dispersion compensation, waveform shaping using a matched filter, nonlinear waveform distortion compensation, and the like to the signals input from the frequency deviation coarse compensation units 621 and 622. Various compensation processes are performed to improve quality.
偏波分離部640は、光送信器において偏波多重された2個の独立した光信号に対応する2個のディジタル信号に受信光信号を分離する。偏波分離処理のアルゴリズムとしては、CMA(Continuous Modulus Algorithm)またはDD-LMS(Decision Decided Least Mean Square)等を用いることができる。
The polarization separation unit 640 separates the received optical signal into two digital signals corresponding to two independent optical signals that are polarization-multiplexed in the optical transmitter. As an algorithm for polarization separation processing, CMA (Continuous Modulus Algorithm) or DD-LMS (Decision Decided Last Mean Square) can be used.
偏波分離部640から出力される信号はそれぞれリサンプリング部651、652によってサンプルタイミングを最適化した状態において1倍でオーバーサンプリングした信号に変換される。なおリサンプリング部651、652は偏波分離部640の直前など、他の位置に配置することも可能であるが、周波数偏差補償部661、662に入力される信号のオーバーサンプリングは1倍である必要がある。
The signal output from the polarization separation unit 640 is converted into a signal that has been oversampled by a factor of 1 in a state where the sample timing is optimized by the resampling units 651 and 652, respectively. The resampling units 651 and 652 can be arranged at other positions such as immediately before the polarization separation unit 640, but oversampling of signals input to the frequency deviation compensation units 661 and 662 is 1 time. There is a need.
周波数偏差補償部661、662は、周波数偏差粗補償部621、622が補償しきれなかった光搬送波周波数偏差を完全に補償する。また、位相偏差補償部671、672は光位相偏差を補償する。
The frequency deviation compensation units 661 and 662 completely compensate for the optical carrier frequency deviation that the frequency deviation coarse compensation units 621 and 622 could not compensate. The phase deviation compensation units 671 and 672 compensate for the optical phase deviation.
図10に、周波数偏差補償部660(661、662)の構成を示す。周波数偏差補償部660の構成は、図7に示した周波数偏差粗補償部620が備える周波数偏差粗推定部623を周波数偏差推定部663に置き換えたものであり、他の構成は同じである。すなわち、周波数偏差補償部660は遅延器625、周波数偏差推定部663、および位相補償量算出部624を備える。
FIG. 10 shows the configuration of the frequency deviation compensation unit 660 (661, 662). The configuration of the frequency deviation compensator 660 is obtained by replacing the frequency deviation coarse estimator 623 included in the frequency deviation coarse compensator 620 shown in FIG. 7 with a frequency deviation estimator 663, and the other configurations are the same. That is, the frequency deviation compensation unit 660 includes a delay unit 625, a frequency deviation estimation unit 663, and a phase compensation amount calculation unit 624.
図11に周波数偏差推定部663の一例を示す。図11に示した周波数偏差推定部663の構成は、M乗法アルゴリズム(M-th Power Algorithm)またはビタビアルゴリズムと呼ばれるアルゴリズムを採用した構成である。このアルゴリズムを用いるためには、サンプルタイミングが最適化された状態において1倍でオーバーサンプリングした信号を入力する必要がある。なお、1倍でオーバーサンプリングした信号を用いるため、補償可能な周波数偏差の範囲に制限がある。
FIG. 11 shows an example of the frequency deviation estimation unit 663. The configuration of the frequency deviation estimator 663 shown in FIG. 11 is a configuration that employs an algorithm called an M-th power algorithm (M-th Power Algorithm) or a Viterbi algorithm. In order to use this algorithm, it is necessary to input a signal that has been oversampled by a factor of 1 in a state where the sample timing is optimized. Since a signal oversampled by 1 is used, there is a limit to the range of frequency deviation that can be compensated.
上記説明した光位相変調方式と偏光多重分離技術を組み合わせたディジタルコヒーレント方式を用いることにより、100Gbpsといった超高速の光通信システムを実現することが可能となる。
By using a digital coherent method combining the optical phase modulation method and polarization multiplexing / demultiplexing technology described above, it is possible to realize an ultra-high-speed optical communication system such as 100 Gbps.
しかしながら、上述した関連する周波数偏差推定部においては、推定可能な周波数偏差の範囲が制限されるため、広範囲にわたって精度よく周波数偏差を推定することが困難であるという問題があった。さらに、多数の演算器を使用する必要があるため、集積回路に実装したときに回路規模が増大し、消費電力が増大するという問題があった。
However, the related frequency deviation estimation unit described above has a problem that it is difficult to accurately estimate the frequency deviation over a wide range because the range of the frequency deviation that can be estimated is limited. Furthermore, since it is necessary to use a large number of arithmetic units, there is a problem that when mounted on an integrated circuit, the circuit scale increases and power consumption increases.
このように、関連する搬送波周波数偏差推定装置においては、消費電力の増大を招くことなく、広範囲にわたって精度よく周波数偏差を推定することが困難であるという問題があった。
Thus, the related carrier frequency deviation estimation apparatus has a problem that it is difficult to accurately estimate the frequency deviation over a wide range without causing an increase in power consumption.
本発明の目的は、上述した課題である、消費電力の増大を招くことなく、広範囲にわたって精度よく周波数偏差を推定することが困難である、という課題を解決する搬送波周波数偏差推定装置および搬送波周波数偏差推定方法を提供することにある。
An object of the present invention is to solve the above-described problem that it is difficult to accurately estimate a frequency deviation over a wide range without causing an increase in power consumption. It is to provide an estimation method.
本発明の搬送波周波数偏差推定装置は、入力信号を、入力信号のシンボルレートの複数倍でオーバーサンプリングしたオーバーサンプリング信号に変換するリサンプリング手段と、オーバーサンプリング信号に含まれる時間的に連続する2個のサンプル信号間の時間変動ベクトルを算出する時間変動ベクトル算出手段と、時間変動ベクトルの平均値を算出するフィルタ手段と、平均値に基づいて搬送波周波数偏差推定値を算出する周波数偏差推定手段、とを有する。
The carrier frequency deviation estimating apparatus according to the present invention includes a resampling unit that converts an input signal into an oversampling signal that is oversampled at a multiple of the symbol rate of the input signal, and two temporally continuous signals included in the oversampling signal. A time variation vector calculating means for calculating a time variation vector between the sample signals, a filter means for calculating an average value of the time variation vectors, a frequency deviation estimating means for calculating a carrier frequency deviation estimated value based on the average value, and Have
本発明の搬送波周波数偏差補償装置は、搬送波周波数偏差推定値を算出する搬送波周波数偏差推定装置と、搬送波周波数偏差推定値に基づいて入力信号の搬送波周波数偏差を補償する周波数偏差補償手段を有し、搬送波周波数偏差推定装置は、入力信号を、入力信号のシンボルレートの複数倍でオーバーサンプリングしたオーバーサンプリング信号に変換するリサンプリング手段と、オーバーサンプリング信号に含まれる時間的に連続する2個のサンプル信号間の時間変動ベクトルを算出する時間変動ベクトル算出手段と、時間変動ベクトルの平均値を算出するフィルタ手段と、平均値に基づいて搬送波周波数偏差推定値を算出する周波数偏差推定手段、とを備える。
The carrier frequency deviation compensation device of the present invention has a carrier frequency deviation estimation device for calculating a carrier frequency deviation estimation value, and a frequency deviation compensation means for compensating the carrier frequency deviation of the input signal based on the carrier frequency deviation estimation value, The carrier frequency deviation estimating apparatus includes a resampling unit that converts an input signal into an oversampling signal that is oversampled at a multiple of a symbol rate of the input signal, and two temporally continuous sample signals included in the oversampling signal. A time variation vector calculating unit that calculates a time variation vector between the filter unit, a filter unit that calculates an average value of the time variation vector, and a frequency deviation estimation unit that calculates a carrier frequency deviation estimated value based on the average value.
本発明の搬送波周波数偏差推定方法は、入力信号を、入力信号のシンボルレートの複数倍でオーバーサンプリングしたオーバーサンプリング信号に変換し、オーバーサンプリング信号に含まれる時間的に連続する2個のサンプル信号間の時間変動ベクトルを算出し、フィルタ処理により時間変動ベクトルの平均値を算出し、平均値に基づいて搬送波周波数偏差推定値を算出する。
According to the carrier frequency deviation estimation method of the present invention, an input signal is converted into an oversampling signal obtained by oversampling at a multiple of the symbol rate of the input signal, and two temporally continuous sample signals included in the oversampling signal are converted. Are calculated, an average value of the time variation vectors is calculated by filtering, and a carrier frequency deviation estimated value is calculated based on the average value.
本発明の搬送波周波数偏差推定装置および搬送波周波数偏差推定方法によれば、消費電力の増大を招くことなく、広範囲にわたって精度よく周波数偏差を推定することができる。
According to the carrier frequency deviation estimation apparatus and the carrier frequency deviation estimation method of the present invention, the frequency deviation can be accurately estimated over a wide range without causing an increase in power consumption.
以下に、図面を参照しながら、本発明の実施形態について説明する。
Hereinafter, embodiments of the present invention will be described with reference to the drawings.
〔第1の実施形態〕
図1は、本発明の第1の実施形態に係る搬送波周波数偏差推定装置100の構成を示すブロック図である。搬送波周波数偏差推定装置100は、リサンプリング手段110、時間変動ベクトル算出手段120、フィルタ手段130、および周波数偏差推定手段140を有する。 [First Embodiment]
FIG. 1 is a block diagram showing a configuration of a carrier frequencydeviation estimation apparatus 100 according to the first embodiment of the present invention. The carrier frequency deviation estimation apparatus 100 includes resampling means 110, time variation vector calculation means 120, filter means 130, and frequency deviation estimation means 140.
図1は、本発明の第1の実施形態に係る搬送波周波数偏差推定装置100の構成を示すブロック図である。搬送波周波数偏差推定装置100は、リサンプリング手段110、時間変動ベクトル算出手段120、フィルタ手段130、および周波数偏差推定手段140を有する。 [First Embodiment]
FIG. 1 is a block diagram showing a configuration of a carrier frequency
リサンプリング手段110は、入力信号を、入力信号のシンボルレートの複数倍でオーバーサンプリングしたオーバーサンプリング信号に変換する。時間変動ベクトル算出手段120は、オーバーサンプリング信号に含まれる時間的に連続する2個のサンプル信号間の時間変動ベクトルを算出する。フィルタ手段130は、時間変動ベクトルの平均値を算出する。そして、周波数偏差推定手段140は、この平均値に基づいて搬送波周波数偏差推定値を算出する。
The resampling means 110 converts the input signal into an oversampling signal that is oversampled at a multiple of the symbol rate of the input signal. The time variation vector calculation means 120 calculates a time variation vector between two temporally continuous sample signals included in the oversampling signal. The filter means 130 calculates the average value of the time variation vectors. Then, the frequency deviation estimating means 140 calculates a carrier frequency deviation estimated value based on this average value.
時間変動ベクトル算出手段120は図1に示すように、遅延器、複素共役器、および積算器を備えた構成とすることができる。ここで遅延器は、リサンプリング手段110が出力する2個のサンプル信号のうち時間的に先行するサンプル信号を1サンプル時間だけ遅延させる。複素共役器は、遅延器が出力するサンプル信号の複素共役を算出し複素共役サンプル信号を出力する。そして積算器は、2個のサンプル信号のうち後続のサンプル信号と、複素共役サンプル信号との積を算出する。
As shown in FIG. 1, the time variation vector calculation means 120 can be configured to include a delay device, a complex conjugate device, and an integrator. Here, the delay unit delays the sample signal that precedes in time among the two sample signals output by the resampling means 110 by one sample time. The complex conjugate unit calculates a complex conjugate of the sample signal output from the delay unit and outputs a complex conjugate sample signal. Then, the accumulator calculates the product of the subsequent sample signal of the two sample signals and the complex conjugate sample signal.
また、周波数偏差推定手段140は図1に示すように、偏角算出部141と周波数偏差算出部142を備えた構成とすることができる。ここで偏角算出部141は、フィルタ手段130が算出した時間変動ベクトルの平均値から偏角を算出する。そして周波数偏差算出部142は、この偏角と搬送波周波数偏差との関係に基づいて搬送波周波数偏差推定値を算出する。
Further, as shown in FIG. 1, the frequency deviation estimating means 140 can be configured to include a declination angle calculating unit 141 and a frequency deviation calculating unit 142. Here, the deflection angle calculation unit 141 calculates the deflection angle from the average value of the time variation vectors calculated by the filter unit 130. Then, the frequency deviation calculation unit 142 calculates a carrier frequency deviation estimated value based on the relationship between the deviation angle and the carrier frequency deviation.
次に、本実施形態による搬送波周波数偏差推定装置100の動作について説明する。
Next, the operation of the carrier frequency deviation estimation apparatus 100 according to this embodiment will be described.
搬送波周波数偏差推定装置100は上述したように、リサンプリング手段110において入力信号を所定のサンプリングレートの信号に変換する。その後に、時間変動ベクトル算出手段120において、ある時刻のサンプルとその直前のサンプルの複素共役の積を算出する。この積の値は1サンプル時間(サンプリングレートの逆数)当たりの光信号の時間変動を表すベクトル、すなわち時間変動ベクトルを意味する。
As described above, the carrier frequency deviation estimating apparatus 100 converts the input signal into a signal having a predetermined sampling rate in the resampling means 110. Thereafter, the time variation vector calculation means 120 calculates the product of the complex conjugate of the sample at a certain time and the immediately preceding sample. The value of this product means a vector representing the time variation of the optical signal per one sample time (reciprocal of the sampling rate), that is, a time variation vector.
フィルタ手段130には、移動平均処理等を行う低域通過フィルタを用いることができる。そして、上述した時間変動ベクトルがこのフィルタ手段130を透過した後に、偏角算出部141において偏角を算出する。この偏角は後述するように周波数偏差と一対一に対応するので、周波数偏差を推定することが可能である。
The filter means 130 can be a low-pass filter that performs moving average processing or the like. Then, after the time variation vector described above passes through the filter means 130, the deflection angle calculator 141 calculates the deflection angle. Since this declination has a one-to-one correspondence with the frequency deviation as will be described later, the frequency deviation can be estimated.
上述した偏角と周波数偏差との関係について、図2を用いて説明する。
The relationship between the aforementioned declination and frequency deviation will be described with reference to FIG.
図2は、本実施形態による搬送波周波数偏差推定装置100を用いて、光信号の周波数偏差と偏角との関係を数値シミュレーションにより算出した結果を示す図である。光信号として、128Gbpsの偏波多重QPSK光信号を用いた。図2には、リサンプリング後のオーバーサンプリングを1倍、2倍、4倍にそれぞれ設定した場合の結果を示す。図2より、2倍および4倍でオーバーサンプリングした場合には周波数偏差と偏角が広い周波数範囲で比例関係にあることがわかる。それに対して、1倍でオーバーサンプリングした場合には比例関係には無いことがわかる。
FIG. 2 is a diagram illustrating a result of calculating the relationship between the frequency deviation and the declination of an optical signal by a numerical simulation using the carrier frequency deviation estimation apparatus 100 according to the present embodiment. A 128 Gbps polarization multiplexed QPSK optical signal was used as the optical signal. FIG. 2 shows the results when oversampling after resampling is set to 1 ×, 2 ×, and 4 ×, respectively. As can be seen from FIG. 2, when the oversampling is performed at 2 times and 4 times, the frequency deviation and the declination are proportional to each other in a wide frequency range. On the other hand, it can be seen that there is no proportional relationship when oversampling is performed by a factor of 1.
図3に、256Gbpsの偏波多重16QAM光信号に対して、本実施形態による搬送波周波数偏差推定装置100を用いて、光信号の周波数偏差と偏角との関係を数値シミュレーションにより算出した結果を示す。図2に示した結果と同様に、2倍および4倍でオーバーサンプリングした場合には、周波数偏差と偏角が広い周波数範囲で比例関係にあることがわかる。それに対して、1倍でオーバーサンプリングした場合には比例関係には無いことがわかる。
FIG. 3 shows the result of calculating the relationship between the frequency deviation and the deflection angle of an optical signal by a numerical simulation for a 256 Gbps polarization multiplexed 16QAM optical signal using the carrier frequency deviation estimation apparatus 100 according to the present embodiment. . Similar to the results shown in FIG. 2, when the oversampling is performed twice or four times, it can be seen that the frequency deviation and the declination are proportional to each other in a wide frequency range. On the other hand, it can be seen that there is no proportional relationship when oversampling is performed by a factor of 1.
図2および図3に示したシミュレーション結果に基づいて、本実施形態による搬送波周波数偏差推定装置100の動作について説明する。
Based on the simulation results shown in FIGS. 2 and 3, the operation of the carrier frequency deviation estimation apparatus 100 according to the present embodiment will be described.
上述したように、本実施形態による搬送波周波数偏差推定装置100においては、時間変動ベクトルに僅かに存在する周波数偏差により生じる位相回転量を、フィルタ手段130としての低域通過フィルタにより抽出している。ここで、サンプリングレートが低い場合、言い換えれば、オーバーサンプリングが1倍のように小さい場合には、時間変動ベクトルに含まれる位相回転量の情報がきわめて少ない。そのため、周波数偏差と偏角の関係が一対一対応とはならず、その結果、周波数偏差を正常に推定することが困難である。それに対して本実施形態による搬送波周波数偏差推定装置100は、リサンプリング手段110において複数倍でオーバーサンプリングする構成としているので、周波数偏差と偏角は比例関係となる。そのため、広範囲にわたって周波数偏差を推定することが可能である。しかも、実装する際に必要となる演算器の個数が増加することはないので、回路規模および消費電力の増大を招くことはない。
As described above, in the carrier frequency deviation estimating apparatus 100 according to the present embodiment, the phase rotation amount caused by the frequency deviation slightly present in the time variation vector is extracted by the low-pass filter as the filter means 130. Here, when the sampling rate is low, in other words, when oversampling is as small as 1 time, information on the amount of phase rotation included in the time variation vector is extremely small. Therefore, the relationship between the frequency deviation and the declination does not have a one-to-one correspondence, and as a result, it is difficult to normally estimate the frequency deviation. On the other hand, the carrier frequency deviation estimation apparatus 100 according to the present embodiment is configured to perform oversampling by multiple times in the resampling means 110, so that the frequency deviation and the deflection angle are in a proportional relationship. Therefore, it is possible to estimate the frequency deviation over a wide range. In addition, since the number of arithmetic units required for mounting does not increase, the circuit scale and power consumption do not increase.
次に、本実施形態による搬送波周波数偏差推定方法について説明する。本実施形態の搬送波周波数偏差推定方法においては、まず、入力信号を、入力信号のシンボルレートの複数倍でオーバーサンプリングしたオーバーサンプリング信号に変換する。続いて、このオーバーサンプリング信号に含まれる時間的に連続する2個のサンプル信号間の時間変動ベクトルを算出する。そして、フィルタ処理により時間変動ベクトルの平均値を算出し、この平均値に基づいて搬送波周波数偏差推定値を算出する。
Next, the carrier frequency deviation estimation method according to this embodiment will be described. In the carrier frequency deviation estimation method of this embodiment, first, an input signal is converted into an oversampling signal that is oversampled at a multiple of the symbol rate of the input signal. Subsequently, a temporal variation vector between two temporally continuous sample signals included in the oversampling signal is calculated. Then, an average value of time variation vectors is calculated by filtering, and a carrier frequency deviation estimated value is calculated based on the average value.
本実施形態による搬送波周波数偏差推定方法によっても、周波数偏差と偏角は比例関係となるため、広範囲にわたって周波数偏差を補償することが可能である。
Also by the carrier frequency deviation estimation method according to the present embodiment, since the frequency deviation and the declination are in a proportional relationship, it is possible to compensate for the frequency deviation over a wide range.
以上説明したように、本実施形態の搬送波周波数偏差推定装置および搬送波周波数偏差推定方法によれば、消費電力の増大を招くことなく、広範囲にわたって精度よく周波数偏差を推定することができる。
As described above, according to the carrier frequency deviation estimation apparatus and the carrier frequency deviation estimation method of the present embodiment, the frequency deviation can be accurately estimated over a wide range without causing an increase in power consumption.
次に、本実施形態による搬送波周波数偏差推定装置100を用いた搬送波周波数偏差補償装置について説明する。本実施形態の搬送波周波数偏差補償装置は、搬送波周波数偏差推定値を算出する搬送波周波数偏差推定装置100と、この搬送波周波数偏差推定値に基づいて入力信号の搬送波周波数偏差を補償する周波数偏差補償手段を有する。搬送波周波数偏差推定装置100の構成は、図1を用いて上述した通りである。
Next, a carrier frequency deviation compensating apparatus using the carrier frequency deviation estimating apparatus 100 according to the present embodiment will be described. The carrier frequency deviation compensating apparatus of this embodiment includes a carrier frequency deviation estimating apparatus 100 that calculates a carrier frequency deviation estimated value, and a frequency deviation compensating unit that compensates the carrier frequency deviation of the input signal based on the carrier frequency deviation estimated value. Have. The configuration of the carrier frequency deviation estimation apparatus 100 is as described above with reference to FIG.
ここで、本実施形態の搬送波周波数偏差補償装置の構成は、図7に示した周波数偏差粗補償部620において、周波数偏差粗推定部623を搬送波周波数偏差推定装置100に置き換えた構成とすることができる。すなわち、本実施形態の搬送波周波数偏差補償装置が備える周波数偏差補償手段は、位相補償量算出部と位相補償部を有する構成とすることができる。
Here, the configuration of the carrier frequency deviation compensator according to the present embodiment may be a configuration in which the frequency deviation rough estimator 623 is replaced with the carrier frequency deviation estimator 100 in the frequency deviation coarse compensator 620 shown in FIG. it can. That is, the frequency deviation compensation means included in the carrier frequency deviation compensation device of the present embodiment can be configured to include a phase compensation amount calculation unit and a phase compensation unit.
位相補償量算出部は、搬送波周波数偏差推定装置100が算出する搬送波周波数偏差推定値と単位サンプリング時間に基づいて入力信号に対する位相補償量を算出する。そして、位相補償部は入力信号の位相を位相補償量だけ回転させることにより搬送波周波数偏差を補償する。
The phase compensation amount calculation unit calculates the phase compensation amount for the input signal based on the carrier frequency deviation estimated value calculated by the carrier frequency deviation estimation device 100 and the unit sampling time. The phase compensator compensates for the carrier frequency deviation by rotating the phase of the input signal by the phase compensation amount.
具体的には図7を用いて説明したように、本実施形態の搬送波周波数偏差補償装置に入力された入力信号のうち、2分岐された入力信号の一方を用いて搬送波周波数偏差推定装置100が周波数偏差を推定し、位相補償量算出部(624)が位相補償量を算出する。他方の信号は位相補償量が算出されるまで位相補償部に含まれる遅延器(625)において待機される。位相補償量は周波数偏差推定値と単位サンプル時間の積の総和として算出される。遅延器(625)において待機していた入力信号は、算出された位相補償量の分だけ時計方向に位相回転されることにより周波数偏差が補償される。
Specifically, as described with reference to FIG. 7, the carrier frequency deviation estimation apparatus 100 uses one of the two branched input signals among the input signals input to the carrier frequency deviation compensation apparatus of the present embodiment. The frequency deviation is estimated, and the phase compensation amount calculation unit (624) calculates the phase compensation amount. The other signal waits in the delay unit (625) included in the phase compensation unit until the phase compensation amount is calculated. The phase compensation amount is calculated as the sum of products of the frequency deviation estimated value and the unit sample time. The input signal waiting in the delay unit (625) is phase-shifted clockwise by the calculated phase compensation amount to compensate for the frequency deviation.
また、本実施形態の搬送波周波数偏差補償装置の構成は、図8に示した周波数偏差粗補償部620において、周波数偏差粗推定部623を搬送波周波数偏差推定装置100に置き換えた構成とすることができる。すなわち、周波数領域において光スペクトルを周波数方向にシフトすることにより周波数偏差を補償する構成を用いることもできる。
Further, the configuration of the carrier frequency deviation compensator of this embodiment can be configured such that the frequency deviation rough compensator 620 shown in FIG. . That is, a configuration in which the frequency deviation is compensated by shifting the optical spectrum in the frequency direction in the frequency domain can also be used.
具体的には、本実施形態の搬送波周波数偏差補償装置が備える周波数偏差補償手段は、FFT部(626)、データシフト部(627)、およびIFFT部(628)を備えた構成とすることができる。ここでFFT部(626)は、搬送波周波数偏差補償装置の入力信号に高速フーリエ変換処理を施す。データシフト部(627)は、FFT部(626)が出力する高速フーリエ変換処理された入力信号を、搬送波周波数偏差推定装置100が算出する搬送波周波数偏差推定値だけ周波数シフトさせる。IFFT部(628)はデータシフト部(627)の出力信号に逆フーリエ変換処理を施す。
Specifically, the frequency deviation compensation means provided in the carrier frequency deviation compensation device of the present embodiment can be configured to include an FFT unit (626), a data shift unit (627), and an IFFT unit (628). . Here, the FFT unit (626) performs a fast Fourier transform process on the input signal of the carrier frequency deviation compensator. The data shift unit (627) shifts the frequency of the fast Fourier transform processed input signal output from the FFT unit (626) by the carrier frequency deviation estimation value calculated by the carrier frequency deviation estimation apparatus 100. The IFFT unit (628) performs an inverse Fourier transform process on the output signal of the data shift unit (627).
上述した本実施形態の搬送波周波数偏差補償装置によれば、搬送波周波数偏差推定装置100によって広範囲な搬送波周波数偏差推定値が得られるので、広範囲にわたって周波数偏差を補償することができる。
According to the carrier frequency deviation compensating apparatus of the present embodiment described above, since a wide range of carrier frequency deviation estimated values can be obtained by the carrier frequency deviation estimating apparatus 100, the frequency deviation can be compensated over a wide range.
〔第2の実施形態〕
次に、本発明の第2の実施形態について説明する。本実施形態では、偏波多重分離技術を用いた場合について説明する。図4は、本発明の第2の実施形態に係る搬送波周波数偏差推定装置200およびそれを用いた搬送波周波数偏差補償装置300の構成を示すブロック図である。 [Second Embodiment]
Next, a second embodiment of the present invention will be described. In the present embodiment, a case where a polarization multiplexing / demultiplexing technique is used will be described. FIG. 4 is a block diagram showing a configuration of a carrier frequencydeviation estimating apparatus 200 and a carrier frequency deviation compensating apparatus 300 using the same according to the second embodiment of the present invention.
次に、本発明の第2の実施形態について説明する。本実施形態では、偏波多重分離技術を用いた場合について説明する。図4は、本発明の第2の実施形態に係る搬送波周波数偏差推定装置200およびそれを用いた搬送波周波数偏差補償装置300の構成を示すブロック図である。 [Second Embodiment]
Next, a second embodiment of the present invention will be described. In the present embodiment, a case where a polarization multiplexing / demultiplexing technique is used will be described. FIG. 4 is a block diagram showing a configuration of a carrier frequency
本実施形態による搬送波周波数偏差推定装置200は、第1の搬送波周波数偏差推定装置211、第2の搬送波周波数偏差推定装置212、および周波数偏差平均手段220を有する。ここで第1の搬送波周波数偏差推定装置211および第2の搬送波周波数偏差推定装置212の構成はそれぞれ、第1の実施形態に係る搬送波周波数偏差推定装置100の構成と同様である。
The carrier frequency deviation estimating apparatus 200 according to the present embodiment includes a first carrier frequency deviation estimating apparatus 211, a second carrier frequency deviation estimating apparatus 212, and a frequency deviation averaging means 220. Here, the configurations of the first carrier frequency deviation estimation device 211 and the second carrier frequency deviation estimation device 212 are the same as the configurations of the carrier frequency deviation estimation device 100 according to the first embodiment.
第1の搬送波周波数偏差推定装置211と第2の搬送波周波数偏差推定装置212はそれぞれ、第1の偏波入力信号(X偏波入力信号)と第2の偏波入力信号(Y偏波入力信号)を入力信号としている。ここで第1の偏波入力信号と第2の偏波入力信号は、中心周波数が同一周波数帯に配置され互いに直交する2個の単一偏波光信号を偏光多重した偏光多重光信号を検波することによって得られる。
The first carrier frequency deviation estimation device 211 and the second carrier frequency deviation estimation device 212 are respectively a first polarization input signal (X polarization input signal) and a second polarization input signal (Y polarization input signal). ) As an input signal. Here, the first polarization input signal and the second polarization input signal detect a polarization multiplexed optical signal obtained by polarization-multiplexing two single polarization optical signals which are arranged in the same frequency band and orthogonal to each other. Can be obtained.
第1の搬送波周波数偏差推定装置211と第2の搬送波周波数偏差推定装置212は、搬送波周波数偏差推定値である第1の搬送波周波数偏差推定値と第2の搬送波周波数偏差推定値をそれぞれ算出し、周波数偏差平均手段220に出力する。そして、周波数偏差平均手段220は第1の搬送波周波数偏差推定値と第2の搬送波周波数偏差推定値の平均である周波数偏差平均値を算出する。
The first carrier frequency deviation estimation device 211 and the second carrier frequency deviation estimation device 212 calculate a first carrier frequency deviation estimation value and a second carrier frequency deviation estimation value, which are carrier frequency deviation estimation values, respectively. Output to the frequency deviation averaging means 220. Then, the frequency deviation averaging means 220 calculates a frequency deviation average value that is an average of the first carrier frequency deviation estimated value and the second carrier frequency deviation estimated value.
次に、本実施形態による搬送波周波数偏差推定方法について説明する。本実施形態の搬送波周波数偏差推定方法においては、第1の偏波入力信号と第2の偏波入力信号を入力信号とする。ここで、第1の偏波入力信号と第2の偏波入力信号は、中心周波数が同一周波数帯に配置され互いに直交する2個の単一偏波光信号を偏光多重した偏光多重光信号を検波することによって得られる。
Next, the carrier frequency deviation estimation method according to this embodiment will be described. In the carrier frequency deviation estimation method of this embodiment, the first polarization input signal and the second polarization input signal are used as input signals. Here, the first polarization input signal and the second polarization input signal are detected from a polarization multiplexed optical signal obtained by polarization multiplexing two single polarization optical signals that are arranged in the same frequency band and orthogonal to each other. It is obtained by doing.
そして、第1の偏波入力信号に対して搬送波周波数偏差推定値である第1の搬送波周波数偏差推定値を算出し、第2の偏波入力信号に対して搬送波周波数偏差推定値である第2の搬送波周波数偏差推定値を算出する。その後に、第1の搬送波周波数偏差推定値と第2の搬送波周波数偏差推定値の平均である周波数偏差平均値を算出する。
Then, a first carrier frequency deviation estimated value that is a carrier frequency deviation estimated value is calculated for the first polarization input signal, and a second carrier frequency deviation estimated value is calculated for the second polarization input signal. The estimated carrier frequency deviation is calculated. Thereafter, a frequency deviation average value that is an average of the first carrier frequency deviation estimated value and the second carrier frequency deviation estimated value is calculated.
上述したように、本実施形態の搬送波周波数偏差推定装置200および搬送波周波数偏差推定方法においては、第1の偏波入力信号と第2の偏波入力信号のそれぞれを用いて搬送波周波数偏差推定値を平均化した周波数偏差平均値を算出する構成としている。それにより、雑音による影響を削減することができるので、搬送波周波数偏差を推定する精度を向上させることができる。
As described above, in the carrier frequency deviation estimation apparatus 200 and the carrier frequency deviation estimation method of the present embodiment, the carrier frequency deviation estimation value is obtained using each of the first polarization input signal and the second polarization input signal. The averaged frequency deviation average value is calculated. Thereby, since the influence by noise can be reduced, the accuracy of estimating the carrier frequency deviation can be improved.
次に、本実施形態による搬送波周波数偏差補償装置300について図4を用いて説明する。
Next, the carrier frequency deviation compensating apparatus 300 according to the present embodiment will be described with reference to FIG.
搬送波周波数偏差補償装置300は、上述した搬送波周波数偏差推定装置200と周波数偏差補償手段を有する。周波数偏差補償手段は、搬送波周波数偏差推定装置200が算出した周波数偏差平均値に基づいて第1の偏波入力信号(X偏波入力信号)および第2の偏波入力信号(Y偏波入力信号)の搬送波周波数偏差を補償する。
The carrier frequency deviation compensating apparatus 300 includes the above-described carrier frequency deviation estimating apparatus 200 and frequency deviation compensating means. The frequency deviation compensation means is configured to output a first polarization input signal (X polarization input signal) and a second polarization input signal (Y polarization input signal) based on the average frequency deviation calculated by the carrier frequency deviation estimation apparatus 200. ) To compensate for the carrier frequency deviation.
搬送波周波数偏差補償装置300について、さらに詳細に説明する。周波数偏差補償手段は、周波数偏差平均値と単位サンプリング時間に基づいて入力信号に対する位相補償量を算出する位相補償量算出部と、入力信号の位相を位相補償量だけ回転させることにより搬送波周波数偏差を補償する位相補償部とを備える。なお、周波数偏差補償手段のX偏波入力信号およびY偏波入力信号に対する構成は同一である。
The carrier frequency deviation compensating apparatus 300 will be described in further detail. The frequency deviation compensation means includes a phase compensation amount calculation unit for calculating a phase compensation amount for the input signal based on the average frequency deviation value and a unit sampling time, and a carrier frequency deviation by rotating the phase of the input signal by the phase compensation amount. A phase compensation unit for compensation. The configuration of the frequency deviation compensating means for the X polarization input signal and the Y polarization input signal is the same.
具体的には、X偏波入力信号およびY偏波入力信号のうち、2分岐されたX偏波入力信号およびY偏波入力信号の一方をそれぞれ用いて、搬送波周波数偏差推定装置200が周波数偏差平均値を算出する。位相補償量算出部311(312)は周波数偏差平均値に基づいて位相補償量をそれぞれ算出する。X偏波入力信号およびY偏波入力信号の他方の信号は、位相補償量が算出されるまで位相補償部に含まれる遅延器321(322)においてそれぞれ待機される。位相補償量は周波数偏差平均値と単位サンプル時間の積の総和として算出される。遅延器321(322)において待機していたX偏波入力信号(Y偏波入力信号)は、算出された位相補償量の分だけ時計方向に位相回転されることにより搬送波周波数偏差がそれぞれ補償される。
Specifically, the carrier frequency deviation estimation apparatus 200 uses the one of the two branched X-polarization input signals and Y-polarization input signals of the X-polarization input signal and the Y-polarization input signal. The average value is calculated. The phase compensation amount calculation unit 311 (312) calculates the phase compensation amount based on the average frequency deviation value. The other signal of the X polarization input signal and the Y polarization input signal waits in the delay unit 321 (322) included in the phase compensation unit until the phase compensation amount is calculated. The phase compensation amount is calculated as the sum of products of the frequency deviation average value and the unit sample time. The X polarization input signal (Y polarization input signal) that has been waiting in the delay device 321 (322) is phase rotated clockwise by the calculated phase compensation amount, so that the carrier frequency deviation is compensated. The
上述した本実施形態の搬送波周波数偏差補償装置300によれば、搬送波周波数偏差推定装置200によって搬送波周波数偏差を推定する精度を向上させることができるので、周波数偏差を高精度で補償することが可能になる。これにより、光通信システムの伝送特性を改善することができる。
According to the carrier frequency deviation compensating apparatus 300 of the present embodiment described above, the accuracy of estimating the carrier frequency deviation by the carrier frequency deviation estimating apparatus 200 can be improved, so that the frequency deviation can be compensated with high accuracy. Become. Thereby, the transmission characteristic of an optical communication system can be improved.
以上、上述した実施形態を模範的な例として本発明を説明した。しかしながら、本発明は、上述した実施形態には限定されない。即ち、本発明は、本発明のスコープ内において、当業者が理解し得る様々な態様を適用することができる。
The present invention has been described above using the above-described embodiment as an exemplary example. However, the present invention is not limited to the above-described embodiment. That is, the present invention can apply various modes that can be understood by those skilled in the art within the scope of the present invention.
この出願は、2013年10月7日に出願された日本出願特願2013-210085を基礎とする優先権を主張し、その開示の全てをここに取り込む。
This application claims priority based on Japanese Patent Application No. 2013-210085 filed on Oct. 7, 2013, the entire disclosure of which is incorporated herein.
100、200 搬送波周波数偏差推定装置
110 リサンプリング手段
120 時間変動ベクトル算出手段
130 フィルタ手段
140 周波数偏差推定手段
141 偏角算出部
142 周波数偏差算出部
211 第1の搬送波周波数偏差推定装置
212 第2の搬送波周波数偏差推定装置
220 周波数偏差平均手段
300 搬送波周波数偏差補償装置
311、312 位相補償量算出部
321、322 遅延器
500 関連するディジタルコヒーレント方式を用いた光受信器
501 局所発振光生成部
510 90度ハイブリッド
521~524 光電変換部
531~534 ADコンバータ(ADC)
540 ディジタル信号処理部
551、552 シンボル識別部
611 X偏波信号生成部
612 Y偏波信号生成部
620、621、622 周波数偏差粗補償部
623 周波数偏差粗推定部
624 位相補償量算出部
625 遅延器
626 FFT部
627 データシフト部
628 IFFT部
631、632 波形歪み補償部
640 偏波分離部
651、652 リサンプリング部
660、661、662 周波数偏差補償部
663 周波数偏差推定部
671、672 位相偏差補償部 100, 200 Carrier frequencydeviation estimation device 110 Re-sampling means 120 Time variation vector calculation means 130 Filter means 140 Frequency deviation estimation means 141 Deflection angle calculation unit 142 Frequency deviation calculation unit 211 First carrier frequency deviation estimation device 212 Second carrier wave Frequency deviation estimator 220 Frequency deviation averaging means 300 Carrier frequency deviation compensator 311, 312 Phase compensation amount calculator 321, 322 Delay 500 Optical receiver using related digital coherent method 501 Local oscillation light generator 510 90 degree hybrid 521 to 524 Photoelectric conversion unit 531 to 534 AD converter (ADC)
540 Digital signal processing unit 551, 552 Symbol identification unit 611 X-polarization signal generation unit 612 Y-polarization signal generation unit 620, 621, 622 Frequency deviation rough compensation unit 623 Frequency deviation rough estimation unit 624 Phase compensation amount calculation unit 625 delay unit 626 FFT unit 627 Data shift unit 628 IFFT unit 631, 632 Waveform distortion compensation unit 640 Polarization separation unit 651, 652 Resampling unit 660, 661, 662 Frequency deviation compensation unit 663 Frequency deviation estimation unit 671, 672 Phase deviation compensation unit
110 リサンプリング手段
120 時間変動ベクトル算出手段
130 フィルタ手段
140 周波数偏差推定手段
141 偏角算出部
142 周波数偏差算出部
211 第1の搬送波周波数偏差推定装置
212 第2の搬送波周波数偏差推定装置
220 周波数偏差平均手段
300 搬送波周波数偏差補償装置
311、312 位相補償量算出部
321、322 遅延器
500 関連するディジタルコヒーレント方式を用いた光受信器
501 局所発振光生成部
510 90度ハイブリッド
521~524 光電変換部
531~534 ADコンバータ(ADC)
540 ディジタル信号処理部
551、552 シンボル識別部
611 X偏波信号生成部
612 Y偏波信号生成部
620、621、622 周波数偏差粗補償部
623 周波数偏差粗推定部
624 位相補償量算出部
625 遅延器
626 FFT部
627 データシフト部
628 IFFT部
631、632 波形歪み補償部
640 偏波分離部
651、652 リサンプリング部
660、661、662 周波数偏差補償部
663 周波数偏差推定部
671、672 位相偏差補償部 100, 200 Carrier frequency
540 Digital
Claims (10)
- 入力信号を、前記入力信号のシンボルレートの複数倍でオーバーサンプリングしたオーバーサンプリング信号に変換するリサンプリング手段と、
前記オーバーサンプリング信号に含まれる時間的に連続する2個のサンプル信号間の時間変動ベクトルを算出する時間変動ベクトル算出手段と、
前記時間変動ベクトルの平均値を算出するフィルタ手段と、
前記平均値に基づいて搬送波周波数偏差推定値を算出する周波数偏差推定手段、とを有する搬送波周波数偏差推定装置。 Resampling means for converting an input signal into an oversampling signal that is oversampled at a multiple of the symbol rate of the input signal;
A time variation vector calculating means for calculating a time variation vector between two temporally continuous sample signals included in the oversampling signal;
Filter means for calculating an average value of the time variation vectors;
A carrier frequency deviation estimator for calculating a carrier frequency deviation estimate based on the average value. - 請求項1に記載した搬送波周波数偏差推定装置において、
前記時間変動ベクトル算出手段は、
前記2個のサンプル信号のうち時間的に先行するサンプル信号を1サンプル時間だけ遅延させる遅延器と、
前記遅延器が出力するサンプル信号の複素共役を算出し複素共役サンプル信号を出力する複素共役器と、
前記2個のサンプル信号のうち後続のサンプル信号と、前記複素共役サンプル信号との積を算出する積算器、とを備える
搬送波周波数偏差推定装置。 In the carrier frequency deviation estimation apparatus according to claim 1,
The time variation vector calculating means includes:
A delayer for delaying a sample signal preceding in time among the two sample signals by one sample time;
Calculating a complex conjugate of the sample signal output by the delay unit and outputting a complex conjugate sample signal; and
A carrier frequency deviation estimating apparatus, comprising: an accumulator that calculates a product of a subsequent sample signal of the two sample signals and the complex conjugate sample signal. - 請求項1または2に記載した搬送波周波数偏差推定装置において、
前記周波数偏差推定手段は、
前記平均値から偏角を算出する偏角算出部と、
前記偏角と搬送波周波数偏差との関係に基づいて前記搬送波周波数偏差推定値を算出する周波数偏差算出部、とを備える
搬送波周波数偏差推定装置。 In the carrier frequency deviation estimation apparatus according to claim 1 or 2,
The frequency deviation estimating means includes
A declination calculator for calculating a declination from the average value;
A carrier frequency deviation estimation device comprising: a frequency deviation calculator that calculates the carrier frequency deviation estimate based on the relationship between the declination and the carrier frequency deviation. - 請求項1から3のいずれか一項に記載した搬送波周波数偏差推定装置である第1の搬送波周波数偏差推定装置と第2の搬送波周波数偏差推定装置と、周波数偏差平均手段、とを有し、
前記第1の搬送波周波数偏差推定装置と前記第2の搬送波周波数偏差推定装置はそれぞれ、中心周波数が同一周波数帯に配置され互いに直交する2個の単一偏波光信号を偏光多重した偏光多重光信号を検波することによって得られる第1の偏波入力信号と第2の偏波入力信号を前記入力信号とし、前記搬送波周波数偏差推定値である第1の搬送波周波数偏差推定値と第2の搬送波周波数偏差推定値をそれぞれ出力し、
前記周波数偏差平均手段は、前記第1の搬送波周波数偏差推定値と前記第2の搬送波周波数偏差推定値の平均である周波数偏差平均値を算出する
搬送波周波数偏差推定装置。 A first carrier frequency deviation estimation device, a second carrier frequency deviation estimation device, and a frequency deviation averaging means, which are the carrier frequency deviation estimation device according to any one of claims 1 to 3,
Each of the first carrier frequency deviation estimation device and the second carrier frequency deviation estimation device is a polarization multiplexed optical signal obtained by polarization multiplexing two single polarization optical signals that are arranged in the same frequency band and orthogonal to each other. The first polarized wave input signal and the second polarized wave input signal obtained by detecting the signal are used as the input signals, and the first carrier frequency deviation estimated value and the second carrier frequency which are the carrier frequency deviation estimated values are used. Output deviation estimates respectively
The frequency deviation averaging means calculates a frequency deviation average value that is an average of the first carrier frequency deviation estimated value and the second carrier frequency deviation estimated value. - 搬送波周波数偏差推定値を算出する搬送波周波数偏差推定装置と、前記搬送波周波数偏差推定値に基づいて入力信号の搬送波周波数偏差を補償する周波数偏差補償手段を有し、
前記搬送波周波数偏差推定装置は、
前記入力信号を、前記入力信号のシンボルレートの複数倍でオーバーサンプリングしたオーバーサンプリング信号に変換するリサンプリング手段と、
前記オーバーサンプリング信号に含まれる時間的に連続する2個のサンプル信号間の時間変動ベクトルを算出する時間変動ベクトル算出手段と、
前記時間変動ベクトルの平均値を算出するフィルタ手段と、
前記平均値に基づいて前記搬送波周波数偏差推定値を算出する周波数偏差推定手段、とを備える
搬送波周波数偏差補償装置。 A carrier frequency deviation estimating device for calculating a carrier frequency deviation estimated value, and a frequency deviation compensating means for compensating the carrier frequency deviation of the input signal based on the carrier frequency deviation estimated value,
The carrier frequency deviation estimating device is:
Resampling means for converting the input signal into an oversampling signal that is oversampled at a multiple of the symbol rate of the input signal;
A time variation vector calculating means for calculating a time variation vector between two temporally continuous sample signals included in the oversampling signal;
Filter means for calculating an average value of the time variation vectors;
A frequency deviation estimating means for calculating the carrier frequency deviation estimated value based on the average value. - 請求項5に記載した搬送波周波数偏差補償装置において、
前記周波数偏差補償手段は、
前記搬送波周波数偏差推定値と単位サンプリング時間に基づいて前記入力信号に対する位相補償量を算出する位相補償量算出部と、
前記入力信号の位相を前記位相補償量だけ回転させることにより、前記搬送波周波数偏差を補償する位相補償部、とを備える
搬送波周波数偏差補償装置。 In the carrier frequency deviation compensating apparatus according to claim 5,
The frequency deviation compensating means is
A phase compensation amount calculation unit for calculating a phase compensation amount for the input signal based on the carrier frequency deviation estimated value and a unit sampling time;
A carrier frequency deviation compensation device comprising: a phase compensation unit that compensates for the carrier frequency deviation by rotating the phase of the input signal by the phase compensation amount. - 請求項5に記載した搬送波周波数偏差補償装置において、
前記周波数偏差補償手段は、
前記入力信号に高速フーリエ変換処理を施すFFT部と、
前記FFT部が出力する高速フーリエ変換処理された入力信号を前記搬送波周波数偏差推定値だけ周波数シフトさせるデータシフト部と、
前記データシフト部の出力信号に逆フーリエ変換処理を施すIFFT部、とを備える
搬送波周波数偏差補償装置。 In the carrier frequency deviation compensating apparatus according to claim 5,
The frequency deviation compensating means is
An FFT unit for performing a fast Fourier transform on the input signal;
A data shift unit that shifts the frequency of the fast Fourier transform processed input signal output by the FFT unit by the carrier frequency deviation estimation value;
An IFFT unit that performs an inverse Fourier transform process on an output signal of the data shift unit. - 請求項5または6に記載した搬送波周波数偏差補償装置において、
前記搬送波周波数偏差推定装置である第1の搬送波周波数偏差推定装置と第2の搬送波周波数偏差推定装置と、周波数偏差平均手段、とを有し、
前記第1の搬送波周波数偏差推定装置と前記第2の搬送波周波数偏差推定装置はそれぞれ、中心周波数が同一周波数帯に配置され互いに直交する2個の単一偏波光信号を偏光多重した偏光多重光信号を検波することによって得られる第1の偏波入力信号と第2の偏波入力信号を前記入力信号とし、前記搬送波周波数偏差推定値である第1の搬送波周波数偏差推定値と第2の搬送波周波数偏差推定値をそれぞれ出力し、
前記周波数偏差平均手段は、前記第1の搬送波周波数偏差推定値と前記第2の搬送波周波数偏差推定値の平均である周波数偏差平均値を算出し、
前記周波数偏差補償手段は、前記周波数偏差平均値に基づいて前記第1の偏波入力信号および前記第2の偏波入力信号の搬送波周波数偏差を補償する
搬送波周波数偏差補償装置。 In the carrier frequency deviation compensating apparatus according to claim 5 or 6,
A first carrier frequency deviation estimation device, a second carrier frequency deviation estimation device, and a frequency deviation averaging means, which are the carrier frequency deviation estimation devices;
Each of the first carrier frequency deviation estimation device and the second carrier frequency deviation estimation device is a polarization multiplexed optical signal obtained by polarization multiplexing two single polarization optical signals that are arranged in the same frequency band and orthogonal to each other. The first polarized wave input signal and the second polarized wave input signal obtained by detecting the signal are used as the input signals, and the first carrier frequency deviation estimated value and the second carrier frequency which are the carrier frequency deviation estimated values are used. Output deviation estimates respectively
The frequency deviation averaging means calculates a frequency deviation average value that is an average of the first carrier frequency deviation estimated value and the second carrier frequency deviation estimated value;
The frequency deviation compensation means compensates carrier frequency deviations of the first polarization input signal and the second polarization input signal based on the frequency deviation average value. - 入力信号を、前記入力信号のシンボルレートの複数倍でオーバーサンプリングしたオーバーサンプリング信号に変換し、
前記オーバーサンプリング信号に含まれる時間的に連続する2個のサンプル信号間の時間変動ベクトルを算出し、
フィルタ処理により前記時間変動ベクトルの平均値を算出し、
前記平均値に基づいて搬送波周波数偏差推定値を算出する
搬送波周波数偏差推定方法。 The input signal is converted to an oversampling signal that is oversampled at a multiple of the symbol rate of the input signal,
Calculating a temporal variation vector between two temporally continuous sample signals included in the oversampling signal;
An average value of the time variation vector is calculated by filtering,
A carrier frequency deviation estimation method for calculating a carrier frequency deviation estimation value based on the average value. - 請求項9に記載した搬送波周波数偏差推定方法において、
中心周波数が同一周波数帯に配置され互いに直交する2個の単一偏波光信号を偏光多重した偏光多重光信号を検波することによって得られる第1の偏波入力信号と第2の偏波入力信号を前記入力信号とし、
前記搬送波周波数偏差推定値である第1の搬送波周波数偏差推定値を前記第1の偏波入力信号に対して算出し、
前記搬送波周波数偏差推定値である第2の搬送波周波数偏差推定値を前記第2の偏波入力信号に対して算出し、
前記第1の搬送波周波数偏差推定値と前記第2の搬送波周波数偏差推定値の平均である周波数偏差平均値を算出する
搬送波周波数偏差推定方法。 In the carrier frequency deviation estimation method according to claim 9,
A first polarization input signal and a second polarization input signal obtained by detecting a polarization multiplexed optical signal obtained by polarization multiplexing two single polarization optical signals whose center frequencies are arranged in the same frequency band and orthogonal to each other As the input signal,
Calculating a first carrier frequency deviation estimate that is the carrier frequency deviation estimate for the first polarization input signal;
Calculating a second carrier frequency deviation estimate which is the carrier frequency deviation estimate for the second polarization input signal;
A carrier frequency deviation estimation method that calculates a frequency deviation average value that is an average of the first carrier frequency deviation estimation value and the second carrier frequency deviation estimation value.
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EP4135277A4 (en) * | 2020-04-07 | 2024-05-22 | Sanechips Technology Co., Ltd. | Frequency offset estimation method and apparatus, electronic device, and computer-readable medium |
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