WO2015015190A1

WO2015015190A1 - Low noise and continuous emission led power supply circuit

Info

Publication number: WO2015015190A1
Application number: PCT/GB2014/052323
Authority: WO
Inventors: Robert Michael TRUNDLE
Original assignee: Trundle Robert Michael
Priority date: 2013-07-30
Filing date: 2014-07-29
Publication date: 2015-02-05
Also published as: GB2516696B; GB2516696A; GB201313623D0

Abstract

There is provided a linear power supply circuit operable to receive a rectified AC voltage supply having an instantaneous voltage which varies between a minimum zero-crossing value and peak value as a function of time. The linear power supply circuit comprises an LED arrangement having a minimum drive voltage above which supply current is operable to flow through said LED arrangement, a drive circuit section operable to control the supply current through the LED arrangement, a load ballast circuit arranged in parallel with said LED arrangement and operable to enable supply current to flow therethrough when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage such that the circuit is operable to provide current control across the whole voltage range of the AC voltage supply, and an energy storage circuit arranged in parallel with the LED arrangement, the energy storage circuit being operable to store energy when the instantaneous voltage across the LED arrangement is greater than or equal to the minimum drive voltage and to power the LED arrangement when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage.

Description

Low Noise and Continuous Emission LED Power Supply Circuit

The present invention relates to a power supply circuit. More preferably, the present invention relates to a power supply circuit for providing continuous LED lighting with low supply current distortions.

Increasingly, light emitting diodes (LEDs) are replacing conventional filament or fluorescent light sources in domestic and commercial lighting applications. LEDs have a low power consumption and a long lifetime. In addition, being solid-state devices, they have an intrinsic resistance to shock and damage. These, and other properties, make LEDs suitable for a variety of lighting applications.

As LED technology improves, LEDs will become more efficient. Therefore, the need to make LED drivers highly efficient to compensate for poor LED efficiency will be reduced. Consequently, the technological demands from the power supply will focus less on power supply efficiency and more on clean power management with reduced exported noise and supply current distortion. The combined effect of thousands of lower power loads (such as in lighting of aircraft cabins, big builds and thousands of homes) may cause a significant problem if exported noise, current harmonics and conducted emissions are not controlled. This may lead to unnecessarily larger power generators and supply cables to reduce current losses in the distribution of electricity.

One such application for clean power management is that of lighting for commercial or private aircraft. In general, commercial aircraft operate on AC power having a frequency which is generally in the range from under 300Hz to over 800Hz.

The requirements for low interference on commercial aircraft demand that onboard electronic devices have extremely low supply current harmonic distortion and conducted emissions, and unity power factor devices. This is to reduce the effect of interference on sensitive aircraft instrumentation. However, known arrangements for powering LEDs from an AC supply suffer from a number of technical problems which prevent such low emission and distortion standards from being reached. Figure 1 shows a conventional linear constant-current circuit 10 for an LED arrangement 12. The linear constant-current circuit 10 comprises an AC power source 14, a rectifier 16, a feedback resistor 18, a current control and detection transistor pair 20 and a current detection resistor 22.

One problem with such a circuit is that, for a considerable portion of a given drive cycle of the circuit, the LED arrangement will not be illuminated. Figure 2 illustrates the problem. An LED requires a particular threshold forward voltage drop V_f across it before it will illuminate. This is due to the bandgap of the semiconductor material from which the LED is formed. Below V_f the LED will not illuminate and no current will flow through the LED. Above V_f the current drawn by the LED is approximately exponentially dependent upon the voltage drop across the LED, limited by the current control and detection transistor pair 20. As shown in Figure 2, when a rectified AC voltage is used to power an LED (or a series array of LEDs, in which case V_f of the array will be the sum of the individual V_f values of the LEDs comprising the array), the LED arrangement is only lit when the voltage is above V_f. Therefore, for the remainder of a voltage cycle, the LED is unlit and no current flows therethrough, the supply current being turned off.

Therefore, the current in the circuit will not follow the voltage directly. This leads to generation of unwanted harmonics in the circuit which renders the circuit unsuitable for use in environments where low harmonics and conducted emissions are required, such as on board aircraft and, increasingly, in domestic and commercial lighting.

Attempts to improve the power efficiency of LED power circuits have been proposed. US-A-2011/0199003 discloses an arrangement whereby, during a power cycle, LED circuits are gradually added in to the circuit to create a stepped current draw EP-A-2 400 819 discloses a power saving LED lighting apparatus which switches LED strings in and out as the rectified mains voltage rises and falls. US-A-7,081 ,722 discloses an arrangement whereby LED strings are progressively switched in/out to improve power efficiency within a power cycle. US-A-201 1/0273103, US-A-2012/0001558 and WO-A- 2012/034102 disclose arrangements whereby strings of LEDs are switched in and out during a cycle. However, with improvements in LED technologies these arrangements to improve power efficiency will become less important.

However, such arrangements would be unsuitable for a commercial aircraft. The switching in/out of LEDs within a circuit generates additional harmonics and current distortion in the form of current spikes which renders this arrangement unsuitable for commercial aircraft, without additional filtering. However, additional filtering adds additional complexity to a power supply circuit and introduces additional current control problems associated with inductive elements.

Further, none of these arrangements is operable to control the supply current when the instantaneous supply voltage is so low that the lowest-voltage LED chain cannot be lit. Furthermore, the above disadvantages aside, switching of LEDs would be impractical for the long strip lighting used in aircraft, where such brightness variations would be readily apparent.

A further problem with the circuit of Figure 1 is that, for the portion of the voltage cycle during which the voltage is below V_f, no light is emitted from the LEDs. This causes an effect known as "pulsing" or "flicker", whereby the LEDs are seen to flicker at the frequency of the AC supply. In many applications, for example internal domestic lighting or commercial aircraft internal lighting, this flicker may be unacceptable. The flicker may be uncomfortable or problematic for individuals, or potentially hazardous for sufferers of medical conditions such as epilepsy. Lighting regulations now specify the level of acceptable flicker in particular circumstances.

It is known to address flicker in switched mode power configurations by storing the energy and delivering it during the AC off time. However, in linear applications that, for example, switch in additional LEDs from other parallel circuits, the power during the off time must be supplied by additional circuits such as, for example, valley fill circuits. However, such valley fill circuits introduce supply current distortion and supply harmonics which make them unsuitable for use in situations where low harmonic distortion is required. Therefore, a technical problem exists in the art that known solutions to improve the power efficiency of, and mitigate the flicker from, LED arrangements are impractical for many lighting purposes and are unable to reduce emissions and harmonic distortion to the required level for applications such as commercial aircraft or even

domestic/commercial lighting. In order to meet future emission requirements, the controlling supply harmonics and emissions of conducted noise will be of significant importance. The present invention addresses, in one aspect, the above issues.

According to a first aspect of the present invention, there is provided a linear power supply circuit operable to receive a rectified AC voltage supply having an instantaneous voltage which varies between a minimum zero-crossing value and peak value as a function of time, the linear power supply circuit being operable to power an LED arrangement continuously throughout an AC voltage cycle and comprising: an LED arrangement having a minimum drive voltage above which supply current is operable to flow through said LED arrangement; a drive circuit section operable to control the supply current through the LED arrangement; a load ballast circuit arranged in parallel with said LED arrangement and operable to enable supply current to flow therethrough when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage such that the circuit is operable to provide current control across an entire power cycle of the AC voltage supply and such that the current is controlled across the whole instantaneous voltage range of the AC voltage supply between the zero crossing point to the maximum instantaneous voltage; and an energy storage circuit arranged in parallel with the LED arrangement, the energy storage circuit being operable to store energy when the instantaneous voltage across the LED arrangement is greater than or equal to the minimum drive voltage and to power the LED arrangement when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage.

By providing such an arrangement, an additional load ballast circuit introduced to a linear circuit design is operable to drain the supply current when the instantaneous mains AC voltage is insufficient to drive the load. This forces the remaining supply current to follow the supply voltage curve, creating a simple unity power factor power supply with extremely low supply current distortion. The present invention thus provides a power supply which is operable at frequencies ranging from DC to over 800 Hz, and from low to high voltages. Further, the linear circuit is operable to drive the LED arrangement continuously throughout an AC voltage cycle.

In one embodiment, the drive circuit section comprises an operational amplifier operable to control the supply current and the load ballast circuit comprises switching means to enable current to flow through the load ballast circuit when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage of said LED arrangement. In one embodiment, the switching means are switched in dependence upon the output of the operational amplifier.

In one embodiment, the switching means comprises a diode. In one embodiment, the load ballast circuit comprises at least one field effect transistor. In one embodiment, the drive circuit section comprises a field effect transistor. In one embodiment, the drive circuit section comprises a transistor pair.

In one embodiment, the drive circuit section comprises a transistor pair including a MOSFET and a bipolar junction transistor.

In one embodiment, the transistor pair comprises a MOSFET and a bipolar junction transistor.

In one embodiment, the energy storage circuit comprises a capacitive element.

In one embodiment, the energy storage circuit further comprises a resistive or current limiting element in series with the capacitive element.

In one embodiment, the capacitor is operable to charge when the instantaneous voltage across the LED arrangement is greater than or equal to the minimum drive voltage and is arranged to discharge through the LED arrangement when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage.

In one embodiment, the energy storage circuit is connected in parallel with the MOSFET and LED Load.

In another embodiment, the energy storage circuit is connected in parallel with the load, with or without a local current regulating device (which may simply be a Resistor) In one embodiment, the LED arrangement comprises a single LED or plurality of LEDs arranged in series connection. The LEDs may be high voltage LEDs.

In one embodiment, the circuit further comprises a rectifier connectable to an AC power source and operable to generate the rectified AC voltage signal therefrom.

In one embodiment, the AC power source comprises an aircraft power supply.

In one embodiment, the AC power source has a frequency in the range of 300 - 800 Hz. In one embodiment, the AC power source has a RMS voltage of 115V.

According to a second aspect of the present invention, there is provided an aircraft lighting apparatus comprising the circuit of the first aspect. In one embodiment, there is provided a power supply network comprising an AC power source connected to a multiplicity of power supply circuits according to the first aspect.

According to a third aspect of the present invention, there is provided a method of controlling the supply current in a linear power supply circuit for powering an LED arrangement continuously through an AC voltage cycle, the method comprising:

receiving a rectified AC voltage supply having an instantaneous voltage which varies between a minimum zero-crossing value and peak value as a function of time; driving an LED arrangement with said rectified AC voltage supply, the LED arrangement having a minimum drive voltage above which supply current is operable to flow through said LED arrangement; charging an energy storage circuit arranged in parallel with the LED arrangement during the period when the instantaneous voltage is equal to or greater than the minimum drive voltage; driving an LED arrangement with charge from said energy storage circuit during the period when the instantaneous voltage is less than the minimum drive voltage; and controlling the supply current by: utilising a drive circuit section operable to control the supply current when the AC voltage drop across the LED arrangement is equal to or greater than the minimum drive voltage; and switching to a load ballast circuit arranged in parallel with said LED arrangement and operable to enable supply current to flow therethrough when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage such that the circuit is operable to provide current control across an entire power cycle of the AC voltage supply and such that the current is controlled across the whole instantaneous voltage range of the AC voltage supply between the zero crossing point to the maximum instantaneous voltage.

In one embodiment, the drive circuit section comprises an operational amplifier operable to control the supply current and the load ballast circuit comprises switching means to enable current to flow through the load ballast circuit when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage of said LED arrangement.

In one embodiment, the switching means are switched in dependence upon the output of the operational amplifier.

In one embodiment, the switching means comprises a diode. In one embodiment, the switching means comprises a Zener diode. In one embodiment, the load ballast circuit comprises at least one field effect transistor. In one embodiment, the drive circuit section comprises a field effect transistor. In one embodiment, the drive circuit section comprises a transistor pair.

In one embodiment, the energy storage circuit comprises a capacitive element and a resistive or current limiting element in series.

In one embodiment, the energy storage circuit is connected in parallel with the MOSFET.

In one embodiment, the LED arrangement comprises a single high voltage LED or a plurality of LEDs arranged in series connection.

In one embodiment, the AC power source comprises an aircraft power supply.

In one embodiment, the AC power source has a frequency in the range of 300 - 800 Hz. In one embodiment, the AC power source has a RMS voltage of 1 15V.

Embodiments of the present invention will now be described in detail with reference to the accompanying drawings, in which:

Figure 1 is a schematic diagram of a known linear power circuit;

Figure 2 is a schematic diagram of a rectified AC voltage signal illustrating the conduction cycle of a driven LED;

Figure 3 shows a circuit according to an embodiment of the present invention; Figure 4 shows a schematic diagram similar to Figure 2 but illustrating the two regimes of operation of the circuit of Figure 3;

Figure 5 shows an example of the total input power to the circuit of Figure 3 across a half cycle;

Figure 6a) shows the current loss across an LED arrangement across the half cycle shown in Figure 5 and for the same input power; Figure 6b) shows the current loss across a load ballast circuit across the half cycle shown in Figure 5 and for the same input power;

Figure 7a) shows the power output from the LED arrangement across a half cycle and for the same input power;

Figure 7b) shows the data of Figures 5, 6a), 6b) and 7a) superimposed; and Figure 8 shows a circuit according to a second embodiment of the present invention. The present invention relates to a linear circuit operable to force the supply current to follow the supply voltage wave shape and frequency. In one embodiment, this is implemented utilising a circuit which does not require expensive filtering. This circuit full fills the requirements specifically for but not limited to aircraft applications where in many cases the supply frequency can vary from as low as 300Hz and up to over 800Hz whilst requiring very low supply current harmonic distortion and very low Conducted Emissions. The circuit is operable down to DC power and thus can cover a range of frequency and voltage requirements. The circuit is, additionally and synergistically, operable to provide continuous light emission from LED arrangement(s), mitigating flicker whilst providing an overall energy efficient circuit with low harmonic distortion.

Figure 3 shows an embodiment of the present invention. A linear power supply circuit 100 is provided. The linear power supply circuit 100 is operable to receive power from an AC supply 102. In this embodiment, the AC supply comprises an aircraft supply of 1 15V 400Hz AC.

A rectifier 104 is provided to rectify the received AC supply 102. A first resistor is located between the positive voltage rail (shown by +V on Figure 3) and Ground (0V). Resistor Ri is operable to provide a minimum load to the circuit during the time when the remainder of the circuit 100 is inactive at very low voltages (in this embodiment, below about 1.5V). Resistor R₂ and Zener diode are also provided in series between the positive voltage rail and ground in parallel with capacitor to provide a fixed micro-power supply for the Op Amp Ai and the bias for the gate of FETs ΤΊ and T₂. In other words, capacitor Ci is used to hold up the 10V supply for the micro-power Op Amp Ai and to provide the bias for the gate of the FETs ΤΊ and T₂.

Resistors R₃ and R₄ are also provided in series between the positive voltage rail and ground and provide the reference signal for the Op Amp A^ In other words, resistors R₃ and R₄ attenuate the rectified rail voltage to derive the signal for the Op Amp A^ . The circuit 100 further comprises a load section 106, a constant current driver section 108, a load ballast circuit section 1 10 and an energy storage section 1 12. In this embodiment, the load section 106 comprises a plurality of LEDs arranged in series (or a string). However, other loads may be used; for example, a single LED or alternative constant current devices.

The constant current driver section 108 comprises an operational amplifier and two transistors and T₂. In this embodiment, comprises an n-channel MOSFET, and T₂ comprises a bipolar junction transistor. Resistors R₅ and R₆ are provided to ensure an equal load is provided to the gate of transistor T₂.

The load ballast circuit section 1 10 is arranged in parallel with the load section 106. In this embodiment, the load ballast circuit section 1 10 comprises a diode D1 , and two transistors T₃ and T₄. In this embodiment, T₃ comprises an n-channel MOSFET, and T₄ comprises a bipolar junction transistor (BJT). Resistors R₇ and R₈ are provided to ensure an equal load is provided to the gate of transistor T₂.

The non-inverting operational amplifier AT and the respective gates of the FETs and T₃ are powered by a reference voltage \Λ provided from the differential of the voltage drop across resistors R₃ and R₄ (the amplifier voltage being taken across R₄) . Resistor R₂, Zener diode and capacitor ensure that the supply received is a stable DC power supply (which is 10V in this embodiment). Operational amplifier AT acts to maintain the voltage V₂ across resistor R₉ equal to voltage \ . Therefore, in a case where V₂ < V A-_\ will generate a greater voltage at the gate of transistor T₂, increasing the collector-emitter current of T₂ flowing from the positive rail through the load section 106. Concomitantly, when V₂≥V_{l t} no bias is applied to the gate of transistor T₂ and so no current flows therethrough.

Since the gate of is powered by a constant DC source, the MOSFET is always open and acts as a current-limiting device in the circuit, controlled by the action of T₂ on the source of Ti . Transistor Ti is also a voltage level shifter to operate over an instantaneous voltage up to 1000V.

In alternative embodiments, further controlling elements may be added; for example, a modified voltage reference Vi to stabilise the current in R₉ against variations in supply RMS voltage. The energy storage section 1 12 is arranged in parallel with the load section 106 and transistor Ti and comprises a capacitor C₂ and the resistor R₁₀ connected in parallel with the load section 106 and transistor Ti . The energy storage section 1 12 is arranged to charge during a cycle such that the LED arrangement can be lit during the portion of the cycle when the supply voltage alone is too low to light the LEDs. In other words, the capacitor C₂ is operable, in use, to charge through resistor R₁₀ during a voltage cycle and to supply that charge to the load section 106 during the remainder of the voltage cycle. Whilst the values of the capacitance and resistance may be varied depending upon the application, typical examples may involve a capacitor C2 having a capacitance of 1 -2 \JF and resistor R10 having a resistance of approximately 5 ΚΩ Further, diode D₂ is also provided in series with the load section 106 and is arranged to prevent the charge stored on Capacitor C₂ from discharging through Transistors T₃ and T₄ . This is to be prevented because this would distort the supply sine wave current, reducing the harmonic efficiency of the circuit. As shown in Figure 3, the components of the energy storage section 1 12 are also arranged in parallel with transistor ^ . The gate of transistor ^ is powered by a constant DC source, and the source of transistor Ti is held low by transistor T₂ during the time there is insufficient supply voltage to drive the LEDs of the load arrangement 106.

Therefore transistor is held ON and also able to function as a current path when the stored charge across capacitor C₂ is used to power the LEDs of the load section 106. The current in the LEDs is now simply limited by the charge on capacitor C₂ and the value of resistor R₁₀.

An advantage of the arrangement of the energy storage section 1 12 is that the capacitor C₂ is charged to a higher voltage than that of the LED arrangement of the load section 106. Therefore, the capacitor C₂ is operable to discharge through the LED arrangement in a controlled manner. Thus, by bypassing the transistor Ti controlling the current in the LED arrangement with the parallel capacitor C₂ /resistor R₁₀ path to the BJT T₂, the BJT T₂ therefore controls the combination of the current through the LED arrangement and the charge current through the capacitor C₂.

In summary, the energy storing circuit 1 12 is operable, in embodiments, to achieve the following functions: to store energy in capacitor C₂ during the time transistor is conducting; to use the stored energy in C₂ to continue driving the LED arrangement during the "off time" between the AC supply cycles; and to improve the efficiency of the circuit by reducing the on time current through transistor Ti . The circuit 100 is operable to provide a supply current within the circuit that follows exactly the supply voltage within the circuit. Therefore, whilst conventional circuits provide an LED current that follows the supply current during the conduction time of the LEDs (i.e. during the shaded portion shown in Figure 2), the present invention enables the supply current to follow the supply voltage throughout a power cycle, resulting in a unity power factor circuit with low noise, low emissions and low harmonics. Further, because there is no requirement to switch LEDs in and out of the circuit, no current spikes (which generate harmonics) will result. The operation of the circuit 100 according to an embodiment of the present invention will now be described with reference to Figures 3 and 4.

Similarly to Figure 2, Figure 4 shows a graph of instantaneous voltage across the load section 106 in the circuit 100 as a function of time. Each cycle is subdivided into two regions - region A and region B. Region A is when the instantaneous voltage is equal to or greater than the forward voltage V_f of the load section 106 (in this case, a series array of LEDs, in which case the forward voltage V_f is the sum of the forward voltages of each of the LEDs in the string). In other words, region A covers the voltage range between V_f and the maximum (or peak) voltage V_max. In region A, the LEDs are directly powered by the AC supply voltage. During this time, the capacitor C₂ of the energy storage section 1 12 also charges directly from the AC voltage supply.

Region B is when the instantaneous voltage is insufficient to drive the load section 106 by means of the supply voltage alone. This is when the instantaneous voltage across the load section 106 is in the range from the zero crossing minimum value up to but not including the forward voltage V_f. Consequently, the LEDs forming the load section 106 cannot be powered by the AC supply. However, during this region of the voltage cycle, the LEDs can be powered by the energy storage section 1 12 as will be described below. Consider first region A. When the instantaneous voltage across the load section 106 is equal to, or greater than, the forward voltage V_f of the load section 106, current flows through the load section 106, through transistor (which is in an "on" state by virtue of the gate of being biased at voltage \Λ) which acts as a current limiting device to the collector of BJT T₂. The base of BJT T₂ is biased by the output of operational amplifier At and controls the current flow through T₂ in order to maintain V₂ = \Λ.

Further, during region A of the voltage cycle (in which the instantaneous voltage is equal to or greater than the forward voltage V_f of the load section 106) the current is split between two paths - a first path through the LEDs of the load section 106 as described above and a second path through the capacitor C₂ and resistor R₁₀ of the energy storage section 1 12. As a result, in region A, current is supplied both to the LEDs of the load section 106 (which will emit light) and to the energy storage section 1 12 to store charge on the capacitor do which is charged through resistor Ri₀.

At the transition between regions A and B, when the instantaneous voltage drops below V_f, no current will flow through the load section 106, and through Jt and T₂. This will cause V₂ to drop. In other words, as the LED current falls the ability of the BJT T₂ to supply the current in resistor R9 to maintain V₂= \A is diminished. Consequently, the output of Operational Amplifier At increases to increase the current flow to the base of BJT T₂. In a conventional arrangement, this will make no difference since the current flowing from the positive rail through the load section 106 is small or zero.

However, by provision of the load ballast section 1 10, the circuit 100 of the present invention enables current to flow to resistor R₉ to increase V₂ seamlessly as follows. As the voltage output of operational amplifier AT increases, the diode Dt will be biased on. This will enable current to flow to the base of BJT T₄, which is switched on. When BJT T₄ switches on, current flows from the positive rail, through current limiting MOSFET T₃ and the collector-emitter region of T₄, to resistor R₉. In other words, in region B, transistors T₃ and T₄ are operable to maintain the current in R₉ when the supply voltage is too low to power the load section 106 (i.e. the LEDs). The operational amplifier At therefore continues to control the current via T₄ and T₃ through the remainder of region B until the supply voltage rises again during the next half cycle. At the same time, when the voltage cycle is in region B and the supply voltage falls and is not sufficient to drive the LEDs of the load section 106, the arrangement of the energy storage section 112 in parallel with the load section 106 will cause the capacitor do to discharge via resistor R₁₀ and through transistor T The diode D₂ prevents the capacitor C₂ from being discharged through transistors T₃ and T₄ as discussed.

As discussed, the voltage across the capacitor C₂ in region B of the voltage cycle will be higher than that across the LED arrangement because the capacitor voltage also includes the MOSFET ΤΊ voltage. This is because the energy storage section 1 12 bypasses the transistor ΤΊ and connects to the collector of the BJT T₂, enabling the capacitor C₂ to discharge slowly into the LED arrangement.

At this stage, as set out above, transistor ΤΊ is now held fully on by transistor T₂.

Therefore, current is supplied to the LEDs of the load section 106 and, as a result, the LEDs of the load section will continue to emit light during region B of the voltage cycle. However, due to the arrangement of the circuit in which the charge is drawn from the capacitor C₂ by the load section 106, sufficient current does not flow through transistor T₂ and so V₂ does not increase materially. When the instantaneous voltage is equal to the forward voltage V_f of the load section 106 (i.e. at the start of region A of the next cycle), then current will flow through ΤΊ and T₂, resulting in an increase in V₂. Concomitantly, the output of the operational amplifier A_t will be reduced, switching off diode Dv In other words, in region A, the combination of ΤΊ and T₂ start to supply current to resistor R₉ again. The process then reverts to the original condition of driving through the LEDs enabling the LED current to rise and fall in line with the instantaneous supply voltage again. Also during this period, the energy storage section 1 12 will begin to store charge to replenish the charge dissipated from the capacitor C₂ during driving of the LEDs in region B of the voltage cycle. As set out above, the change over between the regimes of region A and region B is achieved by the use of Diode Di between the controlling element for the load section 106 (i.e. ΤΊ and T₂) and the controlling elements (i.e. T₃ and T₄) of the load ballast section 1 10. As an additional benefit, the use of the MOSFETs ΤΊ and T₃ enables the voltage range of the current controlling Bipolar Transistors T₂ and T₄ to be increased.

In summary, the additional circuit (the load ballast section 1 10) provided in parallel with the LEDs is operable to take up the supply current waveform when the supply voltage is insufficient to power the LEDs. The load ballast section 1 10 enables the circuit 100 to control the current in the circuit right down to the zero crossing minimum voltage. This reduces or even eliminates harmonic distortion and noise, and power efficiency is concomitantly increased. The provision of the arrangement of the present invention enables this transition to occur inherently and elegantly as the LED current falls off. This, therefore, creates a near seamless transition between the two elements and will happen even as the LED forward voltage drifts.

In other words, because the transition is controlled by the output from the operational amplifier AT and the switch on voltage of the diode D^ the transition is independent of V_f for the load section 106. Consequently, the same circuit can be utilised with different load section 106 components; for example, one or more LEDs; LEDs of different colour; or alternative components. Further, this can be achieved whilst also providing continuous light emission throughout a voltage cycle. In contrast to known arrangements, the present circuit can provide continuous light emission from LEDs driven by an AC voltage supply with little or no harmonic distortion in a single, straightforward circuit design. Figures 5 to 7 illustrate examples of the operational parameters of the circuit 100. For simplicity, these parameters have been calculated without the presence of the energy storage section 1 12. They are intended to provide examples of the benefits of the load ballast section 1 10 in terms of energy efficiency and low harmonic distortion. Figure 5 shows a typical input power into the circuit 100 across a half cycle (i.e. from a given zero-crossing point to the next zero-crossing point). In this example, the average total input power is 9.15 Watts. Figures 6a) and 6b) show the power loss (in Watts) across the half cycle shown in Figure 5. Figure 6a) shows the power loss (in Watts) due to the LED arrangement. There is no power loss when the instantaneous voltage is below V_f (i.e. in region B shown in Figure 4). However, in region A (shown in Figure 4), the LED arrangement is switched on and current can flow. In this region, the magnitude of the power loss is, of course, a function of the magnitude of the instantaneous power shown in Figure 5.

Figure 6b) shows the power loss due to the load ballast section 1 10. Given the nature of the switching as described above, current only flows through the load ballast section 110 when the instantaneous voltage is below V_f (i.e. in region B shown in Figure 4).

Figure 6b) clearly shows the power loss in the load ballast section 110 being right down to the zero crossing point, indicating that the load ballast section 1 10 is dissipating current in the circuit 100 throughout the entire portion of a half cycle in which the instantaneous voltage is below V_f and the LED arrangement is not conducting current. This demonstrates how current control is implemented across an entire power cycle utilising the present invention. In this example, the average power loss across the load ballast section 1 10 is 1.61 Watts. Figure 7a) shows the power output from the LED arrangement, In this example, the total power from the LED is 5.81 Watts. Figure 7b) shows a combined power graph. Figure 7b) thus includes the data from Figures 5, 6a), 6b) and 7a). From this data, the overall power efficiency of the arrangement can be determined and this calculated to be approximately 64%.

However, this value is dependent upon the efficiency of the LED arrangement and this, with development of more efficient LED technology, will rise. Further, by optimising the peak magnitude of the instantaneous Voltage and V_f of the LEDs, the period within a given half cycle for which the LED arrangement is unlit can be shortened. In summary, power efficiency can be improved in numerous ways known in the art. However, the more crucial issue addressed by the present invention is to provide a circuit which is operable to significantly reduce or eliminate harmonic distortions due to a lack of current control across the entire power cycle. Figure 8 shows a second embodiment of the present invention. A linear power supply circuit 200 is provided. The linear power circuit 200 is operable to function in the same manner as the linear power supply circuit 100 of the first embodiment. However, the linear power circuit 200 has a configuration which enables certain components of the linear power circuit 100 to be eliminated. Individual components which serve a function in common with those of the first embodiment have been given the same reference signs for clarity of understanding. The linear power supply circuit 200 is operable to receive power from an AC supply 202. In this embodiment, the AC supply comprises an aircraft supply of 1 15V 400Hz AC.

A rectifier 204 is provided to rectify the received AC supply 202. As for the first embodiment, a first resistor is located between the positive voltage rail (shown by +V on Figure 8) and Ground (0V). Resistor is operable to provide a minimum load to the circuit during the time when the remainder of the circuit 200 is inactive at very low voltages (in this embodiment, below about 1.5V).

As for the first embodiment, Resistor R₂ and Zener diode are also provided in series between the positive voltage rail and ground in parallel with capacitor C-_\ to provide a fixed micro-power supply for the Op Amp A^ In other words, capacitor C-_\ is used to hold up the 10V supply for the micro-power Op Amp Ai .

In common with the first embodiment, resistors R₃ and R₄ are also provided in series between the positive voltage rail and ground and provide the reference signal for the Op Amp AL In other words, resistors R₃ and R₄ attenuate the rectified rail voltage to derive the signal for the Op Amp Α .

The circuit 200 further comprises a load section 206, a constant current driver section 208, a load ballast circuit section 210 and an energy storage section 212. In this embodiment, the load section 206 comprises a plurality of LEDs arranged in series (or a string). However, other loads may be used; for example, a single LED or alternative constant current devices. The constant current driver section 208 comprises an operational amplifier At as for the first embodiment. However, in the second embodiment, a single FET T₅ replaces the pair of transistors Jt and T₂ of the first embodiment. In other words, a single FET fulfils the dual role of current limitation and selective voltage switching which were the functions of separate transistors Jt and T₂ of the first embodiment respectively.

In the second embodiment, T₅ comprises an n-channel MOSFET having a low gate voltage threshold in the region of 2.5 - 4V. Resistors R₅ and R₆ are provided to ensure an equal load is provided to the gate of transistor T₅. R₆ provides a load to the output of the Op Amp At since this cannot adequately be provided by the FETs alone. R₅ is arranged to limit any fault currents at the input to the Op Amp Ai .

The load ballast circuit section 210 is arranged in parallel with the load section 206. In contrast to the load ballast circuit 1 10 of the first embodiment, the load ballast circuit section 210 comprises a Zener diode Z₂, and a single FET T₆ in place of the two transistors of the first embodiment. In this embodiment, T₆ comprises an n-channel MOSFET having a low gate voltage threshold in the region of 2.5 - 4V. Resistor R₇ is provided to ensure an equal load is provided to make sure that the gate charge of transistor T₆ is removed when the Zener diode Z₂ stops conducting.

The non-inverting input to the operational amplifier Aiis fed by a reference voltage Vi provided from the differential of the voltage drop across resistors R₃ and R₄ (the amplifier input reference voltage being taken across R₄) . Resistor R₂, Zener diode∑t and capacitor Ct ensure that the supply voltage to the operational amplifier is a stable DC power supply (which is also 10V in this embodiment).

As for the first embodiment, operational amplifier AT acts to maintain the voltage V₂ across resistor R₉ equal to voltage \ . Therefore, in a case where V₂ < V At will generate a greater voltage at the gate of transistor T₅, increasing the source-drain current of T₅ flowing from the positive rail through the load section 206. Concomitantly, when V₂≥ Vi, no bias is applied to the gate of transistor T₅ and so no current flows therethrough. The energy storage section 212 is arranged in parallel with the load section 206 and comprises a capacitor C₂ connected in series with a local current controlling device such as a resistor in parallel with the load section 206 . Local series current control (such as, for example, resistor R₁₀ in the first embodiment) is not necessarily required in the energy storage section 212.

The energy storage section 212 is arranged to charge during a cycle such that the LED arrangement can be lit during the portion of the cycle when the supply voltage alone is too low to light the LEDs. In other words, the capacitor C₂ is operable, in use, to charge during a voltage cycle and to supply that charge to the load section 206 during the remainder of the voltage cycle.

Whilst the values of the capacitance and resistance may be varied depending upon the application, typical examples may involve a capacitor C2 having a capacitance of 100- 200 iF.

Further, diode D₂ is also provided in series with the load section 206 and is arranged to prevent the charge stored on Capacitor C₂ from discharging through Transistors T₅ and T₆. This is to be prevented because this would distort the supply sine wave current, reducing the harmonic efficiency of the circuit. Additionally, diodes D₃ and D₄ are provided in the constant current driver section 208 to prevent the capacitor C₂ discharging through transistor T₆. As shown in Figure 8, the energy storage section 212 is arranged in parallel with load section 206. During the time there is insufficient supply voltage to drive the LEDs of the load arrangement 206 (where V₂ < V-,). Energy storage section 212 will discharge through the load section 206 maintaining current flow therethrough. The current in the LEDs is now simply limited by the charge on capacitor C₂.

An advantage of this arrangement is that it uses fewer components and uses less power in the control sections, thus improving efficiency. Also that in the energy storage section the capacitor can be selected to maintain a minimum voltage drop during the time it is powering the LEDs, which can be achieved because the Vf of the LED / LED string (i.e. load section 206) reduces slightly as the current falls so the capacitor C₂, through appropriate selection, can maintain conduction in the LED/LED String of the load section 206. .

In summary, the energy storing circuit 212 is operable, in embodiments, to achieve the following functions: to store energy in capacitor C₂ during the time transistor ΤΊ is conducting; and to use the stored energy in C₂ to continue driving the LED arrangement during the "off time" between the AC supply cycles. The circuit 200 is operable to provide a supply current within the circuit that follows exactly the supply voltage within the circuit. Therefore, whilst conventional circuits provide an LED current that follows the supply current during the conduction time of the LEDs (i.e. during the shaded portion shown in Figure 2), the present invention enables the supply current to follow the supply voltage throughout a power cycle, resulting in a unity power factor circuit with low noise, low emissions and low harmonics. Further, because there is no requirement to switch LEDs in and out of the circuit, no current spikes (which generate harmonics) will result.

The operation of the circuit 200 according to the second embodiment of the present invention will now be described with reference to Figure 4.

Consider first region A as shown in Figure 4. When the instantaneous voltage across the load section 206 is equal to, or greater than, the forward voltage V_f of the load section 206 current flows through the load section 206, through transistor T₅. Transistor T₅ is in an "on" state by virtue of the gate of T₅ being biased on by the operational amplifier At to control the current flow through T₅ in order to maintain V₂ = \Λ.

Further, during region A of the voltage cycle (in which the instantaneous voltage is equal to or greater than the forward voltage V_f of the load section 206) the current is split between two paths - a first path through the LEDs of the load section 206 as described above and a second path through the capacitor C₂ of the energy storage section 212. As a result, in region A, current is supplied both to the LEDs of the load section 206 (which will emit light) and to the energy storage section 212 to store charge on the capacitor C₂. At the transition between regions A and B, when the instantaneous voltage drops below V_f, no current will flow through the load section 206, and through T₅. This will cause V₂ to drop. In other words, as the LED current falls the ability of the FET T₅ to supply the current in resistor R₉ to maintain V₂= \A is diminished. Consequently, the output of Operational Amplifier AT increases to increase the current flow to the gate of FET T₅. In a conventional arrangement, this will make no difference since the current flowing from the positive rail through the load section 206 is small or zero.

However, by provision of the load ballast section 210, the circuit 200 of the present invention enables current to flow to resistor R₉ to increase V₂ seamlessly as follows.

As the voltage output of operational amplifier AT increases, the breakdown voltage of the Zener diode Z₂ will be reached and current will flow. This will enable current to flow to the gate of FET T₆, which is switched on. When FET T₆ switches on, current flows from the positive rail, through the source-drain region of T₆, to resistor R₉. In other words, in region B, transistor T₆ is operable to maintain the current in R₉ when the supply voltage is too low to power the load section 206 (i.e. the LEDs). The operational amplifier AT therefore continues to control the current via T₆ through the remainder of region B until the supply voltage rises again during the next half cycle. At the same time, when the voltage cycle is in region B and the supply voltage falls and is not sufficient to drive the LEDs of the load section 206, the arrangement of the energy storage section 212 in parallel with the load section 206 will cause the capacitor C₂ to discharge via through the local current control element or resistor if utilised. The diode D₂ prevents the capacitor C₂ from being discharged through transistor T₆ as discussed.

As discussed the charge on the capacitor C₂ in the region B of the voltage cycle will be sufficient to maintain conduction through the load section 206 with minimal voltage drop because the Vf of the LEDs of the load section 206 fall slightly as the current through them drops. This feature is used advantageously to maintain reasonable conduction through the LED load section 206 during region B of the supply voltage cycle, when capacitor C2 discharges slightly through the load. At this stage, as set out above, current is supplied to the LEDs of the load section 206 and, as a result, the LEDs of the load section 206 will continue to emit light during region B of the voltage cycle. However, due to the arrangement of the circuit in which the charge is drawn from the capacitor C₂ by the load section 206, sufficient current does not flow through transistor T₅ and so V₂ does not increase materially.

When the instantaneous voltage is equal to the forward voltage V_f of the load section 206 (i.e. at the start of region A of the next cycle), then current will flow through T₅, resulting in an increase in V₂. Concomitantly, the output of the operational amplifier Ai will be reduced, switching Zener diode Z₂ below the breakdown voltage. In otherwords, in region A, T₅ starts to supply current to resistor R₉ again. The process then reverts to the original condition of driving through the LEDs enabling the LED current to rise and fall in line with the instantaneous supply voltage again. Also during this period, the energy storage section 212 will begin to store charge to replenish the charge dissipated from the capacitor C₂ during driving of the LEDs in region B of the voltage cycle.

As set out above, the change-over between the regimes of region A and region B is achieved by the use of Zener Diode Z₂ between the controlling element for the load section 206 (i.e. T₅) and the controlling element (i.e. T₆) of the load ballast section 210. In summary, the additional circuit (the load ballast section 210) provided in parallel with the LEDs is operable to take up the supply current waveform when the supply voltage is insufficient to power the LEDs. The load ballast section 210 enables the circuit 200 to control the current in the circuit right down to the zero crossing minimum voltage. This reduces or even eliminates harmonic distortion and noise, and power efficiency is concomitantly increased. The provision of the arrangement of the present invention enables this transition to occur inherently and elegantly as the LED current falls off. This, therefore, creates a near seamless transition between the two elements and will happen even as the LED forward voltage drifts. In other words, because the transition is controlled by the output from the operational amplifier Ai and the breakdown voltage of the Zener diode Z₂, the transition is independent of V_f for the load section 206. Consequently, the same circuit can be utilised with different load section 206 components; for example, one or more LEDs; LEDs of different colour; or alternative components.

Variations of the above embodiments will be apparent to the skilled person. The precise configuration of hardware components may differ and still fall within the scope of the present invention. The circuit of the above-described embodiment is suitable for a solid- state LED cabin lighting power supply. This however is not the limitation of the circuit as the applications can be more general in offering low power outputs not limited to aircraft or LED lighting. The circuit can, for example, be used for the low power applications in commercial and domestic lighting and other applications where high quality and low supply current distortion is required and for providing high quality low power output for many requirements. This becomes increasingly important as LEDs become more efficient and the required power levels are reduced.

Further, any number of LEDs may be utilised, provided they are included as a single series-connected string. Any colour or configuration of LED may be used. A series string of LEDs as described may comprise a plurality of series strings provided that the subsequent strings are matched to the first string in terms of forward voltage and Vf with temperature. The group of LED strings arranged in series then functional electronically as a single string.

Further, a diode or Zener diode need not be utilised and other switching arrangements may be used.

The present invention provides a power supply having negligible or very low power supply current distortion. The present invention, thus, addresses a significant problem associated with supplying power to thousands of low power supply loads. In conventional arrangements, each low power supply load will export supply current distortion and the effect of this on the supply infrastructure (which may supply many thousands of individual loads) is significant and material. The combined effect of thousands of uncontrolled low power lights/power supplies connected to a power generator will cause extra costs in cabling and generators to handle the current distortion.

This problem is addressed, in embodiments, by means of a load ballast circuit arranged in parallel with the LED load device. This provides control of current in the power supply circuit from the zero crossing voltage up to the maximum instantaneous voltage. In contrast, known arrangements are unable to control the current across the entire voltage range and would lead to significant power supply distortions.

Embodiments of the present invention have been described with particular reference to the examples illustrated. While specific examples are shown in the drawings and are herein described in detail, it should be understood, however, that the drawings and detailed description are not intended to limit the invention to the particular form disclosed. It will be appreciated that variations and modifications may be made to the examples described within the scope of the present invention.

Claims

1. A linear power supply circuit operable to receive a rectified AC voltage supply having an instantaneous voltage which varies between a minimum zero-crossing value and peak value as a function of time, the linear power supply circuit being operable to power an LED arrangement continuously throughout an AC voltage cycle and comprising:

an LED arrangement having a minimum drive voltage above which supply current is operable to flow through said LED arrangement;

a drive circuit section operable to control the supply current through the LED arrangement;

a load ballast circuit arranged in parallel with said LED arrangement and operable to enable supply current to flow therethrough when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage such that the circuit is operable to provide current control across an entire power cycle of the AC voltage supply and such that the current is controlled across the whole instantaneous voltage range of the AC voltage supply from the zero crossing point to the maximum

instantaneous voltage; and

an energy storage circuit arranged in parallel with the LED arrangement, the energy storage circuit being operable to store energy when the instantaneous voltage across the LED arrangement is greater than or equal to the minimum drive voltage and to power the LED arrangement when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage.

2. A power supply circuit according to claim 1 , wherein the drive circuit section comprises an operational amplifier operable to control the supply current and the load ballast circuit comprises switching means to enable current to flow through the load ballast circuit when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage of said LED arrangement.

3. A power supply circuit according to claim 2, wherein the switching means are switched in dependence upon the output of the operational amplifier.

4. A power supply circuit according to claim 2 or 3, wherein the switching means comprises a diode.

5. A power supply circuit according to claim 4, wherein the switching means comprises a Zener diode.

6. A power supply circuit according to any one of claims 2 to 5, wherein the load ballast circuit comprises a field effect transistor.

7. A power supply circuit according to any one of the preceding claims, wherein the drive circuit section comprises a field effect transistor.

8. A power supply circuit according to claim 7, wherein the drive circuit section comprises a transistor pair including a MOSFET and a bipolar junction transistor.

9. A power supply circuit according to any one of the preceding claims, wherein the energy storage circuit comprises a capacitive element.

10. A power supply circuit according to claim 9, wherein the energy storage circuit further comprises a resistive or current limiting element in series with the capacitive element.

1 1 . A power supply circuit according to claim 9 or 10, wherein the capacitive element is operable to charge when the instantaneous voltage across the LED arrangement is greater than or equal to the minimum drive voltage and is arranged to discharge through the LED arrangement when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage.

12. A power supply circuit according to claim 9, 10 or 1 1 when dependent upon claim 6, wherein the energy storage circuit is connected in parallel with the both the LED arrangement and the field effect transistor.

13. A power supply circuit according to any one of the preceding claims, wherein the LED arrangement comprises a plurality of LEDs arranged in series connection.

14. A power supply circuit according to any one of the preceding claims, further comprising a rectifier connectable to an AC power source and operable to generate a the rectified AC voltage signal therefrom.

15. A power supply circuit according claim 14, wherein the AC power source comprises an aircraft power supply.

16. A power supply circuit according to claim 15, wherein the AC power source has a frequency in the range of 300 - 800 Hz.

17. A power supply circuit according to claim 16, wherein the AC power source has a RMS voltage of 1 15V.

18. Aircraft lighting apparatus comprising the circuit of any one of the preceding claims.

19. A power supply network comprising an AC power source connected to a multiplicity of power supply circuits as claimed in any one of claims 1 to 17.

20. A method of controlling the supply current in a linear power supply circuit for powering an LED arrangement continuously through an AC voltage cycle, the method comprising:

receiving a rectified AC voltage supply having an instantaneous voltage which varies between a minimum zero-crossing value and peak value as a function of time; driving an LED arrangement with said rectified AC voltage supply, the LED arrangement having a minimum drive voltage above which supply current is operable to flow through said LED arrangement;

charging an energy storage circuit arranged in parallel with the LED arrangement during the period when the instantaneous voltage is equal to or greater than the minimum drive voltage;

driving an LED arrangement with charge from said energy storage circuit during the period when the instantaneous voltage is less than the minimum drive voltage; and controlling the supply current by:

utilising a drive circuit section operable to control the supply current when the AC voltage drop across the LED arrangement is equal to or greater than the minimum drive voltage; and

switching to a load ballast circuit arranged in parallel with said LED arrangement and operable to enable supply current to flow therethrough when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage such that the circuit is operable to provide current control across an entire power cycle of the AC voltage supply and such that the current is controlled across the whole instantaneous voltage range of the AC voltage supply from the zero crossing point to the maximum instantaneous voltage.

21 . A method according to claim 20, wherein the drive circuit section comprises an operational amplifier operable to control the supply current and the load ballast circuit comprises switching means to enable current to flow through the load ballast circuit when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage of said LED arrangement.

22. A method according to claim 21 , wherein the switching means are switched in dependence upon the output of the operational amplifier.

23. A method according to claim 21 or 22, wherein the switching means comprises a diode.

24. A method according to claim 23, wherein the switching means comprises a Zener diode.

25. A method according to any one of claims 20 to 24, wherein the load ballast circuit comprises at least one field effect transistor.

26. A method according to any one of claims 20 to 25, wherein the drive circuit section comprises a field effect transistor.

27. A method according to claim 26, wherein the drive circuit section comprises a transistor pair including a MOSFET and a bipolar junction transistor.

28. A method according to any one of claims 20 to 27, wherein the energy storage circuit comprises a capacitive element.

29. A method according to claim 28, wherein the energy storage circuit further comprises a resistive or current limiting element in series with the capacitive element.

30. A method according to claim 28 or 29, wherein the capacitor is operable to charge when the instantaneous voltage across the LED arrangement is greater than or equal to the minimum drive voltage and is arranged to discharge through the LED arrangement when the instantaneous voltage across the LED arrangement is less than the minimum drive voltage.

31 . A method according to claim 28, 29 or 30 when dependent upon claim 25, wherein the energy storage circuit is connected in parallel with the both the LED arrangement and the field effect transistor.

32. A method according to any one of claims 20 to 31 , wherein the LED arrangement comprises a plurality of LEDs arranged in series connection.

33. A method according to any one of claims 20 to 32, wherein the AC power source comprises an aircraft power supply.

34. A method according to claim 33, wherein the AC power source has a frequency in the range of 300 - 800 Hz.

35. A method according to claim 34, wherein the AC power source has a RMS voltage of 1 15 V.