WO2014036302A1

WO2014036302A1 - Miniaturized antennas

Info

Publication number: WO2014036302A1
Application number: PCT/US2013/057359
Authority: WO
Inventors: Gokhan Mumcu; Saurabh Gupta; Paul A. Herzig
Original assignee: University Of South Florida
Priority date: 2012-08-29
Filing date: 2013-08-29
Publication date: 2014-03-06

Abstract

In one embodiment, a miniaturized antenna inlcudes a substrate, an antenna loop associated with the substrate, and one or more vertically-oriented electrically-conductive elements coupled to the antenna loop that increase the efficiency of the antenna.

Description

MINIATURIZED ANTENNAS

Cross-Reference to Related Application(s)

This application claims priority to co-pending U.S. Provisional Application serial number 61/694,600, filed August 29, 2012, which is hereby incorporated by reference herein in its entirety.

Background

Small antennas are desirable in a variety of applications. One such application relates to global positioning system (GPS) jamming. The relatively weak signal level of the GPS makes it inherently vulnerable to intentional or unintentional jammers. Military systems generally address this drawback by employing multi-antenna GPS arrays to generate pattern nulls in the directions of the jamming signals. Integration of such anti-jam GPS arrays with compact unmanned vehicles and portable devices demands efficient miniature multi-band antennas.

To address the above need, several techniques have been applied for miniaturization of dual-band GPS antennas operating at the L2 (1227 MHz) and LI (1575 MHz) frequencies. For example, a stacked patch antenna implemented in low-temperature cofired ceramic technology (LTCC) has achieved a compact footprint size of ~λο/9 x X¾/9 at the L2 band over a high permittivity = 14) λο/6.8 x λ /6.8 LTCC tape. To achieve a reduced lateral substrate size comparable to the antenna footprint, others have implanted the stacked patch design over a much higher permittivity (s_r = 45) ceramic substrate and realized an overall antenna size of ~λ₀/8 λο/8 around the L2 band. Although the antenna is miniaturized, impedance matching and fabrication of such elements are challenging due to issues associated with machinability and cost of high-contrast ceramic substrates. In another case, a fragmented aperture method was employed but the footprint size of the antenna was still about ~λβ/5 x λο/5 at the L2 band. Others used quadruple inverted-F with a multilayered feed structure to achieve a compact circularly polarized (CP) antenna. However, low gain and radiation efficiency continue to be the design challenges. Recently, design flexibilities offered by the metamaterial inspired antennas have been harnessed to develop small dual band GPS elements without necessitating the use of high permittivity substrates. These antennas have been shown to operate with a λο/6.7 x λο/6.7 footprint area and a λο/13 antenna height at the L2 band. Other traditional antenna miniaturization techniques, such as shorting pins and meandering, have also been employed for GPS antenna miniaturization. Nevertheless, low radiation efficiency and small bandwidth continue to be the major limitation factors.

From the foregoing discussion, it can be appreciated that it would be desirable to have alternative antenna designs that enable miniaturization.

Brief Description of the Drawings

The present disclosure may be better understood with reference to the following figures. Matching reference numerals designate corresponding parts throughout the figures, which are not necessarily drawn to scale. Fig. 1A is a schematic drawing of two unit cells cascaded in a circularly periodic fashion to form a resonator operating at K = π frequencies.

Fig. IB is a dispersion diagram of the antenna unit cells of Fig. 1A.

Fig. 1C is an image of a fabricated dual-band coupled double loop (CDL) GPS antenna.

Figs. ID and IE are L2 and LI band RHCP gain patterns, respectively, measured for the antenna of Fig. 1C over a 10" x 10" ground plane.

Fig. IF is a graph that shows measured broadside RHCP gain performance within the 1.1-1.7 GHz band for the antenna of Fig. 1C.

Figs. 2A-2D are images that show L2 band surface current density on an outer loop of a CDL antenna with substrate sizes of 1.6" x 1.6", 1.4" x 1.4", 1.1" x 1.1", and 1.1" x 1.1", respectively, when the outer loop is loaded with 300 mil tall 39.6 mil diameter vertical pins.

Fig. 2E is a schematic drawing of L2 band current flow directions for port 1 excitation of the antenna of Fig. 1C.

Fig. 3 A is a perspective view of an embodiment of a dual band CDL GPS antenna having a reduced substrate.

Fig. 3B is a top view of the antenna of Fig. 3 A.

Fig. 4A is a graph that shows the variation in L2 and LI band radiation efficiencies as the via locations are changed from the inner to the outer edge of the outer loop of the antenna of Figs. 3A and 3B.

Fig. 4B is a graph that shows L2 band resonance frequency and radiation efficiency as a function of the total number of vias for the antenna of Figs. 3A and 3B.

Fig. 4C is a graph that shows L2 band radiation efficiency versus total number of vias as the shift in resonance frequency is compensated for by varying the via heights for the antenna of Figs. 3A and 3B. Fig. 5A is a perspective view image of a fabricated dual-band CDL GPS antenna having a reduced substrate.

Fig. 5B is a top view of the antenna of Fig. 5 A.

Fig. 5C is an image that provides a size comparison between the dual-band CDL GPS antenna of Figs. 5A and 5B and a conventional L2 band patch.

Fig. 5D is a graph that shows the measured |S11 | performance of the dual-band CDL GPS antenna of Figs. 5A and 5B and the L2 band patch within the 1.1- 1.7 GHz band.

Fig. 6A is a top view image of a fabricated antenna and its feed structure.

Fig. 6B is a side view drawing of a feed mounting for the antenna of Fig. 6A.

Fig. 7A is a graph that shows simulated and measured broadside RHCP and LHCP gains within the 1.1-1.7 GHz band for the antenna of Fig. 6A.

Figs. 7B and 7C are simulated and measured radiation patterns in the x-z plane at the L2 and LI bands, respectively, for the antenna of Fig. 6A.

Fig. 8A is a perspective view of an embodiment of a dual-band CDL GPS antenna having a high-permittivity substrate.

Fig. 8B is a top view of the antenna of Fig. 8A.

Fig. 8C is a graph that shows simulated broadside realized RHCP and LHCP gains for the antenna of Figs. 8 A and 8B.

Fig. 9A is a top view of an embodiment of a 2 x 2 CDL GPS antenna array.

Fig. 9B is a perspective view image of a fabricated 2 2 CDL GPS antenna array.

Figs. 10A and 10B are simulated and measured (respectively) normalized azimuth plane band (Θ = 75°) L2 band total gain patterns of the 2 x 2 CDL GPS array of Fig. 9B. Detailed Description

As described above, it would be desirable to have alternative antenna designs that enable miniaturization. Disclosed herein are miniaturized antennas that include vertically oriented, electrically conductive, elements that increase the efficiency of the antennas to the extent that they can be significantly miniaturized without sacrificing performance. In one embodiment, an antenna is configured as a miniature coupled double loop (CDL) antenna suitable for use as a radiating element of a compact dual-band GPS array. In some embodiments, the antenna comprises electrically conductive pins that extend downward through a substrate of the antenna. In other embodiments, the antenna comprises electrically conductive strips that extend down along the sides of the substrate of the antenna.

In the following disclosure, various specific embodiments are described. It is to be understood that those embodiments are example implementations of the disclosed inventions and that alternative embodiments are possible. All such embodiments are intended to fall within the scope of this disclosure.

Described in this disclosure are miniaturized antennas. In one embodiment, a miniaturized antenna is configured as a modified dual-band CDL antenna capable of providing efficient radiation performance from an overall L2 band lateral size of λο/8.8 ^x λο/8.8. The CDL antenna can be loaded with lumped capacitors and inductive vias to concurrently achieve the goals of miniaturization, high radiation efficiency, and proper L2/L1 resonance frequency spacing. In the discussion that follows, described is an initial 1.6" x 1.6" dual band CDL GPS antenna design and an explanation of its operational principles. The efficiency performance of the antenna is demonstrated as its substrate is gradually decreased down to a 1" x 1" footprint size and volumetric reactive pin loading is identified for alleviating the excessive conductor loss of the outer loop. Subsequently described is a 1.1" x 1.1" (~W8.8 x λ₀/8.8 at L2) reactively loaded dual-band CDL antenna operating with 4.7 dB and 3.3 dB realized right-handed circularly polarized (RHCP) gains at L2 and LI bands, respectively. Furthermore, it is demonstrated that further antenna miniaturization can be achieved if the reactively loaded CDL layouts are realized over higher permittivity ceramic materials. Specifically, a dual-band CDL GPS antenna is presented with an overall size of 0.8" x 0.8" (λο/12 x λο/12 at L2) over an ¾ = 25 substrate. The antenna is shown to operate over an infinite ground plane with simulated 3 dB L2 and 3.2 dB LI band RHCP gains corresponding to >75% radiation efficiency.

Fig. 1 depicts a CDL GPS antenna realized by making use of the mode diversity observed in partially-coupled transmission lines. In this design approach, the antenna footprint is treated as a circularly periodic structure comprising two unit cells (see Fig. 1A). The resonances of the antenna leading to broadside radiation are associated with the K = π frequencies of the dispersion diagram, which depicts the phase shift per unit cell attained by a propagating wave. As shown in Fig. IB, the presence of dual transmission lines within the unit cells enables the antenna to support radiation simultaneously at two different frequencies. Unit cell parameters such as width of the microstrip lines, their lengths, lumped reactive loads, and coupling capacitors (CM) can be utilized to concurrently adjust the separation between the = π frequencies and achieve footprint miniaturization without resorting to high permittivity (ε_Γ > 10) ceramic substrates. For the dual-band GPS antenna shown in Fig. 1C, the antenna footprint was constrained by 1 " x 1" in the design stage to limit its L2 band electrical size to approximately λο/10 x λο/10, where λο is the free space wavelength and is calculated by dividing the speed of light by the frequency at which the antenna operates. In addition, Rogers TMMl Oi (ε_Γ = 9.8, tan8 = 0.002) was selected as the antenna substrate material with 500 mil thickness to realize an efficient GPS antenna with satisfactory bandwidth performance. RHCP gain was achieved by employing two 90° out-of- phase 50 Ω coaxial probes within the rotationally-symmetric antenna layout. Fig. IF depicts the broadside RHCP and left-handed circularly polarized (LHCP) gain of the antenna measured over a 10" x 10" ground plane within the 1.1 to 1.7 GHz band. The RHCP and LHCP gain patters measured in the x-z plane are also demonstrated in Figs. ID and IE. Specifically, this CDL GPS antenna operated with measured 3.4 dB (86% efficiency) and 4.4 dB (95% efficiency) RHCP gains at the L2 and LI bands, respectively. The measured >0 dB gain bandwidths were also found to be satisfactory for the GPS requirements. Nevertheless, the antenna may still be unsuitable for use in a miniature GPS array because its 1.6" x 1.6" substrate is significantly large as compared to its small footprint. In some cases it would be desirable to reduce the substrate size of the CDL GPS antenna to barely fit its footprint in order to utilize it as the element of a miniature anti-jam GPS array. In order to satisfy the desired >0 dB gain bandwidth criteria, the radiation efficiency of the reduced substrate CDL GPS antenna must still be maintained above approximately 70%.

To understand the effect of the substrate size on the performance of the CDL GPS antenna, several computational simulations were carried out by changing the lateral dimensions of the substrate from 2" x 2" to 1" x 1" (in decrements of 0.1"²) while the antenna layout was kept intact (Ansoft HFSSvl l.2 was used as the design tool). As can be expected, the reduction in the substrate size causes the resonances of the antenna to shift to higher frequencies. Specifically, a total of 20 and 19 MHz frequency shifts were observed at the L2 and LI band resonances, respectively. Most importantly, the efficiency of the resonance mode responsible for the L2 band radiation was observed to dramatically decrease from 86% to 26% when the footprint and substrate sizes were made equal to each other. On the contrary, the efficiency of the resonance mode supporting the LI band radiation was found to be always larger than 90% and not degraded by the substrate size reduction.

To understand the loss mechanism of the L2 mode resonance, computational studies were carried out to evaluate the surface current densities. Because L2 mode radiation is primarily associated with the outer loop of the antenna layout, Figs. 2A-D present the outer loop's current density for various substrate sizes. From these plots, it is clearly seen that a reduction in substrate size beyond 1.6" x 1.6" is accompanied by significant increases in surface current densities at the outer loop and coupling capacitors. Consequently, the conductor losses can be identified as the main cause of the low L2 band radiation efficiency observed in the reduced substrate size CDL GPS antenna.

One approach that can be employed for improving the radiation efficiency of the outer loop is resorting to a wider microstrip line. However, this causes a physically and electrically larger antenna structure. Therefore, the surface current density of the outer loop was decreased by distributing the current volumetrically with metallized vertical pins, as depicted in Figs. 2D and 2E. Figs. 3A and 3B illustrate an example antenna embodiment employing such pins. More particularly, these figures illustrate an embodiment of a miniature CDL antenna 10. The antenna 10 comprises a substrate 12 provided on top of a ground plane 14. In some embodiments, the substrate 12 can be made of a thermoset laminate. The substrate 12 has a top surface 16 upon which is formed an inner conductive loop 18 (or "inner loop") and an outer conductive loop 20 (or "outer loop") that surrounds the inner loop. In the illustrated embodiment, the inner loop 18 is a meandered loop having a general cloverleaf shape and the outer loop 20 has a general rectangular (e.g., square) shape. Extending between outer corners of the inner loop 18 and inner corners of the outer loop 20 are lumped coupling capacitors 22.

As illustrated in Fig. 3A, coupled to the outer loop 20 and extending downward from the outer loop into the substrate 12 are multiple electrically conductive pins 24. In some embodiments, the pins 24 are formed by creating vias in the substrate 12 and filling them with a suitable conductive material. In the illustrated example, there are 6 such pins 24 provided along each side of the outer loop 20 so that there are a total of 20 pins extending from the outer loop. Further illustrated in Fig. 3 A are coaxial feed ports 26 that extend up from the ground plane toward the surface 16 of the substrate 12 that can be used to excite the circuitry on the surface.

The pins 24 were electrically connected to the outer loop 20 and therefore did not make any connection with the ground plane 14. Fig. 2E demonstrates the L2 band current distribution on the surface of the pin-loaded antenna 10 when the first feed port 26 is excited. It is observed that the currents at the top surface of each unit cell (i.e., half of the antenna 10) are oriented in the same direction due to the = π resonance, thus resulting in a broadside radiation. The presence of the pins 24 modifies the current distribution in two different ways. Specifically, the pins 24 on the antenna facing parallel to the x-z plane support a current distribution that serves as an extension length for the surface current on the top of the antenna 10 (see current path #2 in Fig. 2E). On the other hand, the pins 24 on the antenna 10 facing parallel to the y-z plane provide a reactive loading effect by supporting a meandered current distribution (see current path #1 in Fig. 2E). Therefore, the loading effect of these pins 24 causes a reduction in L2 band resonance frequency. This effect, in turn, also allows a larger line width for the outer loop 20 without necessitating the enlargement of the antenna size. Because the current is volumetrically distributed, the current densities at the outer loop 20 and capacitors 22 are significantly reduced with the help of the pins 24. Consequently, the radiation efficiency increases from a mere 13% to 88%. It is also important to note that the presence of the pins 24 at the outer loop 20 does not influence the LI band surface current density concentrated at the inner loop 18. Hence, LI band radiation efficiency is minimally affected with this CDL antenna loading scheme.

To identify the effects of the suggested via arrangements on the radiation performance of the CDL GPS antenna, computational studies were carried out to consider various via parameters such as their relative position along the width of the outer loop, radius, and total number. In these studies, the antenna footprint layout was similar to that of the CDL GPS antenna 10 shown in Figs. 3 A and 3B. The substrate 12 was a 1.1" x 1.1" x 0.5" RogersTMMlOi (ε_Γ = 9.8, tan5 = 0.002) substrate and the outer loop 20 had outer dimensions of 1" x 1". There were N = 20, 300-mil pins 24 metalized from copper coupled to the outer loop 20. Each via 24 had a radius of 0.0198". The capacitors 22 were 0.6 pF capacitors and the coaxial feed ports 26 were 50 Ω feed ports.

Fig. 4A presents the variations in the radiation efficiencies of the L2 and LI band resonances as the pins 24 were gradually re-located from the inner to the outer edge of the outer loop 20 in increments of 20 mil. It is observed that the worst-case efficiencies are well above 70% due to the presence of the pins 24. Specifically the LI band radiation efficiency decreases from 78% to 70% as the pins 24 are re-located to the outer edge of the outer loop. On the other hand, the L2 band efficiency increases from 73% to 88% for the same case. Because a standard printed circuit board (PCB) fabrication typically realizes metalized vias from composites having lower conductivities than copper, the outer edge of the outer loop 20 was eventually chosen for the via locations to maximize the efficiency of the L2 band resonance. It is also important to note that, despite the change in via position along the line width, both of the resonance frequencies remain unaltered.

Fig. 4B depicts the change in the L2 band resonance frequency and efficiency as the total number of the pins 24 are changed from N = 0 to N = 36. As can be seen from that figure, the antenna 10 is precisely tuned to 1227 MHz when the outer loop 20 is loaded with N = 20 300 mil long pins 24 and radiates with 85%> efficiency. When the number of pins 24 is gradually decreased from N = 20, the L2 band resonance shifts to higher frequencies, implying an electrically larger antenna structure. On the other hand, increasing the number of pins 24 beyond N = 20 does not provide further pronounced benefits in terms of radiation efficiency and electrical size. Reducing the number of pins 24 without affecting the electrical size, resonance frequency, and radiation efficiency performance of the CDL antenna is important for achieving a lower cost fabrication. The resonance frequency shift observed in the L2 band for N = 20 can be conveniently compensated by increasing the height of the pins 24. As demonstrated in Fig. 4C, longer pins 24 do not significantly degrade the radiation efficiency and the CDL antenna 10 continues to exhibit its miniature size. Nevertheless, the reduction in number of pins 24 is limited by the maximum thickness of the substrate 12. For example, for N = 4, the via heights must be larger than 500 mil to keep the resonance precisely tuned to 1227 MHz (i.e., hence, N = 4 case was modeled for an antenna residing on 550 mil thick substrate). In addition, it may be convenient to restrict the via sizes to increments of available substrate thicknesses for a given PCB fabrication approach. For instance, 500 mil thick CDL antenna can be realized from a stack of five 100 mil thick substrates, thereby allowing a convenient method to realize customized via heights in increments of 100 mil. Consequently, the volumetrically loaded CDL GPS antenna design described below utilized 300 mil long N = 20 vias to accomplish the L2 band resonance frequency tuning. The computational studies also demonstrated that the L2 band resonance frequency does not significantly get affected by the via diameters. Therefore, the diameter of the pins 24 was selected as 39.6 mil based on the availability of the drilling tools in the laboratory.

To achieve >0dB gain bandwidth larger than the minimum GPS criteria of 24 MHz, the thicknesses of the inner and outer loops 18, 20 were modified to be 50 and 215 mil, respectively. The LI band resonance frequency was tuned to 1575 MHz by utilizing coupling capacitors of CM = 0.6 pF modeled with an equivalent series resistance (ESR) of 0.35 Ω. The antenna 10 was excited by 900 offset capacitively coupled coaxial probes. Specifically, the probes were 70 mil below the top surface of the antenna to provide a good impedance match. The computed |S11| < -lOdB bandwidths of 10 and 14 MHz lead to >0dB RHCP gain bandwidths of 52 and 60 MHz at the L2 and LI bands, respectively. As shown in Fig. 4C, the antenna operated with peak realized RHCP gains of 3.8 dB at the L2 band and 3 dB at the LI band, corresponding to 88% and 75% radiation efficiencies, respectively. The computed cross-polarization levels are also at least 15 dB lower than the corresponding peak gains at the L2 and LI bands, implying a <ldB axial ratio performance. The electric field distributions plotted over the top surface of the antenna confirms that the L2 band radiation is primarily associated with the outer loop 20, whereas LI band radiation is controlled by the inner line parameters and coupling capacitor values.

To verify performance of the design, a miniature dual-band CDL GPS antenna was fabricated using two layers of 250 mil thick Rogers TMMlOi substrate, as depicted in Fig. 5A. The initial antenna prototype fabricated using the dimensions of the computational model described above in relation to Figs. 3A and 3B was found to exhibit the L2 and LI band resonances at higher frequencies (i.e., L2 band resonance at 1.3 GHz and LI band resonance at 1.62 GHz). This discrepancy between the simulated and measured resonances can be attributed to the computational model (e.g., accuracy of the dielectric constant, numerical errors), the presence of air gaps in the multilayered structure, capacitor tolerances, and the manual realization of vias from 25 mil diameter wires. To tune the resonance frequencies to the close proximity of L2 and LI frequencies, the antenna prototype was slightly modified through several fabrication iterations. Specifically, the coupling capacitors (obtained from ATC, 0402 size) were increased from 0.6 pF to 1.3 pF to precisely tune the LI band resonance frequency. Subsequently, via heights were increased from 300 mil to 400 mil and a slightly wider outer loop was employed to achieve the L2 band frequency tuning. Fig. 5B shows the top view of the dual-band CDL GPS antenna over the 500 mil thick 1.1" x 1.1" Rogers TMMlOi substrate with its experimentally finalized footprint dimensions. To demonstrate the size reduction performance of the miniature dual-band CDL GPS antenna, a standard L2 band patch was also designed and fabricated over the identical 500 mil thick substrate material, as depicted in Fig. 5C. Fig. 5D presents a comparison of the |S11 | responses of the miniature CDL GPS and the conventional patch antennas measured over a 24" x 24" ground plane. Specifically, the CDL GPS antenna resonates at 1220 and 1580 MHz with [ S 111 < -lOdB bandwidths of 16 and 14 MHz, respectively. On the other hand, the patch provides a much wider bandwidth (28 MHz) at the L2 band due to its 60% larger physical size. Nevertheless, as will be shown in the following (see also Fig. 7A), the >0dB gain bandwidth of the CDL GPS antenna still makes it suitable to be employed in dual-band GPS applications.

The circularly polarized gain of the CDL GPS antenna was measured at an anechoic chamber after integrating the antenna with the feed network shown in Fig. 6A. Fig. 6B illustrates the antenna 30 and its feed network in side view. The antenna 30 comprised a 25 mil thick Rogers 6010.2 LM (ε_Γ = 10.2, tan5 = 0.0022) substrate 32 that was provided on a ground plane 34 and that included top metallizations 36 and vertical pins 38. The feed network included a feed network substrate 40 that was provided on the underside of the ground plane 34 and 50 Ω grounded coplanar waveguide (CPWG) lines 42. The feed network further included a 50 Ω resistive termination (i.e., isolation port), a 50 Ω coaxial probe (i.e., input), and a surface mount quadrature hybrid coupler (Anaren Microwave, Xinger-brand components, part#XC1400P-03S). The antenna 30 was fed through vertical copper pins 44 connected to the CPWG lines 42.

Fig. 7A presents the measured and simulated broadside RHCP and LHCP gains within the 1.1 to 1.7 GHz band when the antenna of Figs. 6 A and 6B was positioned over the 24" x 24" brass ground plane. Because of the addition of the feed network and associated fabrication tolerances, the peak gains were observed at slightly different frequencies as compared to the measured |S11 | performance reported in Fig. 5D. Specifically, the miniature dual-band CDL GPS antenna exhibited 4.7 dB gain at 1250 MHz and 3.3 dB gain at 1579 MHz with 50 and 60 MHz >0dB gain bandwidths, respectively. The cross-polarization levels at both bands were measured to be >10dB lower than the co-polarized gain. The measured increase in the L2 band cross-polarized gain (as compared to simulated) can be attributed to the fabrication tolerances, inequalities in the via heights, and the ground plane shape and size. Figs. 7B and 7C demonstrate the measured and simulated x-z plane radiation patterns at the L2 and LI bands, respectively. It is seen that the measured patterns are in agreement with the simulated ones.

The dual-band CDL antennas can be further miniaturized by employing higher permittivity substrates. Because substrate and conductor losses can become more critical for additional size reductions, the performance of the volumetrically loaded CDL antenna was investigated by carrying out the design over a low-loss high permittivity ceramic material (ε_Γ = 25, tan8 = 0.0002). Figs. 8A and 8B illustrate an example antenna 50 that exhibits dual resonances within the vicinity of the GPS L2 and LI bands. The antenna 50 is similar in many ways to the antenna 10 described above in relation to Figs. 3A and 3B. Accordingly, the antenna 50 comprises a substrate 52 provided on top of a ground plane 54 and having a top surface 56 upon which is formed an inner loop 58 and an outer loop 60. Extending between outer corners of the inner loop 58 and inner corners of the outer loop 60 are lumped coupling capacitors 62.

Instead of comprising electrically conductive pins that extend down from the outer loop 60 through the substrate 52, the antenna 50 comprises electrically conductive strips 64 that extend down from the outer loop along outer sides 66 of the substrate. In some embodiments, the strips 64 are 1.4 mil thick copper strips. In the illustrated example, there are 6 such strips 64 provided along each side 66 of the outer loop 60 so that there are a total of 20 strips extending from the outer loop. Further illustrated in Fig. 8A are coaxial feed ports 68 that extend up from the ground plane 54 toward the surface 56 of the substrate 52 that can be used to excite the circuitry on the surface.

By providing strips 64 that extend over the substrate edges (instead of vias), an additional flexibility is obtained for possible antenna prototyping. The substrate 52 can be realized as a 500 mil thick 0.8" x 0.8" substrate corresponding to an electrical size as small as λο/12 x λο/12 at the L2 band. As shown in Fig. 8C, the simulated broadside realized peak RHCP gains are 3 dB (80% efficiency) and 3.2 dB (75% efficiency) at the L2 and LI bands, respectively. The antenna 50 exhibits a simulated >0dB gain bandwidth of 36 MHz in the L2 band and 44 MHz in the LI band. As compared to the 1.1" x 1.1" antenna described above, the 0.8" x 0.8" antenna provided on the higher permittivity substrate exhibits a 47% smaller footprint area with similar radiation efficiency performance at the expense of a smaller bandwidth. The 36 MHz L2 bandwidth suggests that the antenna size could be further reduced while satisfying the gain bandwidth specifications if alternative materials with ε_Γ > 25 are employed. Nevertheless, tolerances associated with fabrication, material parameters, and numerical simulations can cause major difficulties in realizing antennas with precisely tuned resonances. Therefore, these factors can also be considered in practical antenna applications.

In the foregoing discussion, miniaturized dual-band CDL GPS antennas suitable for compact GPS arrays were described. To achieve the goals of size miniaturization, high radiation efficiency, and proper spacing of the L2 and LI resonances, the antenna layout simultaneously incorporated meandered lines, lumped capacitors, and volumetric reactive loadings. Specifically, a CDL antenna realized over 500 mil thick Rogers TMMlOi (ε_Γ = 9.8) and fed with a CPWG based hybrid was shown to exhibit broadside RHCP gains of 4.7 dB at L2 bands and 3.3 dB at LI bands with 50 and 60 MHz >0dB gain bandwidths, respectively. This CDL antenna was shown to occupy a 60% smaller area with its 1.1" x 1.1" (~λο/8.8 x λο/8.8 at L2) overall footprint as compared to a traditional patch antenna fabricated on the same substrate. Further size reduction without degrading the antenna efficiency can be achieved by using dielectric loading. To demonstrate this, another CDL antenna design was carried out over a high permittivity ceramic substrate (ε_Γ = 25) and was shown to operate with an overall footprint size of 0.8" x 0.8" (λο/12 x λο/12 at L2). Specifically, this CDL antenna exhibited computed broadside RHCP gains of 3 dB at L2 bands and 3.2 dB at LI bands with 36 and 44 MHz >0dB gain bandwidths.

Although the above discussion has focused on CDL antennas suitable for use as radiating elements of compact dual-band GPS arrays, it is emphasized that this is but one configuration and one application for the disclosed miniaturized antennas. In other embodiments, which are intended to fall within the scope of this disclosure, the antennas can support a single band or greater than two bands, and can be used in substantially any application in which an antenna with a small footprint is desired. In embodiments in which multiple antenna loops are employed, the loops need not be provided on the same layer of the antenna.

The miniature antennas described above can be used to form antenna arrays, such as a GPS antenna array. Fig. 9A is an example of such an antenna array 70. As shown in this figure, the array 70 comprises a 2 x 2 array of miniature CDL antennas 72 having constructions similar to those described above. The array 70 includes multiple groups of resonators 74 that are positioned near and between the antennas 72 to create a broadside- coupled split ring resonator-loaded (BC-SRR-loaded) array comprising dual-band (e.g., L2: 1227 MHz, LI : 1575 MHz) CDL antennas. By way of example, each antenna 72 can use a Rogers TMMlOi printed circuit board as its substrate (ε_Γ = 10.2, tan5 = 0.002) and exhibit an overall size of 1.1" x 1.1" x 0.5" (i.e., λ₀/8.5 x W8.5 at L2). This electrically small element size enables construction of a compact GPS array. By way of example, the separation between element edges can be 1.5". This implies a λο/3.7 center-to-center separation at the L2 band. The overall array footprint is 4.3" x 4.3".

Fig. 9B demonstrates the fabricated prototype of the 2 x 2 CDL GPS array loaded with the BC-SRRs. Each antenna element was fed using a quadrature coupler feed network from the back side of the array. The fabricated BC-SRRs were manually glued to the ground plane. Simulations and measurements were performed using the fabricated prototype and demonstrated an improved nulling capability in terms of null depths and accuracy as compared to the conventional array. Specifically, the null depth was improved by more than 10 dB when BC-SRRs were used. As an example, Fig. 10A demonstrates a scenario in which three distinct nulls are desired at the Θ = 75° azimuth plane along the φ = 45°, 150° and φ = 250°, 150°, and 250° directions. The simulated normalized patterns show that both the conventional and BC-SRR-loaded arrays can generate a null with similar depth along the φ = 250° direction. However, the conventional array clearly failed to produce a null along the φ = 45° direction due to its high cross-polarization level. In addition, the pattern null desired along the φ = 150° direction was mis-positioned by 15°. On the other hand, the BC-SRR- loaded array generated all three nulls along the desired directions. Moreover, the nulls were associated with < -20 dB lower realized gain as compared to the maximum realized gain. The measured total normalized patterns presented in Fig. 10B are in good agreement with the performance predicted by the simulations. Specifically, the BC-SRR loaded array can generate the pattern nulls more accurately along the desired directions. Additionally, the nulls are associated with < -11 dB lower measured realized gain as compared to the maximum realized gain by the array.

Claims

CLAIMS Claimed are:

1. A miniaturized antenna comprising:

a substrate;

an antenna loop associated with the substrate; and

one or more vertically-oriented electrically-conductive elements coupled to the antenna loop that increase the efficiency of the antenna.

2. The antenna of claim 1, wherein the substrate is made of a thermoset laminate.

3. The antenna of claim 1, wherein the substrate has lateral dimensions no greater than approximately 1.1 inches.

4. The antenna of claim 1, wherein the substrate has lateral dimensions no greater than approximately 0.8 inches.

5. The antenna of claim 1, wherein the substrate has lateral electrical dimensions no greater than approximately λο/8.8 at the L2 band, where ο is the free space wavelength and is calculated by dividing the speed of light by the frequency at which the antenna operates.

6. The antenna of claim 1, wherein the substrate has lateral electrical dimensions no greater than approximately λο/12 at the L2 band, where λο is the free space wavelength and is calculated by dividing the speed of light by the frequency at which the antenna operates.

7. The antenna of claim 1 , wherein the antenna loop is provided on a top surface of the substrate.

8 The antenna of claim 1, wherein the loop is a meandered loop.

9. The antenna of claim 1 , wherein the loop is generally rectangular.

10. The antenna of claim 1, wherein the vertically-oriented electrically-conductive elements are vertically-oriented electrically-conductive pins that extend downward from the antenna loop and into the substrate.

11. The antenna of claim 10, wherein the pins generally align with an outer edge of the loop.

12. The antenna of claim 1, wherein the vertically-oriented electrically-conductive elements are vertically-oriented electrically-conductive strips that extend downward from the antenna loop and along outer sides of the substrate.

13. The antenna of claim 1, wherein the antenna includes both an inner antenna loop and an outer antenna loop.

14. The antenna of claim 13, further comprising coupling capacitors that couple the inner antenna loop to the outer antenna loop.

15. The antenna of claim 1, further comprising one or more coaxial feed ports that vertically extend through the substrate.

16. A miniaturized antenna comprising:

a substrate having a top surface;

an inner antenna loop provided on the top surface;

an outer loop provided on the top surface and surrounding the inner antenna loop; coupling capacitors provided on the top surface that couple the inner and outer antenna loops; and

multiple vertically-oriented electrically-conductive elements that extend downward from one of the antenna loops that increase the efficiency of the antenna.

17. The antenna of claim 16, wherein the substrate has lateral dimensions no greater than approximately 1.1 inches.

18. The antenna of claim 16, wherein the substrate has lateral dimensions no greater than approximately λο/8.8 at the L2 band, where λο is the free space wavelength and is calculated by dividing the speed of light by the frequency at which the antenna operates.

19. The antenna of claim 16, wherein the vertically-oriented electrically- conductive elements are vertically-oriented electrically-conductive pins that extend downward from one of the antenna loops and into the substrate.

20. The antenna of claim 16, wherein the vertically-oriented electrically- conductive elements are vertically-oriented electrically-conductive strips that extend downward from one of the antenna loops and along outer sides of the substrate.

21. An antenna array comprising:

a ground plane;

multiple antennas provided on the ground plane, the antennas including a substrate, an antenna loop associated with the substrate, and one or more vertically-oriented electrically- conductive elements that increase the efficiency of the antenna; and

resonators provided on the ground plane between the antennas.