WO2014020250A2

WO2014020250A2 - Device for providing electricity

Info

Publication number: WO2014020250A2
Application number: PCT/FR2013/051583
Authority: WO
Inventors: Peyofougou Coulibaly
Original assignee: L'air Liquide,Societe Anonyme Pour L'etude Et L'exploitation Des Procedes Georges Claude
Priority date: 2012-07-31
Filing date: 2013-07-04
Publication date: 2014-02-06
Also published as: FR2994352B1; FR2994352A1; WO2014020250A3

Abstract

Device for providing electricity comprising a fuel cell (100) and a converter (1) of soft-switching type, the converter (1) comprising a DC voltage supply terminal (6) and a reference terminal (4), the converter (1) also comprising two switching cells (30; 31) each comprising an even number of breakers (321, 322; 323, 324) in series, each switching cell (30, 31) comprising an output terminal (361 362), each breaker (321, 322, 323, 324) being connected in parallel on the one hand to a respective capacitive element (341, 342, 343, 344) and on the other hand to a respective diode (101, 102, 103, 104), each switching cell (30, 31) being associated with an auxiliary circuit (50), the converter (1) moreover comprising a transformer (40) whose primary is linked to the output terminals (361, 362) of the switching cells (30; 31) and whose secondary is linked to a rectifier (44), of the type with MOSFET breakers (15, 16), the converter (1) also comprising a control facility (38) linked to the breakers (101, 102, 103, 104) of the switching cells and to the breakers (15, 16) of the rectifier (44), the control facility (38) being configured so as to control on the one hand the breakers (101, 102, 103, 104) of the switching cells (30, 31) according to a logic of "dual thyristor" ("DTL") type and, on the other hand, controlling the breakers (15, 16) of the rectifier (44) in synchronism respectively with the breakers (101, 102, 103, 104) of the switching cells.

Description

Electricity supply device

The present invention relates to a device and a method for supplying electricity using a fuel cell and a converter.

Conventionally, an electrical converter is used to convert the electrical energy of a fuel cell to suit the user load. The converter thus ensures the conversion of the raw voltage output from the battery (fluctuating voltage) into a stable DC voltage (for example 48Vdc or -48Vdc). FIG. 1 illustrates such an arrangement in which a converter 1 is arranged between a fuel cell 100 and a load 200.

In the example of FIG. 1, the device comprises a battery 60 connected to the terminals of the load 200 and to the output of the converter 1. This optional battery 60 allows in particular:

supplying auxiliaries of the battery 100, and in particular a preheating system before starting,

- provide electrical power when starting up or during high power demands.

As illustrated in FIG. 2, several stacks 100 with their respective converter 1 can be connected in parallel with the load 200.

Such a converter should preferably include:

a control system for regulating the input or output current of the converter for controlling the electric current delivered by the battery,

voltage regulation of the output of the converter a system for managing the charge of the battery (constant-current or constant-voltage charge),

a system of smoothing the current by a ramp to limit sudden changes in the current of the battery,

a system of maximum limitation of the current of the battery and the load, a system of safety (protection against overvoltages, overcurrents, short circuits, inversions of polarity, ...).

In fuel cell applications, cost, efficiency, and reliability are important criteria that condition the topology of the static converters used to transform the energy delivered by the battery.

Several solutions are known to improve the efficiency and reduce the cost of converters used in fuel cell applications.

The simplest known structures are generally converters of the "boost" or "buck" type (buck) with a principle called "synchronous recovery". The principle of synchronous rectification is to use low resistance MOSFETs (insulated gate field effect transistors) in place of the diodes (or in parallel) to obtain lower voltage drops at the rectifier in order to reduce losses. by conduction.

This solution effectively improves the efficiency but reduces the reliability of the converter. These converters are generally suitable for low power and do not reduce the switching losses of the switches. Moreover, these structures are not isolated and do not adapt well to a fluctuating battery voltage.

Soft switching is a technique that is generally used to reduce switching losses of semiconductors used as switches in converters.

Indeed, in the case of so-called "hard" conventional switching, the conduction or blocking of a semiconductor can not be instantaneous, switching losses are induced in the components due to the coexistence of voltage and current during switching. An increase in operating frequency causes these switching losses to increase. The operating frequency must be limited in order to maintain switching frequencies compatible with the semiconductors. Moreover, the waveforms of the currents and voltages appearing at the terminals of the components are generally oscillatory with extremely abrupt variations. These oscillations of current and voltage are at the origin of electromagnetic disturbances. The overcurrents and the overvoltages which result from it impose an oversizing of semiconductors.

Solutions such as Switching Assistance Circuits (CALCs) have been considered to address these issues. These circuits do not really reduce switching losses but transfer them to auxiliary elements (inductors, capacitors, resistors, diodes, ...) and are generally dissipative. These circuits are almost indispensable when power semiconductors have poor performance.

Gentle switching improves the efficiency of such converters. Increasing the operating frequency becomes possible and makes it possible to reduce the size of the passive components (transformers, inductors, capacitors, etc.) and thus leads to a reduction in the volume and the cost of the converter.

A structure widely used in the fuel cell application is the quasi-resonant converter known in the Anglo-Saxon literature by the acronym "ZVS-FB-PWM converter": ("Zero-Voltage-Switching Full-Bridge Pulse-Width-Modulation Converter"). It is an isolated structure in which the transformer makes it possible to adapt the voltage levels of the battery and the output of the converter. However, this converter operates in soft switching only from a current level of the load. In addition, synchronous recovery is much more difficult to obtain.

In addition, a disadvantage of known soft switching structures is the limitation of soft switching to low current. Indeed, at low current, the energy stored in an inductance connecting the primary of the transformer to the switching cells is not sufficient to discharge the capacitors of the switching cells. That is, the converter switches to hard switching with a sudden discharge of the capacitors in the switches. This operation generates additional oscillations and losses in the semiconductors.

An object of the present invention is to overcome all or part of the disadvantages of the prior art noted above.

To this end, the device according to the invention, which moreover conforms to the generic definition given in the preamble above, is essentially characterized in that it comprises a fuel cell and a soft-switching type converter for transforming the electrical energy delivered by the battery, the converter comprising two connection ends to a voltage source delivered by the battery, the two connection ends respectively defining a DC voltage supply terminal and a reference terminal, the converter also comprising two switching cells each having an even number of switches connected in series between the supply and reference terminals, each switching cell having a respective output terminal connected between the two halves of series switches of said cell considered, each switch being connected in parallel on the one hand to a respective capacitive element and on the other hand to a respective diode, each switching cell being associated with a respective auxiliary circuit comprising two first passive electrical elements (s) arranged in series between the supply and reference terminals and a second electrical element passive device having a first end connected between the first two electric elements (s) in series and a second end connected to the output terminal of the cell in question, the converter further comprising a transformer whose primary is connected to the output terminals of the cells the secondary is connected to a rectifier, of the MOSFET switch type, the converter also comprising a control member connected to the switches of the switching cells and the switches of the rectifier, the control member being configured to control firstly the switches of the switching cells according to a logic of the "dual thryristor"("LTD") type and secondly , control the rectifier switches in synchronism respectively with the switches of the switching cells.

The device thus makes it possible to combine smooth switching, synchronous rectification and reliable control of the switches from a quasi-resonant basic structure. The smooth switching operation over the entire operating range is obtained by using auxiliary switching circuits in particular. To increase the reliability of operation, a dual thyristor type control is preferably used at the main switches to primary.

With a dual thyristor logic controller, the converter stops safely, the auxiliary circuits are used to overcome the previously noted drawbacks.

In addition, the secondary switches of the synchronous rectifier are synchronously controlled with the primary switches in a simple and reliable way.

Furthermore, embodiments of the invention may include one or more of the following features:

the switches of the switching cells are of the MOSFET type with a blocking type operation controlled by the Zero Voltage Switching (ZVS) control element and device;

the first passive electrical elements of the auxiliary circuits comprise at least one of: a capacitance, a capacitance in parallel with a clipping diode, the second passive electrical element of the auxiliary circuits comprising at least one of: an inductance, an inductance in series with two controlled switches arranged in parallel and in opposite direction,

the switches of the switching cells are each equipped with a circuit provided with a detector of the saturation voltage across the switch and a control circuit of the switch receiving the voltage signal of the detector, said circuit of control being connected to the control member and being configured to perform the priming of the switch only when the controller initiates and that, simultaneously, the voltage measured by voltage detector is zero (V = 0) or between zero and seventeen volts,

an inductive element is placed in series with the primary of said transformer, each switching cell comprises first and second switches connected in series, the rectifier comprising a first and a second switch of the MOSFET type,

- The control member is configured to firstly control the ignition of the first switch of the rectifier at the same time as the ignition of a first switch of a first switching cell and, secondly, to control blocking the first switch of the rectifier at the same time as blocking the second switch of the second switching cell,

the control member is configured, on the one hand, to control the initiation of the second switch of the rectifier at the same time as the initiation of the second switch of the first switching cell and, on the other hand, to control the blocking of the second switch of the rectifier at the same time as the blocking of the first switch of the second switching cell.

The invention also relates to a method of supplying electricity using a device according to the characteristics above or below, comprising:

a first sequence in which the first switch of the first switching cell, the second switch of the second switching cell and the first switch of the rectifier) are conducting while the second switch of the first switching cell, the first switch of the second switching cell and the second rectifier switch are blocked, then

a second sequence in which the first switch of the first switching cell is switched to a blocking state, and then

a third sequence during which the second switch of the rectifier is controlled at initiation, then

a fourth sequence during which the second switch of the second switching cell is switched to the blocking state,

a fifth sequence during which the diode associated with the first switch of the second switching cell starts spontaneously and then the diodes associated with the second switch of the first switching cell and the first switch of the second cell spontaneously lock, then

A sixth sequence during which the first switch of the rectifier is switched to its off state.

According to other possible features:

the process then carries out the six sequences (S1 to S6) again symmetrically with: - The second switch of the first cell and its associated members taking the role (passing / blocked ...) of the first switch of the first cell and vice versa in the sequences; and

the first switch of the second switching cell and its associated members taking on the role (on / off ...) of the second switch of the second switching cell and vice versa, and

the second switch of the rectifier taking on the role (on / off) of the first switch of the rectifier.

The invention may also relate to any alternative device or method comprising any combination of the above or below features.

Other particularities and advantages will appear on reading the following description, made with reference to the figures in which:

FIG. 1 represents a schematic and partial view illustrating a possible example of the installation using an electricity supply device according to the invention,

FIG. 2 represents a schematic and partial view illustrating another possible example of the installation using an electricity supply device according to the invention,

FIG. 3 represents a schematic and partial view illustrating a detail of a converter according to a possible embodiment of the invention,

FIGS. 4 to 6 show diagrammatic and partial views illustrating possible examples of auxiliary circuits that can be used in the converter according to the invention,

FIG. 7 represents a schematic and partial view illustrating a detail of a dual thryristor type switch control system that can be used in the converter according to the invention,

FIGS. 8 to 13 represent schematic and partial views respectively illustrating six successive operating sequences that can be realized when using the converter of the device according to the invention,

FIG. 14 represents a chronogram of different signals during the operation of FIGS. 8 to 13.

FIG. 1 illustrates in greater detail a possible example of a structure of the converter 1 of the device according to the invention.

The converter 1, of the "soft switching" type comprises two connection ends 6, 4 connected to the voltage source P delivered by the fuel cell. The two connecting ends respectively defining a so-called "supply" terminal 6 in DC voltage and a so-called "reference" terminal 4. The converter 1 comprises two cells 30; Switching 31 each having an even number of switches 321, 322; 323, 324 connected in series between the power supply 6 and reference terminals 4. As shown, each cell 30; Switching 31 comprises two switches 321, 322; 323, 324 connected in series between the power supply terminals 6 and 4.

Switches 321, 322; 323, 324 cells 30; Preferably, the switching circuits 31 are preferably MOSFET (insulated gate field effect transistor) components and are preferably driven in a Thyristor Dual (LTD) logic mode. As described below, these switches 321, 322; 323, 324 preferably operate in a ZVS (Zero Voltage Switching) mode, that is to say, controlled blocking and spontaneous ignition under zero voltage.

Each switching cell 30, 31 has a respective output terminal 361 362 connected between the two switches in series of said cell 30; 31 considered. Each switch 321, 322, 323, 324 is connected in parallel on the one hand to a capacitive element 341, 342, 343, 344 (such as a capacitor for example) and on the other hand to a diode 101, 102, 103, 104 respectively.

The converter further comprises a transformer 40 whose primary is connected to the output terminals 361, 362 of the switching cells 30; 31. The secondary of the transformer 40 is connected to a rectifier 44. The rectifier 44 is of the switch type 15, 16 MOSFET, comprising for example a first switch 15 and a second switch 16. A so-called output inductor 17 can be connected in series at one of the switches.

The converter 1 also comprises a control member 38 connected to the switches 321, 322; 323, 324 of the switching cells 30, 31 and to the switches 15, 16 of the rectifier 44.

An inductive element 42 (inductor) is also arranged in series with the primary of said transformer 40. For example an inductor 42 is connected on the one hand to the output terminal 361 of the first switching cell 30 and on the other hand , at the primary of the transformer 40.

In addition, each switching cell 30, 31 is associated with a respective so-called "auxiliary" circuit 50. Each auxiliary circuit 50 comprises two first passive electrical elements 150 arranged in series between the supply and reference terminals 4. That is to say that the ends A and D of the auxiliary circuits 50 are connected to the 6 supply terminal while the ends C and F of the auxiliary circuits 50 connect to the reference terminal 4. Each auxiliary circuit 50 comprises a second passive electric element 250 having a first end connected between the first two elements 150 electrical (s) in series and a second end connected to the output terminal 361, 362 of the cell 30, 31 considered.

As illustrated in FIGS. 4 to 6, the first passive electrical elements 150 of the auxiliary circuits 50 comprise at least one of: a capacitance (FIGS. 4 and 5), a capacitance in parallel with a clipping diode (FIGS. 5 and 6). In addition, the second passive electrical element 250 of the auxiliary circuits 50 may comprise at least one of: an inductor (FIGS. 4 and 5), an inductance in series with two controlled switches arranged in parallel and in opposite directions (FIG. 6) . That is, for example, any of the auxiliary circuits 50 of Figures 4 to 6 may be connected to any of the switching cells 30 or 31 of Figure 3.

Thus, the connection terminals A and D (respectively B, F and L) of the auxiliary circuits 50 of FIGS. 4 to 6 are connected to the terminals A or D (respectively C or F) of the circuit of FIG. and D in FIG. 3 are electrically identical, likewise for the nodes C and F. The terminals C, E, P of the auxiliary circuits 50 of FIGS. 4 to 6 are connected to terminals B or E of FIG.

This auxiliary circuit architecture 50 makes it possible to avoid the loss of soft switching with a low load current (that is to say less than 30 to 50% of the nominal load cost). Indeed, this architecture makes it possible to use an auxiliary current source which is added to the current between the two switches 321, 322, 323, 324 considered so as to facilitate the discharge of the associated capacitors 341, 342, 343, 344. to the switches. With such a circuit, the energy necessary to create the soft switching conditions is not necessarily stored in the inductor 42 connected to the primary of the transformer 40. This necessary energy can therefore be minimized.

This eliminates the loss of duty cycle and significantly reduces the oscillations at the secondary of the transformer 40 (these oscillations are caused by a resonance between the inductor 42 and the capacitors 341, 342, 343, 344 of the rectifying diode junction at the secondary level (nb: these capacitors are intrinsic to the diodes and are not represented for the sake of simplification.) The converter thus has a better performance.

The voltages across the capacitors 341, 342, 343, 344 of the switching cells oscillate between + P / 2 and -P / 2. This produces an additional current whose shape depends on the type of auxiliary circuits. The following detailed description of the operation relates to a converter 1 having auxiliary circuits 50 respectively corresponding to those of Figures 4 and 5 for the cells 30, 31 switching.

As described in more detail below, the control member 38 is configured to control on the one hand the switches of the switching cells 30, 31 according to a "dual thryristor" ("LTD") logic and, on the other hand, to control the switches 15, 16 of the rectifier 44 in synchronism respectively with the switches 321, 322; 323, 324 switching cells. The switches 15, 16 of the rectifier 44 preferably operate as diodes (MOSFETs with their intrinsic diode) controlled in synchronism with the primary switches (that is, in synchronism with the switches 321, 322, 323, 324). switching cells 30, 31).

With a conventional control, the switches of the converter 1 would be controlled in an open loop so that one of the transitions of the control takes place at the appropriate moment (for example, the transistor is triggered during a time interval during which the antiparallel diode is driving). In this type of conventional control, the ignition pulse must be sent to the switch precisely. Failing this, a conventional hard switch occurs if the pulse is sent too late or to a sudden discharge of the switching aid capacitors in the switch if the pulse is sent too early. Thus, with this type of conventional control there may be a shift between the controls of the switches of the same arm with the appearance of short circuits that cause abnormal heating or even destruction. This problem was solved conventionally by adopting "idle time", determined to cover the delays between the orders according to the conditions of load, temperature, drift and dispersion.

On the contrary, the use of thyristor-dual type logic according to the invention makes it possible to overcome these dead time management constraints and allows reliable operation.

For the implementation of a dual thyristor logic, the switches 321, 322; 323, 324 are preferably equipped with a system for detecting and controlling the control (for example, detecting the conduction of the antiparallel diode activating the ignition control of the transistor in question).

As illustrated in FIG. 7, the control circuit of each switch 321, 322; 323, 324 may comprise a circuit 12 for detecting the saturation voltage of the switch (Drain-Source voltage of the MOS) and a circuit 22 of control, receiving on the one hand the voltage signal from the detection circuit and, on the other hand, the control signal 38.

More specifically, the switches 321, 322; 323, 324 switching cells 30, 31 are each equipped with a circuit 2 provided with a detector 12 of the saturation voltage V across the switch and a circuit 22 for controlling the switch receiving the signal of detector voltage 12. The control circuit 22 is also connected to the control member 38 and is configured to perform the priming of the switch only when the control member 38 initiates and simultaneously the measured voltage by detector 12 of the voltage is zero (V = 0) or close to zero, for example between zero and seventeen Volts. In other cases, the circuit provides a blocking control of the switch in question.

This type of command removes any forced priming of the switch. Indeed, at each switching, this circuit validates the boot of the switch when the voltage between the power electrodes thereof is canceled in a natural way. The switching losses at boot are thus completely eliminated.

This "thyristor-dual" function can be synthesized by associating a conventional MOSFET with a specific control logic.

During the blocking, the capacitor 341, 342, 343, 344 for switching assistance placed in parallel with the switch 321, 322; 323, 324 is intended to derive the current flowing through the switch, while limiting the voltage gradient appearing across the switch. This control prohibits the ignition of the switch until the capacitor is fully discharged.

In this way, the switch never undergoes the overcurrent discharge of the capacitor during these turns on. The control does not cause the cancellation of the voltage between the electrodes of the switch. This necessary cancellation must be caused by a circuit external to the switch, either naturally by the structure and the operating conditions of the converter, or via an auxiliary circuit adapted to cause this cancellation as is preferentially the case in the invention. .

Among the remarkable advantages related to this thyristor dual logic control (LTD), there can be mentioned the absence of management of dead time between the switches of the same cell thus making it possible to avoid fugitive short circuits at the level of the components. One of the controls of the switches of the same arm (cell) is replaced by a spontaneous commutation which is naturally complementary to the controlled commutation. Another advantage is the absence of boot control at the switches, avoiding any risk of converter arm short circuit 1. Indeed, during an inadvertent command or a particular malfunction, the biocage of a switch back the circuit in a freewheel sequence if it was in an active sequence. In freewheel mode it is possible to just try to block a switch while it is the connected diode in antiparallel that is busy (as in Figure 10). There is loss of control of the converter 1 in a sequence of freewheel or energy recovery, which limits the consequences on the assembly. On the contrary, with a conventional control according to the prior art for initiation and blocking, there is a risk of fatal short-circuiting of the voltage source through the switches of the cell.

The control according to a dual thyristor logic thus confers a "natural" operating reliability on the converter 1.

In the converter 1 according to the invention, MOSFET elements 15, 16 are used for the switches at the secondary level (rectifier 44). This makes it possible to reduce the conduction losses due to the threshold voltage of the rectifying diodes used according to the prior art.

These switches 15, 16 MOSFET are synchronously controlled (LCS) with the switches 321, 322; 323, 324 primary primary. For example, the first switch 15 (respectively the second 16) of the rectifier 44 is triggered at the same time as the first switch 321 (respectively the second 322) of the first switching cell 30. In addition, the first switch 15 of the rectifier (44 (respectively the second 16) of the rectifier 44 is controlled in blocking at the same time as the second switch 324 (respectively the first switch 323) of the second switching cell 31.

This control mode simplifies control and makes the operation of converter 1 more reliable. An example of operation will now be described with reference to FIGS. 8 to 13 (operating sequences likely to occur during its use). For reasons of simplification, the control member 38 and the auxiliary circuits 50 of the converter 1 of FIG. 3 are not shown in FIGS. 8 to 13. However, the auxiliary circuits 50 of the converter 1 of FIGS. those of Figures 4 and 5.

The operating principle of a quasi-resonant base structure known elsewhere, only the features of the invention will be highlighted. In particular, the following description

In a first sequence S1 illustrated in FIG. 8 and in the timing diagram of FIG. 14, the primary winding of the transformer 40 its terminals a constant voltage (Vi = P). The current in the output inductance 17 is brought back to the primary of the transformer via the first switch 15 of the rectifier 44. The current of the first switch 321 of the first switching cell 30 is equal to the sum of the current on the one hand. output inductance 17 brought back to the primary and, secondly, the current in the inductance 250 of the auxiliary circuit of this first cell 30.

Likewise, the current of the second switch 324 of the second cell 31 is the sum of a part of the current in the output inductance 17 brought back to the primary and, secondly, of the current in the inductance 250 of the auxiliary circuit. of this second cell 31. This first sequence S1 ends when the control member 38 blocks the first switch 321 of the first switching cell 30.

In a second sequence S2 (FIGS. 9 and 14), the first switch

321 of the first cell 30 is first locked and no other change of state of the other switches occurs during a time interval whose duration is the charge / discharge time of the capacitors 341, 342 of the first cell 30 switching.

Due to the blocking of this first switch 321, a current peak starts charging the first capacitor 341 associated with this first switch 321 and, simultaneously, discharges the capacitor 342 of the second switch 322 of the first cell 30. The current produced by the associated auxiliary circuit 50 assists in charging or discharging these capacitors 341, 342. The voltages V ₃₂ i, V ₃₂ 2 in volts at the respective terminals of the two switches 321,

322 of the first cell 30 are given respectively by the following equations:

J ■ l / max

J 250 ^{~ l}

2C 341

_l_ l / max

250

y * 322 = _Γ W_ _t ¹

L, ₃₄₂

with: P = the voltage at the terminals of the battery in Volt, l ₂ so the current in Ampere in the inductance 250 in auxiliary circuit 50 associated; l umax, the current in Ampere in the output inductor 17, C341, the capacitance of the capacitor 341 in Farads (F) associated with the first switch 321, C3 ₄ 2, the capacitance Farad (F) of the capacitor 342 associated with the second switch 322 t = _the current time of operation in second (s), _ m = the transformation ratio of the transformer (without unit).

Due to the presence of the capacitor 341, the voltage V ₃₂ i across the first switch 321 can only grow slowly, allowing the zero voltage or low voltage lock (limitation of the voltage gradient).

Meanwhile, the capacitor 342 of the second switch 322 discharges during this interval. As soon as the latter is completely discharged, the diode 102 which is antiparallel with the second switch 322 closes spontaneously, thus allowing continuity of the current. The voltage V ₃₂₂ at the terminals of the second switch 322 maintained at zero thus creating spontaneous ignition conditions.

In a third sequence S3 (FIGS. 10 and 14), the diode 102 of the second switch 322 is on. As the second switch 324 of the second cell 31 is still on, the primary winding of the transformer 40 sees a zero voltage after the cancellation of the voltage V ₃₂₂ across the second switch 322 of the first cell 30. The second switch 16 of the rectifier 44 can be controlled at the same time as that of the second switch 322 of the first cell 30.

The primary current is kept constant (with losses). The duration of this interval is determined by the phase shift time required for the power setting. The inductance 250 of the auxiliary circuit 50 of the first cell 30 sees a constant positive voltage equal to P / 2. The current l ₂ so that passes through this inductance 250 begins to grow linearly from the negative peak value according to the following expression:

P

- *? sn 9 T t 1

½50

: Where: P, the voltage across the battery Volt, l ₂ so the current in the inductance 250 of the auxiliary circuit 50 associated in Ampere (A); 1250max the maximum current in the inductor 250 of the auxiliary circuit 50 associated in Ampere (A), L ₂₅₀ the value of the inductance of the auxiliary circuit 50 associated with the first cell 30 in Henry (H), t = _the current time of operation in Second (s), _ m = Transformer transformer ratio (without unit).

This third sequence S3 terminates at the controlled blocking of the second switch 324 of the second cell 31.

At the beginning of the fourth sequence S4 (FIGS. 11 and 14), the second switch 324 of the second cell 31 is controlled to block. No other switches change state during this interval. The duration of this interval is defined by the charging / discharging time of the capacitors 343, 344 of the second cell 31. When this second switch 324 of the second cell 31 is blocked, the current l ₂ so the inductance of the auxiliary circuit 50 of the second cell 31 reaches its positive peak value somax- Like the second sequence S2, this current begins then charging the second capacitor 344 of the second cell 31 and discharging the first capacitor 343 of the second cell 31. The voltage V ₃ across the second capacitor 344 of the second cell 31 begins to grow from zero while the voltage V ₃₄ across the first capacitor 343 of the second cell begins to decrease from the value of P. Thanks to the second capacitor 344 of the second cell 31, the voltage V ₃ across the second capacitor 344 can only grow slowly thus ensuring limited voltage gradient blocking switching. The gradual discharge of the first capacitor 343 of the second cell reduces to zero the voltage V ₃₂₃ across the first switch 323 of the second cell 31 during this interval. This allows the spontaneous ignition of the diode 103 associated with the first switch 323 of the second cell 31.

At the beginning of the fifth sequence S5 (FIGS. 12 and 14), the diode 103 associated with the first switch 323 of the second cell 31 spontaneously primes.

Due to the character of this diode 103, the inductance 250 of the auxiliary circuit 50 of the second cell 31 sees a constant negative voltage -P / 2. The current l ₂ so in Ampere (A) in this inductance then begins to decrease linearly according to the following equation:

P

⁷ 250 - _{1 T} ^{Î + 7} 250 _max

½50

with: P, the voltage at the terminals of the battery in Volt, l250max the maximum current in the inductance 250 of the auxiliary circuit 50 associated in Ampere (A), L ₂ so the value of the inductance of the auxiliary circuit 50 associated with the second cell 31 in Henry (H), t = the current running time in Second (s).

During this interval, the currents o ₂ and ₀ 3 (in ampere) in the second diode 102 of the first cell and respectively in the first diode 103 of the second cell 31 are determined by the following equations:

with: P, the voltage at the terminals of the battery in Volt, l250max the maximum current in the inductor 250 of the auxiliary circuit 50 associated in Ampere (A), L ₂ so the value of the inductance of the associated auxiliary circuit 50, bsomax the maximum current in the inductor 250 of the auxiliary circuit 50 associated in Ampere (A), l_i ₇ the value of the inductance (in Henry H) output 17, ₇ the value of the current in the output inductance 17, t = _the current running time in Second (s), m the transformer transformation ratio (without unit),> Vo = the voltage at the load terminals in Volt (V), fc = the operating frequency of the converter in Hertz (Hz).

When these currents I 102 and _O 3 cancel each other out, the corresponding diodes 102 and 103 spontaneously lock and the corresponding switches 322 323 ensure the continuity of the current if their controls are present (soft switch operation condition). This current reversal is done naturally with switches controlled by a dual thyristor type logic.

During a sixth sequence S6 the first switch 15 of the rectifier 44 is blocked. Indeed, when the primary current (in the inductor 42 connected to the primary of the transformer 40) reaches the level of the charge current brought back to the primary, the first switch 15 of the rectifier must be blocked to prevent a malfunction of the converter 1. In the converter 1 according to the invention, this blocking is realized at the same time as the blocking of the second switch 324 of the second switching cell 31. The intrinsic diode of the MOSFET then ensures the continuity of the current for a very short period of time (of the order of the duration of the fourth S4 and fifth S5 sequences). This has no significant effect on conduction losses. The control is then simple and reliable without risk of malfunction of the converter.

The second switch 322 of the first cell 30 and the first switch 323 of the second cell 31 are on while the first switch 321 of the first cell 30 and the second switch 324 of the second cell 31 are blocked (FIGS. 13 and 14). . The second switch 16 of the rectifier 44 is passing. This sequence is symmetrical to the first sequence S1 described above (initial sequence).

Then, another operation cycle symmetrical to the five preceding sequences S1 to S6 can begin (symmetrically because the passing / blocked roles above the first switches 321, 323 of the two cells 30, 31 are reversed and the passing / blocking roles of the second switches 322, 324 of the two cells 30, 31 above are also inverted). The converter 1 can then periodically repeat the operating process.

Fig. 14 illustrates the waveforms in the different components (switches 321, 322, 323, 324, 15, 16) as a function of the operating sequences (S1 to S6).

This architecture and its smooth synchronous rectifying converter operation allows operation with high reliability and high efficiency.

In particular, soft switching has been extended over the entire operating range by the use of auxiliary switching circuits. The control of the switches by a dual thyristor logic makes it possible to overcome the thorny problem of adjusting idle time and to increase the reliability of the converter 1. Synchronous rectification, which consists of using MOSFETs instead of rectifying diodes, reduces conductive losses.

The synchronous control of the MOSFETs (switches 15, 16) of the synchronous rectifier 44 makes the operation reliable.

A sequential study of the operation of this converter 1 made it possible to show the functional characteristics of this converter. Thus, this converter 1 can operate at high switching frequencies (between 100KHz and 200KHz), with high reliability and reduced overall losses. But reliability and dependability are probably the strengths of this converter.

Of course, to further reduce the losses, several MOSFETs can be paralleled for each switch at the primary or secondary level of the converter 44.

The use of such a structure is particularly advantageous in fuel cell applications requiring high efficiency and reliability.

For example, one to three fuel cells of sixty elementary cells each can be used to provide about 2.15 kW per module at nominal under the conditions (ambient 45 ° C and or 2000 meters altitude). A fuel cell can be installed in each battery module ("FCM" = "Fuel Cell Module") and the voltage supplied by it will be transformed into a voltage of 48V stabilized by the converter 1 DC / DC.

The output power of the converter 1 can be used to power the battery charge, a continuous bus that can vary between 42 and 55V depending on the state of charge of the batteries (Figure 2), the auxiliary FCM battery modules and the fuel cell system ("FCS" = "Fuel Cell System") and the provision of power to the customer.

The DC / DC converter 1 of each module may, for example, deliver a power of 2 KW maximum and a maximum current of 50 A maximum for 42 V on the bus.

Such output sizing will result in potential power consumption on the 2150 W stack for 93% converter efficiency. For such a power, the maximum expected current at the input of the converter will be for a voltage of 27V (minimum voltage given for end of life of the battery) of 80A.

The input voltage range of the converter 1 can be between 27 and 60Vdc for a maximum current of 80A and a maximum power of 2150W. At the output of the converter, the voltage can be between 42 and 56Vdc for a maximum current of 50A and a maximum power of 2000W.

Claims

1. An electricity supply device comprising a fuel cell (100) and a soft switching type converter (1) for transforming the electrical energy delivered by the battery, the converter (1) comprising two ends (6, 4) of connection to a voltage source (P) delivered by the battery, the two connection ends respectively defining a DC voltage supply terminal (6) and a reference terminal (4), the converter (1) also comprising two cells (30; 31) each having an even number of switches (321, 322, 323, 324) connected in series between the supply (6) and reference (4) terminals, each cell (30, 31) switching circuit comprising a respective output terminal (361 362) connected between the two halves of switches in series of said cell (30; 31) considered, each switch (321, 322, 323, 324) being connected in parallel with a part in an elem respective capacitive means (341, 342, 343, 344) and a respective diode (101, 102, 103, 104), each switching cell (30, 31) being associated, i.e., connected to a respective auxiliary circuit (50) comprising two first passive electrical elements (150) arranged in series between the supply (6) and reference (4) terminals and a second passive electrical element (250) having a first end connected between the two first elements (150) electrical (s) in series and a second end connected to the output terminal (361, 362) of the cell in question, the converter (1) further comprising a transformer (40) of which the primary is connected to the output terminals (361, 362) of the switching cells (30; 31) and whose secondary is connected to a rectifier (44) of the switch type (15, 16) MOSFET, the converter (1) also comprising a control member (38) connected to the switches (101, 102, 103). , 104) of the switching cells and the switches (15, 16) of the rectifier (44), the control member (38) being configured to control, on the one hand, the switches of the switching cells (30, 31) (101). , 102, 103, 104) according to a logic of the "dual thryristor" ("LTD") type and, secondly, to control the switches (15, 16) of the rectifier (44) in synchronism with the switches (101, 102, 103, 104) of the switching cells.

2. Device according to claim 1, characterized in that the switches (321, 322, 323, 324) of the switching cells (30, 31) are of the MOSFET type with a blocking type operation controlled by the Zero Voltage Switching (ZVS) control element (ZVS).

3. Device according to claim 1 or 2, characterized in that the first passive electrical elements (150) of the auxiliary circuits (50) comprise at least one of: a capacity, a capacitance in parallel with a diode d clipping, the second passive electrical element (250) of the auxiliary circuits (50) comprising at least one of: an inductance, an inductance in series with two controlled switches arranged in parallel and in opposite directions.

4. Device according to any one of claims 1 to 3, characterized in that the switches (101, 102, 103, 104) of the switching cells (30, 31) are each equipped with a circuit (2) provided with a detector (12) of the saturation voltage (V) across the switch and a control circuit (22) of the switch receiving the voltage signal of the detector (12), said control circuit (22) being connected to the control member (38) and being configured to initiate the switch only when the control member (38) activates a priming and, simultaneously, the voltage measured by the detector (12) of the voltage is zero (V = 0) or between zero and seventeen Volts.

5. Device according to any one of claims 1 to 4, characterized in that an inductive element (42) is arranged in series with the primary of said transformer (40).

6. Device according to any one of claims 1 to 5, characterized in that each switching cell (30; 31) comprises first and second switches (321, 322, 323, 324) connected in series, the rectifier ( 44) comprising a first (15) and a second (16) MOSFET type switch.

7. Device according to claim 6, characterized in that the control member (38) is configured to firstly control the priming of the first switch (15) of the rectifier (44) at the same time as the priming of a first switch (321) of a first switching cell (30) and, secondly, for controlling the blocking of the first switch (15) of the rectifier (44) at the same time as the blocking of the second switch (324). ) of the second switching cell (31).

8. Device according to claim 6 or 7, characterized in that the control member (38) is configured to firstly control the ignition of the second switch (16) of the rectifier (44) at the same time as the initiating the second switch (322) of the first cell (30) of switching and, secondly, for controlling the blocking of the second switch (16) of the rectifier (44) at the same time as the blocking of the first switch (323) of the second cell (31) switching.

9. A method of supplying electricity using a device according to claims 7 and 8 taken in combination, comprising:

a first sequence (S1) in which the first switch (321) of the first switching cell (30), the second switch (324) of the second switching cell (31) and the first switch (15) of the rectifier ( 44) are passing while the second switch (322) of the first switching cell (30), the first switch (323) of the second switching cell (31) and the second switch (16) of the rectifier (44) are blocked and then

a second sequence (S2) in which the first switch (321) of the first switching cell (30) is switched to a blocking state, and then

a third sequence (S3) during which the second switch (16) of the rectifier (44) is controlled at initiation, then

a fourth sequence (S4) during which the second switch (324) of the second switching cell (31) is switched to a blocking state,

a fifth sequence (S5) during which the diode (103) associated with the first switch (323) of the second switching cell (31) spontaneously initiates and then the diodes (102, 103) associated with the second switch (322) of the first switching cell (30) and the first switch (323) of the second cell (31) spontaneously lock, then

A sixth sequence (S6) during which the first switch (15) of the rectifier (44) is switched to its off state.

10. A method according to 9, characterized in that it then realizes again the six sequences (S1 to S6) symmetrically with:

the second switch (322) of the first cell (30) and its associated members taking the role (on / off ...) of the first switch (321) of the first cell (30) and vice versa in the sequences (S1 to S6); and

the first switch (323) of the second switching cell (31) and its associated members taking on the role (on / off ...) of the second switch (324) of the second switching cell (31) and vice versa, and the second switch (16) of the rectifier (44) taking (on / off) the first switch (15) of the rectifier (44).