WO2014001406A1

WO2014001406A1 - Method for analysing a scattering sample using a time-resolved measurement and associated device

Info

Publication number: WO2014001406A1
Application number: PCT/EP2013/063426
Authority: WO
Inventors: Jean-Michel Tualle; Eric TINET; Olivier Cussenot
Original assignee: Universite Paris 13
Priority date: 2012-06-28
Filing date: 2013-06-26
Publication date: 2014-01-03
Also published as: FR2992728A1; FR2992728B1

Abstract

The invention relates to a method for analysing a scattering sample (10) using a time-resolved measurement of the light scattered by the sample (10), said method comprising steps of: illuminating the sample (10) at an illumination point (P1) located on the sample (10), using a temporally coherent incident light beam, the wavelength of which is modulated according to a function in which the period is the modulation period (T); and detecting, with a photodetector (44), during a first time interval, the time variation in the intensity of the light scattered by the sample at an exit point (P2) located on the sample (10) at a distance (d) less than 1 millimetre from the illumination point (P1).

Description

METHOD FOR ANALYZING A DIFFUSING SAMPLE BY RESOLVED MEASUREMENT

The present invention relates to a method for analyzing a scattering sample by time-resolved measurement of the light scattered in this sample. The present invention also relates to a device adapted for implementing the method. The invention is particularly applicable for monitoring the oxygenation and perfusion of an organ of a human being.

In the medical field, it is known to use light, especially with wavelengths located in the infrared range (in the vicinity of 800 nanometers), in order to study the nature of biological tissues quite profoundly.

However, biological tissues are highly diffusing at optical wavelengths, which tends to obscure any information about their internal structure. However, this structure influences the data and it is desirable to estimate this influence.

For this, it is interesting to study the dynamics of light propagation in biological tissues. For this purpose, it is known to send a very short pulse of light on the medium to be studied and to carry out determined measurements in time of the diffuse light. Temporal resolution provides valuable insights into the structure of the environment.

Depending on the case, the measurements are made in reflectance or transmittance. Reflectance measurements are used in particular to determine the rate of oxygenation of blood hemoglobin.

However, the temporal resolution must be performed on a very short time, of the order of a few tens of picoseconds (ps). Also, obtaining such a resolution involves the use of very sophisticated and expensive technologies.

In known observation devices, the source producing the light pulse is adapted to provide a very brief pulse.

For example, in the device described in US-A-5 692 51 1, the light source used is a titanium-sapphire laser. Such a laser is very expensive. In addition, the detection implemented is tricky. It is proposed the use of an avalanche photodiode connected to an electronic time gate to determine the number of photons transmitted during a given period of time. The avalanche photodiode has a relatively long response time which deteriorates the potential contrast of the time gate. More generally, in time-resolved, diffuse light measurement analysis devices, a source is commonly used to produce short pulses such as pulse lasers which are expensive.

In addition, the means used for the detection of light pose problems specific to each of them that do not allow satisfactory use of the device.

Among these signal detection means, there are: fast detectors with electronic signal processing from the detector whose relatively long response time considerably deteriorates the contrast of the time gate; time-amplitude converters in the photon counting regime which involve decreasing the signal to fit in the appropriate regime, which is not compatible with the short acquisition times; optical doors with non-linear effects that require powerful lasers with low rates of the order of 10 Hz, which considerably limits the sensitivity; Slot scanning cameras that have very good temporal resolution (about 2 ps) but whose dynamics are not very important; and ultra-fast intensified cameras whose temporal resolution is not very good (of the order of 80 ps) for a low repetition rate (of the order of 1 kHz).

This is why in FR-A-2 809 180 there is provided a device for analyzing a scattering sample by measurement resolved in time of the diffuse light in this sample. The device described in this patent application uses a spectral modulation of light beams and measures time correlation functions of the scattered light for a transit time determined in particular in many applications for medical diagnosis.

However, the device disclosed in the aforementioned patent application remains complex implementation, especially because it involves the use of an interferometer.

There is therefore a need for a device for analyzing a scattering sample by resolute measurement in time of diffuse light are the implementation is easier.

For this purpose, there is provided a method for analyzing a scattering sample by time-resolved measurement of scattered light in this sample, the method comprising the steps of illuminating the sample at an illumination point located on the sample. sample with an incident beam of temporally coherent and wavelength-modulated light according to a function having a period of the modulation period and detecting on a photodetector during a first time interval the temporal evolution of the intensity of the light diffused by the sample at an exit point located on the sample at a distance less than 1 millimeter from the point of illumination. The method also comprises the steps of multiplying the signal detected by the photodetector by a reference signal generating a time gate centered on a predetermined delay to obtain a multiplied signal, extracting demodulated components of the signal detected from the multiplied signal, and applying a non-linear function to the extracted demodulated components.

According to particular embodiments, the method comprises one or more of the following characteristics, taken in isolation or in any technically possible combination:

the extraction step comprises the steps of cutting the first time slot as a function of the modulation period to obtain a succession of integration intervals, and calculating a demodulated component of the signal detected for each integration interval , the demodulated component being defined as the integral of the signal multiplied over the integration interval considered.

at the step of division, each integration interval corresponds to a time interval equal to one half-modulation period.

at the step of illuminating the sample, the incident beam comes from an optical fiber, the optical fiber also serving to collect the light scattered by the sample at the exit point and to bring the collected light to the photodetector .

the fiber is provided at one end with a diffusing probe.

- the non-linear function is the mean of the square of the demodulated component or the average of the product of two distinct demodulated components.

The invention also relates to a device for analyzing a scattering sample by time-resolved measurement of diffuse light in this sample, comprising means for illuminating the sample with an incident beam of temporally coherent light and modulated in length. wave at an illumination point on the sample. The device comprises a photodetector detecting the temporal evolution of the intensity of the light scattered by the sample at an exit point situated on the sample at a distance of less than 1 millimeter from the illumination point and a suitable processing unit. for carrying out the analysis of the signal detected by the photodetector to obtain a temporal correlation of this signal.

According to particular embodiments, the device comprises one or more of the following characteristics, taken separately or in any technically possible combination:

the device is adapted for implementing the method as described above,

the device comprises a circuit comprising the photodetector, a multiplier, an integrator, an analog memory and a unit for estimating a correlation time of the signal detected by the photodetector from the stored components.

The invention also relates to the use of the device as previously described to monitor the oxygenation and perfusion of an organ of a human being.

Other features and advantages of the invention will appear on reading the following detailed description of embodiments of the invention, given by way of example only and with reference to the drawings which are:

FIG. 1, a schematic view of an analysis device according to the invention;

FIG. 2 is a flowchart illustrating the structure of the information processing unit of the detector of the installation of FIG. 1;

FIGS. 3 and 4, curves illustrating the shape of the signals that can be obtained at different points of the information processing unit of FIG. 2;

FIG. 5 is a flowchart illustrating a variant of the structure of the information processing unit of the detector of the installation of FIG. 1; FIG. 6 is a block diagram illustrating the structure of a signal processing circuit;

FIG. 7 is a block diagram illustrating an exemplary embodiment of a high-pass filter;

FIG. 8, a block diagram illustrating a photodetector;

FIG. 9 is a block diagram illustrating an amplification structure; FIG. 10 is a block diagram illustrating a demodulation structure comprising a multiplier and an integrator;

FIG. 11, a block diagram illustrating an embodiment of the resistive devices of the structure of FIG. 10;

FIG. 12 is a block diagram illustrating the structure of means for generating the control voltages FIG. 10;

FIG. 13 is a block diagram illustrating the structure of a reference register;

Fig. 14 is a block diagram illustrating the structure of a data register;

FIG. 15 is a block diagram illustrating the differential input multiplier, and

FIG. 16, a schematic view of a measuring device according to the invention. The device 8 represented in FIG. 1 allows the reflectance analysis of a scattering sample 10 by a time-resolved measurement of the scattered light. With the aid of the device 8, it is possible to reconstruct the temporal evolution of the diffuse energy with temporal resolutions of the order of 200 ps to 20 ps.

The device 8 comprises means 12 for illuminating the sample 10.

More specifically, according to the example of FIG. 1, the means 12 for illuminating comprise a temporally coherent and wavelength-modulated light source 14 so as to simulate, by an appropriate processing of the detected signal, an incoherent source.

The source 14 comprises a laser cavity 16 connected to a control unit 18 able to provide the wavelength modulation of the beam produced by the laser cavity 16. The laser cavity 16 is for example a laser diode or a laser diode mounted in extended cavity. The laser cavity 16 is adapted to produce a temporally coherent monochromatic light beam.

The central wavelength λ ₀ of emission of the source 14 is chosen according to the applications. According to a particular mode of implementation of the device 8, several different central wavelengths are successively implemented, thus making it possible to perform spectroscopic measurements. The wavelengths are for example equal to 780 nm and 850 nm for the analysis of the oxygenation rate of the tissues.

The control unit 18 is adapted for periodic modulation of the wavelength of the beam produced by the laser cavity 16. This modulation is advantageously a sinusoidal modulation. The amplitude noted δλ of this modulation is of the order of a few hundredths of a nanometer.

The temporal resolution At of the device obtained for a modulation amplitude δλ of a few hundredths of a nanometer is approximately given by the following formula: At =

mô

where λ is the emission wavelength of the source and c is the celerity of light.

A temporal resolution At of the order of a few tens of picoseconds is thus obtained.

Advantageously, the modulation of the light produced by the laser cavity 16 is carried out continuously, avoiding mode jumps.

The modulation frequency f is chosen so high that the scattering sample can be considered as immobile during the modulation period. The modulation period corresponds to the inverse of the frequency of modulation. The modulation frequency is advantageously between 1 kilohertz (kHz) and 100 kHz. For example, the modulation frequency is chosen equal to 1 kHz.

The optical beam from the source 14 is injected into an optical fiber.

The injection into the fiber 20 is carried out using a focusing lens 22. A lens 32 may advantageously be used to adapt the divergence of the beam to this focus.

The fiber 20 makes it possible to conduct the beam coming from the source 14 towards the sample 10. A proximal end 20P of the fiber 20 and a distal end 20D of the fiber 20 can thus be defined with respect to the sample 10. distal end 20D is the end of fiber 20 which is closest to sample 10.

In the example of Figure 1, the optical fiber is multimode.

The fiber 20 makes it possible to illuminate the sample 10 at a point called P1.

The light thus introduced into the sample 10 is diffused therein for example along the path 23 shown in dashed lines in FIG.

The scattered light may exit the sample at a point P2 on the sample

10 at a distance d less than 1 millimeter from the illumination point P1. The distance d is visible in FIG.

According to this example, the distance d is sufficiently small for the fiber 20 to harvest some of the light scattered by the sample at the point P2.

In addition, in the illustrated example, the fiber 20 is provided at its distal end 20D with a diffusing probe 24. The probe 24 makes it possible to protect the sample 10. The probe 24 makes it possible to obtain a homogeneous illumination of the sample 10. The probe 24 also makes it possible to accentuate the importance of the short trajectories, of shorter duration than the temporal resolution of the device (a few tens of picoseconds) in the scattered light that is harvested by the fiber 20.

The scattered light collected by the fiber 20 is directed towards a separator 26 allowing the separation of the light from the laser source 14 injected into the fiber 20 of the scattered light which leaves the fiber 20. By way of example, the separator 26 is a polarization splitter cube.

The device 8 further comprises a half-wave plate 28 placed between the laser source 14 and the polarization splitter 26. The half-wave plate 28 serves to control the polarization of the light from the source 14. This makes it possible to control the polarization of the light incident on the polarization splitter 26. Thus, the light injected into the fiber 20 can thus be maximized.

Alternatively, the separator 26 may be any other beam separation system, such as for example a fiber separator. In this case, the system of lenses used for the injection of the laser beam into the fiber is different from the optical system for conveying the diffuse light on the photodetector 44.

The device 8 also comprises an optical isolator 30 placed between the source 14 and the separator 26. This optical isolator 30 serves to block the diffused light to avoid any reinjection of light into the source 14. Such reinjections can, in fact, disturb the operation of the source 14.

The beam obtained after diffusion in the sample 10 is sent to means 42 for detection and analysis.

These means 42 comprise a photodetector 44. The numerical aperture of the detection system in the case of FIG. 1 is not fixed by the photodetector 44 but by the lens 22. This lens has, in effect, the role of increase the numerical aperture of the detection system.

The photodetector 44 is connected to an information processing unit 46 whose principal means are described with reference to FIG.

Advantageously, the characteristics of the optical elements placed downstream of the sample 10, are chosen so that the coherence surface of the scattered beam at the photodetector 44 is of the same order of magnitude as the active surface of this photodetector 44.

In the embodiment described, the information processing unit 46 is formed of a set of analog electronic circuits adapted to perform the functions described with reference to FIG. 2.

As input, the unit 46 comprises a high-pass filter 50 for filtering the signal collected by the photodetector 44. The cutoff frequency of the filter 50 is of the order of one to ten times the frequency f of modulation of the light emitted by the source 14. In this case, the cutoff frequency is of the order of 100 kiloHertz.

The main purpose of the high-pass filter 50 is to suppress the DC component of the signal. It also ensures the elimination of low frequency noise and the possible spurious effects related to source wavelength modulation 14. The useful signal obtained at the output of the filter 50 has a frequency much higher than the modulation frequency .

The signal obtained at the output of the filter is advantageously amplified by an optional amplifier 51.

Although the filter 50 is advantageously implemented, the filter 50 may be omitted. A multiplication stage 52 receives the signal collected by the photodetector 44, if necessary filtered and amplified. I l ensures the multiplication of the filtered signal by a reference signal denoted Ref (t, xo). The signal Ref (t, xo) comes from a signal generation unit 54. The unit 54 is, for example, a programmable signal generator or an analog electronic circuit consisting of a set of oscillators whose characteristic parameters are adjustable.

The output of the multiplication stage 52 is connected to a stage 56 for extracting the DC component of the signal, that is to say for calculating its average value.

The signal Ref (t, io) is such that the DC component of the product of the reference signal Ref (t, xo) and of the signal measured by the photodetector 44 corresponds to a demodulated signal of the latter. The combination of the signal delivered by the detector and the reference signal Ref (t, xo) acts as a time gate centered on the delay τ ₀ .

The signal Ref (t, io) is expressed for example in the form:

Ref (t, r ₀ ) = δλ ₀ τ ₀ ∞ $ (2) + φ Λ in which we have:

a spectral modulation of the light emitted by the source expressing the form δλ {ί) = δλ ₀ cos (2; zt);

the amplitude A of the signal Ref (t, io)

n, an integer greater than one;

- φ, a phase shift of the signal Ref (t, xo).

Several values of A, n and φ are possible. For example, the following values are suitable: A = 1, n = 4 and (p = 0.

According to an alternative embodiment, the signal Ref (t, io) is produced by using an optical interferometer having a delay between the two arms, by filtering the DC component of the signal detected at the output of the interferometer and multiplying the result by a another function generated by an analog electronic unit, this function being canceled at the extremes of the modulation 5A (t). This other function will be for example the function sin ⁿ (2TTft).

The extraction by the stage 56 of the DC component of the product calculated by the stage 52 is obtained for example by implementation of a low-pass filter. In this case, the extracted signal is a continuous signal whose evolution is a slow evolution linked to the movements of the sample 10.

As a variant, the extraction of the DC component is provided by an integrating device which integrates the signal obtained at the output of stage 52 for exactly one half-period T / 2. In this case, the signal from the stage 56 is sampled after each integration of this type.

The information processing unit further comprises a stage 58 for applying a non-linear function to the DC component of the signal obtained at the end of stage 56. This function is, for example, an elevation at square, or any other even power, or the application of the absolute value function.

The application of such a function leads to considering the only amplitude of the signals, independently of their sign and to allow that their average is not null.

The result of the application of the temporal gate implemented during the multiplication performed by stages 52 to 58 is illustrated in FIG. 3. The signal represented in FIG. 3 is the DC component obtained at the output of stage 58. after applying a non-linear function, as a function of the value of the predetermined delay τ ₀ . This signal corresponds only to the diffuse light from the sample 10

The next stage noted 60 of the unit 46 is adapted to average the signals obtained at the output of the stage 58 for a predetermined period. The average calculated by stage 60 is determined over a period of the order of one second. For this purpose, the stage 60 comprises for example a low-pass filter.

Alternatively, the calculation of the average is replaced by a simple summation of the sampled signals from the stage 58 or any other form of linear combination performed on these signals.

The signal obtained at the end of the stage 60 for a given value of the predetermined delay τ ₀ is a measure of the time-resolved diffuse intensity cross-correlation function, for a transit time τ ₀ fixed and determined by the reference function. In the context of the invention, the diffuse light is collected near the illumination point P1 and corresponds essentially to short paths, and this property can be advantageously accentuated by the scattering probe 24. In this case, the function of Cross-correlation of the time-resolved diffuse intensity can be identified at the time-resolved diffuse intensity, thus allowing time-resolved measurement of diffuse light.

According to a particular embodiment, the device 8 comprises several information processing units 46, each associated with a delay value τ ₀ thus making it possible to obtain measurements of the diffuse energy for different delays τ ₀ . The information processing unit 46, described above, consists of an analog electronic device. However, in a variant, this information processing unit 46 comprises an analog high-pass filter 50 at the output of which an analog / digital converter is provided. The set of treatments performed by stages 52 to 60 is then performed completely digitally by implementing an algorithm adapted in a computer. In this case, the different number of values τ ₀ for which the calculation is performed can be very large.

During the measurement, the signal obtained at the output of the stage 58 fluctuates on the same time scale as the signal obtained at the output of the stage 56. These fluctuations are due to the movements in the scattering medium itself when this medium is a biological environment. These movements induce fluctuations on timescales of the order of a millisecond.

However, these fluctuations are desirable to allow the average value obtained at the output of the stage 60 to be as accurate as possible. Indeed, these fluctuations allow the measurement of different values to make the average. This average will be all the closer to its theoretical value, which is the diffuse intensity resolved in time, as the fluctuations make it possible to obtain a large number of different values.

The device 8 which has just been described is suitable for the implementation of a method for analyzing the scattering sample 10 by time-resolved measurement of the light scattered in this sample 10.

This method comprises a step of illuminating the sample 10 at the point P1 with an incident beam of temporally coherent light and wavelength-modulated according to a function having for period the period of modulation T.

The method also comprises a detection step on the photodetector 44 during a first time interval of the temporal evolution of the intensity of the light scattered by the sample 10 at the point P2.

The method also comprises a step of multiplying the signal detected by the photodetector 44 by the reference signal Ref (t, io).

The method also comprises a step of extracting the demodulated components of the detected signal.

The method also includes applying a non-linear function to extracted demodulated components.

FIG. 5 illustrates a variant of the analysis method according to the invention intended to obtain a time-resolved measurement of a quantity relative to the diffuse light in the sample to be analyzed. This variant differs from the process described in FIGS. 1 to 4 only by the structure of the information processing unit 46 at its stages 56 and 58.

The output of the multiplication stage 52 is connected to a stage 56 'of extraction of the respective continuous components of the signals obtained at the output of the stage 52 for a determined duration, substantially equal to that used for the calculation of the calculated average by the stage 60 in the method described with reference to FIG. 2. The continuous components thus obtained at the stage 56 'are denoted S ,, where i designates the i ^th signal at the output of the stage 52 during the above-mentioned duration . The means implemented to obtain each continuous component S, are similar to those explained above.

The output of this stage 56 'is connected to a stage 58' for applying a non-linear function g to at least two variables taken among the continuous components S, coming from the stage 56 '. It should be noted that the term "nonlinear" is considered here in a broad sense, that is to say, in the case of multivariate functions, only linear functions are excluded from their variables taken together under vector form, such as the "sum of two vectors" function.

This nonlinear function is, for example, a product function applied to two distinct S values. Thus, we can consider the function g (S ,, S _{i + a} ) = S, x S _{i + a} , where a is a nonzero positive integer and the sign "x" corresponds to the mathematical product. Such a function is adapted to study the evolution of the value S _{i + a} with respect to the value S _i . Indeed, as explained above, a biological medium is the seat of many movements and, despite the precautions of use including choosing a modulation frequency of the laser source 14 sufficiently high to consider the scattering sample 10 as stationary during the modulation period, the signal S _{i +} i recorded following a signal S, is slightly different from this signal S ,. The signal S _{i + 2} is more so. More generally, we note a decorrelation of the signals S _{i + a} and S, for a> 1. The evolution of this decorrelation as a function of a and of τ ₀ is able to provide useful data on the nature and the intensity movements in the sample 10.

For this purpose, the stage output 58 'is connected to the average stage 60 of the signals obtained at the output of the stage 58', which average is a function, on the one hand, of the delay τ ₀ as previously, and on the other hand the value of the integer a. One way to study the decorrelation of the signals S, between them is to follow the evolution of this average from the stage 60, for a crescent and fixed τ ₀ for example.

According to one variant, the processing unit 46 is adapted to implement another analysis method comprising an estimation of the correlation of the signal detected by the photodetector 44. In this case, the processing unit 46 is the circuit 102 shown in FIG. 6. The various elements of this circuit 102 are more specifically described in FIGS. 7 to 15. This circuit 102 is given by way of example only, knowing that this circuit 102 corresponds to the analog electronic processing circuit of a light signal described in the application FR-A-2 958 430. Thus, the circuit 102 may be according to any one of the various embodiments described in this application for this circuit 102.

The circuit 102 comprises a photodetector 106 adapted to produce an electrical signal 108 from the light signal 104.

According to a particular embodiment of the invention, the circuit 102 comprises an amplifier 1 10 for amplifying the electrical signal 108. The circuit 102 further comprises a multiplier 1 12 adapted to multiply the electrical signal 108, possibly amplified by the amplifier 1 10, by a reference signal f (t) 1 14 of constant sign to obtain a multiplied signal 1 16.

The multiplier 1 12 is connected to an integrator 1 18 adapted to integrate the multiplied signal 1 16 over at least one time interval to obtain at least one integrated signal.

In FIG. 6, two integrated signals 120 and 122 are represented at the output of the integrator 18. The signal 120 is a reference voltage obtained by a particular choice of the integration time interval and the reference signal 1. 14.

The signal 122 is stored in a memory 124 comprising a plurality of data registers Reg_1 126, Reg_2 128, Reg_N 130 and a reference register RegRef 132 adapted to store the reference voltage 120. By way of example, the integrated signal 122 is stored in the Reg_2 data register 128.

The analog memory 124 allows random write access and two random read accesses so that the contents of two different or identical data registers can be read simultaneously.

A differential input multiplier 134 is adapted to multiply the contents of two different or identical registers of the memory 124, for example the contents of the registers 128 and 130.

The result 136 of the multiplication performed by the differential input multiplier 134 represents the temporal correlation of the measurements recorded on two different or identical integration time intervals, which makes it possible to estimate the temporal correlation of the light signal 104. The contents of the register reference reference RegRef 132 determines the reference value of the differential input of the differential input multiplier 134, the result of the multiplication being in this case: Result = Co + K (Reg_i - RegRef) x (ReJ - RegRef), Co and K being two constants depending on the structure of the differential input multiplier 134 and Co being zero.

According to one embodiment of the invention, the photodetector 106 is a simple photodiode.

According to another preferred embodiment of the invention, the photodetector 106 comprises a photodiode associated with a high-pass filter making it possible to filter the low frequency components of the detected signal, in particular the DC component, in order to transmit only the high frequency components which contain the relevant information and thus allow greater momentum.

FIG. 7 shows an exemplary embodiment of such a high-pass filter associated with a photodiode 140. This high-pass filter comprises a voltage-controlled current source 142 which serves to compensate for the low frequencies of the photo-current generated by the photodiode 140.

The high-pass filter further comprises, optionally, an adjustable bandwidth voltage inverter amplifier 144, the input of which is connected to the node N common to the photodiode 140 and the voltage-controlled current source 142.

The high-pass filter also comprises a capacitor 146 inserted between the node N and the output to the multiplier 1 12.

The high-pass filtering is essentially performed by the capacitor 146, the passband of this filter being approximately equal to G / (2nR _d C), G being the gain in absolute value of the amplifier 144, R _d the dynamic transimpedance of the voltage-controlled current source 142 and C the capacitance of the capacitor 146.

The capacitor 146 also makes it possible to isolate the operating point of the amplifier 144, which is also the bias voltage of the photodiode 140, from the operating point of the multiplier 12. Thanks to this arrangement, it is also possible to reduce the cutoff frequency of the high-pass filter, by decreasing the bandwidth of the amplifier 144 which is adjustable.

It may be advantageous to choose the capacitance C of the capacitor 46 greater than the input capacitance of the multiplier 1 12, in order not to reduce the useful signal excessively.

FIG. 8 shows an exemplary embodiment of the photodetector 106 comprising the varicap 148 (constituted by the photodiode 140 and the capacitor 146) and a transistor 150 (of the NMOS type for example) constituting the voltage-controlled current source 142. The transistor 150 operates in saturated mode, and behaves as a current source controlled by the potential of the node N. In this arrangement, the cutoff frequency of the high-pass filter depends only on the bias current of the photodiode 140, FIG. that is, incident light flux, and can not be adjusted by any other means. In addition, this cutoff frequency is not very high.

FIG. 9 illustrates an exemplary embodiment of the amplifier 1 10.

The amplifier 1 10 comprises a transistor M2.1 162, placed between the output and the input of an inverting amplifier 164, controlled by a voltage VCOMa. Transistor 162 converts an incoming current 11 into a voltage, which will then be reconverted to a current 112 by a transistor M2.2 166 placed at the output of the amplifier. A capacitor 168 is also advantageously placed in series with the transistor M2.2 166, so as to perform an additional high-pass filtering function, and to isolate the operating point of the inverting amplifier 164 from the rest of the circuit. The inverting amplifier 164 consists for example of two transistors.

FIG. 10 illustrates the structure of the multiplier 1 12 and the integrator 1 18. This demodulation stage constituted by the multiplier 1 12 and the integrator 1 18 comprises two identical resistive devices 180 and 182 whose conductances are voltage-controlled, a first voltage inverting amplifier 184, a second voltage inverting amplifier 186 whose input operating point is approximately identical to that of the amplifier 184. These two inverting amplifiers 184 and 186 consist, by way of example, of two transistors each.

To ensure that the operating points at the input of the two amplifiers 184 and 186 are identical, it is possible to use two identical structures polarized in the same way.

The input of the amplifier 184 is connected to the node N1 common to the output of the photodetector 106 and to one of the terminals of the resistive device 180. The input of the amplifier 184 is connected to the common node at the output of the amplifier. amplifier 1 and at one of the terminals of the resistive device 180.

The output of the amplifier 184 is connected to the other terminal of the resistive device 180 and to the resistive device 182, the other terminal of the resistive device 182 being connected to the input of the amplifier 186. The voltage (or the voltages VCOM1) of the resistive device 180 is fixed, thereby setting the value of the conductance of this resistive device 180. The output voltage of the amplifier 184 is automatically adjusted by the feedback so that the current flowing through the resistive device 180 compensates the high frequency components of the photocurrent coming from the photodetector 106. By construction, the operating point of the amplifier 184, i.e. the voltage at its input, should be approximately the same as that of the amplifier 186, i.e. the voltage at its amplifier. Entrance. In this case, if by adjusting the control voltage (s) VCOM2 of the resistive device 182 so that its conductance is the product of that of the resistive device 180 by a number f, then the current flowing through the device resistive 182 will be approximately equal to the product of the current flowing through the resistive device 180 through f. The only constraint is that the number f is positive.

By appropriately varying the VCOM2 control voltage (s), it is thus possible to access the product of the photocurrent by an arbitrary positive function f (t). The fact that the function f (t) is positive is not a limitation; just choose:

f (t) = Ref (t) + f ₀ (t)

f ₀ (t) being a positive function to guarantee the positivity of f (t). It is advisable to choose f ₀ (t) in such a way that it contains only low frequencies, and that the contribution of the product of f ₀ (t) by the high frequency components of the photocurrent is negligible after the integrator 1 18 acting as a filter for high frequencies.

The integrator 1 18 is a conventional structure, consisting of the amplifier 186 and a capacitor 188 placed between the input and the output of the amplifier 186. The current flowing through the resistor 182 is simply integrated in the capacitor 188. A transistor 190, also placed between the input and the output of the amplifier 186, allows the resetting of the integrator 1 18. The output of the integrator 1 18, which is also the output of the amplifier 186 , is connected to the memory 124 of FIG.

As illustrated by FIG. 11, each of the resistors 180, 182 may consist of either an NMOS transistor 192 controlled by a voltage VCOM_N.

FIG. 12 presents a means for generating the control signal VCOM_N of the NMOS transistor 192 of FIG. 1 1. This implementation circuit consists of a transistor 196 identical to the NMOS transistor 192 constituting the resistive device 182 or possibly much more broad (a transistor n times wider corresponding to a juxtaposition of n identical transistors) and an operational amplifier 198 and a resistor 200.

The non-inverting terminal of the operational amplifier 198 is attached to a VON potential. The drain of the NMOS transistor 196 is connected to a fixed potential VN, while its source is connected to the inverting input of the operational amplifier 198 and to one of the terminals of the resistor 200. The other terminal of the resistor 200 is connected to a programmable voltage source 202. The output of the operational amplifier 198 is connected to the gate of the NMOS transistor 196, and defines the potential VCOM_N. The signal VCOM_N thus defined sets the resistance of the NMOS transistor 196 so that, under the fixed potential difference VN-VON, the current flowing through the latter is identical to the current flowing through the resistor 200, which is itself determined by the programmable voltage source 202. If the voltage VON is identical to the operating point at the input of the amplifier 186, then the NMOS transistor 192 constituting the resistive device 182 will have the same resistance (or a resistance multiplied by a factor n if the transistor of Figure 12 is n times wider than that which constitutes the resistive device 182). Otherwise, in low inversion mode, an offset error corresponds to a simple multiplicative factor. Finally, in low inversion mode, the NMOS transistor 196 does not necessarily have to operate in linear mode, and it is possible to take VN equal to + avdd.

FIG. 13 illustrates an embodiment of the reference register 132 of the analog memory 124. This reference register 132 is refreshed regularly by recording, by way of example, a measurement result taken with a reference signal 1 14 f ( t) = f ₀ (t) of the level of the multiplier 1 12.

According to the embodiment of FIG. 13, the reference register 132 essentially comprises a follower assembly consisting of transistors M3.1 and M3.2. An M3.3 transistor is also provided to operate in switch mode. According to the control signal WVREF, the voltage to be stored can thus be recorded on the gate of the transistor M3.1 when M3.3 is closed, then stored when M3.3 is open. The VREF output of the follower is, at an offset, a copy of the stored gate voltage. It is directly connected to the differential inputs of the differential input multiplier 134.

Data registers Reg_1 126, Reg_2 128 ..., Reg_N 130 have random write access, and a double read random access. Their implementation, an example of which is shown in FIG. 14, is identical to that of the RegRef register 132, except that the recording operation is controlled by a signal D1 (i) 210, and that the output is isolated from the inputs Vx and Vy of the differential input multiplier 134 by two transistors M4.4 and M4.5, controlled by two signals D2 (i) 212 and D3 (i) 214. It is thus possible, using D2 (i) 212 and D3 (i) 214, to perform the multiplication of any pair of registers of the memory 124.

The differential input multiplier 134 is a differential input four quadrant multiplier, as described in [Gunhee Han and Edgar Sanchez-Sinencio, CMOS Transconductance Multipliers: A Tutorial, IEEE Trans. we wax, and syst., vol. 45 n [deg.] 12, p1550 (1998)], implemented according to the circuit of FIG. 15. The fact of using a differential input and using a RegRef register 132 makes it possible to overcome all offset problems, which may notably be related to a difference between the polarizations of the amplifiers 184 and 186 or to a transfer of charges when opening a transistor operating in switch mode or the operation of the follower assembly constituting the registers of the memory 124.

The operating point of the differential input multiplier 134 is set by a suitable choice of that of the integrator 1 18, that is to say by a suitable choice of the operating point at the input of the amplifier 186 which can - even depend on the choice of supply voltages of this amplifier. It should be noted that in the exemplary implementation of FIG. 15, the reference voltage is directly used at the input of the differential input multiplier 134, which is not the case in the document [Gunhee Han and Edgar Sanchez-Sinencio].

The output of the differential input multiplier 134 consists of two currents I P and IM, the difference IP-IM of which is the result of the multiplication. It should be noted that the potentials UP and UM at the output of the multiplier 134 must, when the multiplier 134 is in operation, be identical and set to a precise value V0.

Such an analog processing circuit 102 can process a very weak light signal, while having a satisfactory signal-to-noise ratio thanks to the integration of this circuit over a very small space of dimension less than or equal to 42 × 44 μηη. It is thus possible to have a large number of pixels so as to maximize the signal-to-noise ratio.

It is thus possible, by this massively parallel processing, to process a large number of images per second (1000 to 100,000) without using specially fast electronics.

The originality of the circuit 102 lies essentially in the fact of using an analog memory. However, the use of such an analog memory presents a difficulty concerning the elimination of offsets on the memorized quantity. This difficulty is solved in the circuit according to the invention, thanks to the use of the reference register 132 and the differential input multiplier 134.

The circuit 102 which has just been described is therefore suitable for implementing an analysis method comprising an estimation of the temporal correlation.

On the basis of the device 8 which has just been described, a measuring device 300 adapted to the implementation of the method as described above can be advantageously implemented for the reflectance measurement of the oxygenation rate of hemoglobin. of a human being. For the measurement of this oxygenation rate, at least two central wavelengths λ-ι and λ ₂ are used.

Thus, the measurement device 300 of the oxygenation rate illustrated in FIG. 16 comprises a first laser diode 16 and a second laser diode 302 whose central emission wavelengths are respectively λ-ι and λ ₂ . The two diodes 16 and 302 are each controlled by a control unit 18 and 304 for modulating the wavelength of the light emitted, as described in the embodiment of FIG.

An automatic shutter 306 is arranged at the output of the laser diodes and in order to selectively illuminate the optical fiber 20 with one or the other of the light beams emitted by these diodes 6 and 302.

Preferably, for each wavelength and λ ₂ , the scattered light is measured for different delays τ ₀ , as described with reference to the embodiment of FIG. 1.

The measuring device 300 comprises, connected to the detection and analysis means

42, a unit 308 for calculating the rate of oxygenation of hemoglobin.

The oxygenation rate of the hemoglobin is deduced from the medium absorption coefficients calculated at the two wavelengths k _-1 and λ ₂ specific to the diodes. The calculation of the absorption coefficient is made, as a function of a relationship known per se, from the exponential decay of light scattering as a function of time, this decay being defined from the different measurements made at delays. τ ₀ different.

Alternatively, if the device 300 implements the method of FIG. 5, the data relating to the decorrelation of the signals S, provide information on the blood circulation in tissues such as the capillaries of the muscle, in addition to the measurement of the blood flow rate. oxygenation as explained with reference to FIG. 16. The time-resolved acquisition of these data makes it possible, for example, to isolate the contribution of the blood circulation in such capillaries compared with that in blood vessels of larger sizes.

The monitoring of the oxygenation rate makes it possible to apply the analysis method to monitoring the perfusion of an organ. For example, the organ is a kidney, prostate, bone marrow or brain.

Thus, the devices illustrated in FIGS. 1 and 16 only make it possible to perform a time-resolved measurement at a point considered in the sample. The devices described here and the method implemented makes it possible to analyze a diffusing sample by time-resolved measurement of diffuse light in this sample at a very low cost.

Indeed, the light source used may be formed of a simple laser diode. Similarly, the detection and analysis means can be made at a lower cost.

In addition, with the wavelength modulated light used, the electromagnetic fields are much less intense than pulsed, as is the case in the state of the art.

Finally, a very high sensitivity is obtained for the analysis.

Even with a simple photodiode as a detector, a very high sensitivity and a very great dynamic can be obtained.

Claims

- A method for analyzing a sample (10) diffusing by time resolved measurement of the light scattered in this sample (10), the method comprising the steps of:

illuminate the sample (10) at an illumination point (P1) located on the sample

(10) with an incident beam of temporally coherent light and wavelength-modulated according to a function having for period the modulation period (T);

detecting on a photodetector (44) during a first time interval the time evolution of the intensity of the light scattered by the sample at an exit point (P2) located on the sample (10) at a distance ( d) less than 1 millimeter from the point of illumination (P1);

multiplying the signal detected by the photodetector (44) by a reference signal (Ref (t, io)) generating a time gate centered on a predetermined delay (τ ₀ ) to obtain a multiplied signal,

extracting demodulated components of the signal detected from the multiplied signal,

- apply a non-linear function to the extracted demodulated components.

The method of claim 1, wherein the extracting step comprises the steps of:

- cutting the first time interval according to the modulation period

(T) to obtain a succession of integration interval,

calculating a demodulated component of the signal detected for each integration interval, the demodulated component being defined as the integral of the signal multiplied over the integration interval considered.

3. The method of claim 2, wherein at the step of cutting, each integration interval corresponds to a time interval equal to half a modulation period.

4. - Method according to any one of claims 1 to 3, wherein at the step of illuminating the sample, the incident beam comes from an optical fiber (20), the optical fiber (20) also serving collecting the light scattered by the sample at the exit point and bringing the collected light to the photodetector (44).

5. - Method according to claim 5, wherein the fiber (20) is provided at one end with a probe (24) diffusing.

6. - Method according to any one of claims 1 to 5, wherein the non-linear function is the average of the square of the demodulated component or the product average of two separate demodulated components.

7. - Device (8) for analyzing a scattering sample by resolved measurement in time of diffuse light in this sample, comprising:

means (12) for illuminating the sample with an incident beam of temporally coherent and wavelength-modulated light at an illumination point (P1) located on the sample (10);

a photodetector (44) detecting the temporal evolution of the intensity of the light diffused by the sample at an exit point (P2) situated on the sample (10) at a distance (d) of less than 1 millimeter from the illumination point (P1);

a processing unit adapted to implement the analysis of the signal detected by the photodetector (44) to obtain a temporal correlation of this signal.

8. - Device according to claim 7, adapted for the implementation of the method according to one of claims 1 to 6.

9. - Device according to claim 7 or 8, wherein the device (8) comprises a circuit (102) comprising the photodetector (44), a multiplier, an integrator, an analog memory; and a unit for estimating a temporal correlation of the signal detected by the photodetector from the memorized components.

10. - Use of the device according to any one of claims 7 to 9 for monitoring the oxygenation and perfusion of an organ of a human being.