WO2011016734A1

WO2011016734A1 - Electromagnetic field energy recycling

Info

Publication number: WO2011016734A1
Application number: PCT/NZ2010/000157
Authority: WO
Inventors: Ashley James Gray; Neville Roy Samuel Illsley
Original assignee: Restech Limited
Priority date: 2009-08-05
Filing date: 2010-08-05
Publication date: 2011-02-10

Abstract

A magnetic field energy recycling circuit comprises a capacitor (C1), an inductive device (L1) and a switching circuit (S1, S2, SC), that is sequentially configurable, firstly to direct a discharge current to flow in a first direction from the capacitor and through the inductive device to establish a magnetic field in association with the inductive device; then secondly to direct a current induced in the inductive device during collapse of the magnetic field to flow into the capacitor in a second direction that is opposite the first direction to thereby charge the capacitor. The circuit is connected to a supply of electrical energy (V1) such that current from the supply flows through the capacitor and the inductive device when the switching circuit is configured in the first and/or the second configuration.

Description

ELECTROMAGNETIC FIELD ENERGY RECYCLING FIELD OF INVENTION The present invention relates to the recycling of electromagnetic field energy. More particularly, the present invention relates to the recycling (i.e. the recovery and re-use) of energy from a magnetic field using electromagnetic circuits having controlled switches. Energy from a collapsing magnetic field of an inductive device is recovered and stored in a capacitor for later use in re-establishing a magnetic field at the inductive device.

BACKGROUND

A common aspect of conventional inductive devices, such as motors, linear actuators, solenoids, transformers and induction coils, is that they rely on the building of a magnetic field to perform a work function either by a motoring, transforming or inducing action, or by a magnetic attraction or repulsion. The energy built up or contained within the magnetic field in these instances is substantial and remains even after a work function is performed. Standard designs of motors, solenoids, linear actuators, transformers and induction coils do not as a general rule use field energy recovery on the primary windings. The propensity of the magnetic field to remain once built up in inductive devices is often treated to some degree as a nuisance. Many control strategies are used to deplete the magnetic field in a way that minimises damage to the inductive device or to other circuit components from excessive voltage spikes and the like. Depletion of the magnetic field, sometimes referred to as 'defluxing', has been achieved by diode clamping, applying reverse voltages and by other field control techniques.

In some cases, rather than merely dissipating the energy, and to avoid nuisance, the energy has been recovered for later re-use. Typically, energy from a collapsing magnetic field has been returned to a capacitor, such as a supply reservoir or supplementary capacitor, for reuse when demand is next placed on the supply. SUMMARY OF INVENTION

The present invention can be used to recover energy from a collapsing magnetic field and efficiently capture this recovered energy for effective re-use.

In broad terms in a first aspect die invention comprises a magnetic field energy recycling circuit comprising a capacitor, an inductive device and a switching circuit; wherein:

the switching circuit is configurable in a first configuration to direct a discharge current to flow in a first direction from the capacitor and through the inductive device to thereby establish a magnetic field in association with the inductive device;

die switching circuit is configurable in a second configuration, after die magnetic field has been established, to direct a current induced in the inductive device during collapse of the magnetic field to flow into the capacitor in a second direction that is opposite die first direction to thereby charge the capacitor; and

die magnetic field energy recycling circuit is connected to a supply of electrical energy such tiiat current from the supply flows tiirough die capacitor and die inductive device when the switching circuit is configured in the first configuration and/or the second configuration. Preferably in die first aspect, when the switching circuit is in the first configuration, the capacitor is connected in series with the supply such that the discharge current flows from the supply, through the capacitor, and through die inductive device to establish the magnetic field in association with the inductive device. Alternatively in the first aspect, when the switching circuit is in the second configuration, the capacitor is connected in series with the supply such that the current induced in the inductive device during collapse of the magnetic field flows through die capacitor and through die supply. In broad terms in a second aspect the invention comprises a magnetic field energy recycling circuit comprising a capacitor, an inductive device and a switching circuit; wherein: the switching circuit is configurable in a first configuration to direct a discharge current to flow in a first direction from the capacitor and through the inductive device to thereby establish a magnetic field in association with the inductive device;

the switching circuit is configurable in a second configuration to direct a current to flow from a supply of electrical energy and through the inductive device to maintain the magnetic field; and

the switching circuit is configurable in a third configuration, after the magnetic field has been established and maintained, to direct a current induced in the inductive device during collapse of the magnetic field to flow into the capacitor in a second direction that is opposite the first direction to thereby charge the capacitor.

In broad terms in a third aspect the invention comprises a magnetic field energy recycling circuit comprising a capacitor, an inductive device and a switching circuit; wherein:

die switching circuit is configurable in a first configuration to direct a discharge current to flow in a first direction from the capacitor and through the inductive device to thereby establish a magnetic field in association with the inductive device;

the switching circuit is configurable in a second configuration, after the magnetic field has been established, to direct a current induced in the inductive device during collapse of the magnetic field to flow into the capacitor in a second direction that is opposite the first direction to thereby charge the capacitor; and

the recycling circuit is connected to a supply of electrical energy such that when the switching circuit is configured in the first configuration and die capacitor is substantially discharged, current flow through the inductive device is maintained by current from the supply.

Preferably, in the third aspect, when the switching circuit is configured in the first configuration, the voltage of the supply of electrical energy is switchable from a first voltage to a second voltage, and the second voltage is lower than the first voltage. The term "comprising" as used in this specification means "consisting at least in part of. That is to say, when interpreting statements in this specification which include "comprising", the features prefaced by this term in each statement all need to be present but other features can also be present. Related terms such as "comprise" and "comprised" are to be interpreted in a similar manner.

This invention may also be said broadly to consist in the parts, elements and features referred to or indicated in the specification of the application, individually or collectively, and any or all combinations of any two or more of said parts, elements or features, and where specific integers are mentioned herein which have known equivalents in the art to which this invention relates, such known equivalents are deemed to be incorporated herein as if individually set forth.

As used herein the term "and/or" means "and" or "or", or both.

As used herein "(s)" following a noun means the plural and/or singular forms of the noun. The term 'inductor' as used in this specification means a passive component that is incorporated in a circuit primarily for its property of inductance.

The term 'inductive device' as used in this specification means a device having inductance but which is incorporated in a circuit primarily for establishing a magnetic field to perform a work function, for example by a motoring, transforming or inducing action, or by a magnetic attraction or repulsion. Inductive devices include, but are not limited to, transformers, electromagnetic motors, linear actuator coils, electromagnets, solenoid coils and induction coils. References herein to a current induced in an inductive device during collapse of a magnetic field can be understood as referring to a current that is driven by a voltage induced in the inductive device by collapse of the magnetic field through the winding inductance of the device. BRIEF DESCRIPTION OF DRAWINGS

The invention will be further described by way of example only and without intending to be limiting with reference to the following drawings, wherein: Figure IA shows a circuit illustrating a first embodiment of the invention;

Figure IB is a switch timing diagram for the circuit of Figure IA, showing one cycle of circuit operation;

Figure 1C is a first magnetising configuration of the circuit of Figure IA during a first stage of a cycle of operation; Figure ID is a second magnetising configuration of the circuit of Figure IA during a first stage of a cycle of operation;

Figure IE is an energy recovery configuration of the circuit of Figure IA during a second stage of a cycle of operation;

Figure IF shows waveforms of the supply current (upper waveform) and inductive device current (lower waveform), for the circuit of Figure IA over several cycles of operation during initial start-up; Figure 1 G shows waveforms of the supply current (upper waveform) and inductive device current (lower waveform), for die circuit of Figure IA over two cycles of operation for a run mode;

Figure IH shows a waveform of the voltage across a recovery capacitor of the circuit- of Figure IA over several cycles of operation during initial start-up;

Figure 11 shows a waveform of the voltage across a recovery capacitor of the circuit of Figure IA over two cycles of operation for a run mode; Figure 2A shows a circuit illustrating a second embodiment of the invention;

Figure 2B is a switch timing diagram for the circuit of Figure 2A, showing one cycle of circuit operation in a run mode; Figure 2C is a first magnetising configuration of the circuit of Figure 2A during a first stage of a cycle of operation; Figure 2D is a second magnetising configuration of the circuit of Figure 2A during a first stage of a cycle of operation;

Figure 2E is a third magnetising configuration of the circuit of Figure 2A during a first stage of a cycle of operation;

Figure 2F is an energy recovery configuration of the circuit of Figure 2A during a second stage of a cycle of operation;

Figure 2G shows waveforms of the supply current (upper waveform) and inductive device current (lower waveform), for the circuit of Figure 2A over several cycles of operation during initial start-up;

Figure 2H shows waveforms of the supply current (upper waveform) and inductive device current (lower waveform), for the circuit of Figure 2A over two cycles of operation for a run mode;

Figure 21 shows a waveform of the voltage between upper and lower rails of the circuit of Figure 2A over several cycles of operation during initial start-up; Figure 2J shows a waveform of the voltage between upper and lower rails of the circuit of Figure 2A over two cycles of operation for a run mode;

Figure 3A shows a circuit illustrating an eighth embodiment of the invention; Figure 3B is a switch timing diagram for the circuit of Figure 3A, showing one cycle of circuit operation; Figure 3C is a first magnetising configuration of the circuit of Figure 3A during a first stage of a cycle of operation;

Figure 3D is a second magnetising configuration of the circuit of Figure 3A during a first stage of a cycle of operation;

Figure 3E is a third magnetising configuration of the circuit of Figure 3A during a first stage of a cycle of operation; Figure 3F is an energy recovery configuration of the circuit of Figure 3A during a second stage of a cycle of operation;

Figure 3 G shows waveforms for the supply current (upper waveform), recovery capacitor current (middle waveform), and inductive device current (lower waveform), for the circuit of Figure 3A over several cycles of operation during initial start-up;

Figure 3H shows waveforms for the supply current (upper waveform), recovery capacitor current (middle waveform) and inductive device current (lower ^*" waveform), for the circuit of Figure 3A over two cycles of operation for a run mode; and

Figure 31 shows voltage waveforms of the circuit of Figure 3A over several cycles of operation during initial start-up, the upper waveform showing the voltage of the dual voltage supplies as applied to the anode of diode D3 and the lower waveform showing the voltage across the recovery capacitor Cl.

DETAILED DESCRIPTION The invention is based on the discovery that energy remains in a magnetic field after the field has been used to perform work, for example the mechanical work performed by the field of an electromagnetic motor. In general terms, the invention allows a magnetic field to be established in association with an inductive device (such as a transformer, motor, solenoid, or induction coil, for example). The field is predominantly established using energy recovered from the collapse of a previously-established magnetic field associated with the inductive device. This recovery and re-use of the energy in the magnetic field allows the inductive device to be operated with improved performance and particularly with improved efficiency. Energy consumed in performing the work through hysteresis, back emf or circuit losses can be replenished on a cycle-by-cycle basis. Significant efficiency gains can be made when these losses are kept low and are a small fraction of the energy needed to create the magnetic field. Energy is recovered from the magnetic field associated with an inductive device, such as a winding, while the field is performing, or has performed, useful work. The recovered energy is stored on a capacitor for re-use when later re-establishing the magnetic field. Controlled switches alternately interconnect the inductive device and the capacitor in the magnetising and energy recovery configurations.

The invention relates to the switching circuit configurations and particularly to:

• series connection of the supply energising the circuit and the capacitor on which the recovered energy is stored (the series connection being either during recovery of energy from the magnetic field, or during re-establishment of the magnetic field); • controlled injection of supply current to the inductive device to meet circuit losses; and

• maintenance of current flow through the inductive device direct from the energy supply, with optional switching of the supply to a lower voltage. The invention relates to a switching circuit which charges a capacitor by energy recovered from a collapsing magnetic field, and subsequently discharges the capacitor to re-establish the magnetic field. This cycle of operation is repeated. Preferably, the capacitor is completely discharged when re-establishing the magnetic field during each cycle of operation. However, the circuits described below can be used without fully depleting the charge on the capacitor. That is, the circuits will operate effectively with a residual charge left on the capacitor after the magnetic field has been re-established. This condition can occur when the timing of the switching of the ckxuit provides a magnetising period that is less than optimum, or when the capacitance of the capacitor is greater than optimum. The current invention relates to circuits for driving electromagnetic devices. The invention relates particularly to such circuits incorporating recovery of energy from a collapsing magnetic field, the storage of that recovered energy as charge on a capacitance, and the subsequent use of the stored recovered energy to establish a magnetic field. The invention makes use of efficient transfer of energy between charge stored on capacitors and magnetic fields associated with inductances of inductive devices, such as in electric motors, generators, transformers, solenoids and induction heating coils, for example. In the current invention, the transfer of energy, from inductance to capacitance, and from capacitance to inductance, behaves similarly to corresponding energy transfers between the inductance and capacitance of a resonant circuit. However, unlike freely oscillating resonant circuits in which energy is continuously and repetitively transferred back and forth between inductance and capacitance without interruption, circuits according to the current invention operate repetitively but with what may be termed interrupted, or dis-continuous, resonant energy transfer. In applications of the current invention, the repetitive but interrupted transfer of energy between capacitance and inductance is performed under the control of a switching circuit, for example using transistors and semiconductor diodes as switch elements.

The repetitive transfer of energy between capacitance and inductance, even when discontinuous, builds energy in- the reactive components (capacitor and inductor) in the same way as in a resonant circuit, such that, after successive cycles, the voltages and circulating currents in the reactive component circuit can be substantially greater than those of the supply feeding the circuit.

In the current invention, the controlled switching circuit effectively connects capacitance and inductance in various circuit configurations to carry out the energy transfers. In a magnetising configuration, the switching circuit effectively connects a capacitance to an inductance to transfer energy stored on the capacitance to the inductance, to establish or assist in establishing a magnetic field. In an energy recovery configuration, the switching circuit effectively connects an inductance to a capacitance to charge the capacitance with energy recovered from the inductance on collapse of the magnetic field. In a third configuration, the switching circuit is configured to hold the recovered energy stored by the capacitance until required for establishing an electromagnetic field.

In many prior art magnetic field energy recovery circuits using a capacitor to store energy recovered from a collapsing magnetic field, and later re-using the energy stored on the capacitor to establish a magnetic field, the voltage across the capacitor is maintained at a relatively high level, usually above or close to a supply voltage. This results in less than optimum efficiency of energy transfer. In these circuits the capacitance is relatively large and acts as ari energy reservoir that is not completely, nor even nearly, depleted during a magnetising period.

In the current invention, the switching circuit is configured in the magnetising and energy recovery configurations for respective magnetising and energy recovery periods. For efficient energy recovery and re-use of recovered energy, these periods are close to, or substantially equal to, one quarter of the natural resonance period of the respective circuit configuration. By correctly controlling these periods to suit the circuit reactances of the respective switching circuit configurations, and/or by designing the circuit to automatically and passively adopt the correct configurations at the correct times, each resonant-like energy transfer action can be dis-continued when the respective energy transfer is at, or close to, a maximum.

Maximum recovery of energy from the magnetic field occurs when current flowing in the inductance falls to zero. In practical switching circuits according to the invention, the recovery period is made sufficient to allow the inductor current that recharges the recovery capacitance to fall to zero. If the inductor current is not zero at the end of the recovery period, and provision is not made to deal with the non-zero current, large and potentially damaging voltages could be generated by the inductance, for example when reconfiguring the switching circuit from the recover_}' configuration to the magnetising configuration. Maximum transfer of energy stored in the charge on the capacitance occurs when voltage on the capacitance falls to zero. However, in practical switching circuits according to the invention, the capacitor voltage does not necessarily need to fall to zero. Unlike the preferred zero inductance current as discussed in the immediately preceding paragraph, there is no necessity for the capacitance voltage to fall to zero during the magnetising period. Voltage remaining on the capacitance can be held, without significant loss, until further charge is added to the capacitance at the next energy recovery period. Substantial energy can be transferred from the capacitance to the inductance even when non-zero voltages remaining on the capacitance at the end of the magnetising period, giving useful performance of circuits according to the current invention. Voltage remaining on the capacitance may result from incomplete discharge of the capacitance or, in some embodiments of the current invention, may result from first discharging the capacitance then recharging the capacitance to an opposite polarity.

The value of the inductance may be substantially constant during magnetising and/or recovery configurations, for example as in transformers, generators or induction heating coils. In some applications of the invention, the inductance may alter dynamically during the periods the switching circuit is configured in these configurations. For example, a switched reluctance motor or a solenoid-driven actuator or pump may present a winding inductance that varies, either linearly or non-linearly, over a wide range during operation. In this case, the capacitance and switching circuit periods can be selected so that even with the dynamically changing inductance, the objective of substantially complete energy transfer is achieved by the end of the respective magnetising or field energy recovery periods. Whether the inductance value is fixed or dynamically varying, the maximum transfer of energy from the capacitance to the inductance still occurs when voltage on the capacitance falls to zero, and the maximum transfer of energy from the inductance to the capacitance occurs when current flowing in the inductance falls to zero. In applications, for example switched reluctance motors, where the inductance is not fixed, an average inductance value can be used in mathematical expressions to determine a relationship between the inductance and the recovery capacitance, and a magnetising or recovery period. Although this average inductance value may not be absolutely mathematically correct, an average value has been found to provide a close approximation for calculation of optimum values of periods and recovery capacitor values for practical circuits. The use of an approximate average inductance value can avoid the need for complex modelling and integration of changing inductance values over magnetising and recovery periods. The values of inductance and capacitance may be substantially the same for the magnetising and recovery configurations. Alternatively, the values of inductance and/or capacitance for the magnetising configuration may differ from die values of inductance and/or capacitance for the recovery configuration. For example, some embodiments of die current invention can employ a plurality of two or more capacitors connected in parallel for die magnetising configuration but connected in series for the recovery configuration. The series connection of the capacitors provides a lower capacitance value than the parallel connection. The lower capacitance of the series-connected capacitors decreases die natural resonance period or circuit time constant and dierefore enables a faster recovery of magnetic field energy. This can be advantageous in applications of die current invention for driving high speed motors. The relatively larger capacitance of the parallel-connected capacitors increases die natural resonance period or circuit time constant and lengthens the duration of the magnetising current pulse.

The changes between parallel connection and series connection of die two or more capacitors can be performed passively, for example by passive switching of semi-conductor diodes by the bias voltage on die diodes. Alternatively, the changes between parallel connection and series connection can be performed actively, for example by controlled switching of transistors. Active control of die series/parallel connection may be used to connect the capacitors solely in parallel, in series and parallel, and solely in series through various phases of start-up or operation of inductive devices to advantageously configure the magnetising and recovery period capacitances to optimise maximum capacitor operating voltages and dierefore energy transfers.

The recover_}' capacitance can also be dynamically varied throughout the operating cycle. In addition to the series/parallel switching arrangements cited above, combinations of capacitors from a bank of parallel capacitors can be switched in and out of circuit to provide a wide range of recovery capacitance values to meet the requirements of specific circuits or applications.

The switching circuit is selectively controlled to commence the magnetising configuration. For example, in applications of circuits according to the current invention to drive a variable reluctance motor, the magnetising configuration may be commenced at a synchronisation time derived from a pick-up or sensor device monitoring the angular position of the rotor of the motor. The duration or period that the switching circuit maintains the magnetising configuration may be actively controlled by controlled switches, for example transistors, or may be determined by passive circuit elements, for example diodes, which respond automatically to polarities of circuit voltages or currents. Similarly, the duration or period that the switching circuit maintains the field energy recovery configuration may be actively controlled by controlled switches, for example transistors, or may be determined by passive circuit elements, for example diodes, which respond automatically to polarities of circuit voltages or currents. Semiconductor diodes are used in some embodiments of the current invention to make automatic changes to the switching circuit configurations. For example, semiconductor diodes are used to react to the fall to zero of the inductor current and to then change the switching circuit from the second configuration to the third configuration at the optimum time of maximum energy transfer, without requiring actively controlled switching.

In the third switching circuit configuration, energy recovered from a magnetic field is stored on an energy recover}' capacitor and held there until required for establishing, or assisting in establishing, a subsequent magnetic field. The third switching circuit configuration ends and the cycle is repeated when the switching circuit is selectively controlled to commence the next magnetising configuration. The next cycle is initiated by actively switching the switching circuit to adopt a magnetising configuration. For example, the initiation of the next cycle may be synchronised with a predetermined position of a rotor in applications where the circuit is used to drive a motor, or synchronised with a clock signal where the circuit is used to provide a predetermined fixed frequency output.

In some circuit topologies according to the invention, the first and second switching circuit configurations may be identical, in which case the first configuration provided by the switching circuit may be maintained to also provide the second configuration. For example, a capacitor charged to a voltage of one polarity is discharged to drive current into an inductor to establish a magnetic field. When the voltage on the capacitor reaches zero, the current in the inductor has reached a maximum and energy transfer from capacitor to inductor is complete. The inductor current continues to flow in the same direction, but starts to drop in amplitude and the magnetic field begins to collapse. The continuing, but falling, current recharges the capacitor to a voltage of the opposite polarity. Energy recovery is complete when the inductor current has dropped to zero. In this circuit there is no change in circuit configuration from the magnetising configuration to the recovery configuration.

The transition from the second, i.e. energy recovery, configuration to the third, i.e. holding, configuration can be achieved by semiconductor diodes which conduct to allow the inductor current to flow in the one direction as described above, but which become non- conductive to block a reverse current from flowing. This blocking prevents discharge of the capacitor when charged to the opposite polarity, at least until actively switched by a switching circuit controller, to commence a new magnetising period, for example.

Some circuits according to the invention may incorporate further switching circuit configurations between the three configurations described above, without departing from the invention.

For example, although in some circuits the second, i.e. recovery, configuration follows immediately after the first, i.e. magnetising, configuration, there may be intermediate configurations by which the inductor current, initiated by transfer of recovered energy from the capacitor, is maintained or extended by passing current, drawn from a supply, through the inductor. At the end of the inductor current extension period the supply is disconnected from the inductor, configuring the switching circuit in a recovery configuration and initiating a field energy recover)' phase. During the recover}' phase, the inductor current falls, the magnetic field collapses and energy is recovered to be stored on the capacitor. In a further example of other switching circuit configurations, the inductor current initiated by transfer of recovered energy from the capacitor may be regulated by switching, or chopping, the discharge of the capacitor into the inductor. By recovering and re-using energy from the magnetic field using the dis-continuous resonant-like energy transfer of the current invention, magnetic fields can be established with increased efficiency, using lower supply voltages and/or providing greater field strengths. For example, in some embodiments of the invention, the voltage stored on the recovery capacitor after recovery of energy from the collapsing magnetic field, is placed in series with the supply to compound the voltage available for subsequently re-establishing the magnetic field. After repetitively recycling energy recovered from the magnetic field over a few cycles, circuits according to the invention can operate with a significantly boosted voltage on the capacitor at the beginning of each magnetising configuration period. The boosted voltage can be many times the voltage of the electrical source supplying the circuit. This voltage boosting or compounding action is similar to that of a resonant circuit, and like the resonant circuit, depends on the quality factor, or Q, of the circuit. The voltage compounding action allows motors and other inductive devices to be operated using relatively high working voltages derived from relatively low supply voltages. Some embodiments of the current invention drive inductive devices harder, i.e. with higher winding currents, and/or operated at higher efficiency, than when operated by prior art circuits using the same supply voltage.

The current invention has particular application to motors where higher mechanical output torque does not necessarily correlate with higher motor winding currents. Motor torque can be affected by the shape of the winding current waveform, and particularly by the steepness of the rise in winding current. A faster rising winding current can give a higher motor torque and is particularly advantageous at high speed operation. The voltage compounding action described above provides a higher voltage that gives a faster rising winding current waveform and a higher motor output torque, than would be achieved from just the supply voltage alone.

Circuits according to the invention can be configured in a wide range of circuit topologies. For example, circuits according to the invention can be configured to establish a magnetic field of one polarity by discharging a capacitor charged to a first polarity, and then recover energy from that magnetic field to recharge the capacitor to the same or opposite polarity. Successive magnetisings of the inductive device may provide magnetic fields of the same or alternating polarities. There may be only a single inductance and capacitance. Alternatively, a pair or a multiple number of capacitors may be alternately charged and discharged to repetitively recover energy from a magnetic field and deliver energy to reestablish a magnetic field, in a single inductor. A single capacitor may be discharged and charged to establish, and recover energy from, magnetic fields alternately in two or more inductances. The two inductances may be from respective inductive devices, or may be respective windings of a single device, or may be mutual inductances of the same inductive device.

In a three-stage closed-loop circuit according to the invention, the energy recovered from a magnetic field in a first inductor, can be transferred to a first capacitor for use in later establishing a magnetic field in a second inductor, and the energy recovered from the magnetic field in the second inductor can be transferred to a second capacitor for use in later establishing a magnetic field in a third inductor, and the energy recovered from the magnetic field in the third inductor can be transferred to a third capacitor for use in later establishing a magnetic field in the first inductor. Such a circuit can be used to efficiendy drive a three phase motor having three stator windings. Similar closed-loop multi-stage circuits can be configured for two or four circuit stages, or any other suitable number of successively connected circuit stages, for example as might be desired for linear motors, according to the invention. The invention utilises energy that remains in a magnetic field after the field has been used to perform work, for example the mechanical work performed by the field of an electromagnetic motor. In general terms, the invention allows a magnetic field to be established in association with an inductive device (such as a transformer, motor, solenoid, or induction coil, for example). The field is predominantly established using energy recovered from the collapse of a previously-established magnetic field that may or may not be associated with the same inductive device. This recover_}' and re-use of the energy from a magnetic field allows inductive devices to be operated with improved performance and particularly with improved efficiency. Energy consumed in the circuits performing the work, through hysteresis, back emf or circuit losses can be replenished on a cycle-by-cycle basis. Significant efficiency gains can be made when these losses are kept low and are a small fraction of the energy needed to establish the magnetic field. Energy is recovered from the magnetic field associated with an inductive device, such as a winding, while the field is performing, or has performed, useful work. The recovered energy is stored on a capacitor for re-use when later re-establishing a magnetic field at that or another inductive device. Controlled switches alternately interconnect the inductive device(s) and the capacitor(s) in the magnetising and energy recovery configurations.

In one aspect the invention relates to a switching circuit for charging a capacitor by energy recovered from a collapsing magnetic field, and discharging the capacitor to re-establish the magnetic field. The voltage on the charged capacitor is compounded, over only one cycle, or over several successive cycles, of circuit operation. The capacitor is charged by the recovered energy to a voltage that is substantially greater, and is typically several times higher, than the supply voltage.

The use of this relatively high compounded voltage on the capacitor provides a steeply rising and higher value magnetising current for the inductive device. This rapid rise in magnetising current is not provided by prior art recycling circuits which recover magnetic field energy for storage on a large value reservoir or supplementary capacitor. The voltage on the reservoir or supplementary capacitor is not compounded but remains close to that of the supply and can only provide a much lower rate of rise of magnetising current at the next cycle.

In another aspect the invention relates to a switching circuit for charging a capacitor by energy recovered from a collapsing magnetic field, and re-establishing the magnetic field using energy obtained from discharging the capacitor. The capacitor may be completely discharged when re-establishing the magnetic field during each cycle of operation. However, the circuits described below can be used without fully depleting the charge on the capacitor. That is, the circuits will operate effectively with a residual charge left on the capacitor after the magnetic field has been re-established. This condition can occur when the timing of the switching of the circuit provides a magnetising period that is less than optimal, or when the capacitance of the capacitor or the inductance of the inductive device is greater than optimal. A similar condition can occur when the timing of the switching of the circuit provides a magnetising period that is greater than optimal, or when the capacitance of the capacitor or the inductance of the inductive device is less than optimal. In this case, the current that initially discharges the capacitor continues to flow without changing direction after the capacitor voltage reaches zero, and recharges the capacitor to the opposite polarity.

Dependent on the application and operating frequency, the switch timing can be controlled to optimise the current amplitude or wave shape in the inductive device, or the percentage of field energy recovered. Typically, 80-85% of the magnetic field energy can be recovered for recycling.

Many of the inductances described and shown ϋi the circuit diagrams are 'ideal' devices of fixed inductance. However, the inductance of many practical inductive devices can be significantly reduced by back emfs from secondary circuit loads or by inductances which vary rapidly over each cycle. These inductance changes may have complex profiles. For ' example, inductance changes may be linear, sinusoidal or trapezoidal, over parts of each operating cycle.

Drawing conventions

It should be noted that _/in the accompanying figures the connection between wires is shown with a dot. Wires that intersect but have no dot at the intersection are not connected.

In general, circuit components providing corresponding functions are labelled similarly throughout the following description and in the accompanying figures. For example, in each of the circuits described, controlled switches Sl and S2 perform a corresponding function of controlling the delivery of energy stored in a capacitor Cl to an inductive device Ll, and diodes Dl and D2 provide a path for a current induced in the inductive device Ll to flow back to charge the capacitor Cl. Controlled switches

The controlled switches in the circuits shown in the accompanying figures are controlled by any suitable controller (labelled SC in the figures). For example, the controller may be a microprocessor, microcontroller or other suitable digital logic or programmable device that can provide the switching devices with control pulses or signals of the required amplitude and timing. In some applications it is envisaged that the control signals provided to the switching devices by the controller will be responsive to one or more operating conditions associated with the inductive device. For example, where the inductive device is a motor, the timing of the control signals provided to the switches may be responsive to the rotational speed or shaft position of the motor, or of a component driven by the motor.

The switches are shown in some of the accompanying figures as simple switches whereas in figures relating to specific applications of some embodiments the switches are shown as field effect transistor (FET) switches.

In some applications the controlled switches may be reed switches, or mechanical switches or contact points operated by mechanical means such as roller cams, lobes, or the like. The controlled switches may be any switch suitable for the currents and voltages encountered, and having suitable switch characteristics such as switching speed, low 'on' or closed resistance, and high 'off or open resistance. Metal oxide semiconductor field effect transistor (MOSFET) switches (for example, International Rectifier IRF740LC, IRFK4HE50 or IRFK4JE50, or IXYS IXTH20N60) have been found suitable for many applications of the circuits described below. The IRFK4JE50 MOSFET (800 volt, 26 ampere, 0.046 ohm) is particularly useful where a higher voltage capability is required.

In many cases the MOSFETs can be replaced by insulated gate bipolar transistors (IGBTs) or other solid state switching devices.

In practical circuits according to some embodiments of the invention, and particularly circuits operating at higher switching frequencies, the controlled switches are preferably matched transistors having closely similar switching speeds, i.e. rise and fall times and switching turn-on and turn-off delay times.

The switches are coupled to the controller by any suitable means. The controller, the type of switches, and the coupling between controller and switches do not form part of the present invention. In some preferred embodiments, FET switches are coupled to the switch controller by optocouplers, for example HCPL-3120 from Hewlett Packard, with gate drives powered by isolated converter supplies, for example from C & D Technologies. Diodes

Some of the switching devices of the invention are semi-conductor diodes which inherently provide a closed state (i.e. a relatively low resistance path) to currents -flowing in one direction but provide an open state (i.e. a relatively high resistance padi) to currents flowing in an opposite direction. The diodes may be used alone or in conjunction with controlled switching devices. In the latter case, diodes can be used in parallel or in series with the controlled switch, depending on the switching required.

Diode switching devices are described and shown in the figures as discrete devices. However, in practice a discrete diode component may not be required. For example, where the switches in the circuits shown in Figure 1OA are MOSFETs, diodes D2A and D2B, shown as discrete diodes in parallel with respective controlled switches SlB and SlA, may each be provided by a diode that is inherently provided in the associated MOSFET switch. Where discrete semiconductor diodes are used, one suitable diode is the Intersil RHRG30120 (1200 volt, 30 ampere, ultra fast).

The semiconductor diodes require a small forward bias voltage to make the diodes conductive. This requirement has generally been ignored in the following description to simplify the explanation of circuit operation.

Although semiconductor diodes are preferred in the positions shown in the circuits of the accompanying figures, the diodes can be substituted by controlled switches. For example, diodes Dl and D2 can be substituted by controlled switches that are opened during the magnetising stage and closed during the magnetic energy recovery stage.

Capacitors

The recovery capacitors described in the following embodiments are preferably "low-loss" capacitors, i.e. capacitors having low equivalent series resistance and low equivalent series inductance. Suitable recovery capacitors are metallised polypropylene pulse capacitors, or metallised polypropylene foil-film capacitors for applications generating high voltages on the recovery capacitors.

The circuit of each embodiment described below includes one or more capacitors that temporarily store energy recovered from the collapsing magnetic field of an inductive device. These recovered energy storage capacitors are, for convenience, generally referred to by the briefer term "recovery capacitor", to help distinguish the function of these capacitors from the power supply reservoir or filter capacitors that are used in some circuits.

Circuit losses

It should also be noted that the inductors and inductive devices represented by the symbols shown in the figures are not perfect or idealised devices. In practice, these inductive components or devices also comprise resistance. Furthermore, the controlled switches and diodes used in the circuits described exhibit 'on' or closed resistances. When current is flowing, energy is dissipated in the 'on' resistance of the closed controlled switches, in the

'on' resistance of the conductive diodes, and in the winding resistance of the inductive devices. These losses are not recovered by the magnetic field energy recycling techniques described herein. In practice, the operation of the circuits described below can be affected by the resistances and other losses associated with the circuit components. For good energy recycling performance, it is preferred that these losses be kept as low as practicable by ensuring that the controllable switch "on" resistances and winding resistances are kept low.

Earthing/ Grounding The circuits shown in the figures have a bottom rail that is earthed or grounded. The earthing or grounding of this rail is optional and does not form part of the invention.

FIRST EMBODIMENT

1.1 Circuit layout

Figure IA is a circuit diagram illustrating a first embodiment of the invention. Energy recovered from an inductive device Ll is returned to a recovery capacitor Cl connected in series with the supply Vl during a second quadrant of the semi-sinusoidal current through the inductive device.

The circuit of Figure IA comprises a DC power supply Vl, two diodes Dl and D2, a capacitor Cl, two controlled switches Sl and S2, and an inductive device Ll. Switch Sl, diode Dl and supply Vl are connected in series to form one leg of an H-bridge connected between upper and lower rails. Diode D2 and switch S2 are connected in series between the upper and lower rails to form the second leg of the H-bridge. The inductive device Ll is connected between the bridge legs. The capacitor Cl is connected between the upper and lower rails. The circuit is operated by periodically switching the controlled switches Sl and S2 between open and closed states to achieve the effective circuit configurations shown in Figures 1C to IE. The opening and closing of the switches Sl and S2 are controlled by a common switch controller SC.

1.2 Switch timing

Figure IB is a switch timing diagram for the controlled switches Sl and S2 showing one cycle of operation from time t, to time t_v Switches Sl and S2 are operated synchronously over each cycle by the switch controller SC.

The switches Sl and S2 are both closed to arrange the circuit of Figure IA for a magnetising stage from time t, to time t₂. During this magnetising stage a current is driven through the inductive device Ll to establish a magnetic field. The magnetising current flows through the inductive device Ll from left to right in the circuits shown in Figures IA, 1C and ID. Switches Sl and S2 remain closed from time t, to time I₂ for a magnetising period. In some applications, it is advantageous that this magnetising period is approximately equal to 0.5 pi V (Ll Cl). However, other magnetising periods can be used. The magnetising stage ends at time t₂ at which time switches Sl and S2 are opened to arrange the circuit of Figure IA for a recovery stage from time t_j to time t₃. This is a magnetic-field-energy recovery stage during which a current induced in the inductive device Ll during collapse of the magnetic field charges capacitor Cl.

Both switches Sl and S2 are closed at time t₃ to arrange the circuit of Figure IA for the next magnetising stage. The operating cycle is repeated with a repetition period equal to

1.3 First magnetising stage circuit

Figure 1 C shows a first effective circuit for the magnetising stage of circuit operation when switches Sl and S2 are closed. This circuit applies during the magnetising stage when diode Dl is non-conductive, i.e. when the voltage on capacitor Cl is greater than the voltage of the supply Vl, reverse biasing diode Dl.

1.4 Second magnetising circuit

Figure ID shows a second effective circuit for the magnetising stage of circuit operation when switches Sl and S2 are closed. This circuit applies when the voltage across the capacitor Cl is less than the voltage of the supply Vl, making diode Dl forward biased and conductive. Magnetising current from the power supply Vl then flows through diode Dl and inductive device Ll, and back through closed switch S2 to contribute to the establishment of the magnetic field in association with the inductive device Ll .

1.5 Magnetising circuit conversion

The conversion of the magnetising circuit of Figure 1C to that of Figure ID occurs automatically during the magnetising stage. The conversion occurs when the voltage on capacitor Cl falls below, or is less than, that of the supply Vl and there is insufficient charge on the recovery capacitor to supply all the magnetising current for the full magnetising period t, to t₂. This conversion occurs immediately on first closing switches Sl and S2 at time t_j of the first cycle of operation, when there is no charge on the capacitor, but can occur progressively later in subsequent cycles. In these subsequent cycles, the recovery capacitor can charge to progressively higher voltages as the circuit builds up to an operating mode.

1.6 Energy recovery circuit

Figure IE shows an effective circuit for the energy recovery stage of circuit operation when switches Sl and S2 are both opened at time t₂. This stage continues from time t₂ to time t₃. 1.7 Start-up mode magnetising - first cycle

Initial start-up of the circuit occurs when the switches Sl and S2 close at time t_j of the first cycle of operation. For the purposes of the immediately-following explanation it is assumed that, prior to time t,, capacitor Cl is uncharged. At time t,, switches Sl and S2 close, effectively arranging the circuit as shown in Figure ID. Magnetising current then flows from the supply Vl through diode Dl, inductive device Ll and closed switch S2 to establish a magnetic field in association with the inductive device Ll. 1.8 Start-up mode magnetising - subsequent cycles

On subsequent cycles during start-up operation, the capacitor Cl will already, at time t, have been charged during a previous recovery stage to a voltage higher than that of the supply Vl. The circuit then adopts the configuration shown in Figure 1C. In this configuration, magnetising current flows from the pre-charged capacitor Cl through closed switch Sl, inductive device Ll and closed switch S2 to establish a magnetic field in association with the inductive device Ll .

This flow of current out of the capacitor Cl depletes the charge on the capacitor which decreases the voltage across the capacitor. If the voltage across the capacitor Cl decreases below that of the supply Vl, diode Dl becomes forward biased and conductive. This switching of the conductive states of diode Dl automatically converts the effective circuit from that shown in Figure 1C to that shown in Figure ID, and magnetising current continues to flow, now from the supply Vl. 1.9 Run mode magnetising

In the run mode, die magnetising current for die inductive device Ll is predominandy derived from the discharge of capacitor Cl by the circuit of Figure 1C.

During a first substantial part of the run mode magnetising stage, the capacitor Cl is connected by closed switches Sl and S2 to the inductive device Ll as seen in the circuit of Figure 1C, to re-establish the magnetic field in the inductive device Ll. When or if the voltage on the discharging capacitor Cl falls below the voltage of the supply Vl, diode Dl conducts to maintain magnetising of the inductive device Ll by current flowing from the supply Vl, through diode Dl to inductive device Ll and back through switch S2. This effective circuit is shown in Figure ID. This continues the magnetising of the inductive device Ll with energy direct from die supply Vl. This automatic replenishment of the circuit occurs during every cycle upon depletion of the capacitor Cl and draws energy from the supply to make up for losses in the circuit or to complete the magnetising current pulse when there is insufficient recovery energy to do so.

1.10 Energy recovery

Figure IE shows an effective circuit for the energy recovery stage of circuit operation when switches Sl and S2 are both opened at time t₂. This stage continues from time t₂ to time t_v At time t,, the current through the inductive device Ll and the associated magnetic field begin to collapse. The collapsing current flows from inductive device Ll through diode D2 to capacitor Cl and back through the supply Vl and diode Dl to inductive device Ll. This current flows through the inductive device Ll in the same direction as the current used to establish the magnetic field (i.e. from left to right in Figure IE), but flows into the capacitor Cl in the opposite direction to the magnetising current flowing from the capacitor Cl during the magnetising stage.

The flow of the induced current, from the inductive device Ll back to the capacitor Cl, recharges the capacitor to effectively transfer energy that is contained in the magnetic field to the capacitor Cl. This recovered energy is used to re-establish the magnetic field during the magnetising stage of the next cycle of operation.

1.11 Voltage multiplication

On initial start-up, the capacitor Cl is charged, in the energy recovery stages of the first few successive cycles of circuit operation, to progressively higher voltages. After only a few cycles of operation the capacitor Cl is recharged at each recovery stage to several times the supply voltage. In the magnetising stages, the inductive device is magnetised from this capacitor voltage.

The recovery of energy from the collapsing magnetic field at each cycle and its re-use to reestablish the field in the magnetising stage at the next cycle effectively multiplies the voltage from which the inductive device is driven to provide a significant improvement in efficiency. The voltage multiplication process is similar to the transient charging phase of a resonant inductance-capacitance (L-C) circuit.

The capacitor Cl discharges over progressively longer times, and with progressively higher peak current values, during respective magnetising stages of each of the first few start-up cycles.

1.12 Energy transfer

When discharge current from the capacitor Cl is magnetising the inductive device Ll during the earlier part of the magnetising stages, before magnetising from the supply alone takes over, the circuit is effectively capacitor Cl series connected by switches Sl and S2 to inductive device Ll .

For optimum operation of the energy recover₎' circuit shown in Figure IA, the recovered energy stored as a charge on capacitor Cl must be efficiently transferred back to the magnetic field associated with the inductive device Ll. Maximum transfer of energy from the capacitor back to the inductive device occurs when the voltage on the capacitor Cl has decreased from a maximum to substantially equal to the voltage of supply Vl, and the current in the inductive device Ll has simultaneously risen from zero to a maximum. The time for this to occur is equal to a quarter of the period of natural resonance of the inductance-capacitance (L-C) circuit, which in this case is equal to 0.5 pi V (Ll Cl).

For optimum operation of the energy recovery circuit of Figure IA, the switches Sl and S2 are closed for each cycle of operation for a time that is approximately equal to 0.5 pi v (Ll Cl) to allow for optimum transfer of energy from the capacitor Cl to the inductive device Ll .

The switches Sl and S2 may be maintained closed for a small additional time period to extend the duration of the magnetising current in the inductive device Ll. During this extension period, the magnetising current can be supplied from the supply to compensate for circuit losses.

1.13 Preferred embodiment

One preferred embodiment of the circuit shown in Figure IA has die following circuit values:

Sl and S2: IRFK4HE50

Dl and D2: RHRG30120

Vl = 48 volts

Cl = 300 μF

Ll = 36 mH (with an effective series resistance of 0.5 ohms)

Switching period t, to t, = 20 mS

Switching frequency— 50 Hz

Duration of magnetising stage t, to t₂ = 5 mS

In this embodiment, the switches Sl and S2 remain closed for 5 mS over the 20 mS period of each cycle. One quarter of the resonant period of the capacitor Cl and inductive device Ll, i.e. 0.5 pi V (Ll Cl), is equal to 5.2 mS, slightly longer than the time period in each cycle that the switches Sl and S2 are closed.

In this embodiment, the capacitor Cl is recharged at each recovery stage to a voltage of over 600 volts, which is more than 12 times the supply voltage after the first 40 cycles of operation, i.e. after only 800 mS from starting. The inductive device is magnetised from this capacitor voltage, giving an effective multiplication of the supply voltage.

Figure IH shows the successive increase in voltage on the recovery capacitor Cl during start-up. Figure II shows a typical waveform of the voltage on the recovery capacitor Cl for two cycles during a run-mode.

1.14 Waveforms

Figures IF, IG, IH and II show typical waveforms of currents and voltages for the preferred first embodiment of the circuit shown in Figure IA. The upper waveforms of

Figures IF and IG shows typical supply current waveforms. The lower waveforms of

Figures IF and IG show typical current waveforms for the inductive device Ll. Figures

IH and II show typical waveforms of the voltage across the recovery capacitor Cl, i.e. between the upper and lower circuit rails. Figure IF and IH show several cycles during start-up. Figures IG and II show run-mode cycles.

In the run-mode of the preferred third embodiment of the Figure IA circuit as described above, the magnetising current in the inductive device Ll rises from zero to a peak of approximately 54 amperes with a waveform that is very close to one quarter cycle of sinusoid. This may be best appreciated from the lower waveform in Figure 1 G, from 800 mS to 805 mS. At the end of the magnetising stage, when the switches Sl and S2 are opened, current induced in the inductive device Ll , by the collapsing magnetic field, falls to zero with a waveform that is very close to the second quarter cycle of a sinusoid. This may be best appreciated from the lower waveform in Figure IG, from 805 mS to 810 mS. The current in the inductive device Ll then remains at zero until the start of the next cycle at 820 mS. In summary, the waveform of the current in the inductive device is substantially a half sinusoid for each cycle of operation.

As shown in the upper waveform of Figure IG, the supply current only flows during the recovery stage, i.e. during the second quarter, or quadrant, of the sinusoid. The supply current waveform is a quarter sinusoid, rising rapidly from zero to approximately 54 amperes (for example, at 805 mS) when the switches Sl and S2 are opened, then falling with a quarter sinusoid shape to zero (for example, between 805 mS and 810 mS). In the run mode, the supply is connected in series with the capacitor Cl to provide a replenishment to make up for circuit losses. SECOND EMBODIMENT

2.1 Circuit layout

Figure 2A is a circuit diagram illustrating a second embodiment of the invention. In diis second embodiment, a capacitor Cl stores energy recovered from an inductive device Ll for providing a re-magnetising current for the inductive device during a first quadrant of a semi-sinusoidal current through the inductive device. The injection of energy into the circuit from a supply Vl is controlled by an additional switch S3 which connects the supply in series with the capacitor just before the end of the first quadrant. Efficiency gains can be made over the second embodiment by limiting the duration of energy injection from the supply.

The circuit of Figure 2A comprises a DC power supply Vl, three diodes Dl, D2 and D5, a capacitor Cl, three controlled switches Sl, S2 and S3, and an inductive device Ll. Switch Sl, diode Dl, switch S3 and supply Vl are connected in series between upper and lower rails to form a first leg of an H-bridge. Diode D2 and switch S2 are connected in series between the upper and lower rails to form the second leg of the H-bridge. The inductive device Ll is connected between the bridge legs. The power supply Vl and the capacitor Cl are connected in series between the upper and lower rails when switch S3 is closed. The circuit is operated by periodically switching the controlled switches Sl, S2 and S3 between open and closed states to achieve the effective circuit configurations shown in Figures 2C to 2F. The opening and closing of the switches Sl, S2 and S3 are controlled by a common switch controller SC.

2.2 Switch timing

The switches Sl, S2 and S3 are controlled over an initial circuit start-up time in a start-up mode and then revert to a run mode. Figure 2B is a switch timing diagram for the controlled switches Sl, S2 and S3 in the run mode. Switches Sl and S2 are closed and opened synchronously under control of switch controller SC over each cycle of operation from time t_t to time t₃ according to the timing shown in Figure 2B to provide respective magnetising and recovery stages in the run mode. Switch S3 is closed during the latter part of the magnetising stage at injection time t, and remains closed until the end of that magnetising stage at time t₂.

In the start-up mode, switches Sl and S2 are closed and opened synchronously over each cycle of operation from time t, to time t₃ according to the timing shown in Figure 2B to provide respective magnetising and recovery stages. Switch S3 is closed for the duration of the start-up mode.

The switches Sl and S2 are both closed to arrange the circuit of Figure 2A for a magnetising stage of the operating cycle from time t, to time t₂. During this magnetising stage a current is driven through the inductive device Ll to establish a magnetic field. The magnetising current flows through the inductive device Ll from left to right in the circuits shown in Figures 2A, 2C, 2D and 2E.

Switches Sl and S2 remain closed from time t, to time t₂ for a period that is approximately equal to 0.5 pi V (Ll Cl). The magnetising stage ends at time t₂ at which time switches Sl and S2 are opened to arrange the circuit of Figure 2A for a recovery stage from time t₂ to time t_v This is a magnetic- field-energy recovery stage during which a current induced in the inductive device Ll during collapse of the magnetic field charges capacitor Cl. Both switches Sl and S2 are closed at time t, to arrange the circuit of Figure 2A for the next magnetising stage. The operating cycle is repeated with a repetition period equal to (t₃ - t,).

In the run mode, switch S3 is closed at inject time t, to inject current from the supply Vl into the circuit during the latter part of the magnetising stage. In the start-up mode, switch S3 remains closed to provide injection of current from the supply Vl over the full duration of the magnetising stage. 2.3 First magnetising circuit

Figure 2C shows a first effective circuit for the first part of the magnetising stage of circuit operation during the run mode, when switches Sl and S2 are closed and switch S3 is open, i.e. from time t, to time t,.

2.4 Second magnetising circuit

Figure 2D shows a second effective circuit for the magnetising stage of circuit operation, when switch S3 is closed (either during the start-up mode, or after supply injection time t, during the run mode) and when the voltage across the capacitor Cl is sufficient to reverse bias diode Dl.

Magnetising current is injected from the power supply Vl to flow through closed switch S3, capacitor Cl, closed switch Sl and inductive device Ll, and back through closed switch S2. This injection of supply current into the circuit contributes to the establishment of the magnetic field in association with the inductive device Ll .

2.5 Third magnetising circuit

Figure 2E shows a third effective circuit for the magnetising stage of circuit operation when switch S3 is closed (either during the start-up mode, or during the run mode after supply injection time t), and when the voltage across the capacitor Cl is insufficient to reverse bias diode Dl. Diode Dl then becomes forward biased and conductive, effectively bypassing the capacitor Cl and closed switch Sl, to provide the circuit shown in Figure 2E. .

Magnetising current injected from the power supply Vl then flows through closed switch S3, diode Dl and inductive device Ll, and back through closed switch S2 to inject supply current into the circuit and maintain establishment of the magnetic field in association with the inductive device Ll . It is to be noted that this effective circuit is only utilised if the voltage on capacitor Cl drops sufficiently to make diode Dl conductive. This may not occur if the duration of the magnetising stage period from time t, to time t, is kept shorter than approximately 0.5 pi v (Ll Cl).

2.6 Magnetising circuit conversion The timing of the conversion of the magnetising circuit of Figure 2C to that of Figure 2D occurs when switch S3 is closed at supply injection time t, under control of switch S3. This control of this circuit conversion is in contrast to the automatic conversion provided by the circuits of the first, second and third embodiments described above.

2.7 Energy recovery circuit

Figure 2F shows an effective circuit for the energy recovery stage of circuit operation when controlled switches Sl, S2 and S3 are opened at time I₂. This stage continues from time t₂ to time t₃.

2.8 Start-up mode magnetising - first cycle

Initial start-up of the circuit occurs when the switches Sl, S2 and S3 close at time t, of the first cycle of operation to effectively arrange the circuit as shown in Figure 2E. For the purposes of the following explanation it is assumed that, prior to this time t,, capacitor Cl is uncharged. Magnetising current flows from the supply Vl through closed switch S3, diode Dl, inductive device Ll and closed switch S2. The current flowing through the inductive device Ll establishes a magnetic field in association with the inductive device Ll .

2.9 Start-up mode magnetising - subsequent cycles

On subsequent cycles during start-up operation, there will, at least initially, be some charge on capacitor Cl. In this case the circuit adopts the configuration shown in Figure 2D and magnetising current flows from the series connection of supply Vl , closed switch S3 and pre-charged capacitor Cl, through closed switch Sl, inductive device Ll and back through closed switch S2 to establish a magnetic field in association with inductive device Ll . This current flow depletes the charge on capacitor Cl, decreasing the voltage across the capacitor.

When, in these subsequent cycles during start-up operation, the voltage across the capacitor Cl becomes insufficient to maintain a reverse bias on diode Dl, diode Dl becomes conductive and the circuit automatically reverts to that shown in Figure 2E. The capacitor Cl discharges over a progressively longer time during the magnetising stage of each of these subsequent start-up cycles. After depletion of voltage on capacitor Cl, magnetising current in the inductive device is provided from supply Vl via closed switch S3 and diode Dl, with a return path to earth or ground through closed switch S2. Diode Dl prevents capacitor Cl from becoming substantially reverse charged.

2.10 Run mode magnetising

In the run mode, the magnetising current from the inductive device Ll is predominantly derived from the discharge of capacitor Cl by the circuit of Figure 2C. During a first substantial part of the run mode magnetising stage of each run mode cycle, the capacitor Cl is connected by switches Sl and S2, and diode D5, to the inductive device Ll, as seen in the circuit of Figure 2C. Current flows from charged capacitor Cl, through closed switch Sl, inductive device Ll and closed switch S2, and back through diode D5, to re-establish the magnetic field in association with the inductive device.

At supply injection time t,, switch S3 is closed to effectively convert the circuit to that shown in Figure 2D. Magnetising current in the inductive device Ll is then maintained by current flowing from the supply Vl, through switch S3, capacitor Cl and switch Sl, to inductor L2, and back through switch S2, as seen in the circuit of Figure 2D. This continues the magnetising current in the inductive device Ll using energy direct from the supply. This replenishment draws energy from the supply during every cycle to make up for losses in the circuit.

The replenishment voltage provided by the supply Vl is less than the voltage provided by the charged capacitor Cl. However, the lower voltage from the supply is sufficient to maintain the level of current in the inductive device Ll and prolong the magnetising current begun by the current flow from the capacitor Cl.

2.11 Energy recovery

Figure 2F shows the effective circuit for the energy recover}' stage of circuit operation when controlled switches Sl, S2 and S3 are opened at time t₂. This stage continues from time t₂ to time t,.

At time t,, the current flowing through the inductive device Ll and the associated magnetic field begin to collapse. The collapsing current flows from inductive device Ll through diode D2 to capacitor Cl and back through diode Dl to inductive device Ll. The current induced by the collapsing magnetic field flows through the inductive device Ll in the same direction as the current used to establish the magnetic field, (i.e. from left to right in Figure 2F) but flows into the capacitor Cl in the opposite direction to the magnetising current flowing from the capacitor Cl during the magnetising stage. The capacitor Cl is recharged 5 by the induced current.

The flow of the induced current from the inductive device Ll back to the capacitor Cl effectively recovers energy that is contained in the magnetic field and transfers the energy to the capacitor Cl. This recovered energy is used to re-establish the magnetic field at the 10 magnetising stage of the next cycle of operation.

2.12 Voltage multiplication

On initial start-up, the capacitor Cl is charged, in the energy recovery stages of the first few successive cycles of circuit operation, to progressively higher voltages. After only a few ^"15 cycles of operation the capacitor Cl is recharged at each recovery stage to several times the supply voltage. In the magnetising stages, magnetising current in the inductive device is driven by this capacitor voltage.

The recovery of energy from the collapsing magnetic field at each cycle and its re-use to re- 20 establish the field in the magnetising stage at the next cycle effectively multiplies the voltage from which the inductive device is driven to provide a significant improvement in efficiency. The voltage multiplication process is similar to the transient charging phase of a resonant inductance-capacitance (L-C) circuit.

25 2.13 Energy transfer

The supply Vl has an effective capacitance that is many times greater than the capacitance of capacitor Cl, giving the series combination of the supply Vl and the capacitor Cl an effective capacitance value substantially equal to the capacitance of capacitor Cl. When the capacitor Cl and supply are together providing the magnetising current for the inductive 30 device Ll during the earlier part of the magnetising stage, before magnetising from the supply alone takes over, the circuit is effectively a capacitance equal to that of capacitor Cl series connected by switches Sl, S2 and S3 to inductive device Ll . For optimum operation of the energy recovery circuit shown in Figure 2A, the recovered energy stored as a charge on capacitor Cl must be efficiently transferred back to the magnetic field associated with the inductive device Ll . Maximum transfer of energy from the capacitor back to the inductive device occurs when the voltage on the capacitor Cl has decreased from a maximum to zero and the current in the inductive device Ll has simultaneously risen from zero to a maximum. The time for this to occur is equal to a quarter of the period of natural resonance of the inductance-capacitance (L-C) circuit, which is equal to 0.5 pi V (Ll Cl). For optimum operation of the energy recovery circuit of Figure 2A, the switches Sl and S2 ^- are closed for each cycle of operation for a time that is approximately equal to 0.5 pi V (Ll Cl) to allow for optimum transfer of energy from the capacitor Cl to the inductive device Ll . If the switches Sl and S2 are kept closed for a small additional time period, the duration of the magnetising current in the inductive device Ll can be extended. During this extension period, the magnetising current can be supplied from the supply to compensate for circuit losses. 2.14 Preferred embodiment

One preferred embodiment of the circuit shown in Figure 2A has the following circuit values:

Sl, S2 and S3: IRFK20450

Dl, D2 and D5: RHRG30120

Vl = 48 volts

Cl = 280 μF

Ll = 36 mH (with an effective series resistance of 0.5 ohms)

Switching period t, to t, = 20 mS

Switching frequency = 50 Hz

Duration of magnetising stage t, to t, = 5 mS

Supply injection delay time t, to t, = 3.5 mS

Duration of supply injection t, to t,— 1.5 mS

Duration of start-up mode 100 mS In this embodiment, the switches Sl and S2 remain closed for 5 mS over the 20 mS period of each cycle to provide the magnetising stage. In the run mode, S3 is closed for 1.5 mS beginning at 3.5 mS after the beginning of each cycle to provide the supply injection. Switch S3 is also closed for the full duration of the start-up period of 100 mS. One quarter of the resonant period of the capacitor Cl and inductive device Ll, i.e. 0.5 pW (Ll Cl), is approximately equal to 5 mS which is the time period in each cycle that the switches Sl and S2 are closed. In this embodiment, the capacitor Cl is recharged at each recovery stage in the run mode to approximately 245 volts, a voltage that is more than 5 times the supply voltage, to give an effective supply voltage multiplication.

Figure 21 shows the successive increase in voltage between upper and lower circuit rails during start-up. Figure 2J shows a typical waveform of the voltage between the upper and lower circuit rails for two cycles during a run-mode. Figure 2J shows the voltage between the upper and lower rails falling as the recovery capacitor Cl discharges into the inductive device Ll, from 180 mS to 183.5 mS. When the switch S3 is closed at 183.5 mS, the voltage between the upper and lower rails initially steps up by 48 volts but then falls steadily, as the recovery capacitor continues to discharge into the inductive device Ll, until 185 mS. Switch S3 is opened at 185 mS causing the voltage between the upper and lower rails to drop suddenly by 48 volts. In this preferred fourth embodiment, a small residual charge remains on the capacitor Cl. Switches Sl and S2 are also opened at 185 mS to convert the circuit to the recovery mode. During the recovery mode, the capacitor Cl is recharged back up to about 245 volts ready for the next cycle which begins at 200 mS.

2.15 Waveforms

Figures 2G, 2H, 21 and 2J show typical waveforms of currents and voltages for the preferred fourth embodiment of the circuit shown in Figure 2A. The upper waveforms of Figures 2G and 2H show typical supply current waveforms. The lower waveforms of Figures 2G and 2H shows typical current waveforms for the inductive device Ll. Figures 21 and 2J show typical waveforms of the voltage between the upper and lower circuit rails. Figure 2G and 21 show several cycles during start-up. Figures 2H and 2J show run-mode cycles. In the run-mode of the preferred fourth embodiment of the Figure 2A circuit having the circuit values described above, the magnetising current in the inductive device Ll rises from zero to a peak of approximately 22.5 amperes with a waveform that is close to one quarter cycle of sinusoid. This may be best appreciated from the lower waveform in Figure 2H, from 180 mS to 185 mS. At the end of the magnetising stage, when the switches Sl, S2 and S3 are opened, current induced in the inductive device Ll, by the collapsing magnetic field, falls to zero with a waveform that is very close to the second quarter cycle of a sinusoid. This may be best appreciated from the lower waveform in Figure 2H, from 185 mS to 190 mS. The current in the inductive device Ll then remains at zero until the start of the next cycle at 200 mS. In summary, the waveform of the current in the inductive device is substantially a half sinusoid for each cycle of operation.

As shown in the upper waveform of Figure 2H, current only flows from the supply when switch S3 is closed, for example from 183.5 mS to 185 mS. As already described above, switch S3 is closed during an initial start-up period. In the preferred fourth embodiment switch S3 is closed for an initial start-up period of 100 mS. As may be seen from the upper waveform of Figure 2G, the continuous closure of switch S3 during the initial 100 mS allows current to flow from the supply during the full 5 mS of each magnetising stage, i.e. from 0 to 5 mS, from 20 to 25 mS, from 40 to 45 mS, from 60 to 65 mS, and from 80 to 85 mS. After that initial 100 mS start-up period, switch S3 is only closed over the latter 1.5 mS of each 5mS magnetising period, i.e. from 103.5 to 105 mS, from 123.5 to 125 mS, from 143.5 to 145 mS, etc. Figure 21, which shows the waveform of the voltage between the upper and lower rails, shows that this voltage steps up very rapidly over the first 5 cycles of operation, i.e. during the 100 mS start-up period when switch S3 is continuously closed.

THIRD EMBODIMENT 3.1 Circuit layout

Figure 3A is a circuit diagram illustrating a third embodiment of the invention. This embodiment has a dual voltage supply controlled by an additional switch S3 which is closed to switch the dual supply from a lower to a higher voltage. The higher voltage is provided by connecting a power supply Vl and a power supply V2 in series by closing switch S3. When switch S3 is open, a diode D5 bypasses supply Vl, leaving only the supply V2 to power the circuit.

Current from the dual voltage supply is injected into the circuit at the higher voltage, on depletion of energy from the recovery capacitor, to replenish energy lost to circuit losses. Supply current is also injected into the circuit at the lower voltage to extend the duration of the peak of the magnetising current through the inductive device. This is useful in motor drive circuits where the increased width of the magnetising current pulse provides more magnetic force for providing mechanical energy.

The Figure 3A third embodiment circuit, which provides near-sinusoidal magnetising current waveforms, provides full field energy recovery and voltage compounding, and efficiencies similar to those achieved by the first embodiment which provides sinusoidal magnetising current waveforms.

The circuit of Figure 3A comprises two DC power supplies Vl and V2, six diodes Dl, D2, D3, D5, D6 and DlO, a capacitor Cl, three controlled switches Sl, S2 ^"and S3, and an inductive device Ll. The power supply V2 is connected in series with the power supply Vl by controlled switch S3. The voltage ratio Vl /V2 of the two supplies typically ranges from about 3/1 to 20/1.

A bypass diode D5 is connected across the series combination of switch S3 and supply Vl to provide a current path for supply V2 when the switch S3 is open. Switch Sl and diodes DlO, Dl and D6 are connected in series between upper and lower rails to form a first leg of an H-bridge. Diode D2 and switch S2 are connected in series between the upper and lower rails to form the second leg of the H-bridge. The inductive device Ll is connected between the bridge legs. The circuit is operated by periodically switching the controlled switches Sl, S2 and S3 between open and closed states to achieve the effective circuit configurations shown in Figures 3C to 3G. The opening and closing of the switches Sl, S2 and S3 are controlled by a common switch controller SC.

3.2 Switch timing

Figure 3B is a switch timing diagram for the controlled switches Sl, S2 and S3 showing one cycle of operation from time t_j to time t₃. The switches Sl, S2 and S3 are closed simultaneously at time t, at the beginning of each cycle to arrange the circuit of Figure 3A in a first magnetising configuration from time t, to time tg₃. Switch S3 is opened at supply switching time tg₃ to arrange the circuit into a second magnetising configuration from time t_j3 to time t_j,. Switch Sl is opened at time tg, to arrange the circuit into a third magnetising configuration from time ^₁ to time ^₂. The magnetising stage ends at time ^₂ when switch S2 is opened to arrange the circuit of Figure 3A in a recovery configuration from time ^₂ to time t₃. This is a magnetic-field-energy recovery stage during which a current induced in the inductive device Ll during collapse of the magnetic field charges a recovery capacitor Cl. Recovery of magnetic field energy may be completed before time t₃.

During the magnetising stage, current is driven through the inductive device Ll to establish a magnetic field. This magnetising current flows through the inductive device Ll from left to right in the circuits shown in Figures 3A, 3C, 3D and 3E.

Magnetising current is drawn from the recovery capacitor Cl during the magnetising stage for one or more periods that in total approximately equal 0.5 pi V (Ll Cl).

After the recovery stage, all three switches Sl, S2 and S3 are closed at time t, to arrange the circuit of Figure 3A for the next magnetising stage. The operating cycle is repeated with a repetition period equal to (t₃ - 1,).

3.3 First magnetising circuit

Figure 3C shows a first effective circuit for the magnetising stage of circuit operation when switches Sl, S2 and S3 are closed. Current flows from charged capacitor Cl, through closed switch Sl, diode DlO, inductive device Ll, closed switch S2 and back to the capacitor Cl through diode D6, to establish a magnetic field in association with the inductive device. The circuit of Figure 3C applies during the magnetising stage when diodes D6 and DlO are conductive and diode D3 is non-conductive, i.e. when the voltage on capacitor Cl is greater than that of the series connection of the two supplies Vl and V2. Diode D5 is non-conductive because of the reverse bias provided by supply Vl through closed controlled switch S3.

The circuit of Figure 3C also applies after switch S3 opens at time tg₃ and before the earlier of the capacitor Cl discharging to a voltage less than that of supply V2, or the switch Sl opening at time ^₁.

3.4 Second magnetising circuit

Figure 3D shows a second effective circuit for the magnetising stage of circuit operation when switches Sl, S2 and S3 are closed. Current flows from the series connection of supplies Vl and V2 (connected in series by closed switch S3), through diode D3, inductive device Ll and closed switch S2, to establish a magnetic field in association with the inductive device. This circuit applies during the magnetising stage when diode D3 is conductive and diodes D6 and DlO are non-conductive, i.e. when the voltage on capacitor Cl is less than that of the series connection of supplies Vl and V2.

3.5 Third magnetising circuit

Figure 3E shows another effective circuit for the magnetising stage of circuit operation. This circuit applies after switch S3 has opened at time t_S3 and capacitor Cl has discharged to a voltage less than that of supply V2 or switch Sl opens at time t_sl. Diode D5 is then forward biased and conductive, bypassing the supply V2 and open switch S3. Diode D3 is forward biased and conductive. The voltage on capacitor Cl is less than that of supply V2, making diodes D6 and DlO reverse biased and non-conductive. Magnetising current from supply V2 flows through diode D3 to inductive device Ll, and back through closed switch S2 and diode D5 to contribute to the establishment of the magnetic field in association with the inductive device Ll . This circuit applies through to time t_S2 when switch S2 opens.

3.6 Magnetising circuit conversion The first magnetising circuit of Figure 3C converts to the second magnetising circuit of Figure 3D automatically when the voltage on capacitor Cl falls below, or is less than, the voltage provided by the series connection of the two supplies Vl and V2. This occurs immediately on first closing switches Sl, S2 and S3 at time t, of the first cycle of operation because capacitor Cl is uncharged, but occurs progressively later in successive subsequent cycles. In the first several subsequent cycles, the recovery capacitor charges to progressively higher voltages as the circuit builds up to an operating or run mode. In the operating mode, capacitor Cl is left charged, after the conversion to the second magnetising circuit of Figure 3D, with a voltage approximately equal to the summation of the voltages of supplies Vl and V2.

The second magnetising circuit of Figure 3D then converts back to the first magnetising circuit of Figure 3C when switch S3 is opened at time t^. At this time, the voltage of the supply is switched from voltage Vl plus voltage V2, to voltage V2 only. The capacitor Cl now discharges further, with the capacitor voltage falling from approximately equal to the summation of the voltages of supplies Vl and V2, to the voltage of supply V2.

When the capacitor voltage falls to, or below, that of the supply V2, the circuit configuration converts from the first magnetising circuit of Figure 3C to the third magnetising circuit of Figure 3E. The lower voltage supply V2 continues to provide magnetising current through to the end of the magnetising period at time t_S2 when switch S2 is opened.

3.7 Energy recovery circuit

Figure 3F shows an effective circuit for the energy recovery stage of circuit operation when switch S2 is opened at time t_S2, (Sl and S3 having earlier been opened at respective times t_S] and t_s,). This energy recovery stage continues from time t_S2 to time t_v

3.8 Start-up mode magnetising - first cycle

Initial start-up of the circuit occurs when the switches Sl, S2 and S3 close at time t, of the first cycle of operation. For the purposes of the immediately-following explanation it is assumed that, prior to this initial time t,, capacitor Cl is uncharged. At time t,, switches Sl, S2 and S3 close to effectively arrange the circuit as shown in Figure 3D. The supply Vl, connected by closed switch S3, makes diode D5 reversed biased and non-conductive. With capacitor Cl uncharged, diode D3 is forward biased and conductive, and diodes Dl, D 6 and DlO are reverse biased and non-conductive. Magnetising current then flows from the series connection of the supplies Vl and V2, through diode D3, inductive device Ll and closed switch S2 to establish a magnetic field in association with the inductive device Ll .

At supply switching time ^₃ switch S3 opens to effectively arrange the circuit as shown in Figure 3E. Magnetising current then flows, from supply V2 only, through diode D3, inductive device Ll, closed switch S2 and diode D5, to maintain the magnetic field established in association with the inductive device Ll. This magnetising current continues until switch S2 is opened at time ^₂. 3.9 Start-up mode magnetising - subsequent cycles

On subsequent cycles during start-up operation, the capacitor Cl will already, at time t, have some charge from energy recovery from one or more previous cycles. If the recovery capacitor is already charged to a voltage higher than the combined voltages of the two supplies Vl and V2, the circuit will adopt the configuration shown in Figure 3C from the beginning of the cycle at time t,. The pre-charged capacitor Cl discharges to provide magnetising current which flows through closed switch Sl, diode DlO, inductive device Ll, closed switch S2 and diode D6.

When the voltage across the discharging capacitor Cl falls below the combined voltage of the two supplies Vl and V2, and therefore becomes insufficient to maintain the reverse bias on diode D3 and the forward bias on diodes D6 and DlO, diode D3 conducts and diodes D6 and DlO become non-conductive, automatically converting the circuit to that shown in Figure 3D. This conversion interrupts the discharge of the recovery capacitor and the voltage on the capacitor then remains at approximately equal to the combined voltage of the two supplies Vl and V2. Magnetising current then continues to flow, but from the series connection of the two supplies Vl and V2. This maintains the rising magnetising current begun by discharge of the recovery capacitor Cl. The automatic conversion from the circuit configuration of Figure 3C to that of Figure 3D occurs progressively later in successive start-up cycles as the recovery capacitor Cl is charged to successively higher voltages and the circuit builds up to a fully operational run mode.

At supply switching time t^ switch S3 is opened to disconnect supply Vl and lower the voltage supplied to the circuit to that of the supply V2 only. The circuit reverts back to that as shown in Figure 3C because the voltage then remaining on the capacitor Cl is greater than the voltage of supply V2, making the diodes D6 and DlO forward biased and conductive, and diodes Dl, D3 and D5 reverse biased and non-conductive. Capacitor Cl again discharges, to provide magnetising current that flows through closed switch Sl, diode DlO, inductive device Ll, closed switch S2 and diode D6.

When the voltage across the discharging capacitor Cl falls below the voltage of the supply V2, and therefore becomes insufficient to maintain the reverse bias on diodes D3 and D5, and the forward bias on diodes D6 and DlO, diodes D3 and D5 conduct and diodes D6 and DlO become non-conductive, automatically converting the circuit to that shown in

Figure 3E. Magnetising current continues to flow, but from the supply V2 only, through diode D3, inductive device Ll, closed switch S2 and diode D5. This maintains the magnetising current, at about the level already established by discharge of the recovery capacitor Cl, until switch S2 is opened at time t_S2.

3.10 Run mode magnetising

In the run mode, the magnetising current for the inductive device Ll is predominantly derived from the discharge of capacitor Cl by the circuit of Figure 3C.

At the start of the run mode cycle, at time t,, the circuit adopts the configuration shown in Figure 3C. The recover_}' capacitor is already charged to a voltage higher than the combined voltages of the two supplies Vl and V2. Magnetising current flows from the pre-charged capacitor Cl, through closed switch Sl, diode DlO, inductive device Ll, closed switch S2 and diode D6. When the voltage across the capacitor Cl falls below the combined voltage of the two supplies Vl and V2, and therefore becomes insufficient to maintain the reverse bias on diode D3 and the forward bias on diodes D6 and DlO, diode D3 conducts and diodes D6 and DlO becomes non-conducting to automatically convert the circuit to that shown in Figure 3D. This conversion interrupts the discharge of the recovery capacitor and the voltage on the capacitor then remains at approximately equal to the combined voltage of the two supplies Vl and V2. Magnetising current continues to flow, but from the two supplies Vl and V2 connected in series by closed switch S3, through diode D3, inductive device Ll and closed switch S2. The magnetising current, begun by discharge of the recovery capacitor Cl, but now supplied from the series-connected supplies Vl and V2, continues until time t^.

At time ts_j switch S3 opens, removing the series connection between the two supplies Vl and V2, to effectively drop the supply voltage to that of supply V2 only. Diode D3 becomes reverse biased and non-conductive, diodes D6 and DlO become forward biased and conductive, and the circuit converts to that shown in Figure 3C. Magnetising current continues to flow from the capacitor Cl, through closed switch Sl, diode DlO, inductive device Ll, closed switch S2 and diode D6. Capacitor Cl discharges and the capacitor voltage falls from a voltage approximately equal to the combined voltage of supplies Vl and V2, down to a voltage approximately equal to supply V2.

When the capacitor Cl voltage falls below that of the supply V2, the circuit configuration converts from the first magnetising circuit of Figure 3C to the third magnetising circuit of Figure 3E. This maintains the magnetising current at about the level already established by discharge of the recovery capacitor Cl. The slope of the magnetising current waveform over this period can be made positive, zero or negative by appropriate selection of the voltage of the supply V2. The supply V2 continues to provide magnetising current through to the end of the magnetising period at time t_s2 when switch S2 opens. 3.11 Energy recovery

Figure 3F shows an effective circuit for the energy recovery stage of circuit operation when switch S2 is opened at time t_S2. At time t_S2, the current through the inductive device Ll and the associated magnetic field begin to collapse. This recovery stage continues from time 1^₂ to time t₃.

The collapsing current flows from the inductive device Ll through diode D2 to capacitor Cl and back through diode Dl to inductive device Ll. This current flows through the inductive device Ll in the same direction as the current used to establish the magnetic field

(i.e. from left to right in Figure 3F), but flows into the capacitor Cl in the opposite direction to the magnetising current flowing from the capacitor Cl during the magnetising stage. This reversal of current direction may be best appreciated from the change in polarity of the capacitor current shown in the middle waveforms of Figures 3G and 3H.

The flow of the induced current, from the inductive device back to the capacitor, recharges the capacitor to effectively transfer energy that is contained in the magnetic field to the capacitor Cl. This recovered energy is used to re-establish the magnetic field during the magnetising stage of the next cycle of operation.

3.12 Voltage multiplication

On initial start-up, the capacitor Cl is charged, in the energy recovery stages of the first few successive cycles of circuit operation, to progressively higher voltages that are significantly higher than that of the supply voltage Vl. After only a few cycles of operation the capacitor Cl is recharged at each recovery stage to several times the supply voltage. In the magnetising stages, the magnetising current in the inductive device is driven by this capacitor voltage. Figure 31 shows, in the lower waveform, a typical waveform for the voltage on capacitor Cl for a preferred third embodiment having circuit values as discussed below. The capacitor voltage rises to just over 120 volts during the first energy recovery stage, from 7 mS to about 12 mS, and progressively rises to greater voltages in successive subsequent recovery stages to reach about 200 volts after about 200 mS of operation. During the magnetising stages, the magnetising current in the inductive device is driven, in part, by this capacitor voltage. The recovery of energy from the collapsing magnetic field at each cycle and its re-use to reestablish the field in the magnetising stage at the next cycle effectively multiplies the voltage from which the inductive device is driven to provide a significant improvement in efficiency. The voltage multiplication process is similar to the transient charging phase of a resonant inductance-capacitance (L-C) circuit.

During the initial part of the magnetising stage of each successive cycle of the first few start-up cycles, when the circuit adopts the configuration shown in Figure 3C, the capacitor Cl discharges with progressively higher peak current values. This may be seen in the middle waveform of Figure 3G which shows the capacitor first discharging in the second cycle, from 20 mS to about 22.5 mS, with a peak current of about 4.8 A. In the initial part of later cycles the capacitor Cl discharges with a peak current of about 14 A. This may be seen in the middle waveform of Figure 3H which shows the capacitor discharging from 200 mS to about 22.5 mS with a peak current of about 4.8 A.

When the voltage on capacitor Cl is below that of the combined voltage of the dual voltage supplies Vl and V2, and diodes D6 and DlO are reverse biased and diode D3 is forward biased, the circuit effectively adopts the supply-fed magnetising circuit configuration as shown in Figure 3D, whereupon the magnetising current in the inductive device is provided from the series connection of the supplies Vl and V2 via diode D3 to inductive device Ll , with a return path to earth or ground through closed switch S2 and diode D6. This may be seen in Figures 3G and 3H in which the upper waveform shows the supply current. In the first cycle, the supply current begins rising from 0 mS, as seen in Figure 3G. In subsequent cycles, the supply current rises when the capacitor discharge current begins to fall, for example at about 21.8 mS, 42.7 mS, 62.8 mS, and 82.5 mS, as seen in Figure 3G.

As described above, the supply voltage is switched to a lower value at supply switching time t_s3 by the opening of switch S3. The supply voltage, as applied to the anode of diode D3, is shown in the upper waveform of Figure 31 which clearly shows the switching of the supply voltage between higher and lower voltages. The voltage of the series connection of the two supplies Vl and V2, and the voltage of the supply V2 only, although less than the much higher run-mode voltages achieved on the capacitor Cl, are sufficient to maintain the level of current in the inductive device Ll and extend the magnetising period of the inductive device through to the end of the magnetising stage.

3.13 Energy transfer

When the capacitor Cl provides the magnetising current for the inductive device Ll the circuit is effectively capacitor Cl series connected to inductive device Ll, by switches Sl and S2 and diode D6.

For optimum operation of the energy recovery circuit shown in Figure 3A, the recovered energy stored as a charge on capacitor Cl must be efficiently transferred back to the magnetic field associated with the inductive device Ll. Maximum transfer of energy from the capacitor back to the inductive device occurs when the voltage on the capacitor Cl has decreased from a maximum to zero and the current in the inductive device Ll has simultaneously risen from zero to a maximum. The time for this to occur is equal to a quarter of the period of natural resonance of the inductance-capacitance (L-C) circuit, which in this case is equal to 0.5 pi ^"V (Ll Cl).

For optimum operation of the energy recovery circuit of Figure 3A, the switch Sl is closed for each cycle of operation for a time that is not less than 0.5 pi V (Ll Cl) to allow for optimum transfer of energy from the capacitor Cl to the inductive device Ll.

The switch Sl is maintained closed after depletion of the charge on capacitor Cl to extend

^" the duration of the magnetising current in the inductive device Ll. During this extension period, the magnetising current can be supplied from supply V2 alone, or from the combined supplies Vl and V2, to compensate for circuit losses. This extension period may be best seen in Figure 3H in which the supply current, shown in the upper waveform, flows from about 205.3 mS to about 207 mS, and from about 225.2 mS to 227 mS, to extend the magnetising current as seen by the flat top to the inductive device current shown in the lower waveform.

3.14 Preferred embodiment

A first preferred embodiment of the circuit shown in Figure 3A has the following circuit values: Sl, S2 and S3: IRFK20450

Dl, D2, D3, D5, D6 and DlO: RHRG30120

Vl = 95 volts

V2 = 5 volts

Cl = 250 μF

Ll = 36 mH (with an effective series resistance of 0.5 ohms)

Switching period t_λ to t₃ = 20 mS

Switching frequency = 50 Hz

Switch Sl closed period t, to tg, = 5.5 mS

Switch S2 closed period t, to ^₂ = 7.0 mS

Switch S3 closed period t, to t_S3 = 4.0 mS

In this embodiment, for the 20 mS period of each cycle, the switch S3 is closed only over the first 4.0 mS, the switch Sl is closed only over the first 5.5 mS, and the switch S2 is closed only over the first 7.0 mS. Switch Sl is closed for 5.5 mS which is longer than one quarter of the resonant period of the capacitor Cl and inductive device Ll, i.e. 0.5 pi V (Ll Cl), which is equal to 4.7 mS. This allows time (0.8 mS) for the extension of the magnetising current from the combined supplies Vl and V2 to occur. Switch S2 is closed for 7.0 mS to allow time for further extension of the magnetising current from the supply V2 alone to occur, after depletion of the charge on the recovery capacitor Cl.

In this embodiment, in the run mode after the first 10 cycles of operation, i.e. after 200 mS from starting, the capacitor Cl is substantially discharged at each cycle to provide re- magnetising current to the inductive device and is recharged at each recovery stage to a voltage that is more than twice the combined supply voltage of supplies Vl and V2. The voltage on the capacitor Cl is shown in the lower waveform of Figure 31.

As best shown by Figure 3H, the waveform of the current in the inductive device (shown in the lower waveform) has a relatively smooth semi-sinusoidal shape. This current is provided by two discrete periods of discharge current from the recovery capacitor (as seen in the positive portion of the middle waveform), interleaved with two distinct periods of current from the supply (as seen in the upper waveform). The semi-sinusoidal waveform is completed by the current flowing back into the recovery capacitor on collapse of the magnetic field, (as seen in the negative portion of the middle waveform).

The current in the inductive device Ll of this first embodiment of the Figure 3A circuit, in the cycle shown in Figure 3H beginning at 200 mS, is made up of the following five components.

1. A first discharge current from the capacitor Cl (charged in the previous cycle to about 200 volts) begins to rise at the beginning of the cycle at 200 mS when switches Sl, S2 and S3 are closed. This first discharge current continues to rise to a peak at about 203 mS, and then falls to zero between 203 mS to 204 mS as the voltage on the discharging capacitor falls below 100 volts, the combined supply voltage.

2. A first supply current (from the series combination of the two supplies Vl and V2) begins to rise at 203 mS, and continues to rise to a peak until suddenly falling at 204 mS when the supply Vl is disconnected by the opening of the switch S3.

3. A second discharge current from the capacitor Cl rises suddenly at 204 mS when the voltage provided by the supply suddenly drops upon opening of the switch S3. This second discharge current continues until depletion of the capacitor when the voltage on the discharging capacitor falls to 5 volts, the voltage of the supply V2 working alone, at about 205.5 mS.

4. A second supply current (from supply V2 working alone) rises at about 205.5 mS upon the depletion of the capacitor Cl, and continues to flow until the switch S2 is opened at 207 mS.

5. A recovery current, flowing from the inductive device and recharging the capacitor Cl, begins to flow at 207 mS when switch S2 is opened and continues to flow until the current in the inductive device has fallen to zero at about 211.8 mS.

In a second preferred version of the third embodiment, the recovery capacitor is not as fully discharged as in the example described above. In this case, the switch Sl closed period (t, to t_si) equals 3.5 mS and the switch S2 closed period (t, to t_S2) equals 5.0 mS with all other circuit and component values remaining as in the first preferred version of the third embodiment. With these two timing changes, the recover)' capacitor Cl is charged to over 280 volts in the run-mode recovery stages, but only discharges to a voltage of about 120 volts, well above the voltage of the combined supplies Vl and V2, during the run- mode magnetisation stages. This circuit provides useful recovery of magnetic field energy, even with the significant residual voltage remaining on the capacitor after discharge to provide the re-magnetising current for the inductive device. In this case, run-mode current in the inductive device is made up of only three components: current from the discharging recovery capacitor, current from the supply, and current induced in the inductive device by the collapsing field and used to recharge the capacitor. These three components roughly correspond to the components 1, 4 and 5 as described above in relation to the first preferred embodiment of the Figure 3A circuit.

3.15 Waveforms

Figures 3G, 3H and 31 show typical waveforms of currents and voltages for the preferred eighth embodiment of the circuit shown in Figure 3A. The upper waveforms of Figures 3G and 3H show typical supply current waveforms. The middle waveforms of Figures 3G and 3H show typical current waveforms for the recovery capacitor Cl. The lower waveforms of Figures 3G and 3H show typical current waveforms for the inductive device Ll . The upper waveform of Figure 31 shows a typical waveform of the voltage provided by the dual voltage supply as applied to the anode of the diode D3. The lower waveform of Figure 31 shows a typical waveform of the voltage on the recovery capacitor Cl. Figures 3G and 31 show several cycles during start-up. Figure 3H shows run-mode cycles.

It can be seen from the lower waveform of Figure 31 that at start-up the voltage on capacitor Cl is initially zero but then increases rapidly over successive start-up cycles to be approximately 200 volts at 200 mS. The lower waveform of Figure 31 shows that the recovery capacitor Cl is substantially discharged by the end of the magnetising stage for each cycle of circuit operation in the run mode.

In the run-mode of the preferred eighth embodiment of the Figure 3A circuit having the circuit values described above, the magnetising current in the inductive device Ll is similar to a flattened half sinusoid with a flat peak value of approximately 18 amperes, as is seen in the lower waveforms of Figures 3G and 3H. The magnetising current in the inductive device rises over the first part of the magnetising stage, is held almost constant for a short period of about 2 mS, then falls to zero over the beginning of the recovery stage to remain at zero until the start of the next cycle. The slope of the flat peak of the current in the inductive device Ll may be made to rise or fall by appropriate selection of the voltage of the supply V2.

APPLICATIONS

4.1 Transformers

Standard transformer designs do not usually use field energy recovery on the primary winding. In switch mode power supplies, it is common to control the magnetic field in the primary winding but energy is not purposely recovered, stored for feeding back to the power source, or used to effectively multiply the voltage used to drive the winding. Bidirectional drive versions of the magnetic field energy recycling circuits described above can be used to drive transformers with improved efficiencies over prior art transformer drives.

These bi-directional drive circuits can be used to drive a transformer in the position of the inductive device Ll with a full near-sinusoidal AC waveform. It is necessary to control secondary winding load current so that currents in the secondary load circuit, and consequent back EMFs, do not deplete the current during the field energy recovery stage. The secondary load circuit is switched on during the magnetising stages, and is switched off (i.e. open circuited) during recovery stages when the magnetic field in the transformer collapses and the field energy is being recovered to recharge the recovery capacitor Cl. This switching allows greater field build and recovery and much better efficiencies than if the secondary winding is loaded through both the rising and falling quadrants of each half cycle.

The primary and secondary windings can be configured in a 1:1 ratio for a close magnetic coupled circuit, and the primary winding magnetising inductance, the capacitor Cl and the magnetising time period selected to optimise the energy recover}' as described above. For loosely coupled transformer circuits, the primary to secondary winding ratio can be 1 :2 and the load is then switched in during the field energy recovery period.

Reduction of back EMFs is an important requirement in maximising energy recycling in transformers. Reduction of back EMFs requires the following.

• Limitation of secondary load 'induced' back EMFs by controlling the period during which the load is applied. —

• A controlled load circuit allowing for controlled switching of the output to load.

• Use of specialised lamination steels with very low magnetic retentitivity.

• Magnetic coupling between primary and secondary windings controlled by an air gap in the magnetic circuit (in some cases).

These same considerations can be applied to energy recycling circuits used for switch mode or resonant mode converters employing pulse transformers with isolated primary and secondary windings. Multiphase transformer circuits can be driven by compiling a number of bridge circuits.

4.2 Solenoids and linear actuators

All conventional solenoid coils and linear electromagnetic actuators build a magnetic field to perform a work function by magnetic attraction or repulsion. The energy built up or contained within the magnetic field can be substantial. The dissipation of this energy, usually after the work function has been performed, has been seen as a nuisance. Many control strategies have been employed to deplete the field while minimising damage to circuit components from voltage spikes. Eddy current shorting rings, diode clamps, application of reverse voltages and other field control techniques have been employed to safely dissipate the field energy.

The invention provides effective supply voltage multiplication and wave shaping permitting a lower supply voltage to be used to power a solenoid or actuator with greater efficiency. The magnetising current can be broken into multiple pulses, or extended in duration to allow field energy recycling techniques to be used when the frequency of operation is much lower than the optimum repetition frequency of the magnetising and recover₎' cycle for a particular winding. The multiple pulse option is particularly suitable for driving -linear actuators and solenoids. This technique allows pulses at a relatively high repetition frequency, for example 160 Hz, to provide effective energy recycling according to the invention while being interrupted at a relatively low frequency, for example 2 Hz. This allows effective energy recycling with the inductance values typical of these devices.

4.3 Motors

In conventional AC motor drives, a sinusoidal waveform is synthesized by an inverter using a complex switching sequence. While this allows for easy frequency variation, it does not provide for the efficiency improvements provided by recycling (i.e. the recovery and reuse) of the magnetic field energy.

The voltage multiplying and wave shaping functions of the invention, when used in full bridge circuits providing alternating magnetising currents suitable for switched reluctance motors, AC synchronous reluctance motors and AC induction motors, and particularly those motors with low or no motional back EMFs, such as those used in hybrid and electric vehicle drives and similar traction applications where lower supply bus voltages can be used to advantage. As well as driving the motor windings with full sinusoidal drive currents, this circuit also provides good start-up and low speed characteristics that are advantageous in providing substantially more torque during motor start-up. The drive circuit can be timed from a motor shaft sensor or run at a frequency varying on start-up up to a set frequency. The connection of the recovery capacitor to magnetise the winding provides the near sinusoidal current waveform provided that there is little or no motional back EMF present during the field recovery stages, i.e. during the second half of each half cycle of the sinusoidal current. With some motional or induced back EMF, such as in AC squirrel cage induction motors, the sinusoidal waveform may be distorted but the motor will still function well while energy savings are obtained.

Reluctance motors of the switched or synchronous reluctance types are particularly suited to magnetic field energy recovery because, unlike squirrel cage induction motors, they do not create 'motional' back EMFs in the stator winding arising from rotation of the rotor. Although there is a motional change of inductance in the motor winding of some reluctance motors, this does not destroy the field energy recovery and in some cases can aid it. For example, if the inductance increases as the winding current is falling, then this will delay the fall and increase the width of the current pulse in some designs.

OPTIMISATION

5.1 Circuit resistances

For maximum efficiency of the energy recycling circuits described above, it is important that circuit losses, and particularly the decrement of the inductance-capacitance recycling circuit, be kept as low as practicable. The decrement is the lost energy over each cycle that needs to be topped up from the supply. The decrement is a similar parameter to the quality factor (Q) which is die ratio of the maximum energy stored to the energy dissipated per cycle. The decrement of an LCR circuit is the energy dissipated per cycle^' and is denoted by:

E_D = RI²/2f where I = maximum peak value of current

R = circuit resistance

f = frequency.

Alternatively, the loaded Q of the recycling circuits described above is advantageously kept at 2 or above. The circuit resistance includes the resistances of the winding, the controlled switches and the diodes, and the equivalent series resistance (ESR) of the recovery capacitor. For optimum or good energy recover}', it is necessary to keep the total circuit resistances as low as possible. One useful rule of thumb is that any winding group or inductive device should have a resistance of less than one ohm, particularly for lower frequency applications running at 50 to 100 Hz. It is desirable that series circuit comprising the inductive device and the recovery capacitor has a total resistance of less than 1 ohm. 5.2 Number of winding turns

Field energy recovery is aided by keeping the ratio of the number of winding turns (N) to the inductance L high, and by keeping as much of the magnetic flux as possible encompassed within the winding cross-sectional area so that on collapse of the flux, the induced currents and voltages are as high as possible.

It is advantageous if the number of turns is not less than 280, in applications operating at 50 Hz.

Inductance is directly proportional to the number of winding turns squared (N²) and the cross-sectional area (A) of the winding. Therefore, it is important to keep the cross- sectional area as small as possible. For practical purposes compact coils or windings with small mean radii will perform best. In electrical machines, the rotor length to diameter ratio (L/D), controlled by the lamination stack length, is best kept around 1.0 to 1.2.

5.3 Magnetising period

The optimum magnetising period has been described as being substantially equal to 0.5 pi V(LC) where L is the inductance value of the inductive device Ll, C is the capacitance value of the recovery capacitor Cl. Preferably the magnetising period is equal to k pi V(LC), where k is substantially between 0.1 and 2.5, preferably between 0.25 and 1.0, more preferably between 0.35 and 0.70, and most preferably approximately equal to 0.5.

However, circuits with combinations of magnetising period, and inductance and capacitance values not satisfying the k pi V(LC) relationship are useful and still provide energy recycling. For example, in the first and second embodiments, the recovery capacitor can be oversized, i.e. made larger than specified by the above relationship, to give good peak magnetising currents but with some 'peaking' in the sinusoidal wave form. The use of an oversized capacitor is suitable for variable or synchronous reluctance motors which operate with changing inductance.

Optimisation of circuit performance depends on the application and on the desired attributes. For example, copper volume of the winding of the inductive device, purity of sinusoidal waveform of the magnetising current, overall energy efficiency, start-up and/or running torque, must be balanced against each other in a practical application.

Claims

1. A magnetic field energy recycling circuit comprising a capacitor, an inductive device and a switching circuit; wherein:

the switching circuit is configurable in a second configuration, after the magnetic field has been established, to direct a current induced in the inductive device during collapse of the magnetic field to flow into die capacitor in a second direction that is opposite the first direction to thereby charge die capacitor; and

the magnetic field energy recycling circuit is connected to a supply of electrical energy such diat current from the supply flows through die capacitor and the inductive device when the switching circuit is configured in the first configuration and/or the second configuration.

2. A magnetic field energy recycling circuit as claimed in claim 1 wherein, when the switching circuit is in the first configuration, die capacitor is connected in series with the supply such diat the discharge current flows from die supply, through the capacitor, and dirough the inductive device to establish the magnetic field in association with the inductive device.

3. A magnetic field energy recycling circuit as claimed in claim 1 wherein, when the switching circuit is in the second configuration, the capacitor is connected in series with the supply such that die current induced in the inductive device during collapse of the magnetic field flows through the capacitor and through the supply.

4. A magnetic field energy recycling circuit comprising a capacitor, an inductive device and a switching circuit; wherein:

the switching circuit is configurable in a first configuration to direct a discharge current to flow in a first direction from the capacitor and dirough the inductive device to thereby establish a magnetic field in association with the inductive device; the switching circuit is configurable in a second configuration to direct a current to flow from a supply of electrical energy and through the inductive device to maintain the magnetic field; and

5. A magnetic field energy recycling circuit comprising a capacitor, an inductive device and a switching circuit; wherein:

the recycling circuit is connected to a supply of electrical energy such that when the switching circuit is configured in the first configuration and the capacitor is substantially discharged, current flow through the inductive device is maintained by current from the supply.

6. A magnetic field energy recycling circuit as claimed in claim 5 wherein, when the switching circuit is configured in the first configuration, the voltage of the supply of electrical energy is switchable from a first voltage to a second voltage, and the second voltage is lower than the first voltage.