WO2010085854A1  Hybrid adaptive antenna array  Google Patents
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 WO2010085854A1 WO2010085854A1 PCT/AU2010/000091 AU2010000091W WO2010085854A1 WO 2010085854 A1 WO2010085854 A1 WO 2010085854A1 AU 2010000091 W AU2010000091 W AU 2010000091W WO 2010085854 A1 WO2010085854 A1 WO 2010085854A1
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 G—PHYSICS
 G01—MEASURING; TESTING
 G01S—RADIO DIRECTIONFINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCEDETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
 G01S3/00—Directionfinders for determining the direction from which infrasonic, sonic, ultrasonic, or electromagnetic waves, or particle emission, not having a directional significance, are being received
 G01S3/02—Directionfinders for determining the direction from which infrasonic, sonic, ultrasonic, or electromagnetic waves, or particle emission, not having a directional significance, are being received using radio waves
 G01S3/023—Monitoring or calibrating

 H—ELECTRICITY
 H01—BASIC ELECTRIC ELEMENTS
 H01Q—ANTENNAS, i.e. RADIO AERIALS
 H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
 H01Q3/26—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture
 H01Q3/30—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array
 H01Q3/34—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array by electrical means
 H01Q3/36—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array by electrical means with variable phaseshifters
Abstract
Description
HYBRID ADAPTIVE ANTENNA ARRAY Technical Field
The present invention relates generally to antenna design and, in particular, to architectures and methods for adaptive antenna arrays.
Background
Adaptive antennas are important subsystems for long range, mobile and ad hoc wireless communications and sensing networks. The most powerful adaptive antenna architecture is the digital beamforming array, where each antenna element has an associated signal chain to convert the received signal from RF (radio frequency) to baseband, and the baseband signals are processed by a digital beamformer using adaptive filtering techniques. Digital beamforming antenna arrays have the capability of generating many antenna patterns simultaneously, performing high precision beamforming such as nulling, and producing output signals with the maximum signal to noise and interference ratio (SNIR). A further advantage of the digital beamforming array architecture is that on line calibration of different branches can be handled automatically by the digital beamformer. However, purely digital beamforming antenna arrays do have a major disadvantage. Since the cost of processing digital data is proportional to the bandwidth and the computational power required for digital signal processing increases at least linearly with the number of elements, a large digitally beamformed antenna array for wideband operation is simply too costly and impractical for most applications. Another problem with digital beamforming arrays is that, due to the limitation of physical space defined by the array element spacing, a pure digital beamformer is impractical at millimetrewave frequencies beyond approximately 55 GHz. The element spacing required for suppression of grating lobes at scan angles within 60 degrees is limited to 2.9 and 1.7 mm at operating frequencies of 55 and 95 GHz respectively, thus making it extremely difficult to physically place the signal chains behind the antenna elements.
The other principal adaptive array architecture is the analogue beamformer, in which "smart beams" are formed using RF or IF phase shifters on each antenna element in the array. In purely analogue systems, cost is a very weak function of bandwidth. Thus, for wideband adaptive antenna systems with a large number of elements, an analogue adaptive antenna array is much more economical than a purely digital adaptive array. However, an analogue beamformer has certain disadvantages. First, the calibration of a large analogue array is labourintensive and online recalibration is extremely difficult. Second, special means must be employed for beam tracking, thus increasing the complexity and the cost. Third, there is no access to the baseband signals of individual analogue branches because the phaseshifted IF or RF signals are combined before conversion to baseband, so the capacity to form selective beam patterns is limited. These problems are particularly severe for large analogue arrays where the imperfection and ageing of components can cause serious degradation in antenna performance.
Summary
Disclosed are arrangements which seek to ameliorate the above disadvantages by means of a hybrid adaptive antenna array, which comprises an array of antenna elements partitioned into multiple analogue beamforming subarrays and a digital beamformer on the subarray outputs, with a control path back to the analogue phase shifters. This reduces the size of the digital beamformer by a factor equal to the average number of elements in each subarray. Also disclosed are calibration methods for both the analogue and the digital beamformers in the hybrid adaptive array. In addition, disclosed are search and tracking methods for the hybrid adaptive array. The disclosed hybrid array provides an efficient means of producing large, highgain adaptive antenna arrays at relatively low cost.
According to a first aspect of the present disclosure, there is provided a hybrid antenna array comprising: (a) a plurality of digital branches, each digital branch including:
(i) an analogue beamforming subarray, each subarray having a plurality of antenna elements, a phase shifter adapted to apply a phase shift to the signal from each antenna element, and a combiner adapted to combine the phaseshifted signals;
(ii) a signal chain adapted to convert the output of the subarray to baseband; and
(b) a digital processing module, including:
(i) an angle of arrival estimation submodule adapted to estimate an angle of arrival of a signal at the antenna elements;
(ii) a phase control submodule adapted to control the phase shift applied by each phase shifter depending on the estimated angle of arrival; and
(iii) a digital beamformer adapted to combine the baseband signals from the digital branches using a weight vector to form an output signal.
According to a second aspect of the present disclosure, there is provided a method of estimating the angle of arrival of a signal at an antenna array comprising a plurality of digital branches and a digital beamformer, the method comprising:
(a) crosscorrelating the baseband signals from adjacent digital branches;
(b) for a plurality of candidate beams,
(i) setting the phase shifts using the arguments of the cross correlation values; and (ii) computing the power of the output signal of the digital beamformer;
(c) determining the candidate beam giving the maximum output power; and
(d) estimating the angle of arrival from the crosscorrelation values of the determined candidate beam.
According to a third aspect of the present disclosure, there is provided a method of calibrating an antenna array comprising a plurality of digital branches and a digital beamformer, the method comprising:
(a) estimating the expected phase difference between the baseband signals from adjacent digital branches;
(b) determining the phase deviation of each digital branch as the difference between the expected phase difference and an actual phase difference between the baseband signals from a digital branch and an adjacent previously calibrated digital branch; and (c) applying each phase deviation to the baseband signal from the corresponding digital branch.
According to a fourth aspect of the present disclosure, there is provided a method of calibrating an antenna array comprising a plurality of analogue branches, each analogue branch comprising an antenna element and a phase shifter, the method comprising: (a) optimising the phase shifts applied by the phase shifters to an input signal to the array so as to obtain a maximumpower output signal from the array;
(b) determining expected values of the phase shifts using an angle of arrival of the input signal; (c) determining the phase deviation for each analogue branch in the array as the difference between the expected phase shift and the optimised phase shift for the analogue branch; and
(d) applying each phase deviation to adjust the phase shift applied by the corresponding phase shifter.
Brief Description of the Drawings
In the drawings:
Fig. 1 shows a receiving hybrid antenna array according to one embodiment of the present invention;
Fig. 2a shows an exemplary linear configuration of the hybrid array of Fig. 1 ; Fig. 2b shows an exemplary planar configuration of the hybrid array of Fig. 1 ; and Fig. 3 is a flowchart illustrating a method of estimating the angle of arrival carried out by the hybrid antenna array of Fig. 1.
Detailed Description
Where reference is made in any one or more of the accompanying drawings to steps and/or features, which have the same reference numerals, those steps and/or features have for the purposes of this description the same function(s) or operation(s), unless the contrary intention appears.
Fig. 1 shows a receiving hybrid antenna array 100 according to one embodiment. The antenna elements, e.g. 120, in the array 100 are grouped into M analogue beamforming subarrays 1101, 1102,..., HOM. A hybrid array is said to have a planar configuration (a planar array) if all the sub arrays are located in a single plane. A linear configuration (a linear array) is a special case of a planar configuration in which all the subarrays are located in a single line.
Fig. 2a shows an exemplary linear configuration 200 of the hybrid antenna array 100 of Fig. 1. The exemplary linear array 200 comprises two subarrays 2101 and 2102 arranged sidebyside, each subarray 210m comprising four evenlyspaced elements, e.g. 220, arranged in a straight line. Fig. 2b shows an exemplary planar configuration 230 of the hybrid antenna array 100 of Fig. 1. The exemplary planar array 230 comprises sub arrays 2401, 2402, 2403, and 2404 laid out in a square, each subarray 240m comprising a fourbyfour square grid of elements, e.g. 250. The hatching of each element 250 indicates the subarray to which it belongs.
Each subarray 110m in Fig. 1 is illustrated with 4 elements 120, which is the preferred subarray size for a linear array, while the preferred subarray size for a planar array is 4 by 4 elements. In general, different subarrays may have different numbers of elements.
A planar array is defined as uniformly spaced if the number M of subarrays can expressed as M = M_{x} x M , where M_{x} and M_{y} are the numbers of subarrays arranged along the xaxis and the yaxis respectively. The Mh element in subarray m of a uniformly spaced planar array is located at coordinates (X_{1} ,_{m} Y_{hm}) such that
Y_{hm} = Y_{l +} m_{y}d;^{{} \ m_{y} = 0XM_{y}  l
where m = m_{y}M_{x} + m_{x} , d_{x} ^{s}^ and d^ are the horizontal and vertical spacings of the sub arrays respectively, and (X_{1}, Y_{1}) is the location of the Mh element of the subarray numbered m = 0. For example, the planar array 230 is uniformly spaced with M_{x} = M_{y} = 2 and d_{x} ^{s)} = d^ .
A uniformly spaced linear array is a special case of a uniformly spaced planar array with M_{y} = 1, so M_{x} = M, m_{y} = 0, and F,_{m} = Y_{1} (d_{y} ^{s}^ is irrelevant). For example, the linear array 200 is uniformly spaced with M_{x} = 2 and M_{y} = 1.
A rectangular subarray has N = N_{χ} χ N_{y} elements, where N_{x} and TV,, are the numbers of elements arranged along the xaxis and the yaxis respectively. The location (X_{1}, Y_{1}) of the zth element of the rectangular subarray numbered m = 0 is given by
Y, = Y_{0} +i_{y}d_{y} ^{e)} i_{y} = 0,\,N_{y} \where i = i_{y}N_{x} + i_{x} , d_{y} and d_{x} are the element spacings along the yaxis and xaxis respectively, and (Xo, Yo) is the location of the element numbered 0. For example, each subarray 240m in the planar array 230 is rectangular, with N_{x} = Ny = A and d[^{e}' = dy .
A linear subarray is a special case of a rectangular subarray with N_{y} = 1 , so N_{x} = N, i_{y} = 0, and Y_{1} = Yo for all i ( d^ is irrelevant). For example, each subarray 210m in the linear array 200 is linear, with N_{x} = 4.
The planar array 230 is a uniformly spaced planar array comprising rectangular subarrays with d^ = N^^ and d^ = N_{y}d_{y} ^{e}K The linear array 200 is also uniformly
spaced, comprising linear subarrays with d^ = N_{x}dy . Each antenna element 120 belongs to an analogue signal branch, e.g. 115, which also includes a phase shifter, e.g. 130, which is independently controllable by an external signal 175, as described below. The phase shifter 130 alters the phase of the received signal from the antenna element 120 by a controllable amount. Each analogue branch 115 may also contain a low noise amplifier, an attenuator, a filter, and a frequency down converter, using a local oscillator, to intermediate frequency (not shown). The phase shifter 130 is preferably implemented as a controllable or 'switched' delay in the radio frequency (RF) signal, the downconverted intermediate frequency (IF) signal, or the local oscillator.
The outputs of each analogue branch 115 in the subarray 110m are combined by a combiner 135m. The phases of each phase shifter 130 are controlled as described below such that each subarray 110m generates a beam in a desired direction, for example the angle of arrival of the RF signal incident on the hybrid array 100. The combination of RF signals from different analogue branches by the combiner 135m is preferably implemented by a conventional corporate network.
In alternative embodiments, the analogue beamforming in a subarray 110m is performed by a Rotman lens or a Butler matrix rather than a phased array as described above. The output of each subarray 110m is passed to a chain 140m, which converts the RF or IF signal to baseband. The combination, e.g. 145, of a subarray 110m and a chain 140m is referred to herein as a "digital branch". The M baseband signals from the digital branches 145 are processed in a digital processing module 150 comprising several submodules: a digital beamformer 165, a calibration submodule 160, an angle of arrival estimation submodule 155, and a subarray phase control submodule 170. The operation of each submodule will be described below.
The digital processing module 150 serves several purposes. First, the digital beamformer 165 ensures that the output signal 180 of the hybrid array 100 has a highest signal to noise and interference ratio (SNIR). Second, the digital processing module 150, through the subarray phase control submodule 170, provides the control signal 175 for each phase shifter 130 in each subarray 110m. The angle of arrival estimation sub module 155 plays a role in this function, as described below. Third, the digital processing module 150, through the calibration submodule 160, provides calibration both for the digital branches 145 and for each analogue branch 115 in the hybrid array 100 as described below. The preferred implementations for the digital signal processing module 150 are embedded firmware on Field Programmable Gate Arrays (FPGAs) or a dedicated ASIC, which is more suitable for mass production.
The digital beamformer 165 generates an output signal z(n) 180 at time n by applying weights w(«) = [w_{Q} (n) w, (n) ■ ■ ■ w_{M}__{γ} (n)]^{τ} to the (complexvalued) baseband signal vector x(n) = [x_{o}(n) x, (n) • ■ ■ x_{M}__{x} (n)^{~}\^{T} output by the digital branches 145 as follows:
Φ) = w"(«)φ) (1) where indicates the conjugate transpose of a vector. Forming the digital beam The weights w(«) vary or adapt with time to cope with changing signal and / or system characteristics. There are a number of adaptive algorithms such as the least mean squares (LMS) algorithm and the recursive least squares (RLS) algorithm which can be employed to update the weight vector in a conventional digital beamformer. The method used by the digital beamformer 165 according to one embodiment uses a reference signal s{ή) that was known to have been transmitted to arrive at time n. This knowledge is available during a training period for the hybrid array 100. The reference signal could be the header portion of a data packet. According to this embodiment, the digital beamformer 165 iteratively adjusts the weight vector as follows: w(n) = αw(n l) + (l α>' (n)φ) (2) where α is a "forgetting factor" that controls the effective window size of the time averaging, and the * superscript indicates complex conjugation.
In an alternative embodiment, to reduce the sensitivity of the digital beamformer 165 to the magnitude of the input signal vector x(n), the input signal vector x(n) is normalized, resulting in the following iterative adjustment equation:
y_{V}(n) = aw(n \) + (\  a)s\n) ^{x(w)} (3)
H x(O II
For slow moving targets, the weights w can be fixed after the training period until the next training period (periodic adaptation). For fast moving target, after the training period, the output signal z(n) can be used as the reference signal s(ή) (known as decision aided beamforming).
Estimating the angle of arrival from the weights
If the subarrays 110m in the hybrid array 100 are uniformly spaced in a linear configuration (e.g. 200), the weights will converge to the following form: w_{m}(n) = exp(jkmd_{x} ^{s)} sin θ) + v_{m} («) (4) where k is the wavenumber of the RF carrier signal, θ is the "zenith" angle of arrival as illustrated in Fig. 2a, and v(n) is a noise vector.
If the subarrays 110m in the hybrid array 100 are uniformly spaced in a planar configuration (e.g. 230), the weights will converge to the following form: w_{m} (n) = exp[jk\m_{x}d_{x} ^{s)} sin θcosφ + m_{y}d_{y} ^{s)} sin #sin φ))+ v_{m} (n) (5) where φ is the "azimuth" angle of arrival as illustrated in Fig. 2b. Note that for a planar array, the "zenith" angle of arrival θ is measured from the normal to the plane of the array. Once the digital beamformer 165 converges, the angle of arrival estimation sub module 155 analyses the weights w(n) according to the model in equation (4) or equation
(5) to form an estimate ψ,φ) of the angle of arrival, given values for k, J^' and d^ .
Search strategies If the angle of arrival is not initially known, the phase shifters 130 cannot be set to the correct values to give each subarray 110/n anything like its maximum gain. The digital beamformer 165 may therefore be unable to form a reasonable beam, and thus the angle of arrival cannot be estimated by the submodule 155 by the method described above. To form a coarse estimate of the angle of arrival, and thereby initialise the hybrid array 100, the digital processing module 150 takes one of several approaches (known as the "search strategy"). One search strategy is to divide the whole hybrid array 100 into multiple groups. The subarray phase control submodule 170 sets the phase shifters 130 in the analogue subarrays 110m using equation (19) or (20) below, such that each group monitors a different "look angle" spanning the field of view. The digital processing module 150 operates on the outputs from each separate group in sequence, keeping the phase shifters 130 fixed. The "look angle" monitored by the group that results in the highestpower output signal z(n) is a coarse estimate of the angle of arrival. The reduced digital beamforming gain caused by the use of fewer than M subarrays at a time can be compensated for by a longer convergence time in the beamforming module 165.
Another search strategy is for the digital processing module 150 to perform a coarse digital beamforming using only one element 120 in each analogue beamforming subarray 1 \0m, making the phase of each element irrelevant. An equivalent approach is to set the initial phases of each analogue subarray such that the beam pattern of the sub array has low directivity (i.e. is close to omnidirectional). A coarse estimate of the angle of arrival may be then obtained using the method described above.
A third search strategy is for the subarray phase control submodule 170 to set all the phase shifters 130 in all the subarrays 110m so as to monitor a particular "look angle" using equation (19) or (20) below. The digital processing module 150 then estimates the angle of arrival using either of the methods described herein, and also records the corresponding output signal power. This process is repeated while the "look angle" is stepped through multiple look angles that cover the field of view. At the end of the search, the coarse estimate of the angle of arrival is the angle estimate corresponding to the highest output signal power.
All the search strategies are executed by the subarray phase control submodule 170 in collaboration with the beamforming module 165.
Estimating the angle of arrival by Correlation An alternative method of estimating the angle of arrival is for the estimation sub module 155 to crosscorrelate the signals x_{m}(n) from adjacent digital branches 145 along the xaxis and the yaxis:
R_{x} = E^_{m>Mχ+m};{n)x_{myMχ+m}^ {n)) (6)
Definingu_{x} = — d_{x} ^{{s)} sin θ cos φ (8)
(9)
where λ_{c} is the wavelength of the RF carrier signal, then provided d^and d^ are lessthan or equal to λ_{c} / 2, u_{x} and u_{y} will be in the range (π, Tt) and are equal to arg{./?_{x} } and arg/?_{r} ) respectively without any ambiguity. The angle of arrival could then be estimated
directly from u_{x} and u_{y}, given values for λ_{c}, d^ and d^ .
However, for a uniformly spaced planar configuration, J^ and d^ are typically greater than λ_{c} / 2, so u_{x} and u_{y} can be outside the range {π, π) and phase ambiguity will be present in R_{x} and R_{y}.
To handle this ambiguity, the estimation submodule 155 carries out the method 300 illustrated in the flow chart in Fig. 3 to estimate the angle of arrival. In step 310, the estimation submodule 155 initialises two indices p and q for a double loop over the following ranges:
P =
In step 320, the estimation submodule 155 estimates the crosscorrelations iteratively over time n using all available subarray outputs (to maximise the signalto noise ratio) as follows:
*{"> = (1  μ)Rr^{{ λ)} + M ∑ ∑ *«_*,♦«>)*(.,♦,>,._{*}., W (11) m_{x}=0m =0
where 0 < μ < 1 is an updating coefficient (typically set to 0.001). The estimation submodule 155 then at step 330 computes u_{x} and u_{y} values from
at the current/? and q respectively, as follows: u_{x}{p) = 2ψ + srg{R_{x}}, (12) where [.] denotes the operation of taking the integer part of a value.Each pair (u_{x}(p),u_{y}(q)) represents a candidate beam. The phase control sub module 170 at step 340 sets the phase shifts of the phase shifters 130 within each subarray for the current (p, q) pair to
The beamforming submodule 165 then at step 350 forms the beam as described above to produce the output signal z_{PA}(ti), after which (at step 360) the estimation sub module 155 computes and records the output signal power for the current (p, q) pair iteratively as where 0 < β < 1 is an updating coefficient (typically set to 0.25).The estimation submodule 155 then at step 370 determines whether the double loop over p and q is complete; if not, p and q are updated at step 375 and the method 300 returns to step 320.
If the double loop is complete, the estimation submodule 155 at step 380 determines the (p, q) pair that yielded the largest output signal power computed at step 360:
Finally, at step 390 the estimation submodule 155 uses the maximising pair (p_{max}, qmax) to estimate the angle of arrival:Due to the delay from the time when the phase shifts are loaded into the phase shifters 130 in subarrays 110m to the time when a change of the beamformed output signal is observed, several (typically 4) iterations per (p, q) pair are preferable for step 360. If the number of iterations for each (p, q) pair is chosen so that the output power P^{{n)}(p,q)of the corresponding beam can be calculated with sufficient accuracy, one pass through the complete double loop will be sufficient to determine the most likely beam and hence estimate the angle of arrival. If fewer than that number of iterations are used, the double loop is repeated until the power of each beam calculated across multiple double loops is obtained with sufficient accuracy. Setting the phase of the analogue branches The subarray phase control submodule 170 uses the estimated angle of arrival
[θ,φj from the estimation submodule 155 to set the phase of each phase shifter 130 in each analogue branch 115 via the control signals 175 as follows:
Ct_{1} = k(x, sin θ cos φ + Y_{1} sin θ sin φ) (19) for the planar array, or α, = kX_{t} smθ (20) for the linear array. In tracking mode, the subarray phase control submodule 170 periodically obtains an estimate of u_{x}(p_{mΑX}) and u_{y}(q_{max}) from the estimation module 155 (which uses steps 320 and 330 of the method 300 with {p, q) set to (p_{max}, gw)), and updates the phase of each phase shifter 130 using equation (14), as in step 340 of the method 300.
Calibrating the array
Equations (4) and (5) hold if the subarrays 110m are uniformly spaced and the signal chains 140w are identical. In practice, there will be small deviations from uniform spacing, and imperfections due to component ununiformity, temperature change, and component ageing in the signal chains 140m, which cause the weight vector to converge to values whose phase deviates slightly from the models in Equations (4) and (5). If these deviations are not corrected by calibration, the estimation of angle of arrival may be affected and the performance of the hybrid array 100 may be suboptimal.
If the subarrays 110m are uniformly spaced, the phases between the signals output from any two adjacent digital branches 145 should have the same difference regardless of the phase deviation. This fact can be used to calibrate the digital branches. The calibration submodule 160 analyses the signals x_{m}{n) from each digital branch 145 to determine the phase deviation of each digital branch 145. To do this, the calibration sub module 160 uses the method 300 to obtain an estimate of the "expected" phase difference, defined by equations (8) and (9), between any two adjacent digital branches 145. This estimate is then used to determine the phase deviation in each digital branch 145 as the difference between the expected phase difference and the actual phase difference between that digital branch 145 and an adjacent, previously calibrated digital branch. The calibration proceeds by induction in each direction from a designated reference digital branch. The phase deviation is used by the beamforming submodule 165 to correct the signals x,(n) before forming the beam.
The calibration submodule 160 also calibrates each analogue branch 115. According to one embodiment, the subarray phase control submodule 170 optimises the phase shifts applied by the phase shifters 130 in a subarray 110m in order to obtain the maximumpower output signal x_{m}(n) from the corresponding digital branch 145. Since the angle of arrival is known from the angle of arrival estimation submodule 155, the expected phase shift values, accurate to the last bit of the phase shifters, can be calculated from equations (19) and (20). The difference between the expected phase shift values and the optimised phase shift values gives the phase deviation for each analogue signal branch. The deviation is applied by the phase control submodule 170 to adjust the phase shifts of the phase shifters 130 in the subarray. Calibration of analogue signal branches is preferably performed a few times a day.
The arrangements described are applicable to the data communication and sensing industries, and particularly to satellite communication and other high data rate communication (mobile, pointtopoint, and pointtomultipoint topologies). The disclosed arrangements are most beneficial for millimetrewave to terahertz adaptive antenna arrays where tight element spacing makes it very difficult or impossible to do conventional digital or analogue beamforming alone. The typical operating range is 10 to 100 GHz, but the arrangements are useful for frequencies as low as 400 MHz.
The foregoing describes only some embodiments of the present invention, and modifications and/or changes can be made thereto without departing from the scope and spirit of the invention, the embodiments being illustrative and not restrictive.
Claims
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EP2392048A4 (en)  20150819  application 
US8754810B2 (en)  20140617  grant 
EP2392048A1 (en)  20111207  application 
US20120033761A1 (en)  20120209  application 
CN102365789A (en)  20120229  application 
CN102365789B (en)  20140611  grant 
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