WO2010061080A1

WO2010061080A1 - Class of dual mode resonators made from a multilayer stack of organic laminates improving the performance and compactness of integrated passive components

Info

Publication number: WO2010061080A1
Application number: PCT/FR2009/001350
Authority: WO
Inventors: Arnaud Sauvaget
Original assignee: Arnaud Sauvaget
Priority date: 2008-11-28
Filing date: 2009-11-27
Publication date: 2010-06-03
Also published as: FR2939270A1; FR2939270B1

Abstract

The invention relates to the design, the topology and the manufacturing of integrated passive components in multilayer organic laminates that can be mounted on printed circuits for processing radiofrequency signals, and that clearly differ from the prior art (LTCC, inter alia) by their advantages in terms of performance, size, yield and reliability. Thanks to the present invention, balun filters, filters and multiplexers can be made according to standard methods in a laminated stripline structure that comprises 3 dielectric layers and 4 metal layers in which the dual mode resonators are made in the form of 2 interdigited transmission lines printed on 2 metal layers provided on either side of a central organic dielectric substrate enabling a strong transverse coupling. Therefore, due to the very strong coupling between the transmission lines, the resonators provide 2 resonance modes well spaced apart and the filter architectures resulting therefrom can be driven in a differential or asymmetrical manner without any space loss or performance degradation. The filters can be designed without vias or isolated members with impedance adaptations implemented by the received lines and with inter-resonator couplings based on electromagnetic field interactions, which improves their compactness.

Description

Title of the Invention: A class of dual-mode resonators constructed from a multilayer stack of organic laminates improving the performance and compactness of integrated passive components.

TECHNICAL FIELD OF THE INVENTION The present invention relates to the design, topology and manufacture of passive filters. It reveals a class of dual-mode resonators built from a multilayer stack of organic laminates improving the performance and compactness of integrated passive components.

//) State of the Art There is currently a growing need in the industry for multi-mode and multi-band radio communication devices. Standards in this area are proliferating rapidly, from applications using frequencies below GHz, such as digital video broadcasting (DVB), cellular telephone networks (GSM, WCDMA) and transmissions. OFDM-based data such as WLAN 802.11, WiMAX, 802.106 or UWB with frequencies up to 10 GHz or higher. The end user expects to have GPS (Global Positioning System) to find his way, Bluetooth ^® to provide short-distance wireless communication between different devices that are earphones and hands-free kits of mobile phones , PDA (Personal Digital Assistant) diaries, laptops or other high-tech equipment. While the active components needed for signal switching and amplification have been seamlessly integrated into solid-state substrates, passive components, essential for removing interference and spurious emissions, continue to face increasingly demanding specifications. in terms of cost, congestion, consumption and radio performance (attenuation and insertion losses). Coaxial cavities and monobloc filters are not satisfactory from a cost and dimension point of view, while the height of the components must systematically be less than 1.5 mm and the manufacturing processes must allow reliable, low cost and reliable production. large volumes. Filters based on multilayer ceramic technologies and Low-Temperature Cofired Ceramic (LTCC), to obtain a smaller footprint, are currently widely integrated in telecommunications equipment (voice and data) while they necessarily entail higher costs. because of the higher temperatures in the manufacturing processes, the cost of the raw material and the possible low economies of scale. Instability, deformation and mechanical shrinkage are other disadvantages leading to failures during assembly or during impact tests. Multilayer organic technology is booming as it seems capable of meeting the industry's current expectations for cost savings through cheap raw material and significant economies of scale (24 "x 18 wide panels). ") and the possible use of standard printed circuit processes.

The prior art of recent patents filed on filters, baluns, balun filter, multiplexers using multilayer organic substrates is centered around the inductances. Inductors and their parasitic capacitances create resonators in filters and duplexers. Inductances across all possible forms of capacitance integration (embedded parallel plates, surface mounted components) form lumped localized filter architectures with localized impedance matching and coupling. They also form localized baluns (classical LC or with high-pass and low-pass branches out of phase by 180 °). Concerning the state of the art on the topologies of RF passive components in organic laminate substrates and based on transmission line resonators such as capacitive half-wave couplings, distributed half-wave, comb-line coupling , pseudo-combline, interdigital or pseudo-interdigitated, hairpin hairpin, U or ladder-line, the work of Hong and Lancaster in Microstrip Filters for RF / Microwave Applications details examples using RT / Duroid ^® , an organic material. Now, these resonators and filters all have a planar geometry, the distributed elements of the resonators are created by patterns on a single metal layer and very often the structures are in microstrip with a single ground plane on the underside. As a result, they suffer from the low dielectric constant of an organic laminate with low losses and thus lead to long lines and bulky filters. There is a trade-off between the dielectric constant and the dielectric losses in the organic laminate. By doping the laminate with ferromagnetic or other powders, to increase the dielectric constant and reduce the size of the filters, the quality factor of the resonators is degraded and thus, consequently, the insertion losses of the filter.

Resonators based on inductors inherently have a lower quality factor than resonators and capacitors of the same substrate. This is due to parasitic coupling between the branches of the inductor or between the branches and the mass, the laces and fittings creating higher modes and resistive losses by the fact of using thin lines to save space. The Multilayer organic laminate technology, the price of which may be one-tenth that of a ceramic planar filter, is therefore meeting the low cost requirements. State-of-the-art RF architectures and components using LCP, PFE, PTFE and other low loss materials are beginning to compete with ceramic multilayer solutions in terms of loss and size performance. By overcoming the need to base the filter resonators on inductors or transmission lines, the topology of filters, balun filters or multiplexers can result in components that occupy even less space and show even lower losses. .

US Patent References:

US 6,987,307 US 6,900,708 US 7,068,124

European patent references:

EP 1 731 006

Other publications:

Hong J.S. and Lancaster M.J., Microstrip Filters for RF / Microwave Applications, Wiley-Interscience, 2001

Mongia R., Bahl I, and Bhartia P., "RF and Microwave Coupled-Line Circuits", Artech House, 1999

Awai L, "Artificial Dielectric Resonators for Miniaturized Filters", IEEE Microwave Magazine, October 2008

Wada, K. et al., ^U A Compact Mutilayered Balanced filter using Resonators basedon

Interdigitally-Coupled Lines ", Proceedings of Asia-Pacific Microwave Conference, 2006

Awai I. et al., Broadside-Coupled Interdigital Resonators, IEICE General Meeting, 2005

Levy R., "Synthesis of Inhomogeneous Distributed Networks", European Microwave Conference, 1984

Allen JL, "Inhomogeneous Coupled-Line Filters with Large Mode-Velocitv Ratios", IEEE Transactions on Micro Wave Theory and Techniques, 1974 - A - III) Presentation of the invention

The present invention features passive components embedded in multilayer organic laminates that can be mounted on printed circuit boards and can handle RF signals with less space and lower losses than any other planar multilayer filter technology. .

The present invention also includes methods and topologies for making these devices that can be literally integrated into multilayer cards (MiniCard, HaIf MiniCard, System-In-Package formats) because both use organic technology, or individually encapsulated in form. surface mounted components (CMS, BGA, CSP) to be mounted on multilayer boards.

In applying this invention, the baluns, filters, and multiplexers are manufactured in a laminated structure of 3 dielectric layers and 4 stripline-type metal layers, in which the resonators are made by 2 transmission lines printed on 2 metal layers separated and arranged on either side of a central dielectric organic substrate and in which the impedance adaptations and the couplings are not made via localized components but by a direct and electromagnetic coupling of the sensed lines. If the desired coupling and impedance matching can not yet be achieved, the inter-resonator capture can be done at any point between the open branch and the shunt branch of the resonator by very short lines, localized capacitances in plates parallel / interdigitated sections of coupled lines (these shunted line sections have their end left open or connected to ground).

In accordance with one aspect of the invention, the dielectric material used to separate the resonator from the lower and upper mass layers of the stripline structure is potentially any low loss, low dielectric constant and low cost organic material such as polytetrafluoroethylene (PTFE) composites, epoxy resins, Liquid Crystal Polymer (LCP) or the like. The dielectric material used to make the balun resonator filters, multiplexers and filters must be a fine material, with high dielectric constant such as MC25ST ^® C-ply ^®, ^® hkl 1, found in 8, 12 or 14 microns thick with dielectric constants of 10 or more. A strong interdigital cross coupling of the 2 lines associated with reduced interaction and low losses with the upper and lower ground planes are essential to create small low loss resonators. The resonators obtained are made by two lines coupled with an incredibly tight coupling which thus makes it possible to obtain two distinct modes of resonance. The stronger the coupling, the greater the frequency separation between the two modes.

Whether it is a ceramic or organic stack, the coupling in a homogeneous medium is not enough and the two modes are too close to each other. In asymmetric (single-ended) architectures, the weak coupling between the two modes indicates that the filter suffers from a parasitic resonance close to the bandwidth: this rules out the use of 2 interdigitated coupled lines as a resonator in the asymmetrical filters at unless the coupling is reinforced using a non-homogeneous stack.

In differential architectures, the highest order mode of the resonator is the common mode and the lowest order mode is the differential mode. This property allows us to transform any asymmetric filter into a balun filter without penalizing performance or clutter. No other filter architecture known today can achieve this.

A weak coupling excludes the use of 2 interdigitally coupled lines as a resonator in the multiplexers where the common mode of a frequency band may interfere with another considered band, unless the coupling is made very strong using a non-stack. -homogeneous.

The coupling is made very important by using a thin organic laminate with a high dielectric constant; ceramic powders, ferroelectric, para-electrical or other materials associated with polyimides, PTFE composites, or epoxy resins can be used to increase the dielectric constant. The organic material thus compound must be available in a thickness of at most 25 microns to ensure strong coupling.

For the first resonance mode (low frequency), the resonator belongs to an organic and artificial environment composed of strip lines (stripline) whose effective dielectric constant is higher than all the other dielectric constants of the stack. Awai in Artificial Dielectric Resonators for Miniaturized Filters presents arrangements between metallic and dielectric layers that offer important permittivities and allow the miniaturization of resonators; the strong coupling in a disk resonator of 2 dielectric layers shows a wider separation between the resonance modes.

Transversally coupled interdigitated lines have been used in LTCC (see Wada et al., In A Compact Multilavered Balanced Filter Factory Resonators based on Interdizitallv-Coupled Lines), but weak coupling allows only applications restricted to differential filters (asymmetric filters and multiplexers being impacted by the limitations described above). There are many aspects of LTCC technology that would prevent a strong coupling suitable for filter applications in the event that someone would want to transfer the concept of this invention to LTCC or ceramic environments.

On the one hand, the accuracy of screen printing screens and "sheet-by-sheet" alignment tolerances will lead to a very poor performance on the central frequency of the resonators. On the other hand, we would need to coexist several materials with different dielectric constants to minimize the coupling with the mass and have a non-homogeneous stack; this is delicate in a ceramic environment because of the varied thermal behavior (expansion) of the different materials leading to reliability problems. If such disadvantages are solved in the future, the costs, manufacturing processes and economies of scale of the ceramic multilayer products will always be in trouble in the face of price reductions demanded by the market.

The transversally coupled lines in a non-homogeneous medium have been studied by many people who have determined their characteristic impedances, their effective permittivities and the propagation constants (the plural is used because of the existence of two modes). Levy in

Svnthesis of Inhomogeneous Disturbed Networks summarizes the equivalent circuits of different arrangements; the interdigitated coupled section is equivalent to two short-circuited lines of the same measurement coupled by two parallel transmission lines having different impedances and electrical lengths and dependent on the two resonance modes.

The filter architectures we can clearly draw from are broadband filters with long quarter-wave resonators, whose coupling is close to critical coupling. That is, the two modes created are very close to one another to form a filter, whereas the filters based on bi-mode resonators disclosed in this invention are small because the mode of low frequency operation of the resonator is at a frequency well below the frequency natural resonance of a single quarter-wave transmission line and the 2 transmission lines are clearly over-coupled, allowing a very wide separation between modes.

Allen in Inhomoeeneous Coupled-Line Filters with Large Mode-Velocity Ratios exploited the case where a digital section leads to two modes, whose propagation constants are very different. These differences are exploited to create elliptic filters with wide bandwidth, well-defined transmission and reflection zeros. But just like the publication mentioned above, the structure composed of 2 interdigitated lines with transverse coupling is not used as band-pass or band-pass resonator, differential or asymmetric, in series or shunt, allowed by a very strong separation between resonance modes.

Thanks to advances in direct laser imaging and automatic optical inspection (AOI), the two interdigitated lines transversely coupled in an organic laminate have a tolerance of misalignment of the order of a few microns which guarantees the repeatability of the results.

The laminated components used in the stack have very similar thermal expansion coefficients and are compatible with the FR4 manufacturing processes for reliable and low cost finished products.

Thus, the dual-mode resonator with transversely coupled interdigitated lines based on an organic stratified structure clearly differs from the existing by its strengths of performance, size, performance and reliability.

In multi-layer laminate (MCM-L) and ceramic (MCM-C) technologies, below 20GHz, the losses of printed resonators are mainly conductive.

Through the use of a common, thick, high-conductivity metal and current etching processes, conductive losses are minimized providing important quality factors to empty dual-mode resonators.

A thick copper, with a RMS roughness much lower than the skin thickness, associated with the latest PWB screen printing / etching processes, allows to obtain lower conductive losses than using the printing and drying of the dough metallic. The dual mode resonator has a first low resonance mode which is the differential mode; Filter architectures can be differentially or asymmetrically controlled without loss of space or performance changes due to the very strong coupling between the transmission lines. In MCM-C filters with lower coupling, differential working results in a loss of 0.5dB. The resonator obtained can also be used in its second mode of work, the common mode, whose frequency is much higher. The reasons that may lead us to prefer this solution are: to design a filter in common mode or to improve the tolerances of a filter whose design would imply very small resonator geometries.

The final encapsulation of the filter with the ground planes at the upper and lower ends of the stripline structure provides shielding against interference and radiation. Side shielding, by coplanar partition, integrated or external, can also be implemented.

The filters constructed from the present invention, have patterns drawn on 2 layers located on either side of an organic dielectric layer, a top and bottom shielding and optionally, if necessary, additional layers for routing or report the components if they are embedded in an RF module or SiP (System in Package). Such a simple 4-layer metal architecture fabricated through printed circuit manufacturing processes costs less than $ 0.30 per square inch (sq. In.) On 24 "xl8" panels: only a fraction of the price of RF ceramic components .

For example, let us take a 3-pole filter built by the present invention, which is supposed to reject 25 dB at cell frequencies up to 1.98 GHz and 20 dB on parasitic emissions at 3.2 GHz, the passband of the filter is between 2.3 and 2, 7GHz. A generalized Chebichev filter constructed from two-mode resonators in an artificial, non-homogeneous, organic stack of 4 layers using only electromagnetic coupling, appears very simple and has a footprint of only 1 x 1.3mm.

The filter has been designed symmetrically and can be controlled in differential mode if necessary and without any noticeable change in insertion losses. The resulting filter can be manufactured on 18 "x24" panels: economy of scale is the basis of its low cost.

The passive built-in components built from the dual low-loss dual-mode resonator disclosed in this invention are not limited to band-pass filters and balun filters but also include multiplexers. The out-of-band susceptance of the resonator-based bandpass filters disclosed in this invention is equivalent to filters based on shunted resonators. The parallel connection to a common terminal therefore follows the same principles known by those skilled in the art including the use in series of high reactance elements to transform susceptance out of bandwidth.

The great separation between the modes and the possibility of using the filters on asymmetric or differential signals without changing their structure makes it possible to design multiplexers without having to remove the spurious signal, the common mode of resonance of the differential filters knowing that the parasitic effect is very far from the frequency band of interest.

The stripline structure described in this invention is suitable for the insertion of transformers, couplers, transformer baluns, merchant baluns, baluns using regressive wave topologies with the advantage that a wide range Even mode impedance combined with a low impedance in odd mode (necessary for the strong coupling of the dual-mode resonator) lead to large bandwidths.

The many advantages of the present invention and its strip-line environment are obvious.

The other systems, methods, functionalities and advantages of the present invention will be or will become obvious to those skilled in the art after examining the following diagrams and the detailed description.

The intention is that any additional systems, methods, features or benefits derived from the invention are included in this specification and are protected by the claims of the invention.

IV) Brief presentation of the different flowers

After describing the invention in general terms, we will now refer to the various accompanying diagrams and figures that are not necessarily drawn to scale and where:

FIG. 1 shows a view of a bi-mode organic resonator according to an embodiment of the present invention. FIG. 2 shows the equivalent circuit diagram of a filter architecture based on two-mode organic resonators according to an embodiment of the present invention.

FIG. 3 shows an exploded view of a first organic filter according to an embodiment of the present invention.

FIG. 4 illustrates a manufacturing methodology for an organic dielectric filter according to the present invention such as the first organic filter of FIG.

FIG. 5 illustrates a design methodology for an organic dielectric filter according to the present invention such as the first organic filter of FIG.

FIG. 6 shows a three-dimensional view of a second organic dielectric filter according to an embodiment of the present invention.

FIG. 7 is a graphical representation of the distribution coefficients for the organic filter of FIG.

FIG. 8 shows a three-dimensional view of a third organic dielectric filter according to an embodiment of the present invention.

FIG. 9 is a graphical representation of the distribution coefficients for the organic filter of FIG.

FIG. 10 shows a three-dimensional view of a fourth organic dielectric filter according to an embodiment of the present invention.

FIG. 1 is a graphical representation of the distribution coefficients for the organic filter of FIG.

FIG. 12 is a graphical representation of the distribution coefficients for the organic filter of FIG.

FIG. 13 is a plan view of an organic triplexer according to an embodiment of the present invention.

FIG. 14 is a graphical representation of the distribution coefficients for the organic filter of FIG. FIG. 15 is a three-dimensional view of a fifth organic dielectric filter according to an embodiment of the present invention.

FIG. 16 is a graphical representation of the distribution coefficients for the organic filter of FIG.

VI) Detailed description

The present invention will now be described in more detail below with reference to the figures in which some, but not all, embodiments of the invention are presented. In fact, it is possible to implement this invention in many ways. It is therefore wrong to interpret that the embodiments presented here are the only ones. The disclosed embodiments are designed to meet the legal requirements of a patent application. There is correspondence between the numbers indicated in the text and those indicated in the figures.

Below and with reference to FIG. 1, the operation of a two-mode resonator designed according to the invention is explained. The dual-mode resonator 10 of FIG. 1 is composed of the transversely coupled transmission lines 12 and 14 printed on the opposite faces of a thin dielectric organic layer 16 having a high dielectric constant. The laminate material 16 has a typical thickness of less than or equal to 25 μm, a dielectric constant, Er 2, greater than or equal to 10 and preferably completely processed with a well controlled thickness. Thickness tolerances of 5% or less are achievable with current technology. The coupled transmission lines 12 and 14 are presented as rectilinear, uniform and perfectly symmetrical according to one embodiment of the invention. Special couplings or requirements on the layout of lines may lead the designers of the filter to use non-rectilinear lines, non-uniform widths and couplings, or to shift the lines, thereby losing symmetry. The resonator is integrated in a stripline environment with an upper lower mass 18 and lower mass of the resonator by the low loss and low dielectric constant laminates 22 and 24. The upper laminate 22 and the lower laminate 24 have a typical thickness greater than or equal to 50 μm, dielectric constants ErI and Er3, respectively, ranging from 2 to 5, and a tangent of the loss angle which can be as low as 0.0005. Mongia Bahl and Barthia in RF and Microwave Coupled-Line Circuits represent such architectures stripline in lines coupled transversely in a homogeneous medium (Er1 = Er2 = Er3) and non-homogeneous (Er2> Er1, Er3) or in an inverted microstrip arrangement (Er2 = 1) and suspended (Er1 = Er3 = 1). The then achieved coupling is used in the coupler topologies but not as a resonator with distinct resonance modes included in the filters. So that the stripline structure in transversely coupled lines described so far becomes a compact and efficient resonator, we couple the lines interdigitally with the terminations 26 and 28 connected to the ground. This can be done by connecting the terminations to a side shield by partition, coplanar integrated or external or by connecting the terminations to the upper and lower ground planes respectively through the vias 26a and 28a. The free terminations 30 and 32 are the inputs and outputs of the resonator if it is controlled in differential. If the resonator is not differential controlled, one of the 2 terminations is left open and only one termination, 30 or 32, can be considered as input / output of the resonator. The resonator described in FIG. 1 has two main resonant frequencies: it is a two-mode resonator. Of course, as with any other distributed element, all the harmonic resonances must also be taken into account, but they are excluded from the description of this invention. The separation between the two resonance frequencies is a function of the coupling between the lines 12 and 14.

Such interdigitated and transversely coupled stripline arrangements have already been used in LTCC but this technology faces many obstacles in achieving the important coupling required by the resonators of the present invention and resonators of which the resonance modes are too close to one another unless the transmission lines are very long and very thick. Inoue and Awai in Broadside-Coupled Interdisital Resonators describe transmission lines of 19mm long and 2mm thick to separate the 2 modes sufficiently from each other. Such dimensions are not realistic in current products. One way to alleviate this problem is to design differential filters and drive the resonator differentially as described by Fukunaga and Wada in A Coupled Multilavered Balanced Filter using Resonators based on Interdigitally-Coupled Lines. The higher frequency mode is the common mode and does not affect the performance of the differential filter unless it is connected to other filters as part of a multiplexer or unless there is a need for rejection common mode. LTCC manufacturing processes have additional limitations. The metal is screen printed on one layer per sheet, the misalignments in the stack of layers and the inaccuracy of the screen printing lead to a transverse coupling with very low tolerance. On the other hand, direct laser imaging and automatic optical inspection (AOI) are very accurate to create patterns metal faces on both sides of the organic dielectric layer 16. The LTCC will face reliability problems for large components, especially when several ceramic materials have to coexist what is necessary to achieve a strong coupling , while organic laminates with several dielectric constants do not have such limitations. The metal used by the organic printed circuit manufacturing processes is a thick copper while the conductive pastes used in LTCC are silver pastes whose overall resistivity (which takes into account the defects of shape and roughness), and the thickness, lead to marginal quality factors for drivers. In printed circuit assembly processes, the metal used for the resonators is obtained using very flat copper sheets whose roughness RMS is less than the skin thickness of the material. In addition, fine engraving allows steep copper edges of 9, 12, 17 microns or more. Although the conductor quality factor is proportional to the square root of the conductivity of the metal, it is preferred to use aluminum, molybdenum, or any other metal to meet specific requirements. The defects of the LTCC process result in less reliable two-mode resonators and higher losses. The cost of the panel multiplied by 10 because of the raw material, the high temperature manufacturing processes and the low economy of scale complete the list of disadvantages of an LTCC solution compared to a solution of stacks of organic laminates for the integration of passive components based on strongly coupled dual mode resonators as disclosed in this invention.

The state of the art based on LTCC used homogeneous media (the difficulty of making different ceramic materials coexist was discussed above). In the present invention a non-homogeneous medium is required to achieve the necessary coupling while maintaining short transmission lines. The dielectric layers 22 and 24 can be made of any low-loss organic material such as Rogers (trademark), Dupont (registered trademark), Nelco (registered trademark), Arlon (trademark), Hitachi (trademark) ), Taconic (trademark) such as LCP, PTFE composites, or any other thermoplastic / thermosetting polymer whose dielectric constant is between 2 and 5, the thickness greater than 50 microns and a tangent of the loss angle from 0.02 to 0.0005 at a frequency of some GHz. The stripline stack can be chosen asymmetric but for optimal coupling and satisfactory operation in differential mode, it is preferable to have dielectric layers 22 and 24 with identical mechanical and electrical characteristics. The central organic layer 16 is a thin material, fully processed at high dielectric constant such as the product MC25ST ^® Oak Mitsui (registered trademark): a charged resin that can be delivered in a thickness as low as 8 or 12 microns with a dielectric constant of about 17 to 1 GHz. Other choices could be a filled polyimide material such as DuPont (trademark) HK1 1 ^® having a thickness of 14 microns, a 3M barium titanate loaded epoxy (registered trademark), a Rogers ceramic filled PTFE composite ( registered trademark) whose thickness can be reduced to 25 microns.

Copper-coated material on both sides (DiClad) will be preferred to the careful copper plating so that the RMS roughness remains well below the skin thickness of the metal which helps to reduce the conductive losses. A thickness of 5 times the skin thickness allows 99% of the electrons to move freely. As the current flows on both sides of the lines, 10 times the skin thickness at the lowest operating frequency can be considered sufficient: this represents 17 microns when talking about the 3rd and 4th wireless communication standards generations. These materials have been developed with the intention of offering a high density of capacitance in order to realize power decoupling and recently, to board capacities for radio and microwave applications. As there is a trade-off between a high dielectric constant and the tangent of the loss angle, these materials have been developed for decoupling, DC blocking, or adaptation applications for boarding mobile phones or PDAs (Personal Digital Assistant) most passive components that do not intervene in filtering where losses are much more critical.

Fine dielectric dielectrics with a high dielectric constant have not been thought to be the central component of small bi-mode resonators but to ship localized components. Advances in thickness, tangent loss angle and dielectric constant in a multi-million dollar industry attempting to switch from MCM-C to MCM-L, are changing the outlook for the world of organic laminates. New hybrid materials whose charges are capable of doping the resins and resulting in high dielectric constants, low losses, thin sheets at high breakdown voltages and other interesting attributes such as compatibility with standard PWB processes or good adhesion of copper, thanks to mechanical or chemical bonding, are the true vectors of the present invention. They are the key to obtain a strong transverse coupling between interdigitated lines in a non-homogeneous organic medium. The susceptance slope of the bi-mode resonators described in this invention is very important which increases the need for coupling between the different resonators in the arrangement. This results in smaller spaces between the resonators resulting in smaller filters. Thanks to the availability of low loss organic dielectric layers with constantly improved mechanical characteristics, the off-load quality factor of the resonator at frequencies below 10GHz depends mainly on the quality factor of the conductors. With 508 microns in thickness, the selected PTFE composite laminates 22 and 24 with a loss angle of loss of less than 0.001, combined with a thin organic core layer with a loss angle of loss of 0.008, lead to a factor the quality factor of the conductors is proportional to the main dimension and the width of the resonator. Then there is a clear compromise between size and performance. The tiniest bi-mode resonators whose implementations are presented later, will have a driver quality factor just below 70, while larger implementations will significantly increase that figure.

The bi-mode resonator described so far is a shunt resonator with high susceptance outside the band. The shunt resonator of FIG. 1 is transformed into a series resonator when terminations 30 and 32 are left open instead of connected to ground. Such resonators are, for example, an ideal means for making very small DC blocking circuits.

The dual-mode shunt resonator of FIG. 1 can be further transformed into a band-shunt differential resonator for the notch filters when terminations 30 and 32 are left open instead of grounded.

The implementation of the series resonator and band shunt differential resonator is shown in FIG. IA; the only notable differences with FIG. 1 being the attribution of terminations. In one embodiment of the present invention and the implementation of the series resonator, the termination 50 is the input and the termination 52 is the output of the resonator while all other ends of the lines are left open.

In the implementation of the band-shunt differential resonator, terminations 50 and 52 are the inputs and outputs of the resonator. Another dual-mode resonator configuration belonging to the class of bi-mode resonators disclosed in this invention is the asymmetric band-shunt resonator. FIG. IB describes an embodiment of the invention for such a resonator. Many similarities with FIG. 1 and the resonator are distinguished only by different termination assignments. Termination 70 is the asymmetric input / output of the shunt resonator, termination 72 is connected to ground via via 74 and all other ends of the lines are left open. Such resonators are very useful in post-amplification filtering where both low losses and attenuation close to the bandwidth are required.

Resonators are the bricks that make up the filters. Their shape, their size, their susceptance slope at resonance, their quality factor impact the shape and performance of the filter. FIG. 2 is the equivalent circuit diagram of the filter 200 constructed from resonators described in FIG. In fact, the shunt resonators 202 with a susceptance Bi (w), i = 1 to N are bi-mode structures in organic laminates as previously described and at the basis of the present invention. All components such as diplexers, duplexers, multiplexers, balun, pseudo-highpass and bandpass filters that are within the scope of this invention are constructed from this resonator.

An expert in the field may not be limited to using only these simple resonators based on lines of different shapes, sizes and widths that constitute the backbone of the invention. If necessary, it will be possible to add sections of bands, whether they are short-circuited or left open, and load capacitors whose effects will influence the resonance frequency. FIG.2 uses the variable N to represent N resonators. It is well known that the rejection of the filters can be improved by increasing the number of resonators: thus N is arbitrary and depends on the particular needs of the designer.

In FIG. 2, the load 204 and the source 206 may be asymmetrical or differential thanks to the fact that the two-mode resonators can be controlled or not in differential mode. The coupling between the load and the resonator 208 and the casting between the source and the resonator 208a are represented by ideal admittance inverters but can also be any other impedance matching element. In concrete applications, these may be captured line couplings, coupled line couplings or localized element coupling such as parallel plate capabilities. In the case where the source and / or the load is coupled only to a single resonator, the impedance matching is then the only function of the couplings 208 and

208a. But in the case where the source and / or the load is coupled to more than one resonator or in the When the source is coupled to the load to generate additional transmission zeros or to correct the group delay, it is also important for the couplings to behave as an admittance inverter with a phase shift of 90 °, or at least as in a narrow band approximation around the bandwidth. The inter-resonator couplings 210 are represented by ideal admittance inverters. The inter-resonator couplings made according to an application mode of the dual-band resonator of FIG. 1 are low-resonance frequency mode magnetic field couplings and high-resonance frequency mode electric field couplings. Direct connections between the resonators (very short connection lines) would allow inductive couplings in the low resonance frequency mode. The degree of coupling will depend on the connection position somewhere between the ends of the resonator transmission lines. Adding to the structure of the single resonator shown above, short-circuit or open-ended band sections and load capacitances would open the door to many types of coupled-line couplings or localized elements and the coupling between the resonators would be magnetic, electrical or a combination of both.

When there is a mixture of inductive / magnetic coupling and capacitive / electrical coupling, the coupling depends on the frequency and cancels out a particular frequency thus creating a finite frequency transmission zero. When the frequency coupling dependency can be neglected, the circuit topology presented in FIG. 2 can not generate a finite transmission zero because there is no triplet, quadruplet or other non-adjacent couplings. between the N resonators. With the right combination of non-adjacent couplings between the N resonators, one can generate (NT) transmission zeros. By adding couplings between the load / source and a non-adjacent resonator, one can generate (NI) transmission zeroes and with a coupling between the source and the load, one can generate N transmission zeros. In the last two cases, we obtain a non-resonant nodal model where the couplings involving the load and the source play a crucial role. Other non-resonant nodal models where resonators based on the present invention are coupled to non-resonant susceptances, as described for example by Amari in Singlets, Cascaded Sinεlets, and the Non-Resonnatins Mode Model for Advanced Modular Design Elliptic Filters, complete the types of architectures based on the resonator of the invention. These variations of architectures allow the addition of transmission zeros which improve the rejection of the filters outside the bandwidth or smooth the group delay of a filter in order to limit the dispersion.

FIG. 3 shows the physical layout of a bandpass filter corresponding to the circuit-equivalent diagram of FIG. The dielectric filter of FIG. 3 is a second-order structure composed of two resonators according to one embodiment of the invention. As a possible application of the invention, FIG. 3 illustrates an embodiment in the form of a surface-mounted component (CMS) of a passive component constructed from the resonator of FIG. The band-pass filter 300 comprises the two-mode resonators 302 and 304, each of which is made from interdigitally coupled interdigital lines situated on the two metal layers 303 and 305, themselves separated by a thin dielectric organic layer with a high dielectric constant. 306. This thin material is preferably a laminate completely treated so that its small thickness has a high tolerance. The material may be a PTFE composite, a resin or polyimide loaded with ceramic material, ferroelectric or paraelectric but is not limited to these hybrids. The ground terminations 308, 308a, 308b and 308c are connected to the side ground shields 310 and 310a but may also be connected to coplanar ground planes, located in an inner layer or on the upper and lower faces 312 and 312a and related through metal through vias or other techniques such as blind vias, buried vias or micro-vias.

Since the two resonators are very close to one another, the electromagnetic coupling can be used to couple the two resonators and thus avoid any other external circuit, which saves space and does not require a component or layers additional. The bi-mode resonators described in this invention allow magnetic coupling at the first resonance mode since the lines are smaller than a quarter wave in electrical length. The small susceptance of two-mode resonators in the first resonance mode is very important, increasing the couplings in the filter making it more compact.

The possibilities of coupling the filter and the geometry of the resonators are not limited to a simple electromagnetic interaction of the lines as explained above. We could link the resonators together at any point between the open and short-circuited ends of the lines. This gives a magnetic coupling all the stronger as the point of capture is far from the mass. Parallel or interdigitated plate capacitors, formed by structures attached to the open ends of the two resonators, overlapping or rub shoulders, can give electrical coupling. Coupling between open or short-circuited line sections attached to the open ends of the two resonators can provide magnetic, electrical, or a combination of the two.

As long as the designer adds additional susceptance created by the shorted line sections or the shorted capacitances, he must take care to maintain the resonant frequency desired for the resonator resulting from these modifications. A mixture of magnetic and electrical coupling leads to frequency-dependent coupling which is eliminated at a particular frequency thereby creating a transmission zero before or after the bandwidth.

The top and bottom shielding electrodes 312 and 312a are respectively placed on organic dielectric layers 314 and 316 which sandwich the organic laminate layer 306. In this embodiment of the invention, the shielding electrodes of the top and bottom 312 and 312a are connected together through the side shield plates 310 and 311a but can also be connected by through metallized holes or a combination of vias in their various forms (blind, buried, microphones).

The dielectric layers 314 and 316 may be organic laminates bare or coated with a copper face, such as PTFE, LCP or other suitable dielectric material with low losses, whose dielectric constant is lower and the thickness greater. The non-homogeneous stripband medium created by this fact is the cause of the strong coupling which favors the bi-mode resonators in size and performance.

The inputs 318, 318a and the output 320 of the filter are coupled to the filter by sensing the resonators for reasons of impedance matching. While this method allows the filter to keep a minimum size we could have also used a coupling by localized elements, coupled lines or line sections. In one embodiment of the invention, one side of the filter is controlled in differential and the other in asymmetric but we could also have any other possible combination. The first mode of resonance of the dual mode resonator is the differential mode. For this operating mode, the filter has the same behavior as the inputs and outputs are controlled in differential or not. The inputs / outputs of the filter are connected to the reception pads of the surface-mounted component through the through-plated holes 322, 322a and 322b or a combination of vias in their various forms (blind, buried, microphones). We can easily generalize the integration of these passive components into System-in- package, BGA, or any other type of encapsulation by connecting the filter's inboard outlets to the top-layer wiring docking lanes, the bottom layer balls, or any other type of circuit.

The manufacture of a filter according to this invention is explained below with reference to the diagrams of FIG. The first step is to correctly select the central organic material. We start the manufacturing process with a thin material, coated on both sides and high dielectric constant. The coating can be annealed "roll-annealed", deposited by redox "electroleaked", vacuum deposited "vacuum deposited", or made by any other process leading to very soft copper surfaces. The starting organic layer can be bare, or coated on one side, but the option of flat-faced double-sided coating allows us to avoid any problem of metal roughness that impacts the losses in the conductors. According to one embodiment of the invention, the bi-mode resonators and the filters that result therefrom do not require drilling to create connections between the two metal layers. Removing this step saves money and space; this eventually makes it possible to select a material that could be fragile during laser drilling. Now, more complex architectures of systems or filters may still require the sub-step of laser drilling.

The next step is metallization of the central laminate. The formation of the circuit can be done using additive processes, semi-additives, substractive. Printing and engraving create the patterns and geometry of the device. Current technologies for direct laser imaging, fine line photolithography and automatic optical inspection help to create geometries with line gaps below 15 microns and very high alignment tolerances between layers not possible to obtain with the screen printing processes.

Once the metal patterns have been defined, the 3 dielectric layers are stacked together. The upper and lower dielectric layers, which may be bare or coated on one side, sandwich the thin core layer. The fourth step is to make internal and external metal connections. According to one embodiment of this invention, SMT terminations, ground and signal connections can be made through mechanically drilled through-through holes. The side shield can be made differently; lateral trenches can also be made by sawing. All the holes obtained are then covered with copper by chemical plating, misting, or any other suitable method of depositing. Then, the metallization of the upper and lower layers is carried out as shown in step 5. The electroplating and direct etching steps create the outer layer surfaces, the CMS docking areas. A finishing varnish, using chemical deposition, dip tin, solder or other suitable methods, helps to protect the filter conductors, while a number of protective layers can also be added, such as than anti-welding or photosensitive for solder balls for example.

While the metal mentioned throughout this specification is copper, it is also possible to use aluminum, molybdenum, or any other metal. Additional steps in the manufacturing process can be envisaged; in the case where a system-in-package requires a hood molded on the top layer, the transfer molding, the "dam-and-fill", or any other suitable method for varnishing the integrated passive component, can be used. The separation process, using a saw or a hot blade, completes the overall process before the final test and control.

The design methodology of the filters is explained here and with reference to FIG. Starting from the low-pass model of the response of an optimum elliptic or generalized Chebichev filter, we can trace the responses in transmission (S21) and in reflection (SU) from the knowledge of the order of the filter, the in-band ripple and the normalized frequency positioning representing the finite transmission zeros.

A simple recursive algorithm helps to determine the normalized frequencies of the reflection zeros of a generalized Chebichev filter that allows the best out-of-band rejection for a given ripple in the given band and to have finite transmission zeros. For a given value of the reflection losses, knowing the position of the transmission and reflection zeros, the distribution coefficients S21 and SI1 are obtained. This information can be plotted for the low pass model and its bandpass equivalent given the bandwidth edges and a projection transformation.

The dual-mode resonators of this invention have a low-frequency resonance behaving almost like the parallel arrangement of a capacitance and an inductance which leads us to use the standard transformation between a low-pass and a pass-band .

For a given filter order and reflection loss value, the designer can change the position of the filter bandwidth edges and the finite transmission zeros until the s-parameters satisfy it. .

The next step is to determine the values of the general coupling matrices of the filter we want to implement. The values of the diagonal are related to the nodal points of the filter as the source, the load, the susceptances of non-resonance and the resonators; the asynchronous filters will have different values for different resonators. The other values of the coupling matrix are the couplings between the source, the load, the non-resonant nodal points and the resonators. These values are 0 if there is no coupling and are non-zero if a coupling exists. The s-parameters can be determined by the inverse of the general coupling matrix.

Since we know the position of transmission zeros, reflection zeros, and band edges, we can calculate the optimization of the coupling matrix with the following error function:

where fi are the frequencies of the normalized transmission zeros, fj are the frequencies of the normalized reflection zeros and ε is related to the RL reflection loss at the band edges by the following equation:

1

RL

V io ¹⁰ - 1

One can easily check if the optimization will converge or not: a determinant of the coupling matrix is a polynomial whose roots are the transmission zeros. Start the optimization with all the non-zero values of the coupling matrix equal to 1. When the optimization has converged, the non-zero final values are the standardized couplings and the frequency offsets of the low-pass model. To obtain the corresponding couplings of the band-pass filter, such as the external coupling Qe and the inter-resonator coupling k, we have to adjust the couplings using the susceptance slope of the resonators and the fractional bandwidth of the filter.

The third step is to perform electromagnetic simulations of the filter. We can run successive simulations with 2 activated resonators to determine Qe and k, knowing that all other unused filter resonators are connected to ground. The books are full of examples of how to calculate k and Qe using group delay, phase cancellations, or separations between resonances.

It is less known that the entire coupling matrix can be calculated from a single electromagnetic simulation using the appropriate system identification software currently available. The extraction of the rational model makes it possible to pass distribution coefficients to the general matrix of coupling in a single step whatever the order of the filter thus saving time in the development of the filter.

VI) Examples

Example 1

An order 3 filter using only electromagnetic coupling has very simple geometries. We have already mentioned the fact that, for the low resonance mode, the coupling is magnetic because the transmission lines are shorter than the quarter wave. Magnetic couplings between resonators tend to show a more aggressive slope after the bandwidth. In many cases borrowed from the RF telecommunications industry such as Bluetooth filter rejections, WLAN 802.1 lb / g or WiMAX, one needs a very abrupt profile before the bandwidth. This can be achieved with the good non-adjacent coupling value in a three resonator architecture.

1

This is the coupling matrix for a third-order asynchronous filter with a bandwidth of 2.3GHz to 2.7GHz and a transmission zero around 1.9GHz to remove cell network interference up to 1.98GHz.

with qel = qe3 = 1,122

FIG. 6 represents a 3D view of the band-pass organic filter based on two-mode resonators according to one embodiment of the invention and characterized by the coupling matrix M1 above. The filter 600 comprises 3 bi-mode resonators 602, 604 and 606 which are magnetically coupled with each other. The two-mode resonators are constructed from interdigitated lines transversely coupled on either side of a dielectric layer 616 in a non-homogeneous stripline stack of 4 metal layers and organic laminates.

The other laminates 618 and 620 separate the resonators from the upper and lower ground electrodes, themselves connected to the side shield, providing a reference to ground for the 4 terminations 622, 622a, 622b and 622c.

The symmetry with respect to plane 608 allows the designer of the filter to differentially control the inputs and / or outputs of the filter. In this example, the input is differential with the input lines 610 and 612 while the output is unbalanced with the output line 614. The presented geometry provides both the necessary symmetry and a strong non-adjacent coupling between resonators to form a triplet with a transmission zero before the bandwidth. This is the only geometry that can do this by a simple electromagnetic coupling between the transmission lines. We need a through metal hole 624 to connect the internal resonator to ground. Shunted band sections or additional capacities could be added to create a non-adjacent coupling while keeping the symmetry of the filter. A transmission zero before the bandwidth is possible for a lower order filter having a frequency-dependent inter-resonator coupling and canceling before the bandwidth.

The performance of the filter is shown in FIG. 7 for both the differential mode and the common mode from 1 to 18 GHz. This makes it possible to visualize the very large separation of the two resonance modes obtained thanks to the strong coupling used to create the two-mode resonators.

The actual dimensions of this filter are: 1.3x1. lmm. Example 2

A very similar example with this third-order filter having a band from [2.37GHz to 2.53GHz] and a transmission zero around 2.1GHz in order to remove near interference like that of CDMA Wideband frequencies up to at 2.17GHz.

Its coupling matrix for an asynchronous implementation is:

with qel = qe3 = 1.0042

FIG. 8 represents a 3D view of the organic band-pass filter 800 based on bi-mode resonators according to one embodiment of the invention and characterized by the coupling matrix M2. The filter 800 comprises 3 bi-mode resonators 802, 804 and 806 which are magnetically coupled with each other. The two-mode resonators are constructed from interdigitated lines transversely coupled on either side of a dielectric layer 814 in a non-homogeneous stripline stack of 4 metal layers and organic laminates. The other laminates 812 and 816 separate the resonators from the upper and lower ground electrodes.

The difference to mention is that the filter is not driven in differential and because of this, there is no need for symmetry in the design. This less constraint allows the designer to have more freedom in the geometries while retaining the necessary inter-resonator couplings including the strong coupling between non-adjacent resonators. In the filter 800 this advantage is exploited to achieve a filter requiring no connection between the layers. The only drilling or sawing required is for the trenches that will accommodate the plans 818 and 820 of the side shield on which will connect the points to the ground: 822, 822a, 822b, 822c, 822d and 822e. Depending on the needs of the system, the asymmetric terminations 808 and 810 can also be connected to other metal layers. Beyond a small advantage in terms of cost, the fact of no longer need connection between the layers impacts very favorably the final dimensions of the filter because the hole, the lozenge and the imposed spacing of the vias, have sizes of the same order as the transmission lines used. The performance of the filter is shown FIG.9. The plot clearly shows that the second resonance mode is very far away because of the strong coupling used to create the two-mode resonators.

The actual dimensions of this filter are: 1.3x1. lmm.

Example 3

A second order filter can generate a transmission zero by having a frequency-dependent coupling that vanishes at a well-chosen frequency. A simple way to do this is to mix magnetic coupling and electrical coupling. The magnetic coupling comes from the electromagnetic field between the resonators. The easiest way to create an electrical coupling is to do it via a capacitive coupling between parallel plates. Now, if we want to avoid using vias between layers to reduce manufacturing costs by removing the drilling operation and reduce space, given the dimensions imposed around the vias, we can implement a coupling capacitive using an interdigitated capacitance.

FIG. 10 represents a 3D view of the organic bandpass filter 1000 based on two-mode resonators according to one embodiment of the invention. The interdigitated capacitances 1002 and 1004, combined with the magnetic coupling between the two-mode resonators 1006 and 1008, create a frequency-dependent coupling. The frequency with which this coupling vanishes is a finite transmission zero. It can be positioned before or after the bandwidth.

The present embodiment of the invention shows an asymmetric architecture with the input 1010 and the output 1012 connected to their respective resonators 1006 and 1008. The symmetry of the plot offers the possibility of differentially controlling the input and / or the output .

FIGs. 11 and 12 show the performance of filters with such a geometry and central frequencies positioned respectively at 3.5GHz and 5.5Ghz.

The finished transmission zero is chosen to be placed before the bandwidth. In the case of the 5.5 GHz, it is necessary to reduce not only the lengths of the resonators but also, in the same order of magnitude, the size of the interdigital capacitance between the resonators. The actual dimensions of this filter for 3.5GHz operation are: 1.2xO.8mm Example 4

The present invention is not limited to bandpass filters and balun filters but also covers any integrated passive device among which the multiplexers.

The possibility of having a difference in the differential and common modes over a very wide band thanks to the strong coupling between interdigitated lines transversely coupled in a non-homogeneous organic medium and the possibility of having the inputs and outputs of the filter driven in differential, without impacting the low resonance mode performance of the resonators described in this invention, simplifies multiplexer architectures in multilayer organic laminates. For differential filters, there is no need to remove the parasites from the common mode resonances because they are very far from the considered bandwidths.

The multiplexer 1300 of FIG. 13 is an embodiment of such an architecture according to the present invention. The filters 1302, 1304 and 1306 have their respective inputs 1310, 1312 and 1314 which can be controlled either in differential or in asymmetric mode; the symmetry of the geometry of the filter ensures a good performance in differential. The center frequencies of the filter are 2.5GHz, 3.5GHz and 5.5GHz. The filters are connected in parallel to a common output 1308. To achieve this, the outputs of the filter must be transformed by a high-reactance serial component. While a small capacity does the trick, 1316 for the high frequency filter 1306, the low frequency filters 1302 and 1304 respectively require the high impedance lines 1318 and 1320. The higher the impedance, the shorter the lines are . This is why the processes of direct laser imaging, "thin-line" photolithography in the PCB industry with increasingly fine line widths, combined with a good separation from the mass in a non-homogeneous medium, help enormously with the implementation of high impedance lines, for the transformation of reactance during the designs of multiplexers.

The overall filter design takes into account the serial reactance transformation components to ensure good bandwidth performance while maintaining low susceptance out of the band required for parallel filter connections. The performance of such an architecture is shown in FIG.

Example 5 This example is a second-order filter with 2 transmission zeros due to the non-adjacent coupling involving the non-resonant nodes of the filter. In this non-resonant nodal model, the source is directly coupled to the load.

FIG. 15 shows a three-dimensional view of the second-order organic band-pass filter 1500 based on two-mode resonators according to one embodiment of the invention. 2 bi-mode resonators 1502 and 1504 are coupled together by their electromagnetic fields. 2 parallel plate capacitors 1506 and 1508 are coupling elements located between the source and the load with their respective adjacent resonators. The capacitive effect of the marginal field, present in the region 1510 between the source and the load, is the fourth coupling present in the filter and is at the origin of the finite transmission zeros that we can observe. The number and sign of the couplings chosen here set a transmission zero on each side of the bandwidth. The geometry of the filter requires no via, thus saving space and cost, since the side shield provides the resonators with the mass reference. The performance of the filter is shown in FIG. 16. The red line indicates the reflection behavior of the filter. The blue line indicates the transmission behavior of the filter in case there is no coupling of the source to the load. The pink line indicates the transmission behavior of the filter with coupling (non-resonant nodal model): in this case, 2 transmission zeros appear. The actual dimensions of this filter are: 1.4xO.8mm.

This concludes the description of the filters designed according to an embodiment of the present invention. However, those skilled in the art will appreciate that the teachings formulated in this invention literally apply to other integrated passive components, to modules built from combinations of these integrated passive components, with or without more than standard ftF topologies. and microwaves, such as localized filters, merchant baluns, couplers, or any other signal processing circuit.

The non-homogeneous 4-layer metal strip-line described in this document can obviously be generalized to a medium with more layers, where the additional layers can be used for more separation between the upper and lower electrodes, for the routing of the system, to report components on the upper and lower layers or any other function requiring to be embedded in the organic substrate.

Claims

1) signal processing module for wireless applications characterized in that it comprises: two transversely coupled and interdigitated transmission lines (12, 14) printed on each side of an organic dielectric layer (16) having a dielectric constant high, greater than 10 and a thickness less than 25 microns, where two terminals (30, 32) are open and the other two (26, 28) are connected to ground; two shielding electrodes (18, 20) separated from the aforementioned transmission lines by two organic dielectric layers (22, 24) having a dielectric constant of less than 5 and a thickness greater than 50 microns; these layers are stacked so as to sandwich the thinner dielectric layer, creating a non-homogeneous "stripline" environment and selected such that the two transmission lines interact with one another to form a bi-directional resonator. mode, where the second resonance mode is at a frequency greater than three times that of the first resonance mode. 2) Module according to claim 1 characterized in that it comprises two transmission lines offset relative to each other, where the lines are not straight or form meanders, where the lines are asymmetrical, the lines do not are not identical in electrical length, where the lines have non-uniform widths, where one or the other line is connected in any way to shunted line segments which are connected to the ground or left open, or mass-connected capabilities, where the lines have non-uniform coupling, or any combination of the previous cases. Terminals connected to the ground are connected to a buried coplanar ground plane, to the lateral shielding, through metallized through holes, to vias whatever their shape (blind, buried or micro-vias), or any combination of the cases mentioned here. -above. 3) signal processing module for wireless applications characterized in that it comprises: two or more modules according to claim 1 or claim 2, acting on each other by means of electromagnetic coupling, coupling of captured lines "tapped -line ", coupling of localized elements, coupling of coupled lines, or any combination of the cases mentioned; one or more of the aforementioned modules are coupled to a source and a load, which are either differential or asymmetric ("single-ended"), by coupling taped lines "tapped-line", coupling localized elements , coupling of coupled lines, or any combination of the cases mentioned. The set of couplings creates a filter based on two-mode resonators. 4) Module according to claim 3, characterized in that the two-mode resonators (302, 304) have a symmetry allowing at least one of them to be controlled differentially, creating a balun filter based on bi-mode resonators.

5) Module according to claim 3, characterized in that the coupling between the two-mode resonators (302, 304) is based solely on the interaction of electromagnetic fields, where the source and the load are coupled to the filter by sensing the resonators and the terminals (308, 308a, 308b, 308c) connected to the ground are connected to insulating faces on two sides (310, 310a); this results in a high frequency filter geometry simple, compact, and requiring no via in its implementation. 6) Module according to claim 3, characterized in that the dual-mode resonators

(602, 604, 606) have a symmetry allowing one of them to be controlled differentially, where the coupling between the resonators is based solely on the interaction of the electromagnetic fields; this results in a very compact "triple", which has a transmission zero at a finite frequency before the bandwidth. 7) Module according to claim 3, characterized in that the coupling between the two-mode resonators (802, 804, 806) is based solely on the interaction of electromagnetic fields, and the terminals connected to ground are connected to faces insulating on two sides (818, 820); this results in a very compact "triplet", which has a transmission zero at a finite frequency before the bandwidth and which requires no via in its realization.

8) Module according to claim 3, characterized in that the coupling between the two-mode resonators (1006, 1008) is a combination of frequency-dependent electrical and magnetic couplings, so that a transmission zero is obtained. at a finite frequency before or after the bandwidth; the combination of an interdigitated capacitance or a coupling by open line sections, with the coupling of electromagnetic fields, makes it possible to produce such a filter in a very compact manner and without via.

9) Module according to claim 8, characterized in that the bi-mode resonators have a symmetry allowing at least one of them to be controlled differentially, creating a balun filter.

10) Module according to claim 3, characterized in that the source, the load, or a non-resonant susceptance are coupled to more than one dual-mode resonator, or in that the source is coupled directly to the load, which extends the present invention to all possibilities of non-resonant nodal models. 11) signal processing module for wireless applications characterized in that it comprises two or more modules (1302, 1304, 1306) based on claims 3 to 10; for each of its modules, the coupling to the load consists of a high-reactance series element, such as a coupled-line circuit having a high "odd" impedance, a transmission line (1318) with a high characteristic impedance, or a low capacitance circuit (capacitance with parallel plates (1316), interdigitated capacitances, "gap capacitor"); all these modules are connected in parallel to a common load (1308); Connecting filters with low out-of-band susceptance creates a multiplexer. One or more filters based on the two-mode resonators described in this invention can be connected to other types of filters.

12) signal processing module for wireless applications characterized in that it comprises a module according to claim 1 or 2, wherein the terminations (26, 28) previously connected to the ground are now left open, creating a series resonator whose input is a termination (30) and for output a termination (32); this arrangement makes it possible to achieve a decoupling capacitance in series, which is very compact.

13) Signal processing module for wireless applications characterized in that it comprises a module according to claim 1 or 2, wherein the terminations (26, 28) previously connected to ground are now left open, creating a shunt resonator cut -band with differential excitation which has for input / output the terminations (50) and

(52)

14) Signal processing module for wireless applications characterized in that it comprises a module according to claim 1 or 2, wherein either the termination (26) or the termination (28) is now left open, creating a shunt resonator band cutter with asymmetric excitation which has the input / output termination (70), a ground termination (72) and the other two ends of the transmission lines are left open.

15) signal processing module for wireless applications characterized in that it comprises a module according to claims 1 to 14, embedded in a stack of more than 4 metal layers for juxtaposing signal processing modules, the addition of embedded resistors, routing to surface mounted components, or any other reason.