WO2010059290A2 - Dispositif à ondes progressives à rétroaction constructive et procédé associé - Google Patents

Dispositif à ondes progressives à rétroaction constructive et procédé associé Download PDF

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WO2010059290A2
WO2010059290A2 PCT/US2009/058172 US2009058172W WO2010059290A2 WO 2010059290 A2 WO2010059290 A2 WO 2010059290A2 US 2009058172 W US2009058172 W US 2009058172W WO 2010059290 A2 WO2010059290 A2 WO 2010059290A2
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Prior art keywords
traveling wave
feedback
gain
stage
translation
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PCT/US2009/058172
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English (en)
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WO2010059290A3 (fr
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James Buckwalter
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The Regents Of The University Of California
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Priority to US13/120,422 priority Critical patent/US8258873B2/en
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Publication of WO2010059290A3 publication Critical patent/WO2010059290A3/fr

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/54—Amplifiers using transit-time effect in tubes or semiconductor devices
    • H03F3/58—Amplifiers using transit-time effect in tubes or semiconductor devices using travelling-wave tubes
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/08—Modifications of amplifiers to reduce detrimental influences of internal impedances of amplifying elements
    • H03F1/22—Modifications of amplifiers to reduce detrimental influences of internal impedances of amplifying elements by use of cascode coupling, i.e. earthed cathode or emitter stage followed by earthed grid or base stage respectively
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/04—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements with semiconductor devices only
    • H03F3/08—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements with semiconductor devices only controlled by light
    • H03F3/087—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements with semiconductor devices only controlled by light with IC amplifier blocks
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/60—Amplifiers in which coupling networks have distributed constants, e.g. with waveguide resonators
    • H03F3/605—Distributed amplifiers
    • H03F3/607—Distributed amplifiers using FET's
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/216—A coil being added in the input circuit, e.g. base, gate, of an amplifier stage
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/301—Indexing scheme relating to amplifiers the loading circuit of an amplifying stage comprising a coil
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/408—Indexing scheme relating to amplifiers the output amplifying stage of an amplifier comprising three power stages

Definitions

  • the subject matter described herein relates to traveling wave circuits that can have amplifying properties.
  • High-frequency and high-speed amplifiers are generally either cascaded or distributed.
  • a cascaded lumped element amplifier can be implemented using narrowband matching elements or broadband matching elements, examples of which are shown in the circuit 100 of FIG. IA.
  • the overall voltage gain, G is the product of the gain of individual stages. In other words, G ⁇ ⁇ G k between the input 102 and the output 104 for k stages.
  • Each stage includes a transistor 106 with the transistor gate connected to either the input 102 or the output of a previous stage drain 110.
  • Narrowband matching as in the circuit 100 of FIG. IA typically allows the highest gain for a given device.
  • a resonant network matches the impedance at the gate, source, or drain of the device to the source and load impedance.
  • These input and output matching networks are well suited to integrated implementations with finite quality factor passives and can alternatively optimize the noise, linearity, or output power of the amplifier stage.
  • a drawback of narrowband matching is that the input and output matching are only acceptable over a narrow bandwidth. Additionally, the bandwidth of the amplifier response generally reduces through multiple stages of amplification. For W-band applications, this technique has been preferred to achieve high gain, low noise amplifiers in any solid-state circuit technology.
  • Feedback techniques such as the shunt resistive feedback circuit 130 of FIG. IB is capable of wideband matching and gain response.
  • resistive shunt feedback inherently reduces the voltage gain of each stage and provides matching determined by the shunt feedback resistance.
  • shunt feedback tends to result in higher noise figure due to the addition of the shunt feedback resistance.
  • the circuit 160 includes a series of stages that each include an active device 162, such as for example a field effect transistor (FET) or a bipolar junction transistor (BJT).
  • An input transmission line 164 runs from an input 166 to a first impedance termination 170 and an output transmission line 172 runs from a second impedance termination 174 to an output 176.
  • the active device 162 of each stage has its gate or base 180 linked to the input transmission line 164, its drain or collector 182 to the output transmission line 172, and its source or emitter 184 terminated.
  • a distributed amplifier 160 absorbs the capacitances of the active devices 162 into an artificial transmission line along the input transmission line 164 and output transmission line 172.
  • the input traveling wave propagates along the input transmission line 164, excites the active devices 162 of each stage along the input transmission line 164, and is absorbed into the first impedance termination 170 at the end of the input transmission line 164.
  • two output traveling waves are generated at the drain 182 of each stage; a forward traveling wave that is constructively amplified across multiple stages and an undesired backward traveling wave that is lost in the second termination impedance 174.
  • Combining the output forward traveling wave at each stage results in a lower overall voltage gain than in a cascaded amplifier configuration.
  • the gains of each stage are summed rather than multiplied, so G ⁇ ⁇ G k for k stages.
  • Distributed amplifiers are generally limited by inherent quality factor losses along the gate and drain artificial transmission lines, so typically no more than six to eight stages can be implemented in a conventional distributed amplifier. The achievable gain therefore remains relatively low.
  • a distributed amplifier does advantageously provide a desirable broadband gain response accompanying the broadband input and output matching.
  • InP indium-phosphide
  • GaAs gallium-arsenide
  • the advance of silicon into millimeter wave regimes has more recently pushed distributed designs into silicon technologies using both n-type metal-oxide- semiconductor (NMOS) devices.
  • NMOS n-type metal-oxide- semiconductor
  • an apparatus in one aspect, includes a transmission line carrying a propagating signal along a direct path from an inlet port to an outlet port, and a feedback stage that samples the propagating signal at the outlet port, generates a feedback signal that comprises a time translation and a gain translation, and routes the feedback signal to the inlet port.
  • the propagating signal includes a forward traveling wave.
  • the time translation causes the gain translation to constructively interfere with the forward traveling wave, thereby increasing a forward traveling wave amplitude of the forward traveling wave.
  • an apparatus in an interrelated aspect, includes a first stage and a second stage.
  • the first stage includes a first transmission line carrying a propagating signal along a first direct path from a first inlet port to a first outlet port and a first feedback stage that samples the propagating signal at the first outlet port, generates a first feedback signal that comprises a first time translation and a first gain translation, and routes the first feedback signal to the first inlet port.
  • the propagating signal includes a forward traveling wave.
  • the first time translation causes the first gain translation to constructively interfere with the forward traveling wave, thereby increasing a forward traveling wave amplitude through the first stage.
  • the second stage includes a second transmission line carrying the propagating signal along a second direct path from a second inlet port to a second outlet port, and a second feedback stage that samples the propagating signal at the second outlet port, generates a second feedback signal that comprises a second time translation and a second gain translation, and routes the second feedback signal to the second inlet port.
  • the second inlet port is directly connected to the first outlet port.
  • the second time translation causes the second gain translation to constructively interfere with the forward traveling wave, thereby further increasing the forward traveling wave amplitude through the second stage.
  • the method also includes sampling the propagating signal at the outlet port in a feedback stage, generating a feedback signal in the feedback stage, and routing the feedback signal to the inlet port such that the time translation causes the gain translation to constructively interfere with the forward traveling wave, thereby increasing a forward traveling wave amplitude of the forward traveling wave.
  • the feedback signal includes a time translation and a gain translation.
  • a 10 can optionally be given by
  • gain translation, A 1 can optionally be within a range of 0 to 0.5.
  • the propagating signal can optionally further include a backward traveling wave, and the time translation can cause the gain to destructively interfere with the backward traveling wave, thereby reducing a backward traveling wave amplitude of the backward traveling wave.
  • the feedback signal can optionally further include a frequency translation.
  • the time translation of the feedback signal can optionally introduce a phase shift of approximately a quarter wavelength relative to the forward traveling wave.
  • the transmission line can optionally introduce a first phase shift while the feedback stage introduces a second phase shift that sum to approximately 180 degrees.
  • the first phase shift and the second phase shift can optionally each be in a range of approximately 50 to 130 degrees.
  • the feedback stage can optionally include a common emitter and an emitter follower.
  • the feedback stage can optionally include one or more transistors selected from a common emitter, a common source, an emitter follower, a source follower, a common base , and a common gate.
  • an apparatus can further include a second feedback stage that samples the propagating signal at the inlet port, generates a second feedback signal that comprises a second time translation and a second gain translation, and routes the second feedback signal to the outlet port.
  • the second time translation can cause the gain translation to constructively interfere with a second forward traveling wave of a second propagating signal carried by the transmission line along the direct path from the outlet inlet port to the inlet port, thereby increasing a second forward traveling wave amplitude.
  • a control signal generator can deactivate the feedback stage when the second feedback stage is active and deactivate the second feedback stage when the feedback stage is active.
  • an input traveling wave can be amplified as it propagates along a single transmission line with broadband matching such as can be provided by distributed amplifier designs. Stages of such an amplifier can be cascaded to provide multiplicative gain over a passband. Unlike a distributed amplifier, the current subject matter has no inherent limitation, other than power and area limitations, on the feasible number of traveling wave stages that can be cascaded. Each stage can compensate passive losses along the transmission line. The gain response and bandwidth of such an amplifier can fundamentally depend on the feedback gain, which can be electronically controlled.
  • a feedback network can unilaterally amplify a traveling wave along a transmission line. An active feedback path constructively contributes energy to the wave traveling in one direction while at least partially canceling the wave traveling in the opposite direction.
  • FIG. 1 is three circuit diagrams showing examples of cascaded amplifiers (FIG. IA and FIG. IB) and a distributed amplifier (FIG. 1C)
  • FIG. 2 is a circuit diagram showing a stage of an amplifier according to an implementation
  • FIG. 3 is a process flow chart illustrating a method according to an implementation
  • FIG. 4 is a chart showing S-parameters for a single stage of an amplifier according to an implementation with transconductance (gm) of 10 mS, a phase ( ⁇ ) of 90 degrees ( ⁇ /2) and a delay time in the feedback path of (Td);
  • FIG. 5 is a chart showing analytical results for dependence of bandwidth and the gain- bandwidth product on the amplifier gain for a single stage;
  • FIG. 6A is a chart showing the stability of an amplifier according to an implementation for a quarter wave transmission line in terms of K;
  • FIG. 6B is a chart showing the stability of an amplifier according to an implementation for a quarter wave transmission line in terms of
  • FIG. 7A is a circuit diagram showing a single stage for an implementation using shunt feedback in a SiGe HBT
  • FIG. 7B is a circuit diagram showing a detail of the feedback path in the single stage of FIG. 7A;
  • FIG. 8 is a chart showing feedback transconductance and delay for a 0.12 ⁇ m SiGe heterojunction bipolar transistor circuit implementation
  • FIG. 9 is a Smith chart showing analytical calculations and circuit simulations of input and output matching for a single stage of an amplifier according to an implementation
  • FIG. 1OA is a chart showing a spectrum of simulations for process and voltage sensitivity of the S21 wave for an amplifier according to an implementation
  • FIG. 1OB is a chart showing a spectrum of simulations for process and voltage sensitivity of the S12 wave for an amplifier according to an implementation
  • FIG. 11 is a diagram showing aspects of a cascaded constructive wave amplifier according to an implementation
  • FIG. 12 is a micrograph of a chip containing a 12-stage cascaded constructive wave amplifier
  • FIG. 13 is a chart showing J-parameter measurements of a 12-stage cascaded constructive wave amplifier according to an implementation
  • FIG. 14 is a chart showing measured stability of a 12-stage cascaded constructive wave amplifier according to an implementation
  • FIG. 15A is a chart showing an example of forward transmission for an increasing emitter follower bias current according to an implementation
  • FIG. 15B is a chart showing an example of backward transmission for an increasing emitter follower bias current according to an implementation
  • FIG. 16A is a chart showing the SIl return losses for an increasing emitter follower bias current according to an implementation
  • FIG. 16B is a chart showing the S22 return losses for an increasing emitter follower bias current according to an implementation
  • FIG. 17 is a chart showing gain and bandwidth tradeoffs for a 12-stage cascaded constructive wave amplifier according to an implementation
  • FIG. 18 is a chart showing measured and simulated noise figures for a 12-stage cascaded constructive wave amplifier according to an implementation
  • FIG. 19 is a chart showing measured gain compression for a 12-stage cascaded constructive wave amplifier according to an implementation
  • FIG. 20 is a circuit diagram showing a system having a Darlington amplifier circuit coupled to a cascaded constructive wave amplifier stage;
  • FIG. 21A is a chart showing gain responses for the Darlington amplifier of the system in FIG. 20;
  • FIG. 21B is a chart showing gain responses for the cascaded constructive wave amplifier stage of the system in FIG. 20.
  • FIG. 22A is a circuit diagram showing a half-duplex radio transceiver; and FIG. 22B is a circuit diagram showing an implementation of a bidirectional amplifier with multiple stages.
  • the subject matter described herein relates to constructive feedback circuits that can be used in a number of applications, including but not limited to wide-band, high-gain millimeter wave amplifiers.
  • a cascaded constructive wavefront amplifier hybridizes traveling wave and cascaded amplifier design to offer high gain and wide bandwidth.
  • the wide -band, high-gain millimeter wave amplifier is described with respect to imaging and communication applications, other applications of the current subject matter are also within the scope of this disclosure.
  • the description of the wide-band, high-gain millimeter wave amplifier uses the example of a 26dB, 100 Gigahertz (GHz) wide-band, high-gain millimeter wave amplifier, this implementation is exemplary.
  • the millimeter-wave (mm-wave) frequency regime offers growing opportunities for silicon integrated circuits in communication, radar, and imaging applications.
  • W-band (75-111 GHz) circuits have relied on GaAs or InP devices, but recent work demonstrates silicon front-ends for W-band imaging and communications in spite of the low intrinsic bandwidth of silicon devices.
  • the availability of W-band silicon integrated circuits could realize medical and security imaging through active and passive imaging arrays.
  • a traveling wave stage enables waves to be propagated from the left-to-right, the forward direction, or right-to-left, the backward direction through a system that includes a transmission line that is formed by linking the output of a successive constructive feedback stage to the output of a preceding constructive feedback stage.
  • the feedback stage can in some examples include one, or in other examples, two or more transistors (Q ⁇ and Q2) that cause a travel delay on the feedback path such that the amplitude of the forward wave is increased while the amplitude of the backward wave is reduced.
  • the feedback stage thus generates a feedback signal that includes a time translation and a gain translation.
  • the time translation is advantageously selected such that the gain translation constructively interferes with a forward traveling wave on the transmission line when the feedback signal is routed back to the beginning of the stage.
  • the amount of amplification per stage can be relatively small to prevent oscillation and maintain stability. However, a very large number of stages can be cascaded to provide very high overall gain across a relatively wide frequency band.
  • the feedback stage can also add a frequency translation to add frequency mixing or multiplying (i.e. doubling, tripling, etc.) functionality to a device.
  • an individual stage 200 which can be part of a cascade of shunt-shunt feedback stages, is created across a transmission line 202 as shown in FIG. 2.
  • Y-parameters of the transmission line 202 and a feedback path or stage 204 can be analyzed separately and combined to determine the response of a single stage.
  • the general Y-parameters for a transmission line 202 through a stage 200 are as follows:
  • ⁇ — a +j ⁇ is the propagation constant that includes the attenuation and phase constant along a transmission line 202 having length, /, with a characteristic admittance of Y j .
  • One or more active devices are part of the feedback path or stage 204. These active devices form a transconductor amplifier 206 in the shunt-shunt feedback network to sample the output of the stage 200 at an output (port 2) 210 and feed current back to an input port (port 1) 212 of the stage 200 via the feedback path 204.
  • the feedback path admittance, Y fl , of the transconductor amplifier 206 is represented mathematically as
  • the transconductance is g n while C 1 and C 2 are the input and output capacitive loading of the feedback contributed by the active devices that make up the transconductor amplifier 206.
  • a time delay, or more generally a time translation, T A through the feedback path or stage 204 corresponds with some fraction of the finite transit time of the active device 206 but could be alternatively considered as a frequency-dependent delay through the feedback path 204.
  • the admittance response, Y,, ⁇ g ⁇ , of a single stage 200 can be determined by summing the admittance, Y 1I1111 , of the transmission line 202 from equation (1) and the admittance, Y ⁇ , of the feedback path 204 from equation (2).
  • the mathematical representation of the combined linear behavior of the transmission line and active feedback signal are represented from the linear combination of these constituent parts as follows:
  • FIG. 3 shows a process flow chart 300 illustrating a method according to an implementation of the current subject matter.
  • a propagating signal that includes a forward traveling wave is received at a transmission line outlet port, such as port 210.
  • the propagating signal is sampled at the outlet port 210 by a feedback stage 204.
  • a feedback signal is generated in the feedback stage at 306 to include a time translation and a gain translation.
  • the feedback signal is routed to the inlet port by the feedback stage such that the time translation causes the gain translation to constructively interfere with the forward traveling wave, thereby increasing a forward traveling wave amplitude of the forward traveling wave.
  • the propagating signal can also include a backward traveling wave with which the gain translation to destructively interferes to thereby decrease a backward traveling wave amplitude of the backward traveling wave.
  • the feedback stage and the transmission line can introduce a first and a second phase shift, respectively.
  • the sum of the first and the second phase shifts can advantageously be approximately ⁇ radians or 180 degrees such that the gain translation of the feedback signal is coherently added to the forward traveling wave and coherently subtracted from the reverse traveling wave. While coherent addition and/or subtraction is optimized at 180 degrees, the constructive and/or destructive interference need not be perfect according to the current subject matter.
  • Real devices can include additional impedances, capacitances, and the like that can modify time or frequency translations from modeled values. Such devices are also within the scope of the current subject matter.
  • ⁇ 0
  • C, 0 and C 2 — 0
  • the ⁇ -parameters depend on the voltage gain through the shunt feedback path 204 as well as the phase, ⁇ , through the transmission line 202 and delay, T 1 ,, through the feedback path 204.
  • Constructive interference or coherent addition occurs when the trans conductor amplifier 206 coherently adds energy to the forward traveling wave and provides forward gain.
  • Concurrently, destructive interference or coherent subtraction occurs when the trans conductor amplifier 206 absorbs energy of the backward traveling wave, reducing the wave amplitude.
  • equation (5b) and equation (5c) the conditions on the transmission line phase, ⁇ , and feedback delay, T 1 , are as follows:
  • the phase through the transmission line 202 and the feedback path 204 are advantageously offset by ⁇ /2 radians or 90° at the amplifier center frequency.
  • scattering matrix reduces to the following equation:
  • the backward transmission, J12 is cancelled completely if A 1 , — Vz while the forward transmission, S21, is 2.
  • the forward wave is amplified while providing isolation of the backward traveling wave.
  • the phase conditions in equation (6) and resulting ⁇ -parameters in equation (7) indicate that the stage amplifies through positive feedback of the wave from the output to the input of the stage.
  • the response in equation (7) resembles an ideal lossless transmission line.
  • the forward transmission for Z j — 50 ⁇ has a maximum of 2.2 dB at the center frequency of 100 GHz, while the backward transmission has a minimum of -4 dB.
  • the input and output return loss is identical, i.e. SIl - S22, and decreases to 9.5 dB.
  • S21 does not change appreciably while Sl 2 increases to -2.5 dB.
  • the primary benefit of reducing the characteristic impedance of the transmission line 202 is a reduction of SIl and S22 to -14 dB across the entire frequency range.
  • the matching of the amplifier stage 200 is wideband over the entire 80 GHz of frequency that is graphed in the chart 400 of FIG. 4. As discussed in greater detail below, the matching is also broadband and independent of the center frequency of the amplifier response. This advantage is not typical of cascaded lumped element amplifier designs where matching networks are inherently narrowband.
  • the feedback voltage gain determines the 3 dB bandwidth around the center frequency, f 0 .
  • the bandwidth 502 and the gain-bandwidth product, GBW 504, as a function of the gain, G n , of the single stage 200 is plotted in the chart 500 of FIG. 5.
  • the bandwidth, BW 502 decreases as G a increases. This trend holds in the limit that the transmission line cutoff frequency is large, in this case, twice the center frequency or 2f 0 , compared to the center frequency of the amplifier.
  • a substantive difference between constructive wave behavior and traditional cascaded amplifiers is the feature that the bandwidth in a constructive wave device, according to some implementations of the current subject matter, tends to approach the cut-off frequency of the transmission line 202 at low gain values since the traveling wave stage degenerates to a simple transmission line when the feedback gain is eliminated.
  • the gain of the stage, G 11 depends on the voltage feedback gain, A 1 , per equation (7), and the bandwidth depends on the voltage feedback gain, A 1 ,, per equation (8)
  • the single stage gain, G 33 , and bandwidth, BW can be electronically controlled. Since gain, G 33 , depends on the transconductance, ⁇ n , in the feedback path 204, the feedback gain, A 11 , can be controlled through current biasing of the stage.
  • FIG. 4 Another notable feature of a constructive wave stage, according to implementations of the current subject matter, is wideband impedance matching reflected in the return loss shown in the chart 400 of FIG. 4.
  • Bode-Fano limit For a shunt RC load impedance, the Bode-Fano limit is determined as follows:
  • equation (10) can double to 2;r ⁇ r /G ss with a one-port matching element, such as a shunt inductance added to the shunt RC load, and quadruples for a two-port matching network, such as a series inductance.
  • the right hand side of the inequality in equation (10) is # ⁇ r /2(l - G 1 , '1 ).
  • the Bode-Fano limit can be compared for a lumped element amplifier with a shunt inductance to resonate the device capacitance.
  • the following relation must hold to demonstrates a higher Bode-Fano bound and therefore superior implementation compared with the conventional cascade:
  • the matching limit of devices according to the current subject matter can increase to the intrinsic matching of the feedforward transmission line 202 as the voltage feedback gain, A 0 approaches zero.
  • the chain matrix determines features of the cascaded constructive traveling wave stages to amplify the forward traveling wave and isolate the backward traveling wave over multiple stages.
  • a cascade of N traveling wave stages can be analyzed using chain matrix multiplication according to the following equation:
  • the feedback voltage gain can be nulled since the feedback current is shunted at the input port 212. If the short is placed at the output port 210, the feedback voltage gain can again be nulled because the input of the transconductor amplifier 206 is shorted.
  • the Rollett stability criteria is shown in equation (18) to be extremely insensitive to the voltage feedback gain, A 1 ,.
  • K s generally improves with additional gain.
  • the denominator becomes negative once the feedback voltage gain exceeds one-half (A 1 , > Va), and changes in the gain can result in potential instability.
  • the stability of the cascade is generally further improved because the losses maintain the stability margins over multiple stages.
  • the stability margins improve for a given quality factor, Q 3 as the number of stages increase. Consequently, a long cascade of stages 200 according to the current subject matter can enhance unconditional global stability without sacrificing significant gain.
  • noise factor of the cascade of constructive wave amplifier stages 200 is determined as follows:
  • the noise factor, F is defined approximately as follows: GL F ss Gl - 1
  • the single stage gain in this example is roughly G 11 — 1:33.
  • the minimum noise figure for a HBT in a 0.12 ⁇ m SiGe BiCMOS is roughly 6 dB at 100 GHz. Consequently, the anticipated noise figure, F, for an infinite cascade of traveling wave stages is around 8.9 dB or equivalently a 3 dB noise figure penalty.
  • F the anticipated noise figure
  • Table I shown below, compares a cascaded constructive traveling wave amplifier according to implementations of the current subject matter with currently available amplifier techniques.
  • the current subject matter is distinct from distributed amplifiers because the cascaded traveling wave provides a passband response at a center frequency determined by the feedforward transmission line length and feedback delay.
  • the current subject matter can provide cascaded gain and is not restricted to distributed amplifier gain limits. However, the gain per stage is generally low due to the trade-off between acceptable return loss and single stage gain. Both the gain and the bandwidth of the response are controlled through the voltage feedback gain. Consequently, implementations of the current subject matter can trade maximum gain for a wideband response.
  • the matching behavior of the stages in the current subject matter can maintain impedance matching outside of the bandwidth of the gain response.
  • the stability of a stage according to the current subject matter can improve with transmission line losses and can be relatively insensitive to voltage feedback gain variations. A moderate noise figure penalty can be provided due to the low maximum achievable gain per stage.
  • the current approach is capable of operating with only a one- quarter wave transmission line per stage and a small overall area consumption. With respect to cascaded narrowband techniques, the current approach may consume more power for a given overall gain but provides advantageous broadband impedance matching characteristics. Table I COMPARISON OF CONVENTIONAL AMPLIFIER CIRCUIT TECHNIQUES
  • a cascaded traveling wave topology can include amplification of the forward traveling wave and opacity, or isolation, of the backward traveling wave.
  • a shunt-shunt active feedback system can be provided with a HBT device in a 0.12 ⁇ m SiGe BiCMOS technology.
  • the active feedback at each stage 700 is based on a cascade of the common emitter device 704 , and the emitter follower devices 706.
  • the feedback path 702 advantageously provides high input and output impedance to prevent loading of the transmission line 712 and further introduces a delay T d that promotes constructive interference with the forward propagating wave 714 and destructive interference with the backward traveling wave 716.
  • the emitter follower 706 increases the collector-emitter voltage of the common emitter device 204 to enhance the transconductance of the feedback stage 702 and reduces capacitive loading of the shunt-shunt feedback on the input port 720 and output port 722 of the transmission line 712. Additionally, the base-collector capacitance of the common emitter device 204 terminates into a high-impedance node between the emitter follower 706 and degenerated common emitter 710.
  • the emitter follower 706 advantageously has, in some implementations, a large input impedance to minimize loading of the capacitance of the base-emitter at the stage output 722.
  • a HBT implementation can offer high impedance and transconductance with a target frequency, / i3 of 200 GHz and a maximum frequency, /, /a ⁇ , of 280 GHz. Consequently, a feedback amplifier using a series of stages 700 can satisfy the desired delay requirements through the feedback path 702 at 100GHz.
  • FIG. 7B shows additional detail of an implementation of an active feedback path, such as stage 702.
  • the mid-band transconductance, ⁇ , of the feedback path 702 can be obtained from:
  • R E represents the resistance value of resistor 710 and r 0 represnets the resistance value of resistor 724.
  • the feedback gain can be obtained by degeneration through R E 710.
  • the common emitter device 204 is biased for a transconductance, g u , of 60 mS while R E is 70 ⁇ . Therefore, the feedback gain is ideally A 1 , — 0:27 for a single stage gain of 1.37. Results of a simulation of the transconductance and phase and group delays through the active feedback path or stage 702 of FIG. 7B are shown in the chart 800 of FIG. 8. As shown in FIG.
  • the magnitude of the transconductance 802 rolls off gradually with frequency and achieves a degenerated transconductance of 10.5 mS at lOOGFIz.
  • the phase delay 804 of the feedback stage also decreases with increasing frequency because of the impact of capacitance between the emitter follower and common emitter stages and is 2.3 ps at 100 GHz.
  • the feedback group delay 806 is also plotted and remains around 2.7ps across the frequency range.
  • the input impedance at the base of the common emitter stage 704 may be determined as follows:
  • C k;J — C M /[1 + gj& ⁇ and Z 0 / 2 is estimated as the impedance at the collector 720 of the common-emitter 704 transistor.
  • the impedance at base of the emitter follower stage 706 may be determined as follows:
  • the total feedback input impedance at the output port (port 2) 722 of the amplifier stage 700 may be determined as follows:
  • the impedance includes resistive and capacitive components.
  • the capacitance at the emitter 726 of the emitter follower stage 706 contributes to negative resistance at the input impedance port 720.
  • the input resistance remains positive because of the relatively high base resistance for the SiGe HBT and contribution of the resistive degeneration 710.
  • the base-collector capacitance of the emitter follower transistor 706 directly loads the input of the feedback stage 702.
  • the output impedance of the feedback stage 702 can then be determined as follows:
  • the output impedance has a relatively large output resistance because of the emitter degeneration 710 and the capacitance seen at this node is limited to the base-collector capacitance of the common emitter 704 transistor.
  • the matching of the feedback stage 702 when shunted by the feedforward transmission line is simulated in the Smith chart 900 of FIG. 9.
  • the input port impedance (SIl) 902 rotates clockwise around the center 904 of the Smith chart as expected from the analysis of the input and output impedance in equation (9a) and equation (9b). From 40 to 140GHz, the magnitude of the input port impedance line 902 remains relatively constant due to the high impedance in equation (25).
  • the output port impedance (S22) 906 also rotates around the center 902 of the Smith chart but illustrates a larger impedance mismatch because of the frequency dependent resistive and admittance components in equation (24).
  • the measured matching 910 for both the input and output ports is compared to the analytical prediction in FIG. 9 at 100GHz and provides reasonable prediction of the matching behavior.
  • the gain sensitivity of a traveling wave stage can be subject to variation of the active devices in the feedback circuit and can be determined from equation (7) as follows:
  • the sensitivity of the feedback circuit to process, voltage and temperature (PVT) variations is mitigated by several factors.
  • the high degeneration of the common emitter amplifier can reduce the PVT variations of the common emitter transistor.
  • the high-frequency, high-current transconductance-limiting mechanisms of the SiGe HBT also improve the robustness of the circuit.
  • the sensitivity of the feedback circuit to PVT can be primarily due to changes in the transit time of the device and, consequently, the time delay through the feedback path as well as the transconductance of the feedback path.
  • the simulated sensitivity to process and voltage variations of the forward transmission coefficient S21 and the reverse transmission coefficient Sl 2 are illustrated in the charts 1000 and 1050 of FIG. 1OA and FIG. 1OB, respectively.
  • the process corners of the HBT model are varied between + 1 ⁇ , where ⁇ is the standard deviation of the device operation.
  • the single stage gain reduces in amplitude and center frequency for the slow corner and increases in amplitude for the fast corner. Shifts in the center frequency and gain occur because the trans conductance and the transit time of the device change inversely with respect to one another.
  • the lower center frequency for the slow corner is expected because the device contributes to more total phase through the feedback loop. Overall the gain changes by 8% and the center frequency by 10% over these process corners.
  • the local stability of the stage can be inferred from FIG. 1OA and FIG. 1OB considering the basic guideline that the magnitude of 521 and Jl 2 be less than unity. Across both the process and the voltage corners, this condition can be maintained and the local as well as global stability can be ensured. Finally, the cascading constructive traveling wave amplifier effectively combines the traveling wave over multiple amplifier stages and, in the presence of PVT variations, stage-to-stage variations will tend to be averaged over multiplied amplifier stages.
  • FIG. 11 shows an overall circuit implementation for a twelve-stage amplifier 1100 with uniform sizing for each stage 700, the biasing of the collector for the common emitter 704 of each stage 700 is provided through the RF input path while the biasing of the collector for the emitter follower 706 is provided through d.c. pads (not shown).
  • a current bias for the emitter follower 706 can be provided through an on-chip mirror.
  • a chip microphotograph of an amplifier chip 1200 according to an implementation is shown in FlG. 12 and measures 330 ⁇ m by 1000 ⁇ m including pads.
  • the area of a single stage 700 measures 160 ⁇ m by 60 ⁇ m.
  • the transmission line 1202 is a shielded microstrip with characteristic impedance of 45 ⁇ .
  • the inset in the upper left of FIG. 11 illustrates the cross-section of the meandering transmission line 1202 and its side shields 1204.
  • the transmission lines 1202 are shielded microstrip lines 1204 where the side shields 1204 are shared between neighboring cells 700. Electromagnetic simulation using HFSS verify that potential coupling between stages 700 through this side shield 1204 was under -20 dB. In particular, the input return loss improves by 5 dB from the single wire model with the shared side shield 1204.
  • the shielded coplanar transmission line 1202 meanders through a length of 220 ⁇ m.
  • the transmission line 1202 length is shorter than the theoretical quarter wave transmission line because of the capacitive loading of the active feedback on the transmission line structures.
  • interstage transmission lines are located between stages and are 40 ⁇ m long. These interstage transmission lines are provided to maintain isolation of the active devices 1206 between stages.
  • J-parameters for the 12-stage amplifier 1200 shown in FIG. 12 were characterized with an Agilent E8361A two-port network analyzer and N5260A mm-wave controller using Cascade ACPIlO-LW GSG probes.
  • the measured J-parameters are shown in the chart 1300 of FIG. 13 for the twelve stage amplifier 1200.
  • the forward and backward transmission illustrates amplification of the forward wave and isolation of the backward wave.
  • the J21 measurement 1302 reaches a gain of 26 dB at 99 GHz over a 3 dB bandwidth of 12 GHz between 93 and 106 GHz.
  • the traveling wave is amplified through twelve cascading stages, the average gain of each stage is 2.2 dB as compared to 2.3 dB from the analysis and design.
  • the S21 measurement 1302 is compared to circuit simulations 1304 performed over twelve stages and demonstrates accurate prediction of the nominal gain and bandwidth.
  • the S12 measurement 1306 is -30 dB at 99 GHz and remains relatively constant at low frequencies and rolls-off above 99 GHz.
  • the simulation 1310 for 512 also agrees reasonably well with the measurement 1306.
  • the measured input return loss 1312 (J 1 Il) across the 3 dB bandwidth is better than -15 dB while the measured output return loss 1314 (522) is better than -12 dB. Both the input and output return loss remain under -10 dB over the measured 40-110 GHz range.
  • the stability at the nominal 26 dB operating point is illustrated in the chart 1400 of FIG. 14.
  • the measured K 1402 is greater than one and the measured I ⁇ I 1404 is less than one at all frequencies indicating unconditional stability for the amplifier 1200.
  • the stability margins are improved by reducing the gain of each stage and, therefore, the overall gain. Reducing the bias current, I b ⁇ as , reduces the emitter follower gain and, hence, A 11 .
  • a gain of 23dB, or roughly 1.92dB per stage, is also graphed in FIG. 14 to illustrate the improved stability margins through increased K 1406 and reduced
  • Control of the gain through the emitter follower bias is predicted from the analysis to trade-off with bandwidth.
  • the forward transmission is plotted in the chart 1500 of FIG. 15A and the reverse transmission is plotted in the chart 1550 of FIG. 15B for bias current, I fea[ , ranging from 0 mA to 2.72 mA.
  • I fea[ bias current
  • the 521 and 512 are reciprocal and are around -6 dB, a loss of 0.5 dB per stage, from 60 to 110 GHz.
  • the forward transmission gain increases to a maximum gain is 35.5 dB while the backward transmission gain decreases to -3OdB.
  • the amplifier Under the highest gain condition, the amplifier is conditionally stable but does not oscillate under 50 ⁇ matching at the input and output.
  • the input return loss is plotted in the chart 1600 of FIG. 16A and the output return loss is plotted in the chart 1650 of FIG. 16B as a function of the bias current.
  • the matching is relatively insensitive to the bias and remains under 10 dB over most operating conditions.
  • the output return loss decreases to 8 dB above 99 GHz. At lower frequencies, the return loss does not change dramatically with the bias current.
  • the gain 1702, bandwidth 1704, and gain-bandwidth product 1706 of the example 12- stage amplifier are plotted in the chart 1700 of FIG. 17 as a function of total power consumption.
  • the gain 1702 is controlled through the bias current, l h ⁇ s , and causes proportionally higher power consumption.
  • the bandwidth 1704 of the amplifier 1200 decreases from 50 GHz to 12 GHz.
  • the bandwidth 1704 does not change significantly for bias currents above 1.8mA.
  • the bandwidth is 12GHz at 26dB gain
  • the bandwidth 1704 at a 35dB gain is 12.1GHz.
  • the achievable gain-bandwidth product 1706 of the twelve stage amplifier 1200 is 260 GHz at 26 dB and increases to 740 GHz at 35 dB.
  • the measured 1802 and simulated 1804 noise figures for the amplifier 1200 is plotted in the chart 1800 of FIG. 18.
  • the noise figure was measured using a W-band noise source which specifies an excess noise ratio in increments of 5 GHz from 75 GHz to 110 GHz.
  • the output noise spectrum of the amplifier 1200 was downconverted using a Millimeter Wave W-band mixer to an IF frequency between 4 GHz and 10 GHz.
  • the LO generated from an RF synthesizer was tripled to 90GHz using a Millimeter W-band frequency tripler and filtered with a 90 to 100 GHz bandpass filter.
  • a two-stage IF amplifier provided 24 dB of gain to compensate the conversion loss of the mixer.
  • the noise figure of the down-conversion stage was initially calibrated with a spectrum analyzer. Additionally, cable and probe losses of 7.5dB were also de-embedded from the noise figure measurement.
  • the noise figure test setup was compared to a QuinStar QLW series 92- 98GHz amplifier and found to have less than 0.2 dB error from the manufacturer specification. Simulation 1804 of the twelve stage amplifier predicts a minimum noise figure of 12.2 dB but the measured noise figure 1802 is even lower — as low as 10.8 dB at 85 GHz. At 99 GHz, the measured noise figure 1802 is 12.3 dB.
  • the gain compression is plotted in the chart 1900 of FIG. 19 under the nominal biasing conditions using a power meter with a power sensor.
  • the input power is swept by controlling the attenuation of the network analyzer millimeter wave head.
  • the output-referred IdB compression point for the amplifier was recorded at -O.ldBm and the input-referred compression point is approximately -25dBm.
  • the amplifier 1200 was measured at a nominal current consumption of 32.5 mA for the emitter follower stages from a 2.2V supply and 21 mA for the common emitter stages each from a 2V supply.
  • the emitter follower bias current is provided through an external current bias.
  • the total power consumption was 114mW, which guarantees unconditional stability at 26 dB and provides a maximum achievable gain of 35 dB.
  • a system 2000 can include a cascaded constructive wave amplifier stage 700 as described herein coupled to a Darlington amplifier 2002.
  • a signal input at the input port 2004 of the Darlington amplifier 2002 is amplified according to the response of the Darlington amplifier 2002 and output via the output port 2006.
  • the Darlington amplifier 2002 provides good gain on the S21 band at lower frequencies as shown in the chart 2100 of FIG. 21 A but has very Kttle impact at higher frequencies.
  • a cascading constructive wave amplifier according to the current subject matter has complementary performance as shown in the chart 2150 of FIG. 21B.
  • FIG. 22A and FIG. 22B Another implementation of the current subject matter is shown in FIG. 22A and FIG. 22B.
  • FIG. 22A shows a half-duplex radio transceiver 2200 that includes a duplexer 2202 to allow switching from a transmit path 2204 between a power amplifier (PA) 2206 and an antenna 2210 and a receive path 2212 between the antenna 2210 and a low noise amplifier (LNA) 2214.
  • the duplexer 2202 typically incurs substantial losses.
  • a device 2250 can include multiple bidirectional stages 2252 that are cascaded to provide the desired two-way gain without the need for a duplexer.
  • Each bidirectional stage 2252 includes a transmission line segment 2254 between a first port 2256 and a second port 2260.
  • Each bidirectional stage 2252 further includes a forward feedback path 2262 and a backward feedback path 2264, only oe of which is in operation at a time.
  • the forward feedback paths are active.
  • the reverse direction for example from port 1 to port 2 in the device 2250
  • the forward feedback paths are active.
  • a control signal provided by a control signal generator can be used to turn on or off either the forward feedback paths 2262 or the backward feedback paths 2264.
  • the forward feedback path 2262 is disabled and traveling waves are amplified from port 2 to port 1.
  • the forward feedback path 2262 can be enabled and the backward feedback path 2264 disabled allowing for traveling wave amplification from port 1 to port 2.
  • Each of the forward feedback path 2262 and the backward feedback path 2264 of each bidirectional stage 2252 can be a feedback stage according to the current subject matter.
  • each of the forward feedback path 2262 and the backward feedback path 2264 can be similar to feedback stage 702 in FIG. 7.
  • the backward feedback path 2264 is reversed relative to the forwards feedback path 2262.
  • circuitry according to the current subject matter are described herein with reference to amplifier circuits, other circuits can also use this subject matter.
  • the subject matter described herein may be used for 60GHz video transmission (e.g., SiBeam, Panasonic, MediaTek), 77GHz radar applications (e.g., MaCOM, Infineon), and 80-90GHz point-to-point wideband communication (e.g., Gigabeam).
  • the subject matter described herein may be extremely useful to provide systems using silicon-based chip technologies rather than the more expensive InP and GaAs technologies.
  • the subject matter described herein can be embodied in systems, apparatus, methods, and/or articles depending on the desired configuration.
  • various implementations of the subject matter described herein can be realized in digital electronic circuitry, integrated circuitry, specially designed application specific integrated circuits (ASICs), computer hardware, combinations thereof, and the like.
  • ASICs application specific integrated circuits
  • NMOS n-type metal- oxide-semiconductor
  • PMOS p-type metal-oxide-semiconductor

Abstract

Appareil et procédé associé utilisant une ligne de transmission transportant un signal se propageant entre un point d’entrée et un point de sortie. Le signal se propageant peut comprendre une onde progressive directe et éventuellement une onde progressive inverse. Un étage de rétroaction échantillonne le signal se propageant au niveau du point de sortie, génère un signal de rétroaction incorporant une translation temporelle et une translation de gain dans l’énergie de rétroaction, et achemine le signal de rétroaction jusqu’au point d’entrée de telle sorte que la translation de gain interfère de façon constructive avec l’onde progressive directe pour en accroître l’amplitude.
PCT/US2009/058172 2008-09-24 2009-09-24 Dispositif à ondes progressives à rétroaction constructive et procédé associé WO2010059290A2 (fr)

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US10367463B2 (en) 2016-06-09 2019-07-30 Keysight Technologies, Inc. Variable gain distributed amplifier systems and methods
US10803220B2 (en) 2017-06-05 2020-10-13 International Business Machines Corporation Transient and AC simulations with traveling wave probe circuit
US10985699B2 (en) 2018-03-02 2021-04-20 North Carolina A&T State University Differential constructive wave oscillator device
CN114553155B (zh) * 2022-04-22 2022-08-16 成都嘉纳海威科技有限责任公司 一种覆盖基频的超宽带射频放大器

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EP0600548A1 (fr) * 1992-12-03 1994-06-08 Philips Electronics Uk Limited Circuit amplificateur à micro-onde à forme de construction compacte utilisable en cascade
US20020014920A1 (en) * 2000-07-14 2002-02-07 Masami Ohnishi High frequency power amplifier
US20060055464A1 (en) * 2002-11-19 2006-03-16 Koninklijke Philips Electronics , N.V. Travelling-wave amplifier
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EP0600548A1 (fr) * 1992-12-03 1994-06-08 Philips Electronics Uk Limited Circuit amplificateur à micro-onde à forme de construction compacte utilisable en cascade
US20020014920A1 (en) * 2000-07-14 2002-02-07 Masami Ohnishi High frequency power amplifier
US7142052B2 (en) * 2001-03-27 2006-11-28 Qinetiq Limited Travelling wave amplifiers
US20060055464A1 (en) * 2002-11-19 2006-03-16 Koninklijke Philips Electronics , N.V. Travelling-wave amplifier

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