WO2009148278A2

WO2009148278A2 - Channel estimation and equalization method and system

Info

Publication number: WO2009148278A2
Application number: PCT/KR2009/002991
Authority: WO
Inventors: So Ra Park; Se Bin Im; Hyung Jin Choi; Sung Ik Park; Yong Tae Lee; Jong Soo Lim; Soo In Lee
Original assignee: Electronics And Telecommunications Research Institute; Sungkyunkwan University
Priority date: 2008-06-04
Filing date: 2009-06-04
Publication date: 2009-12-10
Also published as: KR20090126420A; WO2009148278A3; EP2289183A2; EP2289183A4; WO2009148278A8; KR100964396B1

Abstract

Disclosed is a channel estimation and equalization method and system. The method inserts a pilot signal in the time domain into a guard interval, uses a hard decision signal in the frequency domain to estimate a frequency response of the channel, and equalizes a radio channel distortion through the estimated channel value.

Description

TITLE OF THE INVENTION

CHANNEL ESTIMATION AND EQUALIZATION METHOD AND SYSTEM

BACKGROUND OF THE INVENTION

(a) Field of the Invention

The present invention relates to a channel estimation and equalization method and system.

(b) Description of the Related Art The orthogonal frequency division multiplexing (OFDM) system safely provides various multimedia services to a user when a receiver moves. Attention to the OFDM system has increased all over the world because of the merit. Hence, the OFDM system is anticipated to be applicable to various fields such as vehicles, high-fidelity (HiFi) home theaters, and mobile receivers. In Korea, the Ministry of Information and Communication has determined the Eureka-147 digital audio broadcasting (DAB) for the terrestrial DMB (T-DMB) standard in 2002, and the terrestrial digital broadcasting systems have been provided and various types of software have been developed. Accordingly, services started to be provided to users in December 2005. As the domestic commercialization and outreach of the T-DMB service to foreign countries have accelerated, it has become required for the T-DMB system to generate plans for generating comparative advantage over the digital video broadcasting-handheld (DVB-H) system, the media forward link only (MediaFLO) system, or the integrated service digital broadcasting terrestrial (ISDB-T) system that are competitive with the T-DMB system concerning skills and services. One of demerits of the T-DMB system compared to other systems is its low data rate. The advanced T-DMB (AT-DMB) solves this problem to upgrade the T-DMB service quality. In addition, when a high-quality service is exemplified, an AT-DMB transmitter separates a standard-definition (SD) video source into high priority (HP) video signals and low priority (LP) video signals to encode them, and provides them to the receiver through a single T-DMB channel by using a layered modulation skill. When the layered modulation skill is applied, error protection performance for the high priority video signals and the low priority video signals are realized differently.

Therefore, the conventional T-DMB receiver restores the high priority video signal from the AT-DMB signals transmitted by the transmitter, and uses the quarter video graphics array (QVGA) quality T-DMB service that is similar to the existing quality. The AT-DMB receiver restores the high priority video signals and the low priority video signals included in the AT-DMB signals transmitted by the transmitter to use an SD quality video service.

However, the AT-DMB system may substantially degrade the performance of receiving data signals that are additionally transmitted according to the layered modulation scheme in the real radio channel condition. Also, the signals may be easily distorted by the channel.

The above information disclosed in this Background section is only for enhancement of understanding of the background of the invention and therefore it may contain information that does not form the prior art that is already known in this country to a person of ordinary skill in the art. SUMMARY OF THE INVENTION

The present invention has been made in an effort to provide a channel estimation and equalization method and system for increasing data rates by reducing channel estimation errors when a receiver moves at a high speed.

An exemplary embodiment of the present invention provides a system for estimating and equalizing a channel of a signal, including: a serial/parallel converter for dividing the signal into a guard interval and an useful data signal, and outputting the signals; a first channel estimator for receiving the guard interval, estimating an impulse response of the channel, and outputting an impulse response value; a channel equalizer for equalizing channel distortion of the useful data signal by using the impulse response value output by the first channel estimator, and outputting an equalized signal; and a second channel estimator for receiving the useful data signal and the equalized signal output by the channel equalizer, estimating a frequency response to the useful data signal, and outputting a frequency response value.

Another embodiment of the present invention provides a method for estimating and equalizing a channel of a signal, including: converting the signal into a guard interval and a useful data signal; estimating an initial impulse response value of the channel through the guard interval, and outputting an impulse response value; estimating a frequency response on the channel through the useful data signal, and outputting a frequency response value; correcting the output impulse response value and outputting the impulse response value; and equalizing a channel of the signal based on the corrected impulse response value, and outputting an equalized signal. BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows an AT-DMB system according to an exemplary embodiment of the present invention. FIG. 2 shows a configuration diagram of a transmitting system according to an exemplary embodiment of the present invention.

FIG. 3 to FIG. 6 show a constellation according to an exemplary embodiment of the present invention.

FIG. 7 and FIG. 8 show a frequency domain pilot subcarrier arrangement according to an exemplary embodiment of the present invention.

FIG. 9 shows a configuration diagram of a receiving system according to an exemplary embodiment of the present invention.

FIG. 10 shows a time domain subcarrier arrangement according to an exemplary embodiment of the present invention. FIG. 11 shows a frequency domain subcarrier arrangement according to an exemplary embodiment of the present invention.

FIG. 12F and FIG. 13 show performance according to symbol positions according to an exemplary embodiment of the present invention.

FIG. 14 and FIG. 15 show performance according to the received signal power according to an exemplary embodiment of the present invention.

FIG. 16 and FIG. 17 show performance according to the moving speed of a receiver according to an exemplary embodiment of the present invention.

FIG. 18 and FIG. 19 show performance according to the number of repeated estimation according to an exemplary embodiment of the present invention.

DETAILED DESCRIPTION OF THE EMBODIMENTS

In the following detailed description, only certain exemplary embodiments of the present invention have been shown and described, simply by way of illustration. As those skilled in the art would realize, the described embodiments may be modified in various different ways, all without departing from the spirit or scope of the present invention. Accordingly, the drawings and description are to be regarded as illustrative in nature and not restrictive. Like reference numerals designate like elements throughout the specification. Throughout the specification, unless explicitly described to the contrary, the word "comprise" and variations such as "comprises" or "comprising" will be understood to imply the inclusion of stated elements but not the exclusion of any other elements.

In the specification, a receiver may indicate a mobile station (MS), a mobile receiver (MR), a mobile terminal (MT), a subscriber station (SS), a portable subscriber station (PSS), user equipment (UE), and an access receiver (AT), and may include partial or entire functions of the mobile station, the mobile receiver, the subscriber station, the portable subscriber station, the user equipment, and the access receiver. A channel estimation and equalization method according to an exemplary embodiment of the present invention will be described with reference to accompanying drawings. A mobile communication system for channel estimation and equalization will be described by using an AT-DMB system as an example, to which an exemplary embodiment of the present invention is not restricted.

FIG. 1 shows an AT-DMB system according to an exemplary embodiment of the present invention.

As shown in FIG. 1, a transmitting system of an AT-DMB system according to an exemplary embodiment of the present invention includes a video/audio source encoder 10, a high priority modulator 20, a low priority modulator 30, a subcarrier mapper 40, an OFDM modulator 50, and a transmitter 60. A receiving system includes a basic quality video receiver 70 and a high quality video receiver 80. The video/audio source encoder 10 encodes an input video source and an audio source into high priority (HP) signals and low priority (LP) signals having digital binary codes.

The high priority modulator 20 and the low priority modulator 30 code and multiplex channels of respective bits of the high priority signals and the low priority signals that are encoded and output by the video/audio source encoder 10, and an adder outputs the multiplexed signals as a signal. The output signal is input to the subcarrier mapper 40 to be mapped into a predetermined constellation. In this instance, the high priority modulator 20 modulates high priority signals by a predetermined data modulation rate. The low priority modulator 30 modulates low priority signals by a predetermined data modulation rate based on the high priority signals. Here, the predetermined data modulation rate will use quadrature phase shift keying (QPSK) constellation as an example, but the embodiment is not restricted thereto.

The OFDM modulator 50 modulates the mapping signal to OFDM signal. The transmitter 60 amplifies the OFDM signal with appropriate power by up-converting and transmits the signals. The basic quality video receiver 70 of the receiving system receives and demodulates the high priority signals. The high quality video receiver 80 receives the high priority signals and the low priority signals and demodulates them.

The transmitting system will now be described with reference to FIG. 2.

FIG. 2 shows a configuration diagram of a transmitting system according to an exemplary embodiment of the present invention.

As shown in FIG. 2, the high priority modulator 20 of the transmitting system includes a first channel encoder 21 , a first QPSK mapper 22, a symbol delay unit 23, and a first phase rotator 24, and the low priority modulator 30 includes a second channel encoder 31 , a second QPSK mapper 32, and a second phase rotator 33. The OFDM Modulator 50 includes an inverse fast

Fourier transformer (IFFT) 51 , a pilot signal generator 52, and a parallel/serial converter 53. The transmitter 60 includes an RF unit 61 and a base station antenna 62.

The first channel encoder 21 and the second channel encoder 31 respectively receive the high priority signals and the low priority signals having digital binary codes from the video/audio source encoder 10, and encode channels in order to correct transmission errors in preparation for the case in which the digital binary codes may generate transmission errors because of the radio channel. Here, the first channel encoder 21 and the second channel encoder 31 will be described to be included in the high priority modulator 20 and the low priority modulator 30, and are not restricted thereto. The first QPSK mapper 22 QPSK maps the high priority data that are channel encoded by the first channel encoder 21 on four phases. The first phase rotator 24 rotates the phase of the QPSK mapped signal by using a delayed signal for an OFDM symbol interval. Simultaneously, the first phase rotator 24 transmits phase information of the mapped signal to the second phase rotator 33 of the low priority modulator 30.

The second QPSK mapper 32 QPSK maps the channel encoded low priority data on four phases. The second phase rotator 33 rotates the phase of the signal mapped by the second QPSK mapper 32 by using phase information transmitted through the first phase rotator 24.

The adder 25 adds the output signal of the high priority modulator 20 and the output signal of the low priority modulator 30, and outputs them as a layer modulation signal to the subcarrier mapper 40. In this instance, the constellations for the output signal of the high priority modulator 20 are shown in FIG. 3 and FIG. 4, and the constellations for the output signal of the low priority modulator 30 are shown in FIG. 5 and FIG. 6.

FIG. 3 to FIG. 6 show constellations according to an exemplary embodiment of the present invention.

FIG. 3 shows the constellation of the high priority signal of the even OFDM symbol according to an exemplary embodiment of the present invention, and FIG. 4 shows the constellation of the high priority signal of the odd OFDM symbol. FIG. 5 shows the constellation of the low priority signal of the even OFDM symbol according to an exemplary embodiment of the present invention, and FIG. 6 shows the constellation of the low priority signal of the odd OFDM symbol.

The subcarrier mapper 40 shown in FIG. 2 inputs the input layered modulation signal to the IFFT 51 in order to insert the layered modulation signal into a predetermined subcarrier position within the range of N_used used

subcarriers. In this instance, the layered modulation signal ^{u \} J j_{nput to}

the IFFT 51 is expressed in Equation 1 , and the output signal x^(m)( ^vn ^}) of the

IFFT 51 is expressed in Equation 2. (Equation 1)

Here, ^HP represents the k-th high priority modulation signal of

X^{m~l)(k) the m-th OFDM symbol, and HP ^\ > indicates the k-th high priority

modulation signal of the (m-1)-th OFDM symbol. Also, LP V ^/ _rep_resent_s the k-th low priority modulation signal of the m-th OFDM symbol. (Equation 2)

1 ^N used ^'I χ^im) W = 77 Σ ^Xu^m) (*) ^• exp {j2πn(k + Ic₀)I N) N *=°

Here, N shows the IFFT dimension, and ko represents the first subcarrier position that is the reference within the used subcarrier range. The lower case u indicates the used subcarrier.

The pilot signal generator 52 generates a pilot signal c(n) by passing the constant amplitude zero autocorrelation code (CAZAC) sequence C(k) having the length N_c,_Used through the IFFT having the dimension Nc. Here, the CAZAC sequence C(k) is expressed in Equation 3, and the pilot signal generated by the pilot signal generator 52 is expressed in Equation 4. (Equation 3)

(Equation 4)

Here, ^ is a factor for controlling power of the pilot signal. In order to prevent an increase of power caused by inserting the pilot signal in the

exemplary embodiment of the present invention,

p ^ = so that the average power of the pilot signal may correspond to the average power of the time domain OFDM symbol.

The parallel/serial converter 53 receives the pilot signal generated by the pilot signal generator 52 and the OFDM signal by the IFFT 51 and outputs an

OFDM signal by performing a parallel/serial conversion process.

The RF unit 61 converts the OFDM signal into a radio frequency (RF) signal so that the RF signal may be transmitted to the receiver through the base station antenna 62. A frequency domain pilot subcarrier arrangement when transmitting signals through the above-described transmitter will now be described with reference to FIG. 7 and FIG. 8.

FIG. 7 and FIG. 8 show a frequency domain subcarrier arrangement according to an exemplary embodiment of the present invention. FIG. 7 shows a subcarrier arrangement for a layered modulated data signal according to an exemplary embodiment of the present invention, and FIG. 8 shows a subcarrier arrangement for a pilot signal generated by a CAZAC sequence according to an exemplary embodiment of the present invention. The pilot signal occupies the same bandwidth as the data signal, but has a greater subcarrier interval than the data signal, and its ratio is set to be N/N_c as shown in FIG. 8.

The first subcarrier of the data signal is given as k₀. The first subcarrier of the pilot signal is given as k₀N_c/N in consideration of the reduced number of the subcarriers. For reference, when the subcarrier interval is increased to N/Nc in the frequency domain, the length of the signal is reduced by N_c/N that is an inverse number of the subcarrier interval in the time domain.

A receiving system according to an exemplary embodiment of the present invention will now be described in detail with reference to FIG. 9.

FIG. 9 shows a configuration diagram of a receiver according to an exemplary embodiment of the present invention.

As shown in FIG. 9, the receiving system includes an RF unit 100, a serial/parallel converter 110, a first channel estimator 120, a second channel estimator 200, a first fast Fourier transformer (FFT) 130, a second FFT 140, a high priority demodulator 180, a first parallel/serial converter 190, a channel equalizer 150, a low priority demodulator 160, and a second parallel/serial converter 170. Here, the first channel estimator 120 includes a pilot signal extractor 121 , a channel impulse response estimator 122, and a first impulse response corrector 123. The second channel estimator 200 includes a third FFT 205, a second impulse response corrector 204, an inverse fast Fourier transformer (IFFT) 203, a channel frequency response estimator 202, and a hard decision unit 201.

The RF unit 100 converts the signal received through the base station antenna into a digital signal, and the serial/parallel converter 110 transforms the digital signal into an OFDM symbol that is a parallel layered modulation signal

and outputs it. In this instance, m-th OFDM symbol ^{y v} ' is expressed in Equation 5.

(Equation 5)

h^(m)(n) Here, ^v ' is a time domain impulse response of a channel, and

^ ' is additive white Gaussian noise (AWGN) in the time domain. L_h represents a number of delay paths in the multipath channel environment, and τ i is the l-th delay path.

Also, the serial/parallel converter 110 divides the OFDM symbol into a

guard interval

^{GI ~} ' and a useful data signal V^(IB)(Λ), (0 ≤ n < N) v ' . In this instance, the guard interval divided from the

OFDM symbol is input to the first channel estimator 120, and the useful data signal is input to the second Fourier transformer 140.

The pilot signal extractor 121 in the first channel estimator 120 receives

the guard interval and extracts a used pilot signal ^{y p v J} from the guard interval as expressed in Equation 6. (Equation 6)

= /">(„ - N₀₁ + N_C>OI), (0 ≤ n < N_c)

The channel impulse response estimator 122 estimates an initial impulse

response h^{m\ ^vn\ ^y' ( V0 ≤ n < N_c ^c ) / of . a ch . anne ,l t .h. roug _hh _t t_hhe used . p .i,lo _tt signal extracted by the pilot signal extractor 121 as expressed in Equation 7. (Equation 7)

When the channel impulse response is estimated as expressed in Equation 7, an estimation error is generated because of the limited signal characteristic of the noise and bandwidth. Therefore, the first impulse response corrector 123 corrects the error of the estimated impulse response as expressed in Equation 7. Since the process for correcting the impulse response value is performed for each OFDM symbol independently, the superscript (m) representing channel impulse response estimation of the m-th OFDM symbol in the equation developing process will be omitted for ease of description.

In order to correct the impulse response value, a position of the impulse response value of the channel having the maximum power is selected as expressed in Equation 8.

(Equation 8)

|2

«£L = mκ|A(«)| , (0 < » < JV_c)

Here, the superscript [1] represents the number that is selected first in the process for selecting the position having the consecutive maximum power. When the channel impulse response estimator 122 selects the position of the impulse response value of the channel having the maximum power as

shown in Equation 8, the initial impulse response value of h( ^vή ^}) is corrected with reference to the impulse response value of the first maximum power position as expressed in Equation 9. Here, g'(n) is defined as Equation 10, and

h^m(ή) c^{\ \ J} represents the first result value of the first error correction.

(Equation 9)

AJ, = A(«) - Λ(«H) ^• g'(n - nil), (« ≠ «il )

(Equation 10) g'(n) = g(n)/g(0) g(n) = subcarrier index

ise

In Equation 10, S_usecι subcamer index indicates a used subcarrier index set.

After the impulse response value is corrected, the position of the i-th maximum power impulse response value is selected. The position is selected by using Equation 11.

(Equation 11)

(n ≠ n:^[ l, l ≤ iκ ι - 1, 2 < i ≤ L_c)

The impulse response value of h ^cυl~ιH ^vn ^mmax )^/ is corrected with reference to the position of the i-th maximum power impulse response value that is selected by using Equation 11. In this instance, the impulse response value is corrected by using Equation 12. (Equation 12) / a V^Λ*max "max )^•> V c /

Here, the maximum value L_c of i is set to be L₀ ≥ U, and the process for selecting the position of the i-th maximum power impulse response value and

[i] the process for correcting the impulse response value ooff ^ccll ^ ^v ' m^ma^ax^x .^/ are repeated until the value i reaches L_c.

The resultant value h ^c{lL<Hn ^mmax ) that is acquired by repeated performance is a primary error correction value, and secondary error correction is performed through the primary error correction value. In detail, the error of

the initial impulse response ⁿ\ⁿ) is re-corrected with reference to the primary h^[LΛ(n^[ι] ) error correction resultant value ^{cλ v max /} as expressed in Equation 13.

(Equation 13)

*£(*)

)

Here, ϊfi ^c2H ^vή ^y) represents the first resultant value of the secondary error correction process.

Next, impulse response values of h ^cl2'^~l](ri) are corrected with reference to the i-th maximum power position as expressed in Equation 14. (Equation 14)

Ag(O = e»-i(«D- £'(«-»!)> («≠«L 2^<< ^<4, 0≤n<N_c)

The process for correcting the impulse response values of h ^cl2'^{~1]( v}ri ') is repeated within the range of 2 < i < Lc, and a resultant value is input to the

final impulse response error correction value h'^(m)( ^vn) ^y as expressed in

Equation 15.

(Equation 15)

The first fast Fourier transformer 130 shown in FIG. 9 receives the useful

data signal h^Km)( ^vή >) in the interval 0 ≤ n < Nc output by the first impulse response estimator 120 and performs a fast Fourier transform process. In the other interval N_c ≤ n < N, the value 0 is input to the first fast Fourier transformer 130. The signal output by the first fast Fourier transformer 130 is a frequency response value of the channel, and is expressed in Equation 16. (Equation 16)

The second fast Fourier transformer 140 receives the useful data signal

in the interval of 0 < n < N output by the serial/parallel transformer 110, transforms it into a parallel signal, and outputs a frequency domain signal

" ^ expressed in Equation 17. (Equation 17)

Y_u ^m)W =

The output signal of the second fast Fourier transformer 140 shown in

FIG. 9 is input to the high priority demodulator 180, the channel equalizer 150, and the second channel estimator 200. The high priority demodulator 180 demodulates the high priority data bits of the frequency domain signal, and outputs them to the first parallel/serial converter 190.

The first parallel/serial converter 190 transforms the demodulated high priority data bits input by the high priority demodulator 180 into serial signals, and outputs them to a source decoder (not shown). The channel equalizer 150 equalizes the frequency domain signal

Y ^u{m\k) by using the frequency response value H "^{m)( ^vk) ' of the channel output by the first channel estimator 120, as expressed in Equation 18. The channel equalizer 150 re-receives a frequency response value of the channel estimated by the second channel estimator 200, and re-equalizes the frequency

domain signal ^{u v} ' . In this instance, the channel equalization performance by the channel equalizer 150 and the second channel estimator 200 is improved as the number of repetitions is increased. However, the number of repeated channel estimations can be limited in consideration of the processing delay time by the receiver. (Equation 18)

Here, the superscript * represents a conjugate complex, and σ ² is noise power of a frequency domain received signal. The second channel estimator 200 re-estimates the frequency response

of the channel by using the frequency domain received signal Y ^uim)( ^\k J) _anc|

the output signal X «^im)( ^kλ ' of the channel equalizer 150. In this instance, the receiving without the first channel estimator 120 can use the channel estimate of the OFDM symbol. The hard decision unit 201 in the second channel estimator 200 performs

a hard decision process on the hard decision performed signal X u^(m}( ^\k ^/) that is output by the channel equalizer 150 in a like manner of FIG. 5 and FIG. 6.

The channel frequency response estimator 202 estimates a frequency response value of a channel by using the hard decision performed signal

u

' and the frequency domain signal as expressed in

Equation 19.

(Equation 19)

The inverse fast Fourier transformer 203 receives the frequency response value estimated by the channel frequency response estimator 202,

and transforms it into an impulse response h^{m)( ^vn ^y) of a channel as expressed in Equation 20. (Equation 20)

H_u ^(m)(k) - Qxp(j2πn(k + k₀)/N), (θ ≤ n < N_c )

The second impulse response corrector 204 receives an impulse

response value ^{v J} of a channel output by the inverse fast Fourier transformer 203, and corrects the impulse response value through the impulse response correction process. In this instance, the impulse response correction process corresponds to the impulse response value correction process performed by the first impulse response corrector 123.

The third Fourier transformer 205 receives an impulse response value

*" _of a channel output by the second impulse response corrector 204 during the interval 0 < n < Nc, receives 0 during the remaining interval N_c ≤ n < N, and performs a fast Fourier transform process. An output signal that is a fast Fourier transform result is a frequency response value of a channel, and is expressed in Equation 21. (Equation 21)

The low priority demodulator 160 receives an output signal Y «^{m)( Vk /λ of the channel equalizer 150, demodulates low priority data bits, and outputs resultant data. The data that are output by demodulating the low priority data bits are input to the second parallel/serial converter 170 to be transformed into serial signals, and the serial signals are transmitted to a source decoder (not shown).

A subcarrier arrangement according to an exemplary embodiment when the transmitter of the AT-DMB system transmits data signals will now be described with reference to FIG. 10 and FIG. 11. Here, the first channel estimator 120 of FIG. 9 estimates the channel of the signal transmitted by the method of FIG. 10, and the second channel estimator 200 of FIG. 9 estimates the channel of the signal transmitted by the method of FIG. 11. A time domain subcaπϊer arrangement will now be described with reference to FIG. 10.

FIG. 10 shows a time domain subcarrier arrangement according to an exemplary embodiment of the present invention. In general, a guard interval of an OFDM signal is in the format of reproducing the latter part of the used symbol interval and inserting the latter part thereof into the former part. However, the guard interval according to the exemplary embodiment of the present invention arranges part of the signal of the existing guard interval at the latter part and arranges a pilot signal at the former part as shown in FIG. 10.

That is, in the exemplary embodiment of the present invention, the guard interval length N_Gι is divided into N_Gι₍₁₎ and N_Gi(2), and a pilot signal having the length N₀ and a pilot guard interval having the length N_C,G_I are arranged in the interval N_Gι i- A signal of a general guard interval is arranged in the interval N_GI₍2₎- In this instance, the latter part of the pilot signal is reproduced and inserted into the pilot guard interval. The time domain OFDM signal including the guard interval follows Equation 22. (Equation 22) N_C - N_{C GI} I (-N₀₁ ≤ n < -N₀₁ + N_{C G/} ) χ(«)_(n) ^NCGI)> (^~NGI + New ^{≤ n < ~N}GI + ΛW , (-N_G/ + N_G/(l) < « < 0)

(0 ≤ n < N)

FIG. 11 shows a frequency domain subcarrier arrangement according to an exemplary embodiment of the present invention. The general T-DMB system arranges 1 phase reference symbol (PRS) for each group of 76 OFDM symbols. Further, when considering high priority transmission, transmission of a frequency domain pilot signal except the PRS is not allowed. However, when considering additional low priority transmission, it is required to transmit an additional pilot signal together with a phase reference symbol for the purpose of in-phase demodulation through channel estimation.

Here, the pilot signal represents a pilot signal that is transmitted within the used symbol interval of the OFDM signal, and it is different from the time domain pilot signal according to the exemplary embodiment of the present invention inserted into the guard interval.

Additional transmission of the pilot signal reduces data transmission efficiency. Further, when power of the pilot signal is greater than power of the data signal, interference between adjacent subcarriers may occur. Therefore, as shown in FIG. 11 in the exemplary embodiment of the present invention, the signal including a phase reference symbol and a layered modulated data signal is controlled to be transmitted.

FIG. 11 shows the same structure of the existing transmission signal exclusively for high priority. However, when the low priority signal is transmitted, no additional pilot signal is transmitted within the used symbol interval.

Channel estimation performance according to an exemplary embodiment of the present invention will now be described with reference to FIG. 12 to FIG. 19. A simulation is performed by recording statistical performance numbers repeatedly in the condition of the random multi-path Rayleigh fading channel and white Gaussian noise. The radio channel model is based on the COST 207 typical urban (TU), and expresses the radio channel condition in the real urban environment. Each performance estimation is based on the symbol error rate (SER) of the high priority and the low priority, and it is assumed that the receiver is perfectly synchronized since it is for checking channel estimation performance.

The explanatory notes shown in FIG. 12 to FIG. 19 have the following meanings.

Coh. (coherent demodulation): in-phase demodulation. Diff. Coh (differentially coherent demodulation): differential in-phase demodulation.

Conv. CE (conventional channel estimation): conventional decision directed channel estimation (DDCE) method.

Prop. CE (proposed channel estimation): channel estimation method according to an exemplary embodiment of the present invention.

Prop. CE1 : channel estimation method using a time domain pilot signal according to an exemplary embodiment of the present invention.

Prop. CE2: channel estimation method using a frequency domain hard decision signal according to an exemplary embodiment of the present invention. Prop. CE3: channel estimation method using a time domain pilot signal and a frequency domain hard decision signal according to an exemplary embodiment of the present invention.

Ideal CE (ideal channel estimation): ideal channel estimation method.

FIG. 12, FIG. 13, FIG. 14, and FIG. 15 show graphs for comparing performance between the exemplary embodiment of the present invention and the prior art assuming that the receiver moves at the speed of 60km/h. In this instance, the number of repeatedly estimating Prop. CE2 and Prop. CE3 will be given as 1 , and it is not restricted thereto. FIG. 12 and FIG. 13 show performance for respective symbol positions according to an exemplary embodiment of the present invention.

FIG. 12 shows a symbol error rate of the high priority according to an exemplary embodiment of the present invention, and FIG. 13 shows a symbol error rate of the low priority. During the process for demodulating the layered modulated signal, high priority demodulation performance of the prior art is not significantly changed by the OFDM symbol position. However, low priority demodulation performance is degraded with respect to OFDM symbol position that is time, which is called the error propagation characteristic.

However, in the channel estimation method according to the exemplary embodiment of the present invention, the high priority and the low priority have no symbol error rate change with respect to time. Also, the channel estimation generates performance near Ideal CE in the order of Prop. CE3, Prop. CE2, and

Prop. CE1.

FIG. 14 and FIG. 15 show performance according to the received signal power according to an exemplary embodiment of the present invention. In this instance, FIG. 14 shows a symbol error rate of the high priority, and FIG. 15 shows a symbol error rate of the low priority.

When the channel estimation error is less, the in-phase demodulation method generally outperforms the differential in-phase demodulation method. However, when the high priority symbol error rate of the in-phase demodulation method is greater than that of the differential in-phase demodulation method, degradation may occur because the DDCE depending on the hard decision error has a large estimation error and is not adaptive to the channel change caused by the OFDM symbol position.

However, performance caused by the received signal power according to the exemplary embodiment of the present invention outperforms the prior art, and particularly, performance of Prop. CE3 almost reaches that of Ideal CE.

FIG. 16 and FIG. 17 show performance according to the moving speed of a receiver according to an exemplary embodiment of the present invention. Here, the number of repeatedly estimating Prop. CE2 and Prop. CE3 is given as 1 , but it is not restricted thereto.

FIG. 16 shows a symbol error rate of the high priority, and FIG. 17 shows a symbol error rate of the low priority. In general, when the receiver's moving speed is increased, the change rate of the channel is also increased, and channel estimation performance and demodulation performance are degraded.

Therefore, since the degree of performance degradation caused by an increase of a moving speed by the receiver must be small in order to maintain stable signal receiving performance, performance under the condition in which the receiver moves fast is very important. Since the prior art uses the channel estimate of the OFDM symbol, it increases performance degradation caused by the increase of the moving speed of the receiver.

However, in the cases of Prop. CE1 and Prop. CE3 for directly estimating the channel of the OFDM symbol according to the exemplary embodiment of the present invention, the degree of performance degradation caused by the increase of the moving speed of the receiver is reduced. Particularly, Prop. CE3 maintains the performance approaching that of Ideal CE in the high-speed environment of 180km/h. FIG. 18 and FIG. 19 show performance according to the number of repeated estimations according to an exemplary embodiment of the present invention. FIG. 18 shows a symbol error rate of the high priority, and FIG. 19 shows a symbol error rate of the low priority. In this instance, since estimation using the prior art degrades performance in the condition in which the receiver moves at the speed of 180km/h, prior art performance is not considered in FIG. 18 and FIG. 19.

Prop. CE2 and Prop. CE3 including repeated estimation improve demodulation performance as the number of repetitions is increased, and the demodulation performance reaches a predetermined level of performance over a predetermined repeated number. In detail, Prop. CE2 reaches stable performance when the repeated number becomes greater than 6 under the condition in which the receiver moves at the speed of 180km/h. Prop. CE3 generates excellent performance through at least 2 repeated numbers.

Therefore, the channel estimation method according to the exemplary embodiment of the present invention solves the performance degradation caused by the receiver's moving speed by increasing the number of repeatedly estimating the channel.

In addition, a channel estimation error is reduced by inserting a pilot signal into a guard interval of an OFDM symbol, and estimating the channel by using a sequence included in the pilot signal.

Also, stable signal receiving performance is maintained in the mobile radio channel condition since additional performance improvement is guaranteed through a repeated channel estimation method using a frequency domain soft decision signal.

Further, performance improvement in the fast moving object speed condition is substantial, and the maximum performance can be generated when linked with channel estimation using a time domain pilot signal.

The above-described embodiments can be realized through a program for realizing functions corresponding to the configuration of the embodiments or a recording medium for recording the program in addition to through the above-described device and/or method, which is easily realized by a person skilled in the art.

While this invention has been described in connection with what is presently considered to be practical exemplary embodiments, it is to be understood that the invention is not limited to the disclosed embodiments, but, on the contrary, is intended to cover various modifications and equivalent arrangements included within the spirit and scope of the appended claims.

Claims

WHAT IS CLAIMED IS:

1. A system for estimating and equalizing a channel of a signal, comprising: a serial/parallel converter for dividing the signal into a guard interval and a useful data signal, and outputting the signals; a first channel estimator for receiving the guard interval signal, estimating an impulse response of the channel, and outputting an impulse response value; a channel equalizer for equalizing channel distortion of the useful data signal by using the impulse response value output by the first channel estimator, and outputting an equalized signal; and a second channel estimator for receiving the useful data signal and the equalized signal output by the channel equalizer, estimating a frequency response to the useful data signal, and outputting a frequency response value.

2. The system of claim 1 , further comprising: a first fast Fourier transformer for establishing a predetermined interval, receiving an impulse response value output by the first channel estimator within the interval, and receiving a predetermined value and outputting it as a frequency response value of the channel outside the interval; a second fast Fourier transformer for receiving the useful data signal output by the serial/parallel converter, applying Fourier transform to the same, and outputting it as a frequency domain signal; a high priority demodulator for receiving the frequency domain signal, demodulating high priority data bits, and outputting them as parallel data; and a low priority demodulator for receiving an equalized signal from the channel equalizer, demodulating low priority data bits, and outputting them as parallel data.

3. The system of claim 1 , further comprising: a first parallel/serial converter for receiving the parallel data from the high priority demodulator, and outputting them as serial data; and a second parallel/serial converter for receiving the parallel data from the low priority demodulator, and outputting them as serial data, wherein the first channel estimator includes: a used pilot signal extractor for extracting a used pilot signal in a guard interval from the input guard interval signal; a channel impulse response estimator for using the used pilot signal extracted by the used pilot signal extractor, estimating an impulse response of the channel, and outputting an initial impulse response value; and an impulse response corrector for correcting the initial impulse response value, and outputting the corrected impulse response value.

4. The system of claim 1 , further comprising: a hard decision unit for performing a hard decision process on the equalized signal output by the channel equalizer, and outputting a resultant signal; a channel frequency response estimator for estimating a frequency response value of a channel by using the hard decision performed signal and the frequency domain received signal; an inverse fast Fourier transformer for performing an inverse fast Fourier transform on the estimated frequency response value to generate and output an impulse response value of the channel; an impulse response corrector for correcting the output impulse response value, and outputting it as a corrected channel impulse response value; and a fast Fourier transformer for establishing a predetermined interval, receiving the output corrected channel impulse response value within the interval, and receiving a predetermined value, performing a Fourier transform process thereon, and outputting it as a frequency response value of a channel outside the interval.

5. A method for estimating and equalizing a channel of a signal, comprising: converting the signal into a guard interval and a useful data signal; estimating an initial impulse response value of the channel through the guard interval signal, and outputting an impulse response value; estimating a frequency response on the channel through the useful data signal, and outputting a frequency response value; correcting the output impulse response value and outputting the impulse response value; and equalizing a channel of the signal based on the corrected impulse response value, and outputting an equalized signal.

6. The method of claim 5, wherein the outputting of the impulse response value includes: selecting a position of a first impulse response value of a channel having maximum, power from among the initial impulse response value including a plurality of response values; correcting the initial impulse response value based on the selected first impulse response value; selecting a position of a second impulse response value having the maximum power at a random position; primarily correcting an impulse response value provided before the random position with reference to the position of the second impulse response value; secondarily correcting an error of the initial impulse response value based on the primarily corrected result value; correcting the secondarily corrected initial impulse response value with reference to the position of the second impulse response value; and outputting the corrected initial impulse response value as the impulse response value.

7. The method of claim 5, wherein the guard interval signal includes a signal acquired by dividing an entire guard interval length into a first guard interval length and a second guard interval length that are generated by partially reproducing the pilot signal interval and the used symbol interval and inserting former part of the useful data signal, the pilot signal interval includes a pilot guard interval and a specific sequence, and the pilot guard interval is generated by reproducing a partial interval of the specific sequence.

8. The method of claim 5, wherein the method includes, after the outputting of the impulse response: performing a hard decision process on the equalized signal, and outputting it as a hard decision signal; estimating a frequency response value of the channel by using the hard decision signal and the frequency domain signal; transforming the estimated frequency response value into an impulse response value of the channel; correcting the transformed impulse response value, transforming the corrected impulse response value into a frequency response value of the channel, and outputting the response value.