WO2008072556A1 - Digital modulator - Google Patents

Digital modulator Download PDF

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Publication number
WO2008072556A1
WO2008072556A1 PCT/JP2007/073626 JP2007073626W WO2008072556A1 WO 2008072556 A1 WO2008072556 A1 WO 2008072556A1 JP 2007073626 W JP2007073626 W JP 2007073626W WO 2008072556 A1 WO2008072556 A1 WO 2008072556A1
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Prior art keywords
coefficient
frequency
modulation
value
coefficients
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PCT/JP2007/073626
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French (fr)
Japanese (ja)
Inventor
Naoki Takahashi
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Nsc Co., Ltd.
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Publication of WO2008072556A1 publication Critical patent/WO2008072556A1/en

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/10—Frequency-modulated carrier systems, i.e. using frequency-shift keying
    • H04L27/12—Modulator circuits; Transmitter circuits

Definitions

  • the present invention relates to a digital modulator, and is particularly suitable for use in a circuit that performs digital modulation using a coefficient stored in a memory.
  • Digital modulation that directly modulates a carrier wave with a digital signal is used instead of modulation with a conventional analog signal.
  • Digital modulation methods include those that change the frequency of the carrier according to the digital signal (modulated data) (FSK: Frequency Shift Keying), those that change the phase (PSK: Phase Shift Keying), and amplitude changes.
  • Reminder Amplitude Shift
  • a carrier wave signal having a predetermined frequency is used for signal processing.
  • the carrier wave signal is generated by a PLL (Phase Locked Loop) circuit (see, for example, Patent Document 1).
  • Patent Document 1 Japanese Patent Application Laid-Open No. 6-3 5 0 6 5 6
  • the present invention has been made to solve such a problem.
  • a digitally modulated signal is improved.
  • the purpose is to make it possible to suppress the occurrence of turbulence in the waveform and to easily generate a digitally modulated signal.
  • the coefficient represents the angular frequency according to the modulation frequency and the sampling frequency, and the first and second coefficients according to the first modulation frequency.
  • the coefficient memory storing the third and fourth coefficients corresponding to the second modulation frequency, and the first coefficient according to the value of the modulation data among the multiple types of coefficients stored in the coefficient memory.
  • the second coefficient set or the third and fourth coefficient set, and one of the two coefficients in the selected set is used as long as the cumulative added value does not exceed the threshold.
  • the other coefficient is used, and the coefficient value is added sequentially for each sampling frequency to obtain the angular frequency for each sample point.
  • a trigonometric function operation unit for generating amplitude by Ri digital modulation signal to the calculation child of the corresponding sine wave angular frequency of Rupoi down bets each.
  • the coefficient representing the angular frequency according to the modulation frequency and the sampling frequency storing two types of coefficients prepared according to the modulation frequency, and the modulation frequency and Represents irrelevant angular frequency
  • Select a coefficient memory that stores fixed coefficients and a fixed coefficient when the value of the modulation data changes, and select two types of coefficients and select two types of coefficients at other times. If the cumulative added value does not exceed the threshold value, one coefficient is used, and when the cumulative added value exceeds the threshold value, the other coefficient is used to calculate the coefficient value stored in the coefficient memory.
  • the cumulative addition operation unit that calculates the angular frequency for each sample point, and the angular vibration for each sample point obtained by the cumulative addition operation unit It has a trigonometric function calculation unit that generates a digital modulation signal by calculating the amplitude of the sine wave corresponding to the number.
  • a digital modulation signal can be generated by digital calculation from only a small number of two types of coefficients prepared for one modulation frequency.
  • a digital modulation signal can be generated by digital computation using coefficients without using a PLL circuit that requires a predetermined convergence time to change the frequency and phase. For this reason, it is not necessary to wait for the convergence time when switching the frequency and phase, and the frequency and phase can be changed instantaneously and smoothly. As a result, it is possible to suppress the disturbance of the waveform of the digital modulation signal. Also, it is only necessary to prepare a plurality of types of coefficients in accordance with the frequency to be changed, and it is not necessary to prepare a plurality of carriers, so that a digital modulation signal can be easily generated.
  • FIG. 1 is a diagram illustrating a configuration example of an FSK modulator according to the first embodiment.
  • FIG. 2 is a diagram illustrating an operation example of a cumulative addition operation unit according to the first and second embodiments.
  • FIG. 3 is a diagram illustrating a waveform of an FSK modulated signal according to the first embodiment.
  • FIG. 4 is a diagram illustrating a configuration example of a PSK modulator according to the second embodiment.
  • FIG. 5 is a diagram illustrating the second embodiment. It is a figure which shows the waveform of the PSK modulation signal by
  • FIG. 1 is a diagram showing a configuration example of an FSK modulator according to the first embodiment in which the digital modulator of the present invention is implemented.
  • the FSK modulator according to the first embodiment includes a coefficient R OM l, a cumulative addition operation unit 2, a CORDIC (Coordinate Rotation Digital Computer) 3, a DZA converter 4, a mixer 5, and a mixer.
  • the antenna 6 is provided.
  • the start pulses for the integrator 15 and C ORD I C 3 included in the cumulative addition operation unit 2 are input at intervals of the sampling frequency f s.
  • the coefficient ROM 1 corresponds to the coefficient memory of the present invention. That is, the coefficient R OM 1 is the modulation frequency (the two types of frequencies f 2 f 2 of the FSK modulation signal to be obtained) and the sampling frequency fs ( ⁇ s> 2 f i f s> 2 f 2 , and fs>> 2 f !, fs>> 2 f 2 ( preferably a factor) representing the angular frequency according to the first and second coefficients for the first modulation frequency. Is stored, and the third and fourth coefficients are stored for the second modulation frequency f2.
  • the first coefficient is 2 f, ⁇ / fs
  • the second coefficient is (S ft — fs) 7c / fs
  • the third coefficient is 2 ⁇ 2 ⁇ / fs
  • the fourth coefficient is ( 2 f 2 _ fs) Tt Z fs.
  • the cumulative addition operation unit 2 sequentially adds the coefficient values stored in the coefficient ROM 1 for each sampling frequency fs to obtain the cumulative addition value of the angular frequency.
  • This cumulative addition operation unit 2 sets the first and second sets of coefficients ⁇ 2 ft ⁇ / fs, (2) according to the value (0 or 1) of the modulation data among the multiple types of coefficients stored in the coefficient ROM 1.
  • the cumulative addition operation unit 2 includes selectors 11 to 14, an integrator 15, and a comparator 16.
  • the first selector 1 1 selects one of the first and second coefficients ⁇ 2 f, ⁇ / fs, (2 ⁇ ,- ⁇ s) ⁇ / fs ⁇ stored in the coefficient ROM 1.
  • To the third selector 13 The second selector 1 2 selects the third and fourth coefficients ⁇ 2 f 2 ⁇ / fs, (2 f 2 — fs) stored in the coefficient ROM 1 and selects the third Selector 1
  • the third selector 13 selects either the coefficient output from the first selector 1 1 or the coefficient output from the second selector 1 2 and outputs it to the integrator 15. .
  • the third selector 13 performs selection according to the value of the input modulation data. That is, when the value of the modulation data is “0”, the output from the first selector 1 1 (the first and second coefficient pairs ⁇ 2 f! ⁇ / f s,
  • the integrator 15 sequentially adds the coefficients supplied from the third selector 13. As a result, the cumulative added value of the angular frequency is obtained. The obtained cumulative addition value is output to CORDIC 3 and comparator 16. Comparator 16 compares the cumulative added value of the angular frequency supplied from integrator 15 with a predetermined threshold value, and outputs a signal corresponding to the comparison result to fourth selector 14. To do.
  • the predetermined threshold value to be compared by the comparator 16 is used when the modulation data value is “0” and when the modulation data value is “1”.
  • the first threshold is (fs— 4 f,) ⁇ / 2 fs (where fs> 4 f.
  • the second threshold is (fs— 4 f 2 ) ⁇ / 2 fs (where fs > 4 f 2 ).
  • the fourth selector 14 selects the comparison result with the first threshold value when the value of the modulation data is “0”, and sends a signal corresponding to the comparison result to the control terminal of the first selector 11. Output to.
  • the fourth selector 14 selects a comparison result with the second threshold value when the value of the modulation data is “1”, and outputs a signal corresponding to the comparison result of the second selector 1 2. Output to the control terminal.
  • the first selector 11 receives a comparison signal indicating that the cumulative added value of the angular frequency supplied from the integrator 15 is not more than the first threshold value.
  • a comparison signal indicating that the accumulated addition value is larger than the first threshold is a comparator.
  • the second selector 12 is supplied with a comparison signal from the comparator 16 indicating that the cumulative added value of the angular frequency supplied from the integrator 15 is less than or equal to the second threshold value. When one of them is selected, select one coefficient ⁇ 2 f 2 ⁇ / fs ⁇ . Also
  • FIG. 2 is a diagram illustrating an operation example of the cumulative addition operation unit 2 configured as described above.
  • 2A is a sine wave signal having a frequency f Jk Hz shown for reference
  • FIG. 2B shows an operation example of the cumulative addition operation unit 2.
  • FIG. In Fig. 2 (a) the black circles indicate the sample points.
  • [f j k H z] the amplitude value of each sample point on the sine wave signal returns to the same value for one cycle (one cycle is 1 msec).
  • Cumulative addition operation unit 2 uses one coefficient ⁇ 2 f! For each sampling frequency f s. Since ⁇ / f s ⁇ is added sequentially, the cumulative added value of the angular frequency gradually increases at a constant rate as shown in Fig. 2 (b). When the cumulative addition value exceeds the first threshold value ⁇ (f s-4 f,) ⁇ / 2 fs ⁇ , the first selector 1 1 sets the value to be added to the other coefficient ⁇ (2 f! -fs) ⁇ / fs ⁇ .
  • the sampling frequency fs is sufficiently larger than the first modulation frequency f ((2 f, ⁇ fs). Therefore, the other coefficient ⁇ (2 ffs) ⁇ / fs ⁇ is a negative value, and its absolute value is also sufficiently larger than the first threshold. Therefore, by adding this other coefficient ⁇ (2 ffs) ⁇ / fs ⁇ , as shown in Fig. 2 (b), the cumulative added value is reduced to a small value at once. After that, one coefficient again ⁇ 2 f! As ⁇ / fs ⁇ is added sequentially, the cumulative added value rises again. If this operation is repeated f [times], it returns to the same value as the cumulative added value at the f t previous sample points.
  • CORDIC 3 corresponds to the trigonometric function calculation unit of the present invention, and calculates the amplitude of the sine wave and cosine wave corresponding to the angular frequency for each sample point obtained by the cumulative addition calculation unit 2. Calculations generate digitally modulated signals (sine and cosine signals with frequency f t or sine and cosine signals with frequency f 2 ). This is as shown in Fig. 2 (b). This corresponds to generating a sine wave signal or cosine wave signal with the waveform shown in Fig. 2 (a) from the waveform data.
  • CORDIC 3 is an algorithm that realizes plane rotation of a two-dimensional vector by elementary function operations such as trigonometric functions, multiplication, and division, and iteratively performs operations by shifting, adding and subtracting, and reading constants from a table.
  • the value of trigonometric function can be obtained with. For example, if the coordinates of the point P (X, y) rotated by the angular frequency of a certain sample point is obtained using the point P 0 (1, 0) on the X axis as the reference point, the X coordinate Can be obtained as the value of the cos function corresponding to the angular frequency, that is, the amplitude of the cosine wave signal to be obtained. Also, the y coordinate can be obtained as the value of the sin function corresponding to the angular frequency, that is, the amplitude of the sine wave signal to be obtained.
  • the D-no A converter 4 converts the frequency or f 2 sine wave signal and cosine wave signal output from CORDIC 3 from a digital signal to an analog signal, respectively.
  • the mixer 5 performs frequency conversion by mixing the sine wave signal and cosine wave signal having the frequency fi or f 2 converted to the analog signal by the D / A converter 4 and the carrier wave signal having the frequency ⁇ .
  • the mixer 5 includes multipliers 2 1 and 2 2 and an adder 2 3.
  • the first multiplier 2 1 multiplies the sine wave signal (sin f or sin f 2 ) output from CORDIC 3 and the cosine wave signal (cos co) as the carrier wave of the predetermined frequency ⁇ . Output to adder 2 3.
  • the second multiplier 2 2 multiplies the cosine wave signal (c os f or cos f 2 ) output from CORDIC 3 and the sine wave signal (sin ⁇ ) as a carrier wave of a predetermined frequency ⁇ . Output to adder 2 3.
  • the adder 23 adds the signals output from the first and second multipliers 2 1 and 2 2 and outputs the result to the antenna 6.
  • the digital modulation signal is radiated from the antenna 6 as, for example, an FM radio wave.
  • FIG. 3 is a diagram showing a digital modulation signal generated by the FSK modulator according to the first embodiment configured as described above, that is, a waveform of the FSK modulation signal.
  • the modulation data when the value of the modulation data is “0”, the frequency (fi + ⁇ ) is a sine wave, and the value of the modulation data is When “1”, a frequency-modulated signal that is a sine wave of frequency (f 2 + ⁇ ) can be obtained.
  • an F S ⁇ modulation signal can be generated by digital calculation from only four types of coefficients stored in the coefficient R ⁇ ⁇ 1.
  • an FSK modulated signal can be generated by a digital operation using coefficients without using a PLL circuit that requires a predetermined convergence time to change the frequency according to the value of the modulation data.
  • the capacity of the coefficient ROM l can be significantly reduced compared to the case where a sine wave signal (sinf t or sinf 2 ) or cosine wave signal (co s or co sf 2 ) is generated by a lookup table. Yes, it has a reputation for reducing the hardware scale.
  • FIG. 4 shows the present invention.
  • FIG. 5 is a diagram illustrating a configuration example of a PSK modulator according to a second embodiment in which a digital modulator is implemented.
  • the PSK modulator according to the second embodiment includes a coefficient ROM 41, a cumulative addition calculation unit 4 2, a CORDIC 3, a mixer 5, and an antenna 6.
  • the coefficient ROM 4 1 is a coefficient representing the angular frequency corresponding to the modulation frequency f and the sampling frequency fs, and two types of coefficients prepared for the modulation frequency f (described in the embodiment of the third embodiment). Corresponding to the first and second coefficients) and a fixed coefficient (one coefficient of " ⁇ / 2") representing the angular frequency independent of the modulation frequency f. ing.
  • the cumulative addition calculation unit 4 2 selects the fixed coefficient “ ⁇ / 2” when the value of the input modulation data changes from “0” to “1” or “1” to “0”. In other cases, select a pair of two types of coefficients ⁇ 2 f 7r / fs, (2 f-fs) ⁇ / fs ⁇ . When two types of coefficients are selected, one coefficient ⁇ 2 f ⁇ / fs ⁇ is used as long as the cumulative added value does not exceed the threshold, and the other coefficient ⁇ Use (2 f-fs) ⁇ / fs ⁇ .
  • the cumulative addition calculation unit 42 obtains the angular frequency for each sample point by sequentially adding the coefficient values selected in this way for each sampling frequency.
  • the cumulative addition operation unit 4 2 includes a first selector 11, a third selector 13, an integrator 15, and a comparator 3 6.
  • the first selector 1 1 selects one of the first and second coefficients ⁇ 2 f ⁇ / ⁇ s, (2 f-fs) ⁇ / fs ⁇ stored in the coefficient ROM 4 1 And output to the third selector 13.
  • the third selector 13 is a fixed coefficient stored in the coefficient ROM 41 and the coefficient output from the first selector 11. Select one of the coefficients and output it to the integrator 15. That is, when the value of the modulation data does not change, the output from the first selector 11 1 (the first and second coefficient pairs ⁇ 2 f ⁇ / fs, (2 f-fs) ⁇ / fs ⁇ ) And select a fixed coefficient ⁇ , 2 ⁇ when the value of the modulation data changes.
  • the integrator 15 obtains the cumulative added value of the angular frequency by sequentially adding the coefficients supplied from the third selector 13. The obtained cumulative addition value is output to CORDIC 3 and comparator 36.
  • Comparator 36 compares the cumulative addition value of the angular frequency supplied from integrator 15 with a predetermined threshold value, and outputs a signal corresponding to the comparison result to the control terminal of first selector 11. Output.
  • the predetermined threshold is (fs ⁇ 4 f) ⁇ / 2 fs (where fs> 4 f).
  • the first selector 11 receives a comparison signal from the comparator 3 6 indicating that the cumulative added value of the angular frequency supplied from the integrator 15 is less than a predetermined threshold value.
  • the comparison signal indicating that the cumulative addition value is larger than the predetermined threshold is supplied from the comparator 36, the other coefficient ⁇ (2 f ⁇ f s) ⁇ / f s ⁇ is selected.
  • CORDIC 3 calculates the amplitude of the sine wave and cosine wave corresponding to the angular frequency for each sample point obtained by the cumulative addition operation unit 42, thereby obtaining a sine wave of frequency f. Generates a signal cosine wave signal.
  • the mixer 5 performs frequency conversion by mixing the sine wave signal (sin f) and cosine wave signal (cos f) output from the CORDIC 3 with the carrier signal having the frequency ⁇ .
  • a mixer 5 ⁇ , sin f co cos ⁇ + cos f ⁇ 3 ⁇ ⁇ 3 ⁇ ⁇ (f + ⁇ ;
  • the digital modulation signal is radiated from the antenna 6 as an FM radio wave, for example.
  • FIG. 5 is a diagram showing a digital modulated signal generated by the PSK modulator according to the second embodiment configured as described above, that is, a waveform of the PSK modulated signal.
  • a phase modulation signal whose phase changes by ⁇ 2 according to the value of the modulation data can be obtained.
  • a PSK modulation signal can be generated by a digital operation using a coefficient without using a PLL circuit that requires a predetermined convergence time to change the phase according to the value of the modulation data.
  • phase change is 90 degrees, simply prepare one type of “ ⁇ 2” as a fixed coefficient, and simply add (or subtract) this in the cumulative addition operation unit 42.
  • the phase can be changed, and the PS modulation signal can be generated easily.
  • the capacity of the coefficient ROM 41 can be significantly reduced compared to the case where a sine wave signal (sin f) or cosine wave signal (cos f) is generated by a look-up table. It also has a remedy if it can be made smaller.
  • two-phase PSK modulation is described as an example, but the present invention is not limited to this.
  • three types of fixed coefficients “ ⁇ Z 2, ⁇ , 3 ⁇ // 2” may be prepared.
  • an ASK modulator is shown in Figure 4.
  • the circuit can be configured by arranging as follows.
  • the coefficient R OM 4 1 stores only two types of the first coefficient ⁇ 2f ⁇ / fs ⁇ and the second coefficient ⁇ (2f ⁇ fs) ⁇ / fs ⁇ .
  • the third selector 13 is omitted, and the output of the first selector i 1 is input to the integrator 15 in a direct manner. Then, for example, an amplifier is provided in the subsequent stage of the mixer 5, and the amplification factor of the modulation signal output from the mixer 5 is changed according to the value of the input modulation data.
  • the present invention is useful for a circuit that performs digital modulation using coefficients stored in a memory, such as an FSK modulator, a PSK modulator, an ASK modulator, and a QAM modulator.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
  • Transmitters (AREA)

Abstract

A digital modulator includes: a coefficient ROM (1), an accumulative addition operation unit (2), and a CORDIC (3). The ROM (1) contains two types of coefficients concerning a first modulation frequency f1 and two types of coefficients concerning a second modulation frequency f2. The accumulative addition operation unit (2) selects coefficients concerning one of the frequencies according to the modulation data value, uses one of the two types of coefficients concerning the selected frequency while the accumulative addition value does not exceed a threshold value and the other type of coefficient when the accumulative addition value exceeds the threshold value, thereby successively adding a coefficient value for each sampling frequency. The CORDIC (3) calculates an amplitude of a sinusoidal wave corresponding to an angular frequency for each sample point obtained by the accumulative addition operation unit (2). Thus, it is possible to generate an FSK modulation signal by a digital operation by using a small number of coefficients without using a PLL circuit which requires a predetermined converging time for changing the frequency.

Description

デジタル変調器 Digital modulator
技術分野 Technical field
本発明はデジタル変調器に関し、 特に、 メモリ に格納された係数を使 つてデジタル変調を行う回路に用いて好適なものである。  The present invention relates to a digital modulator, and is particularly suitable for use in a circuit that performs digital modulation using a coefficient stored in a memory.
明 田  Akita
背景技術 Background art
近年、 無線通信のデジタル化が進行している。 デジタル無線通信では 、 従来のアナログ信号による変調に代わり、 デジタル信号で直接搬送波 を変調するデジタル変調が用いられる。 デジタル変調の方式には、 デジ タル信号 （変調データ） に応じて搬送波の周波数を変化させるもの （ F S K ： Frequency Shift Keying) 、 位相を変化させるもの ( P S K ： Pha se Shift Keying) 、 振幅を変ィ匕させるもの (A S K : Amplitude Shift In recent years, digitization of wireless communication has progressed. In digital wireless communication, digital modulation that directly modulates a carrier wave with a digital signal is used instead of modulation with a conventional analog signal. Digital modulation methods include those that change the frequency of the carrier according to the digital signal (modulated data) (FSK: Frequency Shift Keying), those that change the phase (PSK: Phase Shift Keying), and amplitude changes. Reminder (ASK: Amplitude Shift
Keying) 、 振幅と位相を変化させるもの '( Q A M ： Quadrature Amplitu de Modulation) 等力 Sめる。 Keying), which changes amplitude and phase (QAM: Quadrature Amplitu de Modulation).
一般に、 これらのデジタル変調では、 信号処理の際に所定周波数の搬 送波信号を使用する。 多く の場合、 搬送波信号は、 P L L (Phase Locke d Loop) 回路によって発生する （例えば、 特許文献 1参照） 。  Generally, in these digital modulations, a carrier wave signal having a predetermined frequency is used for signal processing. In many cases, the carrier wave signal is generated by a PLL (Phase Locked Loop) circuit (see, for example, Patent Document 1).
特許文献 1 ： 特開平 6 — 3 5 0 6 5 6号公報  Patent Document 1: Japanese Patent Application Laid-Open No. 6-3 5 0 6 5 6
発明の開示 Disclosure of the invention
しかしながら、 上記特許文献 1 のよ うな P L L方式の場合、 ロ ック状 態に収束して動作が安定するまでに複雑な信号処理が必要で、 所定の時 間がかかってしま う。 そのため、 変調データの値の変化に応じて搬送波 の周波数や位相を変化させる際のレスポンスが悪く 、 デジタル変調され た信号の波形に乱れが生じてしま う という問題があった。 また、 周波数 変調を行う F S Kでは、 周波数が異なる複数の搬送波を用意する'必要が あり、 デジタル変調された信号を簡単に発生させることができないとい う問題もあった。 However, in the case of the PLL system as described in Patent Document 1 above, complicated signal processing is required until the operation is stabilized by converging to the locked state, and a predetermined time is required. Therefore, the carrier wave There was a problem that the response when changing the frequency and phase of the signal was poor and the waveform of the digitally modulated signal was disturbed. In addition, in FSK that performs frequency modulation, it is necessary to prepare a plurality of carriers having different frequencies, and there is a problem that it is not possible to easily generate a digitally modulated signal.
本発明は、 このよ うな問題を解決するために成されたものであり、 変 調データに応じて搬送波信号の周波数や位相を変化させる際のレスボン スを改善することで、 デジタル変調された信号の波形における乱れの発 生を抑制できるよ うにすると と もに、 デジタル変調された信号を簡単に 発生させるこ とができるよ う にすることを目的とする。  The present invention has been made to solve such a problem. By improving the response when changing the frequency and phase of a carrier signal according to modulation data, a digitally modulated signal is improved. The purpose is to make it possible to suppress the occurrence of turbulence in the waveform and to easily generate a digitally modulated signal.
上記した課題を解決するために、 本発明では、 変調周波数およびサン プリ ング周波数に応じた角振動数を表した係数であって、 第 1 の変調周 波数に応じた第 1および第 2の係数と第 2の変調周波数に応じた第 3お ょぴ第 4の係数とを格納した係数メ モ リ と、 係数メ モ リ に格納された複 数種類の係数のうち、 変調データの値に従って第 1および第 2の係数の 組または第 3および第 4の係数の組の何れかを選択し、 選択した組の 2 種類の係数のうち、 累積加算値が閾値を超えない間は一方の係数を使用 し、 累積加算値が閾値を超えたときには他方の係数を使用して、 係数値 をサンプリ ング周波数毎に順次加算していく ことによ り、 サンプルボイ ン ト毎の角振動数を求める累積加算演算部と、 累積加算演算部に.よ り求 められたサンプルポイ ン ト毎の角振動数に対応する正弦波の振幅を計算 するこ とによ りデジタル変調信号を発生する三角関数演算部とを備えて いる。  In order to solve the above-described problem, in the present invention, the coefficient represents the angular frequency according to the modulation frequency and the sampling frequency, and the first and second coefficients according to the first modulation frequency. And the coefficient memory storing the third and fourth coefficients corresponding to the second modulation frequency, and the first coefficient according to the value of the modulation data among the multiple types of coefficients stored in the coefficient memory. And the second coefficient set or the third and fourth coefficient set, and one of the two coefficients in the selected set is used as long as the cumulative added value does not exceed the threshold. When the cumulative added value exceeds the threshold value, the other coefficient is used, and the coefficient value is added sequentially for each sampling frequency to obtain the angular frequency for each sample point. Calculated by the calculation unit and the cumulative addition calculation unit And a trigonometric function operation unit for generating amplitude by Ri digital modulation signal to the calculation child of the corresponding sine wave angular frequency of Rupoi down bets each.
本発明の他の態様では、 変調周波数およびサンプリ ング周波数に応じ た角振動数を表した係数であって当該変調周波数に応じて用意された 2 種類の係数を格納すると と もに、 変調周波数と無関係の角振動数を表し た固定の係数を格納した係数メモ リ と、 変調データの値が変化したとき に固定の係数を選択する と ともに、 それ以外のときに 2種類の係数を選 択し、 2種類の係数を選択した場合には、 累積加算値が閾値を超えない 間は一方の係数を使用し、 累積加算値が閾値を超えたときには他方の係 数を使用して、 係数メモ リ に格納された係数値をサンプリ ング周波数毎 に順次加算していく ことによ り、 サンプルポイン ト毎の角振動数を求め る累積加算演算部と、 累積加算演算部によ り求められたサンプルポイ ン ト毎の角振動数に対応する正弦波の振幅を計算するこ とによ りデジタル 変調信号を発生する三角関数演算部とを備えている。 In another aspect of the present invention, the coefficient representing the angular frequency according to the modulation frequency and the sampling frequency, storing two types of coefficients prepared according to the modulation frequency, and the modulation frequency and Represents irrelevant angular frequency Select a coefficient memory that stores fixed coefficients and a fixed coefficient when the value of the modulation data changes, and select two types of coefficients and select two types of coefficients at other times. If the cumulative added value does not exceed the threshold value, one coefficient is used, and when the cumulative added value exceeds the threshold value, the other coefficient is used to calculate the coefficient value stored in the coefficient memory. By sequentially adding each sampling frequency, the cumulative addition operation unit that calculates the angular frequency for each sample point, and the angular vibration for each sample point obtained by the cumulative addition operation unit It has a trigonometric function calculation unit that generates a digital modulation signal by calculating the amplitude of the sine wave corresponding to the number.
上記のよ うに構成した本発明によれば、 一の変調周波数に対して 2種 類用意した僅かな数の係数だけからデジタル演算によ りデジタル変調信 号を発生することができる。 つま り、 周波数や位相を変えるのに所定の 収束時間を要する P L L回路を使わず、 係数を用いたデジタル演算によ つてデジタル変調信号を発生することができる。 このため、 周波数や位 相の切替時に収束時間を待つ必要がなく 、 瞬時にしかも滑らかに周波数 や位相を変化させることができる。 これによ り、 デジタル変調信号の波 形に乱れが生じるのを抑制するこ とができる。 また、 変化させる周波数 に合わせて複数種類の係数を用意するだけでよく 、 複数の搬送波を用意 する必要がないので、 デジタル変調信号を簡単に発生させるこ とができ る。 図面の簡単な説明  According to the present invention configured as described above, a digital modulation signal can be generated by digital calculation from only a small number of two types of coefficients prepared for one modulation frequency. In other words, a digital modulation signal can be generated by digital computation using coefficients without using a PLL circuit that requires a predetermined convergence time to change the frequency and phase. For this reason, it is not necessary to wait for the convergence time when switching the frequency and phase, and the frequency and phase can be changed instantaneously and smoothly. As a result, it is possible to suppress the disturbance of the waveform of the digital modulation signal. Also, it is only necessary to prepare a plurality of types of coefficients in accordance with the frequency to be changed, and it is not necessary to prepare a plurality of carriers, so that a digital modulation signal can be easily generated. Brief Description of Drawings
図 1 は、 第 1 の実施形態による F S K変調器の構成例を示す図である 図 2は、 第 1 、 第 2の実施形態による累積加算演算部の動作例を示す 図である 図 3は、 第 1 の実施形態による F S K変調信号の波形を示す図である 図 4は、 第 2の実施形態による P S K変調器の構成例を示す図である 図 5 は、 第 2の実施形態による P S K変調信号の波形を示す図である FIG. 1 is a diagram illustrating a configuration example of an FSK modulator according to the first embodiment. FIG. 2 is a diagram illustrating an operation example of a cumulative addition operation unit according to the first and second embodiments. FIG. 3 is a diagram illustrating a waveform of an FSK modulated signal according to the first embodiment. FIG. 4 is a diagram illustrating a configuration example of a PSK modulator according to the second embodiment. FIG. 5 is a diagram illustrating the second embodiment. It is a figure which shows the waveform of the PSK modulation signal by
発明を実施するための最良の形態 BEST MODE FOR CARRYING OUT THE INVENTION
(第 1 の実施形態）  (First embodiment)
以下、 本発明の一実施形態を図面に基づいて説明する。 図 1 は、 本発 明のデジタル変調器を実施した第 1 の実施形態による F S K変調器の構 成例を示す図である。 図 1 に示すよ うに、 第 1 の実施形態による F S K 変調器は、 係数 R OM l、 累積加算演算部 2、 C O R D I C (Coordinat e Rotation Digital Computer) 3、 DZA変換器 4、 ミキサ 5およぴァ ンテナ 6 を備えている。 累積加算演算部 2が有する積分器 1 5および C O R D I C 3に対するスター トパルスは、 サンプリ ング周波数 f sの間隔 で入力される。  Hereinafter, an embodiment of the present invention will be described with reference to the drawings. FIG. 1 is a diagram showing a configuration example of an FSK modulator according to the first embodiment in which the digital modulator of the present invention is implemented. As shown in FIG. 1, the FSK modulator according to the first embodiment includes a coefficient R OM l, a cumulative addition operation unit 2, a CORDIC (Coordinate Rotation Digital Computer) 3, a DZA converter 4, a mixer 5, and a mixer. The antenna 6 is provided. The start pulses for the integrator 15 and C ORD I C 3 included in the cumulative addition operation unit 2 are input at intervals of the sampling frequency f s.
係数 R O M 1 は、 本発明の係数メモ リ に相当するものである。 すなわ ち、 係数 R OM 1 は、 変調周波数 （求める F S K変調信号の 2種類の周 波数 f い f 2) およびサンプリ ング周波数 f s ( ί s〉 2 f い f s> 2 f 2 であり 、 f s > > 2 f ！, f s> > 2 f 2であることが好ま しレヽ） に応じた 角振動数を表した係数であって、 第 1 の変調周波数 に対して第 1およ ぴ第 2の係数を格納すると と もに、 第 2の変調周波数 f 2に対して第 3お よび第 4の係数を格納している。 本実施形態において、 第 1 の係数は 2 f , π / f s, 第 2の係数は （ S f t— f s) 7c / f s、 第 3の係数は 2 ί 2π / f s、 第 4の係数は （ 2 f 2_ f s) Tt Z f sで表される。 累積加算演算部 2は、 係数 R O M 1 に格納された係数値をサンプリ ン グ周波数 f s毎に順次加算していき、 角振動数の累積加算値を求めるもの である。 この累積加算演算部 2は、 係数 R O M 1 に格納された複数種類 の係数のうち、 変調データの値 （ 0または 1 ) に従って第 1および第 2 の係数の組 { 2 f t π / f s， ( 2 f !- f s) π f s} または第 3および 第 4の係数の組 { 2 f 2 π / f s, ( 2 f 2- f s) π / f s} の何れかを選 択する。 そして、 選択した組の 2種類の係数のう ち、 累積加算値が閾値 を超えない間は一方の係数を使用し、 累積加算値が閾値を超えたときに は他方の係数を使用して、 係数値をサンプリ ング周波数 f s毎に順次加算 していく ことによ り、 サンプルポイン ト毎の角振動数を求める。 The coefficient ROM 1 corresponds to the coefficient memory of the present invention. That is, the coefficient R OM 1 is the modulation frequency (the two types of frequencies f 2 f 2 of the FSK modulation signal to be obtained) and the sampling frequency fs (ί s> 2 f i f s> 2 f 2 , and fs>> 2 f !, fs>> 2 f 2 ( preferably a factor) representing the angular frequency according to the first and second coefficients for the first modulation frequency. Is stored, and the third and fourth coefficients are stored for the second modulation frequency f2. In this embodiment, the first coefficient is 2 f, π / fs, the second coefficient is (S ft — fs) 7c / fs, the third coefficient is 2 ί 2 π / fs, and the fourth coefficient is ( 2 f 2 _ fs) Tt Z fs. The cumulative addition operation unit 2 sequentially adds the coefficient values stored in the coefficient ROM 1 for each sampling frequency fs to obtain the cumulative addition value of the angular frequency. This cumulative addition operation unit 2 sets the first and second sets of coefficients {2 ft π / fs, (2) according to the value (0 or 1) of the modulation data among the multiple types of coefficients stored in the coefficient ROM 1. f!-fs) π fs} or the third and fourth coefficient pair {2 f 2 π / fs, (2 f 2 -fs) π / fs}. Then, one of the two types of coefficients in the selected set is used as long as the cumulative added value does not exceed the threshold, and when the cumulative added value exceeds the threshold, the other coefficient is used. By sequentially adding the coefficient values for each sampling frequency fs, the angular frequency for each sample point is obtained.
この累積加算演算部 2は、 セ レク タ 1 1〜 1 4、 積分器 1 5およぴ比 較器 1 6 を備えて構成されている。 第 1 のセレクタ 1 1 は、 係数 R O M 1 に格納された第 1およぴ第 2の係数 { 2 f , π / f s, ( 2 ί ,- ί s) π / f s} の何れかを選択して第 3のセレクタ 1 3に出力する。 第 2 のセ レ クタ 1 2は、 係数 R O M 1 に格納された第 3およぴ第 4の係数 { 2 f 2 π / f s, ( 2 f 2— f s) の何れかを選択して第 3のセレクタ 1The cumulative addition operation unit 2 includes selectors 11 to 14, an integrator 15, and a comparator 16. The first selector 1 1 selects one of the first and second coefficients {2 f, π / fs, (2 ί,-ί s) π / fs} stored in the coefficient ROM 1. To the third selector 13. The second selector 1 2 selects the third and fourth coefficients {2 f 2 π / fs, (2 f 2 — fs) stored in the coefficient ROM 1 and selects the third Selector 1
3に出力する。 Output to 3.
第 3 のセ レク タ 1 3 は、 第 1 のセ レク タ 1 1 から出力される係数と第 2のセレクタ 1 2から出力される係数との何れかを選択して積分器 1 5 に出力する。 この第 3のセレクタ 1 3は、 入力される変調データの値に 応じて選択を行う。 すなわち、 変調データの値が " 0 " のときは第 1 の セレクタ 1 1からの出力 （第 1および第 2の係数の組 { 2 f ！ π / f s, The third selector 13 selects either the coefficient output from the first selector 1 1 or the coefficient output from the second selector 1 2 and outputs it to the integrator 15. . The third selector 13 performs selection according to the value of the input modulation data. That is, when the value of the modulation data is “0”, the output from the first selector 1 1 (the first and second coefficient pairs {2 f! Π / f s,
( 2 f f s) π / f s} ) を選択し、 変調データの値が " 1 " のと きは 第 2のセレクタ 1 2からの出力 （第 3およぴ第 4の係数の組 { 2 f 2 π / f s, ( 2 f 2— f s) π / f s} ) を選択する。 (2 ffs) π / fs}) and when the modulation data value is "1", the output from the second selector 1 2 (the third and fourth coefficient pairs {2 f 2 π / fs, (2 f 2 — fs) π / fs}).
積分器 1 5は、 第 3 のセ レク タ 1 3 よ り供給される係数を順次加算し ていく ことによ り、 角振動数の累積加算値を求める。 求めた累積加算値 は、 C O R D I C 3および比較器 1 6 に出力する。 比較器 1 6は、 積分 器 1 5 よ り供給される角振動数の累積加算値と所定の閾値とを大小比較 し、 その比較結果に応じた信号を第 4 のセ レク タ 1 4に出力する。 The integrator 15 sequentially adds the coefficients supplied from the third selector 13. As a result, the cumulative added value of the angular frequency is obtained. The obtained cumulative addition value is output to CORDIC 3 and comparator 16. Comparator 16 compares the cumulative added value of the angular frequency supplied from integrator 15 with a predetermined threshold value, and outputs a signal corresponding to the comparison result to fourth selector 14. To do.
こ こで、 比較器 1 6 にて比較する所定の閾値は、 変調データの値が " 0 " のと きに使用する第 1 の閾値と、 変調データの値が " 1 " のと きに 使用する第 2の閾値との 2種類がある。 第 1 の閾値は、 （ f s— 4 f ,) π / 2 f s (ただし、 f s > 4 f とする。 また、 第 2の閾値は、 （ f s— 4 f 2) π / 2 f s (ただし、 f s〉 4 f 2) とする。 Here, the predetermined threshold value to be compared by the comparator 16 is used when the modulation data value is “0” and when the modulation data value is “1”. There are two types of thresholds to be used. The first threshold is (fs— 4 f,) π / 2 fs (where fs> 4 f. The second threshold is (fs— 4 f 2 ) π / 2 fs (where fs > 4 f 2 ).
第 4のセレクタ 1 4は、 変調データの値が " 0 " のときには第 1 の閾 値との比較結果を選択し、 その比較結果に応じた信号を第 1 のセ レク タ 1 1 の制御端子に出力する。 また、 第 4のセレクタ 1 4は、 変調データ の値が " 1 " のときには第 2の閾値との比較結果を選択し、 その比較結 果に応じた信号を第 2のセ レク タ 1 2 の制御端子に出力する。  The fourth selector 14 selects the comparison result with the first threshold value when the value of the modulation data is “0”, and sends a signal corresponding to the comparison result to the control terminal of the first selector 11. Output to. The fourth selector 14 selects a comparison result with the second threshold value when the value of the modulation data is “1”, and outputs a signal corresponding to the comparison result of the second selector 1 2. Output to the control terminal.
これによ り、 第 1 のセレクタ 1 1 は、 積分器 1 5 よ り供給される角振 動数の累積加算値が第 1 の閾値以下であることを示す比較信号が比較器 As a result, the first selector 11 receives a comparison signal indicating that the cumulative added value of the angular frequency supplied from the integrator 15 is not more than the first threshold value.
1 6 よ り供給されたとさは、 一方の係数 { 2 f , π / f s} を選択する。 また、 累積加算値が第 1 の閾値よ り大きいこ とを示す比較信号が比較器When supplied from 1 6, select one coefficient {2 f, π / f s}. In addition, a comparison signal indicating that the accumulated addition value is larger than the first threshold is a comparator.
1 6 よ り供給されたとさは、 他方の係数 { ( 2 f f s) π / f s} を選 択する。 When supplied by 1 6, select the other coefficient {(2 f f s) π / f s}.
同様に、 第 2 のセ レクタ 1 2は、 積分器 1 5 よ り供給される角振動数 の累積加算値が第 2 の閾値以下であるこ とを示す比較信号が比較器 1 6 よ り供給されたときは 一方の係数 { 2 f 2π / f s} を選択する。 またSimilarly, the second selector 12 is supplied with a comparison signal from the comparator 16 indicating that the cumulative added value of the angular frequency supplied from the integrator 15 is less than or equal to the second threshold value. When one of them is selected, select one coefficient {2 f 2 π / fs}. Also
、 累積加算値が第 2 の閾値よ り大きいこ とを示す比較信号が比較器 1 6 よ り供給されたときは、 他方の係数 { ( 2 f 2- f s) π / f s} を選択す る。 6 When the comparison signal indicating that the cumulative added value is larger than the second threshold is supplied from the comparator 16, the other coefficient {(2 f 2 -fs) π / fs} is selected. . 6
7  7
図 2は、 以上のよ うに構成した累積加算演算部 2の動作例を示す図で ある。 なお、 図 2 ( a ) は参照のために図示した周波数 f Jk H z ]の正 弦波信号であり、 図 2 ( b ) が累積加算演算部 2 の動作例を示している 。 図 2 ( a ) において、 黒丸は各サンプルポイン トを示している。 f j k H z ]の場合、 正弦波信号上の各サンプルポイ ン トの振幅値は、 個 の波形を 1サイクル （ 1サイクルは l m秒） と して同じ値に戻る。  FIG. 2 is a diagram illustrating an operation example of the cumulative addition operation unit 2 configured as described above. 2A is a sine wave signal having a frequency f Jk Hz shown for reference, and FIG. 2B shows an operation example of the cumulative addition operation unit 2. FIG. In Fig. 2 (a), the black circles indicate the sample points. In the case of [f j k H z], the amplitude value of each sample point on the sine wave signal returns to the same value for one cycle (one cycle is 1 msec).
累積加算演算部 2は、 サンプリ ング周波数 f s毎に一方の係数 { 2 f ！ π / f s} を順次加算していく ので、 図 2 ( b ) のよ うに角振動数の累積 加算値は一定の割合で徐々に大き く なつていく。 そして、 その累積加算 値が第 1 の閾値 { ( f s- 4 f ,) π / 2 f s} を超える と、 第 1 のセレク タ 1 1 によって、 加算する値が他方の係数 { ( 2 f !- f s) π / f s} に 切り替えられる。  Cumulative addition operation unit 2 uses one coefficient {2 f! For each sampling frequency f s. Since π / f s} is added sequentially, the cumulative added value of the angular frequency gradually increases at a constant rate as shown in Fig. 2 (b). When the cumulative addition value exceeds the first threshold value {(f s-4 f,) π / 2 fs}, the first selector 1 1 sets the value to be added to the other coefficient {(2 f! -fs) π / fs}.
いま、 サンプリ ング周波数 f sは第 1 の変調周波数 f 〖に比べて充分に 大きい ( 2 f ,< < f s) 。 よって、 他方の係数 { ( 2 f f s) π / f s } は負の値で、 その絶対値も第 1 の閾値に比べて充分に大きいものとな る。 そのため、 この他方の係数 { ( 2 f f s) π / f s} を加算するこ とによって、 図 2 ( b ) に示すよ うに、 累積加算値が一気に小さい値に 落と される。 その後は、 再び一方の係数 { 2 f ！ π / f s} が順次加算さ れていく ので、 累積加算値が再び上昇していく。 このよ うな動作を f 【回 繰り返すと、 f t個前のサンプルポイン トでの累積加算値と同じ値に戻る Now, the sampling frequency fs is sufficiently larger than the first modulation frequency f ((2 f, <<fs). Therefore, the other coefficient {(2 ffs) π / fs} is a negative value, and its absolute value is also sufficiently larger than the first threshold. Therefore, by adding this other coefficient {(2 ffs) π / fs}, as shown in Fig. 2 (b), the cumulative added value is reduced to a small value at once. After that, one coefficient again {2 f! As π / fs} is added sequentially, the cumulative added value rises again. If this operation is repeated f [times], it returns to the same value as the cumulative added value at the f t previous sample points.
C O R D I C 3は、 本発明の三角関数演算部に相当するものであり 、 累積加算演算部 2によ り求められたサンプルポイ ン ト毎の角振動数に対 応する正弦波および余弦波の振幅を計算するこ とによ り 、 デジタル変調 信号 （周波数 f tの正弦波信号および余弦波信号、 または周波数 f 2の正 弦波信号および余弦波信号） を発生する。 これは、 図 2 ( b ) のよ うな 波形のデータから、 図 2 ( a ) のよ うな波形の正弦波信号または余弦波 信号を生成することに相当する。 CORDIC 3 corresponds to the trigonometric function calculation unit of the present invention, and calculates the amplitude of the sine wave and cosine wave corresponding to the angular frequency for each sample point obtained by the cumulative addition calculation unit 2. Calculations generate digitally modulated signals (sine and cosine signals with frequency f t or sine and cosine signals with frequency f 2 ). This is as shown in Fig. 2 (b). This corresponds to generating a sine wave signal or cosine wave signal with the waveform shown in Fig. 2 (a) from the waveform data.
C O R D I C 3は、 三角関数、 乗算、 除算などの初等関数演算によつ て 2次元ベク トルの平面回転を実現するアルゴリ ズムで、 シフ ト、 加減 算、 テーブルからの定数読み出しによる演算を繰り返し行う ことで三角 関数の値を得ることができる。 例えば、 X軸上の点 P 0 (1, 0)を基準点と して、 あるサンプルポイ ン トの角振動数だけ回転させた点 P (X, y)の座標 を求めれば、 その X座標を当該角振動数に対応する cos関数の値、 つま り求める余弦波信号の振幅と して得ることができる。 また、 y座標を当 該角振動数に対応する sin関数の値、 つま り求める正弦波信号の振幅と して得るこ とができ る。  CORDIC 3 is an algorithm that realizes plane rotation of a two-dimensional vector by elementary function operations such as trigonometric functions, multiplication, and division, and iteratively performs operations by shifting, adding and subtracting, and reading constants from a table. The value of trigonometric function can be obtained with. For example, if the coordinates of the point P (X, y) rotated by the angular frequency of a certain sample point is obtained using the point P 0 (1, 0) on the X axis as the reference point, the X coordinate Can be obtained as the value of the cos function corresponding to the angular frequency, that is, the amplitude of the cosine wave signal to be obtained. Also, the y coordinate can be obtained as the value of the sin function corresponding to the angular frequency, that is, the amplitude of the sine wave signal to be obtained.
Dノ A変換器 4は、 C O R D I C 3 よ り 出力された周波数 または f 2の正弦波信号および余弦波信号をそれぞれデジタル信号からアナログ信 号に変換する。 ミキサ 5は、 D / A変換器 4によ りアナログ信号と され た周波数 f iまたは f 2の正弦波信咅および余弦波信号と周波数 ωの搬送 波信号とを混合して周波数変換を行う。 このミキサ 5は、 乗算器 2 1， 2 2および加算器 2 3 を備えている。 第 1 の乗算器 2 1 は、 C O R D I C 3 よ り 出力された正弦波信号 （sin f ,または sin f 2) と所定周波数 ω の搬送波と しての余弦波信号 （cos co ) とを乗算して加算器 2 3に出力す る。 第 2 の乗算器 2 2は、 C O R D I C 3 よ り 出力された余弦波信号 （c os f ,または cos f 2) と所定周波数 ωの搬送波と しての正弦波信号 （sin ω ) とを乗算して加算器 2 3 に出力する。 The D-no A converter 4 converts the frequency or f 2 sine wave signal and cosine wave signal output from CORDIC 3 from a digital signal to an analog signal, respectively. The mixer 5 performs frequency conversion by mixing the sine wave signal and cosine wave signal having the frequency fi or f 2 converted to the analog signal by the D / A converter 4 and the carrier wave signal having the frequency ω. The mixer 5 includes multipliers 2 1 and 2 2 and an adder 2 3. The first multiplier 2 1 multiplies the sine wave signal (sin f or sin f 2 ) output from CORDIC 3 and the cosine wave signal (cos co) as the carrier wave of the predetermined frequency ω. Output to adder 2 3. The second multiplier 2 2 multiplies the cosine wave signal (c os f or cos f 2 ) output from CORDIC 3 and the sine wave signal (sin ω) as a carrier wave of a predetermined frequency ω. Output to adder 2 3.
加算器 2 3は、 第 1およぴ第 2 の乗算器 2 1 , 2 2から出力される信 号を加算してアンテナ 6に出力する。 つま り、 以上のよ うに構成したミ キサ 5は、 sin f i ' cos co + cos f i ' sin o - sin C f i+ co ) または sin f 2 - cos ω + cos f 2 ' sinew = sin 、 f 2+ o ) なる ^を if了つ こ と によって正 弦波によるデジタル変調信号を発生し、 これをアンテナ 6に出力するよ うになつている。 これによ り 、 デジタル変調信号は、 例えば F M電波と してアンテナ 6から放射される。 The adder 23 adds the signals output from the first and second multipliers 2 1 and 2 2 and outputs the result to the antenna 6. In other words, the mixer 5 configured as above is sin fi 'cos co + cos fi' sin o-sin C f i + co) or sin f 2 -cos ω + cos f 2 'sinew = sin, f 2 + o) It generates a digitally modulated signal using a string wave and outputs it to the antenna 6. As a result, the digital modulation signal is radiated from the antenna 6 as, for example, an FM radio wave.
図 3は、 以上のよ うに構成した第 1 の実施形態による F S K変調器に よ り発生されるデジタル変調信号、 つま り F S K変調信号の波形を示す 図である。 図 3に示すよ うに、 本実施形態の F S K変調器に変調データ を入力すること よって、 変調データの値が " 0 " のときは周波数 （ f i + ω ) の正弦波となり 、 変調データの値が " 1 " のと きは周波数 （ f 2 + ω ) の正弦波となるよ うな周波数変調信号を得ることができる。 FIG. 3 is a diagram showing a digital modulation signal generated by the FSK modulator according to the first embodiment configured as described above, that is, a waveform of the FSK modulation signal. As shown in FIG. 3, by inputting the modulation data to the FSK modulator of this embodiment, when the value of the modulation data is “0”, the frequency (fi + ω) is a sine wave, and the value of the modulation data is When “1”, a frequency-modulated signal that is a sine wave of frequency (f 2 + ω) can be obtained.
以上詳しく説明したよ うに、 第 1 の実施形態によれば、 係数 R Ο Μ 1 に格納した 4種類の係数だけからデジタル演算によ り F S Κ変調信号を 発生することができる。 つま り、 変調データの値に応じて周波数を変え るのに所定の収束時間を要する P L L回路を使わず、 係数を用いたデジ タル演算によって F S K変調信号を発生するこ とができる。  As described above in detail, according to the first embodiment, an F S Κ modulation signal can be generated by digital calculation from only four types of coefficients stored in the coefficient R Ο Μ 1. In other words, an FSK modulated signal can be generated by a digital operation using coefficients without using a PLL circuit that requires a predetermined convergence time to change the frequency according to the value of the modulation data.
このため、 周波数の切替時に収束時間を待つ必要がなく 、 瞬時にしか も滑らかに周波数を変化させるこ とができる。 これによ り、 F S K変調 信号の波形に乱れが生じるのを抑制することができる。 また、 変化させ る周波数に合わせて複数種類の係数を用意するだけでよく 、 複数の搬送 波を用意する必要がないので、 F S K変調信号を簡単に発生させるこ と ができる。  Therefore, it is not necessary to wait for the convergence time when switching the frequency, and the frequency can be changed smoothly and instantaneously only. As a result, the disturbance of the waveform of the FSK modulation signal can be suppressed. Further, it is only necessary to prepare a plurality of types of coefficients in accordance with the frequency to be changed, and it is not necessary to prepare a plurality of carrier waves, so that an FSK modulation signal can be easily generated.
また、 正弦波信号 （s i n f tまたは s i n f 2) や余弦波信号 （co s また は co s f 2) をルックアップテーブルで発生する場合に比べて、 係数 R O M l の容量を格段に少なく するこ とができ、 ハー ドウエア規模を小さ く することができるとレヽぅ メ リ ッ トも有する。 In addition, the capacity of the coefficient ROM l can be significantly reduced compared to the case where a sine wave signal (sinf t or sinf 2 ) or cosine wave signal (co s or co sf 2 ) is generated by a lookup table. Yes, it has a reputation for reducing the hardware scale.
(第 2の実施形態）  (Second embodiment)
次に、 本発明の第 2の実施形態について説明する。 図 4は、 本発明の デジタル変調器を実施した第 2の実施形態による P S K変調器の構成例 を示す図である。 この図 4において、 図 1 に示した構成要素と同一の機 能を有する構成要素には同一の符号を付している。 図 4に示すよ うに、 第 2の実施形態による P S K変調器は、 係数 R O M 4 1、 累積加算演算 部 4 2 、 C O R D I C 3、 ミキサ 5およびアンテナ 6 を備えている。 係数 R O M 4 1 は、 変調周波数 f およびサンプリ ング周波数 f sに応じ た角振動数を表した係数であって、 当該変調周波数 f に対して用意され た 2種類の係数 （第 ίの実施形態で説明した第 1およぴ第 2の係数に相 当） と、 変調周波数 f と無関係の角振動数を表した固定の係数 （こ こで は " π / 2 " の 1種類の係数） を格納している。 Next, a second embodiment of the present invention will be described. FIG. 4 shows the present invention. FIG. 5 is a diagram illustrating a configuration example of a PSK modulator according to a second embodiment in which a digital modulator is implemented. In FIG. 4, components having the same functions as those shown in FIG. 1 are denoted by the same reference numerals. As shown in FIG. 4, the PSK modulator according to the second embodiment includes a coefficient ROM 41, a cumulative addition calculation unit 4 2, a CORDIC 3, a mixer 5, and an antenna 6. The coefficient ROM 4 1 is a coefficient representing the angular frequency corresponding to the modulation frequency f and the sampling frequency fs, and two types of coefficients prepared for the modulation frequency f (described in the embodiment of the third embodiment). Corresponding to the first and second coefficients) and a fixed coefficient (one coefficient of "π / 2") representing the angular frequency independent of the modulation frequency f. ing.
累積加算演算部 4 2は、 入力される変調データの値が " 0 " から " 1 " または " 1 " から " 0 " へと変化したときに固定の係数 " π / 2 " を 選択すると と もに、 それ以外のときに 2種類の係数の組 { 2 f 7r / f s, ( 2 f - f s) π / f s} を選択する。 そして、 2種類の係数を選択した 場合には、 累積加算値が閾値を超えない間は一方の係数 { 2 f π / f s} を使用し、 累積加算値が閾値を超えたときには他方の係数 { ( 2 f - f s ) π / f s} を使用する。 累積加算演算部 4 2は、 このよ うにして選択し た係数値をサンプリ ング周波数毎に順次加算していく こ と によ り 、 サン プルポイン ト毎の角振動数を求める。  The cumulative addition calculation unit 4 2 selects the fixed coefficient “π / 2” when the value of the input modulation data changes from “0” to “1” or “1” to “0”. In other cases, select a pair of two types of coefficients {2 f 7r / fs, (2 f-fs) π / fs}. When two types of coefficients are selected, one coefficient {2 f π / fs} is used as long as the cumulative added value does not exceed the threshold, and the other coefficient { Use (2 f-fs) π / fs}. The cumulative addition calculation unit 42 obtains the angular frequency for each sample point by sequentially adding the coefficient values selected in this way for each sampling frequency.
この累積加算演算部 4 2 は、 第 1 のセ レク タ 1 1、 第 3 のセ レク タ 1 3、 積分器 1 5および比較器 3 6 を備えて構成されている。 第 1 のセ レ グタ 1 1 は、 係数 R O M 4 1 に格納された第 1およぴ第 2の係数 { 2 f κ / ί s, ( 2 f - f s) π / f s} の何れかを選択して第 3 のセ レク タ 1 3に出力する。  The cumulative addition operation unit 4 2 includes a first selector 11, a third selector 13, an integrator 15, and a comparator 3 6. The first selector 1 1 selects one of the first and second coefficients {2 f κ / ί s, (2 f-fs) π / fs} stored in the coefficient ROM 4 1 And output to the third selector 13.
第 3のセレクタ 1 3は、 入力される変調データの値に応じて、 第 1 の セレク タ 1 1から出力される係数と係数 R O M 4 1 に格納された固定の 係数との何れかを選択して積分器 1 5に出力する。 すなわち、 変調デー タの値が変化していないときは第 1 のセレクタ 1 1からの出力 （第 1お よび第 2の係数の組 { 2 f π / f s, ( 2 f - f s) π / f s} ) を選択し 、 変調データの値が変化したときは固定の係数 { π , 2 } を選択する。 積分器 1 5は、 第 3 のセレクタ 1 3 よ り供給される係数を順次加算し ていく こ とによ り、 角振動数の累積加算値を求める。 求めた累積加算値 は、 C O R D I C 3および比較器 3 6 に出力する。 比較器 3 6は、 積分 器 1 5 よ り供給される角振動数の累積加算値と所定の閾値とを大小比較 し、 その比較結果に応じた信号を第 1 のセレクタ 1 1 の制御端子に出力 する。 こ こで、 所定の閾値は、 （ f s— 4 f ) π / 2 f s (ただし、 f s > 4 f ) とする。 According to the value of the input modulation data, the third selector 13 is a fixed coefficient stored in the coefficient ROM 41 and the coefficient output from the first selector 11. Select one of the coefficients and output it to the integrator 15. That is, when the value of the modulation data does not change, the output from the first selector 11 1 (the first and second coefficient pairs {2 f π / fs, (2 f-fs) π / fs }) And select a fixed coefficient {π, 2} when the value of the modulation data changes. The integrator 15 obtains the cumulative added value of the angular frequency by sequentially adding the coefficients supplied from the third selector 13. The obtained cumulative addition value is output to CORDIC 3 and comparator 36. Comparator 36 compares the cumulative addition value of the angular frequency supplied from integrator 15 with a predetermined threshold value, and outputs a signal corresponding to the comparison result to the control terminal of first selector 11. Output. Here, the predetermined threshold is (fs− 4 f) π / 2 fs (where fs> 4 f).
これによ り、 第 1 のセ レク タ 1 1 は、 積分器 1 5 よ り供給される角振 動数の累積加算値が所定の閾値以下であることを示す比較信号が比較器 3 6 よ り供給されたときは、 一方の係数 { 2 f 7c Z f s} を選択する。 ま た、 累積加算値が所定の閾値よ り大きいこ とを示す比較信号が比較器 3 6 よ り供給されたときは、 他方の係数 { ( 2 f — f s) π / f s} を選択 する。  As a result, the first selector 11 receives a comparison signal from the comparator 3 6 indicating that the cumulative added value of the angular frequency supplied from the integrator 15 is less than a predetermined threshold value. When supplied, select one coefficient {2 f 7c Z fs}. When the comparison signal indicating that the cumulative addition value is larger than the predetermined threshold is supplied from the comparator 36, the other coefficient {(2 f − f s) π / f s} is selected.
C O R D I C 3は、 累積加算演算部 4 2によ り求められたサンプルポ ィン ト毎の角振動数に対応する正弦波および余弦波の振幅を計算するこ とによ り 、 周波数 f の正弦波信号おょぴ余弦波信号を発生する。 ミキサ 5は、 C O R D I C 3 よ り出力された正弦波信号 （sin f ) および余弦波 信号 （cos f ) と周波数 ωの搬送波信号とを混合して周波数変換を行う。 すなわち、 ミ キサ 5 ίま、 sin f · cos ω + cos f · 3Ϊη ω = 3ίη ( f + ω ; な る演算を行う こ とによって正弦波によるデジタル変調信号を発生し、 こ れをアンテナ 6に出力するよ うになっている。 これによ り、 デジタル変 調信号は、 例えば F M電波と してアンテナ 6から放射される。 図 5は、 以上のよ うに構成した第 2の実施形態による P S K変調器に よ り発生されるデジタル変調信号、 つま り P S K変調信号の波形を示す 図である。 図 5に示すよ うに、 本実施形態の P S K変調器に変調データ を入力するこ とよって、 変調データの値に応じて位相が π 2だけ変わ るよ うな位相変調信号を得るこ とができる。 CORDIC 3 calculates the amplitude of the sine wave and cosine wave corresponding to the angular frequency for each sample point obtained by the cumulative addition operation unit 42, thereby obtaining a sine wave of frequency f. Generates a signal cosine wave signal. The mixer 5 performs frequency conversion by mixing the sine wave signal (sin f) and cosine wave signal (cos f) output from the CORDIC 3 with the carrier signal having the frequency ω. In other words, a mixer 5 ί, sin f co cos ω + cos f Ϊ 3 η ω = 3 ί η (f + ω; As a result, the digital modulation signal is radiated from the antenna 6 as an FM radio wave, for example. FIG. 5 is a diagram showing a digital modulated signal generated by the PSK modulator according to the second embodiment configured as described above, that is, a waveform of the PSK modulated signal. As shown in FIG. 5, by inputting modulation data to the PSK modulator of this embodiment, a phase modulation signal whose phase changes by π 2 according to the value of the modulation data can be obtained.
以上詳しく説明したよ うに、 第 2の実施形態によれば、 係数 R Ο Μ 4 1 に格納した 3種類の係数だけからデジタル演算によ り P S Κ変調信号 を発生することができる。 つま り、 変調データの値に応じて位相を変え るのに所定の収束時間を要する P L L回路を使わず、 係数を用いたデジ タル演算によって P S K変調信号を発生することができる。  As described above in detail, according to the second embodiment, it is possible to generate a P S Κ modulation signal by digital computation from only three types of coefficients stored in the coefficient R Ο Μ 41. In other words, a PSK modulation signal can be generated by a digital operation using a coefficient without using a PLL circuit that requires a predetermined convergence time to change the phase according to the value of the modulation data.
このため、 位相の切替時に収束時間を待つ必要がなく 、 瞬時にしかも 滑らかに位相を変化させるこ とができる。 これによ り 、 P S Κ変調信号 の波形に乱れが生じるのを抑制することができる。 また、 9 0度の位相 変化であれば、 固定の係数と して " π 2 " の 1種類を用意し、 これを 累積加算演算部 4 2にて加算 （減算でも良い） するだけで簡単に位相を 変えることができ、 P S Κ変調信号を簡単に発生させることができる。  Therefore, it is not necessary to wait for the convergence time when switching the phase, and the phase can be changed instantaneously and smoothly. As a result, it is possible to suppress the occurrence of disturbance in the waveform of the PS modulation signal. If the phase change is 90 degrees, simply prepare one type of “π 2” as a fixed coefficient, and simply add (or subtract) this in the cumulative addition operation unit 42. The phase can be changed, and the PS modulation signal can be generated easily.
また、 正弦波信号 （sin f ) や余弦波信号 （cos f ) をルックアップテ 一ブルで発生する場合に比べて、 係数 R O M 4 1 の容量を格段に少なく するこ とができ、 ハー ドウェア規模を小さ くすることができるとレヽぅ メ リ ッ ト も有する。  In addition, the capacity of the coefficient ROM 41 can be significantly reduced compared to the case where a sine wave signal (sin f) or cosine wave signal (cos f) is generated by a look-up table. It also has a remedy if it can be made smaller.
なお、 ここでは 2相の P S K変調を例に挙げて説明したが、 これに限 定されない。 例えば、 4相の P S K変調器を構成する場合は、 固定の係 数と して " π Z 2 , π， 3 π // 2 " の 3種類を用意すればよい。  In this example, two-phase PSK modulation is described as an example, but the present invention is not limited to this. For example, when configuring a four-phase PSK modulator, three types of fixed coefficients “π Z 2, π, 3 π // 2” may be prepared.
また、 第 1 の実施形態で F S Κ変調、 第 2の実施形態で P S Κ変調を 行う例について説明したが、 A S K変調器や Q AM変調器なども同様の 思想で構成するこ とが可能である。 例えば A S K変調器は、 図 4に示し た回路を以下のよ うにアレンジすることによって構成することが可能で ある。 In addition, the example in which the FS modulation is performed in the first embodiment and the PS modulation is performed in the second embodiment has been described. However, an ASK modulator, a QAM modulator, and the like can be configured based on the same idea. is there. For example, an ASK modulator is shown in Figure 4. The circuit can be configured by arranging as follows.
すなわち、 係数 R OM 4 1 には第 1 の係数 { 2 f π / f s} およぴ第 2 の係数 { ( 2 f - f s) π / f s} の 2種類のみを格納しておく。 また、 第 3のセ レク タ 1 3は省略し、 第 1のセ レク タ i 1 の出力をダイ レク ト に積分器 1 5に入力する。 そして、 例えばミキサ 5の後段に増幅器を設 け、 入力される変調データの値に応じて、 ミキサ 5から出力される変調 信号の増幅率を変える。  In other words, the coefficient R OM 4 1 stores only two types of the first coefficient {2fπ / fs} and the second coefficient {(2f−fs) π / fs}. Also, the third selector 13 is omitted, and the output of the first selector i 1 is input to the integrator 15 in a direct manner. Then, for example, an amplifier is provided in the subsequent stage of the mixer 5, and the amplification factor of the modulation signal output from the mixer 5 is changed according to the value of the input modulation data.
その他、 上記第 1および第 2の実施形態は、 何れも本発明を実施する にあたっての具体化の一例を示したものに過ぎず、 これらによって本発 明の技術的範囲が限定的に解釈されてはならないものである。 すなわち 、 本発明はその精神、 またはその主要な特徴から逸脱することなく 、 様 々な形で実施することができる。 産業上の利用可能性  In addition, each of the first and second embodiments described above is merely an example of a specific example for carrying out the present invention, and the technical scope of the present invention is interpreted in a limited manner. It must not be. In other words, the present invention can be implemented in various forms without departing from the spirit or the main features thereof. Industrial applicability
本発明は、 メモリ に格納された係数を使ってデジタル変調を行う回路 、 例えば F S K変調器、 P S K変調器、 A S K変調器、 Q AM変調器な どに有用である。  The present invention is useful for a circuit that performs digital modulation using coefficients stored in a memory, such as an FSK modulator, a PSK modulator, an ASK modulator, and a QAM modulator.

Claims

1 . 入力される変調データの値に従ってデジタル変調信号を発生するデ ジタル変調器であって、 変調周波数およびサンプリ ング周波数に応じた角振動数を表した係数 であって、 第 1 の変調周波言数に応じた第 1およぴ第 2 の係数と第 2 の変 調周波数に応じた第 3およぴ第 4の係数とを格納した係数メモリ と、 上記係数メモ リ に格納された複数種類の係数のう ち、 上記変調データ の 1 1. A digital modulator that generates a digital modulation signal according to the value of the input modulation data, a coefficient representing an angular frequency corresponding to the modulation frequency and the sampling frequency, and the first modulation frequency A coefficient memory storing the first and second coefficients according to the number and the third and fourth coefficients according to the second modulation frequency, and a plurality of types stored in the coefficient memory. 1 of the above modulation data
4  Four
の値に従って上記第 1およぴ第 2の係数の組または上記第 3およぴ第 4 の係数の組の何れかを選択し、 選択した組の 2種類の係数のう ち、 累積 囲 Select one of the first and second coefficient sets or the third and fourth coefficient sets according to the value of, and select either of the two types of coefficients in the selected set.
加算値が閾値を超えない間は一方の係数を使用し、 累積加算値が上記閾 値を超えたときには他方の係数を使用して、 係数値を上記サンプリ ング 周波数毎に順次加算していく ことによ り、 サンプルポイン ト毎の角振動 数を求める累積加算演算部と、 One coefficient is used while the added value does not exceed the threshold value, and when the accumulated added value exceeds the above threshold value, the other coefficient is used, and the coefficient values are sequentially added for each sampling frequency. Thus, a cumulative addition operation unit for obtaining the angular frequency for each sample point,
上記累積加算演算部によ り求められたサンプルボイ ン ト毎の角振動数 に対応する正弦波の振幅を計算することによ り上記デジタル変調信号を 発生する三角関数演算部とを備えたことを特徴とするデジタル変調器。 A trigonometric function calculation unit that generates the digital modulation signal by calculating the amplitude of the sine wave corresponding to the angular frequency for each sample point obtained by the cumulative addition calculation unit. A digital modulator characterized by
2. 上記第 1 の変調周波数を f い 上記第 2 の変調周波数を f 2、 上記サン プリ ング周波数を f s ( f s> 2 f い f s> 2 f 2) と して、 上記第 1 の係 数は 2 ί ι π Ζ ί 3、 上記第 2の係数は （ 2 f — f s) π / f s, 上記第 3 の係数は S f sTt Z f s, 上記第 4 の係数は （ 2 f 2— f s) π / f s で表さ れることを特徴とする請求の範囲第 1項に記載のデジタル変調器。 2. If the first modulation frequency is f, the second modulation frequency is f 2 , and the sampling frequency is fs (f s> 2 f or f s> 2 f 2 ), then the first coefficient Is 2 ί ι π Ζ ί 3, the second coefficient is (2 f — fs) π / fs, the third coefficient is S f sTt Z fs, the fourth coefficient is (2 f 2 — fs) 2. The digital modulator according to claim 1, expressed by π / fs.
3. 入力される 調データの値に従ってデジタル変調信号を発生するデ ジタル変調器であって、  3. A digital modulator that generates a digital modulation signal according to the value of the input key data,
変調周波数およびサンプリ ング周波数に応じた角振動数を表した係数 であって上記変調周波数に応じて用意された 2種類の係数を格納すると 15 A coefficient that represents the angular frequency according to the modulation frequency and the sampling frequency, and stores two types of coefficients prepared according to the modulation frequency. 15
と もに、 上記変調周波数と無関係の角振動数を表した固定の係数を格納 した係数メモ リ と、 In addition, a coefficient memory storing a fixed coefficient representing the angular frequency unrelated to the modulation frequency, and
上記変調データの値が変化したときに上記固定の係数を選択すると と もに、 それ以外のときに上記 2種類の係数を選択し、 上記 2種類の係数 を選択した場合には、 累積加算値が閾値を超えない間は一方の係数を使 用し、 累積加算値が上記閾値を超えたときには他方の係数を使用して、 上記係数メモリ に格納された係数値を上記サンプリ ング周波数毎に順次 加算していく こ とによ り、 サンプルポイン ト毎の角振動数を求める累積 加算演算部と、  If the fixed coefficient is selected when the value of the modulation data changes, and if the two types of coefficients are selected at other times and the two types of coefficients are selected, the cumulative added value One coefficient is used as long as the value does not exceed the threshold value, and when the cumulative added value exceeds the threshold value, the other coefficient is used, and the coefficient value stored in the coefficient memory is sequentially changed for each sampling frequency. By accumulating, an accumulative addition operation unit that calculates the angular frequency for each sample point,
上記累積加算演算部によ り求められたサンプルボイ ン ト毎の角振動数 に対応する正弦波の振幅を計算することによ り上記デジタル変調信号を 発生する三角関数演算部とを備えたこ とを特徴とするデジタル変調器。  A trigonometric function calculation unit that generates the digital modulation signal by calculating the amplitude of the sine wave corresponding to the angular frequency for each sample point obtained by the cumulative addition calculation unit. A digital modulator characterized by
4 . 上記変調周波数を f 、 上記サンプリ ング周波数を f s ( f s> 2 f ) と して、 上記一方の係数は 2 f w Z f s、 上記他方の係数は （ 2 f — f s ) π / f sで表されるこ とを特徴とする請求の範囲第 3項に記載のデジタ ル変調器。 4. The modulation frequency is f and the sampling frequency is fs (fs> 2 f), where one coefficient is 2 fw Z fs and the other coefficient is (2 f — fs) π / fs. 4. The digital modulator according to claim 3, wherein the digital modulator is provided.
PCT/JP2007/073626 2006-12-11 2007-11-30 Digital modulator WO2008072556A1 (en)

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Cited By (1)

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CN102123064A (en) * 2010-01-12 2011-07-13 华为技术有限公司 Method and device for processing loop

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JPH04152736A (en) * 1990-10-17 1992-05-26 Hitachi Ltd Orthogonal signal generating circuit
JPH06152675A (en) * 1992-11-05 1994-05-31 N T T Idou Tsuushinmou Kk Digital modulator
JPH09200012A (en) * 1995-12-26 1997-07-31 Tektronix Inc Phase modulator and phase modulation method

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH04152736A (en) * 1990-10-17 1992-05-26 Hitachi Ltd Orthogonal signal generating circuit
JPH06152675A (en) * 1992-11-05 1994-05-31 N T T Idou Tsuushinmou Kk Digital modulator
JPH09200012A (en) * 1995-12-26 1997-07-31 Tektronix Inc Phase modulator and phase modulation method

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN102123064A (en) * 2010-01-12 2011-07-13 华为技术有限公司 Method and device for processing loop
CN102123064B (en) * 2010-01-12 2013-11-06 华为技术有限公司 Method and device for processing loop

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