WO2007098766A1 - Modulateur numérique auto-oscillant - Google Patents

Modulateur numérique auto-oscillant Download PDF

Info

Publication number
WO2007098766A1
WO2007098766A1 PCT/DK2007/000104 DK2007000104W WO2007098766A1 WO 2007098766 A1 WO2007098766 A1 WO 2007098766A1 DK 2007000104 W DK2007000104 W DK 2007000104W WO 2007098766 A1 WO2007098766 A1 WO 2007098766A1
Authority
WO
WIPO (PCT)
Prior art keywords
digital
output
block
modulator
transfer function
Prior art date
Application number
PCT/DK2007/000104
Other languages
English (en)
Inventor
Michael A. E. Andersen
Lars Tønnes JAKOBSEN
Original Assignee
Danmarks Teknishe Universitet
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from EP06388014A external-priority patent/EP1830464A1/fr
Application filed by Danmarks Teknishe Universitet filed Critical Danmarks Teknishe Universitet
Publication of WO2007098766A1 publication Critical patent/WO2007098766A1/fr

Links

Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/32—Modifications of amplifiers to reduce non-linear distortion
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F3/217—Class D power amplifiers; Switching amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/351—Pulse width modulation being used in an amplifying circuit

Definitions

  • the present invention relates to a digital self-oscillating modulator having a digital reference signal as input and comprising a forward loop with a first output and a feedback loop.
  • the switching-mode power conversion technology has over the years radically changed the appearance of commercial products, making them far smaller than a few decades ago and leaving the designers with possibilities to experiment with their look and feel without being limited by technology barriers. These benefits are a direct result of the improved efficiency of the power supplies and power amplifiers that use a switching approach instead of a linear one. As a side effect, the amount of heat-sinking material needed is reduced by at least an order of magnitude, which improves the level of integration between the various components, so that the overall board space, weight and volume are significantly reduced and power density is improved.
  • DE 198 38 765 discloses a power amplifier employing a hysteresis control for generating pulse width modulated voltages.
  • the difference between the input voltage and the output voltage is integrated in this power amplifier, said difference being stepped down by a factor corresponding to the ratio of the maximum level of the input voltage to the maximum level of the output voltage.
  • the difference between the scaled output signal and the input signal corresponds to the instantaneous amplitude error of the output signal with the result that the integration corresponds to the accumulated error on the output.
  • the output signal of the integrator is triangular, and when the power amplifier is running idle, the slope of said triangle is of the same value for both the positive and the negative flanks.
  • these flanks change in such a manner that the positive flank discloses a slope differing from the slope of the negative flank.
  • the curve shape remains triangular with straight flanks.
  • the switch-frequency decreases as well.
  • the input signal to the power amplifier is sinusoidal, and then the switch-frequency is at maximum at the zero-pass for the sine curve and significantly lower at the maximum and the minimum value, respectively, of said sine curve.
  • the switch-frequency becomes very low, almost zero.
  • the triangular signal from the integrator is transferred to a comparator, typically a Schmitt-trigger, which converts the triangular signal into square pulses of a varying pulse width.
  • These square pulses are the switching on signals and the switching off signals, respectively, for the transistors in the power amplifier.
  • These switching on pulses are transferred to the output stage of the power amplifier, viz. to the transistors in the output, and therefore these pulses are up- scaled by the relation between said pulse voltages and the internal DC-voltage of the power amplifier.
  • the resulting voltage includes square pulses and is typically of a higher amplitude than the signal voltage.
  • the square voltage is then transmitted to the output filter of the power amplifier, said output filter typically being a second order filter, which is often referred to as a reconstruction filter.
  • the voltage applying after the filter is the output voltage of the power amplifier.
  • the voltage returned to the integrator is the voltage applying before the output filter.
  • a modulator of this type is often referred to as an Astable Integrating Modulator or an AIM. Such a modulator is encumbered with the problem that the distortions of the output filter have not been taken into account.
  • the operational amplifier used to construct the integrator has to be of high qual- ity.
  • WO 02/25357 discloses a controlled oscillation modulator, also called a COM.
  • the COM ensures that the open-loop-phase characteristics involve a phase shift of exactly 180° at the frequency where the open-loop-amplification is 0 dB.
  • the latter is rendered possible by the feedback voltage from the output stage of the power amplifier being forwarded through function blocks causing a phase shift of 180° and/or through function blocks with time delays.
  • the desired phase shift of 180° is obtained by including said phase shift in the function blocks, such as in form of a cascade coupling of poles, and/or by choosing a suitable time delay.
  • the modulator oscillates at the frequency, where the amplification is 0 dB.
  • the resulting signal is a substantially pure sine.
  • the oscillation is superimposed by the input signal.
  • a comparator is subsequently used for generating the switching pulses of the output stage.
  • An increasing loading of the amplifier has the effect that the pure sine resulting from the phase shift of 180° is altered into being something between the pure sine and the triangular voltage known from AIM.
  • the linearity of a modulator depends on variations in the inclination of this signal. As this signal is not a pure triangular curve unlike AIM, but instead something between a sine curve and a triangular curve, the modulator according to the COM principle is nonlinear, and the modulation per se dis- torts the output signal.
  • WO 98/19391 describes a way of improving a class D power amplifier.
  • the amplifier includes an internal modulator generating the well-known pulse-width modulated output signal. This signal is transmitted to an output filter, and the resulting filtered signal is the output voltage of the power amplifier.
  • additional feedback loops have been included, and the characteristics of these feedback loops can compensate for the distortions of said output filter.
  • the described system includes several cascade-coupled feedback loops for compensating the distortions.
  • the system shows an improved procedure structure with respect to power amplifiers without such feedbacks, but the system is per se very complex and requires much design work in order to achieve the desired effect.
  • a system of this type is often referred to as being Multivariable Enhanced Cascade Controlled or MECC.
  • WO2004/086629 discloses a pulse width modulator for converting a digital signal into a PWM signal, comprising a plurality of integrators (11) with integrator gains (12) arranged in series, a comparator (17) for comparing the output of the last integrator (11') with a reference, and thereby creating the PWM signal.
  • the modulator further has means (13) for realizing self-oscillation at a desired switching frequency, and a feedback path (14) connected to a point down stream said comparator and leading to a plu- rality of summing points, each preceding one of said integrators.
  • WO 04/100356 by the present applicant discloses a switch-mode modulator operating at a two-level voltage and including an alternating output stage, an optional output filter and a feedback including a function block with a transfer function.
  • the modulator fur- thermore includes a forward block provided with means for calculating the difference between the signal originating from the function block and a reference signal as well as with a transfer function.
  • the output of the forward block is the input of a Schmitt-trigger, which generates switch on signals for changing the output stage.
  • the output voltage of the modulator applying either after the optional output filter or the output stage is fed back through the function block so as to generate the signal fed back.
  • the transfer function of the function block and of the forward block is chosen both in response to the transfer function of the output filter and in response to the desired total open-loop- transfer function of the modulator.
  • the presented modulator is analogue in the sense that the signal input to the modulator is analogue.
  • US provisional application no. 60/714,077 discloses a self-oscillating modulator operating with a two-level output.
  • the modulator comprises an alternating output stage, an optional output filter, a feedback including a function block with a transfer function, a forward block having a transfer function and is provided with means for calculating the difference between the signal originating from said function block and a reference signal.
  • the output voltage of the modulator either from the alternating output stage and/or the output filter is fed back through the function block, and a multiplication element is placed together with the forward block before the alternating output stage.
  • the multiplication element can thus be utilised to change the po- larity of the signal sent to the alternating output stage, thereby compensating for the polarity of the power converter, in which the modulator is used.
  • Self-oscillating modulators represent an interesting alternative to the conventional fixed frequency pulse width modulators (PWM) for all kinds of power converters as found in almost all electronics equipment from audio power amplifiers, converters for renewable energy sources, uninterruptible power supplies to general isolated power converters and power supplies.
  • PWM pulse width modulators
  • the object of the invention is to provide a new and improved digital self-oscillating modulator for all kinds of power converters as found in almost all electronics equipment from audio power amplifiers, converters for renewable energy sources, uninterruptible power supplies to general isolated power converters and power supplies.
  • the feedback loop comprising a feedback block having a transfer function and a digital output
  • the forward loop comprising an alternating output stage
  • a forward block comprising a filter with a transfer function (MFW) and having a digital output
  • said digital output from the forward block being input to the alternating stage
  • said forward block being provided with means for calculating the difference between the digital output from the feedback block and the digital reference signal, with the proviso that the first output in digital form being fed back to the feedback block and wherein provided that the transfer function of the forward block is formed by a plurality of integrators, the transfer function of the feedback block is not a unity transfer function.
  • self-oscillating is meant a modulator, in which the phase shift of the open- loop transfer function is 360 degrees at the oscillation frequency of the modulator.
  • the bandwidth of the modulator is limited by the switching frequency of the output stage, and in digital modulator with fixed frequency pulse width modulation, the bandwidth is limited by the sampling frequency, which is smaller than or equal to the switching frequency.
  • the sampling frequency is not limited by a fixed switching frequency.
  • the bandwidth is comparable to that of an analogue self-oscillating modulator, but the bandwidth is increased compared to conventional digital modulators.
  • the digital self-oscillating modulator as such overcomes the Nyquist criteria imposed by the switching frequency. Additionally, an increased loop gain can be achieved.
  • the digital implementation of the self-oscillating modulator makes it possible to achieve a better performance by providing a simple modulator structure with internally created carrier, high bandwidth equal to the switching frequency, high power supply rejection ratio through intrinsic power supply voltage feedback within the modulator, as well as reduced electromagnetic interference through spread spectrum techniques in variable switching frequency mode.
  • unity transfer function is meant a non-frequency com- pensating transfer function, which normally is obtained by a direct feedback via a conductor, and possibly with a gain from an amplifier.
  • the transfer function of the forward block (12) is formed by a plurality of integrators, the transfer feedback is not transferred to a plurality of summing points, each preceding one of the integrators.
  • the transfer function of the feedback block is a unity transfer function
  • the transfer function of the forward block is not formed by a plurality of integrators.
  • the modulator additionally comprises an output filter, wherein said first output is the output from the alternating stage, said first output both being fed to the feedback block and the output filter.
  • the forward loop additionally comprises an output filter, which as input has the output from the alternating stage and as output has said first output.
  • the modulator additionally comprises an output filter, wherein both the output from the alternating stage and the output filter in digital form are fed to the feedback block. Including the output filter within the modulator feedback makes it possible to increase open load stability.
  • the transfer function of the forward block and/or the transfer function of the feedback block are frequency compensating, i.e. the transfer function has one or more poles or zero points.
  • the transfer functions are preferably transfer functions of digital filters.
  • the forward block includes a comparator block.
  • the forward block includes a comparator with hysteresis, or alternatively a comparator block and a hysteresis block.
  • modulators with a higher basic linearity and thus lower distortion are obtained.
  • the comparator block includes a multiplexer and a digital comparator, said multiplexer being adapted to multiplex sig- nals from the positive and negative sides of the hysteresis block, the output from the multiplexer being sent to the digital comparator, which compares the output from the multiplexer with a digital reference signal.
  • This reference signal can for instance be the output from a filter of the forward block.
  • the comparator block includes a digital comparator and an S-R latch. Thereby a particular simple implementation of the comparator block is introduced.
  • the feedback loop additionally comprises an A/D-converter. If the signal fed back is analogue, which will for instance often be the case, if the output from the output filter is fed back, this signal must be converted to a digital signal, which is processed by the feedback block.
  • the A/D- converter thus produces the digital form of said first output signal.
  • the forward block includes an integrator.
  • an integrator In another embodiment of the modulator according to the invention, the forward block includes an integrator.
  • the integrator comprises an adder and a clocked latch. This provides a particularly simple digital implementation of the integrator.
  • the transfer function of the forward block, the transfer function of the filter of the feedback block, as well as the transfer function of the optional output filter depend on the desired open-loop transfer function.
  • the transfer function of the forward block can be chosen to compensate the error signal (difference between the signal fed back and the reference signal) in an optimised way dependent of the specific modulator, and the transfer function of the output filter can be optimised to reconstruct the signal.
  • the transfer function of the feedback block can in turn be determined from the desired open-loop transfer function.
  • the alternating output stage is a switching amplifier with bidirectional switches.
  • the means for comparing the signal fed back with the reference signal comprises of two identical digital comparator sections, which have inverted inputs.
  • a digital multiplication element is placed together with the forward block before the alternating output stage.
  • the multiplication element can thus be utilised to change the digital polarity of the signal sent to the alternating output stage, thereby compensating for the polarity of for instance the power converter, in which the modulator is used.
  • the alternating output stage is supplied with an AC-voltage from an AC-link converter and the digital multiplication element works by switching between the two identical digital comparator sections with inverted inputs, said switching being controlled by the sign of the AC-voltage from the AC-link converter.
  • one or more of the zero points or poles of the feedback block and of the forward block are coinciding or approximately coinciding with the pole or zero points of the optional output filter.
  • the modulator further includes one or more additional output filters, one or more additional feedbacks with transfer functions coupled either after the output stage, after the output filter or after one or more of the additional output filters, as well as one or more forward blocks, which include both means for calculating the difference between one of the signals fed back from the additional function blocks and a reference signal, as well as a transfer function.
  • the transfer function of the function block, the transfer function of the forward block, the additional output filters, the transfer functions of the additional feedback blocks and the transfer function of the additional forward blocks together are generating the desired total transfer function.
  • the desired total transfer function is similar to a first order low-pass-characteristic or integrator characteristic.
  • a synchronised modulator comprising two of the afore-mentioned digital modulators, wherein a reference voltage is input to a first digital modulator and a digital inverted reference voltage is input to a second digital modulator, the outputs from said first and said second digital modulator being synchronised by use of a synchronisation block.
  • Fig. 1 shows a block diagram of a digital self-oscillating modulator (DISOM) according to the invention
  • Fig. 2 a block diagram of a DISOM according to the invention with an additional outer control loop
  • Figs. 3-7 different embodiments of DISOMs comprising comparators with hysteresis
  • Figs. 22-24 different embodiments of hysteresis modulators with substantially constant switching frequency
  • Fig. 43 a block diagram of a prototype for proofing the DISOM concept
  • Fig. 44 a plot of the switching frequency as a function of the duty cycle of a prototype for proofing the DISOM concept
  • Fig. 45 a DC/DC converter implementing a DISOM
  • Fig. 48 a synchronised modulator comprising two DISOMs
  • Fig. 49 a digital implementation of a SICAM type DISOM modulator
  • Fig. 50 a digital comparator with hysteresis and synchronisation input
  • Fig. 51 a DISOM modulator type 1.1 with variable hysteresis window.
  • FIG. 1 A block diagram of a digital self-oscillating modulator (DISOM) according to the inven- tion is shown in Fig. 1.
  • the structure is very general and can as such include digital versions of any of the conventional analogue self-oscillating modulators found in the power converters or switching audio power amplifiers with DC-link or HF-link power supplies.
  • the digital error signal from a summing junction 11 is fed to a modulator feed forward block 12 comprising a filter 12' having a transfer function MFW.
  • the feed forward block 12 will optionally include a hysteresis block 13 or a comparator with hysteresis as well as other optional components, which will later be explained.
  • the digital output from the forward block 12 is then fed to the switching amplifier 14, which is supplied with the voltage from a power supply PS.
  • the output from the switching stage or the switching amplifier 14 can optionally be passed through an output filter 15 and to the load 16, which can be a loudspeaker or a general load for a power converter.
  • the output from the switching stage or the switching amplifier 14 can be regarded as either analogue or digital depending on the particular implementation of the switching stage or the switch- ing amplifier 14, which according to a preferred embodiment is a switching power stage.
  • the output filter 15 is here depicted as a second order low-pass filter with passive LC components, said low-pass filter 15 for instance reconstructing a sinusoidal signal or input signal V ref from the square voltage from the switching stage or the switching amplifier 14.
  • the output filter 15 can have any desired frequency compensating transfer function.
  • the output from either the switching stage or the switching amplifier 14 or the output filter 15 is fed back through the modulator feedback block 18 having a transfer function MFB.
  • the input to the feedback block 18 must be digital. Therefore, the feedback loop can optionally include an A/D-converter 17, which converts an ana- logue signal to a digital signal. The A/D-converter 17 is thus necessary, if the signal fed back from the switching stage or the switching amplifier 14 and/or the output filter 15 is analogue.
  • the digital output from the feedback block 18 is fed back to a summing junction 11 , which extracts the error signal as the difference between the digital signal fed back and a digital reference signal V r ⁇ f .
  • the transfer functions of the feed forward block 12, the feedback block 18 and the output filter 15 can have any desired frequency dependency.
  • the filters can inter alia be high-pass filters, low-pass filters, band-pass filters, band-stop filters or unity filters based on the desired frequency compensation of the particular self-oscillating modulator and the open loop transfer function.
  • the transfer function MFW of the forward block 12 and/or the transfer function MFB of the feedback block 18 have a frequency compensating effect, i.e. one or more poles or zeros.
  • the digital self-oscillating modulator 1 can have one or more outer control loops, which improve the performance of the modulator 1 even further by introducing additional gain and optimising bandwidth within a control feed forward block 20 with a transfer function CFW and a control feedback block 22 with a transfer func- tion CFB as shown in Fig. 2.
  • the difference between the signal fed back through the control feedback block 22 and the reference signal V r ⁇ f is extracted at an additional summing junction 19. Since the output from the DISOM 1 is typically an analogue signal, the additional feedback loop can optionally include an additional A/D-converter 21.
  • Figs. 3-7 show different embodiments of DISOMs, which include a comparator with hysteresis in the forward block 12.
  • Fig. 3 shows a first embodiment of a DISOM version of a band-pass current control type. This embodiment will typically be combined with an outer control loop (not shown). The illustrations of these transfer functions cor- respond to Bode plots, which are well known to a person skilled in the art.
  • Fig. 4 shows a second embodiment of a DISOM version of a band-pass current control type. The current measurement is here replaced with a current estimation, for instance by directly coupling the voltage over the inductor of the LC output filter, by measuring the potential of a winding of the inductor of the output filter or via a transformer.
  • Fig. 3-7 show different embodiments of DISOMs, which include a comparator with hysteresis in the forward block 12.
  • Fig. 3 shows a first embodiment of a DISOM version of a band-pass current control type.
  • FIG. 5 shows a third embodiment, which corresponds to the embodiment shown in Fig. 4, but where the output from the amplifier stage can be regarded as a digital signal.
  • the fourth embodiment shown in Fig. 6 corresponds to the embodiment shown in Fig. 5 combined with an outer control loop.
  • a fifth embodiment is shown in Fig. 7.
  • This embodiment has a current estimation by use of the voltage at the output.
  • the embodiment does not have an output filter and can for instance be used to drive a speaker directly, also called an ac- tive transducer (ACT) or a pulse-modulated transducer (PMT).
  • ACT ac- tive transducer
  • PMT pulse-modulated transducer
  • Fig. 8 shows a general block diagram of a digital version of the GLIM type 1 modulators (see also WO 04/100356), and Figs. 9-14 show different embodiments of this type of modulator.
  • This type of digital GLIM has a first order low-pass characteristic for the open loop transfer function.
  • the A/D-converter can be neglected, since both the output from the hysteresis comparator, which optionally can be fed back to the feedback block, and the output from the switching stage or the switching ampli- fier are digital.
  • the embodiment shown in Fig. 9 corresponds to the AIM- implementation (see also DE 198 38 765). All the GLIM type 1 embodiments are simple and cheap to implement.
  • the embodiments can be part of a power modulator with an output filter and an outer control loop as shown in Fig. 14.
  • Fig. 15 shows a general block diagram of a digital version of the GLIM type 2 modulators (see also WO 04/100356), and Figs. 16-21 show different embodiments of this type of modulator.
  • This type of digital GLIM also has a first order low-pass characteristic for the open loop transfer function.
  • the A/D-converter must be used in this type of modulator, since the output from the output filter is analogue.
  • the GLIM type 2 modulators are more expensive to implement, since a fast A/D-converter is necessary.
  • these embodiments can be part of a power modulator with an output filter and an outer control loop as shown in Fig. 21.
  • Figs. 22-24 show different embodiments of modulators with a substantially constant switching frequency, all of them being current mode types of hysteresis modulators.
  • Fig. 22 shows a DISOM version of an AIM modulator with a variable hysteresis window, which is carried out in order to achieve a constant switching frequency. This em- bodiment is very simple to implement, since the input of the hysteresis window of the digital hysteresis comparator shown can be used directly without the use of an A/D- converter, since the signal is digital.
  • Fig. 22-24 show different embodiments of modulators with a substantially constant switching frequency, all of them being current mode types of hysteresis modulators.
  • Fig. 22 shows a DISOM version of an AIM modulator with a variable hysteresis window, which is carried out in order to achieve a constant switching frequency.
  • This em- bodiment is very simple to implement, since the input of the hy
  • FIG. 23 shows a similar embodiment, which incorporates a digital multiplicator, which makes the implementation slightly more complicated and expensive.
  • Fig. 24 shows another embodiment, in which a digital comparator is used for comparing the signal fed back of the power converter output current and the reference signal and thus forms the summing junction 11. This embodiment must include an A/D-converter in the feedback loop.
  • Figs. 25-31 show different embodiments of phase shift types of modulators, which use a digital power stage without hysteresis, i.e. the hysteresis window is equal to zero.
  • Fig. 25 shows a DISOM version of the COM implementation (see also WO 98/19391 ).
  • the A/D-converter can be neglected since the output from the switching stage or the switching amplifier is digital. However, an A/D-converter can be used to compensate for errors of the power stage.
  • Fig. 26 shows a similar COM-type power modulator with an output filter and an outer control loop. The outer loop must include an A/D- converter.
  • Fig. 27 shows a DISOM version of a GCOM (see also WO 2004/047286).
  • the A/D-converter must be of a fast type.
  • Fig. 28 shows a similar GCOM-type power modulator with an outer control loop. A single A/D-converter can be used for both the inner feedback loop and the outer control loop.
  • Fig. 29 shows a DISOM version of a CCOM (see also WO 02/25357). The embodiment uses a current measurement at the output of the switching stage or the switching amplifier, and the feedback loop must include a fast A/D-converter.
  • Fig. 30 shows a similar CCOM-type modulator with an outer control loop, which must include a separate A/D-converter.
  • Fig. 31 shows yet another embodiment of the phase-shift type modulator.
  • Figs. 32-42 show different embodiments of hysteresis comparators for use in DISOMs.
  • X "in”, and “input” denote the signal from for instance the summing junction or from a filter of the forward block
  • V a and V b are the signals from the positive and negative sides of the hysteresis windows, respectively, V b usually being a negative number.
  • the embodiment shown in Fig. 32 incorporates a 2-1 multiplexer and a digital comparator, both of which are well known to a person skilled in the art.
  • the digital comparator compares the input X and the output Y from the 2-1 multiplexer. The output of the digital comparator is fed back to the SEL terminal of the 2-1 multiplexer.
  • a and B are input to the 2-1 multiplexer, and the output of the 2-1 multiplexer is either A or B depending on the state of SEL.
  • Fig. 33 shows a similar embodiment, but where the reference point of the input is shifted (with the value B).
  • Fig. 34 shows yet another embodiment, which incorporates two digital comparators, and an S-R latch, the functions of which all are well known to a person skilled in the art.
  • Fig. 35 is substantially the same embodiment as shown in Fig. 34.
  • the hysteresis window HW input to the digital comparator is the difference between A and B.
  • Fig. 36 shows a simplified embodiment with only one comparator.
  • Fig. 37 shows a digital power hysteresis comparator including a digital comparator with a hysteresis comparator output (HC Output), which is input to a power stage, the output of which is fed to a feedback loop comprising an A/D-converter and a constant gain K.
  • This embodiment compensates for the voltage drop in the transistors of the power stage.
  • Fig. 38 shows an alternative embodiment of a digital power hysteresis compara- tor, which includes a 2-1 multiplexer in the feedback loop. As long as the output from the power stage is fed to the SEL terminal of the 2-1 multiplexer, this embodiment functions identical to the embodiment shown in Fig. 37, where both timing errors and voltage supply errors of the power stage are compensated.
  • Figs. 39- 41 show similar embodiments.
  • the hysteresis window in Fig. 41 is constant.
  • Fig. 42 shows yet another implementation, where Z 1 is a constant and Z 2 is the hysteresis window, which can be constant or externally controlled.
  • Fig. 49 shows a digital implementation of a SICAM modulator (see also US provisional application 60/714,077).
  • the input (in) is fed to two identical digital comparators, which compare the input with the signal fed back through the feedback loop.
  • the outputs from the two digital comparators are digitally inverted.
  • the outputs are fed to a 2-1 multiplexer, the output of which is determined by the state of SEL.
  • the output from the 2-1 multiplexer is fed to the alternating output stage, which preferably comprises a switch- ing amplifier with bidirectional switches.
  • the alternating output stage is supplied with an AC-voltage from an HF-link converter, the AC-voltage also being fed to the SEL.
  • the polarity of the HF-link converter thereby determines the output from the 2-1 multiplexer.
  • the digital output from the alternating output stage is fed back through the feedback loop comprising an A/D-converter and a feedback block with gain K.
  • the transfer functions of the feedback block and forward block can of course be any desired function.
  • the SICAM modulator can also contain an output filter and/or one or more additional feedback loops.
  • FIG. 43 A block diagram of the main modulator is shown in Fig. 43.
  • the shown embodiment is one of the simplest forms of a DISOM modulator.
  • the main modulator is digital and has been implemented in a Complex Programmable Logic Device (CPLD).
  • CPLD Complex Programmable Logic Device
  • An input REF from an external controller is fed to a summing junction 31 , which is also fed by the signal from the feedback loop.
  • the summing junction 31 extracts the difference between these two signals.
  • the output from the sum- ming junction 31 is sent to a latch 32, which also receives a latch enable signal LE.
  • the output from the latch 32 is sent to an integrator 40, which consists of an adder 33 and a clocked latch 34, the output of the clocked latch 34 being fed back to the adder 33.
  • the output of the integrator 40 is sent to a comparator 45 with hysteresis.
  • the comparator 45 comprises two separate comparators 35, 36 and an S-R latch 37.
  • the output from the integrator is fed to both comparators 35, 36, one comparator 35 comparing the signal to the signal from the hysteresis window HW and the other comparator 36 comparing the signal to zero.
  • the output from the two comparators 35, 36 are fed to the R and S terminals of the S-R latch 37, respectively.
  • the output from the S-R latch 37 is fed back to a multiplexer 38, which sets the output to zero if the input is zero and the output to T if the input is one.
  • the output of the multiplexer 38 is fed back to the summing junction 31.
  • the switching frequency f sw of the main modulator is given as a function of the duty cycle of the modulator output given by the following expression:
  • f sw is the switching frequency
  • n is number of bits of the reference input REF
  • T c i ock is the period of time for the clocked latch
  • HW is the hysteresis window
  • D is the duty cycle.
  • the overall control loop including the DC/DC converter is shown in Fig. 45.
  • the CFB block 22 is not included in the feedback loop, since the CFB block simply has a gain of one in the shown embodiment.
  • the CFW block has a transfer function with three zeros and an integrator. The placement of the three zeros determines the overall transfer function of the DC/DC converter.
  • fig. 46 and 47 Some experimental results measured on the prototype are shown in fig. 46 and 47.
  • the output voltage is 12V and the load current is 1OA in Fig. 46.
  • the lower trace 42 shows the gate signal for the two MOSFETs in the offline converter and the upper trace 41 shows the output voltage.
  • the switching frequency is approximately 80 kHz.
  • the output voltage is 9V and the load current is 1OA in Fig. 47.
  • the duty cycle of the MOSFET gate signal has been decreased and the switching frequency has fallen to 59 KHz.
  • the lower trace 77 shows the gate signal for the two MOSFETs in the offline converter and the upper trace 76 shows the output voltage.
  • the measurements shown in Fig. 46 and 47 correspond well with the theoretical curve shown in Fig. 44, which shows that the switching frequency is decreased by a decrease in duty cycle.
  • Fig. 48 shows a block diagram of a DISOM version for synchronising two modulators.
  • the DISOM+ 51 and DISOM- 52 can be any type of DISOM modulator, for instance of the phase shift type or the hysteresis type.
  • the reference signal X in is fed to DISOM+
  • the two modulators 51 , 52 are synchronised by the synchronisation block 53.
  • synchronisation for instance by an optional external synchronisation signal ES, via the capacitors of the output filter, by summing part of the high frequency part of the carrier waves, by summing parts of the hysteresis window, or by master/slave control.
  • the hysteresis com- parator output will be reset to logic O', when the comparator input (Input) becomes lar- ger than the hysteresis window.
  • the positive transition of the switching output has been synchronised to the positive transition of the Sync, input.
  • the second method for making a DISOM modulator operate with a fixed switching fre- quency is to change the hysteresis window according to the modulation index/duty cycle of the switching output.
  • An example of how this can be achieved is shown in Fig. 51FeJI! Henvisningskilde ikke fundet., which gives a more detailed explanation of the embodiment shown in Fig. 22.
  • a hysteresis window calculator block HWC calculates the hysteresis window size based on the input to the DISOM modulator and sets the hysteresis window.
  • the HWC can be implemented as either a mathematical function that actively calculates the hysteresis window or as a lookup table, where the hysteresis window values have been precalculated and stored.
  • An optional digital lowpass filter can be inserted between the hysteresis window calculator HCW and the modulator input to avoid large changes in the switching frequency during transient conditions.

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Physics & Mathematics (AREA)
  • Nonlinear Science (AREA)
  • Amplifiers (AREA)

Abstract

Modulateur (1) numérique auto-oscillant ayant comme entrée (Vref) un signal de référence numérique et comprenant une boucle directe avec une première sortie et une boucle de contre-réaction. La boucle de contre-réaction comprend un bloc (18) de contre-réaction ayant une fonction de transfert (MFB) et une sortie numérique. La boucle directe comprend un étage (14) de sortie alternatif et un bloc (12) direct comprend un filtre (12') ayant une fonction de transfert (MFW) et une sortie numérique. La sortie numérique du bloc (12) direct est acheminée à l'entrée de l'étage (14) alternatif. Le bloc (12) direct est doté de moyens pour calculer la différence entre la sortie numérique du bloc (18) de contre-réaction et le signal de référence numérique (Vref). La première sortie est sous forme numérique renvoyée au bloc (18) de contre-réaction. Sous réserve que la fonction (MFW) de transfert du bloc (12) direct est formée par une pluralité d'intégrateurs, la fonction (MFB) de transfert du bloc (18) de contre-réaction n'est pas une fonction de transfert unitaire.
PCT/DK2007/000104 2006-03-01 2007-03-01 Modulateur numérique auto-oscillant WO2007098766A1 (fr)

Applications Claiming Priority (4)

Application Number Priority Date Filing Date Title
EP06388014A EP1830464A1 (fr) 2006-03-01 2006-03-01 Modulateur numérique auto-oscillant
EP06388014.0 2006-03-01
US78045806P 2006-03-07 2006-03-07
US60/780,458 2006-03-07

Publications (1)

Publication Number Publication Date
WO2007098766A1 true WO2007098766A1 (fr) 2007-09-07

Family

ID=38038697

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/DK2007/000104 WO2007098766A1 (fr) 2006-03-01 2007-03-01 Modulateur numérique auto-oscillant

Country Status (1)

Country Link
WO (1) WO2007098766A1 (fr)

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2008075318A3 (fr) * 2006-12-21 2008-08-28 Bang & Olufsen Icepower As Système de correction d'erreur pour un étage de puissance de classe d
JP2015512609A (ja) * 2012-03-30 2015-04-27 クアルコム，インコーポレイテッド バックレギュレータの外部ｌｃフィルタ内のピーキングを抑圧するためのシステムおよび方法

Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO1998019391A2 (fr) * 1996-10-31 1998-05-07 Bang & Olufsen A/S Amplificateur de puissance module en impulsions presentant un procede ameliore de commande en cascade
DE19838765A1 (de) * 1998-08-26 2000-05-11 Elbo Gmbh Selbstschwingender Digitalverstärker
WO2002025357A2 (fr) * 2000-09-19 2002-03-28 Bang & Olufsen Ice Power A/S Modulateur a auto-oscillation controlee et systeme de conversion de puissance utilisant un tel modulateur
WO2004086629A1 (fr) * 2003-03-24 2004-10-07 Bang & Olufsen Icepower A/S Modulateur numerique a oscillations a largeur d'impulsions controlee
WO2004100356A2 (fr) * 2003-05-09 2004-11-18 Danmarks Tekniske Universitet Modulateur integrant une boucle globale

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO1998019391A2 (fr) * 1996-10-31 1998-05-07 Bang & Olufsen A/S Amplificateur de puissance module en impulsions presentant un procede ameliore de commande en cascade
DE19838765A1 (de) * 1998-08-26 2000-05-11 Elbo Gmbh Selbstschwingender Digitalverstärker
WO2002025357A2 (fr) * 2000-09-19 2002-03-28 Bang & Olufsen Ice Power A/S Modulateur a auto-oscillation controlee et systeme de conversion de puissance utilisant un tel modulateur
WO2004086629A1 (fr) * 2003-03-24 2004-10-07 Bang & Olufsen Icepower A/S Modulateur numerique a oscillations a largeur d'impulsions controlee
WO2004100356A2 (fr) * 2003-05-09 2004-11-18 Danmarks Tekniske Universitet Modulateur integrant une boucle globale

Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2008075318A3 (fr) * 2006-12-21 2008-08-28 Bang & Olufsen Icepower As Système de correction d'erreur pour un étage de puissance de classe d
US7839211B2 (en) 2006-12-21 2010-11-23 Bang & Olufsen Icepower A/S Error correction system for a class-D power stage
JP2015512609A (ja) * 2012-03-30 2015-04-27 クアルコム，インコーポレイテッド バックレギュレータの外部ｌｃフィルタ内のピーキングを抑圧するためのシステムおよび方法

Similar Documents

Publication Publication Date Title
US7652603B2 (en) ΔΣ-type AD converter, class-D amplifier, and DC-DC converter
US7612608B2 (en) Sigma-delta based Class D audio or servo amplifier with load noise shaping
US7683707B2 (en) Self-oscillating modulator
US7262658B2 (en) Class-D amplifier system
US20080272945A1 (en) Control system using a nonlinear delta-sigma modulator with nonlinear process modeling
US7605653B2 (en) Sigma-delta based class D audio power amplifier with high power efficiency
TWI477064B (zh) A system and method for reducing distortion in an audio amplification system
EP1865597B1 (fr) Appareil amplificateur
TW200709542A (en) Control circuit of DC-DC converter and its control method
US7075803B2 (en) Frequency stabilization technique for self oscillating modulator
US8928301B2 (en) Pulse width modulation based controller
KR100771854B1 (ko) 전원 전압의 변동을 보상할 수 있는 저 노이즈 직류-직류변환기
Lam et al. Class-D amplifier power stage with PWM feedback loop
CN109478846A (zh) 具有鲁棒稳定的反馈的功率转换器
WO2017067174A1 (fr) Procédé et système d'amélioration de la réponse aux transitoires de charge d'un convertisseur abaisseur de tension en mode tension
CN103378812A (zh) D类放大器
CN104868701A (zh) 电源转换器的混合式补偿电路
US8558610B2 (en) Integrator input error correction circuit and circuit method
WO2007098766A1 (fr) Modulateur numérique auto-oscillant
US11159132B2 (en) Class D amplifier stereo to mono converter
US8937507B2 (en) Self oscillating modulator
JP2013102638A (ja) スイッチング電源回路
EP1830464A1 (fr) Modulateur numérique auto-oscillant
Trescases et al. A digital predictive on-line energy optimization scheme for dc-dc converters
JP2013009516A (ja) スイッチング電源回路

Legal Events

Date Code Title Description
121 Ep: the epo has been informed by wipo that ep was designated in this application
NENP Non-entry into the national phase

Ref country code: DE

122 Ep: pct application non-entry in european phase

Ref document number: 07711255

Country of ref document: EP

Kind code of ref document: A1