WO2006018692A2

WO2006018692A2 - Apparatus and method for conditioning signals produced by physico-chemical sensors

Info

Publication number: WO2006018692A2
Application number: PCT/IB2005/002342
Authority: WO
Inventors: Alessandra Flammini; Daniele Marioli; Andrea Taroni
Original assignee: Universita' Degli Studi Di Brescia
Priority date: 2004-08-18
Filing date: 2005-08-05
Publication date: 2006-02-23
Also published as: ITMI20041654A1; WO2006018692A3

Abstract

An apparatus and method are described for conditioning signals coming from a physico-chemical sensor for the purpose of performing the calculation of one or more of the sensor's electrical characteristics. The sensor is powered by a voltage set by a microcontroller that also provides for setting a threshold voltage to be compared with the output voltage of an integrator circuit connected to the sensor.

Description

"APPARATUS AND METHOD FOR CONDITIONING SIGNALS PRODUCED BY PHYSICO-CHEMICAL SENSORS" FIELD OF THE INVENTION

This invention relates to an apparatus and a method for conditioning signals from physico-chemical sensors.

In particular, this invention proposes an apparatus for estimating at least the resistive and capacitive components of sensors with resistance value within over seven decades (from a few kΩ to tens of GΩ) and small capacitive component (from about 0.1 pF to tens of pF).

BACKGROUND ART

In the field of physical measurement there are a wide range of resistive variation sensors, such as, for example, thermistors, photodetectors and chemical sensors. Suitable interface circuits are used for conditioning the signals coming from these sensors and estimating their variable electrical characteristic as a function of the substances and/or conditions involving each sensor.

One example of an interface module for measuring the electrical characteristics of various types of sensors is described in patent no. US-5854564. This document proposes to evaluate the charging time of a reference capacitor up to a reference voltage for the purpose of determining the value of the desired characteristic, whether it is, for example, a resistance, a capacitance or a current intensity. A "current mirroring" circuit is interposed between the sensor and the reference capacitor.

In particular, in the field of measuring the concentration of gaseous mixtures, MOS technology sensors using innovative materials such as titanium dioxide and molybdenum have recently been the subject of research. These new types of sensors find application in the i realization, for example, of the so-called "electronic noses".

The innovative materials mentioned above provide high sensitivity but exhibit very high resistive values, variable within wide ranges (from a few kΩ up to a few tens of GΩ). Moreover, due to their physical structure, these sensors exhibit a stray capacitance of a few pF that cannot be disregarded with high resistive values.

Currently, the measurement of variable resistive values within wide ranges requires the aid of laboratory instruments, typically a reference voltage generator and a picoammeter capable of measuring the current in the sensor, i.e. highly expensive solutions. If then, in addition to the resistive component, one wishes to measure the capacitive component of the sensor in the presence of the analyte, the impedance-meters currently available do not cover the measurement range of interest. So there is a particularly felt need to make available an instrument that allows characterizing new types of sensors and, in particular, an instrument that allows the realization of compact "electronic noses" consisting of various types of sensors, at a reasonable cost. SUMMARY OF THE INVENTION

With these as the premises, the general object of this invention is to provide efficient and economical apparatus for conditioning signals for the purpose of realizing interfaces with various types of sensors that can be represented by a resistor and a capacitor connected in parallel.

In particular, the object of this invention is to provide an apparatus and a method for measuring at least the resistive component of a sensor with resistive value which varies within a wide range. A further object of this invention is to provide an apparatus and a method for estimating also the value of a possible capacitive component, even if modest, in addition to the sensor's resistive value only. According to a first aspect of this invention, these objects are achieved thanks to an apparatus according to claim 1.

The apparatus according to the invention, is based on an integrator circuit, with programmable excitation and threshold, that converts the resistance to be measured into the period of a square wave. In particular, it is possible to evaluate the pure resistive component of a sensor by using simple microcontrollers for processing the measurements.

The apparatus can be directly interfaced to a sensor without the need for other conditioning stages or A/D converters. The use of special circuit solutions and low-leakage operational amplifiers allows the integration of currents from a few tens of picoamperes up to I mA, while a circuit to optimize the discharge phase of the integration capacitor allows a rapid and accurate restart of the integration procedure. The apparatus is regulated by a microcontroller, such as, for example, a common 8-bit microcontroller that, in addition to managing the logic circuit for activating the phases of measuring and estimating the period, can also manage the heater typically used for the types of sensors to which it is primarily destined, as well as all processing functions.

In the apparatus according to the invention, it is possible to modify the principal parameters during operation, such as the excitation voltage (or power supply) of the sensor, the measurement time (which is, at any rate, maintained below a maximum, even for high resistive values of the sensor), to activate procedures for compensating for the leakage currents of the operational amplifier in the integrator circuit and, if necessary, to exclude the estimation of the capacitive component for a more rapid measurement. The apparatus turns out to be particularly versatile and, if necessary, the same apparatus can also be used as a picoammeter capable of evaluating currents of even a few tens of pA with good precision.

In accordance with a second aspect of this invention, a method is proposed according to claim 15.

The solution identified essentially consists in a measurement method that includes a first phase in which the sensor's resistance value is calculated and, if necessary, a second phase in which the capacitive component is calculated for the sensor that is assumed to be in parallel with the resistive one.

In the first phase, the sensor is powered exclusively with a rigorously constant voltage to estimate the pure resistive component while, during the second phase, the capacitive component is estimated by powering the sensor with a voltage that switches between ground and the above-mentioned constant power supply voltage. A suitable logic circuit, realized, for example, using a Programmable Logic Device (PLD), regulates the alternation between the two phases and, if necessary, the disabling of the second phase in favour of a faster measurement. The measurement time is kept to the order of a few seconds, such as to a maximum of about 3s, even in the presence of very high resistive values (>10 GOhm), suitably varying the threshold voltage that is set in comparison to the output voltage of the integrator circuit.

By implementing suitable prescaling circuits, together with a careful management of the threshold of comparison with the output of the integrator circuit, it is possible to simplify the measurement of the period and significantly limit the variability of the observation time. It is, thus, possible to maintain the observation time between approximately two decades (such as between 10ms and 3s) in the face of more than seven decades of resistance measurement (from less than 10 kOhm to more than 100 GOhm).

By appropriately varying the sensor's excitation voltage, it is also possible to activate a procedure for estimating the leakage current of the integrator circuit.

The same resistance estimation method can also be advantageously used for estimating variable direct currents within wide ranges (from a few pA to some mA).

The estimation of the capacitive component, as well as the procedures for self-compensating for the leakage currents, can constitute useful diagnostic tools. The circuits used must provide good measurement accuracy, a variable sampling time within preset limits over the entire range and the possibility of configuring the measurement process (sampling period, estimate of the capacitive component and measurement parameters). BRIEF DESCRIPTION OF THE DRAWINGS

Additional features and advantages of this invention will be clearer in the following description, which is provided for illustrative and not limiting purposes, with reference to the attached drawings, in which:

- Figure 1 is a block diagram of a signal conditioning apparatus according to this invention;

- Figure 2 is a temporal diagram that shows the performance of consecutive measurement phases of the sensor's resistive component only according to the method of this invention; and

- Figure 3 is a temporal diagram that shows the execution of the consecutive measurement phases of the sensor's resistive component and capacitive component according to the method of this invention.

MODES FOR CARRYING OUT THE INVENTION

Figure 1 shows a block diagram of the apparatus that explains the measurement principle used and the various blocks that comprise it. In the following considerations, we assume a realistic electrical model equivalent to a MOS sensor 1 , i.e. a pure resistor in parallel to a capacitor of a few pF.

A management and control unit 18, consisting, for example, of an 8-bit microcontroller, sets the excitation voltage V_eCc by writing a suitable value (typically on the order of about 1 Volt) to the D/A converter 16 (Digital/Analog converter). In the same way, the microcontroller 18 sets the threshold voltage Vt through a D/A converter 17. The relationship between the threshold voltage Vt and the power supply voltage V_ecc of the sensor 1 is expressed by the formula Vt=-G-V_ecc For the moment, we assume that the apparatus's function is to evaluate only the resistive component of the sensor 1 , a situation illustrated, for example, by the diagram in Figure 2.

During this measurement phase, it is assumed that switch Sl 4 is constantly open and that switch Sl 5 is constantly closed. A Programmable Logic Device (PLD) 12 provides the respective CMIS and RMIS signals necessary for setting these conditions following suitable commands received from the microcontroller 18.

In any case, the representation of switches Sl 4 and Sl 5 is purely for exemplary purposes and has only the aim of making the following explanation more clear. In a real embodiment, switches Sl 4 and Sl 5 do not physically exist and the open and closed conditions are obtained by modifying the output voltage of the D/A converter 16 between V_eCc and zero. In the phase of measuring the resistive component only, the sensor 1 is connected between the continuous voltage V_eCc and the virtual ground of an integrator circuit including the operational amplifier 2, thus eliminating the effect of any capacitive component in parallel with the sensor. The integrator circuit can be implemented with products commonly available on the market, such as, for example, that identified by the code ACF2101 of Texas Instruments, in which the same chip contains the operational amplifier 2, the integration capacitor 3 and switches S4, S5 and S6.

The phase of measuring the resistive component starts at time 0 with switches S4, S5 and S6 open and the integrator output V₀ to ground (V₀(O) = 0). Thanks to the current flowing in the sensor 1 and the presence of the integration capacitor 3 with capacitance O, the output V₀ of the integrator circuit begins to decrease following a linear negative ramp and has the trend shown in the following equation 1 :

(1)

where Rsens is the resistance of the sensor 1.

When the voltage V₀ goes down below the value Vt set through the D/A converter 17, an instant that is indicated with T_c, then the comparator 13 switches, emitting a short pulse F_Out that activates a reset phase through the PLD 12.

The reset phase is activated for a time T_reSet, on the order of about ten microseconds, set by the microcontroller 18. In its turn, the reset phase consist of two phases: - during the first reset phase (time interval TM in Figure 2), switch S4 is closed (RESET signal output by the PLD 12 and acting on switch SA which almost immediately discharges the integration capacitor 3, with capacitance Ci typically on the order of 10OpF, substantially bringing the voltage V₀ to zero;

- during the second phase (time interval T_r2 in Figure 2) a circuit is activated to optimize the discharge of the integration capacitor 3 (HOLD signal output by the PLD 12 and acting on switches S5 and S6).

In fact, the closing resistance of the low-leakage switches S4, S5 and S6, electronically implemented in an integrated circuit, is not insignificant and, for this reason, there are two limitations: a) the capacitor 3 takes a certain amount of time (estimated to be a few μs) to discharge, so switch S4 must be kept closed for a sufficient time; b) the voltage V₀ is not completely canceled, since the impedances of switch S4 and sensor 1 form an inverting configuration on the operational amplifier 2 in which the voltage V_eCc acts as an input. We, thus, have: _τ- _. Rsw

Vo = -Vecc

Rsens where R_sw is the closing resistance of switch S4 that can have values between 1 and 2 kOhm.

The first limitation is remedied by setting an adequate value for Treset with the microcontroller 18, while the second limitation is handled by enabling an optimization circuit for the discharge of the integration capacitor 3, which acts in the second reset phase.

The optimization circuit consists of an operational amplifier 8 in a non-inverting low-pass filter configuration, with gain set by the resistors 9, 10 ad the capacitor 1 1 with capacitance C_r of about 100 pF. The non-inverting pin of the amplifier 8 is connected to switch S6 and the fixing resistor 7 with resistance Rf, while the output is connected to switch S5. When, in the second reset phase, switches S5 and S6 are also closed, the optimization circuit is, thus, enabled and the voltage V₀ is reduced according to the following equation 2:

Rsw

Vo = -Vecc ^^- &

1 + A where Av = 1 + R2/R1 and Ri, R2 are the resistances of the respective resistors 9, 10.

It should be noted that setting a high value of Av limits many of the effects of the non-ideality of the switches. For example, assuming Av = 100, Vecc = I V, a closed switch resistance of 1 kOhm and sensor resistance Rsens of 10 kOhm, the residual voltage V₀ is lower than I mV and, for this reason, can be ignored or, if necessary compensated for by the software of the microcontroller 18 with a table to correct the linearity.

The equation 2 shown above and the relationships set for it, constitutes a simplification of the calculations that should take into account a higher number of factors.

The simplification adopted is based on the fact that the closing and opening resistances of switches S4 and S5 can be considered substantially identical. Switch S6, which does not require a particularly high opening resistance, has a closing resistance of about 100 Ohm and the resistor 7 is selected with a higher value with respect to the closing resistance of switch S6 (for example, Rf = 10 kOhm, approximately). It can, thus, be assumed that the voltage V₀ output from the operational amplifier 2 is substantially equal to the voltage entering the operational amplifier 8 and that the gain Av is essentially defined by the relationship A_v = 1 + R2/Ri without further weighting the calculation with the resistance values of all the switches S4-S6, as well as with the resistance Rf of the resistor 7.

At the end of the reset phase, with a total duration of T_reset, switches S4, S5 and So are opened and the apparatus is ready for another measurement.

By observing the signal F_out output by the comparator 13, a short pulse is obtained every T, where T is linked to sensor resistive component Rsens by the equation 3a shown below:

T - T R = ϊ^ (^3a) sens QQ i and in which T is equal to the sum of T_c and T_reset (T = T_c + T_reset).

Moreover, the following relationships are valid for instant T_c

V T

V (T ) = 5CC_C_ ₌ _Qγ _(3b)

<Λ c' _{R C} ecc sens i

T = GCR (3c)

C i sens As can be seen in equation 3c, the time T_c is linearly linked to sensor resistance Rsens, while G (with G = -Vt/V_eCc) acts a scale factor of the integration capacitor 3, whose value can, thus, be virtually regulated by acting on the voltage values Vt and/or Vecc.

Still with reference to the principle diagram shown in Figure 1 , the management of the CMIS, RMIS, HOLD and RESET signals is entrusted to logical circuits activated by the pulse from Fout and implemented in the PLD 12. Preferably, the same PLD 12 will have a prescaling circuit that, suitably programmed by the microcontroller 18, allows to obtain a signal Fscaied whose period will have sufficient duration to be able to be easily measured from the microcontroller in such a way as to provide a value that is representative of the characteristic measured by the sensor 1.

In addition to estimating the resistance of the sensors, the same apparatus can be used in the case of estimating variable direct currents over a wide range. A direct current l_sens can easily be estimated by injecting the current to be measured in the virtual ground node of the operational amplifier 2 and equations 3a-3c are modified, as a consequence, according to equations 4a-4c shown below:

GCV

I = ' ^ecc (4a) sens T - T reset

- VC GCV

T = —L± = —i-ec^c _(4c) ^c I sens I sens in which T is still equal to the sum of T_c and TVeset (T = T_c + Treset).

Equations 4a-4c can also be used to calibrate the apparatus in the case of estimating the resistance. In fact, varying V_eCc at the beginning of the reset phase and maintaining the voltage threshold Vt fixed, it is varied the current lsens that flows in the sensor 1.

The algebraic sum of the current l_sens and the leakage currents Ii of the operational amplifier 2 is integrated by the capacitor 3. Taking into account h, equations 3a-3c are modified as shown in the following equations 5a-5c:

T T - T

R sens = V ^e = V ^ (5a)

— I 1TC - v tc i ^ecc i l (^vT - T reser ) - v tc i V T I₁T

V (T ) = ^ej^- + -¹^- = V (5b)

^{0 c} R sens C i C i ^{l ;}

T C (R sens 11 - V ecc ) = V tR sens C i ^(5c) in which T is still equal to the sum of T_c and Treset (T = T_c + Treset).

Assuming that we repeat the measurements setting two different values for excitation voltage (V_ΘCc(l ) and V_eCc(2)) and measuring two different values of T (T(I ) and T(2)) and, on the assumption that the value of R_sens remains constant, the estimate of Ii can be calculated in accordance with the following equation 6, keeping in mind that the threshold voltage Vt is negative and assuming leakage current h as entering the terminal of the operational amplifier.

! , _{c v} V_ecc(l)T<:(l) -V_ecc(2)T_c(2) ₍₆₎

^{1 j l} T_c(l)T_c(2)[V_ecc(l)-V_ecc(2)]

It should be noted that the two values of V_eCc should not be too different, since it can't be assumed that the sensor 1 behaves linearly at high variations of power supply voltage V_eCc Moreover, it is better to vary the value of V_ΘCc at the beginning of the reset phase, when switch S4 is closed, so as to limit the effects due to the capacitive component of the sensor 1 , as will be better described below. In any case, it is should be noted that a similar type of correction weighs down the calculation, considering that it must be repeated frequently to take into account the variation of Ii with the temperature.

Concerning the estimate of the capacitive component C_sens of the sensor 1 assumed to be in parallel with Rsens, the only difference, which is managed by the PLD 12, consists in the closing of switch Sl 4 (CMIS signal) and the simultaneous opening of the switch SI 5 (RMIS signal) for the entire duration of the reset phase.

Figure 3 illustrates a diagram showing the two consecutive phases of measuring the resistance and capacitance of the sensor 1.

In this way, by assuming that the resistance output by the operational amplifier 19 is equal to R₀, the shape of the voltage wave Vo output by the integrator circuit is expressed by the following equation 7:

in which the value of τ is given by:

Tec is defined as the instant of reaching the threshold Vt (the period measured during this phase is indicated by T' = T_cc - Treset); disregarding the effect of R₀ with respect to R_sens (Ro+Rsens≡Rsens) and considering that T_cc»Csens-Ro, we obtain the equations 8a, 8b shown below:

T_cc = GQR_sens-C_3ensR_sens (8b)

Taking into account that from the measurement relative to the phase of estimating the resistive component only, we obtained T_c = G-Cj-Rsens, we obtain equations 9a and 9b, which express the estimate of the capacitive and resistive component of the sensor 1 starting from the period measurements of the first phase T, the second phase T' and from the knowledge of the duration of the reset phase Treset settable from the microcontroller, the value of G (G = -Vt/V_eCc) and the capacitance Ci of the integration capacitor 3.

CL se_nns. = (9a)

T T - T R = —£-. = ^ireset (9b)

GC_j GC₁

It is important to emphasize how the method allows to free one from having to know the excitation voltage Vecc In addition, the value G-Ci can be estimated by measuring the resistive component only of a resistor of known value R' by calculating the relation G-Ci = (T-

T_Reset)/R' ) .

Claims

1. An apparatus for conditioning signals from a physico-chemical sensor, including circuits for measuring one or more electrical characteristics of said sensor, characterized by including at least one integrator circuit, to which said sensor is connected, and at least one microcontroller for the programmable setting of at least the power supply voltage of said sensor and at least one threshold voltage to be compared with the output voltage of said integrator circuit.

2. An apparatus according to claim 1 , wherein D/A converters are provided connected to said microcontroller for setting said power supply and threshold voltages.

3. An apparatus according to claim 1 , wherein said integrator circuit includes at least one operational amplifier and at least one integration capacitor.

4. An apparatus according to claim 1 , wherein at least one circuit is provided for the optimization of the discharge phase of said integration capacitor.

5. An apparatus according to claim 1 , wherein switching means are provided to selectively enable or disable at least said circuit for the optimization of the discharge phase of said integration capacitor and wherein one or more of said switching means have a significant closing resistance during the discharge phase of said integration capacitor.

6. An apparatus according to claim 4 or 5, wherein said optimization circuit includes one operational amplifier set with a high gain value to compensate for the closing resistance of said switching means.

7. An apparatus according to claim 1 , wherein the electrical characteristics measured include at least the resistance value of said sensor.

8. An apparatus according to claim 7, wherein the resistance value measured for said sensor varies within a range of at least seven decades with values between less than 10 kOhm and more than 10 GOhm.

9. An apparatus according to claim 7 or 8, wherein the time for measuring the resistance value of said sensor is kept within an interval of two decades with times between about 10ms and 3s.

10. An apparatus according to claim 1 , wherein the electrical characteristics measured include at least the capacitance value of said sensor.

1 1. An apparatus according to claim 1 , wherein the electrical characteristics measured include at least the value of the current flowing in said sensor.

12. An apparatus according to claim 1 , wherein a programmable logic device is provided to control the measurements of at least two distinct electrical characteristics in at least two consecutive measurement phases.

13. An apparatus according to claim 12, wherein said two distinct electrical characteristics measured in at least two consecutive phases include the resistance and capacitance of said sensor.

14. An apparatus according to claim 1 , wherein a programmable logic device is provided to supply said microcontroller with a scaled signal representative of the characteristic measured for said sensor.

15. A method for conditioning signals coming from a physico- chemical sensor, wherein the measurement of one or more electrical characteristics of said sensor is provided, characterized by providing for the connection of said sensor to at least one integrator circuit and the programmable setting, by means of at least one microcontroller, of at least one power supply voltage of said sensor and at least one threshold voltage to be compared with the output voltage of said integrator circuit.

16. A method according to claim 15, wherein said power supply and threshold voltages are set by said microcontroller.

17. A method according to claim 15, wherein charge and discharge phases are provided for a capacitor in said integrator circuit.

18. A method according to any of claim 15 to 17, wherein the time necessary for the discharge of said capacitor is set at least in part by said microcontroller.

19. A method according to claim 15, wherein the optimization of the discharge phase of said integration capacitor is provided.

20. A method according to claim 15, wherein said optimization circuit is selectively enabled or disabled by switching means and wherein one or more of said switching means have a significant closing resistance during the discharge phase of said integration capacitor.

21. A method according to claim 19 or 20, wherein the optimization of the discharge phase of said capacitor includes compensating for the closing resistance of said switching means.

22. A method according to claim 15, wherein at least measuring the resistance value of said sensor is provided.

23. A method according to claim 22, wherein the resistance value measured for said sensor varies within a range of at least seven decades for values between less than 10 kOhm and more than 10 GOhm.

24. A method according to claim 22 or 23, wherein the time for measuring the resistance value of said sensor is kept within a range of two decades with times between about 10ms and 3s.

25. A method according to claim 15, wherein at least measuring the capacitance of said sensor is provided.

26. A method according to claim 15, wherein at least measuring the current flowing in said sensor is provided.

27. A method according to claim 15, wherein calculating the leakage currents in said integrator circuit is provided to calibrate the apparatus for conditioning signals coming from said sensor.

28. A method according to claim 27, wherein the calculation of the leakage currents in said integrator circuit is made by successively setting two distinct values for the power supply voltage of said sensor.

29. A method according to claim 15, wherein the measurements of at least two distinct electrical characteristics are made in at least two consecutive measurement phases.

30. A method according to claim 15, wherein said two distinct electrical characteristics measured in at least two consecutive phases include the resistance and capacitance values of said sensor.

31. A method according to claim 15, wherein the representative value of the characteristic measured for said sensor is a function of the period of a square wave.

32. A method according to claim 15, wherein the value of the characteristic of said sensor is calculated by said microcontroller on the basis of a scaled signal representative of the characteristic measured for said sensor.