WO2005064877A1

WO2005064877A1 - Digital rf modulator comprising a phase or frequency return path

Info

Publication number: WO2005064877A1
Application number: PCT/EP2004/014342
Authority: WO
Inventors: Gérard Marque-Pucheu
Original assignee: Eads Telecom
Priority date: 2003-12-24
Filing date: 2004-12-16
Publication date: 2005-07-14
Also published as: FR2864737B1; FR2864737A1

Abstract

A digital modulator comprises a phase or frequency modulation path providing for the upconversion of a phase or frequency modulation component (FM), and a phase or frequency return path providing for the downconversion of the phase or frequency modulation component. The modulator relies on a technique similar to the technique of scrambling (or 'dithering'). The technique uses a purely digital frequency conversion method, which creates noise (Sdiv) outside the band of the useful signal, which is subsequently filtered (28), but which makes it possible to reveal signals (N - No) exhibiting a very low frequency excursion with respect to the frequency of the radiofrequency carrier (Fo).

Description

DIGITAL RF MODULATOR COMPRISING A PHASE OR FREQUENCY RETURN PATH

BACKGROUND OF THE INVENTION

1. Technical Field

The present invention pertains in a general manner to digital modulation and relates more particularly to a digital radiofrequency modulator (RF) comprising a phase or frequency return path.

The function of such a modulator is to generate a phase- or frequency-modulated RF signal, suitable for RF transmission via an antenna or a cable.

The invention finds applications, in particular, in the RF transmitters of mobile stations or fixed stations of a radiocommunication system, for example a Professional Mobile Radiocommunication system (PMR system) .

Related Art

Current radiocommunication systems conventionally use, for the transmission of digital data coding an audio signal or, more generally, information of any kind, so- called constant envelope modulations. With such modulations, the data transmitted are not carried by the amplitude of an RF carrier but by its phase or its frequency. In order to send more information within a frequency band of given width, assigned to a transmission channel, and hence to increase the spectral efficiency of the system while complying with the constraints related to the splitting of the frequency spectrum, there may also be provision for amplitude modulation, in addition to phase or frequency modulation. Thus, the signal generated by the modulator may exhibit a composite modulation, comprising both a Phase Modulation component (PM component) or Frequency Modulation component (FM component), and an Amplitude Modulation component (AM component) .

Digital modulators are now a well known technology in the state of the art. For linear modulations like π/4- DQPSK (standing for "Differential Quadrature Phase- Shift Keying") modulation or QAM (standing for "Quadrature Amplitude Modulation") modulations, the in- phase and quadrature modulations (or I-Q modulations) are used most. These I-Q modulations use two baseband signals called the in-phase component (I) and the quadrature component (Q) respectively, which are in quadrature.

As a variant, for constant envelope modulations such as GMSK (standing for "Gaussian Minimum Shift Keying") or other variants of CPM (standing for "Continuous Phase Modulation") , it is frequent practice to use a modulation of frequency by numerical control of a VCO (standing for "Voltage Controlled Oscillator") . It is also known to combine this frequency modulation with an amplitude modulation according to techniques known by the name of EER techniques (standing for "Envelope Elimination and Restoration") . In this latter type of modulator, two techniques are commonly used.

In the first technique, the PM or FM component, as well as a continuous component corresponding to the carrier frequency of the RF signal, are generated digitally and then converted into an analogue signal by a digital/analogue converter (or DAC) . This signal is injected onto the (analogue) control input of the VCO through a low-pass filter. The pass band of this filter is greater than the width of the modulation, so as to avoid the latter being distorted by the filtering. This technique is very suitable as long as the slope of the VCO is not too big. Otherwise, the wideband noise characteristics of the DAC may become difficult to comply with under good cost and consumption conditions. Moreover, it is then necessary to employ a phase or frequency return loop for the centring of the carrier frequency of the modulation.

The second technique, known by the name of two-point injection, divides the difficulties. Specifically, the PM or FM component is generated digitally and then converted into analogue by a DAC, while the continuous component is generated separately. The continuous component is then filtered in a very narrow band, limited only by the switching times required for changing channel. Conversely, the PM or FM component is filtered by a filter whose bandwidth is much bigger, but whose gain is small since the VCO control voltage excursion corresponding to the PM or FM modulation width is much smaller than the voltage excursion corresponding to the continuous component, for a scan of all the channels. The voltage corresponding to the continuous component may either be generated by a device of the analogue kind such as a Phase Locked Loop (or PLL) , or generated digitally then converted into analogue by an auxiliary DAC called a continuous component DAC. In the latter case, the difficulty is divided since the noise generated by the modulation DAC is hardly filtered, but strongly attenuated, while the noise generated by the continuous component DAC is hardly attenuated, but strongly filtered.

In all cases, it is necessary to tailor the gain of the means for injecting the PM or FM component so as to take account of the variations of the slope of the VCO (voltage/frequency characteristic) which is not generally constant over the entire band covered, and which furthermore may vary as a function of the supply voltage, of the temperature and of the aging of the components.

Moreover, in the case of a modulation of the EER type, the AM component is generated digitally, then converted by a DAC, and finally introduced in general by varying the gain of a Power Amplifier (or PA) amplifying the output from the modulator. Other imperfections may then come in such as a time shift between the AM component and the PM or FM component, as well as imperfections of the amplifier itself like the AM-PM or AM-FM conversions .

It is easy to demonstrate that all of these imperfections may be corrected by digital modulation of the modulating signal, according to an amplitude and/or phase- or frequency-adaptive predistortion technique. This modification presupposes the existence of a return path making it possible to compare the output signal from the modulator with the desired signal, this comparison making it possible to deduce the modulating signal necessary for the obtaining of the desired output signal.

An exemplary implementation of this return path known in the state of the art consists in performing an I-Q demodulation of the output signal, followed by a digitization of the baseband signal thus obtained. The digitized signal obtained makes it possible, by comparison with the desired signal, to deduce the predistortions to be imposed on the modulating signal via suitable digital processing.

This method nevertheless has several practical drawbacks. In particular, it requires the effecting of a frequency transposition to bring the output signal to baseband (or to low intermediate frequency) . Now, this transposition requires the presence of an oscillator whose frequency generally varies with the frequency of the channel, and whose spectral purity must be compatible with the imperfections that one seeks to correct. In the case of narrowband modulations, whose spectral specifications in the vicinity of the carrier are comparatively much more severe than relatively wider- band modulations, the specifications of this transposition oscillator may become incompatible with economic implementation, compactness and low consumption.

A solution which is at first sight more economical for implementing the return path consists in digitizing the output signal not in an I-Q form, but in a Phase- Amplitude form. This option is particularly attractive in the case of modulations exhibiting a low depth of amplitude modulation. In this case in fact, the imperfections of AM-AM conversion type are generally negligible, if the gain of the output power amplifier is modulated by the drain voltage of an MOS (standing for "Metal Oxide Semiconductor") power transistor. An Analogue/Digital Converter (or ADC) of fairly mediocre quality is then sufficient for the digitization of the AM component at the output of the amplifier. Specifically, the only goal sought is • correct synchronism between the AM component and the PM or FM component, that is to say a determined time shift between the AM component measured at the output of the amplifier and the desired AM component. This comparison (of phase shift) requires no superior performance on the part of the ADC, neither from the point of view of linearity, nor from the point of view of noise since the latter subsequently undergoes extremely severe (digital) filtering.

The invention concentrates exclusively on the phase or frequency return path. The latter is the most useful since it makes it possible to correct most imperfections of the output signal: error in the central frequency of the radio channel, error in the slope of the VCO and hence error in the width of the phase or frequency modulation, time shift between the AM component and the PM or FM component, and AM-PM or AM-FM conversion in the output power amplifier.

The techniques based on a prior frequency conversion must be excluded for the same reason as that mentioned earlier. The analogue/digital conversion must therefore be done solely with the aid of pulse counting techniques, so as to allow the direct injection of the output signal into the return path of the digital modulator. Techniques of this kind have been described in patent US 4,310,795 and in patent US 6,269,135.

These techniques are not however suitable for direct determination of frequency making it possible to reveal the sought-after imperfections.

Specifically, with a modulation of the type, for example, of the modulation used for the APCO Phase 2 standard (TIA-905.BAAA) , the frequency deviation is ± 4 kHz for a carrier frequency of 800 MHz. An error of 2% in the slope of the VCO, which has prejudicial effects on the quality of the spectrum, therefore corresponds to a frequency shift of 80 Hz, that is to say to 10^~7 with respect to the carrier frequency. It is therefore necessary to reveal a difference of counting of one pulse every 10 million (0.1 ppm). Accordingly, the quantization stepsize proposed in patent US 6,269,135 is inappropriate.

To allay this drawback, a frequency transposition technique making it possible to decrease the value of the carrier frequency without thereby decreasing the deviation is desirable in order to increase the fineness of the modulation.

BRIEF DESCRIPTION OF THE INVENTION

To this end, a first aspect of the invention proposes a digital modulator comprising a phase or frequency modulation path providing for the upconversion of a phase or frequency modulation component of the baseband on a radiofrequency carrier, and a phase or frequency return path providing for the downconversion of the phase or frequency modulation component from the modulated radiofrequency carrier. The phase or frequency return path comprises: a coupler designed to obtain a first signal reflecting the modulated radiofrequency carrier; - a frequency divider with variable division ratio, with an input receiving the said first signal, an input for controlling the division ratio receiving a scrambling signal, and an output delivering a second signal corresponding to the said first divided signal; a pulse counter, with a counting input receiving the said second signal, an initialization input receiving a periodic reference signal, and an output delivering a counting value at each period of the said reference signal, the said counting value corresponding to the number of pulses of the said second signal that are received during a period of the said reference signal; a digital subtractor, with a first input receiving the said counting value, a second input receiving a constant value (No) associated with the central frequency of the radio channel, and an output delivering a measured signal.

The scrambling signal is generated by a Sigma-Delta generator so that the following relation is satisfied: Fo = Frefx No x n where Fo is the central frequency of the radio channel, where Fref is the said reference frequency, where No is the said constant value associated with the central frequency of the radio channel, and where n is the mean value of the division ratio of the frequency divider with variable division ratio. The invention thus relies on a technique similar to the technique of scrambling (or "dithering") , which is conventional for the analogue/digital conversion of voltages. By adding noise outside the band of the useful signal, it is possible to reveal signals whose amplitude is much smaller than the quantization stepsize of an analogue/digital converter.

The technique described by the present invention uses a purely digital frequency conversion method, which creates noise outside the band of the useful signal, which is subsequently filtered, but which makes it possible to reveal signals exhibiting a very low phase or frequency deviation with respect to the frequency of the radiofrequency carrier.

In one embodiment the phase or frequency return path furthermore comprises a digital processing module, with an input receiving the measured signal, and at least one output delivering a control of at least one parameter of the phase or frequency modulation path.

Preferably, the phase or frequency return path furthermore comprises at least one decimation filter, disposed between the output of the digital divider and the digital processing module, and adapted for extracting the continuous component of the measured signal.

The digital processing module may be adapted for calculating a correlation function relating the measured signal and the phase or frequency modulation component .

When the phase or frequency modulation path comprises a phase- or frequency-adaptive predistortion module, the control of the correction of the imperfections in the phase or frequency modulation path may be adapted for modifying at least some of the predistortion coefficients of the predistortion module, so as to correct the phase or frequency nonlinearities.

As a variant or as a supplement, the digital processing module may be adapted for modifying the value No and/or the value n , so as to ensure the centring of the radiofrequency carrier on the desired central frequency of the channel.

When the modulator comprises an amplitude modulation path providing for the upconversion of an amplitude modulation component of the baseband on the radiofrequency carrier, the digital processing module may be adapted for calculating a correlation function relating the measured signal and the amplitude modulation component, and for generating accordingly a command for correcting at least one parameter of the phase or frequency modulation path and/or of the amplitude modulation path, so as to compensate for a time shift between the phase or frequency modulation component on the one hand, and the amplitude modulation component on the other hand.

In this case, the phase or frequency modulation path and the amplitude modulation path are designed according to an EER technique presented hereinabove.

Preferably, the Sigma-Delta generator is a Sigma-Delta generator of order 5, with 1 bit.

A second aspect of the invention relates to a radiofrequency transmitter comprising a modulator according to the first aspect.

According to a third aspect of the invention, there is proposed a mobile terminal of a radiocommunications system, comprising a radiofrequency transmitter according to the second aspect. Finally, a fourth aspect of the invention pertains to a base station of a radiocommunications system, comprising a radiofrequency transmitter according to the second aspect.

BRIEF DESCRIPTION OF THE DRAWINGS

Other characteristics and advantages of the invention will become further apparent on reading the description which follows. The latter is purely illustrative and should be read in conjunction with the appended drawings, in which:

Figure 1 is a block diagram of an exemplary embodiment of a modulator according to the present invention;

Figure 2 and Figure 3 are graphs of the scrambling signal generated by the spectrum of the Sigma- Delta generator;

Figure 4 and Figure 5 are graphs of the signal measured by the phase or frequency return path in an exemplary modulator according to the present invention;

Figure 6 and Figure 7 are graphs of the correlation function relating to the signal measured by the phase or frequency return path and the original frequency or phase modulation component, after an operation of filtering and decimation of the said measured signal, for an exemplary modulator according to the present invention;

Figure 8 is a graph of the same function, but after several operations of filtering and decimation of the measured signal; and Figure 9 is a graph showing the variation of the correlation function mentioned above as a function of time, when the slope of the VCO is equal to a nominal value and when it is modified by 2% with respect to this nominal value.

DESCRIPTION OF PREFERRED EMBODIMENTS

The block diagram in Figure 1 illustrates an exemplary modulator according to an embodiment of the present invention.

The modulator comprises a frequency modulation path and an amplitude modulation path that are designed according to an EER technique. These two paths receive for example the modulation information in the form of an in-phase digital signal I and a quadrature digital signal Q. These signals are converted by a conversion module 11, into an amplitude modulation component AM and a frequency modulation component FM (or a phase modulation component PM, which amounts to the same) .

An amplitude- and frequency-adaptive predistortion module 12 receives the components AM and PM as input, and outputs components AM' and FM' respectively, corresponding to the predistorted components AM and FM. The module 12 carries out a digital predistortion. This predistortion is said to be adaptive in the sense that the predistortion coefficients used are adaptive.

An interpolation module 13 thereafter receives the components AM' and FM' as input, and outputs components AM" and FM" respectively, corresponding to the components AM' and FM' oversampled by interpolation. The function of the module 13 is to increase the speed of operation of the downstream part of the modulator with respect to the upstream part.

The component AM" is converted into an analogue signal S_AM, by means of a digital/analogue converter 14a followed by a low-pass filter 15a. The converter 14a is for example a 1-bit Sigma-Delta converter. The filter 15a is for example a second or third order RLC analogue filter.

Likewise, the component FM" is converted into an analogue signal S_FM by means of a digital/analogue converter 14b followed by a low-pass filter 15b. The converter 14b is for example a 1-bit Sigma-Delta converter. The filter 15b is for example a second or third order RLC analogue filter.

The signal S_FM is provided on a first input of a summator 17 (analogue adder) .

The modulator furthermore comprises a register 10 storing a digital value Fo. The value Fo corresponds to the central frequency of a radio channel managed by the modulator. In practice, the register 10 stores a plurality of values such as the value Fo, corresponding respectively to the central frequencies of the various radio channels managed by the modulator.

The value Fo is converted into an analogue signal S_Fo, by means of a digital/analogue converter 14c followed by a low-pass filter 15c. The converter 14c is for example a 1-bit Sigma-Delta converter. The filter 15c is for example a second or third order RLC analogue filter. This signal S_Fo is delivered on a second input of the summator 17.

The summator 17 outputs a signal Svco which thus contains the continuous component corresponding to the central frequency of the radio channel as well as the frequency modulation component.

The signal Svco is provided as input to a VCO 18, which provides for the upconversion of the frequency modulation path, that is to say the frequency conversion from the baseband to the RF carrier. Stated otherwise, the VCO 18 associated with the summator 17 carries out the injection of the frequency modulation component FM into an RF signal corresponding to the RF carrier for the radio channel considered, whose central frequency is determined by Fo.

The signal delivered by the VCO 18 is input to a power amplifier 19 which outputs a signal Sout. The signal Sout is an RF signal which is suitable for radio transmission via an antenna 20 to which it is input.

The amplitude modulation component is injected into the signal Sout by controlling the gain of the amplifier 19. Accordingly, the signal S_m is provided on a gain control input of the amplifier 19.

The modulator also comprises a frequency return path, which provides for the downconversion, that is to say from the RF carrier to the baseband. A function of this return path is to recover the frequency modulation from the signal Sout which corresponds to the modulated RF carrier. The aim is to compare the frequency modulation actually present in the signal Sout with the desired frequency modulation, so as to deduce therefrom commands for correcting imperfections to be applied to one and/or the other of the modulation paths. These corrections make it possible in particular to correct phase nonlinearities due to variations of the slope of the VCO, AM/PM conversions, and a delay between the amplitude modulation component and the frequency modulation component, which would be generated in the modulation paths.

It will be noted that the modulator may also comprise an amplitude return path, which is not the subject of the present description. In the description of the operation of the frequency return path that follows, the signals and their frequency are spoken of interchangeably.

The frequency return path comprises a coupler 21 designed in such a way as to tap off a part of the power of the signal Sout at the output of the modulator. One thus obtains a radiofrequency signal F reflecting the signal Sout and which contains in particular the frequency modulation. This signal F is input to a frequency divider with variable division ratio 23, through an attenuator 22. A function of the attenuator 22 is to bring the signal F to a power level suitable for the operation of the divider 23. The attenuator 22 can also eliminate the amplitude variations of the signal F due to the amplitude modulation that it additionally contains.

Typically, the divider 23 divides the frequency of the signal received at input by n or n+1, where n is an integer determined, as a function of the value for example 1 or 0 respectively, from a control signal Sdiv received on a division ratio control input.

In an example, n is equal to 4. Thus, the divider 23 divides either by 4, or by 5, depending on the value of the signal Sdiv. The mean value of the division ratio of the divider 23 is denoted n .

F The signal — output by the divider 23 is provided on n the counting input of a pulse counter 25. The pulse counter 25 receives a periodic signal Fref at a frequency Fref on an initialization input. The pulse counter 25 is adapted for counting, without overflow, F the number of pulses of the signal — during a period n of the signal Fref. The counting value N thus obtained is stored, at each period of the signal Fref, in a register 26. The number N is given by the following relation: F N = -—" (I⁾ n x Fref

It will be noted that the number of bits of the pulse counter 25 is sufficient to count without overflow all the pulses at the maximum frequency Fmax permitted at the output of the modulator after division by the minimum ratio n of the divider 23, doing so for a period of the reference frequency Fref. For example, for a reference frequency Fref equal to 6.5 MHz and for n equal to 4, a 5-bit counter will make it possible to process a frequency of the output signal Sout of up to: Fmax = 32 x 4 x 6.5 MHz = 832 MHz (2)

At each pulse of the reference signal Fref, the content of the pulse counter 25 is loaded into a register. At each instant, the value N stored in the register 26 is provided on a first input of a digital subtractor 27.

As trie pulses leaving the divider 23 are asynchronous with respect to the reference signal Fref, an anticoincidence device, well known to the person skilled in the art, will be used to generate the pulse for transferring the value N stored in the register 26 to the subtractor 27. For example, this pulse may be generated after a fixed time interval following the first complete pulse, following the rising edge of the pulse Fref. This fixed time interval is chosen in such a way that the state of the pulse counter 25 is stable. Thus the subtractor 27 contains a valid data item on the falling edge of the pulse Fref.

To allow the implementation of the invention, the channel frequency Fo is related to the frequency Fref and to the mean value n of the division ratio of the divider 23 by the following relation: Fo = Fref x No x n ( 3)

where No is a value, corresponding preferably to an integer, stored in the register 10.

At each instant, the number No is provided on a second input of the digital subtractor 27. The value No is then subtracted from the value N contained in the register 26. The output of the digital subtractor 27 delivers a numerical value N - No such that:

N - No = ^~F° (4) n Fref

Stated otherwise, the result N - No of the -subtraction is a measured signal which corresponds, to within the factor — xFref, to the frequency deviation F - Fo, that n is to say to the frequency modulation actually present in the RF output signal Sout. The means, described hereinabove, of the frequency return path therefore make it possible to recover the frequency modulation actually present in the output signal Sout.

It is then sufficient to compare the measured signal N - No with the original frequency modulation component FM to detect frequency imperfections of the modulator.

In the case where a phase modulation rather than a frequency modulation is considered, it is sufficient to provide a prior step of integration of the measured signal N - No in the processing module 29. The integrated signal N - No may then be compared with the phase modulation component PM to detect phase imperfections of the modulator.

The signal Sdiv, which controls the value of the division ratio n or n + 1 of the divider 23, is produced with the aid of a device 24 which here is a 1-bit Sigma-Delta (Σ - Δ) generator. This device 24 receives as input the value Fref x No x n stored in the register 10.

Stated otherwise, the value N - No obtained as output from the subtrator 27 is the Σ - Δ approximation of the frequency modulation actually present in the RF output signal Sout .

The signal Sdiv is a scrambling signal (or "dithering signal"), which makes it possible, in a manner known per se, to reveal small effects present in a signal. In the application which is made thereof here, the scrambling effect thus obtained makes it possible to reveal the frequency modulation and the corresponding distortions in the signal F. Specifically, the pseudorandom (and hence nonregular) nature of the time F intervals between the pulses of the signal ■=■ at the n output of the divider 23 causes the appearance of a variation in the statistical distribution of the counting values N, which makes it possible to detect much lower variations in frequency than those that can be detected by a process of regular counting such as those employed in the prior art cited in the introduction.

Figure 2 gives the spectrum of the signal Sdiv generated by the Sigma-Delta device 24, and Figure 3 shows the central part of this spectrum for further details, in a given example.

In this example, the central frequency Fo of the channel is equal to 807 MHz, the reference frequency Fref is equal to 6.5 MHz, No is equal to 31, n is equal to 4 and n is therefore equal to 4.004962779. As a variant, it would also be possible to choose a value No equal to 30, n then being equal to 4.138461538. It may be seen in particular in the graph of Figure 3 that the spectrum of the Sigma-Delta device 24 exhibits a depth of around -140 dB, with nevertheless a useful band whose width is around 8 kHz. These characteristics make it possible to detect a distortion of the order of 80 Hz, representing 1% of the bandwidth of 8 kHz of the radio channel, on an RF carrier at around 800 MHz.

The value F - Fo delivered by the output of the digital divider 27 is input to a digital processing module 29 through a set of decimation filters 28. The filters 28 perform successive operations of filtering and decimation, so as to eliminate the out-of-band noise introduced by the Sigma-Delta device 24 via the divider 23. Stated otherwise, these operations are aimed at extracting the continuous component (at f = 0) of the measued signal N - No.

A function of the digital processing module 29 is to detect imperfections of the modulator and to generate commands CTRL for correcting parameters of the frequency modulation path and/or of the amplitude modulation path accordingly. The detections are carried out, on the basis of the value F - Fo on the one hand, and of the desired modulation components FM and/or AM on the other hand. In particular, the module 29 is adapted for modifying the adaptive predistortion coefficients of the module 12. In case of shift with respect to the central frequency Fo of the radio channel, it may also modify the value No and/or the value n, which are stored in the register 10.

A simple counting technique such as that envisaged in the prior art cited in the introduction shows up a single difference of counting only after 10 million pulses (the inverse of 0.1 ppm), that is to say at a recurrence frequency of around 80 Hz when Fo (and hence Sout) is of the order of 800 MHz. It is therefore very difficult to reveal in this way the slightest distortion in a modulating signal S_M whose frequency is 8 kilo-symbols per second.

On the other hand, the good performance in this regard of the modulator according to the invention will now be commented on with reference to the graphs of Figures 4 to 9.

The graph of Figure 4 gives the spectrum of the measured signal N - No, and Figure 5 shows the central part of this spectrum. In this particular configuration, the divider 23 is controlled by a Σ - Δ generator of order 5. The input signal is a signal frequency-modulated by a sinusoidal signal of 2 kHz with a maximum deviation of 3 kHz.

The slope error of the oscillator 18 can be determined by performing in the module 29 the calculation of the correlation function, denoted (N - No)®FM, relating the measured signal N - No and the modulating signal FM.

Figures 6 and 7 show the result of the correlation (N - No)®FM after the first step of filtering and decimation implemented in the set of filters 28, in a specific example. In this example, the total frequency deviation is ± 3 kHz. The continuous component of the correlation function (N - No)®FM is proportional to the slope of the VCO. Figure 7, which shows the curve of Figure 6 in detail, makes it possible to note the good signal-to- noise ratio that is obtained.

Figure 8 gives a curve comparable to that illustrated by Figures 6 and 7, but corresponds to the correlation

(N - No)®FM after the entirety of the various steps of filtering and decimation implemented in the set of filters 28.

Finally, the curve in Figure 9 represents the result of the correlation (N - No)<S>FM (after all the steps of filtering and decimation) when the slope is the nominal slope, and when this slope increases by 2%, that is to say when the deviation is ± 3.06 kHz. This curve illustrates the alacrity with which a variation as minor as 2% of the slope of the VCO is detected. Speci ically, considering that the change of value of the slope of the VCO occurs at the instant t=0, it may be seen that this variation can be sensed after only a few milliseconds.

It will be appreciated that other imperfections of the modulation path may be detected from the signal N - No.

For example, the AM/PM conversions can be determined by performing in the module 29 the calculation of the correlation function, denoted (N - No)®AM, relating the measured signal N - No and the amplitude modulation signal AM.

Likewise, the delay of the frequency modulation component in the radiofrequency output signal can be determined by performing in the module 29 the calculation of the correlation function, denoted (N-NO)<8> , relating the measured signal N - No and the derivative of the frequency modulation signal FM.

Claims

1. Digital modulator comprising a phase or frequency modulation path providing for the upconversion of a phase or frequency modulation component (FM) of the baseband on a radiofrequency carrier (Fo), and a phase or frequency return path providing for the downconversion of the phase or frequency modulation component from the modulated radiofrequency carrier (Sout) , wherein the phase or frequency return path comprises: - a coupler designed to obtain a first signal (F) reflecting the modulated radiofrequency carrier; a frequency divider (23) with variable division ratio, with an input receiving the said first signal, an input for controlling the division ratio receiving a scrambling signal (Sdiv) , and an F output delivering a second signal ( — ) n corresponding to the said first divided signal; a pulse counter (25) , with a counting input receiving the said second signal, an initialization input receiving a periodic reference signal (Fref) , and an output delivering a counting value (N) at each period of the said reference signal, the said counting value corresponding to the number of pulses of the said second signal that are received during a period of the said reference signal; a digital subtractor (27), with a first input receiving the said counting value, a second input receiving a constant value (No) associated with the central frequency of the radio channel, and an output delivering a measured signal (N - No) , and wherein the scrambling signal is generated by a Σ-Δ generator (24) so that the following relation is satisfied: Fo = Fref x No x n where Fo is the central frequency of the radio channel, where Fref is the said reference frequency, where No is the said constant value associated with the central frequency of the radio channel, and where n is the mean value of the division ratio of the frequency divider with variable division ratio.

2. Digital modulator according to Claim 1, wherein the phase or frequency return path furthermore comprises a digital processing module, with an input receiving the measured signal, and at least one output delivering a control of at least one parameter of the phase or frequency modulation path.

3. Digital modulator according to Claim 2, wherein the phase or frequency return path furthermore comprises at least one decimation filter (28), disposed between the output of the digital divider and the digital processing module, and adapted for extracting the continuous component of the measured signal.

4. Digital modulator according to Claim 2 or Claim 3, wherein the digital processing module is adapted for calculating a correlation function relating the measured signal and the phase or frequency modulation component.

5. Digital modulator according to any one of Claims 2 to 4, wherein the phase or frequency modulation path comprises a phase- or frequency-adaptive predistortion module (12), and in which the control of the correction of the imperfections in the phase or frequency modulation path adapted for modifying at least some of the predistortion coefficients of the predistortion module.

6. Digital modulator according to any one of Claims 2 to 5, wherein the digital processing module is adapted for modifying the value No and/or the value n .

7. Digital modulator according to any one of Claims 2 to 6, further comprising an amplitude modulation path providing for the upconversion of an amplitude modulation component (AM) of the baseband on the radiofrequency carrier, and wherein the digital processing module is adapted for calculating a correlation function relating the measured signal and the amplitude modulation component, and for generating accordingly a command for correcting at least one parameter of the phase or frequency modulation path and/or of the amplitude modulation path.

8. Digital modulator according to Claim 7, wherein the phase or frequency modulation path and the amplitude modulation path are designed according to an EER technique.

9. Digital modulator according to any one of the preceding claims, wherein the Σ-Δ generator is a Sigma- Delta generator of order 5, with 1 bit.

10. Radiofrequency transmitter comprising a modulator according to any one of the preceding claims.

11. Mobile terminal of a radioco munications system, comprising a radiofrequency transmitter according to Claim 10.

12. Base station of a radiocommunications system, comprising a radiofrequency transmitter according to Claim 10.