WO2004019656A2 - Modulation spatiale de canal audio - Google Patents

Modulation spatiale de canal audio Download PDF

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Publication number
WO2004019656A2
WO2004019656A2 PCT/US2003/024570 US0324570W WO2004019656A2 WO 2004019656 A2 WO2004019656 A2 WO 2004019656A2 US 0324570 W US0324570 W US 0324570W WO 2004019656 A2 WO2004019656 A2 WO 2004019656A2
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WIPO (PCT)
Prior art keywords
input
variable
input signals
correlation
process according
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PCT/US2003/024570
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English (en)
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WO2004019656A3 (fr
Inventor
Mark Franklin Davis
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Dolby Laboratories Licensing Corporation
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Priority to US10/522,515 priority Critical patent/US7660424B2/en
Priority to JP2004530877A priority patent/JP4434951B2/ja
Priority to CN03817877XA priority patent/CN1672464B/zh
Priority to BRPI0305746-1A priority patent/BRPI0305746B1/pt
Application filed by Dolby Laboratories Licensing Corporation filed Critical Dolby Laboratories Licensing Corporation
Priority to CA2494454A priority patent/CA2494454C/fr
Priority to DK03770229T priority patent/DK1527655T3/da
Priority to AU2003278704A priority patent/AU2003278704B2/en
Priority to MXPA05001413A priority patent/MXPA05001413A/es
Priority to DE60308876T priority patent/DE60308876T2/de
Priority to BR0305746-1A priority patent/BR0305746A/pt
Priority to EP03770229A priority patent/EP1527655B1/fr
Publication of WO2004019656A2 publication Critical patent/WO2004019656A2/fr
Publication of WO2004019656A3 publication Critical patent/WO2004019656A3/fr
Priority to IL165941A priority patent/IL165941A/en
Priority to HK05106538A priority patent/HK1073963A1/xx

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04S—STEREOPHONIC SYSTEMS 
    • H04S5/00—Pseudo-stereo systems, e.g. in which additional channel signals are derived from monophonic signals by means of phase shifting, time delay or reverberation 
    • H04S5/005—Pseudo-stereo systems, e.g. in which additional channel signals are derived from monophonic signals by means of phase shifting, time delay or reverberation  of the pseudo five- or more-channel type, e.g. virtual surround
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04S—STEREOPHONIC SYSTEMS 
    • H04S3/00—Systems employing more than two channels, e.g. quadraphonic
    • H04S3/002—Non-adaptive circuits, e.g. manually adjustable or static, for enhancing the sound image or the spatial distribution
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04S—STEREOPHONIC SYSTEMS 
    • H04S3/00—Systems employing more than two channels, e.g. quadraphonic
    • H04S3/02—Systems employing more than two channels, e.g. quadraphonic of the matrix type, i.e. in which input signals are combined algebraically, e.g. after having been phase shifted with respect to each other

Definitions

  • the invention relates to audio signal processing. More particularly the invention relates to translating M audio input channels representing a soundfield to N audio output channels representing the same soundfield, wherein each channel is a single audio stream representing audio arriving from a direction, M and N are positive whole integers, and M is at least 2 and N is at least 3, and N is larger than M.
  • a spatial translator in which N is greater than M is usually characterized as a "decoder”.
  • Dolby "Dolby", "Pro Logic” and “Surround EX” are trademarks of Dolby Laboratories Licensing Corporation. To one degree or another, these systems provide enhanced spatial reproduction compared to monophonic presentation. However, mixing a larger number of channels incurs larger time and cost penalties on content producers, and the resulting perception is typically one of a few scattered, discrete channels, rather than a continuum soundfield. Aspects of Dolby Pro Logic decoding are described in U.S. Patent 4,799,260, which patent is incorporated by reference herein in its entirety.
  • AC-3 Digital Audio Compression Standard
  • ATSC Advanced Television Systems Committee
  • Document A/52 December 20, 1995 (available on the World Wide Web of the Internet at www.atsc.org/Standards/A52/a_52.doc). See also the Errata Sheet of July 22, 1999 (available on the World Wide Web of the Internet at www.dolby.com/tech/ATSC_err.pdf.
  • any output channel with a location that does not correspond to the position of one of the input channels will be referred to as an "intermediate” channel.
  • An output channel may also have a location coincident with the position of an input channel.
  • a process for translating M audio input signals, each associated with a direction, to N audio output signals, each associated with a direction, wherein N is larger than M, M is two or more and N is a positive integer equal to three or more comprises providing an M:N variable matrix, applying the M audio input signals to the variable matrix, deriving the N audio output signals from the variable matrix, and controlling the variable matrix in response to the input signals so that a soundfield generated by the output signals has a compact sound image in the direction of the nominal ongoing primary direction of the input signals when the input signals are highly correlated, the image spreading from compact to broad as the correlation decreases and progressively splitting into multiple compact sound images, each in a direction associated with an input signal, as the correlation continues to decrease to highly uncorrelated.
  • the variable matrix may be controlled in response to measures of: (1) the relative levels of the input signals, and (2) the cross- correlation of the input signals.
  • the soundfield may have a compact sound image when the measure of cross-correlation is the maximum value and may have a broadly spread image when the measure of cross-correlation is the reference value
  • the soundfield may have the broadly spread image when the measure of cross-correlation is the reference value and may have a plurality of compact sound images, each in a direction associated with an input signal, when the measure of cross correlation is the minimum value.
  • a process for translating M audio input signals, each associated with a direction, to N audio output signals, each associated with a direction, wherein N is larger than M, and M is three or more comprises providing a plurality of m:n variable matrices, where m is a subset of M and n is a subset of N, applying a respective subset of the M audio input signals to each of the variable matrices, deriving a respective subset of the N audio output signals from each of the variable matrices, controlling each of the variable matrices in response to the subset of input signals applied to it so that a soundfield generated by the respective subset of output signals derived from it has a compact sound image in the direction of the nominal ongoing primary direction of the subset of input signals applied to it when such input signals are highly correlated, the image spreading from compact to broad as the correlation decreases and progressively splitting into multiple compact sound images, each in a direction associated with an input signal applied to it, as the correlation continues to
  • variable matrices may also be controlled in response to information that compensates for the effect of one or more other variable matrices receiving the same input signal.
  • deriving the N audio output signals from the subsets of N audio output channels may also include compensating for multiple variable matrices producing the same output signal.
  • each of the variable matrices may be controlled in response to measures of: (a) the relative levels of the input signals applied to it, and (b) the cross- correlation of the input signals.
  • a process for translating M audio input signals, each associated with a direction, to N audio output signals, each associated with a direction, wherein N is larger than M, and M is three or more comprises providing an M:N variable matrix responsive to scale factors that control matrix coefficients or control the matrix outputs, applying the M audio input signals to the variable matrix, providing a plurality of m:n variable matrix scale factor generators, where m is a subset of M and n is a subset of N, applying a respective subset of the M audio input signals to each of the variable matrix scale factor generators, deriving a set of variable matrix scale factors for respective subsets of the N audio output signals from each of the variable matrix scale factor generators, controlling each of the variable matrix scale factor generators in response to the subset of input signals applied to it so that when the scale factors generated by it are applied to the M:N variable matrix, a soundfield generated by the respective subset of output signals produced has a compact sound image in the nominal ongoing primary direction of the
  • variable matrix scale factor generators may also be controlled in response to information that compensates for the effect of one or more other variable matrix scale factor generators receiving the same input signal.
  • deriving the N audio output signals from the variable matrix may include compensating for multiple variable matrix scale factor generators producing scale factors for the same output signal.
  • each of the variable matrix scale factor generators may be controlled in response to measures of: (a) the relative levels of the input signals applied to it, and (b) the cross- correlation of the input signals.
  • M audio input channels representing a soundfield are translated to N audio output channels representing the same soundfield, wherein each channel is a single audio stream represents audio arriving from a direction, M and N are positive whole integers, and M is at least 2 and N is at least 3, and N is larger than M.
  • Each input and output channel has an associated direction ⁇ e.g., azimuth, elevation and, optionally, distance, to allow for closer or more distant virtual or projected channel).
  • One or more sets of output channels are generated, each set having one or more output channels.
  • Each set is usually associated with two or more spatially adjacent input channels and each output channel in a set is generated by determining a measure of the cross-correlation of the two or more input channels and a measure of the level interrelationships of the two or more input channels.
  • the measure of cross-correlation preferably is a measure of the zero-time- offset cross-correlation, which is the ratio of the common energy level with respect to the geometric mean of the input signal energy levels.
  • the common energy level preferably is the smoothed or averaged common energy level and the input signal energy levels are the smoothed or averaged input signal energy levels.
  • multiple sets of output channels may be associated with more than two input channels and a process may determine the correlation of input channels, with which each set of output channels is associated, according to a hierarchical order such that each set or sets is ranked according to the number of input channels with which its output channel or channels are associated, the greatest number of input channels having the highest ranking, and the processing processes sets in order according to their hierarchical order. Further according to an aspect of the present invention, the processing takes into account the results of processing higher order sets.
  • each of the M audio input channels representing audio arriving from a direction was generated by a passive- matrix nearest-neighbor amplitude-panned encoding of each source direction ⁇ i.e., a source direction is assumed to map primarily to the nearest input channel or channels), without the requirement of additional side chain information (the use of side chain or auxiliary information is optional), making it compatible with existing mixing techniques, consoles, and formats.
  • source signals may be generated by explicitly employing a passive encoding matrix, most conventional recording techniques inherently generate such source signals (thus, constituting an "effective encoding matrix").
  • the playback or decoding aspects of the present invention are also largely compatible with natural recording source signals, such as might be made with five real directional microphones, since, allowing for some possible time delay, sounds arriving from intermediate directions tend to map principally to the nearest microphones (in a horizontal array, specifically to the nearest pair of microphones).
  • a decoder or decoding process may be implemented as a lattice of coupled processing modules or modular functions (hereinafter, “modules” or “decoding modules”), each of which is used to generate one or more output channels (or, alternatively, control signals usable to generate one or more output channels), typically from the two or more of the closest spatially adjacent input channels associated with the decoding module.
  • the output channels typically represent relative proportions of the audio signals in the closest spatially adjacent input channels associated with the particular decoding module.
  • the decoding modules are loosely coupled to each other in the sense that modules share inputs and there is a hierarchy of decoding modules.
  • Modules are ordered in the hierarchy according to the number of input channels they are associated with (the module or modules with the highest number of associated input channels is ranked highest).
  • a supervisor or supervisory function presides over the modules so that common input signals are equitably shared between or among modules and higher-order decoder modules may affect the output of lower-order modules.
  • Each decoder module may, in effect, include a matrix such that it directly generates output signals or each decoder module may generate control signals that are used, along with the control signals generated by other decoder modules, to vary the coefficients of a variable matrix or the scale factors of inputs to or outputs from a fixed matrix in order to generate all of the output signals.
  • Decoder modules emulate the operation of the human ear to attempt to provide perceptually transparent reproduction.
  • Signal translation according to the present invention may be applied either to wideband signals or to each frequency band of a multiband processor, and depending on implementation, may be performed once per sample or once per block of samples.
  • a multiband embodiment may employ either a filter bank, such as a discrete critical-band filterbank or a filterbank having a band structure compatible with an associated decoder, or a transform configuration, such as an FFT (Fast Fourier Transform) or MDCT (Modified Discrete Cosine Transform) linear filterbank.
  • FFT Fast Fourier Transform
  • MDCT Modified Discrete Cosine Transform
  • Another aspect of this invention is that the quantity of speakers receiving the N output channels can be reduced to a practical number by judicious reliance upon virtual imaging, which is the creation of perceived sonic images at positions in space other than where a loudspeaker is located.
  • virtual imaging is the creation of perceived sonic images at positions in space other than where a loudspeaker is located.
  • virtual imaging may include the rendering of phantom projected images that provide the auditory impression of being beyond the walls of a room or inside the walls of a room.
  • Virtual imaging is not considered a viable technique for group presentation with a sparse number of channels, because it requires the listener to be equidistant from the two speakers, or nearly so.
  • the problem with signal distribution is that there is often too little information to recover the original signal amplitude distribution, much less the signals themselves.
  • the basic info ⁇ riation available is the signal levels at each module input and the averaged cross product of the input signals, the common energy level.
  • the zero-time offset cross-correlation is the ratio of the common energy level with respect to the geometric mean of the input signal energy levels.
  • the significance of cross-correlation is that it functions as a measure of the net amplitude of signal components common to all inputs. If there is a single signal panned anywhere between the inputs of the module (an "interior" or “intermediate” signal), all the inputs will have the same waveform, albeit with possibly different amplitudes, and under these conditions, the correlation will be 1.0.
  • Lt/Rt left total and right total
  • a two-input, five-output module might feed only the output channel corresponding to the dominant direction (C in this case) and the output channels corresponding to the input signal residues (L, R) after removing the C energy from the Lt and Rt inputs, giving no signals to the MidL and MidR output channels.
  • C dominant direction
  • L, R input signal residues
  • Such a result is undesirable - turning off a channel unnecessarily is almost always a bad choice, because small perturbations in signal conditions will cause the "off channel to toggle between on and off, causing an annoying chattering sound ("chattering " is a channel rapidly turning on and off), especially when the "off channel is listened to in isolation.
  • the conservative approach from the point of view of individual channel quality is to spread the non-dominant signal components as evenly as possible among the module's output channels, consistent with the signal conditions.
  • An aspect of the present invention is evenly spreading the available signal energy, subject to the signal conditions, according to a three-way split rather than a "dominant" versus "all the rest" two-way split.
  • the three-way split comprises dominant (common) signal components, fill (even-spread) signal components, and input signal components residue.
  • a correlation value is calculated corresponding to all output channels receiving the same signal amplitude.
  • This correlation value may be referred to as the"random_xcor" value.
  • the random- xcor value may calculate as 0.333.
  • the random-xcor value may calculate as 0.483.
  • scaled_xcor (correlation — random xcor) / (1 - random_xcor)
  • scaled_xcor (correlation — random xcor) / (1 - random_xcor)
  • the amount of spread energy should either be progressively reduced if equal distribution to all output channels is maintained or, alternatively, the amount of spread energy should be maintained but the energy distributed to output channels should be reduced in relation to the "off centeredness" of the dominant energy - in other words, a tapering of the energy along the output channels.
  • additional processing complexity may be required to maintain the output power equal to the input power.
  • the current correlation value is less than the random-xcor value, the dominant energy is considered to be zero, the evenly-spread energy is progressively reduced, and the residue signal, whatever is left over, is allowed to accumulate at the inputs.
  • correlation zero, there is no interior signal, just independent input signals that are mapped directly to output channels.
  • each input is fed to one output position only (generally one of the outputs, but it could be a panned position between two of them).
  • channel translation may be considered to involve a lattice of "modules". Because multiple modules may share a given input channel, interactions are possible between modules and may degrade performance unless some compensation is applied. Although it is not generally possible to separate signals at an input according to which module they "go with", estimating the amount of an input signal used by each connected module can improve the resulting correlation and direction estimates, resulting in improved overall performance.
  • modules at a common or lower hierarchy level t.e., modules with a like number of inputs or fewer inputs
  • neighbors modules at a higher hierarchy level (having more inputs) than a given module but sharing one or more common inputs, referred to as “higher-order neighbors”.
  • the L/R input amplitude ratios of each module are offset because the common input has more signal amplitude (A+B) than either outer input, which causes the direction estimate to be biased toward the common input.
  • the correlation value of both modules is now something less than 1.0 because the waveforms at both pairs of inputs are different. Because the correlation value determines the degree of spreading of the non- common signal components and the ratio of the dominant (common signal component) to non-dominant (non-common signal component) energy, uncompensated common-input signal causes the non-common signal distribution of each module to be spread.
  • the analysis of a module's input signals does not allow directly solving for the common input level at each input, only a proportion of the overall common energy, which is the geometric mean of the common input energy levels. Because the common input energy level at each input cannot exceed the total energy level at that input, which is measured and known, the overall common energy is factored into estimated common input levels proportional to the observed input levels, subject to the qualification below.
  • the three-input common signal effects should be subtracted from the inputs before the two-input calculation can be performed properly.
  • the higher-order common signal elements should be subtracted not only from the lower-level module's input levels, but from its observed measure of common energy level as well, before proceeding with the lower level calculation. This is different from the effects of common input levels of modules at the same hierarchy level that do not affect the measure of common energy level of a neighboring module.
  • the higher-order neighbor levels should be accounted for, and employed, separately from the same-order neighbor levels.
  • FIG. 1 is a top plan view showing schematically an idealized decoding arrangement in the manner of a test arrangement employing a sixteen channel horizontal array around the walls of a room, a six channel array disposed in a circle above the horizontal array and a single overhead channel.
  • FIGS. 4A-4C show a functional block diagram of a module according to an aspect of the present invention.
  • FIG. 5 is a schematic view showing a hypothetical arrangement of a three input module fed by a triangle of input channels, three interior output channels, and a dominant direction. The view is useful in understanding the distribution of dominant signal components.
  • FIGS. 6 A and 6B are functional block diagrams showing, respectively, one suitable arrangement for (1) generating the total estimated energy for each input of a module in response to the total energy at each input, and (2) in response to a measure of cross- correlation of the input signals, generating an excess endpoint energy scale factor component for each of the module's endpoints.
  • FIG. 7 is a functional block diagram showing a preferred function of the "sum and/or greater of block 367 of FIG. 4C.
  • FIG. 1 is a top plan view showing schematically an idealized decoding arrangement in the manner of the just-described test arrangement.
  • Five wide range horizontal input channels are shown as squares 1', 3', 5', 9' and 13' on the outer circle.
  • a vertical channel which may be derived from the five wide range inputs via correlation or generated reverberation, or separately supplied (as in FIG. 2), is shown as the broken square 23' in the center.
  • the twenty-three wide range output channels are shown as numbered filled circles 1-23.
  • the outer circle of sixteen output channels is on a horizontal plane, the inner circle of six output channels is forty-five degrees above the horizontal plane.
  • Output channel 23 is directly above one or more listeners.
  • FIG. 1 Although the arrangement of FIG. 1 employs five modules (24-28) (each having two inputs) and five inputs (l', 3', 5', 9' and 13') to derive sixteen horizontal outputs (1-16) representing locations around the four walls of a room, similar results may be obtained with a minimum of three inputs and three modules (each having two inputs, each module sharing one input with another module).
  • each module has output channels in an arc or a line (such as the example of FIGS. 1 and 2)
  • decoding ambiguities encountered in prior art decoders in which correlations less than zero are decoded as indicating rearward directions may be avoided.
  • input and output channels may be characterized by their physical position, or at least their direction, characterizing them with a matrix is useful because it provides a well-defined signal relationship.
  • Each matrix element (row i, column j) is a transfer function relating input channel i to output channel j.
  • Matrix elements are usually signed multiplicative coefficients, but may also include phase or delay terms (in principle, any filter), and may be functions of frequency (in discrete frequency terms, a different matrix at each frequency). This is straightforward in the case of dynamic scale factors applied to the outputs of a fixed matrix, but it also lends itself to variable-matrixing. either by having a separate scale factor for each matrix element, or, for matrix elements more elaborate than simple scalar scale factors, in which matrix elements themselves are variable, e.g., a variable delay.
  • signal commonality may extend to three or more channels.
  • Use and detection of signal commonality may also be used to convey additional signal information.
  • a vertical signal component may be represented by mapping to all five full range channels of a horizontal five-channel array.
  • the "initial mapping" (before processing) derives a passive "master" matrix that relates the input/output channel configurations to the spatial orientation of the channels.
  • the processor or processing portion of the invention may generate time- varying scale factors, one per output channel, which modify either the output signal levels of what would otherwise have been a simple, passive matrix or the matrix coefficients themselves.
  • the scale factors in turn derive from a combination of (a) dominant, (b) even- spread (fill), and (c) residue (endpoint) signal components as described below.
  • a master matrix is useful in configuring an arrangement of modules such as shown in the example of FIG. 1 and described further below in connection with FIG. 2.
  • the master matrix By examining the master matrix, one may deduce, for example, how many decoder modules are needed, how they are connected, how many input and output channels each has and the matrix coefficients relating each modules' inputs and outputs. These coefficients may be taken from the master matrix; only the non-zero values are needed unless an input channel is also an output channel (t.e., an endpoint).
  • Each module preferably has a "local" matrix, which is that portion of the master matrix applicable to the particular module.
  • the module may use the local matrix for the purpose of producing scale factors (or matrix coefficients) for controlling the master matrix, as is described below in connection with FIGS. 2 and 4A-4C, or for the purpose of producing a subset of the output signals, which output signals are assembled by a central process, such as a supervisor as described in connection with FIG. 2.
  • a supervisor in the latter case, compensates for multiple versions of the same output signal produced by modules having a common output signal in a manner analogous to the manner in which supervisor 201 of FIG. 2 determines a final scale factor to replace the preliminary scale factors produced by modules that produce preliminary scale factors for the same output channel.
  • modules may continually obtain the matrix information relevant to itself from a master matrix via a supervisor rather than have a local matrix.
  • less computational overhead is required if the module has its own local matrix.
  • the module has a local matrix, which is the only matrix required (in effect, the local matrix is the master matrix), and that local matrix is used to produce output signals.
  • descriptions of embodiments of the invention having multiple modules are with reference to the alternative in which modules produce scale factors.
  • Any decode module output channel with only one nonzero coefficient in the module's local matrix (that coefficient is 1.0, since the coefficients sum-square to 1.0) is an endpoint channel.
  • Output channels with more than one nonzero coefficient are interior output channels.
  • output channels 01 and O2 are both derived from input channels II and 12 (but with different coefficient values)
  • one of the decoder modules has inputs II and 12 and feeds outputs Ol and O2 such that:
  • the decoder may have two inputs (II and 12), two outputs, and the scale factors relating them are:
  • Either the master matrix or the local matrix may have matrix elements that function to provide more than multiplication.
  • matrix elements may include a filter function, such as a phase or delay term, and/or a filter that is a function of frequency.
  • filtering is a matrix of pure delays that may render phantom projected images.
  • FIG. 2 is a functional block diagram providing an overview of a multiband transform embodiment implementing the example of FIG. 1.
  • a PCM audio input for example, having multiple interleaved audio signal channels is applied to a supervisor or supervisory function 201 (hereinafter “supervisor 201") that includes a de-interleaver that recovers separate streams of each of six audio signal channels (1', 3', 5', 9', 13' and 23') carried by the interleaved input and applies each to a time-domain to frequency-domain transform or transform function (hereinafter "forward transform").
  • the audio channels may be received in separate streams, in which case no de-interleaver is required.
  • signal translation according to the present invention may be applied either to wideband signals or to each frequency band of a multiband processor, which may employ either a filter bank, such as a discrete critical-band filterbank or a filterbank having a band structure compatible with an associated decoder, or a transform configuration, such as an FFT (Fast Fourier Transform) or MDCT (Modified Discrete Cosine Transform) linear filterbank.
  • a filter bank such as a discrete critical-band filterbank or a filterbank having a band structure compatible with an associated decoder
  • a transform configuration such as an FFT (Fast Fourier Transform) or MDCT (Modified Discrete Cosine Transform) linear filterbank.
  • FIGS. 2, 4A-4C and other figures are described in the context of a multiband transfo ⁇ n configuration.
  • an LFE channel may be derived by using a lowpass filter (for example, a fifth-order Butterworth filter with a 120 Hz corner frequency) applied to the sum of the channels, or, to avoid cancellation upon addition of the channels, a phase-corrected sum of the channels may be employed.
  • the LFE channel may be added to one or more of the output channels.
  • modules 24-34 receive appropriate ones of the six inputs l', 3', 5', 9', 13' and 23 ' in the manner shown in FIG. 1.
  • Each module generates a preliminary scale factor ("PSF") output for each of the audio output channels associated with it as shown in FIG. 1.
  • PSF preliminary scale factor
  • module 24 receives inputs 1 ' and 3' and generates preliminary scale factor outputs PSF1, PSF2 and PSF3.
  • each module may generate a preliminary set of audio outputs for each of the audio output channels associated with it.
  • Each module also may communicate with a supervisor 201, as explained further below.
  • Information sent from the supervisor 201 to various modules may include neighbor level information and higher-order neighbor level information, if any.
  • Information sent to the supervisor from each module may include the total estimated energy of interior the outputs attributable to each of the module's inputs.
  • the modules may be considered part of a control signal-generating portion of the overall system of FIG. 2.
  • the system may apply windowing and a filterbank ⁇ e.g., FFT, MDCT, etc.) to each channel (so that multiple modules do not perform redundant transforms that substantially increase the processing overhead) and pass streams of transform values to each module for processing.
  • a filterbank ⁇ e.g., FFT, MDCT, etc.
  • Each module passes back to the supervisor a two-dimensional array of scale factors: one scale factor for all transform bins in each subband of each output channel (when in a multiband transform configuration, otherwise one scale factor per output channel), or, alternatively, a two-dimensional array of output signals: an ensemble of complex transform bins for each subband of each output channel (when in a multiband transform configuration, otherwise one output signal per output channel).
  • the supervisor may smooth the scale factors and apply them to the signal path matrixing (matrix 203, described below) to yield (in a multiband transform configuration) output channel complex spectra.
  • the supervisor may derive the output channels (output channel complex spectra, in a multiband transform configuration), compensating for local matrices that produce the same output signal. It may then perform an inverse transform plus windowing and overlap-add, in the case of MDCT, for each output channel, interleaving the output samples to form a composite multichannel output stream (or, optionally, it may omit interleaving so as to provide multiple output streams), and sends it on to an output file, soundcard, or other final destination.
  • a supervisor may be performed by a supervisor, as described herein, or by multiple supervisors, one of ordinary skill in the art will appreciate that various ones or all of those functions may be performed in the modules themselves rather than by a supervisor common to all or some of the modules. For example, if there is only a single, stand-alone module, there need be no distinction between module functions and supervisor functions. Although, in the case of multiple modules, a common supervisor may reduce the required overall processing power by eliminating or reducing redundant processing tasks, the elimination of a common supervisor or its simplification may allow modules to be easily added to one another, for example, to upgrade to more output channels.
  • Matrix 203 may be considered a part of the signal path of the system of FIG. 2.
  • Matrix 203 also receives as inputs from supervisor 201 a set of final scale factors SF1 through SF23 for each of the 23 output channels of the FIG. 1 example.
  • the final scale factors may be considered as being the output of the control signal portion of the system of FIG. 2.
  • the supervisor 201 preferably passes on, as final scale factors to the matrix, the preliminary scale factors for every "interior” output channel, but the supervisor determines final scale factors for every endpoint output channel in response to information it receives from modules.
  • An "interior" output channel is intermediate to the two or more "endpoint" output channels of each module.
  • the modules produce output signals rather than scale factors, no matrix 203 is required; the supervisor itself produces the output signals.
  • output channels 2, 4, 6-8, 10-12, 14-16, 17, 18, 19, 20, 21 and 22 are interior output channels.
  • Interior output channel 21 is intermediate or bracketed by three input channels (input channels 9', 13' and 23'), whereas the other interior channels are each intermediate (between or bracketed by) two input channels.
  • the supervisor 201 determines the final endpoint scale factors (SF1, SF3, etc.) among the scale factors SF1 through SF23.
  • FIG. 3 is a functional block diagram useful in understanding the manner in which a supervisor, such as supervisor 201 of FIG. 2, may determine an endpoint scale factor.
  • the supervisor does not sum all the outputs of the modules sharing an input to get an endpoint scale factor. Instead, it additively combines, such as in a combiner 301, the total estimated interior energy for a input from each module that shares the input, such as input 9', which is shared by modules 26 and 27 of FIG. 2. This sum represents the total energy level at the input claimed by the interior outputs of all the connected modules.
  • the final scale factor (SF9, in this example) for that output is obtained. Note that the supervisor derives a single final scale factor for each such shared input regardless of how many modules share the input. An arrangement for dete ⁇ nining the total estimated energy of the interior outputs attributable to each of the module's inputs is described below in connection with FIG. 6 A.
  • the levels are energy levels (a second-order quantity), as opposed to amplitudes (a first-order quantity), after the divide operation, a square-root operation is applied in order to obtain the final scale factor (scale factors are associated with first-order quantities).
  • the addition of the interior levels and subtraction from the total input level are all performed in a pure energy sense, because interior outputs of different module interiors are assumed to be independent (uncorrelated). If this assumption is not true in an unusual situation, the calculation may yield more leftover signal at the input than there should be, which may cause a slight spatial distortion in the reproduced soundfield ⁇ e.g., a slight pulling of other nearby interior images toward the input), but in the same situation, the human ear likely reacts similarly.
  • the interior output channel scale factors such as PSF6 through PSF 8 of module 26, are passed on by the supervisor as final scale factors (they are not modified).
  • FIG. 3 only shows the generation of one of the endpoint final scale factors.
  • Other endpoint final scale factors may be derived in a similar manner.
  • the variability may be complicated (all coefficients variable) or simple (coefficients varying in groups, such as being applied to the inputs or the outputs of a fixed matrix).
  • coefficients varying in groups such as being applied to the inputs or the outputs of a fixed matrix.
  • one of the simpler approaches that is, a fixed matrix followed by a variable gain for each output (the gain of each output controlled by scale factors) has been found to produce satisfactory results and is employed in the embodiments described herein.
  • a variable matrix in which each matrix coefficient is variable is usable, it has the disadvantage of having more variables and requiring more processing power.
  • Supervisor 201 also performs an optional time domain smoothing of the final scale factors before they are applied to the variable matrix 203.
  • output channels are never "turned off, the coefficients are arranged to reinforce some signals and cancel others.
  • a fixed-matrix, variable-gain system does turn channels on and off, and is more susceptible to undesirable "chattering" artifacts. This may occur despite the two-stage smoothing described below ⁇ e.g., smoothers 319/325, etc.). For example, when a scale factor is close to zero, because only a small change is needed to go from 'small' to 'none' and back, transitions to and from zero may cause audible chattering.
  • the scale factor smoother time constants may also scale with frequency as well as time, in the manner of frequency smoothers 413, 415 and 417 of FIG. 4A, described below.
  • the variable matrix 203 preferably is a fixed decode matrix with variable scale factors (gains) at the matrix outputs.
  • Each matrix output channel may have (fixed) matrix coefficients that would have been the encode downmix coefficients for that channel had there been an encoder with discrete inputs (instead of mixing source channels directly to the downmixed array, which avoids the need for a discrete encoder.)
  • the coefficients preferably sum-square to 1.0 for each output channel.
  • the matrix coefficients are fixed once it is known where the output channels are (as discussed above with regard to the "master" matrix); whereas the scale factors, controlling the output gain of each channel, are dynamic.
  • Inputs comprising frequency domain transform bins applied to the modules 24-34 of FIG. 2 may be grouped into frequency subbands by each module after initial quantities of energy and common energy are calculated at the bin level, as is explained further below.
  • the frequency-domain output channels 1 -23 produced by matrix 203 each comprise a set of transform bins (subband-sized groups of transform bins are treated by the same scale factor).
  • the sets of fi-equency-domain transform bins are converted to a set of PCM output channels 1-23, respectively, by a frequency- to time-domain transform or transform function 205 (hereinafter "inverse transform"), which may be a function of the supervisor 201, but is shown separately for clarity.
  • inverse transform may be a function of the supervisor 201, but is shown separately for clarity.
  • the supervisor 201 may interleave the resulting PCM channels 1-23 to provide a single interleaved PCM output stream or leave the PCM output channels as separate streams.
  • FIGS. 4A-4C show a functional block diagram of a module according an aspect of to the present invention.
  • the module receives two or more input signal streams from a supervisor, such as the supervisor 201 of FIG. 2.
  • Each input comprises an ensemble of complex- valued frequency-domain transform bins.
  • Each input, 1 through m is applied to a function or device (such as function or device 401 for input 1 and function or device 403 for input m) that calculates the energy of each bin, which is the sum of the squares of the real and imaginary values of each transform bin (only the paths for two inputs, 1 and m, are shown to simplify the drawing).
  • Each of the inputs is also applied to a function or device 405 that calculates the common energy of each bin across the module's input channels.
  • this may be calculated by taking the cross product of the input samples (in the case of two inputs, L and R, for example, the real part of the complex product of the complex L bin value and the complex conjugate of the complex R bin value).
  • Embodiments using real values need only cross-multiply the real value for each input.
  • the special cross-multiplication technique described below may be employed, namely, if all the signs are the same, the product is given a positive sign, else it is given a negative sign and scaled by the ratio of the number of possible positive results (always two: they are either all positive or all negative) to the number of possible negative results. Pairmse Calculation of Common Energy
  • a 2 0.5X 2 + Y 2 i.e., Because X and Y are uncorrelated, the total energy in input channel A is the sum of the energies of signals X and Y.
  • the averaged cross-product of the signals is equal to the energy of the common signal component in each channel. If the common signal is not shared equally, i.e., it is panned toward one of the inputs, the averaged cross-product will be the geometric mean between the energy of the common components in A and B, from which individual channel common energy estimates can be derived by normalizing by the square root of the ratio of the channel amplitudes. Actual time averages are computed subsequent smoothing stages, as described below.
  • This problem may be resolved by employing a variant of the averaged product technique.
  • the sign of the each product is discarded by taking the absolute value of the product.
  • the signs of each term of the product are examined. If they are all the same, the absolute value of the product is applied to the averager. If any of the signs are different from the others, the negative of the absolute value of the product is averaged. Since the number of possible same-sign combinations may not be the same as the number of possible different-sign combinations, a weighting factor comprised of the ratio of the number of same to different sign combinations is applied to the negated absolute value products to compensate.
  • This compensation causes the integrated or summed product to grow in a positive direction if and only if there is a signal component common to all inputs of a decoding module.
  • the individual input energies of a module can be calculated as the average of the square of the corresponding input signal, and need not be first raised to the kth power and then reduced to a second order quantity.
  • the transform bin outputs of each of the blocks may be grouped into subbands by a respective function or device 407, 409 and 411.
  • the subbands may approximate the human ear's critical bands, for example.
  • the remainder of the module embodiment of FIGS. 4A-4C operates separately and independently on each subband. In order to simplify the drawing, only the operation on one subband is shown.
  • Each subband from blocks 407, 409 and 411 is applied to a frequency smoother or frequency smoothing function 413, 415, and 417 (hereinafter “frequency smoother”), respectively.
  • frequency smoother or frequency smoothing function 413, 415, and 417
  • the purpose of the frequency smoothers is explained below.
  • Each frequency- smoothed subband from a frequency smoother is applied to optional "fast” smoothers or smoothing functions 419, 421 and 423 (hereinafter “fast smoothers”), respectively, that provide time-domain smoothing.
  • the fast smoothers may be omitted when the time constant of the fast smoothers is close to the block length time of the forward transform that generated the input bins (for example, a forward transform in supervisor 201 of FIG. 1).
  • the fast smoothers are “fast” relative to the “slow” variable time constant smoothers or smoother functions 425, 427 and 429 (hereinafter “slow smoothers") that receive the respective outputs of the fast smoothers. Examples of fast and slow smoother time constant values are given below.
  • the time constants of the slow smoothers preferably are in synchronism with each other within a module. This may be accomplished, for example, by applying the same control information to each slow smoother and by configuring each slow smoother to respond in the same way to applied control information. The derivation of the information for controlling the slow smoothers is described below.
  • each pair of smoothers are in series, in the manner of the pairs 419/425, 421/427 and 423/429 as shown in FIGS. 4A and 4B, in which a fast smoother feeds a slow smoother.
  • a series arrangement has the advantage that the second stage is resistant to short rapid signal spikes at the input of the pair.
  • similar results may be obtained by configuring the pairs of smoothers in parallel. For example, in a parallel arrangement the resistance of the second stage in a series arrangement to short rapid signal spikes may be handled in the logic of a time constant controller.
  • Each stage of the two-stage smoothers may be implemented by a single-pole lowpass filter (a "leaky integrator") such as an RC lowpass filter (in an analog embodiment) or, equivalently, a first-order lowpass filter (in a digital embodiment).
  • the first-order filters may each be realized as a "biquad" filter, a general second- order IIR filter, in which some of the coefficients are set to zero so that the filter functions as a first-order filter.
  • the two smoothers may be combined into a single second- order biquad stage, although it is simpler to calculate coefficient values for the second (variable) stage if it is separate from the first (fixed) stage.
  • the two-stage smoothers thus provide a time average for each subband of each input channel's energy (that of the 1 st channel is provided by slow smoother 425 and that of the m" 1 channel by slow smoother 427) and the average for each subband of the input channels' common energy (provided by slow smoother 429).
  • the average energy outputs of the slow smoothers (425, 427, 429) are applied to combiners 431, 433 and 435, respectively, in which (1) the neighbor energy levels (if any) (from supervisor 201 of FIG. 2, for example) are subtracted from the smoothed energy level of each of the input channels, and (2) the higher-order neighbor energy levels (if any) (from supervisor 201 of FIG.
  • each module receiving input 3 ' (FIGS. 1 and 2) has two neighboring modules and receives neighbor energy level information that compensates for the effect of those two neighboring modules. However, neither of those modules is a "higher-order" module (t.e., all modules sharing input channel 3' are two-input modules).
  • module 28 (FIGS. 1 and 2) is an example of a module that has a higher-order module sharing one of its inputs. Thus, for example, in module 28, the average energy output from a slow smoother for input 13' receives higher-order neighbor level compensation.
  • the resulting "neighbor-compensated" energy levels for each subband of each of the module's inputs are applied to a function or device 437 that calculates a nominal ongoing primary direction of those energy levels.
  • the direction indication may be calculated as the vector sum of the energy- weighted inputs. For a two input module, this simplifies to being the L/R ratio of the smoothed and neighbor-compensated input signal energy levels.
  • a planar surround array in which the positions of the channels are given as 2-ples representing x, y coordinates for the case of two inputs.
  • the listener in the center is assumed to be at, say, (0, 0).
  • the left front channel in normalized spatial coordinates, is at (1, 1).
  • the spatial direction may be expressed in matrix coordinates, rather than physical coordinates.
  • the input amplitudes, normalized to sum-square to one are the effective matrix coordinates of the direction.
  • the left and right levels are 4 and 3, which normalize to 0.8 and 0.6.
  • Block 337 produces the same number of outputs, indicating a spatial direction, as there are inputs to the module (two in this example).
  • the neighbor-compensated smoothed energy levels for each subband of each of the module's inputs applied to the direction-determining function or device 337 are also applied to a function or device 339 that calculates the neighbor-compensated cross-correlation ("neighbor-compensated_xcor").
  • Block 339 also receives as an input the averaged common energy of the module's inputs for each subband from slow variable smoother 329, which has been compensated in combiner 335 by higher-order neighbor energy levels, if any.
  • the neighbor-compensated cross-correlation is calculated in block 339 as the higher-order compensated smoothed common energy divided by the M th root, where M is the number of inputs, of the product of the neighbor-compensated smoothed energy levels for each of the module's input channels to derive a true mathematical correlation value in the range 1.0 to - 1.0.
  • values from 0 to -1.0 are taken to be zero.
  • Neighbor-compensated_xcor provides an estimate of the cross-correlation that exists in the absence of other modules.
  • the neighbor-compensated xcor from block 339 is then applied to a weighting device or function 341 that weights the neighbor-compensated_xcor with the neighbor-compensated direction information to produce a direction-weighted neighbor-compensated cross- correlation ("direction- weighted_xcor").
  • the weighting increases as the nominal ongoing primary direction departs from a centered condition. In other words, unequal input amplitudes (and, hence, energies) cause a proportional increase in direction-weighted xcor.
  • Direction- weighted xcor provides an estimate of image compactness.
  • the weighting increases as the direction departs from center toward either left or right ⁇ i.e., the weighting is the same in any direction for the same degree of departure from the center).
  • the neighbor-compensated_xcor value is weighted by an L/R or R/L ratio, such that uneven signal distribution urges the direction- weighted xcor toward 1.0.
  • calculation of the direction- weighted xcor from the neighbor- weighted_xcor requires, for example, replacing the ratio L/R or R/L in the above by an "evenness" measure that varies between 1.0 and 0.
  • evenness measure that varies between 1.0 and 0.
  • the direction- weighted_xcor is weighted further by its application to a function or device 443 that applies a "random xcor" weighting to produce an "effective_xcor".
  • Effective_xcor provides an estimate of the input signals' distribution shape. Random_xcor is the average cross product of the input magnitudes divided by the square root of the average input energies. The value of random_xcor may be calculated by assuming that the output channels were originally module input channels, and calculating the value of xcor that results from all those channels having independent but equal-level signals, being passively downmixed.
  • random xcor calculates to 0.333,and for the case of a five-output module (three interior outputs) with two inputs, random xcor calculates to 0.483.
  • the random_xcor value need only be calculated once for each module. Although such random_xcor values have been found to provide satisfactory results, the values are not critical and other values may be employed at the discretion of the system designer. A change in the value of random_xcor affects the dividing line between the two regimes of operation of the signal distribution system, as described below. The precise location of that dividing line is not critical.
  • Information for controlling the slow smoothers 325, 327 and 329 is derived from the non-neighbor-compensated slow and fast smoothed input channels' energies and from the slow and fast smoothed input channels' common energy.
  • a function or device 345 calculates a fast non-neighbor compensated cross-correlation in response to the fast smoothed input channels' energies and the fast smoothed input channels' common energy.
  • a function or device 347 calculates a fast non-neighbor compensated direction (ratio or vector, as discussed above in connection with the description of block 337) in response to the fast smoothed input cham el energies.
  • a function or device 349 calculates a slow non-neighbor compensated cross-correlation in response to the slow smoothed input channels' energies and the slow smoothed input channels' common energy.
  • a function or device 351 calculates a slow non-neighbor compensated direction (ratio or vector, as discussed above) in response to the slow smoothed input channel energies.
  • the fast non-neighbor compensated cross- correlation, fast non-neighbor compensated direction, slow non-neighbor compensated cross- correlation and slow non-neighbor compensated cross-correlation, along with direction- weighted_xcor from block 341, are applied to a device or function 353 that provides the information for controlling the variable slow smoothers 325, 327 and 329 to adjust their time constants (hereinafter "adjust time constants").
  • the same control information is applied to each variable slow smoother.
  • the direction-weighted xcor preferably is used without reference to any fast value, such that if the absolute value of the direction- weighted_xcor is greater than a threshold, it may cause adjust time constants 353 to select a faster time constant. Rules for operation of "adjust time constants" 353 are set forth below.
  • a module's time constants may speed up and rapidly assume a new control state as desired.
  • the basic decoding process within each module depends on a measure of energy ratios of the input signals and a measure of cross-correlation of the input signals, (in particular, the direction-weighted correlation (direction-weighted xcor), described above; the output of block 341 in FIG. 4B), which, together, control signal distribution among the outputs of a module.
  • Derivation of such basic quantities requires smoothing, which, in the time domain, requires computing a time- weighted average of the instantaneous values of those quantities.
  • the range of required time constants is quite large: very short (1 msec, for example) for fast transient changes in signal conditions, to very long (150 msec, for example) for low values of correlation, where the instantaneous variation is likely to be much greater than the true averaged value.
  • a common method of implementing variable time constant behavior is, in analog terms, the use of a "speed-up" diode.
  • the instantaneous level exceeds the averaged level by a threshold amount, the diode conducts, resulting in a shorter effective time constant.
  • a drawback of this technique is that a momentary peak in an otherwise steady-state input may cause a large change in the smoothed level, which then decays very slowly, providing unnatural emphasis of isolated peaks that would otherwise have little audible consequence.
  • the correlation calculation described in connection with the embodiment of FIGS. 4A-4C makes the use of speedup diodes (or their DSP equivalent) problematical. For example, all smoothers within a particular module preferably have synchronized time constants, so that their smoothed levels are comparable.
  • embodiments of aspects of the present invention preferably use two-stage smoothing without a diode-equivalent speedup. Estimates of correlation and direction may be derived at least from both the first and second stages of the smoothers to set the time constant of the second stage. For each pair of smoothers ⁇ e.g., 319/325), the first stage, the fixed fast stage, time constant may be set to a fixed value, such as 1 msec.
  • the second stage, variable slow stage, time constants may be, for example, selectable among 10 msec (fast), 30 msec (medium), and 150 msec (slow). Although such time constants have been found to provide satisfactory results, their values are not critical and other values may be employed at the discretion of the system designer. In addition, the second stage time constant values may be continuously variable rather than discrete. Selection of the time constants may be based not only on the signal conditions described above, but also on a hysteresis mechanism using a "fast flag", which is used to ensure that once a genuine fast transition is encountered, the system remains in fast mode, avoiding the use of the medium time constant, until the signal conditions re- enable the slow time constant. This may help assure rapid adaptation to new signal conditions.
  • direction- weighted_xcor is less than a first reference value (0.5, for example) and the absolute difference between fast non-neighbor-compensated_xcor and slow non-neighbor-compensated xcor is less than the same first reference value, and the absolute difference between the fast and slow direction ratios (each of which has a range +1 to -1) is less than the same first reference value, then the slow second stage time constant is used, and the fast flag is set to True, enabling subsequent selection of the medium time constant.
  • a first reference value 0.5, for example
  • the fast flag is True, the absolute difference between the fast and slow non-neighbor-compensated_xcor is greater than the first reference value and less than a second reference value (0.75, for example), the absolute difference between the fast and slow temporary L/R ratios is greater than the first reference value and less than the second reference value, and the absolute value of direction- weighted xcor is greater than the first reference value and less than the second reference value, then the medium second stage time constant is selected. Else, the fast second stage time constant is used, and the fast flag is set to False, disabling subsequent use of the medium time constant until the slow time constant is again selected.
  • the slow time constant is chosen when all three conditions are less than a first reference value
  • the medium time constant is chosen when all conditions are between a first reference value and a second reference value and the prior condition was the slow time constant
  • the fast time constant is chosen when any of the conditions are greater than the second reference value
  • a time delay may be built into the system to allow control signals to adapt before applying them to a signal path.
  • this delay may be realized as a discrete delay (5 msec, for example) in the signal path.
  • the delay is a natural consequence of block processing, and if analysis of a block is performed before signal path matrixing of that block, no explicit delay may be required.
  • Multiband embodiments of aspects of the invention may use the same time constants and rules as wideband versions, except that the sampling rate of the smoothers may be set to the signal sampling rate divided by the block size, ⁇ e.g., the block rate), so that the coefficients used in the smoothers are adjusted appropriately.
  • the time constants preferably are scaled inversely to frequency.
  • this is not possible inasmuch as there are no separate smoothers at different frequencies, so, as partial compensation, a gentle bandpass/preemphasis filter may be applied to the input signal to the control path, to emphasize middle and upper-middle frequencies.
  • This filter may have, for example, a two-pole highpass characteristic with a corner frequency at 200 Hz, plus a 2-pole lowpass characteristic with a corner frequency at 8000 Hz, plus a preemphasis network applying 6 dB of boost from 400 Hz to 800 Hz and another 6 dB of boost from 1600 Hz to 3200 Hz.
  • multiband versions of aspects of the invention preferably also employ frequency-domain smoothing, as described above in connection with FIG. 4A (frequency smoothers 413, 415 and 417).
  • the non-neighbor- compensated energy levels may be averaged with a sliding frequency window, adjusted to approximate a 1/3-octave (critical band) bandwidth, before being applied to the subsequent time-domain processing described above.
  • the width of this window increases with increasing frequency, and is usually only one transfo ⁇ n coefficient wide at low frequencies (below about 400 Hz). Therefore, the total smoothing applied to the multiband processing relies more on time domain smoothing at low frequencies, and frequency-domain smoothing at higher frequencies, where rapid time response is likely to be more necessary at times.
  • preliminary scale factors (shown as "PSFs" in FIG. 2), which ultimately affect the dominant/fill/endpoint signal distribution, may be produced by a combination of devices or functions 455, 457 and 459 that calculate "dominant" scale factor components, "fill” scale factor components and “excess endpoint energy” scale factor components, respectively, respective normalizers or no ⁇ nalizer functions 361, 363 and 365, and a device or function 367 that takes either the greatest of the dominant and fill scale factor components and/or the additive combination of the fill and excess endpoint energy scale factor components.
  • the preliminary scale factors may be sent to a supervisor, such as supervisor 201 of FIG. 2 if the module is one of a plurality of modules.
  • Preliminary scale factors may each have a range from zero to one.
  • the nominal ongoing primary direction may be assumed to be (0.8, 0.6), between the Middle Left ML channel (.92, .38) and the center C channel (.71, .71). This may be accomplished by finding two consecutive channels where the L coefficient is larger than the nominal ongoing primary direction L coordinate, and the channel to its right has an L coefficient less than the dominant L coordinate.
  • the dominant scale factor components are apportioned to the two closest channels in a constant power sense.
  • a system of two equations and two unknowns is solved, the unknowns being the dominant-component scale factor component of the channel to the left of the dominant direction (SFL), and the corresponding scale factor component to the right of the nominal ongoing primary direction (SFR) (these equations are solved for SFL and SFR).
  • first dominant coord SFL * left-channel matrix value 1 + SFR * right-channel matrix value 1
  • second_dominant_coord SFL * left-channel matrix value 2 + SFR * right-channel matrix value 2
  • left- and right- channel means the channels bracketing the nominal ongoing primary direction, not the L and R input channels to the module.
  • the solution is the anti-dominant level calculations of each channel, normalized to sum-square to 1.0, and used as dominant distribution scale factor components (SFL, SFR), each for the other channel.
  • SFL, SFR dominant distribution scale factor components
  • the anti-dominant value of an output channel with coefficients A, B for a signal with coordinates C, D is the absolute value of AD - BC.
  • Antidom (first chan) abs (AB - BA)
  • Antidom (second chan) abs (CB - DA)
  • the first antidom value is zero.
  • the second antidom value is 1.0.
  • the first channel receives a dominant scale factor component of 1.0 (times square root of effective_xcor) and the second channel receives 0.0, as desired.
  • block 337 of FIG. 4B calculates the nominal ongoing primary direction coordinates by taking the input amplitudes, after neighbor compensation, and normalizing them to sum-square to one.
  • the three nearest channels to the nominal ongoing primary direction are those tliree interior channels at the bottom, but they do not sum to the dominant coordinates using scale factors between 0 and 1, so instead one chooses two from the bottom and the top endpoint channel to distribute the dominant signal, and solve the three equations for the three weighting factors in order to complete the dominant calculation and proceed to the fill and endpoint calculations.
  • device or function 357 (“calculate fill scale factor components") receives random_ xcor, direction-weighted_xcor from block 341, "EQUIAMPL” ("EQUIAMPL” is defined and explained below), and information regarding the local matrix coefficients from the local matrix (in case the same fill scale factor component is not applied to all outputs, as is explained below in connection with FIG. 14B).
  • the output of block 457 is a scale factor component for each module output (per subband).
  • effective_xcor is zero when the direction- weighted xcor is less than or equal to random_xcor.
  • fill scale factor component sqrt (direction- weighted_xcor / random_xcor) *
  • decoder module Associated with every decoder module is not only a value of random xcor but also a value of "EQUIAMPL", which is a scale factor value that all the scale factors should have if the signals are distributed equally such that power is preserved, namely:
  • Endpoint Scale Factor Components receives the respective 1 st through the m* input's smoothed non-neighbor-compensated energy (from blocks 325 and 327) and, optionally, information regarding the local matrix coefficients from the local matrix (in case either or both of the endpoint outputs of the module do not coincide with an input and the module applies excess endpoint energy to the two outputs having directions closest to the input's direction, as discussed further below).
  • the output of block 359 is a scale factor component for each endpoint output if the directions coincide with input directions, otherwise two scale factor components, one for each of the outputs nearest the end, as is explained below.
  • the excess endpoint energy scale factor components produced by block 359 are not the only "endpoint" scale factor components.
  • There are three other sources of endpoint scale factor components two in the case of a single, stand-alone module: First, within a particular module's preliminary scale factor calculations, the endpoints are possible candidates for dominant signal scale factor components by block 355 (and normalizer 361).
  • the endpoints are treated as possible fill candidates, along with all the interior channels. Any non-zero fill scale factor component may be applied to all outputs, even the endpoints and the chosen dominant outputs.
  • a supervisor performs a final, fourth, assignment of the "endpoint" channels, as described above in connection with FIGS. 2 and 3.
  • the total energy at all interior outputs is reflected back to the module's inputs, based on neighbor-compensated_xcor, to estimate how much of the energy of interior outputs is contributed by each input (“interior energy at input 'n'"), and that energy is used to compute the excess endpoint energy scale factor component at each module output that is coincident with an input ⁇ i.e., an endpoint).
  • FIGS. 6A and 6B are functional block diagrams showing, respectively, in a module, such as any one of modules 24-34 of FIG. 2, one suitable arrangement for (1) generating the total estimated interior energy for each input of a module, 1 through m, in response to the total energy at each input, 1 through m, and (2) in response to the neighbor- compensated_xcor (see FIG.
  • FIG. 6A The total estimated interior energy for each input of a module, (FIG. 6A) is required by the supervisor, in the case of a multiple module arrangement, and, in any case, by the module itself in order to generate the excess endpoint energy scale factor components.
  • the arrangement of FIG. 6A uses the scale factor components derived in blocks 455 and 457 of FIG. 4C, along with other information, the arrangement of FIG. 6A calculates the total estimated energy at each interior output (but not its endpoint outputs).
  • the calculated interior output energy levels uses the calculated interior output energy levels to multiples each output level by the matrix coefficient relating that output to each input ["m" inputs, "m” multipliers], which provides the energy contribution of that input to that output. For each input, it sums all the energy contributions of all the interior output channels to obtain the total interior energy contribution of that input. The total interior energy contribution of each input is reported to the supervisor and is used by the module to calculate the excess endpoint energy scale factor component for each endpoint output. Referring to FIG. 6A in detail, the smoothed total energy level for each module input
  • FIG. 6A shows two inputs, "1" and “m” and two interior outputs "X" and "Z”.
  • the smoothed total energy level for each module input is multiplied by a matrix coefficient (of the module's local matrix) that relates the particular input to one of the module's interior outputs (note that the matrix coefficients are their own inverses because matrix coefficients sum square to one). This is done for every combination of input and interior output.
  • the smoothed total energy level at input 1 (which may be obtained, for example at the output of the slow smoother 425 of FIG.
  • multiplier 601 that multiplies that energy level by a matrix coefficient relating interior output X to input 1, providing a scaled output energy level component X ⁇ at output X.
  • multipliers 603, 605 and 607 provide scaled energy level components X m , Z ⁇ and Z m .
  • each interior output ⁇ e.g., X] and X m ; ⁇ ⁇ and Z m ) are summed in combiners 611 and 613 in an amplitude / power manner in accordance with neighbor-compensated_xcor. If the inputs to a combiner are in phase, indicated by a neighbor- weighted cross correlation of 1.0, their linear amplitudes add. If they are uncorrelated, indicated by a neighbor-weighted cross correlation of zero, their energy levels add. If the cross-correlation is between one and zero, the sum is partly an amplitude sum and partly a power sum.
  • both the amplitude sum and the power sum are calculated and weighted by neighbor-compensated_xcor and (1 - neighbor-weighted_xcor), respectively.
  • the summation products ⁇ X ⁇ + X m ; Z ⁇ + Z m ) are multiplied by the scale factor components for each of the outputs, X and Z, in multipliers 613 and 615 to produce the total energy level at each interior output, which may be identified as X' and Z'.
  • the scale factor component for each of the interior outputs is obtained from block 467 (FIG. 4C),. Note that the "excess endpoint energy scale factor components" from block 459 (FIG. 4C) do not affect interior outputs and are not involved in the calculations performed by the FIG. 6A arrangement.
  • the total energy level at each interior output, X' and Z' is reflected back to respective ones of the module's inputs by multiplying each by a matrix coefficient (of the module's local matrix) that relates the particular output to each of the module's inputs. This is done for every combination of interior output and input.
  • the total energy level X' at interior output X is applied to a multiplier 617 that multiplies the energy level by a matrix coefficient relating interior output X to input 1 (which is the same as its inverse, as noted above), providing a scaled energy level component X ⁇ ' at input 1.
  • a second order weight is required. This is equivalent to taking the square root of the energy to obtain an amplitude, multiplying that amplitude by the matrix coefficient and squaring the result to get back to an energy value.
  • multipliers 619, 621 and 623 provide scaled energy levels X m ', Z and Z m '.
  • the energy components relating to each output ⁇ e.g., ⁇ and Zi', X m ' and Z m ') are summed in combiners 625 and 627 in an amplitude / power manner, as described above in connection with combiners 611 and 613, in accordance with neighbor-compensated_xcor.
  • the outputs of combiners 625 and 627 represent the total estimated interior energy for inputs 1 and m, respectively. In the case of a multiple module lattice, this information is sent to the supervisor, such as supervisor 201 of FIG. 2, so that the supervisor may calculate neighbor levels.
  • the supervisor solicits all the total interior energy contributions of each input from all the modules connected to that input, then informs each module, for each of its inputs, what the sum of all the other total interior energy contributions was from all the other modules connected to that input. The result is the neighbor level for that input of that module.
  • the generation of neighbor level information is described further below.
  • the total estimated interior energy contributed by each of inputs 1 and m are also required by the module in order to calculate the excess endpoint energy scale factor component for each endpoint output.
  • FIG. 6B shows how such scale factor component information may be calculated. For simplicity in presentation, only the calculation of scale factor component information for one endpoint is show, it being understood that a similar calculation is performed for each endpoint output.
  • the total estimated interior energy contributed by an input, such as input 1 is subtracted in a combiner or combining function 629 from the smoothed total input energy for the same input, input 1 in this example (the same smoothed total energy level at input 1, obtained, for example at the output of the slow smoother 425 of FIG. 4B, which is applied to a multiplier 601).
  • the result of the subtraction is divided in divider or dividing function 631 by the smoothed total energy level for the same input 1.
  • the square root of the result of the division is taken in a square rooter or square rooting function 633. It should be noted that the operation of the divider or dividing function 631 (and other dividers described herein) should include a test for a zero denominator. In that case, the quotient may be set to zero.
  • the endpoint preliminary scale factor components are thus determined by virtue of having determined the dominant, fill and excess endpoint energy scale factors.
  • all output channels including endpoints have assigned scale factors, and one may proceed to use them to perform signal path matrixing.
  • each one has assigned an endpoint scale factor to each input feeding it, so each input having more than one module connected to it has multiple scale factor assigmnents, one from each connected module.
  • the supervisor (such as supervisor 201 of the FIG. 2 example) performs a final, fourth, assignment of the "endpoint" channels, as described above in connection with FIGS. 2 and 3. that the supervisor determines final endpoint scale factors that override all the scale factor assignments made by individual modules as endpoint scale factors.
  • an output matrix such as the variable matrix 203 of FIG. 2, may map an output channel to one or more appropriate output channels if there is no actual output channel that directly corresponds to an input channel.
  • each of the "calculate scale factor component" devices or functions 455, 457 and 459 are applied to respective normalizing devices or functions 461, 463 and 465.
  • Such normalizers are desirable because the scale factor components calculated by blocks 455, 457 and 459 are based on neighbor-compensated levels, whereas the ultimate signal path matrixing (in the master matrix, in the case of multiple modules, or in the local matrix, in the case of a stand-alone module) involves non- neighbor-compensated levels (the input signals applied to the matrix are not neighbor- compensated).
  • scale factor components are reduced in value by a no ⁇ nalizer.
  • Each normalizer receives the neighbor-compensated smoothed input energy for each of the module's inputs (as from combiners 331 and 333), the non-neighbor-compensated smoothed input energy for each of the module's inputs (as from blocks 325 and 327), local matrix coefficient information from the local matrix, and the respective outputs of blocks 355, 357 and 359.
  • Each normalizer calculates a desired output for each output channel and an actual output level for each output channel, assuming a scale factor of 1. It then divides the calculated desired output for each output channel by the calculated actual output level for each output channel and takes the square root of the quotient to provide a potential preliminary scale factor for application to "sum and/or greater of 367.
  • the "sum or and/or greatest of 367 preferably sums the corresponding fill and endpoint scale factor components for each output channel per subband, and, selects the greater of the dominant and fill scale factor components for each output channel per subband.
  • the function of the "sum and/or greater of block 367 in its preferred form may be characterized as shown in FIG. 7. Namely, dominant scale factor components and fill scale factor components are applied to a device or function 701 that selects the greater of the scale factor components for each output ("greater of 701) and applies them to an additive combiner or combining function 703 that sums the scale factor components from greater of 701 with the excess endpoint energy scale factors for each output.
  • acceptable results may be obtained when the "sum and/or greatest of 467: (1) sums in both Region 1 and Region 2, (2) takes the greater of in both Region 1 and Region 2, or (3) selects the greatest of in Region 1 and sums in Region 2.
  • FIG. 8 is an idealized representation of the manner in which an aspect of the present invention generates scale factor components in response to a measure of cross-correlation.
  • the figure is particularly useful for reference to FIGS. 9 A and 9B through FIGS. 16A and 16B examples.
  • the generation of scale factor components may be considered as having two regions or regimes of operation: a first region, Region 1, bounded by "all dominant” and “evenly filled” in which the available scale factor components are a mixture of dominant and fill scale factor components, and a second region, Region 2, bounded by "evenly filled” and "all endpoints” in which the available scale factor components are a mixture of fill and excess endpoint energy scale factor components.
  • the "all dominant” boundary condition occurs when the direction-weighted_xcor is one.
  • Region 1 dominant plus fill
  • Region 2 fill plus endpoint
  • the "evenly filled” boundary point may be considered to be in either Region 1 or Region 2. As mentioned below, the precise boundary point is not critical.
  • the fill scale factor components increase in value, reaching a maximum as the dominant scale factor component(s) reach a zero value, at which point as the fill scale factor components decline in value, the excess endpoint energy scale factor components increase in value.
  • the result when applied to an appropriate matrix that receives the module's input signals, is an output signal distribution that provides a compact sound image when the input signals are highly correlated, spreading (widening) from compact to broad as the correlation decreases, and progressively splitting or bowing outward into multiple sound images, each at an endpoint, from broad, as the correlation continues to decrease to highly uncorrelated.
  • FIGS. 9 A and 9B through FIGS. 16A and 16B illustrate the output scale factors of a module for various examples of input signal conditions.
  • a single, stand-alone module is assumed so that the scale factors it produces for a variable matrix are the final scale factors.
  • the module and an associated variable matrix have two input channels (such as left L and right R) that coincide with two endpoint output channels (that may also be designated L and R).
  • there are three interior output channels such as left middle Lm, center C, and right middle Rm).
  • the preliminary scale factors produced by block 467 are the same as the normalized dominant scale factor components produced by block 361.
  • the input energy levels are equal, but direction- weighted_xcor is less than 1.0 and more than random xcor. Consequently, the scale factor components are that of Region 1 -mixed dominant and fill scale factor components.
  • the greater of the normalized dominant scale factor component (from block 361) and the normalized fill scale factor component (from block 363) is applied to each output channel (by block 367) so that the dominant scale factor is located at the same central output channel C as in FIG. 10B, but is smaller, and the fill scale factors appear at each of the other output channels, L, LM, RM and R (including the endpoints L and R).
  • the scale factors, FIG. 1 IB are that of the boundary condition between Regions 1 and 2 - the evenly filled condition in which there are no dominant or endpoint scale factors, just fill scale factors having the same value at each output (hence, "evenly filled"), as indicated by the identical a ⁇ ows at each output.
  • the fill scale factor levels reach their highest value in this example.
  • fill scale factors may be applied unevenly, such as in a tapered manner depending on input signal conditions.
  • the input energy levels remain equal, but direction-weighted_xcor is zero. Consequently, the scale factors, shown in FIG. 13B, are that of the all endpoints boundary condition. There are no interior output scale factors, only endpoint scale factors.
  • the direction- weighted_xcor (such as produced by block 441 of FIG. 4B) is the same as the neighbor-compensated_xcor (such as produced by block 439 of FIG. 4B).
  • the input energy levels are not equal (L is greater than R).
  • the resulting scale factors are not fill scale factors applied evenly to all channels as in the example of FIGS. 11 A and 1 IB. Instead, the unequal input energy levels cause a proportional increase in the direction-weighted_xcor (proportional to the degree to which the nominal ongoing primary direction departs from its central position) such that it becomes greater than the neighbor-compensated_xcor, thereby causing the scale factors to be weighted more towards all dominant (as illustrated in FIG. 8).
  • This is a desired result because strongly L- or R- weighted signals should not have broad width; they should have a compact width near the L or R channel endpoint.
  • the resulting output shown in FIG. 14B, is a non-zero dominant scale factor located closer to the L output than the R output (the neighbor- compensated direction information, in this case, happens to locate the dominant component precisely at the left middle LM position), reduced fill scale factor amplitudes, and no endpoint scale factors (the direction weighting pushes the operation into Region 1 of FIG. 8 (mixed dominant and fill)).
  • the outputs may be expressed as:
  • output_channel_sub_i sf (i) * (Lt_Coeff (i) * Lt + Rt_Coeff (i) * Rt)
  • output amplitude (output_channel_sub_i) sf (i) * (Lt_Coeff (i) * Lt + Rt_Coeff (i) * Rt)
  • the fill scale factors may be equally distributed to the output channels as shown in the examples of FIGS. 10B, 11B, 12B and 14B.
  • the fill scale factor components may be varied with position in some manner as a function of the dominant (correlated) and/or endpoint (uncorrelated) input signal components (or, equivalently, as a function of the direction- weighted_xcor value.)
  • the fill scale factor component amplitudes may curve convexly, such that output channels near the nominal ongoing primary direction receive more signal level than channels farther away.
  • the fill scale factor component amplitudes may flatten to an even distribution, and for direction- weighted xcor ⁇ random_xcor, the amplitudes may curve concavely, favoring channels near the endpoint directions. Examples of such curved fill scale factor amplitudes are set forth in FIG. 15B and
  • FIG. 16B The FIG. 15B output results from an input (FIG. 15 A) that is the same as in FIG. 10A, described above.
  • Each module in a multiple-module arrangement requires two mechanisms in order to support communication between it and a supervisor, such as supervisor 201 of FIG. 2:
  • the information required by the supervisor is the total estimated interior energy attributable to each of the module's inputs as generated, for example, by the arrangement of FIG. 6A.
  • the neighbor levels are subtracted in respective combiners 431 and 433 from the smoothed energy levels of each input, and the higher-order neighbor levels (if any) are subtracted in respective combiners 431, 433 and 435 from the smoothed energy levels of each input and the common energy across the channels.
  • HO neighbor levels are neighbor levels of one or more higher-order modules that share the inputs of a lower-level module.
  • the above calculation of neighbor levels relates only to modules at a particular input that have the same hierarchy: all the three- input modules (if any), then all the two-input modules, etc.
  • An HO-neighbor level of a module is the sum of all the neighbor levels of all the higher order modules at that input, ⁇ i.e., the HO neighbor level at an input of a two-input module is the sum of all the third, fourth, and higher order modules, if any, sharing the node of a two-input module).
  • a module Once a module knows what its HO-neighbor levels are at a particular one of its inputs, it subtracts them, along with the same-hierarchy-level neighbor levels, from the total input energy level of that input to get the neighbor-compensated level at that input node. This is shown in FIG. 4B where the neighbor levels for input 1 and input m are subtracted in combiners 431 and 433, respectively, from the outputs of the variable slow smoothers 425 and 427, and the higher- order neighbor levels for input 1, input m and the common energy are subtracted in combiners 431, 433 and 435, respectively, from the outputs of the variable slow smoothers 425, 427 and 429.
  • HO-neighbor levels also are used to compensate the common energy across the input channels ⁇ e.g., accomplished by the subtraction of an HO-neighbor level in combiner 435).
  • the rationale for this difference is that the common level of a module is not affected by adjacent modules of the same hierarchy, but it can be affected by a higher-order module sharing all the inputs of a module.
  • the total common signal level observed by the two- input module includes common elements of the three input module that do not belong to the latter output channel, so one subtracts the square root of the pairwise products of the HO neighbor levels from the common energy of the two-input module to dete ⁇ nine how much common energy is due solely to its interior channel (the latter one mentioned).
  • the smoothed common energy level from block 429) has subtracted from it the derived HO common level to yield a neighbor-compensated common energy level (from combiner 435) that is used by the module to calculate (in block 439) the neighbor-compensated_xcor.
  • the present invention and its various aspects may be implemented in analog circuitry, or more probably as software functions performed in digital signal processors, programmed general-purpose digital computers, and/or special purpose digital computers. Interfaces between analog and digital signal streams may be perfonned in appropriate hardware and/or as functions in software and/or firmware. Although the present invention and its various aspects may involve analog or digital signals, in practical applications most or all processing functions are likely to be performed in the digital domain on digital signal streams in which audio signals are represented by samples.

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Abstract

Une matrice variable M:N permet de moduler des signaux d'entrée audio M, chaque signal étant associé à une direction, pour les convertir en signaux de sortie audio N, chaque signal étant associé à une direction, N étant supérieur à M, M étant supérieur ou égal à 2, et N étant un nombre entier positif supérieur ou égal à 3. La matrice variable est commandée en réaction à des mesures : (1) des niveaux relatifs des signaux d'entrée, et (2) de la corrélation croisée des signaux d'entrée, de sorte qu'un champ acoustique généré par les signaux de sortie présente une image sonore compacte dans la direction primaire continue nominale des signaux d'entrée, lorsque les signaux d'entrée sont hautement corrélés, l'image s'étale du compact au large, lorsque la corrélation diminue et se divise progressivement en plusieurs images sonores compactes, chacune dans une direction associée à un signal d'entrée, lorsque la corrélation continue à diminuer pour devenir hautement non corrélée.
PCT/US2003/024570 2001-02-07 2003-08-06 Modulation spatiale de canal audio WO2004019656A2 (fr)

Priority Applications (13)

Application Number Priority Date Filing Date Title
DK03770229T DK1527655T3 (da) 2002-08-07 2003-08-06 Rumlig translation af audiokanal
CN03817877XA CN1672464B (zh) 2002-08-07 2003-08-06 音频声道空间转换
BRPI0305746-1A BRPI0305746B1 (pt) 2002-08-07 2003-08-06 Tradução espacial de canal de áudio
MXPA05001413A MXPA05001413A (es) 2002-08-07 2003-08-06 Conversion espacial de canales de audio.
CA2494454A CA2494454C (fr) 2002-08-07 2003-08-06 Modulation spatiale de canal audio
JP2004530877A JP4434951B2 (ja) 2002-08-07 2003-08-06 オーディオチャンネルの空間的変換
AU2003278704A AU2003278704B2 (en) 2002-08-07 2003-08-06 Audio channel spatial translation
US10/522,515 US7660424B2 (en) 2001-02-07 2003-08-06 Audio channel spatial translation
DE60308876T DE60308876T2 (de) 2002-08-07 2003-08-06 Audiokanalumsetzung
BR0305746-1A BR0305746A (pt) 2002-08-07 2003-08-06 Tradução espacial de canal de áudio
EP03770229A EP1527655B1 (fr) 2002-08-07 2003-08-06 Modulation spatiale de canal audio
IL165941A IL165941A (en) 2002-08-07 2004-12-22 Audio channel spatial translation
HK05106538A HK1073963A1 (en) 2002-08-07 2005-08-01 Audio channel spatial translation

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