WO2003094293A1 - Antenne a fente - Google Patents

Antenne a fente Download PDF

Info

Publication number
WO2003094293A1
WO2003094293A1 PCT/US2002/013821 US0213821W WO03094293A1 WO 2003094293 A1 WO2003094293 A1 WO 2003094293A1 US 0213821 W US0213821 W US 0213821W WO 03094293 A1 WO03094293 A1 WO 03094293A1
Authority
WO
WIPO (PCT)
Prior art keywords
antenna
slot
line
resonant
dipole
Prior art date
Application number
PCT/US2002/013821
Other languages
English (en)
Inventor
Reza Azadegan
Kamal Sarabandi
Original Assignee
The Regents Of The University Of Michigan
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by The Regents Of The University Of Michigan filed Critical The Regents Of The University Of Michigan
Priority to PCT/US2002/013821 priority Critical patent/WO2003094293A1/fr
Priority to US10/511,858 priority patent/US7075493B2/en
Publication of WO2003094293A1 publication Critical patent/WO2003094293A1/fr

Links

Classifications

    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/36—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith
    • H01Q1/38—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith formed by a conductive layer on an insulating support
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q13/00—Waveguide horns or mouths; Slot antennas; Leaky-waveguide antennas; Equivalent structures causing radiation along the transmission path of a guided wave
    • H01Q13/10—Resonant slot antennas
    • H01Q13/106—Microstrip slot antennas
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q13/00—Waveguide horns or mouths; Slot antennas; Leaky-waveguide antennas; Equivalent structures causing radiation along the transmission path of a guided wave
    • H01Q13/10—Resonant slot antennas
    • H01Q13/16—Folded slot antennas

Definitions

  • the present invention relates to efficient miniaturized resonant slot antennas, and more particularly to loaded resonant slot antennas, or folded resonant narrow slot antennas.
  • the present invention builds on the concept of a class of miniaturized, planar, re-configurable antennas, which take advantage of antenna topology for miniaturization. Using this concept, design of a miniaturized antenna as small as 0.05 ⁇ 0 x 0.05 ⁇ 0 and a fairly high efficiency of -3dBi can be accomplished. Since there are neither polarization nor mismatch losses, the antenna efficiency is limited only by the dielectric and Ohmic losses of the substrate on which the antenna is made. The bandwidth of this antenna is rather small as is the case for all miniaturized antennas.
  • Resonant antennas in general, and slot-dipoles in particular are inherently narrowband.
  • the physical aperture of the antenna is reduced and therefore, the radiation conductance of miniaturized slot antenna becomes very small.
  • an infinitesimal dipole can have an effective aperture, which is as high as that of a half wavelength dipole under the impedance matched condition.
  • One way to match the impedance of the miniaturized slot antenna is to tune it slightly off resonance, whether capacitively, or inductively. A smaller capacitance or larger inductance is needed depending on whether the antenna is tuned below or above the resonance. However, a smaller capacitance, or conversely a larger inductance, results in a narrower bandwidth.
  • the physical aperture can be increased without increasing the overall size of the antenna.
  • the present invention takes advantage of the topology of the antenna.
  • This miniaturization for a resonant slot dipole is achieved by noting that a slot dipole can be considered as a transmission line resonator, where at the lowest resonant frequency the magnetic current (transverse electric field in the slot) goes to zero at each end of the dipole antenna.
  • the antenna length / ⁇ g /2 where ⁇ g is the wavelength of the quasi-TEM mode supported by the slot line.
  • ⁇ g is the wavelength of the quasi-TEM mode supported by the slot line.
  • transmission line resonators one can also make a quarter-wave resonator by creating a short circuit at one end and an open circuit at the other end. However, creating a physical open circuit for slot lines is not practical.
  • a spiral slot of a quarter wavelength and shorted at one end behaves as an open circuit at the resonant frequency.
  • the size of the slot dipole can be reduced by approximately 50%. Further reduction can be accomplished by bending the radiating section. This bending procedure should be done so that no section of the resulting line geometry carries a magnetic current opposing the current on any other sections.
  • Figure 1 A is a magnetic current distribution on a ultra high frequency
  • Figure IB is an electric current distribution on a microstrip feed of the slot antennae of Figure 1A at the resonant frequency
  • Figure 2A is a simulated reflection co-efficient of the miniaturized UHF antennae on an infinite ground plane using Smith chart representation
  • Figure 2B is a simulated reflection co-efficient of the miniaturized UHF antennae on an infinite ground plane with magnitude of /S n / in logarithmic scale;
  • Figure 3 is a photograph of three miniaturized UHF antennas with similar geometry and dimensions while differing only in the size of the ground plane;
  • Figure 4 is a graph illustrating measured magnitude of reflection coefficient for the three miniaturized UHF slot antennas shown in Figure 3 having the same size in geometry while having different ground plane sizes;
  • Figure 5A is a graph illustrating the co-polarized and cross-polarized pattern of the miniaturized UHF antennae in H-plane;
  • Figure 5B is a graph illustrating the co-polarized and cross-polarized pattern of the miniaturized UHF antennae in E-plane;
  • Figure 7 is a simplified schematic view illustrating E-plane and H-plane of the slot antennae being tested experimentally with co-polarized and cross-polarized pattern measurements performed in the indicated principle planes;
  • Figure 8 is a graph illustrating magnetic current distribution of a half wave length and inductively terminated miniaturized slot antennae;
  • Figure 9A is a simplified schematic diagram of a transmission line model of a half wave slot antennae;
  • Figure 9B is a simplified schematic diagram of a transmission line model of an inductively terminated slot antennae;
  • Figure 9C is a simplified schematic diagram of a transmission line model of a slot antennae with two series inductive termination
  • Figure 21 is a simplified schematic view of a miniaturized folded slot antennae
  • Figure 22A is a graph illustrating impedance of a center fed miniaturized folded-slot antennae
  • Figure 22B is a graph illustrating impedance of a miniature slot antennae for comparison with Figure 12 A
  • Figure 23 is a simplified schematic diagram of a capacitively fed miniaturized folded slot antennae geometry
  • Figure 24 is a graph illustrating measurement and simulation of a miniaturized folded slot antennae return loss
  • Figure 25 A is a graph illustrating radiation pattern for the miniaturized folded slot antennae in the E-plane
  • Figure 25B is a graph illustrating the radiation pattern for the miniaturized folded slot antennae in the H-plane
  • Figure 26 is a simulated radiation pattern of the total field for the miniaturized folded slot antennae;
  • a major reduction in size is achieved by noting that a slot dipole can be considered as transmission line resonator where at the lowest resonant frequency the magnetic current (transverse electric field in the slot) goes to zero at each end of the dipole antenna.
  • the antenna length where ⁇ g is the wavelength of the quasi-TEM mode supported by the slot line.
  • ⁇ g is a function of substrate thickness, dielectric constant, and the slot width, which is shorter than the free-space wavelength.
  • the present invention incorporates the idea of non-radiating tightly coiled slot spiral.
  • a spiral slot of a quarter wavelength and shorted at one end behaves as an open circuit at the resonant frequency. Therefore a quarter-wave slot line short-circuited at one end and terminated by the non-radiating quarter-wave spiral should resonate and radiate electromagnetic waves very efficiently.
  • the size of the slot dipole can be reduced by approximately 50%. Further reduction can be accomplished by bending the radiating section. This bending procedure should be done so that no section of the resulting line geometry carries a magnetic current opposing the current on any other sections.
  • Figs. 1 A and IB shows the geometry of a typical ⁇ g /4 compact resonating slot antenna.
  • the radiating section is terminated with two identical quarter- wave non-radiating spiral slots to maintain the symmetry. It was found that by splitting the magnetic current at the end into equal and opposing magnetic currents the radiation efficiency is enhanced. Since the magnetic current distribution attains its maximum at the end of the quarter-wave line, the magnetic current in the beginning segments of a single (unbalanced) quarter-wave spiral reduces the radiation of the radiating section. But the opposite magnetic currents on two such spirals simply cancel the radiated field of each other and as a result the radiated field of the radiating section remains intact. Some additional size reduction can also be achieved, by noting that the strength of the magnetic current near the short-circuited end of the radiating section is insignificant.
  • FIG. 1 A the T-top represents a small reduction in length of the line without affecting the radiation efficiency.
  • This antenna is fed by an open ended microstrip line.
  • a quarter wavelength line corresponds to a short-circuit line under the slot, however, using the length of the microstrip line as an adjustable parameter, the reactive part of the antenna input impedance can be compensated for.
  • FIGS 1 A and IB respectively, show the electric current distribution on the microstrip feed and the magnetic current distribution on the slot of the compact UHF antenna designed to operate at 600 MHZ.
  • PiCASSO TM software was used for the simulations of this antenna.
  • the microstrip feed is constructed from two sections: 1) a 50 ⁇ line section, and 2) an open-ended 80 ⁇ line. The 80 ⁇ line is thinner which allows for compact and localized feeding of the slot. The length of this line is adjusted to compensate for the reactive component of the slot input impedance.
  • a line length of less than ⁇ m /4 compensates for an inductive reactance and a line length of longer than ⁇ m /4 compensates for a capacitive reactance.
  • ⁇ m is the guided wavelength on the microstrip line.
  • First a quarter wavelength section was chosen for the length of the microstrip line feeding the slot. In this case the simulation predicts the impedance of the slot antenna alone. Through this simulation it was found that the slot antenna fed near the edge is inductive. So a length less than ⁇ m /4 is chosen for the open-ended microstrip line to compensate for the inductive load.
  • the real part of input impedance of a slot dipole depends on the feed location along the slot and increases from zero at the short-circuited end to about 2000 ⁇ at the center (quarter wavelength from the short circuit). This property of the slot dipole allows for matching to almost all practical transmission lines.
  • the crossing of the microstrip line over the slot was determined using the full-wave analysis tool, (PiCASSOTM) and by trial-and-error.
  • the uniform current distribution over the 50 ⁇ line section indicates no standing wave pattern, which is a result of a very good input impedance match.
  • the quarter-wave radiating section of the slot dipole is composed of three slot line sections, two vertical and one horizontal. Significant radiation emanates from the middle and lower sections. Polarization of the antenna can be chosen by changing the relative size of these two sections. In this design the relative lengths of the three line sections were chosen in order to minimize the area occupied by the slot structure.
  • the slot width of the first section can be varied in order to obtain an impedance match as well. When there is a limitation in moving the microstrip and slot line crossing point, the slot width may be changed. At a given point from the short-circuited end an impedance match to a lower line impedance can be achieved when the slot width is narrowed.
  • the magnetic current over the T-top section is very low and does not contribute to the radiated field but its length affects the resonant frequency.
  • Half the length of the T-top section originally was part of the first vertical section, which is removed and placed horizontally to lower the vertical extent of the antenna.
  • the slot line sections were chosen so that a resonant frequency of
  • Figs. 2A and 2B respectively, show the simulated input impedance and return loss of the miniaturized UHF antenna as a function of frequency. It is shown that the 1.2 VSWR (-10 dB return loss) bandwidth of this antenna is around 6 MHZ which corresponds to a 1% fractional bandwidth. This low bandwidth is a characteristic of miniaturized and resonant slot dipoles.
  • the simulation also shows a weak resonance, which may be caused by the interaction between the radiating element and the non-radiating spirals.
  • FIG. 4 An antenna based on the layout shown in Figs. 1 A and IB was made on a FR4 printed-circuit-board.
  • the size of the ground plane was chosen to be 8.5cm xllcm .
  • the return loss of this antenna was measured with a network analyzer and the result is shown by the solid line in Fig. 4. It is noticed that the resonant frequency of this antenna is at 568MHz, which is significantly lower than what was predicted by the simulation. Also the measured return loss for the designed microstrip feed line was around -lOdB. To get a better return loss the length of the microstrip line had to be extended slightly.
  • Fig. 4 shows the measured return loss after the modification.
  • the gain of Antenna 3 is almost as high the gain of an ideal dipole considering the loss-tangent of the substrate used in these experiments.
  • the gain reduction as a function of ground plane size can be explained by noting that the equivalent magnetic currents that are flowing in the upper and lower side of the ground plane are in opposite directions. In the case of infinite ground plane, the upper and lower half-spaces are electromagnetically decoupled. However, when the ground plane is finite and small compared to the wavelength the radiated field from the lower half-space can reduce the radiated field from the magnetic current in the upper half-space. The level of back-radiation depends on the size of the ground plane.
  • Fig. 7 shows the direction of maximum radiation and the direction of electric field (polarization) and magnetic field at the antenna boresight.
  • Figs. 5A and 5B show the co- and cross-polarized antenna patterns in the H-plane and E-plane, respectively. It is shown that the antenna polarization remains linear on these principal planes.
  • a topology for an electrically small resonant slot antenna is demonstrated.
  • a major size reduction was achieved by constructing a ⁇ JA resonant slot rather than the traditional XJ2 antenna. This is accomplished by generating a virtual open circuit at one end of the slot. Further miniaturization was achieved by bending the slot into three pieces in order to use the area of the board more efficiently.
  • the antenna is very efficient and shows a gain as high as a dipole antenna and a 1% bandwidth. It is also shown that if the antenna is made on a small ground plane its gain will be reduced and its radiation pattern changes slightly.
  • a novel procedure according to the present invention allows the design of a miniaturized slot antenna where its dimensions (relative to wavelength) can be arbitrarily chosen depending on the application without any adverse effects on the impedance matching.
  • the antenna is first fed by a two-port microstrip line, and then the location of the null in the insertion loss (S 21 ) is found and adjusted.
  • S 21 null in the insertion loss
  • an equivalent circuit for the antenna is proposed and its parameters are extracted using a genetic algorithm in conjunction with a full-wave simulation tool.
  • a prototype antenna is designed, fabricated and its performance is evaluated experimentally.
  • BC boundary conditions
  • These two conditions are chosen to enforce zero electric current (open circuit) for a wire antenna or zero voltage (short circuit) for the slot antenna and yield a half-wave resonant antenna.
  • these alternative BCs result in a smaller resonant length than a half wavelength antenna.
  • One choice which is conducive to antenna miniaturization is the combination of a short circuit and an open circuit, which allows a shorter resonant length of A/4.
  • the choice of the two BCs is not restricted to the above conditions, whereas the effect of reactive BCs in reducing the resonant length and antenna miniaturization is investigated in what follow.
  • M 0 represents the amplitude of the magnetic current density (electric field across the slotline). This approximate form of the current distribution satisfies the short circuit boundary conditions at the end of the slot antenna. If by using an appropriate boundary condition, the magnetic current density at any arbitrary point ⁇ z' ⁇ ⁇ ( ⁇ s /4) along the length of a modified slot antenna can be maintained the same as the A/2 slot antenna, then it is possible to make a smaller slot antenna. Any size reduction of interest can be achieved so long as the appropriate BCs are in place at the proper location on the slot.
  • Figure 8 illustrates the idea where it is shown that by imposing a finite voltage at both ends of a slot, the desired magnetic current distribution on.
  • a short slot antenna can be established.
  • a series inductive element at the end of the slot antenna.
  • terminating the slot antenna with a lumped inductance or capacitance is not practical since the slot is embedded in a ground plane, which can in fact short-circuit any termination.
  • a lumped inductor could be physically realized by a compact short-circuited slotted spiral.
  • the length of the spiral slot must be less than a quarter wavelength.
  • it is preferred to use two inductive slotlines opposite of each other see Fig. 9A- 9C and Fig. 10).
  • microstrip feed is based on the ease of fabrication and stability. This feed structure is also more amenable to tuning by providing the designer with an additional parameter. Instead of short-circuiting the microstrip line over the slot, an open-ended microstrip line with an appropriate length extending beyond the microstrip-slot crossing point (additional parameter) can be used.
  • a Coplanar Waveguide (CPW) can also be used to feed the antenna providing ease of fabrication, whereas it is more difficult to tune. Usually, a metallic bridge is needed to suppress the odd mode in the CPW.
  • CPW lines also reduces the effective aperture of the slot antenna, especially when a very small antenna is to be matched to a 50 ⁇ line.
  • the center conductor in the CPW lines at 50 ⁇ is rather wide and the gap between the center conductor and the ground planes is relatively narrow. Hence, feeding the slot antenna from the center blocks a considerable portion of the miniaturized slot antenna.
  • a procedure according to the present invention provides for designing a novel miniaturized antenna with the topology discussed in the previous section.
  • a miniaturized slot antenna at 300MHz is designed. This frequency is the lowest frequency at which accurate antenna measurements can be performed in the anechoic chamber, and yet, the miniature antenna is large enough so that standard printed circuit technology can be used in the fabrication of the antenna.
  • the basic transmission line model is employed to design the antenna and then, a full-wave Moment Method analysis is used for fine tuning.
  • a slot antenna whose radiating slot segment is of a length i, should be terminated by a reactance given by
  • the required terminating reactance of X t can be constructed by two smaller series slotlines. Denoting the length of a terminating slotline by fi ", as shown in Fig. 9A-9C, the relationship between the required reactance and ⁇ " is given by
  • Z' 0s and A' 0s are the characteristic impedance and the guided wavelength of the terminating slotline.
  • a narrower slot is used to construct the terminating slotlines so that a more compact configuration can be achieved.
  • the narrower slotline has a smaller characteristic impedance and guided wavelength which results in a slightly shorter length of the termination (fi ")• Although . " is smaller than . ' the actual miniaturization is obtained by winding the terminating line into a compact spiral as seen in Fig. 10.
  • the vertical dimension (along axis) of the rectangular spiral should not exceed half of the length of the radiating slot segment ( ⁇ >).
  • This constraint on the inductive rectangular spiral is imposed so that the entire antenna structure can fit into a square area of 55mm x 55mm, which is about 0.05 ⁇ 0 x 0.05 ⁇ 0 .
  • the miniaturization is mainly achieved by the proper choice of the antenna topology. It is worth mentioning that further size reduction can be obtained once a substrate with higher permittivity is used.
  • the transmission line model was employed for designing the proposed miniature antenna. Although this model is not very accurate, it provides the intuition necessary for designing the novel topology.
  • the transmission line model ignores the coupling between the adjacent slot lines and the microstrip to slot transition.
  • IE3D a commercially available Moment Method code is used for required numerical simulations.
  • Figure 10 shows the proposed antenna geometry fed by a two-port 50 ⁇ microstrip line.
  • the two-port structure is constructed to study the resonant frequency of the antenna as well as the transition between microstrip and the slot antenna.
  • the microstrip line is extended well beyond the slot transition point so that the port terminals do not couple to the slot antenna.
  • the resonance at the desired frequency is indicated by a deep null in the frequency response of S 21 .
  • the simulated S-parameters of this two-port structure are shown in Fig. 11. This figure indicates that the antenna resonates at around 304MHz, which is close to the desired frequency of 300MHz.
  • Fig. 12 shows an equivalent circuit model for the two-port device when the transition between microstrip and slot line is represented by an ideal transformer with a frequency dependent turn ratio (n 2 ), and the slot is modeled by a second order shunt resonant circuit near its resonance.
  • the radiation conductance G s which is also referred to as the slot conductance, attains a low value that corresponds to a very high input impedance at the resonant frequency. However, this impedance would decrease considerably, when the frequency moves off the resonance. The 4MHz offset in the resonant frequency of the antenna is maintained for this purpose.
  • an equivalent circuit model for the proposed antenna is developed.
  • This model is capable of predicting the slot radiation conductance and the antenna input impedance near resonance. This approach provides a very helpful insight as to how this antenna and its feed network operate. As mentioned before, this model is also needed to find a proper matching network for the antenna.
  • the slot antenna can be modeled by a simple second order RLC circuit. Since the voltage across the slot excites the slot antenna at the feed point, it is appropriate to use the shunt resonant model for the radiating slot as shown in Fig. 12. The coupling between the microstrip and the slot is modeled by a series ideal transformer with a turn ratio n.
  • FIG. 13 shows the de-embedded Y-parameters of the two-port micro strip-fed slot antenna where the location of de-embedded ports are shown in Fig. 10. Note that these two ports are now defined at the microstrip-slot junction. According to the lumped element model of Fig. 12, the Y-parameters are given by:
  • Algorithm (GA) optimization code has been developed and implemented.
  • the sum of the squares of relative error for real and imaginary parts of Y-parameters over 40 frequency points around the resonance is used as the objective (fitness) function of the optimization problem.
  • the program can be run with different random number seeds to ensure the best result over the entire domain of the parameters space. Also, the parameters were constrained only to physical values in the region of interest.
  • the parameters of the GA optimizer are shown in Table 3. Table 4 shows the extracted equivalent circuit parameters after fifty thousands iterations.
  • the antenna's matching network can readily be designed.
  • a purely reactive admittance is sought to terminate the feed line, which in fact is the load for the second port of the two-port equivalent circuit model.
  • the explicit expression for a termination admittance (7,) to be placed at the second terminal of the two-port model in order to match the impedance of the antenna is given by:
  • Fig. 15 shows the spectral behavior of Y t for a standard 50i2 line
  • Figure 17 shows the antenna geometry matched to a 50 ⁇ line.
  • the feed line has been extended a short distance beyond the slot line.
  • the width of the microstrip where it crosses the slot is reduced so that it may block a smaller portion of the radiating slot. It is worth mentioning that the effect of the feed line width on its coupling to the slot was investigated, and it was found that as long as the line width is much smaller than the radiating slot length, the equivalent circuit parameters do not change considerably.
  • Figure 18 shows a photograph of the fabricated antenna.
  • the return loss (S n ) of the antenna was measured using a calibrated vector network analyzer and the result is shown in Fig. 16.
  • the measured results show a slight shift in the resonant frequency of the antenna ( « 1%) from what is predicted by the numerical code.
  • the errors associated with the numerical code could contribute to this frequency shift.
  • This deviation can also be attributed to the finite size of the ground plane, 0.2l ⁇ 0 * 0. ⁇ SA 0 for this prototype, knowing that an infinite ground plane is assumed in the numerical simulation.
  • the far field radiation patterns of the antenna were measured in the anechoic chamber of The University of Michigan.
  • the gain of the antenna was measured at the bore-sight direction under polarization-matched condition using a standard antenna whose gain is known as a function of frequency.
  • the gain of -3 dB, (relative to an isotropic radiator) was measured.
  • the simulated radiation efficiency is the ratio of the total radiated power to the input power of the antenna.
  • the reduction in the directivity of the slot antenna with a finite ground plane can also be attributed to the radiation from the edges and surface wave diffraction.
  • the same antenna with a slightly larger ground plane (0.58i 0 x 0.43 ⁇ 0 ) was fabricated and measured. Table 6 shows the comparison between the radiation characteristics of these two antennas and simulated results. As explained, when the size of the antenna ground plane increases, the gain of the antenna increases from -3.0 dB t to 0.6 dB f , which is almost equal to the gain of a half wavelength dipole and very close to the simulated value for the antenna gain.
  • E- and H-plane were measured and compared with the theoretical ones.
  • the simulated radiation patterns of this antenna are shown in Fig. 19. It is seen that the simulated radiation patterns of the proposed antenna with an infinite ground plane is almost the same as that of an infinitesimal slot dipole.
  • Figures 20A and 20B show the normalized co- and cross-polarized radiation patterns of the H- and E-plane, respectively, for two different ground planes. As expected, the null in the H-plane radiation pattern is filled considerably owing to the finite ground plane size.
  • E ⁇ tangential E-field
  • ⁇ 90°
  • this cut of the pattern is constant except at the dielectric-air interface where the normal E-field is discontinuous.
  • This null in the E-plane is the result of the cancellation of fields, which are radiated by the two opposing magnetic currents.
  • the equivalent magnetic currents, flowing in the upper and lower side of the ground plane are in opposite directions and consequently, their radiation in the point of symmetry at the E-plane cancel each other.
  • the contribution of the anechoic chamber, giving rise to the cross- polarized component at the low frequency of 300MHz is also a factor.
  • the radiated field of the antenna is always capable of inducing currents on the feeding cable, especially when the ground plane size is very small compared to the wavelength. Then, the induced currents re-radiate and give rise to the cross polarization. Nevertheless, both of the above mentioned sources for the cross-polarization can be eliminated by increasing the ground plane size.
  • a procedure for designing a new class of miniaturized slot antennas according to the present invention has been disclosed.
  • the area occupied by the antenna can be chosen arbitrarily small, depending on the applications at hand and the trade-off between the antenna size and the required bandwidth.
  • an antenna with the dimensions 0.05 ⁇ 0 x0.05 ⁇ 0 was designed at 300MHz and perfectly matched to a 50 ⁇ transmission line.
  • a substrate with a low dielectric constant of ⁇ r 2.2 was used to ensure that the dielectric material would not contribute to the antenna miniaturization.
  • An equivalent circuit for the antenna was developed, which provided the guidelines necessary for designing a compact loss-less matching network for the antenna. To validate the design procedure, a prototype antenna was fabricated and measured at 300MHz.
  • a perfect match for this very small antenna was demonstrated with a moderate gain of -3.0dBj when the antenna is fabricated on a very small ground plane with the approximate dimensions of 0.2 ⁇ 0 x0.2 ⁇ 0 .
  • the gain of this antenna can increase to that of a half- wave dipole when a slightly larger ground plane of about 0.5 ⁇ 0 x 0.5 ⁇ 0 is used.
  • the fractional bandwidth for this antenna was measured to be 0.4%.
  • a new miniaturized antenna structure according to the present invention is disclosed with a larger radiation conductance (physical aperture), bandwidth, and efficiency, while maintaining the size of the antenna. Conversely, maintaining the bandwidth and efficiency, this structure can be further miniaturized (0.03 ⁇ 0 x 0.03 ⁇ 0 ).
  • the structure according to the present invention is based on a folded slot design whose geometry is shown in Fig. 21.
  • the physical aperture of the miniaturized folded slot is twice as large as that of the miniaturized slot illustrated in Fig. 1A, and therefore, should demonstrate a radiation conductance four times as high as the design of Fig. 1 A.
  • the antenna was center-fed with a CPW line and was simulated using a commercially available Moment Method code.
  • Figures 22A and 22B show a comparison between the input impedance of the folded design, and the single slot of Fig.
  • Figure 23 shows the miniaturized folded slot antenna matched to a 50 ⁇ CPW line.
  • the proper value of the capacitance to be inserted in the feed is determined from a second order resonant equivalent circuit model. These model parameters can be extracted using a full wave simulation of the antenna structure.
  • the folded slot has a resonance at 337.9 MHZ with a radiation resistance of about 5K ⁇ , as shown in Fig. 22A.
  • the antenna is matched to 50 ⁇ at a slightly lower frequency of 336.1 MHZ. (See Fig.24).
  • Fig. 24 shows the same data sets for a miniaturized single slot antenna, having approximately the same size. Comparison of Figs. 24 and 16, clearly indicates an increase in the -10 dB return-loss bandwidth of the antenna. Table 7, shows a comparison between both simulated and measured bandwidths of these two antennas.
  • Figs. 25 A and 25B depict the E- plane and H-plane radiation patterns of the antenna and the results are shown in Figs. 25 A and 25B.
  • Fig. 26 depicts the simulated radiation pattern of the total field and shows that this structure has a pattern very similar to that of a small dipole.
  • the cross polarization components are negligible in the principal planes.
  • the observed cross-polarized radiation is believed to emanate from feeding cables rather than from the antenna itself.
  • a miniaturized folded slot antenna according to the present invention presents an improved configuration for miniaturized slot antennas, which demonstrates wider bandwidth and higher radiation efficiency.
  • the bandwidth of the antenna was increased by 100% with a slight increase in the gain of the antenna.

Landscapes

  • Waveguide Aerials (AREA)

Abstract

L'invention concerne des aspects de conception et les résultats mesurés d'une antenne à fente étroite résonante et miniaturisée. Des éléments de rayonnement à fente étroite résonante présentent une géométrie plane et sont capables d'émettre une polarisation verticale quand ils sont placés presque à l'horizontal. Une antenne à fente étroite résonante selon l'invention simplifie l'adaptation d'impédance. Des dipôles de fente peuvent être excités par une ligne microruban et peuvent être adaptés à des impédances de ligne arbitraires par déplacement du point d'alimentation le long de la fente. Une miniaturisation de l'antenne peut être obtenue par mise en oeuvre d'un substrat à permittivité ou perméabilité élevée et de matériaux de superstrate et/ou par mise en oeuvre d'une topologie d'antenne appropriée. Une miniaturisation est obtenue par utilisation d'une géométrie unique pour une antenne à fente étroite résonante. Un élément de rayonnement très efficace est prévu. L'exécution virtuelle de la condition limite requise au niveau de l'extrémité d'une antenne à fente permet de réduire la zone occupée par l'antenne résonante. Afin d'obtenir les conditions limites virtuelles requises, deux courts-circuits situés au niveau de l'extrémité de la fente résonante sont remplacés par des conditions limites réactives, notamment des conditions limites inductives ou capacitives, notamment des charges inductives ou capacitives.
PCT/US2002/013821 2002-05-01 2002-05-01 Antenne a fente WO2003094293A1 (fr)

Priority Applications (2)

Application Number Priority Date Filing Date Title
PCT/US2002/013821 WO2003094293A1 (fr) 2002-05-01 2002-05-01 Antenne a fente
US10/511,858 US7075493B2 (en) 2002-05-01 2002-05-01 Slot antenna

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
PCT/US2002/013821 WO2003094293A1 (fr) 2002-05-01 2002-05-01 Antenne a fente

Publications (1)

Publication Number Publication Date
WO2003094293A1 true WO2003094293A1 (fr) 2003-11-13

Family

ID=29398918

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/US2002/013821 WO2003094293A1 (fr) 2002-05-01 2002-05-01 Antenne a fente

Country Status (1)

Country Link
WO (1) WO2003094293A1 (fr)

Cited By (10)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1628359A1 (fr) 2004-08-21 2006-02-22 Samsung Electronics Co., Ltd. Petite antenne planaire ayant une bande passante améliorée et petite antenne micro-ruban
EP1628360A1 (fr) 2004-08-21 2006-02-22 Samsung Electronics Co., Ltd Petite antenne planaire ayant une bande passante améliorée et petite antenne redresseuse pour RFID et transpondeur détecteur sans fil
WO2006065693A1 (fr) * 2004-12-14 2006-06-22 Intel Corporation Antenne a fente comportant une diode a capacite variable de microsysteme electromecanique pour l'accord de frequence de resonance
GB2429845A (en) * 2005-09-05 2007-03-07 Motorola Inc Antenna and RF terminal incorporating the antenna
CN103022654A (zh) * 2011-09-23 2013-04-03 深圳光启高等理工研究院 一种双极天线及移动多媒体广播装置
WO2013093466A1 (fr) 2011-12-23 2013-06-27 The University Court Of The University Of Edinburgh Elément et dispositif d'antenne comportant de tels éléments
WO2015114004A1 (fr) * 2014-01-31 2015-08-06 Thomson Licensing Résonateur à ligne de fentes
WO2018201675A1 (fr) * 2017-05-05 2018-11-08 深圳市景程信息科技有限公司 Procédé d'analyse de mode de résonance pour antenne à fentes à double fréquence
CN112821052A (zh) * 2021-02-08 2021-05-18 福耀玻璃工业集团股份有限公司 天线、交通工具玻璃板组件和交通工具
CN113964513A (zh) * 2021-10-25 2022-01-21 国网天津市电力公司电力科学研究院 一种无线通信微波天线及其成型方法

Non-Patent Citations (3)

* Cited by examiner, † Cited by third party
Title
AZADEGAN R ET AL: "DESIGN OF MINIATURIZED SLOT ANTENNAS", IEEE ANTENNAS AND PROPAGATION SOCIETY INTERNATIONAL SYMPOSIUM. 2001 DIGEST. APS. BOSTON, MA, JULY 8 - 13, 2001, NEW YORK, NY: IEEE, US, vol. 4 OF 4, 8 July 2001 (2001-07-08), pages 565 - 568, XP001072153, ISBN: 0-7803-7070-8 *
PEROULIS D ET AL: "A PLANAR VHF RECONFIGURABLE SLOT ANTENNA", IEEE ANTENNAS AND PROPAGATION SOCIETY INTERNATIONAL SYMPOSIUM. 2001 DIGEST. APS. BOSTON, MA, JULY 8 - 13, 2001, NEW YORK, NY: IEEE, US, vol. 1 OF 4, 8 July 2001 (2001-07-08), pages 154 - 157, XP001072179, ISBN: 0-7803-7070-8 *
SARABANDI K ET AL: "Design of an efficient miniaturized UHF planar antenna", IEEE ANTENNAS AND PROPAGATION SOCIETY INTERNATIONAL SYMPOSIUM. 2001 DIGEST. APS. BOSTON, MA, JULY 8 - 13, 2001, NEW YORK, NY: IEEE, US, vol. 1 OF 4, 8 July 2001 (2001-07-08), pages 446 - 449, XP010564673, ISBN: 0-7803-7070-8 *

Cited By (18)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7355559B2 (en) 2004-08-21 2008-04-08 Samsung Electronics Co., Ltd. Small planar antenna with enhanced bandwidth and small strip radiator
US7289076B2 (en) 2004-08-21 2007-10-30 Samsung Electronics Co., Ltd. Small planar antenna with enhanced bandwidth and small strip radiator
EP1628359A1 (fr) 2004-08-21 2006-02-22 Samsung Electronics Co., Ltd. Petite antenne planaire ayant une bande passante améliorée et petite antenne micro-ruban
EP1628360A1 (fr) 2004-08-21 2006-02-22 Samsung Electronics Co., Ltd Petite antenne planaire ayant une bande passante améliorée et petite antenne redresseuse pour RFID et transpondeur détecteur sans fil
US7262740B2 (en) 2004-08-21 2007-08-28 Samsung Electronics Co., Ltd. Small planar antenna with enhanced bandwidth and small rectenna for RFID and wireless sensor transponder
WO2006065693A1 (fr) * 2004-12-14 2006-06-22 Intel Corporation Antenne a fente comportant une diode a capacite variable de microsysteme electromecanique pour l'accord de frequence de resonance
US7348928B2 (en) 2004-12-14 2008-03-25 Intel Corporation Slot antenna having a MEMS varactor for resonance frequency tuning
GB2429845B (en) * 2005-09-05 2008-02-13 Motorola Inc Antenna and RF terminal incorporating the antenna
GB2429845A (en) * 2005-09-05 2007-03-07 Motorola Inc Antenna and RF terminal incorporating the antenna
CN103022654B (zh) * 2011-09-23 2015-02-04 深圳光启创新技术有限公司 一种双极天线及移动多媒体广播装置
CN103022654A (zh) * 2011-09-23 2013-04-03 深圳光启高等理工研究院 一种双极天线及移动多媒体广播装置
WO2013093466A1 (fr) 2011-12-23 2013-06-27 The University Court Of The University Of Edinburgh Elément et dispositif d'antenne comportant de tels éléments
US9899737B2 (en) 2011-12-23 2018-02-20 Sofant Technologies Ltd Antenna element and antenna device comprising such elements
WO2015114004A1 (fr) * 2014-01-31 2015-08-06 Thomson Licensing Résonateur à ligne de fentes
WO2018201675A1 (fr) * 2017-05-05 2018-11-08 深圳市景程信息科技有限公司 Procédé d'analyse de mode de résonance pour antenne à fentes à double fréquence
CN112821052A (zh) * 2021-02-08 2021-05-18 福耀玻璃工业集团股份有限公司 天线、交通工具玻璃板组件和交通工具
CN113964513B (zh) * 2021-10-25 2024-01-26 国网天津市电力公司电力科学研究院 一种无线通信微波天线及其成型方法
CN113964513A (zh) * 2021-10-25 2022-01-21 国网天津市电力公司电力科学研究院 一种无线通信微波天线及其成型方法

Similar Documents

Publication Publication Date Title
US7075493B2 (en) Slot antenna
Fallahpour et al. Antenna miniaturization techniques: A review of topology-and material-based methods
Zhu et al. A compact transmission-line metamaterial antenna with extended bandwidth
Emadian et al. Very small dual band-notched rectangular slot antenna with enhanced impedance bandwidth
Sarabandi et al. Design of an efficient miniaturized UHF planar antenna
Mitra et al. A miniaturized ring slot antenna design with enhanced radiation characteristics
US7911386B1 (en) Multi-band radiating elements with composite right/left-handed meta-material transmission line
Rezaeieh et al. Bandwidth and directivity enhancement of loop antenna by nonperiodic distribution of mu-negative metamaterial unit cells
Patil Enhancement of bandwidth of rectangular patch antenna using two square slots techniques
Shih et al. A compact, broadband spiral antenna with unidirectional circularly polarized radiation patterns
Jung et al. Electromagnetically coupled small broadband monopole antenna
Shao et al. A single-layer miniaturized patch antenna based on coupled microstrips
Rahman et al. Circularly polarised patch antenna with periodic structure
WO2003094293A1 (fr) Antenne a fente
Shankar et al. A Fractal Monopole Antenna With Dual Polarization Reconfigurable Characteristics for X-Band Applications
Wang et al. Low-Profile Planar Ultrawideband Modular Antenna Array Loaded with Parasitic Metal Strips
Nath et al. A compact flower-shaped printed monopole MIMO antenna for wideband applications
Xiong et al. A broadband low profile patch antenna of compact size with three resonances
Mukherjee et al. Dielectric resonator antennas
Abdel-Rahman Design and development of high gain wideband microstrip antenna and DGS filters using numerical experimentation approach
Rao et al. A novel complementary slotted split ring resonator loaded truncated arc patch antenna with enhanced performance
Abdelaziem et al. High gain and efficiency dual-band antenna using meta-surface
Denidni et al. High gain microstrip antenna design for broadband wireless applications
Jung et al. Design and performance of small electromagnetically coupled monopole antenna for broadband operation
Sheeja et al. Compact tri-band metamaterial antenna for wireless applications

Legal Events

Date Code Title Description
AK Designated states

Kind code of ref document: A1

Designated state(s): CA US

WWE Wipo information: entry into national phase

Ref document number: 10511858

Country of ref document: US