WO2001093463A2

WO2001093463A2 - A method for synchronizing a radio receiver to a received data bit stream

Info

Publication number: WO2001093463A2
Application number: PCT/US2001/017080
Authority: WO
Inventors: Thomas J. Campana, Jr.; Al Jewer; Gordon Smith; Robert Fries; Michael P. Ponschke, Sr.
Original assignee: Sophia Communications, Inc.
Priority date: 2000-05-31
Filing date: 2001-05-30
Publication date: 2001-12-06
Also published as: WO2001093463A3; AU2001265011A1

Abstract

A method is provided for continuous received data bit stream synchronization in a receiver/transmitter in a wireless communications system through the use of a fixed frequency reference tone, which is simultaneously included in a same channel as a data signal being transmitted. The reference signal comprises a sinosoidal signal having fixed amplitude, fixed phase, and fixed frequency. This signal has a periodic association with the transmitted data signals and can be used to align the data sampling circuits of the receiver with the demodulated data signal. By comparing the received tone signal with a known reference set of parameters stored in memory, the relative amplitudes and timing alignment errors can be quickly determined, and appropriate corrective signals can be sent to data extraction modules. This allows accurate reception of large messages without the need to send multiple bit-synchronization preambles.

Description

A METHOD FOR SYNCHRONIZING A RADIO RECEIVER TO A RECEIVED

DATA BIT STREAM

FIELD OF THE INVENTION

This invention relates to the field of data reception in wireless data

communications systems, and more particularly to a method for synchronizing

received data bit streams in radio receivers.

BACKGROUND OF THE INVENTION

In a wireless data communications system that is characterized by high

speed serial data bursts that are addressed to a unique radio receiver followed by

long periods of inactivity, there is a need to synchronize the receiver to the

timing characteristics of a data bit stream to enable the receiver to accurately

extract information contained in the data bit stream. This precision timing

alignment requirement is well known in the art, and typically is solved using

techniques which allow the alignment of the receiver sampling circuit to center

on each data bit time period. All such techniques occupy a portion of the

transmission time resulting in loss of usable bandwidth.

Typically, a transmission will include a bit-synchronization preamble

pattern of a fixed-length sequence of alternating ones and zeroes. A receiver,

which is conditioned to receive and synchronize with these signals, will adjust its

internal sampling circuit timing to bring the sampling circuit into alignment with

the timing of data bits in this preamble. The drawback is that although a single preamble period will synchronize the receiver timing circuits, internal receiver

timing tolerances and deviations will over time cause the receiver timing to

become misaligned with the incoming data by the end of the message, resulting

in sampled data errors. Thus, for a synchronization preamble of 256 bit periods

for example, typical receiver architectures will limit the number of data bits that

can be received based on that 'single preamble, typically 500 to 2000 data bits.

The larger the data block, the higher the probability of errors due to timing error

buildup. This type of synchronization is the easiest and least expensive to

implement and thus is widely used.

An alternate technique for synchronization is to allow a predetermined

number of data bits to be included in the data stream which will be of a known

state and used solely for synchronization. For example, such pilot symbols may

be every fourth or every tenth bit. Such architectures provide for continuous

synchronization and allow for the transmission and reception of large blocks of

data. However, as can be seen/such a technique also consumes large amounts of

available bandwidth through the inclusion of non-data in the message, e.g 25%.

To overcome ^' the drawbacks of conventional implementations of bit

synchronization techniques in wireless data communications systems, the

present invention provides for a fixed sinusoidal reference signal to be imbedded

within a transmitted message. After such reference signal is extracted at a

receiver, the receiver timing circuits are then calibrated to optimize data extraction using time-synchronization information contained in the reference

tone. Since such a reference signal is continuous during a data transmission, data

bit synchronization is also continuous, and longer messages can be used than

would heretofore be possible using conventional techniques. Since any

transmitted message requires .the bit-synchronization preamble, a single large

message will have significant overhead reductions when compared with several

small messages (including several preambles) that would be required to send

identical information. ■ ^'

An exemplary reference signal is a sinusoidal signal having fixed

amplitude, fixed phase, and fixed^* frequency. The signal has a periodic

association with the data signals and can be used to align the data sampling

circuits with the demodulated signal. By comparing the received tone signal

with a known reference set of parameters stored in memory, the relative

amplitudes and timing alignment errors can be quickly determined.

SUMMARY OF THE INVENTION

It is an object of the current invention to provide a method for accurately

and continuously-synchronizing data sampling timings to an incoming signal in

a wireless communications system. It is a specific object of the current invention

to provide such a system easily and inexpensively, with a minimum modification

to the existing systems. According to one aspect of the current invention, the method is used for

calibrating a remote transmitter/ receivers in a communication system including

at least one base stations and several remote transmitter/ receivers. In such

systems a number of transmissions are transmitted and received, and

communications contain both a data signal and a reference signal. Each such

signal is addressed to one or. more such receiver/ transmitter at any particular

time. The method includes embedding calibration parameters in each base

station reference signal, after which the calibration parameters are extracted at

each transmitter/ receiver. The next step is the calibrating of the

transmitter/ receiver after extracting of the calibration parameters.

According to a third aspect of the invention, the communications are

transmitted as modulated signals. As a result, the reference signals must be

demod ilated in each transmitter/ receiver.

According to a fourth aspect of the invention, the reference signal is a

sinusoidal signal having a fixed frequency, a fixed amplitude and a fixed phase

angle.

According to a fifth aspect of the invention, the data signal has a different

frequency from that of the reference signal.

According to a sixth aspect of the invention, the modulated data signal

has a pair of passband frequencies located in a communications channel, whereas the reference signal has a center frequency that is located outside the pair of

passband frequencies.

According to a seventh aspect of the invention, the reference signal has a

center frequency that is located inside a pair of passband frequencies of the

communications channel.

According to an eighth aspect of the invention, the processing may be

done by either a digital signal processor and/ or a microprocessor.

According to a ninth aspect of the invention, the reference signal contains

any or all of the following characteristics: a voltage amplitude; a frequency; and

a phase angle.

According to an tenth aspect of the invention, the modulated signal is

encoded using a phase shift modulation technique which may include any or all

of the following: binary phase shift keying, quadrature phase shift keying, and

quadrature phase shift keying, and quadrature amplitude modulation

According to a final aspect of the invention, a method for received data bit

stream synchronization in a receiver in a wireless data communications system

includes transmitting a modulated signal containing a data signal and a reference

signal. The signals are received in a remote receiver, and are demodulated. The

next step is the extracting several characterization signals from the demodulated

reference signal. A number of calibration signals are similarly generated.

Finally, the receiver is calibrated using these calibration signals, and the data - extraction circuits of the receiver are synchronized with the received data bit

stream.

BRIEF DESCRIPTION OF THE DRAWINGS

Figure 1 shows a conventional wireless data communications system.

Figure . 2 shows a timing diagram of a transmission signal having

conventional bit-synchronization preamble. ^' ,

Figure 3 shows an amplitude vs frequency plot showing a reference tone

relative and a modulated data signal implemented according to^' the present

invention.

Figure 4 shows the details of the fixed frequency reference tone.

Figure 5 shows a block diagram of the steps used to transmit and receive

timing information and to provide continuous calibration of a receiver.

Figure 6 shows a diagram of the steps used to create a signal using two

distinct signals each having a separate frequency.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

In a wireless data communications system a method is disclosed for

synchronizing data extraction circuitry in a receiver to the timings of the data in •

an incoming signal. Further, the method provides for continuous data

synchronization over the entirety of a message. Figure 1 shows a conventional wireless data communications system

having a base station 10 which has a transmitter/ receiver 12 in communications

with a plurality of RTs 14. Each message transmitted by base station 10 includes

an identifier address that directs the message to one or more of the plurality of

RTs 14. Each one of the plurality of RTs 14 is configured to receive messages

transmitted by base station 10 at a predetermined frequency, modulation

scheme, and data rate.

Figure 2 shows a timing diagram of an exemplary data message of a

conventional data communications system comprising two portions. An

unmodulated first portion, or preamble 16, represents the time required for the

receiver to iteratively sample and determine the unique timings associated with

the incoming data bit stream, during which no data can be accurately processed.

A second portion 18 is a modulated data message. A unique data word 20

separates the first and second portions of the message. Word 20 is used for

notifying a computing device in the receiver of the location of the beginning of

valid data. As previously discussed, the length of preamble 16 can typically be

up to 25% of the total time of the message.

The bit-synchronization of a receiver typically uses a process of over-

sampling an incoming bit stream to determine accurate timings of a sequence of

"zero crossings" of an alternating incoming preamble signal. These zero

crossings then define the duration of a single bit period, and corrections are rnade to the timing and sampling circuits of the receiver to align an internal

receiver clock so as to ideally sample the data in the center of the data period. As

previously discussed this timing determination will remain valid for only a

limited time due to .errors associate with the internal clock tolerances and to

errors associated with the initial signal sampling due to signal jitter and other

over-the-air disturbances which mutate a received signal.

Figure 3 shows a frequency plot of the magnitude of a reference tone

relative to the modulated^* data signal. Within an FCC spectral mask 22 for a

I given channel 24, a frequency roll-off of filters typically requires a data signal 26

to be centered within mask 22 and to begin attenuation almost immediately, so

as to insure that harmonics frequencies of a transmitted signal will not spill over

and interfere with adjacent channels. This leaves an area at the shoulders 28 that

can be used for additional frequency tones, such as tone 30, providing that the

added signal is contained within mask 22 using a high performance filter.

A Digital Signal Processor (DSP) can provide sufficient processing power

that enables a pair of high performance digital filters to be implemented on a

single integrated^' circuit that can filter both tone 30 and data signal 26. Such

filtering can take the form of a finite impulse response (FIR) filter and can be

typically configured as a low pass filter addressing the lower-frequency data

signal 26 and a band-pass filter for filtering the higher-frequency tone signal 30.

While a design choice to implement digital filters in the DSP is a more practical embodiment in light of the advances in DSPs, alternative filtering techniques

involving both external hardware and software and both digital and analog can

be used. For example, external filtering integrated circuits are widely available

that operate simply by loading a plurality of filter coefficients into the chip.

Further, computing devices other than DSPs, such as powerful microprocessors

and microcontrollers, can be used in lieu of, or in conjunction with, a DSP and/ or ^'

such external filters to perform the filtering and calibration processing.

For example, assuming an exemplary channel width of 5 KHz, or ± 2500

Hz, with the upper limits of spectral mask 22 established at +2000 Hz, data signal

26 can be attenuated to a level of -3db by +1700 Hz and -30db by +2000 Hz. The

margin 28 between signal 26 at a representative 1700 Hz and spectral mask 22

allows the inclusion of tone 30 at a lower amplitude than the peak value of signal

26 at the channel midpoint. The requirements for such inclusion is that the

filtering be accomplished using a filter typically having a much higher number of

poles, 128 poles for example, so that the magnitude attenuation vs. frequency

curve of tone 30 is either identical to or much steeper than the attenuation curve

of signal 26.

While tone 30 can have a voltage amplitude equal to 'that of signal 26, an

implementation of such a filter is impractical for -■a tone situated so close to the

spectral mask, even if in digital form. The optimum amplitude of tone 30 is

determined by the trade-off between the signal magnitude necessary to be easily detected and interpreted at a remote RT and the signal that can be practically

implemented without undue system complexity and cost. It should be noted

that a two-way communications system requires that tone 30 to be imbedded at

both ends of the communication link. So while a single base station can have

complex and expensive implementation circuitry,. a plurality of inexpensive RTs

cannot practically -and economically use the same circuitry and components.

Further, although signal 26 is ideally located in the center of a channel,

alternative embodiments allow it to be located at other frequencies . in the

channel, or even in another channel. For example, due to the presence of tone 30

in the upper sideband at an exemplary frequency of 2 KHz, moving the center

frequency lower in the band by a few hundred hertz would provide better

frequency separation and thus better insulation against destructive inter-symbol

interference. Alternatively, having tone 30 in an adjacent channel that is within

the frequency range of an RT having simultaneous dual channel processing

capability can enable identical filtering processing of tone 30, although such an

approach is less practical and would not be an efficient use^' of a valuable

communications channel.

Figure 4 shows the details of the fixed frequency reference tone 30 vs time.

An exemplary reference tone 30 is a fixed sinusoid having a one volt Amplitude,

a 0° Phase angle, and a 2 KHz Frequency." This Amplitude-Phase-Frequency

(APF) signal 30 is based on a master clock located at base station 10 and is transmitted with each message originating at base station 10. An exemplary RT

(not shown) located near base station 10 might receive APF signal 30 and sample

it as having 0.9 volts amplitude. Since the RT has an internal table of values, and

knows that the signal has been attenuated by the amount of 0.1 volt, it can either

adjust its receiver sensitivity or a plurality of mathematical gain coefficients in

the DSP. • ^'

While an exemplary modulation scheme of a preferred embodiment

would be a phase shift keying architecture, such as binary phase shift keying

(BPSK) and quadrature phase shift keying (QPSK), having the ability to calibrate

the amplitude of a received signal allows the use of modulation schemes

incorporating amplitude modulation, such as Quadrature Amplitude

Modulation (QAM.) In QAM architectures having a higher number of

constellation points (e.g. 16QAM and 24QAM), the ability to differentiate

between minute changes in amplitude becomes critical. Additionally, in some

applications, such a detected attenuation can be used to proportionately adjust

the output transmitter power of the RT to compensate for an identical signal

degradation that can be expected on the uplink communications path.

A principal advantage of the fixed frequency of APF signal 30 is the ability

to calibrate a highly accurate frequency for use in* both receiving and

transmitting sections of the RT according to the timing contained in APF signal

30. Since the received 2 KHz APF signal 30 is superimposed on a carrier frequency of several hundreds of megahertz, minute adjustments made to bring

the 2KHz frequency signal into alignment determines the amount of offset of the

carrier frequency and the RT can then make any necessary calibration

corrections. This capability is of less importance in the receiver portion of the RT

since a DSP-based system typically, has wide frequency capture capability.

However, the fixed frequency reference provided by APF signal 30, does provide

the capability ^' of tuning to a transmit frequency that is referenced to a high

stability clock in base station 10, thereby reducing exposure to out-of-band

transmissions that would typically result from component tolerances and drift in

the RT.

In using the phase information contained in APF signal 30, there is a

condition that a numerical relationship exists between the frequency of the data

and the frequency of the reference signal, and that this relationship is known by

all elements of the data communications system. Specifically, at any given zero

crossing of either the data or the reference signal, a phase angle can be

determined that identifies a difference error between the two signals, such that

corrective actions can be taken to bring the signal timings back into alignment.

. In an example to show the use of the fixed phase of APF signal 30, if an

exemplary RT (not shown) is located far from base station 10 and receives an

APF signal 30 having data signal 26 that is not easily discernible, a monitoring of

the phases of APF signal 30 can still identify the appropriate data sampling times. If, for example, data signal 26 has a corner frequency of 1725 Hz in a

5KHz channel and a data rate of 3450 symbols per second, a data clock occurs

every 290 microseconds. If APF signal 30 has a fixed frequency of 2300 Hz and a

period of 435 microseconds, the two signals will coincide at every third cycle of

data signal 26- and every second cycle, of APF signal 30, or every 870

microseconds. Thus, a sequence of clock pulses in the RT based on APF signal 30

will start at the positive a zero crossing 32 of APF signal 30, with a second clock

at a phase angle of 240° (2/3 of 360°) at location 34, a third clock pulse at 120°

(4/3 of 360°) at location 36, and a fourth clock pulse repeating at 0° (6/3 of 360°)

at location 38. For clarity purposes, data bits are shown as blocks in the timing

diagram in figure 4 rather than as more typical I-Q constellation vectors.

However, the center of each block represents an idealized data sampling time.

In this way an accurate data clock can be constructed from APF signal 30

even though data signal 26 is unable to provide, the clocking information. Again,

the processing power of the DSP can easily extract the data information from

data signal 26 if the samples are made at the appropriate times, typically in the ^■

center of a data time period. The foregoing examples are for illustration

purposes only and are not intended to be limiting to the scope of the present

invention.

Figure 5 shows a block diagram of the steps used to transmit and receive

timing information and to provide continuous calibration to a receiver. By •extracting the timing information in parallel with the processing of the data

contained in a message, a continuous correction of the timing for data sampling

of the receiver can be made. Since the DSP processes both portions of the

message, comparisons made at any point in time will yield a phase angle of the

reference sinusoidal waveform, which indicates the data location within the

waveform over time.

Initially, a computer in a .base station 10 or other transmitting device

combines in step 40 a data signal 26 and a reference signal 30 in memory and

modulates the resultant signal according to the modulation technique being used

in step 42. The transmitter transmits the modulated signal in step 44, which is

received in a remote receiver in step 46. The receiver then demodulates the

combined signal in step 48 and separates the two signals, with the reference

signal being processed in step 50 to extract the APF signal parameters to enable

the continuous calibration of the receiver in step 52 and the data signal being

processed using the re-calibrated receiver in step 54. A dashed line separates the

actions performed in the transmitting device (40, 42 and 44) and the_. steps

performed in the receiving device (46, 48, 50, 52 and 54.)

Figure 6 shows a diagram of the steps used to create a signal using two

distinct signals each having a separate modulation frequency. In step 56, a data*

signal is digitally modulated in the DSP at a first frequency. In step 58, the tone

signal 30 is digitally modulating in the DSP as the addition of a modulated sine wave and a modulated cosine wave, both at a representative value of 2KHz. In

step 60, the data and the two signals are added so as to produce a combined third

signal, which is digitally filtered in step 62 to eliminate spectral components that

are not contained within a predetermined spectral mask.

In the foregoing discussion, all dependencies on component tolerances

and temperature and life drift in each one of the plurality of RTs 14 are removed

through the referencing of all receiver and transmitter timing activities to the

stable precision clock of base station .10: Such referencing is accomplished

through the inclusion of APF signal 30 in the same transmission channel that

heretofore would have been solely used for data and lead-in bit-synchronization

sequences. This APF signal 30 improves system messaging capacity over

systems using conventional methods by through significant reductions in the

transmission time used for conventional bit-synchronization sequences and,

more specifically, by allowing the transmission of larger blocks of data that are

continuously ^{^}synchronized. This large block communications . capability

eliminates the separate preambles that would be appended to each one of the

smaller messages.

Numerous modifications to and alternative embodiments of the present

invention will be apparent to those skilled in the art in view of the foregoing

description. Accordingly, this description is to be construed as illustrative only

and is for the ^■ purpose of teaching those skilled in the art the best mode of carrying out the invention. Details of the methods may be varied without

departing from the spirit of the invention, and the exclusive use of all

modifications which come within the scope of the appended claims is reserved.

Claims

What is claimed is:

1. A method for received data bit stream synchronization in a receiver in a

wireless data communications system, comprising the steps of:

transmitting a complex signal comprising a data signal and a reference

signal, wherein the reference signal further comprises a synchronization signal,

said synchronization signal being present during the substantially entire

duration of the data signal; ^' ,

receiving said complex signal in a remote receiver;

extracting, the • reference signal and the data signal from said received

complex signal;

extracting the synchronization signal from the reference signal; and

. synchronizing the data signal with the synchronization signal during the

duration of the received data signal.

2. The method of claim 1, wherein:

the complex signal is a modulated signal;

the extracting of the reference signal further comprises demodulation and

filtering; and

the extracting of the data signal further comprises demodulation and

filtering.

3. The method of claim 2, wherein the synchronization signal further comprises

an amplitude, a frequency, and a phase; and wherein the synchronizing of the data signal further comprises processing dependent upon said amplitude,

frequency, and phase.

4. The method according to claim 1, wherein said reference. signal is a sinusoidal

signal having a fixed frequency, a fixed amplitude and a fixed phase angle.

5. The method according to claim 4, wherein said data signal comprises a

spectrum about a center frequency and wherein said center frequency is

different from said reference signal frequency.

6. The method according to claim 5, wherein said reference signal has a center

frequency that is located outside a pair of passband frequencies of the

modulated data signal.

7. The method according to claim 6, wherein said reference signal has a center

frequency that is located inside a pair of passband frequencies of a frequency

channel. •

8. The method according to claim 1, wherein the transmitter is a base station

and the receiver is one of a plurality of receivers in communications with said

base station.

9. The method according to claim 3, wherein the processing means comprises at

least one from the group consisting of a digital signal processor and a

microprocessor.

10. The method according to claim 1, wherein said modulated signal is encoded

using a phase shift modulation technique consisting of one or more from the group consisting of: binary phase shift keying, quadrature phase shift keying,

and quadrature amplitude, modulation.

11. The method according to claim 1, wherein the modulated signal incorporates

at least two ^' independent signals having different frequencies in a narrow

frequency spectral mask using a digital signal processor and is created by a

process comprising the steps of:

digitally modulating in the^', digital .signal processor a first signal at a first

frequency;

digitally modulating in the digital signal processor a second signal at a

second frequency;

adding the first and second signals so as to produce a combined third

signal; and

digitally filtering said third signal to eliminate spectral components that

are not contained within a predetermined spectral mask.

12. The process according to claim 11, wherein the first signal comprises a data

signal to be transmitted.

13. The process according to claim 11, wherein the second signal ^'comprises a

sinusoidal signal that has a fixed frequency, a fixed phase angle, and a fixed

amplitude.

•14. The method according to claim 2, wherein the modulated reference signal is

included in a frequency spectral mask that includes a modulated data portion of the transmission signal, but is offset in frequency from the modulated data

portion so as to be separately detectable.

15. A method for calibrating the timings of data extraction modules in a remote

transmitter/ receivers in a communication system comprising at least one base

stations and a multiplicity of remote transmitter/ receivers, in which a

multiplicity of communications* are transmitted and received, and in which each

communication comprises a data signal and a reference signal, and each

communication is addressed to one or more such receiver/ transmitter at any

particular time, the method comprising:

embedding a plurality of calibration parameters in each transmitted

reference signal, said reference signal being present during the substantially

entire duration of the data signal;

receiving said communications signal in a remote transmitter/ receiver;

extracting the calibration parameters at each transmitter/ receiver; and ^'

calibrating the transmitter/ receiver after extracting of the calibration

parameters.

16. The method of claim 15, wherein the communications are transmitted as

modulated signals, and further comprising demodulating said reference signal in

said transmitter/ receiver.

17. The method according to claim 16, wherein said demodulated reference

signal is a fixed frequency- sinusoid, having a fixed amplitude and a fixed phase

angle.

18. The method according to claim 17, wherein said data signal comprises a

spectrum about^* a center frequency and wherein said center frequency is different

from said reference signal frequency. ..

19. The method according to claim.18, wherein said modulated data signal

further is located inside a pair of passband frequencies in a communications

channel, and wherein the reference signal has a center frequency that is located

outside the pair of passband frequencies.

20. The method according to claim 19, wherein said reference signal has a center

frequency that is located inside a pair of passband frequencies of the

communications channel.

21. The method according to claim 15, wherein the processing means further

comprises at least one from the group comprising a digital signal processor and a

microprocessor.

22. The method according to claim 15, wherein the reference signal contains a

plurality of characterization signals, said characterization signals containing at

least one member selected from the group consisting of:

a voltage amplitude;

a frequency; and a phase angle.

23. The method according to claim 15, wherein the reference signal further

contains a plurality of calibration signals at least one member of which is selected

from the group consisting of:

a timing signal;

a voltage amplitude; and

a phase angle.

24. The method according to claim 3, Wherein said modulated signal is. encoded

using a phase shift modulation technique consisting of one or more from the

group consisting of: binary phase shift keying, quadrature phase shift keying,

and quadrature amplitude modulation.