WO2001001511A1 - Filtre ligne a retard - Google Patents

Filtre ligne a retard Download PDF

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Publication number
WO2001001511A1
WO2001001511A1 PCT/US2000/016257 US0016257W WO0101511A1 WO 2001001511 A1 WO2001001511 A1 WO 2001001511A1 US 0016257 W US0016257 W US 0016257W WO 0101511 A1 WO0101511 A1 WO 0101511A1
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WO
WIPO (PCT)
Prior art keywords
delay
filter
circuit
recited
resonators
Prior art date
Application number
PCT/US2000/016257
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English (en)
Other versions
WO2001001511A9 (fr
Inventor
Rafi Hershtig
Frederic J. Moore
Original Assignee
K & L Microwave
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by K & L Microwave filed Critical K & L Microwave
Priority to AU58724/00A priority Critical patent/AU5872400A/en
Priority to EP00944661A priority patent/EP1186071A4/fr
Priority to CN00810467.0A priority patent/CN1399805A/zh
Publication of WO2001001511A1 publication Critical patent/WO2001001511A1/fr
Publication of WO2001001511A9 publication Critical patent/WO2001001511A9/fr

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Classifications

    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P9/00—Delay lines of the waveguide type
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/20—Frequency-selective devices, e.g. filters
    • H01P1/201—Filters for transverse electromagnetic waves
    • H01P1/205—Comb or interdigital filters; Cascaded coaxial cavities
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/32—Modifications of amplifiers to reduce non-linear distortion
    • H03F1/3223—Modifications of amplifiers to reduce non-linear distortion using feed-forward
    • H03F1/3229—Modifications of amplifiers to reduce non-linear distortion using feed-forward using a loop for error extraction and another loop for error subtraction
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/198—A hybrid coupler being used as coupling circuit between stages of an amplifier circuit

Definitions

  • the present invention is directed generally to systems and methods for filtering and/or delaying transmission signals, and, more particularly, to systems and methods for providing integrated filters with a flat delay response.
  • a particular effective high power amplifier is a feed forward design which typically includes one or more delay lines along a main path and one or more delay lines along a secondary feed forward path.
  • Fig. 1 shows an exemplary feed forward high power amplifier 1.
  • the main path includes a hybrid splitter 2, a phase and amplitude adjustment 3, a main amplifier 4, an output coupler 5, a delay line 6, and an error injection coupler 7, feeding output 13.
  • the secondary path in the embodiment shown in Fig. 1 includes a delay line 8, a hybrid combiner 9, phase and amplitude adjustment circuits 10, 11, and an error amplifier 12.
  • the high power amplifier shown in Fig. 1 is suitable for use as a multi-carrier power amplifier.
  • the delay lines ideally have a uniform, temperature stable, and fixed amount of insertion delay and constant phase over a predetermined frequency range.
  • the delay lines 6 along the main path is implemented using multi- path delay equalization cavity filters.
  • the multi-path delay equalization cavity filters are configured to utilize cross coupling techniques available due to the greater isolation of cavity filters to minimize delay variations across the passband while handling very high levels of power, e.g., in excess of 1,000 watts for most designs, and over 2,000 watts on a custom basis, even for small cavity diameters. Examples of this type of delay line are available from the assignee of the present application as a standard product line. Filters of this type are typically large measuring from around 15 in 2 to more than 40 in 2 .
  • the conventional implementation of the delay lines is a coiled coaxial cable or micro-strip printed on a high dielectric material. Examples of devices falling within this category are shown in U.S. Patents 4,409,568, 5,252,934, and 4,218,664 and in conventional components such as Murata LDH33, LDH36, and LDH46 series delay lines.
  • conventional printed micro-strip delay lines are disadvantageous in that there is coupling between the various turns across the high dielectric material and a high insertion loss.
  • Coaxial cable type delay lines have little cross coupling but still are characterized by high insertion loss, and require a large amount of area to implement. Accordingly, a suitable secondary delay circuit is not now available conventionally.
  • Dr. S.B. Cohn proposed using a four port coupler or a three port circulator to achieve equalization of non-linear phase angle or time delay characteristics of other components. See, for example, U.S. Patent 3,277,403, herein incorporated by reference. U.S. Patents 4,197,514 and 4,988,962 cite examples of Dr. Cohn's earlier work as prior art in the background portions of the respective specifications.
  • U.S. Patents 4,197,514 and 4,988,962 cite examples of Dr. Cohn's earlier work as prior art in the background portions of the respective specifications.
  • none of these devices has been practical for low cost applications such as the secondary delay line in the multi-carrier power amplifiers discussed in connection with Fig. 1.
  • the present state of the art is to use a coaxial delay line in the secondary path of the feed forward power amplifier.
  • the coaxial delay line in the secondary path has many of the problems discussed above and thus is not satisfactory. Accordingly, the present invention seeks to take an altogether new approach to equalizing the delay across the passband using lumped discrete components. The present invention further seeks to provide improved high and low power delay lines in the main and secondary paths, respectively. Summary of the Invention
  • the goal of one or more aspects of the invention is to define a lumped element delay line which has a flat delay characteristic over the passband in a similar manner to how the cavity delay line has a relatively flat delay characteristic over the passband.
  • the present invention utilizes a new configuration for lumped elements previously thought not to be practical in a miniaturized structure.
  • the new architecture utilizes conventional theories to provide a highly improved structure for delay equalization using lumped components to form a flat delay response. This is particularly useful in the secondary feed forward path of the linear amplifier.
  • One or more aspects of the present invention may solve one or more of the above problems and/or provide improved techniques for implementing delay lines using lumped components.
  • a miniaturized delay line assembly is configured using lumped components comprising a miniaturized 3dB quadrature hybrid coupler (having resonators using lumped components) coupled to two ports and connected in series with a discrete implementation of a band pass filter.
  • the lumped components are tuned and mounted in a metal canister prior to shipping to a customer in order to provide the proper equalization.
  • Proper equalization is difficult to achieve using the tolerances normally associated with printed circuit boards assembled on a production basis.
  • the delay line may be provided as a pre-assembled part to various manufacturers such as for use in the secondary path of power amplifiers.
  • a miniaturized four port hybrid quadrature coupler is configured with discrete L-C resonators as the lumped components at two ports coupled in series with a filter, preferably a bandpass filter itself formed from lumped components.
  • the miniaturized 3dB quadrature hybrid coupler is mounted to the same circuit board as the filter with the resonators mounted in close proximity to the miniaturized 3dB quadrature hybrid coupler, e.g., within X A inch and more preferably within 1/4 inch.
  • the resonator components are discrete components mounted on the miniaturized 3dB quadrature hybrid coupler.
  • the resonator components are mounted on top of the miniaturized 3dB quadrature hybrid coupler.
  • the filter is implemented using miniature ceramic resonators such as the KEL-FIL ® class of ceramic filters.
  • multiply delay circuits are included along with a filter circuit.
  • the delay gain of the circuit is split between the delay circuits and the filter circuits such that the delay circuits comprise 60% or less of the total delay and preferably 50% or less of the total delay and most preferably about 40% of the total delay. Further aspects of the invention provide for:
  • a capacitive probe providing coupling between a pair of resonators having an odd number of resonators in a signal path therebetween.
  • the pair of resonators may preferably include a resonator disposed at a turn in the signal path.
  • a cavity delay arrangement utilizing mixed capacitive and inductive coupling between at least some of a plurality of resonators.
  • An all-pole Chebychev filter may be cascaded with the high power multicoupler.
  • a pre-tuned delay line module The module may be used as either a main delay line or a secondary delay line in a feed forward high power amplifier.
  • Another object of the present invention is to provisde a filter, for use with a delay compensating circuit in a delay line filter, formed from co-located capacitors, in which the capacitors may be formed as traces disposed on a circuit card and co-located to provide capacitive coupling to each other and ground, and proximity coupling wires are connected at one end to capacitive traces, and at the other end hang in mid-air above the next adjacent capacitive trace, so that the distance between each proximity coupling wire and the neighboring capacitive trace can be varied to control the capacitance coupling between neighboring capacitive traces.
  • Still another object of the present invention is to provide a filter that can be tuned by controlling the distances between the proximity coupling wires, and by squeezing or stretching the inductors
  • the shunt coils control the frequency of the filter, while the series of capacitive traces control the bandwidth.
  • Fig. 1 is a block diagram of one exemplary embodiment of an amplifier having a feed forward design.
  • Fig. 2 is a block diagram in accordance with an embodiment of the invention.
  • Fig. 3 is a partial pictorial/partial schematic view of the first exemplary embodiment of the invention.
  • Fig. 4 is a schematic diagram of the embodiment shown in Fig. 3.
  • Fig. 5 is a top view of a completed circuit board in accordance the first exemplary embodiment of the invention.
  • Fig 6. is a top view of a completed circuit board in accordance with a second exemplary embodiment of the invention.
  • Fig. 7 is a frequency versus delay chart showing the operation of embodiments of the invention.
  • Fig. 8 is a frequency versus delay chart of the operation of other embodiments of the invention.
  • Fig. 9 shows additional embodiments of the invention.
  • Fig. 10 shows a preferred packaging of embodiments of the invention.
  • Fig. 11 shows typical operating characteristics of embodiments of the invention.
  • Figs. 12A and 12B show a pictorial view of exemplary miniaturized 3dB quadrature hybrid coupler.
  • Fig. 13 is an exploded perspective view of an apparatus including a filter according to aspects of the present invention.
  • Figs. 14-18 are a plan views of filters having various exemplary embodiments of capacitive probes according to aspects of the present invention.
  • Fig. 19 is a side view from perspective A-A of the filter shown in Fig. 14.
  • Figs. 20-29 are photographs of exemplary filters having capacitive probes according to aspects of the present invention.
  • Fig. 30 is a functional block diagram of an exemplary embodiment of a main delay line according to aspects of the present invention.
  • Figs. 31 A-32A are top plan views of an exemplary embodiment of a stripline
  • Figs. 32B is a side plan view of the loaded multicoupler of Figs. 31A-32A.
  • Fig. 33 is a functional block diagram of an exemplary high power pre-tuned main delay line module according to aspects of the present invention.
  • Fig. 34 is a functional block diagram of an exemplary low power pre-tuned secondary delay line module according to aspects of the present invention.
  • Fig. 35 is partial pictorial/partial schematic view of an additional embodiment of the present invention.
  • Fig. 36 is a schematic view the embodiment shown in Fig. 35.
  • Fig. 37 is a perspective view of the embodiment shown in Fig. 35.
  • Fig. 38 is a side view of the embodiment shown in Fig. 35.
  • a delay circuit 28 may be variously configured to include a delay compensating circuit 29 coupled in series with a filter circuit 30.
  • Fig. 3 shows a partial pictorial and partial schematic view of a first exemplary embodiment of the invention.
  • Fig. 4 shows a schematic view of the first exemplary embodiment of the invention shown in Fig. 3.
  • Fig. 5 shows a top view of a circuit card containing the first exemplary embodiment of the invention.
  • Figs. 3-5 all utilize a miniaturized 3dB quadrature hybrid coupler which was previously thought not to be possible for use in delay equalization circuits with lumped elements since the small size of these couplers conventionally was thought to result in prohibitive cross-coupling problems when used with lumped elements.
  • Figs. 12 and 13 show pictorial representations of two exemplary miniaturize 3dB quadrature hybrid coupler. As shown in Figs. 12 and 13, the miniaturized 3dB quadrature hybrid coupler 40 typically has a volume of less than about 0.03 in 3 for high power devices (e.g., 100-200w) and about .01 in 3 for standard power devices (e.g., about 60 watts).
  • suitable miniaturize 3dB quadrature hybrid couplers are typically formed using hybrid microstrip technology, hybrid multilayer technology, or an integrated circuit technology such as chip monolithic implementations.
  • the miniaturized 3dB quadrature hybrid couplers are typically formed using hybrid microstrip technology, hybrid multi
  • 3dB quadrature hybrid couplers formed with hybrid multilayer technology and hybrid microstrip technology are formed using, for example, ceramic filled PTFE composites or other materials compatible with typical expansion characteristics of circuit boards formed from common substrate materials such as FR4, G-10, and polyamide.
  • a Xinger produced by Anaren, part number 1A1306, available at Anaren.com, Anaren Microwave, Inc., 6635 Kirkville Road, East Syracuse, NY 13057 may be utilized.
  • Other examples of miniaturized 3db quadrature hybrid couplers include models made by Mid- Atlantic RF Systems Inc., P.O. Box, 745, Forest Hill, MD 21050, EMC Technology LLC, 1971 Old Cuthbert Road, Cherry Hill, NJ 08034, and RF Power Components, Inc., 125 Wilbur
  • a miniaturized 3db quadrature hybrid coupler 40 has an input coupled to node PI, an output coupled to P4, and resonators coupled to P2 and P3.
  • the resonator 41 comprises micro inductor LI, capacitors Cl, C2, and coupling strap 45.
  • the resonator 46 comprises micro inductor L2, capacitors C3, C4, and coupling strap 50.
  • the resonators are preferably mounted close to the miniaturized 3db quadrature hybrid coupler 40, e.g., within one inch, and more preferably within Vi inch, and even more preferably within 1/4 inch.
  • the mounting of the resonators in close proximity to the miniaturized 3db quadrature hybrid coupler 40 decreases the variances caused by the coupling straps 45, 50, decreases cross coupling, and increases the ease of tuning of the resonators.
  • space may be minimized by locating the miniaturized 3db quadrature hybrid coupler 40 over or under the resonators. In this manner, the area required by the 3db quadrature hybrid coupler 40/resonators combination may be minimized while also minimizing the length of the straps 45, 50.
  • the resonators are disposed on top of and physically coupled to the miniaturized 3db quadrature hybrid couplers.
  • the miniaturized 3db quadrature hybrid couplers may be utilized to support and electrically isolate the resonators, increasing tuning ease and reducing cross coupling.
  • An example of the resonators disposed on and supported by the miniaturized 3db quadrature hybrid couplers is shown in Figs. 3 and 5.
  • the miniaturized 3db quadrature hybrid coupler may be customized to include some or all of the resonators 41, 46.
  • the resonators may be completely integrated into the miniaturized 3db quadrature hybrid coupler. This allows the entire delay compensating circuit to be manufactured on a production line for high volume, low cost applications.
  • one or more of the capacitors may be integrated into the miniaturized 3db quadrature hybrid coupler.
  • the miniaturized 3db quadrature hybrid coupler include capacitors Cl, C2, C3, and/or C4 formed integrally with the miniaturized 3db quadrature hybrid coupler.
  • the delay compensating circuit 29 may be tuned to a particular frequency, e.g., by manually tuning the resonators 41, 46.
  • the devices are suitable for manufacture in low volume, high mix environments.
  • the resonators are soldered together and to the top of the miniaturized 3db quadrature hybrid coupler 40. This arrangement provides greater reliability, reduced cross-coupling and smaller area. Further, the capacitors may be easily interchanged to provide maximum flexibility for custom applications (e.g, low volume, high mix applications).
  • Fig. 5 shows one exemplary embodiment of the miniaturized 3db quadrature hybrid coupler 40 mounted on a circuit card 50.
  • the drawing shown in Fig. 5 is three times actual size.
  • the exact overall packaged dimensions of the exemplary delay circuits 28 are specified in Fig. 11.
  • delay compensating circuit 29 may be coupled to filter 30 as described above in connection with Fig. 2.
  • the filter 30 may be disposed remote from the delay compensating circuit 29 or disposed proximate the delay compensating circuit 29. In many applications, it is desirable to mount the filter 30 close to the delay compensating circuit such as less than one inch, preferably less than Vi inch and more preferably less than 1/4 inch.
  • the filter 30 is mounted directly adjacent to the delay compensating circuit and co-located in the same housing (e.g., the housing shown in Fig. 10) and/or mounted on the same circuit board (e.g. circuit board 50 shown in Fig. 5).
  • Co- locating the delay compensating circuit 29 with the filter 30 has the advantage of protecting both circuits from cross coupling with other circuits, e.g., other elements in a feed forward amplifier. Additionally, the delay characteristics of both the filter 30 and the delay compensating circuit 29 can be closely controlled and matched. Additionally, an integrated assembly can be tested, tuned, and provided with proper quality assurance prior to being delivered.
  • the filter 30 may be formed of any suitable lumped element components, e.g., micro dielectric resonators such as ceramic elements, or combined resistive, inductive, and/or capacitive elements .
  • the filter 30 is formed from co-located capacitors formed from lines of the circuit board with discrete micro inductors soldered between the capacitors and ground. The micro resonators are preferred since, in the disclosed embodiment, cross coupling can be minimized. As shown in Figs.
  • the capacitors C5-C9 may be formed as square traces disposed on a circuit card and co- located to provide capacitive coupling to each other and ground. Schematically, the capacitors C5-C9 are shown in Fig. 4 as C5A, C5B, C6A, C6B, C7A, C7B, C8A, C8B, and C9A, C9B.
  • the filter 30 may also be configured to include one or more inductors such as micro-inductors L3, L4A, L4B, L5A, L5B, L6A, L6B, L7A, L7B, L8A, L8B, and L9 as shown in Figs. 3 and 5.
  • the micro inductors are shown schematically in Fig. 4 as inductors L3-L9.
  • the circuit of Figs. 3-5 may be configured to provide a flat delay response for a any suitable filter such as a band-pass filter.
  • a any suitable filter such as a band-pass filter.
  • the overall delay circuit 28 may exhibit, for example, only a minimal delay variation of 250 ps (.25 nanoseconds) over the operational range of the linear amplifier i.e., the over the preferred bandwidth. Accordingly, it is possible with delay equalization to provide a precise delay element using lumped components that exhibits a flat delay response over of the appropriate bandwidth. This was previously thought not to be possible at high frequencies using miniaturized components due to cross coupling between the components.
  • the miniaturized lumped elements may be configured to provide excellent delay elements for high frequency communications in the disclosed configurations. This results in a major cost savings over conventional configurations which heretofore have utilized either bulky and costly cavity filters, microstrips, and/or coaxial delay lines to accomplish the delay function. Circuits in accordance with the present invention are particularly useful in the feed forward path of a feed forward amplifier.
  • Figs. 3-5 Signals input into port 1 of the miniaturized 3db quadrature hybrid coupler are split into two signals into ports two and three with each signal having a -3dB loss.
  • the signal output at port 3 is - 180° out of phase and the signal output at port two is -90° out of phase. Accordingly, the difference between the port 2 signal and the port 3 signal is a 90° phase shift, with the port 2 signal being shifted -90° and the port 3 signal being shifted -180°.
  • Figs. 3-5 Signals input into port 1 of the miniaturized 3db quadrature hybrid coupler are split into two signals into ports two and three with each signal having a -3dB loss.
  • the signal output at port 3 is - 180° out of phase and the signal output at port two is -90° out of phase. Accordingly, the difference between the port 2 signal and the port 3 signal is a 90° phase shift, with the port 2 signal being shifted -90° and the port 3 signal being shifted
  • a signal entering the miniaturized 3db quadrature hybrid coupler at node 1 progresses to node 2 (90° out of phase), is reflected back via resonator 41, travels from node 2 to node 1 (180° out of phase), travels from node 1 to node 4 and is 270 degrees out of phase at node 4.
  • the signal entering at node 1 of the miniaturized 3db quadrature hybrid coupler also travels from node 1 to node 4 (90° out of phase), from node 4 to node 3 (180° out of phase), is reflected back via resonator 46, passes from node 3 to node 4 and is 270° out of phase at node 4.
  • the signals reflected back from both resonators 41 and 46 are both 270° out of phase at node 4 and additively combine to form the delay signal shown as curve B in Fig. 7. Since the signals are in-phase at node 4, the only losses through the miniaturized 3db quadrature hybrid coupler are the losses associated with the extremely low impedance of the coupler itself, e.g., on the order of 0.3 - 0.4 db or less per nanosecond.
  • the insertion loss may be further reduced by selection of appropriate coupling components to be 0.25db or less, 0.20 db or less, 0.15 db or less, and/or 0.13 db or less per nanosecond.
  • a reduction in the insertion loss has the advantage in that more power is output from the delay circuit 28. This is advantageous over other delay circuit implementations which have high insertion losses per volume ratios.
  • the power handling capabilities of the circuit may be variously configured by selection of appropriate components.
  • the delay circuit may be configured to handle, 25, 30, 35, 45, 65, 75, 100, or 200 or more watts.
  • the delay circuit 28 may be configured to handle 200 or more watts while still maintaining the compact size and low cost associated with embodiments of the present invention.
  • Embodiments of the invention employing high power, low insertion loss, and compact size open entire new applications for the invention such as for use in the main path of some low power feed forward amplifiers.
  • the delay through the coupling circuit as it enters and exits the resonators 41, 46 is such that it forms an inverse parabola as follows: the delay in the frequency at resonance enters the resonator, is reflected within the resonator several times and then is reflected back out. Signals entering resonators 41, 46 are delayed within the resonance frequency range of the resonators while signals outside of the resonate frequency range are immediately reflected. In other words, signals which lie outside of the resonance frequency range are bounced back immediately while signals within the resonance frequency range are circulated within the resonator. Thus, for frequencies increasingly further away from the center resonance frequency, the reflection is immediate and not delayed. Therefore, the delay path at resonance is much longer than for frequencies outside of the resonance.
  • the center frequency and shape of the inverted parabola may be modified by selectively choosing reactive elements and configurations of resonator 41, 46. In this manner, the characteristics and shape of curve B may be modified to closely compensate for the delay response of filter 30.
  • the delay compensating circuit 29 may be configured to provide a matched impedance of 50 ohms and a flat transmission loss over the bandwidth.
  • the miniaturized 3db quadrature hybrid coupler is desirable in many applications over coupling arrangements because it has been found to provide a superior flat amplitude response such that some frequencies are not preferred over others. This provides superior performance characteristics for the overall delay circuit 28.
  • curve C represents the output of an exemplary delay circuit 28 in accordance with embodiments of the invention.
  • the delay circuit 28 may exhibit small oscillations in the passband (e.g., on the order of about a quarter of a nanosecond of delay deviation) with the overall delay determined based on the value of the components selected. In the embodiment shown in Fig. 7, the overall delay is about 10ns.
  • Curve A indicates the unequalized normal filter response of a bandpass filter as, for example, the bandpass filter 30 shown in Figs.
  • the normal unequalized response of the bandpass filter may be compensated by a delay compensation circuit 29 (e.g., as shown in Figs. 3-5) to compensate for the delay variations in the normal bandpass filter response.
  • the delay compensation circuit may be coupled serially with the filter as discussed above.
  • the delay compensation circuit may have a frequency response which has a delay characteristic opposite the accompanying filter, e.g., an inverted parabola as shown in curve B in Fig. 7.
  • Fig. 8 shows a second exemplary plot of a delay response curve in accordance with embodiments of the present invention. As shown in Fig.
  • the delay compensating circuit 29 represents a contribution to the overall delay of between 4ns and approximately just slightly over 6 ns over the desired bandwidth, with an average delay of 5ns.
  • the filter circuit 30 represents a delay contribution of between 8 and just over 10 ns over the desired bandwidth, with an average delay of 9 ns.
  • Curve C represents the output of the overall delay circuit 28 showing an average delay of 14ns with a delay variation of less than .25 ns over the desired bandwidth.
  • the relative proportion of the overall delay (curve C) attributable to the delay circuit is such that the overall delay is split between the delay circuit and the filter circuit in a predetermined amount.
  • the minimum delay deviation may be achieved where the delay circuit comprises 60% or less of the overall delay, and preferably 50% or less of the total delay and most preferably about 40% of the total delay.
  • the delay circuit provides about 36% of the overall delay associated with the delay circuit 28.
  • Fig. 6 shows a second exemplary embodiment delay circuit 28 configured in a similar manner as the circuits of Figs. 3-5 with the exception that KEL FILTM resonator components (miniaturized ceramic/dielectric resonators) are utilized to form the filter and the resonators, but is otherwise similar to Fig. 3-5.
  • the ceramic resonator 61 is utilized to form resonator 46 while the ceramic resonator 62 is utilized to form resonator 41.
  • the ceramic resonators 61 , 62 may be mounted in close proximity to (e.g. adjacent to or on) the miniaturized 3db quadrature hybrid coupler as discussed above with respect to resonators 41 , 46.
  • the filter 30 may similarly be configured using miniaturized ceramic resonators 63-67 in place of filter stages L4C5, L5C6, L6C7, L7C8, L8C9 respectively.
  • the circuit board 50 may be used to form coupling board array 68 or in addition to coupling board array 68.
  • it may be desirable to utilize a standard KEL FILTM filter supplied by K&L Microwave, Incorporated, 2250 Northwood Drive, Salisbury MD 21801.
  • the general block diagram shown in Fig. 2 may be alternately configured to include any number of delay compensating circuits 29 and/or filter circuits 30.
  • one or more delay compensating circuits 29 e.g., first
  • the delay compensating circuit 29 may be coupled in series with one or more filter circuits 30 (e.g., first 30A, second 30B (not shown), and/or third 30B (not shown) filter circuits).
  • the delay compensating circuit 29 may be positioned either before or after the filter circuit 30.
  • the filter circuit may be positioned before, after, or between (as shown in Fig. 9) the delay compensating circuits.
  • the filters may be located before, after, and/or interspersed with the delay compensating circuits.
  • Fig. 10 is a pictorial view of the metal case encompassing the delay circuit 28 including both the delay compensating circuit 29 and the filter circuit 30.
  • the metal case is advantageous in that it isolates the delay circuit 28 from other extraneous electromagnetic interference.
  • the delay circuit 28 may be shipped as a tested, tuned, and mounting ready unit for operation in any application requiring a delay circuit 28 without the need for special electromagnetic engineering on the circuit card containing the delay circuit.
  • the overall ease of design and reliability of various feed forward amplifiers is increased because the lumped element delay circuit comprising the miniaturized 3db quadrature hybrid coupler may be pretested and assembled prior to mounting on a main board containing other elements such as the remaining components of a feed forward amplifier as shown in Fig. 1.
  • Delay circuits may be formed using various methods.
  • the delay circuit may be assembled by mounting on a circuit board at least a first miniaturized 3db quadrature hybrid coupler using known techniques such as a surface mount connection.
  • a plurality of first micro inductors and micro capacitors may also be coupled to the circuit board using known techniques, e.g. surface mounting.
  • the plurality of first micro inductors and micro capacitors may be configured to form one or more resonators, e.g., first and second resonators respectively coupled to at least two ports of the miniaturized 3db quadrature hybrid coupler.
  • a plurality of second micro inductors and capacitors may be configured to form a filter.
  • the filter may utilize either discrete micro capacitors and/or micro capacitors from directly from traces on the circuit board. Where capacitors are formed directly from traces on the circuit board, manufacturing costs may be substantially reduced while sizing and ease of tuning are substantially increased.
  • Surface mount couplers are conducive to conventional reflow methods such as (IR, Vapor Phase, Wave Solder, Manual Hot Air.).
  • delay circuits may be formed using various methods.
  • the delay circuit may be assembled by mounting on a circuit board at least a first miniaturized 3db quadrature hybrid coupler using known techniques such as a surface mount connection.
  • a plurality of first micro inductors and micro capacitors may also be coupled to the circuit board using known techniques, e.g. surface mounting.
  • the plurality of first micro inductors and micro capacitors may be configured to form one or more resonators, e.g., first and second resonators respectively coupled to at least two ports of the miniaturized 3db quadrature hybrid coupler.
  • a plurality of second micro inductors and capacitors may be configured to form a filter.
  • the filter may utilize either discrete micro capacitors and/or micro capacitors from directly from traces on the circuit board. Where capacitors are formed directly from traces on the circuit board, manufacturing costs may be substantially reduced while sizing and ease of tuning are substantially increased.
  • Fig. 11 provides a table showing exemplary performance characteristics for the delay circuit 28 including the delay compensating circuit 29 and the filter circuit 30.
  • Figs. 12A and 12B show exemplary embodiments of various commercially available miniaturized 3db quadrature hybrid couplers for use with embodiments of the present invention.
  • the high power amplifier 1 typically includes the main delay line 6.
  • These types of filters are well-known, such as is shown by U.S. Patent No. 5,739,733 to Cameron, issued April 14, 1998, and U.S. Patent No. 4,216,448 to Kasuga et al., issued August 5, 1980.
  • U.S. Patent No. 5,739,733 to Cameron issued April 14, 1998
  • U.S. Patent No. 4,216,448 to Kasuga et al. issued August 5, 1980.
  • a problem arises in that these filters do not have sufficiently flat delay response. Accordingly, an improved high power delay line filter is needed.
  • a first exemplary embodiment of the main delay line 6 is shown in Fig. 14 as delay arrangement 1400.
  • the delay arrangement 1400 may have a housing 1401 and/or a lid (not shown in Fig. 14) that may be screwed down and secured to the housing 1401.
  • the housing 1401 and/or the lid may preferably be partially or completely made of a conductive material such as aluminum.
  • the delay arrangement 1400 may further have a plurality of inductively- coupled resonators 1402a-1402h, each of which may be disposed in the housing 1401. Although eight resonators are shown, any number of resonators may be used, e.g., four to 16 resonators, including any increment between these ranges. In certain embodiments, more than 16 resonators may be utilized.
  • the resonators 1402a-1402h may be partially or completely made of a conductive material such as aluminum and/or silver.
  • each of the resonators 1402a-1402h may have an inner portion 1407a- 1407h that may be empty (i.e., filled with air) and/or solid. Where the inner portions 1407a-1407h are empty, then the resonators may be hollow cylinders.
  • the resonators 1402a-1402h may optionally be made of aluminum with hollow inner portions 1407a-1407h, and externally coated with silver (as shown in Figs. 20-29) to reduce loss within the delay arrangement 1400.
  • the delay arrangement 1400 may further have an input port 1405 coupled via a wire 1403 or other conductive material, and/or an output port 1404 coupled via a wire 1406 or other conductive material.
  • the resonators may be arranged within the housing to provide for a U-shaped path for the signal to be delayed, as is well-known in the art.
  • the path may start at the input port 1405, then to the resonator 1402a, then to the resonator 1402b, then to the resonator 1402c, then to the resonator
  • the path may be defined according to the relative distances between the resonators 1402a-1402h and/or walls and apertures (not shown in Fig. 14) appropriately placed between the resonators 1402a-1402h.
  • the delay arrangement 1400 would thereby provide a mix of both inductive and capacitive coupling.
  • inductive and capacitive coupling is well known for filters, such an arrangement has not heretofore been previously utilized in a cavity delay arrangement as discussed in this application.
  • the flatness of cavity delay arrangements may be substantially improved by capacitively coupling resonators that have an odd number of resonators in the main signal path there between.
  • the resonators 1402c and 1402e may be capacitively coupled with each other.
  • an odd number of resonators i.e., one resonator - the resonator 1402d
  • the delay arrangement may have the most flat delay response where the capacitive coupling is made with a resonator located at the turn of the U-shaped path (i.e., in the embodiment shown in Fig. 14, the resonators 1402d or 1402e are at the turn of the U-shaped path).
  • the capacitive coupling that would provide the flattest delay response would be between either the pair of resonators 1402c, 1402e, or the pair of resonators 1402d, 1402f.
  • a capacitive probe may be used.
  • the inductance and capacitance may be reversed.
  • the resonators may normally capacitively couple with each other, and the capacitive probe may instead be an inductive probe.
  • the cavities may be capacitively coupled together with the angular cross coupling being an inductive window.
  • each of the window apertures may be replaced by a capacitive probe and the capacitive probe may be replaced with a window aperture.
  • this arrangement is not the most preferred.
  • a capacitive probe may include one or more of the following in any combination: a wall 1450, a conductive bar 1451 disposed through the wall 1450, and/or conductive extensions 1452, 1453 disposed proximate to each of the resonators 1402e, 1402c, respectively. Because the extensions 1452, 1453 and the bar 1451 are each conductive, and because the extensions 1452, 1453 are disposed proximate to the resonators, the resonators are capacitively coupled with each other.
  • the wall 1454 may include a hole 1454 through which a screw (screw 1460 in Fig. 19) may be screwed to lock the bar 1451 into place relative to the wall 1450.
  • the capacitive probe may be insulated from the wall. For a better understanding of the embodiment shown in Fig. 14, please refer to Fig. 19 which shows the embodiment of Fig. 14 from view A-A.
  • the extensions 1452, 1453 may be soldered or welded to the bar 1451 (as shown in Fig. 29), or the bar and the extensions 1452, 1453 may be one continuous piece that is appropriately bent. In one exemplary embodiment, the bar 1451 and the extensions 1452, 1453 together form a "z" shape, as shown in Fig. 14.
  • the extensions 1452, 1453 may each be bent to adjust the amount of distance Dl, D2 between the respective extensions 1452, 1453 and the resonators 1402e, 1402c, thereby adjusting the amount of capacitive coupling that occurs between the resonators 1402e, 1402c and the extensions 1452, 1453.
  • Fig. 15 shows an alternative embodiment of the delay arrangement 1400.
  • the resonators 1402d and 1402f are capacitively coupled with each other via a capacitive probe.
  • Fig. 16 shows yet another embodiment where the resonators 1402b and 1402f are capacitively coupled with each other via a capacitive probe.
  • Fig. 17 shows the delay arrangement 1400 having an yet another alternative embodiment of a capacitive probe.
  • the capacitive probe may include an insulated wall section 1450 and/or a conductive bar 1455 bent into a "z" shape (a "z- bar”) or other appropriate capacitive coupling arrangement.
  • the z-bar 1455 may be a single piece or a plurality of sections conductively connected together.
  • Fig. 18. shows the delay arrangement 1400 having yet another alternative embodiment of a capacitive probe.
  • a capacitive probe 1456 is shown being an elongated structure that capacitively couples the resonators 1402c and 1402e with each other.
  • the main delay line 6 may be implemented as one or more delay equalization filters, such as delay arrangement 1300.
  • the delay arrangement 1300 may include a housing 1301 and/or a lid 1303 that may be screwed down and secured to the housing 1301.
  • the housing 1301 and/or the lid 1303 may preferably be partially or completely made of a conductive material such as aluminum.
  • the delay arrangement 1300 may further have a plurality of inductively-coupled resonators 1302a-1302h, each of which may be disposed in the housing 1301.
  • the resonators 1302a-1302h may be partially or completely made of a conductive material such as aluminum and/or silver.
  • a plurality of tuning screws 1304a-1304d may be disposed in the lid 1303, and conductively coupled with the lid 1303, for adjusting capacitive coupling between the resonators 1302a-1302h and the lid 1303.
  • Figs. 20-29 are photographs showing various views of exemplary embodiments of delay arrangements having capacitive probes according to aspects of the present invention.
  • the main delay line 6 may be embodied as a hybrid quadrature 90 degree phase shift coupler multicoupler loaded by two resonators (loaded multicoupler) 3001 cascaded with an all-pole Chebychev filter 3002.
  • the all-pole Chebychev filter 3002 may preferably be a cavity filter having zeros at DC and/or infinity.
  • the loaded multicoupler 3001 may be electrically configured in the same way as the delay compensating circuit 29, except that the loaded multicoupler 3001 may preferably be built to be more suitable for high power.
  • the loaded multicoupler 3001 is depicted functionally as a multicoupler loaded with resonators. The first resonator in Fig.
  • the first and second resonators may be variously configured. In exemplary embodiments, the first and second resonators may have a resonance value which is about the same, substantially the same, and/or identical. In the most preferred embodiments, the first and second resonators are identical.
  • the loaded multicoupler 3001 may include two stripline circuits 3101, 3102 overlaid with each other such that the "A" labeled sides match up and the "B" labeled sides match up.
  • the stripline circuit 3101 may have a dielectric substrate 3121 with conductive strips 3150, 3151 disposed thereon as shown, and the stripline circuit 3102 may have a dielectric substrate 3122 with conductive strips 3152, 3153 disposed thereon as shown.
  • the stripline circuits 3101, 3102 are properly overlaid, the conductive strips 3150, 3151, 3152,
  • the dielectric substrates may be back-to- back, and the sides of the stripline circuits 3101, 3102 having the conductive strips 3150, 3151, 3152, 3153 may be facing outward.
  • the two substrates prevent the conductive strips 3150, 3151 from directly contacting the conductive strips 3152,
  • the overlaid stripline circuits 3101, 3102 may be housed as suspended substrate boards in a housing 3201.
  • the housing 3201 may include an input port 3203 conductively coupled to the conductive strip 3152 and an output port 3204 conductively coupled to the conductive strip 3151.
  • the housing 3201 may include two halves 3201a, 3201b that may be screwed together.
  • Figs. 32C and 32D may use a multi-layer configuration employing, for example, low temperature co- fired ceramic.
  • the multi-layer configuration may use a one to 18 layer stack. In exemplary embodiments, a 15-18 layer stack may be utilized as shown in Fig. 32C.
  • the individual layers may comprise one or more of an interdigital configuration, hair pin configuration, combline configuration, or edge coupled configuration.
  • Fig. 32D shown a interdigital configuration of a bandpass filter coupled to the output of a multicoupler.
  • the loaded multicouplers shown in Figs. 32 C and 32D are available from Dielectric Lab (DLI) (www.dilabs.com) and American Technical Ceramic (ATC) (www.atceramics.com).
  • a consistent problem with the construction of a feed forward high power amplifier is the tuning of the delay lines coupled to one or more couplers. Conventional delay lines in the low power feed forward path could not be tuned.
  • the present invention allows for pre-tuned modules to be sold for both the high-power and low-power delay arrangements.
  • the main delay line 6 and/or the secondary delay line 8 may each be manufactured as pre-tuned modules ready for insertion into an amplifier.
  • the problem with present delay lines is that they must be tuned, which is a painstaking process. It is preferable to have the delay line modules available already tuned so that manufacturers of amplifiers may buy pre-tuned delay modules either custom-tuned or off-the-shelf and insert the pre-tuned delay modules into their amplifier products.
  • the main delay line 6 may be embodied as a pre-tuned high power delay-line module 3301, and may include a first high power coupler 3302, cascaded with a high power delay line filter 3303, and/or further cascaded with a second high power coupler 3304.
  • the couplers 3302 and/or 3304 may be standard couplers or multicouplers.
  • the high power delay line filter 3303 may be any of the high power delay line filters discussed herein.
  • the secondary delay line 8 may be embodied as a pre-tuned low power delay- line module 3401, and may include a first low power coupler 3402, cascaded with a low power delay line filter 3403, and/or further cascaded with a second low power coupler 3404.
  • the couplers 3402 and/or 3404 may be standard couplers or they may be multicouplers.
  • an alternate embodiment of the filter 30 is shown formed from once again from any suitable lumped element components, and in the embodiment shown in Figs. 35-38, the filter 30 is formed from co-located capacitors formed from lines of the circuit board. Unlike the embodiment shown in Figs. 3-5, the embodiment shown in Figs. 35-38 requires only one capacitor and inductor per shunt. The second discrete micro-inudctor between the capacitor and ground is not needed. As a consequence, the filter 30 is easier to tune due to the reduction in the number of parts.
  • the capacitors C5-C9 may be formed as square traces disposed on a circuit card 3500 and co-located to provide capacitive coupling to each other and ground.
  • the capacitors C5-C9 are shown in Fig. 36 as C5A, C5B, C6A, C6B, C7A, C7B, C8A, C8B, and C9A, C9B.
  • the filter 30 is also be configured to include one or more inductors such as micro-inductors L3, L4, L5, L6, L7, L8, and L9 as shown in Figs. 35-38.
  • the proximity coupling wires C 56 , C 67 , and C 7g are connected at one end to capacitive traces Cl, C2, and C3, respectively.
  • the other ends of proximity coupling wires C 56 , C 67 , and C 7g hang in mid-air above the next adjacent capacitive trace, C2, C3, and C4, respectively.
  • the distance between each proximity coupling wire and the neighboring capacitive trace can be varied to control the capacitance coupling between neighboring capacitive traces.
  • the filter 30 is thus tuned by controlling the distances between the proximity coupling wires, and by squeezing or stretching the inductors (shunt coils) L3, L4, L5, L6, L7, L8, and L9.
  • the shunt coils control the frequency of the filter 30, while the series of capacitive traces control the bandwidth.
  • Figs. 35, 37 and 38 show three capacitive traces Cl, C2 and C3.
  • Fig. 36 shows a schematic diagram with five capacitive traces Cl, C2, C3, C4, and C5, with an additional coupling wire, represented as capacitor C g9 .
  • the number of capacitors in the filter 30 is determined by the delay that is desired.
  • the filter 30 will have between 3 and 7 shunts, however, the present invention is not intended to be limited to that range and greater and fewer shunts may be employed. It Is expected that once the filter has been assembled and tuned, so that all of the proximity coupling wires are located the desire distance from the neighboring capacitive trace, that the unit will be encased in a housing to be employed as a pre- tuned unit as described previously with regard to other embodiments.

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  • Physics & Mathematics (AREA)
  • Nonlinear Science (AREA)
  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Electromagnetism (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)

Abstract

La présente invention concerne un circuit à retard (28) conçu pour contenir un circuit de compensation de retard (29) couplé en série avec un circuit à filtre intégré (30). Le circuit de compensation de retard (29) comprend un coupleur (40) hybride de quadrature de 3db miniaturisé présentant un première, une deuxième, une troisième et une quatrième borne, et un premier résonateur (41) et un deuxième résonateur (46) à constantes localisées, couplé chacun aux deuxième et troisième bornes; le circuit filtrant (30) comprend une pluralité de constantes localisées couplées en série avec le circuit de compensation de retard (29) à la première et/ou à la deuxième borne.
PCT/US2000/016257 1999-06-14 2000-06-14 Filtre ligne a retard WO2001001511A1 (fr)

Priority Applications (3)

Application Number Priority Date Filing Date Title
AU58724/00A AU5872400A (en) 1999-06-14 2000-06-14 Delay line filter
EP00944661A EP1186071A4 (fr) 1999-06-14 2000-06-14 Filtre ligne a retard
CN00810467.0A CN1399805A (zh) 2000-06-14 2000-06-14 延迟线滤波器

Applications Claiming Priority (6)

Application Number Priority Date Filing Date Title
US13865199P 1999-06-14 1999-06-14
US60/138,651 1999-06-14
US14869099P 1999-08-16 1999-08-16
US60/148,690 1999-08-16
US19524600P 2000-04-07 2000-04-07
US60/195,246 2000-04-07

Publications (2)

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WO2001001511A1 true WO2001001511A1 (fr) 2001-01-04
WO2001001511A9 WO2001001511A9 (fr) 2002-06-13

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PCT/US2000/016257 WO2001001511A1 (fr) 1999-06-14 2000-06-14 Filtre ligne a retard

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AU (1) AU5872400A (fr)
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Cited By (2)

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EP1343218A2 (fr) * 2002-03-07 2003-09-10 Murata Manufacturing Co., Ltd. Egalisateur intrabande de temps de propagation de groupe et amplificateur de compensation de distortion
CN104538711A (zh) * 2014-09-13 2015-04-22 南京理工大学 微型微波毫米波i/q正交滤波器

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DE2242535A1 (de) * 1972-08-30 1974-03-07 Siemens Ag Filter fuer sehr kurze elektromagnetische wellen
US4197514A (en) * 1976-09-21 1980-04-08 Nippon Electric Co., Ltd. Microwave delay equalizer comprising a pair of distributed-constant_ elements as a directional coupler
US4472725A (en) * 1982-02-01 1984-09-18 Century Iii Electronics Inc. LC Delay line for feedforward amplifier
US4890078A (en) * 1988-04-12 1989-12-26 Phase Devices Limited Diplexer
US5146191A (en) * 1990-06-13 1992-09-08 Murata Manufacturing Co., Ltd. Delay line device and a method for producing the same
US5252934A (en) * 1992-08-25 1993-10-12 At&T Bell Laboratories Microwave delay assembly
US5481231A (en) * 1994-06-21 1996-01-02 Motorola, Inc. Lumped element four port coupler
US5748058A (en) * 1995-02-03 1998-05-05 Teledyne Industries, Inc. Cross coupled bandpass filter
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Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1343218A2 (fr) * 2002-03-07 2003-09-10 Murata Manufacturing Co., Ltd. Egalisateur intrabande de temps de propagation de groupe et amplificateur de compensation de distortion
EP1343218A3 (fr) * 2002-03-07 2004-03-31 Murata Manufacturing Co., Ltd. Egalisateur intrabande de temps de propagation de groupe et amplificateur de compensation de distortion
US6958663B2 (en) 2002-03-07 2005-10-25 Murata Manufacturing Co., Ltd. In-band group delay equalizer and distortion compensation amplifier
CN104538711A (zh) * 2014-09-13 2015-04-22 南京理工大学 微型微波毫米波i/q正交滤波器

Also Published As

Publication number Publication date
EP1186071A4 (fr) 2002-09-04
EP1186071A1 (fr) 2002-03-13
AU5872400A (en) 2001-01-31
WO2001001511A9 (fr) 2002-06-13

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