WO2000062415A1 - Means and method for increasing performance of interference-suppression based receivers - Google Patents

Abstract

In a bidirectional data chanel between a central station and a multiple of remote stations, a method (fig. 6) for equalizing interference over a synchronized packet or frame based baseband transmission system wherein the crosstalk on the system is cyclostationary or periodic with a period equal to a symbol interval, the method (fig. 6) comprising the steps of synchronizing the trannsmitters and receivers using the uncorrelated transmit signals; generating the cyclostationary NEXT and FEXT interference (fig. 6) along with ISI using the uncorrelated symbols at the synchronized transmitters at one or more remote stations and the centrally station; using cascaded fractionally spaced linear equalizer (FSLE) and decision feedback equalizer (DFE) for both interference supression and equalization to minimize excess bandwidth at central receivers at the central station; (fig. 6) increasing the receiver's FSLE filter taps (NT) to maximize signal to noise ratio; (fig. 6) combining FSLE/DFE and proper phase sampling adjustment, enabling use of the spectral correlation properties peculiar to the modified signals (fig. 6).

Description

MEANS AND METHOD FOR INCREASING PERFORMANCE OF INTERFERENCE-SUPPRESSION BASED RECEIVERS

I. INTRODUCTION

A. RELATED APPLICATIONS

The subject matter of this application is related to the subject matter of the commonly owned applications and Serial Number 60/109,340, attorney docket number 3697, titled "Means and Method of Precursor ISI Cancellation" filed on November 20, 1998, by Francois Trans; Serial No. 60/085,605, attorney docket number 3432 titled "System And Method For Scalable Com2000 Gigabit Ethernet CAT5 Physical Layer (GPHY4)" May 15, 1998, by

Francois Trans; and Serial Number 60/054,406, attorney docket number 2960, titled "Method And Means For A Synchronous Network Communication System" filed on July 31, 1997, by Francois Trans; and Serial Number 60/054,415 attorney docket number 2961, titled "Method And Means For A Universal Information Technological System" filed on July 31, 1997, by Francois Trans; and Serial Number 60/089,526, attorney docket number 3480, titled

"Simulation Findings For The Scalable Com2000 Gigabit Ethernet CAT5 Physical Layer (GPHY4)" filed on June 15, 1998, by Francois Trans; and Serial Number 09/127,383, attorney docket number 3476, titled "Means and Method for a Synchronous Network Communications System" filed on July 31, 1998 by Francois Trans; and Serial Number 60/104,316, attorney docket number 3659, titled "Means And Method For Increasing Performance Of Interference- Suppression Based FSLE/DFE xDSL Receiver Over Pots Cables" filed on October 13, 1998 by Francois Trans; Serial Number 60/129,314, attorney docket number 3926, titled "Means And Method Of High-Speed Data Transmission Over Existing Mil-STD-1553" filed on April 14, 1999, by Francois Trans; and Serial Number 09/417,528, attorney docket number 4511, titled "Means And Method For Increasing Performance Of Interference-Suppression Based Receivers" filed on October 13, 1999 by Francois Trans, all of which are incorporated by reference.

B. RELATED ART

This invention is related to maximizing the transmission of data over communication networks. More specifically, this invention is related to suppression of cross-talk and interference over communications channels.

C. BACKGROUND Interference (echo and crosstalk) is one of the major performance-limiting impairments on UTP cables. In this application, various receiver structures suitable to the transmission of Gigabit Ethernet over 4 pairs of UTP cables are described. Furthermore, the performance of the invention is disclosed through use of a receiver structure that uses a cascade of FSLE and DFE for interference suppression. Some of these results are based on the assumption that the interference can be cyclostationary, i.e., interference statistics are periodic with a period equal to a symbol interval. This is true as long as all transmitter symbol timing clocks are synchronized in frequency. These assumption are meant to explain the results provided and should not be itnerpreted as a limitation on the scope of the claims.

Suppression of cyclostationary interference by linear equalizers has been considered previously. Prior studies have identified that linear processing of cyclostationary interfering signals can exploit spectral correlation properties peculiar to these signals. In Section II, we describe the channel characteristics and model the communication channels. In Section III, we present embodiments of the invention including the different receiver structures suitable to the transmission of Gigabit Ethernet over 4 pairs of cat-5 UTP cables. In Section IV, we present the analytical model for the cascaded FSLE/DFE receiver structure using interference suppression approach and its performance analysis. Numerical results on the SNR and numbers of taps required for the FSLE and DFE are discussed in Section V. D. SUMMARY OF THE INVENTION

The present invention provides a method for equalizing interference over a synchronized packet or frame based baseband transmission system wherein the crosstalk on the system is cyclostationary or periodic with a period equal to a symbol interval. The method comprises the steps of : synchronizing the transmitters and receivers using the uncorrelated transmit signals; generating the cyclostationary NEXT and FEXT interference along with ISI using the uncorrelated symbols at the synchronized transmitters at one or more remote stations and the centrally station ;using cascaded Fractionally Spaced Linear Equalizer (FSLE) and Decision Feed back Equalizer (DFE) for both interference suppression and equalization to minimize excess bandwidth at central receivers at the central station; increasing the receiver's FSLE filter taps (NT) to maximize Signal to noise ration; combining FSLE/DFE and proper phase sampling adjustments, enabling use of the spectral correlation properties peculiar to the modified signals. Brief Description of the Drawings Figure 1 illustrates the performance degradation for transceivers operating over UTP wiring caused by propagation loss and crosstalk generated between pairs;

Figure 2a illustrates the worst-case insertion loss associated with data transmitted over a 100 meter, Cat-5 cable; Figure 2b illustrates the cable impulse response over a 100 meter, Cat-5 cable;

Figure 3 shows the plot of the measured return loss and the return loss limit ;

Figure 4a illustrates the worst-case return loss between pairs of Cat-5 cable pairs;

Figure 4b illustrates the worst-case NEXT loss between pairs of Cat-5 cable pairs;

Figure 5 illustrates the worst case FEXT loss between pairs of Cat-5 cable pairs; Figure 6 illustrates the channel model including the effects of partial response, DAC and hybrid filtering in the transmitter, the main and coupling channel characteristics, and the filtering in the receiver front-end;

Figure 7 illustrates the received pulse responses including both the mail pulse response and the

NEXT pulse responses; Figure 8 illustrates the Echo pulse response using the present method;

Figure 9a illustrates the NEXT pulse response achieved using the present method;

Figure 9b illustrates the FEXT pulse response using the present method;

Figure 10a provides a sample receiver structure using Interference cancellers prior to equalisation; Figure 10b provides a sample receiver structure using Interference cancellers after equalisation;

Figure 10c provides a sample receiver structure using cascaded FSLE/DFE for both interference suppression and equalisation;

Figure lOd illustrates a C17 Cable Impulse Response Graph; Figure 20 provides a System Model illustrating the described system;

Figure 30: Existing 1553 Transceiver Structure

Figure 40: Before and After Fractional Space Equalizer for Precursor ISI

Figure 50A : Intersymbol Interference (ISI) At High Transmission Rate Over MIL-C17 Cable

Figure 50B : Decision Feedback Equalization To Remove Postcursor ISI Figure 60: Proposed High Level MDI-1553+ Transceiver Structure High Level

Figure 70: Proposed Detailed MDI-1553+ Transceiver Structure using DPIC

Detailed Description of the Invention The two major causes of performance degradation for transceivers operating over UTP wiring are propagation loss and crosstalk generated between pairs, as shown in Figure 1. Each UTP supports a 250Mb/s full-duplex channel using a 5-level 125Mbaud transmission scheme. Consider the transmission on pair#l. With respect to the Receiver #1L on the left, its wanted signal is sent by the Transmitter #1R on the right. The transmitter #1L sends a signal to the Receiver #1R, but also generates spurious signal (called echo) to its own Receiver#lL. The interference signals generated by Transmitters 2L-4L appear at the input of the Receiver #1L are called near-end crosstalk (NEXT) interferers, NEXT_21 to NEXT 41. The interference signals generated by Transmitters 2R-4R on the right appear at the input of the Receiver #1L are called far-end crosstalk (FEXT) interferers, FEXT_21 to FEXT_41.

A. Propagation Loss:

The models for the propagation loss of a loop that are presented in this section are valid for frequencies that are larger than about 500 kHz. The signals considered in this paper have a very small amount of energy below this frequency. Thus, for simplicity, we will assume that the propagation loss models discussed here are valid at all frequencies.

The transfer function H(d, f) of a perfectly terminated loop with length d can be written as follows:

H d,f) = e-^d^ = e-^da e-MW where γ (f) is the propagation constant, a (f) is the attenuation constant, and β (f) is the phase constant. The quantity that is usually specified in practice is the propagation loss for a given cable length (e.g., d = 100 meters). The propagation loss (or insertion loss) limit Lp (f) for category 5 (cat-5) 100m cable is a positive quantity expressed in dB

L_P(f) = - 20 1og|H(J = 100m, )|

²⁰ -a (f) - 2.1/°-⁵²⁹ ₊0.4//

In 10 B. ECHO Loss: The Echo loss is indicated by the return loss. Figure 3 shows the plot of the measured return loss and the return loss limit which is 15dB for frequency from 1 to 20MΗz and 15- 101og(_720) for frequency from 20 to 100MHz. C. NEXT Loss:

The wavy curves in Fig. 4 give the measured pair-to-pair NEXT loss characteristics for three different combinations of twisted pairs in 100m cat-5 cables. The existence of the minima (small loss) and maxima (large loss) in these curves is due to the fact that the frequencies considered here correspond to wavelengths that are in the same length range as the distance between points of unbalance in the NEXT coupling path. Notice that the minima and maxima usually occur at different frequencies for the three pair combinations. Notice also that the NEXT loss corresponding to the minima decreases with increasing frequency and tends to follow the smooth dotted curve on the bottom in the figure, which is defined as the worst-case pair-to-pair NEXT loss (or NEXT loss limit) as a function of frequency. The worst-case TIA/EIA-568-A NEXT loss model shown in Figure 4 is 27.1-16.81og(f/100) in dB.

D. Channel Modeling Figure 6 shows the channel model including the effects of partial response, DAC and hybrid filtering in the transmitter, the main and coupling channel characteristics, and the filtering in the receiver front-end. The DAC and hybrid filtering is represented by the cascade of two identical first-order Butterworth sections with a corner frequency of 180MHz. This introduces a 4ns rise/fall time. The receiver front-end is modelled as a fifth-order Butterworth filter with a corner frequency of 80MHz. The main channel, echo coupling and NEXT coupling channels are represented by C(ω), E(ω), N₂(ω), N₃(ω), and N₄(ω), respectively. The models for the FEXT's are similar to those of the NEXT's except the coupling channels will be F,(ω), F₃(ω), and F₄(ω), instead of N₂(ω), N₃(ω), and N₄(ω). The pulse responses of the main, echo, NEXT's and FEXT's at the input of the RECEIVER shown in Figure 6 are shown in Figures 7, 8, and 9, respectively.

III. INTERFERENCE CANCELLATION AND SUPPRESSION (EQUALIZATION)

Reliable duplex operation at 250Mb/s over two pairs of a CAT-5 UTP cable requires the usage of some kind of technique to combat interference including echo, NEXT and FEXT. Since the FEXT has a small contribution in interference level, we can neglect FEXT's and focus on the echo and NEXT's. Since the transmission on all four pairs uses the same Tx clock, the crosstalk can be shown to be cyclostationary, i.e., crosstalk statistics are periodic with period equal to a symbol interval. The two techniques that are presently being used are NEXT cancellation and NEXT equalization (or suppression). Figures lOa-c show three general receiver structures. The structures shown in Figures 10a and b are based on interference cancellation. A NEXT canceller synthesizes, in an adaptive fashion, a replica of the NEXT interferer. The interferer is then cancelled out by subtracting the output of the canceller from the signal appearing at the receiver. A NEXT canceller has the same principle of operation as an echo canceller, and all the familiar structures used for echo cancellers can also be used for NEXT cancellers. The cancellers needs to have access to the local transmitters from which they get their input signals. Typically, this input signal is the stream of symbols generated by the transmitter's encoder. In Fig. 10a the output signal of the canceler is subtracted from the received signal immediately after the AID. With such an approach, the canceler generates outputs at the same rate as the sampling rate of the A/D. An alternative embodiment is to make the subtraction at the input of the sheer as shown in Fig. b. In this case, the outputs of the canceller need only be generated at the symbol rate.

The FFE (feed-forward equalizer) in Figures 10a and b can be a symbol-spaced (SS) or fractionally spaced (FS) FFE or an analog equalizer. It is used to equalize the precursor ISI. The DFE is used to remove the post cursor ISI. Note that the performance of the DFE is also dependent on the reliability of the symbols detected by the slicer and influenced by the error propagation. For this, one may replace the simple slicer by a sequence detector (such as Viterbi decoder) for a better performance. In that case, the long processing delay of the decoder can be an issue.

With NEXT equalisation shown in Figure 10c, no attempt is made to cancel out the NEXT interferer and there is no need to have access to the transmitter generating the interferer. Rather, the interfering NEXT signals are equalised at the receiver in such a way that it passes through zero at all the sampling instants of the slicer. In Fig. 10b, the FSFFE or DFE used by the receiver equalises the desired signal received from the other end of the cable and the echo and NEXT interferers, but in a different fashion. Let f(t) be the impulse response of the in- phase component of the desired signal and r(t) be the impulse response of the in-phase component of the interferer. The conditions for perfectly equalising the desired signal and interferer in the desired fashion can then be written as f(kT) = δ(k) and r(kT) = 0 where k is an integer, T is the symbol period, and δ(.) is the Dirac delta function, i.e., δ(0) = 1 and δ(k) = 0 for k ≠ 0. These conditions also guarantee that the impulse responses of the quadrature component of the far signal and NEXT interferer satisfy f(kT) = r'(kT = 0 for all k. Interference equalisation is optimally feasible if the transceiver uses a large excess bandwidth. Specifically, it can be shown that, with one cyclostationary interferer, these conditions can be satisfied if the transmitter uses an excess bandwidth of at least 100%. Heuristically, the need for such a large excess bandwidth can be explained as follows. With 0% excess bandwidth, an adaptive equaliser has just enough degrees of freedom to perfectly equalise one signal, but cannot do anything else. In order to equalise two signals, the number of degrees of freedom available to the equaliser has to be doubled with respect to what is required for one signal. This is achieved by doubling the bandwidth of the transmitted signal, which results in an excess bandwidth of 100%. Theoretically, it is possible to perfectly equalise the two interferers, but this requires the usage of an excess bandwidth of 200%, and, in general, perfect equalisation of n interferers requires an excess bandwidth of n times 100%. For most applications of bandwidth-efficient digital transmission schemes, the usage of excess bandwidth would be considered as a waste of bandwidth.

IV. ANALYSIS OF THE RECEIVER USING CASCADED FSLE/DFE:

Figure 6 shows the overall system that is used to study the performance of the receiver structure using a FSLE cascaded with a DFE in the presence of interference (echo and NEXT's), ISI, and additive white noise (AWN). The AWN has power spectral density of NJ2. The waveform received by the receiver is:

where ♦ the first term of r(t) is the desired signal (i.e., sequence to be detected), while the second term represent N interferers, and n(t) is the AWN at the input of the FFE.

♦ 0<Ω, <T is the 1th interferer's delay. Φ₀(t) is the overall end-to-end pulse response (e.g., Figure 7), and Φ,(t) is the pair-to-pair pulse response of the 1th interferer (e.g., Figures 8-9).

♦ a_k is the transmitted symbol, b_kI is the interfering symbol. It is assumed that: 1) all a_k and b_H are uncorrelated;

2) E(_ak) = 0; E(a_k ²) = l;

3) E(b_kl) = 0; E(b_k] ²) = l; and

4) there are no decision errors (i.e., practically negligible). The input to the slicer (in Figure 10c) is

N Nf

Y_n = 2 >v_mr(«r - (mE> + ^)) + 2 , m^an-m-j. (2) m=0 m=\ where θ is the sampling phase representing time shift in a symbol period, D=T/M is the delay element used in the FFΕ (M=l for symbol-spaced FFΕ and M>1 for fractionally spaced FFΕ). w_m' s and ξ„'s are the tap settings of the FFΕ and DFΕ, respectively, and p is the delay in the receiver's decision relative to the receiver's input. The FFΕ and DFΕ coefficients are optimized to minimize the mean squared error (MSΕ), where the error is:

e = Y - a (3)

and includes interference, ISI, and AWN. Equation (2) for the output of the slicer can be expressed as:

Y„ = U^T _n , where T is the transpose operator

U^τ = [w₀ w, ... w_Nw f, f₂ ... f_Nf]

with

R_n ^T = [r(nT - θ)r(nT -D -θ) -- r(nT - N_wD

and

The MSE we need to minimize is:

MSE = [E[(Y_n - a_n._p)²] = U^TAU - 2U^TV + 1 (5) where A = E [X_X ], and V = E [X_na_n._p].

Setting the derivative of the MSE to zero, we find the optimum weight of the forward and feedback coefficients, which is: U_opt = A^"!V (6) and the corresponding minimum MSE is

MMSE = 1 - VA-'V = 1 - V^TU_opt (7) where V and A are obtained by taking expectations, using (4):

= E[r(nT -θ) --- r(nT - N_wD -θ) a„__x__p a_n__p ^■■^■ a„__Nf__p a_n__p This matrix A could be written as

where A, = E(R_₁R_n ^T) , A2 = E(R-a^τ _n._ι._p) , and I is the identity matrix.

Under our assumptions, we find that A₂ = [x_y], where

x_u = Φ₀((p + j)T -iD -θ), 0 ≤ i ≤ N_w, and \ < j ≤ N_f (11) and A, = [q(ij)], where

q(i,j) = ∑ Φ₀(kT - iD)Φ₀(kT -jD) + ∑ ∑Φ_l(kT-iD -Ω_l) k=-∞ I k=-∞

R(t) is the autocorrelation function oFrhe- ower spectral density of AWN at the output of the receiver filter. Note that for stationary interference with power spectrum equal to that of the cyclostationary interference, the results are the same except the q(i, j) term becomes:

q(i,j) =

+ R((i -j)D). (12b)

The above model and analysis are used with pulse shapes shown in Figures 7-9 to compute the SNR at the slicer input for different values of taps and D.

We assume a small contribution of AWGN, i.e. in the absence of NEXT, the receiver signal- to-noise ratio is about 60 dB The choice of a low AWN level ensures that crosstalk is the dominant additive impairment. The performance measure used in the evaluations of this and the next section is output SNR, defined as SNR = 10 * log ₁₀ (1 /MSE) where the mean squared error (MSE) expression is shown by Equation (7) above. Thus, the transmitted data symbol levels are normalized so as to have unit variance.

With the above channel models, SNR, maximized with respect to receiver sampling phase, has been evaluated for two interferers for various numbers of DFE feedforward and feedback tap coefficients. Preliminary results on the SNR performance as a function of receiver sampling phase are summarized in Figures 11 and 12. In these figures,

♦ NT is the span of the FFE in terms of the number of symbol intervals, and D is the delay 5 element used in the FFE. Hence, the number of taps of the FFE is given by the product of

(NT)(T/D); and

♦ NF is the number of DFE taps.

As shown in Figure 11, for the case of symbol-spaced FFE (i.e., D=T), the SNR is quite sensitive to the sampling phase with a variation of 6dB. With a proper choice of sampling 0 phase we can achieve an SNR of 27.36dB. It also indicates that with a symbol-spaced FFE

(i.e., D=T), the increase in NT from 16 to 32 does not make any noticeable improvement in

SNR. In other words, NT=16 is sufficient.

For the case of fractionally spaced FFE (i.e., D=T/M with M>1), we can make the following observations: 5 ♦ The SNR is very much less sensitive to the sampling phase.

♦ For NT>5, increasing NT from 6 to 32 introduces an increase in SNR of around 2.5dB. As NT increases, the SNR increase get smaller and smaller. An SNR of 30dB is achievable.

♦ With NF=20, changing D=T/2 to D/4 will not provide a noticeable increase in SNR. 0 For NT=16 or larger, increasing NF from 20 to 80 does not provide a noticeable increase in

SNR.

The results indicates that combinations of NF=20, NT=16 and D=T or T/2 are good choices:

The combination of NF=20, NT=16 and D=T with a proper sampling provides an SNR of

27.36dB with symbol-spaced 16-tap FFE and 5 The combination of NF=20, NT=16 and D=T/2 can achieve an SNR of 29.63dB with a fractionally space 32-tap FFE, i.e. an increase of 2.27dB in SNR at the expense of doubling the sampling rate and number of FFE taps.

Numerical Results 0

SAMPLE SYSTEM: Using The Present Invention to Enhance High-Speed Data Transmission Over Existing Mil-STD-1553

Background of the System:

The AS1553 standard, commonly referred to as MIL-STD-1553, was introduced in the early 1970's to define a digital communications bus for the interconnection of different subsystems that were required to share or exchange information in a multi-drop configuration. Since its introduction, the AS 1553 standard has been evolving to incorporate functional and user community enhancements. However, the basic communications and architectural characteristics of the bus have not varied from its original release. Message-based communications over the multi-drop bus make use of the Manchester II bi-phase coding for IMb/s transmission in a half-duplex mode over Twisted-Shielded Pair (TSP) with 90% shield coverage. The largest message is 32 word long where a word has 20 bits. Transmission performance is specified for a word error rate (WER) of 10^"7 or better for an equivalent worst- case Additive White Noise (AWG) of 140mVrms in a bandwidth from 1kHz to 4MHz, and a signal level of 2.1 Vpp .

Problem:

Over the last 2 years, the Society of Automotive Engineers (SAE) Avionics Systems Subcommittee (AS-1A) has been investigating the use of different technologies to increase the data transfer capacity of existing AS 1553 networks. The SAE initiated this investigation in response to a tri-service request from the United States Department of Defense (DoD). The DoD request was driven by present and projected future needs for retrofitting existing weapon system platforms with subsystems that would demand more data transfer bandwidth. The main research objective is to find a solution to support robust, deterministic, and reliable transmission at higher data transfer rates over the existing physical cable plants. The primary goal of the transfer rate is 100 Mb/s. The desirable aim is the interoperability with existing AS1553 terminals and transformer assemblies.

lOOft-cable shows an insertion loss of 2dB or less for frequency range from 100kHz to 8MHz and the insertion loss increases rapidly beyond 8MHz. An insertion loss of 8dB was measured at 100MHz. A group delay variation within 3ns was measured for frequencies from 75kHz to 100MHz.

Many vendors had performed several tests on the data bus couplers to determine their operating characteristics in high speed applications. The following are the findings of the tests performed by RayChem. First, Amplitudes received at Stub 8 (8 Data bus Couplers in the loop), which is the last coupler, indicate that the bus tested may operate at up to 6MHz. Secondly, The stub cables cause little attenuation up to 10MHz. Thirdly, The signals are subject to little attenuation in passing through couplers. Fourthly, The presence of stubs has little effect and the main cause of attenuation is believed to be the bus cable.

High speed data transmission of digital data over C-17 cables requires adaptive equalization to equalize channel distortion and adaptive interference cancellation to remove both echo and crosstalk interference (NEXT's and FEXT's). Channel distortion includes mainly amplitude distortion and delay dispersion. It causes the smearing and elongation of the duration of each symbol. High speed 1553 network communications where the data symbols closely follow each other, particularly at multiple hundred megabit speeds, time dispersion results in an overlap of successive symbols, an effect known as inter-symbol interference (ISI). An Equalization system, in concert with a synchronous communication environment, alleviates the relative phase dispersion of the interfered and interfering signals that greatly reduces ISI. This is a critical factor affecting the C-17 or Mil-Std 1553 receiver performance. Interference (echo and crosstalk) is another major performance-limiting impairments on STP C-17 cables at the high speed communication. In many systems, perfect equalization and interference cancellation are not possible and residual ISI, NEXT's and FEXT's are present at the decision device (slicer). To deliver a robust Multi-Gigabit data stream over C-17 or Mil-Std 1553 cable in the

Advance High Speed 1553 embodiment of this system, the sources of interference and noise for a system were analyzed in order to provide methods for removing interference and increasing the Signal to Noise Ratio (SNR).

Summary of Application: This portion of the application describes the implementation details and performance of the system and method which enable High Speed Data Transmission over existing Mil-STD-1553 wireline communications, called MDI-1553+ which utilizes the Com2000™ Signal Equalization of Decision Precursor ISI Canceller (DPIC) (described above), and Com2000™ Signal Coding of Coded Synchronous M-PAM with the emphasis of backward compatibility with existing Mil-STD-1553 standard.

The above discussions indicate that it is desired to find advanced signaling techniques for high-speed data transmissions over the multi-drop bus using the existing MIL-C-17 Cable. The present invention provides a method and system, hereinafter referred to as the MDI-1553+, that create an enhanced 1553 System for supporting new terminals with data rate up to lOOMb/s using enhanced coupler. The invention also provides interoperability with existing low-speed AS 1553 terminals at rate IMb/s using the existing AS 1553 transformer assemblies.

As discussed in the previous section, the cable channel has a severe frequency-selective attenuation at frequencies beyond 1MHz, which limits the transmission at higher rate. The transmission using Manchester coding is limited by the bandwidth of IMHz in which the attenuation is relatively flat. However, the present invention further provides equalization techniques and advanced combined coding and modulation schemes enabling transmissions at lOOMb/s or above.

In the preferred embodiment, the system uses a baseband bandwidth up to 30MHz. Based on the previously mentioned study on lOOm-cable, the insertion loss variation is about 2dB, i.e., the frequency-selective attenuation has a depth of 2dB. Our DPIC equalization technique will be used to remove the inter-symbol interference and crosstalk due to such frequency-selective attenuation. Furthermore, adaptive equalization will be applied to adapt to a particular bus in use.

And, finally, for a larger bandwidth in use, multi-level modulation combined with advanced coding methods are provided to enhance both the bandwidth and power efficiencies. Multilevel modulation, such as baseband Synchronous Pulse Amplitude Modulation (SPAM) will increase the bandwidth efficiency required to support transmission of lOOMb/s over a bandlimited channel of up to 30MHz. However, it will require higher signal level to maintain the WER of 10^"7 for the specified noise floor.

Note that a specified noise floor of 140mVrms in a frequency range from 1kHz to 4MHz is equivalent to 383mVrms in a frequency range from 1kHz to 30MHz. In other words, the use of multi-level modulation scheme for high bandwidth efficiency alone will require a much larger signal level to maintain the same WER of 10^"7, especially when the bandwidth is also increased.

In order to reduce the signal level, combined power-efficient coding and modulation techniques are provided. The combined coding and modulation technique takes into account the frequency-selective attenuation of the cable. This is achieved by using a new signaling scheme that combines modulation, coding and advanced equalization for noise suppression to achieve a high performance and high capacity suitable to support lOOMb/s over the existing MIL-C-17-Cable.

MDI-1553+ signaling scheme for lOOMb/s speed does not require new coupler. When MDI- 1553+ transmits and receives in IMb/s and lOOMb/s speed, the current legacy passive coupler supports the new transceiver chip operations. However, an active coupler, which is provided power by the new MDI-1553+ node via a new stub wire, is optional. Some of challenges, such as additional power requirements, cooling concerns, and system reliability are all addressed.

Sample System Description Organization: In Section II, we describe the channel characteπstics and modeling used to evaluate the performance of various receiver structures. In Section III, we present the receiver structures currently for the IMb/s transmission of Mιl-STD-1553 over single Twisted Pairs cable and their limited performance. Section IV describes the MDI-1553+ Interference and Noise suppression via DPIC techniques and its applications in the design of various receiver structures using both Echo and NEXT cancellers Their performance and complexity as compared to the existing schemes are discussed. Section V descπbes an embodiment of the MDI-1553+ Receiver Architecture

SECTION II: CHANNEL CHARACTERISTICS AND MODELING

Propagation Loss:

The models for the propagation loss of a loop that are presented in this section are valid for frequencies that are larger than about 500 kHz. The signals considered in this description have a very small amount of energy below this frequency. Thus, for simplicity, we will assume that the propagation loss models discussed here are valid at all frequencies.

The transfer function H(d, f) of a perfectly terminated loop with length d can be written as follows:

H(d,f) = e'^d^ = _e-*^{« )} _e-J^dβ(f⁾ ₍₁₎| where γ (f)\ is the propagation constant, (/)|is the attenuation constant, and β(f) | is the phase constant. The quantity that is usually specified in practice is tb propagation loss for a given cable length (e.g., d = 100 meters). The propagation loss (or insertion loss) limit Lp (f) for C-17 100m cable is a positive quantity expressed in dB

L_P(f) = -201og|H(l,/)||

Channel Modeling:

Figure 20 shows the channel model including the effects of partial response, DAC and hybrid filtering in the transmitter, the main and coupling channel characteristics, and the filtering in the receiver front-end. The DAC and hybrid filtering is represented by the cascade of two identical first-order Butterworth sections with a corner frequency of 180MHz. This introduces a 4ns rise/fall time. The receiver front-end is modelled as a fifth-order Butterworth filter with a corner frequency of 80MHz. The main channel, echo coupling and NEXT coupling channels are represented by C(ω), E(ω), N₂(ω) respectively. The models for the FEXT's are similar to those of the NEXT's except the coupling channels will be F₂(ω) instead ofN₂(ω).

SECTION III : CURRENT 1553 TRANSCEIVER STRUCTURE

Figure 30 shows the data coding scheme and bus cabling architecture for the 1553 . The symbol timing recovery is shown in this figure. The receiver is a standard Bi-Phase Manchester Signaling Receiver.

SECTION IV: ADDITIONAL INTERFERENCE CANCELLATION AND

EQUALIZATION TECHNIQUES

The Challenge of Noise Suppression Substandard cabling attenuates signal strength and generates cross talk and noise that propagates over distance. This noise weakens the strength of the true signal decreasing available bandwidth. Traditional approaches to mitigate noisy lines have focused on static filtering techniques using DSP to condition signals. Because of their static nature, these techniques fall short of conditioning signals and suppressing noise adequately to guarantee specified bandwidth over the popular data signaling technologies: such as 1553 and Arinc MDI's Com2000^tm noise suppression methods intelligently seek out the highest amplitude signals within the sub-symbol and dynamically filter noise by locking in on the highest amplitude signal. This enables SSI's solution to condition both pre- and post-Inter Symbol Interference, and Near and Far End Interference, effectively improving noise suppression capacity by multiple orders of magnitude. The chart below illustrates the difference between the traditional approach and the Com2000™ approach. Reliable duplex operation at 300Mb/s over single pair of a C-17 Twisted Pairs cable requires the usage of some kind of technique to remove inter-symbol interference (ISI) and to combat interference including echo, NEXT and FEXT.

A NEXT canceller synthesizes, in an adaptive fashion, a replica of the NEXT interferer. The interferer is then cancelled out by subtracting the output of the canceller from the signal appearing at the receiver. A NEXT canceller has the same principle of operation as an echo canceller, and all the familiar structures used for echo cancellers can also be used for NEXT cancellers. The cancellers preferably have access to the local transmitters from which they get their input signals. Typically, this input signal is the stream of symbols generated by the transmitter's encoder.

The FFE (feed-forward equalizer) can be a symbol-spaced (SS) or fractionally spaced (FS) FFE or an analog equalizer. It is used to equalize the precursor ISI. The DFE is also used to remove the post cursor ISI. Note that the performance of the DFE is also dependent on the reliability of the symbols detected by the slicer and influenced by the error propagation. One may optionally replace the simple slicer by a sequence detector (such as Viterbi decoder) for a better performance. Figure 50A provides a graph demonstrating INTERS YMBOL INTERFERENCE (ISI) at high transmission rates over C-17 cable without modification. Figure 50B demonstrates the results of using Feedforward filtering and Equalization to suppress the precursor intersymbol interference in the signal.

SECTION V: THE MDI- 1553+ TRANSCEIVER

In the preferred embodiment, the MDI- 1553+ transceiver includes a 20-tap fractionally-spaced (T/2) equalizer, an 128-tap DFE and an 165-tap symbol-spaced echo canceller. As discussed above, the performance margin is very tight and a rate ³Λ 512-state Trellis Code is used in order to provide 6dB of coding gain required for a proper operation. It is therefore desired to enhance the transceiver

PROPOSED PHYSICAL LAYER

• Short command/signaling messages use 1 Mb/s so that all low- speed and high-speed devices can "understand". New highspeed devices can exchange data at rate of 100 b/s.

• The proposed physical layer protocol will automatically handle this arrangement so that it becomes transparent to upper layer protocols.

CONFIDENTIAL performance so that a larger margin can be provided. Additional margin can be used to increase the range (distance) or transmission rate. Figure 70 shows the proposed transceiver structure using DPIC.

The critical issues were that the required performance would include a BER of 1E-7 and that the margin used in theoretical and simulation studies would be 12dB, while the margin on a measured piece of equipment need only be 6dB. The crosstalk model has a NEXT loss of about 57dB at 80kHz and decreases at about 15dB per decade for frequencies above about 20kHz.

An embodiment of the system using a single-pair Advanced 1553 achieved a 17dB margin using coded Synchronous 16PAM on existing 1553 test loops for 300ft. The basics of the proposed MDI- 1553+ standard include the following recommendations:

• performance margin of 17 dB for IMb/s speeds and 2dB of margin on all 1553 loops (measured by increasing the crosstalk noise until a BER of 1E-7 is reached) at the speed of 300Mb/s.

• use of symmetrical Tx power spectrum S-PAM (Synchronous Overlap PAM Transmission with Interlocking Partial Response Spectra)

REDESIGN THE PHYSICAL LAYER using the available BW of up to 50MHz and a combined coding/modulation/equalization scheme: to provide high-speed data transfer at 100Mb/s (with the provision of nx100Mb/s) over the existing cabling system. maintaining the support of the current low data rate (1Mb/s, Manchester II Bi-phase), i.e., backward compatibility to support existing low-speed devices.

CONFIDENTIAL • use of a programmable encoder for rate-3/4, 512-state trellis codes for extra 6dB of coding gain. While the present system and method have been described with reference to specific embodiments, those skilled in the art will recognize that these procedures may be applied to all kinds communications channels and filtering mechanisms. Thus, the scope of this invention and claims should not be limited by the described implementations.

Claims

I claim:

1. In a bi-directional data channel between a central station and a multiple of remote stations, a method for equalizing interference over a synchronized packet or frame based baseband transmission system wherein the crosstalk on the system is cyclostationary or periodic with a period equal to a symbol interval, the method comprising the steps of : synchronizing the transmitters and receivers using the uncorrelated transmit signals; generating the cyclostationary NEXT and FEXT interference along with ISI using the uncorrelated symbols at the synchronized transmitters at one or more remote stations and the centrally station ; using cascaded Fractionally Spaced Linear Equalizer (FSLE) and Decision Feed back Equalizer (DFE) for both interference suppression and equalization to maximize signal to noise ratio with minimum excess bandwidth at the receivers; while holding the DFE contstant, increasing the receiver's FSLE filter taps (NT) to minimize pre-inter-symbol interference cursor and noise propogation; while holding the FSLE constant, increasing the receiver's DFE filter taps (NT) to minimize post- inter-symbol interference cursor; combining FSLE/DFE with proper phase sampling adjustments, to maximize the signal to noise ratio thereby enabling use of the spectral correlation properties peculiar to the modified signals.

2. The method as recited in claim 1, wherein the step of combining further comprises the step of increasing the receiver's FSLE filter taps to 16, wherein the communciations channel comprises a 100 meter lOOObaseT ethernet cable.

3. The method of claim 2, wherein the step of combining further comprises the step of doubling the sampling rate of the signal.

4. The method of claim 3, wherein the step of combining further comprises the step of increasing the receiver's DFE filter tap to 20.

5. The method of claim 4, wherein the step of combining comprises the step of increasing the FSLE and DFE filter taps to 32.

6. The method as recited in claim 1, wherein the step of synchronizing comprises transferring synchronization information, wherein the information comprises sending the time of a precision clock between the remote station and central station.

7. The method as recited in claim 6 wherein the step of synchronizing further comprises transferring synchronization information, wherein the synchronization information includes transferring the frequency and phase of the signal between the remote station and central station.

8. The method of claim 7, wherein the synchronization information is transmitted over the signal itself.

9. The method of claim 8, wherein the synchronization information is transferred over the signal without using overhead subchannels.

10. A method as recited in claim 1, wherein the method further comprises the steps of: measuring and calibrating the communication channel by sending control signals between the synchronized remote stations and central station; characterizing the communications channel so that imperfections, in frequency, phase and time distortions, can be identified; using the communication channel measurements to enhance signal quality by controlling the measured errors and imperfections of the channel.