WO2000036799A2 - Linearisation d'emetteur - Google Patents

Linearisation d'emetteur Download PDF

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Publication number
WO2000036799A2
WO2000036799A2 PCT/FI1999/001050 FI9901050W WO0036799A2 WO 2000036799 A2 WO2000036799 A2 WO 2000036799A2 FI 9901050 W FI9901050 W FI 9901050W WO 0036799 A2 WO0036799 A2 WO 0036799A2
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WO
WIPO (PCT)
Prior art keywords
transmitter
linearization parameters
parameters
linearization
signal
Prior art date
Application number
PCT/FI1999/001050
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English (en)
Other versions
WO2000036799A3 (fr
Inventor
Mikko Huttunen
Original Assignee
Nokia Networks Oy
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Nokia Networks Oy filed Critical Nokia Networks Oy
Priority to AU19859/00A priority Critical patent/AU1985900A/en
Publication of WO2000036799A2 publication Critical patent/WO2000036799A2/fr
Publication of WO2000036799A3 publication Critical patent/WO2000036799A3/fr

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
    • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
    • H04L27/36—Modulator circuits; Transmitter circuits
    • H04L27/366—Arrangements for compensating undesirable properties of the transmission path between the modulator and the demodulator
    • H04L27/367—Arrangements for compensating undesirable properties of the transmission path between the modulator and the demodulator using predistortion
    • H04L27/368—Arrangements for compensating undesirable properties of the transmission path between the modulator and the demodulator using predistortion adaptive predistortion
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/16—Multiple-frequency-changing
    • H03D7/165—Multiple-frequency-changing at least two frequency changers being located in different paths, e.g. in two paths with carriers in quadrature
    • H03D7/166—Multiple-frequency-changing at least two frequency changers being located in different paths, e.g. in two paths with carriers in quadrature using two or more quadrature frequency translation stages
    • H03D7/168—Multiple-frequency-changing at least two frequency changers being located in different paths, e.g. in two paths with carriers in quadrature using two or more quadrature frequency translation stages using a feedback loop containing mixers or demodulators

Definitions

  • the invention relates to linearization of a radio transmitter.
  • the scarcity of radio frequencies leads to a need to use spectrum- efficient modulation methods in new radio systems.
  • PMR Professional Mobile Radio
  • TETRA Terestrial Trunked Radio
  • ⁇ /4-DQPSK ⁇ /4-shifted Differential Quadrature Phase Shift Keying
  • IM InterModulation
  • IM products spread the spectrum of the transmitted signal, and thus tend to reduce the benefits of using a linear modulation method.
  • the IM products cannot usually be filtered as they form very close to the desired signal.
  • spectrum spreading does not occur, and consequently the signal may be amplified by a nonlinear amplifier.
  • a trunked PMR system in which different user groups share same radio channels, has strict requirements regarding adjacent channel interference caused by a transmitter. These requirements call for a good linearity in the transmitter of the used radio system.
  • nonlinearities of an amplifier were known in advance, it would be possible to form inverse functions of the nonlinearities, and use them to convert the input signal, whereby the nonlinearities would be cancelled.
  • the properties of the amplifier do not, however, remain the same but vary due to ageing, warming up, and according to the radio channel and transmit power used.
  • the amplifiers have individual differences. Need exists for linearization methods that must in an adaptive way be capable of adapting to changing conditions. Development work has been targeted into a number of different linearizing methods, and three of them have been found to possess qualities suitable for practical radio systems. These methods are feedforward, cartesian feedback and predistortion.
  • a linearization method can also be adaptive.
  • the signal to be transmitted can be linearized by applying a suitable transfer function causing a predistortion to the signal.
  • a suitable transfer function causing a predistortion This way, the signal coming out of the amplifier can be made linear.
  • This method is called predistortion.
  • Predistortion is usually performed on baseband using a lookup table into which the conversion parameters causing the predistortion, i.e. the predistortion parameters, are stored.
  • the invention is based on the idea that predistortion parameters used by a transmitter predistorter are formed by means of the normal signal being transmitted, i.e. the information payload.
  • Information payload thus mainly refers to a signal which is not transmitted for the purpose of linearization only, but would be transmitted anyway. According to the invention, this information payload can also be used for transmitter linearization.
  • the invention provides the advantage that no separate training sequence is needed and information payload can be continuously transmitted. Another advantage is that the time for defining the linearization parameters can be decided on according to the requirements of the transmitter, for instance, without being bound to prior art linearization time slots defined by the system.
  • Figure 1 shows a block diagram of a transmitter of the invention according to one embodiment
  • Figure 2 shows characteristic curves of the amplitudes of amplifiers and a predistorter
  • Figure 3 shows a characteristic curve of phase difference.
  • FIG. 1 shows a block diagram of a transmitter of the invention according to one embodiment. It should be noted that the figure shows only the elements that are essential for understanding the invention.
  • the transmitter receives an I and Q signal l_IN and QJN for transmission.
  • Predistortion has been implemented using a digital signal processor (DSP) 1.
  • DSP digital signal processor
  • Predistortion of the amplitude occurs in the multiplier elements 3A and 3B in accordance with an amplitude correction parameter obtained from an amplitude table 19.
  • a possible phase predistortion occurs in a phase shifter element 4 in accordance with a phase correction parameter obtained from a phase table 20.
  • the predistorted signals are D/A-converted using D/A converters 5A and 5B into analogue signals which are also preferably filtered using low-pass reconstruction filters 6A and 6B. These signals are forwarded on to an l/Q modulator 7.
  • the l/Q modulator 7 and an l/Q demodulator 15 operate according to the quadrature modulation principle. With this principle, it is possible to combine two independent signals in the transmitter, to transmit them on the same transmission band and then to separate them from each other in a receiver.
  • the quadrature modulation principle is that two separate signals, I and Q (Inphase and Quadrature phase), are modulated using the same carrier frequency, but the phases of the carriers differ in that the carrier of the signal Q lags 90° behind the carrier of the signal I.
  • the signals are summed. Due to phase difference, the signals I and Q can be separated from each other when the sum signal is demodulated.
  • the signals are modulated and combined.
  • the l/Q modulator is synchronized by a local oscillator 10.
  • the l/Q-modulated signal is forwarded to a power amplifier PA 8 and onward to an OUT antenna 2 for transmission.
  • feedback is formed by means of a sampling arrangement 9, for instance a directional coupler.
  • a radio frequency (e.g. 400 MHz) feedback signal is preferably down-converted to an intermediate frequency of 450 kHz, for instance, using a down-converter 12.
  • the down-conversion is synchronized by a local oscillator 11.
  • the intermediate frequency signal can, if necessary, be filtered using a broadband filter 13 and subsequently attenuated.
  • the intermediate frequency signal is sampled using an A/D converter 14 for baseband (or intermediate frequency) processing.
  • the l/Q demodulator 15 has been implemented using a digital signal processor 1.
  • the A/D-converted intermediate frequency feedback signal is l/Q-demodulated with the l/Q demodulator 15 by digital multiplication into baseband I and Q signals.
  • the l/Q demodulator is implemented by program in the digital signal processor 218, and a separate analogue l/Q demodulator is not needed. l/Q demodulation could also be performed as an analogue process without affecting the invention.
  • the baseband I and Q feedback signals are preferably converted from an orthogonal presentation to a polar presentation in a converter 16B, whereby the amplitude and phase are directly obtained from the converted signals and forwarded to a calculation unit 17 in which the predistortion tables 19 and 20 are generated, which generation is described later in this description.
  • the signals IJN and QJN received by the transmitter for transmission, which are preferably also converted to a polar presentation in the converter 16, are also forwarded to the calculation unit 17.
  • the actual predistortion is done by means of the created predistortion tables
  • phase table 19 and 20 An absolute value of the complex signal formed by the signals IJN and QJN, i.e. the amplitude of the signal, is defined in an element 18.
  • This amplitude data is entered into the amplitude predistortion table 19 which provides a corresponding amplitude correction parameter to the multiplier elements 3A and 3B on the basis of the amplitude data.
  • a corrected, i.e. predistorted amplitude data, formed using a multiplier element 21 is entered into the phase predistortion table 20. On the basis of this data, the phase table
  • the amplitude data is preferably used as control data in the phase table 20, because a phase error in the transmitter depends on the amplitude of the signal.
  • the amplitude data received by the phase table 20 is preferably also predistorted and thus corresponds to the amplitude of the signal coming out of the predistorter, whose magnitude determines the nonlinearity in the transmitter, and the phase predistortion required can be defined as accurately as possible. It is also possible to use amplitude predistortion only, which means that the phase predistortion table 20 and phase shifter element 4 are not needed.
  • samples 22 and 23 of the transmitter input signals IJN and QJN and samples 24 and 25 of the (complex) transmitter output signal OUT are used as the inputs of the calculation element 17 in Figure 1.
  • the signals 22 and 23 together form a complex reference signal REF in polar presentation.
  • the reference signal REF is ideal in that it does not have any nonlinearity caused by the transmitter.
  • the signals 24 and 25 together form a complex feedback signal FB in polar presentation.
  • the signals REF and FB are normalized so that the highest amplitude in both signals is 1.
  • the reference signal REF and the corresponding feedback signal FB arrive at different times at the calculation element 17, because the transmitter causes a certain delay in the transmission of the signal, i.e.
  • the value of the reference signal REF corresponding to a certain sample point of the incoming complex signal IJN and QJN arrives at the calculation element 17 earlier than the value of the corresponding feedback signal FB. Because of this, the reference signal REF is buffered in order to be able to compare in the calculation element 17 the value of a certain input complex signal IJN and QJN which was input into the transmitter and the corresponding value which comes out of the transmitter.
  • the predistortion parameters in tables 19 and 20 are set to values in which no predistortion occurs. After this, a suitable signal is fed into the transmitter and a predefined number of samples (e.g.
  • the amplitude values of the mutually corresponding sample points of the sample signals REF and FB are categorized into a required number of classes (e.g. 128 to 16,384) on the basis of the amplitude of the reference signal REF.
  • the samples are preferably categorized on the basis of the amplitude, because the nonlinearity of the transmitter depends on the transmit power which, for its part, depends on the amplitude of the signal.
  • the values of the sample points of the feedback signal FB are compared with the corresponding buffered values of the reference signal REF, and the correction parameters are defined based on this comparison.
  • Straight line 31 shows the dependence of the normalized output amplitude of an ideal transmitter on the normalized input amplitude.
  • Straight line 31 also shows the dependence of the reference signal REF on the input signal IJN and QJN of the input signal, i.e. they are equal.
  • Curve 32 shows the nonlinear characteristic curve of the transmitter (when no predistortion is used). The nonlinearity of the transmitter is mainly caused by the power amplifier 8. The transmitter can also have several power amplifiers 8 in series.
  • Curve 32 is defined on the basis of the sample signals REF and FB for instance as follows: an average of the amplitudes of the sample points of the feedback signal FB in every class is calculated.
  • an average of the sample points of the reference signal REF corresponding to the sample points of the feedback signal FB in every class is calculated.
  • the value of curve 32 at the centre point of the class in question in relation to the ideal curve 31 is obtained with the ratio of the calculated averages.
  • the centre points of classes are marked with circles.
  • the ratio can also be calculated by first defining the ratio of the amplitudes of each sample point of the feedback signal FB and the corresponding sample points of the reference signal REF and then defining the average of the ratios of the sample point pairs in each class.
  • Curve 33 is the characteristic curve of the predistorter obtained by mirroring the characteristic curve 32 of the transmitter defined without predistortion in relation to the ideal curve 31.
  • the counterpart of point 41 is determined by first finding a point of the ideal curve 31 that corresponds to the point 41 on the basis of the output amplitude value. In the example, where the output amplitude is 0.4, the result is the point 42 of the ideal curve 31. Next, a point is searched whose input amplitude value is the same as that of the defined ideal curve point 42 (0.4) and whose output amplitude value is the same as the input amplitude value (0.2) of the point 41 of the characteristic curve 32. In this case, the counterpart of the point 41 is the point 43. A counterpart is defined for each point of the characteristic curve 32 in the same way.
  • the counterparts form the characteristic curve 33 of the predistorter.
  • the characteristic curve 32 of the transmitter is discrete in relation to the input amplitude (the x axis)
  • the characteristic curve 33 of the predistorter defined on the basis of it is discrete in relation to the output amplitude (the y axis), i.e. they have been defined in the marked points only (the circles and the x's) due to the categorization of the samples into classes.
  • the more classes are used, the more points are obtained, and, further, the closer one gets to the continuous curves 32 and 33 marked with a continuous line in the figure.
  • the amplitude predistortion table 19 can be created. Because the predistorter characteristic curve 33 is discrete in relation to the output amplitude, i.e. the amplitude correction parameter has been defined for certain output amplitude values only, as described above, and because, on the other hand, the predistorter input amplitude can obtain any values, the correction parameter to be used for a certain input amplitude is defined by finding the point closest to the input amplitude on the discrete curve 33 and using the amplitude correction parameter corresponding to this point.
  • the amplitude correction parameter refers here to a value by which the input amplitude should be multiplied to obtain the required output amplitude according to the characteristic curve 33 of the predistorter.
  • the correction parameter is thus obtained at a certain point of curve 33 by dividing the value of the output amplitude by the value of the input amplitude at the point in question.
  • the amplitude predistortion table 19 can be formed by defining for a certain defined correction parameter a certain input amplitude value range within which the correction parameter is used. This can be done by means of the characteristic curve 33 by dividing curve 33 into ranges in relation to the input amplitude, the centre points of the ranges (marked with x's) being the points at which the output amplitude and correction parameter have been defined.
  • FIG. 3 illustrates the definition of the correction parameters of the phase difference (between the sample point of the feedback signal FB and the sample point of the corresponding reference signal REF).
  • the figure shows an example of phase difference as a function of a normalized amplitude (input amplitude) of the reference signal REF, i.e. curve 51.
  • the characteristic curve 51 of the phase is formed by defining the average phase differences of the sample point pairs of the feedback signal FB and the reference signal REF in each class into which they have been categorized on the basis of the amplitude when defining the amplitude correction parameters as described above.
  • the phase differences are categorized into classes on the basis of the normalized amplitude of the reference signal REF and the average of the phase differences is calculated for each class.
  • the characteristic curve values of the amplifier phase at the centre point of each class i.e. the points marked with circles in Figure 3 forming the discrete characteristic curve of the phase.
  • the solider curve 51 becomes.
  • the phase predistortion table 20 can be created. Because the characteristic curve 51 of the phase is discrete, i.e.
  • the phase difference has been defined for certain input amplitude values (class averages) only, as described above, and, on the other hand, because the input amplitude can obtain any values, the correction parameter for a certain input amplitude is defined by finding the point closest to the input amplitude on the discrete characteristic curve 51 of the phase and using the phase correction parameter corresponding to this point.
  • the phase correction parameter refers here to the value which defines how much the phase of a signal arriving at the predistorter must be shifted and into which direction to achieve the required phase predistortion. The correction parameter is thus obtained at a certain point of the characteristic curve 51 by multiplying the phase difference value by -1 at the point in question, i.e. the phase is predistorted into the opposite direction by the amount of the phase difference.
  • the phase predistortion table 20 can be formed, for instance, by defining for a certain defined correction parameter a certain input amplitude value range within which the correction parameter is used. This can be done by means of the characteristic curve 51 by dividing curve 51 into ranges in relation to the input amplitude, the centre points of the ranges (marked with circles) being the points at which the output amplitude and, thus also, the correction parameter have been defined. The table can then be used for checking, into which range the value of the input amplitude belongs, and the correction parameter corresponding to the range can then be used.
  • the format of the data in the predistortion tables 19 and 20 can differ from the above without making any difference to the basic idea of the invention.
  • the creation of the predistortion tables can differ from what is described above.
  • a combined predistortion table can be used in which a certain input amplitude value has a complex correction parameter which includes both the amplitude and phase correction parameters. It is also possible to use only amplitude predistortion, which reduces memory use, since a phase predistortion table is not needed.
  • the update of the linearization parameters of the invention is mainly done as follows: the linearization parameters are maintained in the transmitter and the information payload to be transmitted is predistorted using the linearization parameters. New linearization parameters are defined by means of the transmitted information payload as described above, for instance.
  • the predistortion tables 19 and 20 are set to values which cause no predistortion, whereby the nonlinearity in the transmitter can be defined directly by comparing the incoming IJN and QJN signal and the outgoing OUT signal, as described above. It is also possible to define the linearization parameters while predistortion is being used. In such a case, the comparison is made between the predistorted signal fed into the transmitter and the signal going out of the transmitter.
  • the new linearization parameters have been defined, the old parameters are replaced by the new ones and the signal to be transmitted is predistorted according to the new parameters.
  • the update of the linearization parameters can be done at predefined intervals, for instance, or when the spectrum of the transmitted signal spreads over a predefined limit, or in response to a prompt from an outside instance when for instance the accumulator voltage, temperature, used frequency or some other corresponding element of the transmitter changes.
  • the new linearization parameters can be compared with the ones being used, and the parameters being used can be replaced by the new parameters only if the new ones differ significantly from the ones being used. Only a part of the parameters can be defined for such a comparison, and the remaining will be defined only, if necessary.
  • the linearization parameters are preferably stored in the memory 19 and 20 of the transmitter, whereby in the beginning of a new transmission, the linearization parameters stored during the previous transmission can be used. Also, the memory 19 and 20 of the linearization parameters is preferably non-volatile in that the data remains even though the transmitter is switched off. This way, when the transmitter is switched on again, the linearization parameters stored earlier can be used in the beginning.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Power Engineering (AREA)
  • Transmitters (AREA)
  • Amplifiers (AREA)

Abstract

L'invention a trait à un procédé de mise à jour des paramètres de linéarisation d'un dispositif de distorsion préalable d'émetteur, et à un émetteur. Dans le procédé, des paramètres de linéarisation sont stockés dans un dispositif de mémoire (19, 20), la charge utile d'informations (I IN, Q IN) à transmettre fait l'objet d'une distorsion préalable effectuée à l'aide des paramètres de linéarisation dans un dispositif de distorsion préalable (3A, 3B, 4), la charge utile d'informations ayant fait l'objet d'une distorsion préalable est transmise dudit émetteur vers un conduit hertzien, ce qui permet de définir de nouveaux paramètres de linéarisation dans un moyen de définition (17) au cours de ladite transmission à l'aide de ladite charge utile d'informations transmise (OUT), et de remplacer les paramètres de linéarisation utilisés par ces nouveaux paramètres de linéarisation.
PCT/FI1999/001050 1998-12-17 1999-12-16 Linearisation d'emetteur WO2000036799A2 (fr)

Priority Applications (1)

Application Number Priority Date Filing Date Title
AU19859/00A AU1985900A (en) 1998-12-17 1999-12-16 Transmitter linearization

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
FI982739A FI982739A (fi) 1998-12-17 1998-12-17 Lähettimen linearisointi
FI982739 1998-12-17

Publications (2)

Publication Number Publication Date
WO2000036799A2 true WO2000036799A2 (fr) 2000-06-22
WO2000036799A3 WO2000036799A3 (fr) 2000-09-14

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FI (1) FI982739A (fr)
WO (1) WO2000036799A2 (fr)

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2003009464A2 (fr) * 2001-07-20 2003-01-30 University Of Bristol Melangeur linearise utilisant la retransposition de frequences

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5093637A (en) * 1987-12-27 1992-03-03 Mitsubishi Denki Kabushiki Kaisha Modulation device with input signal modification for correction of amplifier nonlinearities
EP0658975A1 (fr) * 1993-12-14 1995-06-21 ALCATEL ITALIA S.p.A. Système à prédistorsion pour signaux en bande de base pour la linéarisation adaptative d'amplificateurs de puissance
EP0731556A1 (fr) * 1995-03-06 1996-09-11 NEC Corporation Amplificateur de puissance linéaire utilisant des valeurs de correction mises à jour en amplitude et en phase
EP0907276A1 (fr) * 1997-06-30 1999-04-07 Harris Corporation Dispositif de prédistorsion adaptatif pour la linéarisation d'une sortie d'amplicateur dans un système de transmission de données

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5093637A (en) * 1987-12-27 1992-03-03 Mitsubishi Denki Kabushiki Kaisha Modulation device with input signal modification for correction of amplifier nonlinearities
EP0658975A1 (fr) * 1993-12-14 1995-06-21 ALCATEL ITALIA S.p.A. Système à prédistorsion pour signaux en bande de base pour la linéarisation adaptative d'amplificateurs de puissance
EP0731556A1 (fr) * 1995-03-06 1996-09-11 NEC Corporation Amplificateur de puissance linéaire utilisant des valeurs de correction mises à jour en amplitude et en phase
EP0907276A1 (fr) * 1997-06-30 1999-04-07 Harris Corporation Dispositif de prédistorsion adaptatif pour la linéarisation d'une sortie d'amplicateur dans un système de transmission de données

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2003009464A2 (fr) * 2001-07-20 2003-01-30 University Of Bristol Melangeur linearise utilisant la retransposition de frequences
WO2003009464A3 (fr) * 2001-07-20 2003-09-25 Univ Bristol Melangeur linearise utilisant la retransposition de frequences

Also Published As

Publication number Publication date
FI982739A (fi) 2000-06-18
WO2000036799A3 (fr) 2000-09-14
FI982739A0 (fi) 1998-12-17
AU1985900A (en) 2000-07-03

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