WO1999027647A2 - Symmetrical biasing architecture for tunable resonators - Google Patents
Symmetrical biasing architecture for tunable resonators Download PDFInfo
- Publication number
- WO1999027647A2 WO1999027647A2 PCT/US1998/025373 US9825373W WO9927647A2 WO 1999027647 A2 WO1999027647 A2 WO 1999027647A2 US 9825373 W US9825373 W US 9825373W WO 9927647 A2 WO9927647 A2 WO 9927647A2
- Authority
- WO
- WIPO (PCT)
- Prior art keywords
- resonating
- tunable
- bias
- reactance
- electrically
- Prior art date
Links
Classifications
-
- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
- H01P7/00—Resonators of the waveguide type
- H01P7/08—Strip line resonators
- H01P7/088—Tunable resonators
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03J—TUNING RESONANT CIRCUITS; SELECTING RESONANT CIRCUITS
- H03J1/00—Details of adjusting, driving, indicating, or mechanical control arrangements for resonant circuits in general
Definitions
- the present invention is related to tunable resonators
- an electrically tunable element e.g., a voltage
- biasing means e.g., a power source and bias lines
- the first and second positions are the positions of
- the electrically tunable element can include a dielectric material having an index of refraction that is a function of
- the index of refraction can be dielectric permittivity for the capacitor (which is a function of bias voltage) cr magnetic permeability for the inductor (which is
- the dielectric material can be a bulk or thin film ferroelectric
- the maintaining means can include a second
- the first and second voltage nodes are
- the elements could be
- the tunable elements are located
- the tunable elements are connected in series with the resonating element.
- the biasing means includes first and second connections between the biasing means and the resonating element. Each of the first and second connections are located at a voltage node. A voltage node can be located between the electrically tunable elements on either side of
- the voltage nodes are typically about
- the resonating element is symmetrical about a line passing
- the resonating element performs the
- first and second biases have different magnitudes
- the resonating element can include a second
- step can include maintaining the reactance and the second
- Fig. 1 depicts an electrically tunable resonating element
- Fig. 2 depicts an electrically tunable resonating element
- Fig. 3 is a plot of RF voltage versus position along the
- Fig. 4 depicts an electrically tunable resonating element
- FIG. 5 is another view of the circuit of Fig. 4;
- Fig. 6 is a plot of transmission resonance versus
- Fig. 7 depicts various plots of capacitance versus bias
- Fig. 8 is a plot of K- factor versus dc voltage
- Fig. 9 is a plot of RF voltage versus inverse tunability.
- qu-a ⁇ cr-wavc impedance transformers are usually incorporated into the bias circuitry to make the bias circuits imped-ance iargc as seen from the resonator, and contact between the bias circuitry and resonator is made ut the resonutor's voltuge node where the resonunt mode's impedance is zero.
- These impedances will vary when the bias circuitry is employed to lune the resonator, i.e. the location of the voltage-node will shift and the l.arge impedance of the quarter- wave transformers will drop.
- the symmetrical biasing architecture ensures that the voltage nodes remain stationury when the resonator is tuned. These stationary voltage-nodes arc used as the contact points to the resonator for the bias circuitry to ensure that no resonant energy will escape the resonator into the bias circuitry.
- Two examples of the symmetrical bias architecture are drawn below, In the first example, a loop resonator has two identical, tunable capacitors inserted in series into it at positions 180° from pach other. This makes the frequency of the lowest order resonant mode (as well as all the odd ordered modes) dependent on the value of the tunable capacitors.
- the points 90° from the cupacitors axe voltage-nodes of the lowest frequency resona- nt mode (and for the next-to-lowest frequency mode which is even-ordered und thus non-tunable). These points remain voltage nodes regardless of the value of the capacitors as long as that value is the same for both capacitors. Bias circuitry may be connected to these voltage nodes without disturbing the resonunt mode.
- two identical, series capacitors arc placed together in holes with a loop resonator. The point between these capacitors and the point 180° from that point are v ⁇ llage-nodes and remain voltage nodes regardless of the value of the capacitors as long as that value is the same for both cnpacitors.
- the first example has the advantage that the microwave voltage gradient is less steep near the voltage nodes and thus is mode tolerant to dissimilarities between the two cap.acitors while the second example has the advantage of a smaller overall size.
- Thin films of SrTi ⁇ 3 and Bao. Sro.fi i 3 have been pulse laser ablated onto La ⁇ lO.1 substrates. Normal metal coplanar capacitor electrodes were patterned on top of these films and the capacitors were incorporated into weakly coupled microstrip resonators. Resonant frequencies und Q's were measured as a function of bias at room temperature and at 77 K. The microwave frequency capacitance and loss is calculated from the resonant properties and compur ⁇ d with the simultaneously measured 1 MIJ7, capacitance and dis.sipation. Two-tone intermodular on distortion products were measured .and the third-order intercept is referenced to the microwave voltage across the capacitors. Commercially available semiconductor v ⁇ ractors were tested in a similar manner.
- Tuning quality (the ratio of the relative capacitance tuning to dissipation), frequency dispersion, and power handling of these capacitors is compared. Although there appears to be no intrinsic power handling capability of the paraelectrics over the semiconductor varactors, the paraelectrics can offer tuning quality advantages.
- the nonlinear dielectric constant of paraelectric films such as SrTiO, and Ba,. x Sr,TiO ⁇ can be used to tune microwave signal processing components such as phase shifters, 1 ' * rcsonators ' 1 ' 5,6 filters, 78 and voltage controlled oscillators.
- Tunable capacitors mude from these films may offer some advantages over semiconductor v ⁇ ractors for tunable microwave applications, especially at X band and higher microwave frequencies. Possible advantages include a better tuning to loss ratio and the possibility of designing for higher power handling capability. This paper directly compares the two technologies at frequencies near 1.5 GHz.
- Coplanar, intcrdigital electrodes (separated by 1 ⁇ m) of e-beam ' evaporated normal metals (10 nm Ti / 100 nm ⁇ u / 6 ⁇ ra Ag / ] 50 nm ⁇ u) were formed on top of these films by lift-off. The samples were then diced into individual capacitors. Abrupt junction varactors based on G ⁇ As (cat. M ⁇ 46506-1056) and Si (cut. it M ⁇ 45234-1056) were purchased from M/A-COM, Inc. for comparison. Abrupt junction GaAs varactors were also purchased from MDT, Inc. (cat. U MV2008-36).
- Capacitors were reflow soldered (52% In, 48% Sn) into a microstrip resonator oj * characteristic impedance Z, depicted in Fig.5.
- Transmission resonances ( .. shown in Fig. ⁇ ) were measured by connecting a vector network analyzer (HP8510C) to the resonator's ports 1 und 2 which are weakly coupled to the resonant line so that the measured, loaded Q, ⁇ , , is approximately equal to the unloaded Q.
- Tine resonator design is such that the lowest order mode is highly sensitive to the capacitance and dissipation of inserted capacitors (5- > 0.4 for C > 1.35 pF).
- the relation between j, and C for the resonator of Fig. ' (for the tunable, odd-ordered mode) is
- L eJ . is calibrated by measuring linear capacitors of normal metal, interdigital electrodes on AO and .assuming no frequency dispersion between the 1 MHz capacitance and the microwave frequency capacitance.
- the even-ordered, non-tunable mode may also be used to calibrate ttr .
- the phase velocity, v ⁇ , and Z 0 are calculated from the resonant line geometry and L ⁇ O's dielectric constant 9 using standard microstrip design equations.
- the microwave voltage across the capacitor can be calculated as a reference point for the third order intercept of the power measurements.
- the resonant line is formed by lift-off of e-beam evaporated melals (10 nm Ti / 100 nm ⁇ u / 6 ⁇ m Ag / 2 ⁇ m ⁇ u) on LAO.
- the losses due to it are dominated by the conductor losses.
- Values of Q m in this study range from 20 to 200 indicating they are dominated by the varactor loss.
- Tuning quality, K is the ratio of the element's "tunability" to loss. 12 H Equivalent differential forms of it can be defined as
- the STO capacitors were measured at 77 K while the BST, Si, and GaAs capacitors were measured at room temperature. Their capacitance and dissipation near 1.5 GHz is displayed in Fig. 7. Data from the MDT, Inc. varactor is not shown here for brevity. (Raw resonant data for the MDT, Inc. varactor is displayed in Fig. B) The simultaneously measured 1 MHz capacitance is similar but not indistinguishable from the high frequency capacitance for all samples. Capacitance frequency dispersion is greater at low bias than at high bias for all samples.
- the high frequency capacitance is 8% lower than the 1 Mlfo capacitance but the difference decreuses to less than 1% at biases greater than 15 V. (This trend has previously been observed in STO varactors. 5,15 )
- the BST's high and low frequency capacitances are within 3% of each other at all biases.
- the ⁇ aAs sample's high frequency capacitance is 13% larger than the 1 Milz capacitance at zero bias decreasing to 11% at high bi ⁇ .
- the Si sample's hi h frequency capacitance is 12% larger than the 1 MHz capacitance at .zero bias decreasing to a 10% difference at high bias. The apparent -10% .
- Figure 8 is a comparison of the various varactors' integrated -factors where K. is plotted versus the bias range employed. Note that the x-axis is geometry dependent and that if the paraelectric capacitor gaps were reduced from, for example, 10 ⁇ m to 5 ⁇ m, then roughly the same K.-factor would be achieved at half the bias.
- the MDT, Inc. varactor is quoted to break down near 48 V limiting its K to 55 at its measurement
- the MDT, Inc GaAs varactor is of a lower capacitance (0.35 pF ⁇ C ⁇ 1.13 pF) than the M/A-COM GaAs sample (1.4 pF ⁇ C ⁇ 6.27 pF) .and is thus measured at a higher frequency (2.1 GHz ⁇ f 0 ⁇ 3 GHz) than the M/A-COM GaAs sample (1.0 GHz ⁇ f ⁇ ⁇ 1.9 GHz) resulting in a larger measured dissipation.
- BST would provide significant tuning quality advjintagcs over the semiconductors at higiier microwave frequencies and STO would provide vastly better tuning quality if one is willing lo employ cryogenics. Higher frequency measurements arc needed to ascertain the accuracy of these speculations.
- V m/ is the TOl of the resonator referenced to the microwave voltage across the capacitor and g is a function independent of the material upon which the varactor is based.
- Figure 9 is a plot of V lvl versus the inverse tunability of the various varactors. Error b.ars are derived from the deviation of the third order slope from 3 und they reflect the sign of the deviation. No other errors arc considered.
Abstract
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Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
AU19024/99A AU1902499A (en) | 1997-11-26 | 1998-11-24 | Symmetrical biasing architecture for tunable resonators |
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US6656497P | 1997-11-26 | 1997-11-26 | |
US60/066,564 | 1997-11-26 |
Publications (2)
Publication Number | Publication Date |
---|---|
WO1999027647A2 true WO1999027647A2 (en) | 1999-06-03 |
WO1999027647A3 WO1999027647A3 (en) | 1999-09-16 |
Family
ID=22070295
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
PCT/US1998/025373 WO1999027647A2 (en) | 1997-11-26 | 1998-11-24 | Symmetrical biasing architecture for tunable resonators |
Country Status (2)
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AU (1) | AU1902499A (en) |
WO (1) | WO1999027647A2 (en) |
Citations (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4500854A (en) * | 1981-04-20 | 1985-02-19 | John Fluke Mfg. Co., Inc. | Voltage-controlled RF oscillator employing wideband tunable LC resonator |
US4799034A (en) * | 1987-10-26 | 1989-01-17 | General Instrument Corporation | Varactor tunable coupled transmission line band reject filter |
US5021757A (en) * | 1988-11-28 | 1991-06-04 | Fujitsu Limited | Band pass filter |
-
1998
- 1998-11-24 AU AU19024/99A patent/AU1902499A/en not_active Abandoned
- 1998-11-24 WO PCT/US1998/025373 patent/WO1999027647A2/en active Application Filing
Patent Citations (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4500854A (en) * | 1981-04-20 | 1985-02-19 | John Fluke Mfg. Co., Inc. | Voltage-controlled RF oscillator employing wideband tunable LC resonator |
US4799034A (en) * | 1987-10-26 | 1989-01-17 | General Instrument Corporation | Varactor tunable coupled transmission line band reject filter |
US5021757A (en) * | 1988-11-28 | 1991-06-04 | Fujitsu Limited | Band pass filter |
Also Published As
Publication number | Publication date |
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AU1902499A (en) | 1999-06-15 |
WO1999027647A3 (en) | 1999-09-16 |
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