WO1999013590A1

WO1999013590A1 - Rf transmitter

Info

Publication number: WO1999013590A1
Application number: PCT/GB1998/002716
Authority: WO
Inventors: Marcus Antony Westall; William Stancer
Original assignee: Gec-Marconi Communications Limited
Priority date: 1997-09-09
Filing date: 1998-09-09
Publication date: 1999-03-18
Also published as: GB2329085A; GB9719224D0; AU9083698A; GB2329085B

Abstract

An RF transmitter employs negative feedback to reduce non-linearities in power amplifying means (18 to 21) and, since the transmitter received as input I and Q components, a Cartesian Loop is provided, by means of which a signal derived from the output of the amplifier by a probe (22) is split into de-modulated I and Q components in order to provide feedback signals at (27 and 28). Unfortunately the negative feedback cannot correct for errors in the feedback loop itself, and component imperfections introduce a DC offset into the feedback loop. The modulation is suppressed carrier but the DC offset re-introduces the carrier. The invention provides a method of reducing the DC offset to reduce the carrier breakthrough by injecting known DC offsets into the I and Q inputs and measuring the resulting ripple in the output of the power amplifier means corresponding to the re-introduced carrier, enabling calibration offset DC levels to be calculated and injected into inputs (27, 28).

Description

RF TRANSMITTER

This invention relates to RF transmitters.

The invention particularly relates to RF transmitters for generating suppressed carrier

RF signals from in-phase and quadrature baseband input signals.

To prevent energy being carried over into sidebands which would contravene

broadcasting regulations, one technique used is for the transmitter to employ linear

power amplifiers.

Accordingly, a feedback loop means is used so that the in-phase and quadrature input

signals are compared with signals derived from the output of the power amplifier

means, de-modulated using quadrature RF reference signals, to produce respective in-

phase and quadrature feedback signals, which are used to produce error signals, and

the error signals are modulated onto respective RF quadrature reference signals and

combined to produce the suppressed carrier RF signal for input to the power amplifier

means such that in the process the feedback is responsible for pre-distorting the input

to reduce non-linearities in the power amplifier means.

Unfortunately this can only correct for non-linearities in the power amplifier and

modulator means and not for errors in the feedback loop itself. It is found that such

errors are actually responsible for introducing DC offsets into the feedback loop which are then responsible for re-introducing a component of the carrier into the output.

It has been proposed to reduce this DC offset error in an iterative manner by making

a series of adjustments to the DC input level of the in-phase and quadrature input

signals and maintaining those which were responsible for reducing the amount of re-

introduced carrier. However, it was found that this procedure took a relatively long

time in order to be effective.

The invention provides an RF transmitter for generating a suppressed carrier RF signal

from in-phase and quadrature baseband input signals, comprising power amplifier

means, feedback loop means arranged so that the in-phase and quadrature input signals

are compared with respective signals demodulated using quadrature RF reference

signals from signals derived from the output of the power amplifier means, to produce

respective in-phase and quadrature error signals, means for modulating the error signals

onto respective quadrature RF reference signals and for combining them to produce a

suppressed carrier RF signal for input to the power amplifier means, the input thus

being pre-distorted to reduce non-linearities in the power amplifier means, and means

for applying a calibration DC component to the in-phase and quadrature input signals

based on measurements of signals derived from the output of the power amplifier

means when a known DC offset is injected into the in-phase and quadrature input

signals, to reduce carrier breakthrough.

The calibration can be performed relatively rapidly, because the pre-distorting DC component is calculated from the measurements made during a defined period.

The values can be measured by using a detector such as a diode detector to sense a

signal derived from the output of the power amplifier means in order to detect the

envelope amplitude variation corresponding to the transmitted output, and correlation

means or filtering means can be employed to quantify the amount of the amplitude

variation which relates to the carrier being re- introduced.

An RF transmitter constructed in accordance with the invention will now be described,

by way of example, with reference to the accompanying drawings, in which:

Figure 1 is a block diagram of a base station for private mobile radio system, in which

base station the transmitter is incorporated;

Figure 2 is a block diagram of the RF transmitter;

Figure 3 is a block diagram of a Cartesian Loop RF Stage of the transmitter;

Figure 4 is a phase diagram showing the possible phase changes of the transmitted

carrier;

Figure 5 is a spectral diagram showing the generation of a tone offset with respect to

the carrier frequency; Figure 6 is a vector diagram showing the vectors of the tone illustrated in Figure 5 and

a small component at carrier frequency;

Figure 7 is a spectral diagram showing the two vectors shown in Figure 6;

Figure 8 shows the construction of the detector 31 of Figure 3;

Figure 9 illustrates signals M₀. Mj. M₂, M₃, M₄ used to reduce the breakthrough of a

component at the carrier frequency;

Figure 10 is a phase diagram illustrating DC offsets applied to the I input of the

Cartesian Loop RF Stage;

Figure 11 is a phase diagram illustrating the addition of DC offsets to the Q input of

the Cartesian Loop RF Stage; and

Figure 12 illustrates the calibration process.

Referring to Figure 1, the RF transmitter forms part of a base station of a digital

private mobile radio network. The network typically consists of base stations, each

connected by land lines to switching centres, which interface with the national

telephone system. The users of the system employ mobile stations which communicate

with their nearest base station. Two mobile stations which are in the range of the same base station may communicate via that base station. Mobile stations in the range of

different base stations may communicate via those base stations and an intermediate

switching centre. In addition, mobile stations can, in certain circumstances,

communicate with other mobile stations. The system is suited to users such as the

emergency services and public transport. The messages are communicated in digital

form so that a range of facilities including voice, circuit mode data, short data

messages and packet mode services are possible.

Referring to Figure 1 , the base station consists of a site controller 1 , a digital signal

processor (DSP) 2, a transmitter output stage 3 and a receiver 4, together with

respective synthesisers 5 and 6. The site controller is the highest level of control

within the base station. Its function is to control the supervisory functions of the base station, including establishing, maintaining and terminating connections to the DSP 2,

which includes typically carrier processor cards. Each carrier processor card processes

signals for transmission and reception at one carrier frequency. A signal received from

a mobile station is demodulated in a carrier processor card and decoded at the receiver

4, whereupon it is retransmitted via a carrier processor card and I (in-phase) and Q

(quadrature) channels are output from the carrier processor card to the transmitter

output stage 3. A separate transmitter is associated with each carrier processor card

(only one transmitter is shown), and the respective synthesiser 5 generates the local

oscillator. Similarly there will be a separate receiver 4 for each carrier processor card,

and a respective synthesiser unit 6. The transmission is time division multiple access (TDM A), and there are four time

slots per carrier so that, for each carrier processor card, four communication channels

can be maintained. It is necessary, however, to reserve one communication channel

for control purposes.

The modulation performed in the DSP 2 of Figure 1 is differential quadrature phase

shift keying (DQPSK). What this means is that, for each consecutive pair of input data

bits (each dibit) to the modulator 7 of Figure 2. a change is made in the phase of the

transmitted carrier. Referring to Figure 4. which represents in simplified form the

possible phases of the transmitted carrier, the phase can be only one of the eight phase

angles shown ie. 0, ± π/4, + π/2, ± 3π/4, π . Further, each dibit can change the

transmitted carrier only by ± π/4, or ± 3π/4. The actual modulation of the I and Q

channels onto the carrier takes place in the Cartesian Loop RF Stage (the transmitter

output stage). Each dibit input into the modulator 7 produces an output on each of the

I and Q channels. If the position was (which it is not) that the I and Q channels

maintained a constant phase, it can be visualised how successive dibits could produce,

say, on the I channels, successive digital values as to represent a sinusoid and, on the

Q channel similar values but 90° out of phase. In fact, the modulation is differential

phase shift keying, which means that changes in phase of the transmitted carrier

represent the information bits. For example, if the input to the modulator consisted of

successive pairs of dibits each representing 1, 1, the values output by the modulator 7

on each channel corresponding to each input digit could be responsible for advancing

the phase of the I and the Q channel each by 45°. This would be equivalent to advancing by 45° along the sequence of digital values representing a sinusoid referred

to above. Because the filtering which takes place in the I and Q channels, the phase

changes linearly with time rather than moving in steps. Thus in reality each straight

line in Figure 4 would actually be curved: one curved line is shown as an example.

This linear phase change is manifested in a frequency shift from the original

unmodulated carrier frequency f_c. The frequency offset is proportional to the bit rate

of the incoming data string. The gross data rate in the DSP 2 is 36k bits per second,

which gives a rate for the dibits of 18k symbols per second. Since eight π/4 transitions

represents one cycle, the symbol rate results in a shift of the carrier frequency of

2.25kHz.

Referring to Figure 2, the samples being output from the modulator 7 are I and Q

baseband samples and these are fed into channel filters 8, 9 which are root raised

cosine filters implemented using a linear phase digital Finite Impulse Response (FIR)

filter. The digital samples are then converted into analogue signals by means of two

12 bit DACs 10, 11 for the I and Q channels. The analogue signals are then fed into

anti-aliasing filters 12, 13 to remove the sampling frequency component of 144kHz and

fed into the Cartesian Loop RF Stage 3.

The circuit elements 7-13 may be provided by dedicated hardware, but are more

conveniently carried out by the DSP 2.

The analogue signals entering the Cartesian Loop RF Stage on the I and Q channels are baseband signals, and are modulated on to a carrier in the Cartesian Loop RF Stage,

described in more detail now with reference to Figure 3. The modulation is double

sideband, suppressed carrier (DSB-SC). The I channel is modulated by an in-phase

local oscillator via mixer 14, and the Q channel is modulated by a quadrature local

oscillator via mixer 15. Quadrature phase shift device 16 produces the signals from an

input from the local oscillator from synthesiser 5 (Figure 1) operating at carrier

frequency (in this case 396.5MHz). The I and Q signals thus modulated onto

quadrature carriers are then combined at adder 17 and a DSB-SC waveform results.

Figure 5 shows the spectrum of the transmitted signal, and it will be noted that no

component is shown at the carrier frequency f_c.

The Cartesian Loop RF Stage includes a 25W class AB power amplifier illustrated by linear amplifiers 18 to 21 in cascade.

It is necessary to correct for non-linearities in the amplifier chain 18-21, and this is

done by means of the (negative feedback) Cartesian Loop.

A probe 22 picks off a part of the output signal and feeds it back to the input I and Q

channels. Of course, the signal picked off is modulated and so must be demodulated,

again using quadrature signals derived from the same local oscillator by means of

quadrature phase shift device 23 and mixers 24 and 25 which act on part of the

feedback signal which is split at splitter 26. The demodulated I and Q channels are fed back to nodes 27 and 28, where the feedback signals are subtracted from the input signals. If there is sufficient gain around the loop, the feedback signal will strive to

track closely the input signal, so that the signal leaving the nodes 27 and 28 and being

fed to the frequency changes 14 to 17 are error signals and the input to the amplifiers

29 and 30 behave as virtual earths.

It will thus be seen that the Cartesian Loop is a closed loop system where a part of the

transmitted signal is fed back and compared with the input to produce an error signal.

The error signal is then used to pre-distort the input to the power amplifier so that a

linear response is achieved.

The Cartesian Loop only cancels out distortions if the feedback loop is perfect.

Unfortunately, this is not the case, and in fact both the splitter 26 and the mixers 24,

25 introduce a small DC offset into the signal picked off from the RF output 22. The

DC offset is responsible for introducing into the transmitted output a small component

at the carrier frequency f_c (Figure 7), since the local oscillator mixed with DC produces

the carrier.

The result of this so-called carrier breakthrough can be considered by reference to

Figure 5, 6 and 7. It will be remembered that the large vector in Figure 7 is the tone

generated at the frequency offset by 2.25kHz from the carrier. The small vector is the

re-introduced carrier component. The carrier component is therefore modulated by a

much larger component at 2.25kHz. It is easier to visualise this by referring the

modulation to the tone, in which case the carrier modulates this tone at 2.25kHz. The tone is thus amplitude modulated at 2.25kHz.

Detector 31 picks off the feedback signal using probe 32 and detects the envelope

amplitude of the signal. The detector 31 may be configured like the diode detector

shown in Figure 8. The resistor R2 and capacitor Cl provide the necessary time

constant to detect amplitude variations in the envelope of the amplitude modulated

(AM) wave.

The carrier breakthrough causes a ripple at 2.25kHz, and a correlation is performed by

means not shown to detect the amplitude of that ripple.

The ripple detected is a combination of the I and Q channels, and may be represented

by a vector M₀ at 2.25kHz. In order to achieve cancellation of the DC offset

introduced in the negative feedback path, it is necessary to calculate this vector M₀.

The vector M₀ has been illustrated in Figures 10 and 11 and is a complex quantity.

The correlations at 2.25kHz are performed using quadrature waveforms to detect the

I and Q components of vector M₀. The vectors I_dc and Q^ must be calculated in order

to cancel the DC offset which brings vector M₀ into existence. They cannot be worked

out from a knowledge of M₀ alone, since they appear at different parts of the circuit

and the relation between them is not known.

The Applicants have previously attempted to reduce the vector M₀ in an iterative manner, by making changes of the DC level in the feedback path and then noting

whether M₀ was increased or reduced. Nevertheless, this is a very time-consuming

process.

In accordance with the invention, a calculation is made after five measurements in

order to obtain a measure of the ripple vector M₀. In each of five time slots, an initial

period is allowed to elapse before measurements are made to allow transients to decay.

Then the correlation is performed with quadrature 2.25kHz waveforms. In the first

measurement period the I and Q components of vector M₀ are calculated. In the second

measurement period ("Measure 1") a positive offset current is generated by DAC 33

and injected into the I input at node 27, and the ripple is measured to calculate vector

M_x. In the next measurement period ("Measure 2") a negative offset of the same value

is generated by DAC 33 and is subtracted from the I channel, and the corresponding

vector M₂ is measured. Since the positive DC current offset (+ Δ_dc) and the negative

DC current offset (- Δ_dc) can be plotted on the vector diagram 10 and are applied at

the same point in the circuit at which the error I_dc is to be cancelled, the unknown

quantity I_dc can now be calculated.

In the next measurement slot ("Measure 3"), a positive offset current is added to the Q

channel at node 28 by means of DAC 33 and vector M₃ representing the 2.25kHz

component is measured, and in the final measuring period ("Measure 4") a negative DC offset current is added to node 28 by means of DAC 33, and the ripple M₄ is measured.

In the same way, it is then possible to calculate Q_dc.

The calculation of I_dc and Q_dc is performed after "Measure 4" in the DSP 2.

DSP can now, via DAC 33 inject I_dc and Q_dc current offsets into respective nodes 27,

28, and it is found that this will reduce the 2.25kHz carrier breakthrough component

significantiy. It does not reduce it all in one go, and, but this can be reduced to a low

level in a matter of three of four such adjustments.

Of course the correction is advantageously performed on a continuous basis. The

telecommunication time slots last 14.2 μs, and the whole measurement and calculation

process illustrated in Figure 12 can take place in one time slot.

It will be appreciated that the injected currents ± Δ_dc are such as to increase and

decrease the DC level at the corresponding node ie. it could be that after injection of

+ Δ_dc, the DC level moves from one negative level to another, less negative DC level.

The correlation process is performed in DSP 2 and, to this end, an ADC 34 is

provided. Of course variations may be made without departing from the scope of the invention.

Thus, the invention is not restricted to base stations of private mobile radio networks,

and is also applicable to transmitters with Cartesian Loops in switching centres, or to

transmitters with Cartesian Loops used in base stations or switching centres of other

mobile radio systems or, indeed, more generally to any RF transmitter employing a

Cartesian Loop.

Claims

1. An RF transmitter for generating a suppressed carrier RF signal from in-phase

and quadrature baseband input signals, comprising power amplifier means, feedback

loop means arranged so that the in-phase and quadrature input signals are compared

with respective signals demodulated using quadrature RF reference signals from signals

derived from the output of the power amplifier means, to produce respective in-phase

and quadrature error signals, means for modulating the error signals onto respective

quadrature RF reference signals and for combining them to produce a suppressed

carrier RF signal for input to the power amplifier means, the input thus being pre-

distorted to reduce non-linearities in the power amplifier means, and means for

applying a calibration DC component to the in-phase and quadrature input signals based

on measurements of signals derived from the output of the power amplifier means when

a known DC offset is injected into the in-phase and quadrature input signals, to reduce

carrier breakthrough.

2. An RF transmitter as claimed in Claim 1, including means for applying a known

increase and a known decrease to the DC level of each of the in-phase and the

quadrature signals.

3. An RF transmitter as claimed in Claim 1 or Claim 2, in which the means for

applying a calibration DC component is arranged to base the values on measurements

when the known DC offset is injected as well as when no DC offset is being injected.

4. An RF transmitter as claimed in any one of claims 1 to 3. including detector

means for measuring the envelope amplitude of a signal derived from the output of the

power amplifier means.

5. An RF transmitter as claimed in Claim 4, including correlation means to

measure the amount of envelope amplitude modulation corresponding to the non-

suppressed carrier.

6. An RF transmitter substantially as herein described with reference to and as

shown in the accompanying drawings.

7. A base station of a mobile radio system which includes an RF transmitter as

claimed in any one of claims 1 to 6.

8. A private mobile radio network employing a base station as claimed in Claim

7.

9. A method of reducing carrier breakthrough in an RF transmitter for generating

a suppressed carrier RF signal from in-phase and quadrature baseband input signals,

comprising the steps of comparing the in-phase and quadrature input signals with

respective signals demodulated using quadrature RF reference signals from signals

derived from the output of power amplifier means, to produce respective in-phase and

quadrature error signals, modulating the error signals onto respective quadrature RF reference signals and combining them to produce a suppressed carrier RF signal input

to the power amplifier means, the input thus being pre-distorted to reduce non-

linearities in the power amplifier means, and applying a calibration DC component to

the in-phase and quadrature input signals based on measurements of signals derived

from the output of the power amplifier means when a known DC offset is injected into

the in-phase and quadrature input signals, to reduce carrier breakthrough.

10. A method as claimed in Claim 9, which includes the steps of injecting known

DC offsets into the in-phase and quadrature input signals, and measuring signals

derived from the output of the power amplifier means both when the offsets are

injected and when no offsets are injected, and calculating the calibration DC component

required to be applied to the in-phase and quadrature input signals to reduce carrier

breakthrough.

11. A method of reducing carrier breakthrough in an RF transmitter for generating

a suppressed carrier RF signal substantially as herein described with reference to the

accompanying drawings.