WO1998037671A1 - Pre-egaliseur adaptable pour utilisation dans un equipement de transmission de donnees - Google Patents

Pre-egaliseur adaptable pour utilisation dans un equipement de transmission de donnees Download PDF

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Publication number
WO1998037671A1
WO1998037671A1 PCT/US1997/002758 US9702758W WO9837671A1 WO 1998037671 A1 WO1998037671 A1 WO 1998037671A1 US 9702758 W US9702758 W US 9702758W WO 9837671 A1 WO9837671 A1 WO 9837671A1
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WO
WIPO (PCT)
Prior art keywords
signal
error
values
coefficients
equalizer
Prior art date
Application number
PCT/US1997/002758
Other languages
English (en)
Inventor
Hanan Herzberg
Ehud Langberg
Jin-Der Wang
Jean-Jacques Werner
Original Assignee
Globespan Semiconductor Inc.
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Globespan Semiconductor Inc. filed Critical Globespan Semiconductor Inc.
Priority to PCT/US1997/002758 priority Critical patent/WO1998037671A1/fr
Publication of WO1998037671A1 publication Critical patent/WO1998037671A1/fr

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/38—Synchronous or start-stop systems, e.g. for Baudot code
    • H04L25/40—Transmitting circuits; Receiving circuits
    • H04L25/49—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems
    • H04L25/497—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems by correlative coding, e.g. partial response coding or echo modulation coding transmitters and receivers for partial response systems
    • H04L25/4975—Correlative coding using Tomlinson precoding, Harashima precoding, Trellis precoding or GPRS
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03012—Arrangements for removing intersymbol interference operating in the time domain
    • H04L25/03019—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03343—Arrangements at the transmitter end
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/38—Synchronous or start-stop systems, e.g. for Baudot code
    • H04L25/40—Transmitting circuits; Receiving circuits
    • H04L25/49—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems
    • H04L25/497—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems by correlative coding, e.g. partial response coding or echo modulation coding transmitters and receivers for partial response systems
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/03777—Arrangements for removing intersymbol interference characterised by the signalling
    • H04L2025/03802—Signalling on the reverse channel
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/03777—Arrangements for removing intersymbol interference characterised by the signalling
    • H04L2025/03802—Signalling on the reverse channel
    • H04L2025/03808—Transmission of equaliser coefficients

Definitions

  • the present invention relates to data communications equipment, e.g., modems, and, more particularly, to the equalization of signals in a data communications system.
  • a receiver employs an adaptive decision feedback equalizer (DFE) to compensate for distortion introduced by the communications channel.
  • DFE adaptive decision feedback equalizer
  • the use of a DFE introduces "error propagation" effects in the receiver.
  • pre-equalization with modulo arithmetic e.g., Tomlinson filtering
  • This pre-equalizer uses equalizer coefficient values communicated from the receiver, typically over a reverse channel. These coefficient values are generated in the receiver as the result of an initialization phase, or training, between the far-end transmitter and the receiver.
  • the pre-equalizer will not be able to compensate for the error propagation problem in the receiver.
  • a re-train is required so that the receiver can generate a new set of pre-equalizer coefficients, which must be then sent back to the far-end transmitter.
  • each re-train takes time to both calculate the pre- equalizer coefficients anew and to communicate them back to the far-end transmitter over what is typically a low bandwidth reverse channel.
  • the foregoing error propagation problem is solved by using the communications channel to adapt a set of coefficients of a pre- equalizer of a transmitter.
  • a transmitter includes a pre- equalizer.
  • the latter adapts to changes in the communications channel by using an error signal that is communicated over a reverse channel by a corresponding receiver.
  • no re-trains are required and the error signal typically requires less bandwidth than a set of coefficient values.
  • FIG. 1 is a block diagram of a prior art DFE
  • FIG. 2 is a block diagram of a prior art precoder
  • FIG. 3 is an illustrative signal point constellation for use in the precoder of FIG. 2;
  • FIG. 4 is a block diagram of a communications system that embodies the principles of the invention;
  • FIG. 5 is a block diagram of a receiver, embodying the principles of the invention.
  • FIG. 6 is a block diagram of a transmitter embodying the principles of the invention.
  • FIG. 7 is an illustrative flow diagram for generating an adaptation signal in accordance with the principles of the invention.
  • FIG. 8 is another illustrative flow diagram for generating an adaptation signal in accordance with the principles of the invention.
  • FIG. 9 is another illustrative flow diagram for generating an adaptation signal in accordance with the principles of the invention.
  • FIG. 1 shows a prior art DFE that includes feedforward filter (FF) 50, sampler 53, adder 55, slicer 60, adder 80, and feedback filter (FB) 65.
  • feedforward filter (FF) 50 sampler 53, adder 55, slicer 60, adder 80, and feedback filter (FB) 65.
  • FF feedforward filter
  • FB feedback filter
  • a received data signal 49 is applied to feed forward filter 50 for processing.
  • Feedforward filter 50 whitens the noise present in the received data signal.
  • the output signal from feedforward filter 50 is applied, via sampler 53, to adder 55, which, theoretically, subtracts the inter- symbol interference (ISI) estimated by feedback filter 65 (described further below).
  • Adder 55 provides a signal, 56, to slicer 60.
  • the latter selects a particular data symbol as a function of the mapping of the signal, 56, into a predefined constellation of data symbols (not shown) to provide x(n), which is an estimate of a transmitted data symbol, x(n).
  • the signal x ⁇ n) typically represents a stream of data symbols occurring at a symbol rate of 1/T seconds and is provided for processing by feedback filter 65 and by other receiver circuitry (not shown) to recover the actually transmitted data.
  • Feedback filter 65 is a finite-impulse-response (FIR) having an impulse response represented by vector /( «).
  • FIR finite-impulse-response
  • Adaptation of feedback filter 65 is performed by using e ⁇ as an error signal, which is developed by adder 80. For illustration, it is assumed that a least-mean-square
  • the DFE structure of FIG. 1 is based on the assumption that x(n) is a good estimation of the transmitted da.tax(n). As long as this estimate of the transmitter symbol currently received is, in fact, correct, there is no problem. However, if the estimate of the currently transmitted symbol is wrong, then the feedback section adds this error to the next received symbol and error propagation occurs. As a result, as known in the art, a form of non-linear precoding is typically used in the far-end transmitter to minimize error propagation.
  • the DFE of the receiver In precoding there are two phases of receiver operation.
  • the "initialization” phase the DFE of the receiver, illustrated in FIG. 1 , adapts to a standard test signal, or training sequence, received from a transmitter (described below). This phase is also referred to in the art as a “start-up,” or “training” phase. Typically, there is no precoding of this test signal by the transmitter.
  • the transmitter now precodes the data before transmission using any of the well-known precoding techniques, e.g., Tomlinson precoding.
  • Tomlinson precoding An example of which is shown in FIG. 2.
  • a data signal is applied to a Tomlinson precoder comprising adder 605, mod-2L element 610, and filter 615.
  • Adder 605 subtracts a signal developed by filter 615, described below, from the data signal, x(n).
  • the output signal 606 of adder 605 is applied to mod-2L element 610, which performs as known in the art, to provide an output data symbol stream 611.
  • mod-2L element 610 maps the output signal 606 to a position in a signal point constellation. This mapping is performed using modulo 2L arithmetic, where L is the size of a signal point constellation.
  • the output data symbol stream 611 is applied to transmitter 620, which develops a signal for transmission.
  • the latter is transmitted from the corresponding receiver after the above-described training phase.
  • the precoding technique utilizes the above- mentioned coefficient values as determined during the initialization phase.
  • the receiver processes any received signal in a complementary fashion to remove the precoding, e.g., now incorporating a Tomlinson decoder. If the response of the communications channel remains constant for the transmission period, no further adaptation will be required since the precoding in the transmitter is equivalently performing the feedback function. As such, typically, the DFE section of the receiver is no longer used during the communications phase. However, in case of small changes in the response of the communications channel during the communications phase, a DFE feedback filter can be added to the receiver, initially set to zero.
  • FIG. 4 An illustrative communications system embodying the principles of the invention is shown in FIG. 4.
  • the communications system comprises data communications equipment (DCE) 11, communications channel 15, and DCE 21.
  • DCE data communications equipment
  • Transmitter 10 includes precoding and transmits a data signal to receiver 20, via communications channel 15, e.g., over primary channel 17.
  • Receiver 20 communicates an adaptation signal, in accordance with the principles of the invention, to transmitter 10 over reverse channel 16.
  • primary channel 17 and reverse channel 16 are shown as separate channels for simplicity, they are not so limited and represent any single, or plurality, of communications channels that enables transmission in both directions whether half-duplex, or full-duplex, over any number of different types of facilities (such as is found in the public-switched-telephone network).
  • reverse channel 16 can be a control channel that exists on a full-duplex primary communications link between transmitter 10 and receiver 20, thus enabling the inventive concept to also be practiced in the corresponding receiver (not shown) associated with DCE 1 1 and a transmitter (not shown), associated with DCE 21.
  • FIG. 5 is an illustrative block diagram of receiver 20 in accordance with the principles of the invention.
  • receiver 20 has been simplified to focus on the inventive concept, e.g., typically there is other receiver circuitry between feedforward filter 105 and communications channel 15.
  • FF feedforward filter
  • Tomlinson decoder/slicer 110 infinite size slicer 115
  • adder 120 processor 125 and transmitter 130.
  • a received data signal for processing is applied to feed forward filter 105, from primary channel 17.
  • Feedforward filter 105 whitens the noise present in the received data signal to generate the output signal y( ⁇ ).
  • the latter is applied to Tomlinson decoder/slicer 110, infinite size slicer 115, and adder 120.
  • Tomlinson decoder/slicer includes circuitry that performs in a complementary fashion to the Tomlinson precoder of transmitter 10 to provide an estimate, x ⁇ n ⁇ , of the actually transmitted data symbol, x(n).
  • y(n) is, ideally, ISI free
  • transmitter 10 since transmitter 10 incorporates precoding, transmitter 10 has an ideal reference for x(n) - namely x(n) itself. Therefore, and in accordance with the inventive concept, samples of y(n) contain sufficient information for adapting the pre-equalizer of transmitter 10 and can simply be communicated back to transmitter 10 over reverse channel 16 (ignoring for the moment adder 120 and infinite size slicer 115). For example, a few bits per sample o ⁇ y(n) can be transferred to transmitter 10 every AT time instants.
  • the convergence rate will be slower than that of an adaptive DFE directly located in the receiver.
  • This convergence rate can be increased to a degree by either increasing the data rate on the reverse channel (typically not an attractive systems option), or, where the data rate for the reverse channel is fixed, by reducing the number of bits required for each sample oiy(n).
  • an error signal is developed for transmission from DCE 21 to DCE 11 such that the number of bits required for representing the error signal is less than the number of bits required ⁇ o ⁇ y(n).
  • adder 120 of receiver 20 develops an approximation (denoted by e(n)) of the error signal e(n) by using an estimation of the transmitted data, x ⁇ ), which is developed by infinite size slicer 115.
  • the latter is required since a form of modulo precoding is used.
  • the estimate x' ⁇ n) developed by Tomlinson decoder/slicer 110 may generate a large error at the boundary of the signal point constellation due to the modulo nature of the precoding
  • the received signal point may be on one side of the constellation but the sliced signal point is on the opposite side, which would yield a large error value. Therefore, infinite size slicer 115 is configured to mathematically represent an infinite signal point constellation.
  • FIG. 6 is an illustrative block diagram of a portion of DCE 11 in accordance with the principles of the invention. FIG. 6 is similar to FIG. 2 described above except for the addition of receiver 230 and processor 225.
  • FIG. 7 shows a generalized method in accordance with the principles of the invention as described above.
  • processor 125 of DCE 21 generates an adaptation signal in step 305 and transfers this signal to DCE 11 via reverse channel 16 in step 310.
  • processor 125 of DCE 21 generates the sign of e(«)in step 405 and transfers the sign of e( «) to DCE 11 via reverse channel 16 in step 410.
  • the value of the sign of e ⁇ ) is based upon one sample per data block of length K.
  • Processor 225 of DCE 11 recovers the sign of e( «)from reverse channel 16 in step 415 and using any well-known sign algorithm calculates the changes to the coefficient values in step 420.
  • ign algorithms are known in the art. For example, see V.J. Mathews and S.H.
  • processor 125 statistically processes ei ) to generate at least one statistical parameter in step 505 and transfers the statistical parameter(s) to DCE 11 via reverse channel 16 in step 510.
  • This method is aimed at utilizing the data available in receiver 20 for the K samples (whereas the adaptation of the sign algorithm is based on one sample per data block of length JK , and avoiding the need for synchronization between ein) and x(n - i).
  • ⁇ be the variance of the error e(n). Assuming the e(n) is ergodic, ⁇ can be evaluated by:
  • V , -- 5; e( ,i ) x (» - - (5)
  • step 505 hereafter referred to as gradient estimation
  • the variance of the minimum error in the estimation of the gradient is reduced by a factor ⁇ K.
  • the misadjustment is proportional to this variance.
  • the adaptation step size can be increased for obtaining a desired misadjustment, provided that the increased step size would ensure convergence.
  • Increasing the step size would also accelerate the convergence of the pre-equalizer.
  • gradient estimation is mostly effective for slow varying channels, where the channel is quasi- stationary for a period of K transmitted symbols.
  • a set of values of V, for each coefficient is calculated by processor 125 in step 505 and then transmitted to DCE 11 in step 510.
  • a disadvantage of the above-described gradient estimation method is the need to transfer through the reverse channel different information for each coefficient, whereas in the sign algorithm only one bit is required for the adaptation of all the coefficients.
  • the gradient estimation method uses more bandwidth than the above-mentioned use of a sign algorithm.
  • the gradient estimation method can be further modified so as to reduce the bandwidth required over the reverse channel. For example, requirements of the reverse channel data rate can be reduced by transferring over the reverse channel only a few bits per coefficient for representing V, . Indeed, even one bit may be considered (due to the accuracy of the estimation which is based on an average error).
  • Yet another variation is to only adapt a few dominant coefficients of the pre- equalizer in the transmitter. Recall that the aim of performing adaptation of the pre- equalizer is eliminating error propagation, mostly caused by a few ISI coefficients. The remaining coefficients can be used to adapt a DFE in the receiver during the communications phase.
  • the mean square error, E ⁇ e (n) ⁇ is small after the initialization process.
  • the mean squared error may serve as a figure-of-merit for the adaptation step size.
  • Increase in the mean squared error may be interpreted as an increase in the misadjustment.
  • the pre-equalizer can increase the step size (using the reverse channel) for a short period in order to accelerate the adaptation.
  • the adaptation signal can take many forms and only a few illustrative suggestions were described above.
  • the adaptation signal can represent a sequence of k-bit size words, where each k-bit size word represents the location of the first non-zero bit in a corresponding value of, e.g., e(«), as opposed to the value ofe( «) itself. This approach, in effect, sends the most significant, non-zero bit(s) of the signal used for adaptation.
  • any precoding scheme can be used in conjunction with the inventive concept.
  • the precoding specified by CCITT modulation standard V.34 could also be used with correspondingly straightforward changes in the receiver structure.
  • This proposed scheme can be used in either an uncoded or coded communications system.
  • any one or more of those building blocks can be carried out using one or more appropriate programmed processors, e.g., the above-described pre-equalizer and processor of transmitter 10 can be implemented together in a suitably programmed digital signal processor.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Physics & Mathematics (AREA)
  • Spectroscopy & Molecular Physics (AREA)
  • Power Engineering (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)

Abstract

L'invention vise à résoudre le problème de la propagation d'erreurs, au moyen d'une voie de communication (15) servant à adapter un pré-égaliseur d'un émetteur (10) aux modifications de réponse de la voie de communication (15). En particulier, le pré-égaliseur s'adapte aux modifications dans la voie de communication (15), par traitement d'un signal d'erreur qui est transmis sur une voie de retour (16) par un récepteur correspondant (20).
PCT/US1997/002758 1997-02-25 1997-02-25 Pre-egaliseur adaptable pour utilisation dans un equipement de transmission de donnees WO1998037671A1 (fr)

Priority Applications (1)

Application Number Priority Date Filing Date Title
PCT/US1997/002758 WO1998037671A1 (fr) 1997-02-25 1997-02-25 Pre-egaliseur adaptable pour utilisation dans un equipement de transmission de donnees

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
PCT/US1997/002758 WO1998037671A1 (fr) 1997-02-25 1997-02-25 Pre-egaliseur adaptable pour utilisation dans un equipement de transmission de donnees

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Cited By (11)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
GB2376391A (en) * 2001-06-07 2002-12-11 Cambridge Broadband Ltd Method for setting up a precoder in a subscriber unit of a wireless transmission system
WO2003094460A2 (fr) * 2002-04-30 2003-11-13 Ericsson Inc. Traitement des signaux de retour de stations mobiles
EP1492289A2 (fr) * 2003-06-25 2004-12-29 Texas Instruments Incorporated Egalisation à décision récursive pour des liaisons série à haute vitesse
US6996375B2 (en) 2001-07-26 2006-02-07 Ericsson Inc. Transmit diversity and separating multiple loopback signals
US6996380B2 (en) 2001-07-26 2006-02-07 Ericsson Inc. Communication system employing transmit macro-diversity
WO2006073326A1 (fr) * 2004-12-30 2006-07-13 Intel Corporation Precodage de reponses prescrites pour des canaux avec brouillage intersymbole
US7209511B2 (en) 2001-08-31 2007-04-24 Ericsson Inc. Interference cancellation in a CDMA receiving system
US7224942B2 (en) 2001-07-26 2007-05-29 Telefonaktiebolaget Lm Ericsson (Publ) Communications system employing non-polluting pilot codes
US8036305B2 (en) 2005-09-30 2011-10-11 Intel Corporation Precoder design for different channel lengths
US8681849B2 (en) 2005-06-29 2014-03-25 Intel Corporation Precoder construction and equalization
EP2259519B1 (fr) * 2003-04-09 2015-06-03 Rambus Inc. Récepteur à réponse partielle

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US4577329A (en) * 1982-10-11 1986-03-18 Telecommunications Radioelectriques Et Telephoniques T.R.T. Self-adaptive equalizer for baseband data signals
US4866736A (en) * 1987-06-09 1989-09-12 U.S. Philips Corporation Data transmission system comprising a decision feedback equalizer and using partial-response techniques
US5008903A (en) * 1989-05-25 1991-04-16 A.T. & T. Paradyne Adaptive transmit pre-emphasis for digital modem computed from noise spectrum
US5251328A (en) * 1990-12-20 1993-10-05 At&T Bell Laboratories Predistortion technique for communications systems
US5291520A (en) * 1991-02-06 1994-03-01 General Datacomm, Inc. Methods and apparatus employing distribution preserving Tomlinson precoding in transmission of digital data signals

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Publication number Priority date Publication date Assignee Title
US4577329A (en) * 1982-10-11 1986-03-18 Telecommunications Radioelectriques Et Telephoniques T.R.T. Self-adaptive equalizer for baseband data signals
US4866736A (en) * 1987-06-09 1989-09-12 U.S. Philips Corporation Data transmission system comprising a decision feedback equalizer and using partial-response techniques
US5008903A (en) * 1989-05-25 1991-04-16 A.T. & T. Paradyne Adaptive transmit pre-emphasis for digital modem computed from noise spectrum
US5251328A (en) * 1990-12-20 1993-10-05 At&T Bell Laboratories Predistortion technique for communications systems
US5291520A (en) * 1991-02-06 1994-03-01 General Datacomm, Inc. Methods and apparatus employing distribution preserving Tomlinson precoding in transmission of digital data signals

Cited By (19)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
GB2376391B (en) * 2001-06-07 2004-05-05 Cambridge Broadband Ltd Wireless transmission system and method
GB2376391A (en) * 2001-06-07 2002-12-11 Cambridge Broadband Ltd Method for setting up a precoder in a subscriber unit of a wireless transmission system
US6996380B2 (en) 2001-07-26 2006-02-07 Ericsson Inc. Communication system employing transmit macro-diversity
US7224942B2 (en) 2001-07-26 2007-05-29 Telefonaktiebolaget Lm Ericsson (Publ) Communications system employing non-polluting pilot codes
US7197282B2 (en) 2001-07-26 2007-03-27 Ericsson Inc. Mobile station loop-back signal processing
US6996375B2 (en) 2001-07-26 2006-02-07 Ericsson Inc. Transmit diversity and separating multiple loopback signals
US7209511B2 (en) 2001-08-31 2007-04-24 Ericsson Inc. Interference cancellation in a CDMA receiving system
GB2404824B (en) * 2002-04-30 2006-03-01 Ericsson Inc Mobile station loop-back signal processing
GB2404824A (en) * 2002-04-30 2005-02-09 Ericsson Inc Mobile station loop-back signal processing
WO2003094460A3 (fr) * 2002-04-30 2004-02-05 Ericsson Inc Traitement des signaux de retour de stations mobiles
WO2003094460A2 (fr) * 2002-04-30 2003-11-13 Ericsson Inc. Traitement des signaux de retour de stations mobiles
EP2259519B1 (fr) * 2003-04-09 2015-06-03 Rambus Inc. Récepteur à réponse partielle
EP1492289A3 (fr) * 2003-06-25 2006-12-20 Texas Instruments Incorporated Egalisation à décision récursive pour des liaisons série à haute vitesse
EP1492289A2 (fr) * 2003-06-25 2004-12-29 Texas Instruments Incorporated Egalisation à décision récursive pour des liaisons série à haute vitesse
WO2006073326A1 (fr) * 2004-12-30 2006-07-13 Intel Corporation Precodage de reponses prescrites pour des canaux avec brouillage intersymbole
JP2008527790A (ja) * 2004-12-30 2008-07-24 インテル・コーポレーション シンボル間干渉があるチャネルのための所定の応答プリコーディング
US7986744B2 (en) 2004-12-30 2011-07-26 Intel Corporation Prescribed response precoding for channels with intersymbol interference
US8681849B2 (en) 2005-06-29 2014-03-25 Intel Corporation Precoder construction and equalization
US8036305B2 (en) 2005-09-30 2011-10-11 Intel Corporation Precoder design for different channel lengths

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