WO1995029498A1

WO1995029498A1 - Dc actuator control circuit with voltage source sag compensation and fast dropout period

Info

Publication number: WO1995029498A1
Application number: PCT/US1995/005152
Authority: WO
Inventors: G. Stephen Perreira
Original assignee: Kilovac Corporation
Priority date: 1994-04-26
Filing date: 1995-04-26
Publication date: 1995-11-02

Abstract

An improved economizer for DC actuator coil current control in spring-biased DC actuators is accomplished by control of power from a voltage source Vs to an actuator coil circuit by a chopper circuit (104) incorporating a switch S1 in the coil circuit which is closed responsive to a control signal generated by a power switching circuit. The power switching circuit incorporates a threshold detector VTH providing a triggering signal to initiate operation of the actuator. The control circuit is responsive to a predetermined source voltage, a gate signal generator VGATE responsive to the triggering signal and having a time constant sufficient to drive the coil through the pick-up interval, and a variable duty cycle oscillator (106) for generating the switch control signal.

Description

DC ACTUATOR CONTROL CIRCUIT WITH VOLTAGE SOURCE SAG COMPENSATION AND FAST DROPOUT PERIOD

Background of the Invention

A. Field of the Invention

The present invention is directed generally to the field of economizers for spring- biased DC actuators. In particular, the present invention provides a control circuit for power reduction using an electronic chopper incorporating an oscillator having a variable duty cycle responsive to the source voltage to maintain average coil holding current at a level to assure minimum required magnetomotive force (MMF) for the actuator coil. Additionally, the invention includes an electronic dropout switch providing high impedance for rapid current decay upon deactivation of the actuator.

B. Prior Art

Spring-biased DC actuators are employed for numerous applications including high power contactors requiring fail-safe capability. Such actuators typically require significant MMF to overcome armature and contactor inertia and spring force for closing the contactor. However, once closed, the contactor may be retained in a closed position at significantly lower MMF. Various prior art methods have been applied to provide initial high currents to the actuator coil to obtain contact closure with subsequent reduction of the current to maintain MMF at a level required to resist the spring force and maintain the contact in the closed position. To assure minimum MMF in conditions of high temperature, wherein coil resistance may be significantly altered, and power changes due to voltage sagging in the power supply, most designs require a significant safety factor to be applied to the current level maintained in the coil, precluding optimizing of performance. Prior art systems employing chopper circuits to reduce average current in the coil must provide sufficient MMF to accommodate worst-case conditions. Higher currents which need to be maintained exacerbate the situation by contributing to heating of the coil.

In addition, the use of most prior art choppers as an economizer circuit results in difficulties during de-actuation of the circuit. Addition of a chopper controller normally increases dropout time due to slow decay of coil current through the low-impedance return circuit employed in chopper economizers. Regeneration in the coil created by motion of the armature while current is still flowing in the coil impacts the velocity and kinetic energy of the armature, thereby extending the dropout time and weakening the action that breaks open sticky contacts. The present invention overcomes the difficulties in the prior art to allow high efficiency in conservation of coil power, enhancement of mechanical performance, and potential reduction in size of actuators for given applications.

Summary of the Invention The present invention provides an improved economizer for DC actuator coil current control in spring-biased DC actuators. The actuator coil circuit receiving power from a voltage source is controlled by a chopper circuit which incorporates a switch in the coil circuit which is closed responsive to a control signal generated by a power-switching circuit. The power-switching circuit incorporates a threshold detector which provides a triggering signal to initiate operation of the actuator and the control circuit responsive to a predetermined source voltage, a gate signal generator responsive to the triggering signal and having a time constant sufficient to drive the coil through a pickup interval, and a variable duty cycle oscillator for generating the switch control signal. The oscillator passes the gate signal during the pickup interval and subsequently varies its duty cycle responsive to the voltage of the voltage source.

Modification of duty cycle for the oscillator is accomplished in a first embodiment under voltage control and in a second embodiment under current control, both operated in response to the power source for the coil. The change in the duty cycle increases or decreases the period in which the switch for control of the coil circuit remains on, providing energy pulses to sustain average coil current at the desired MMF. The ability to compensate for voltage sag and resultant current decrease in the coil reduces the safety factor necessary to assure minimum MMF to adequately resist spring force. Environmental temperature variations become the only consideration for safety margin in the voltage controlled embodiments while the current controlled embodiment compensates for temperature variation as well.

The present invention also incorporates a dropout switch for opening the sustaining current return circuit for the coil, which operates in concert with the power switching circuit to maintain coil current. The dropout switch is opened by removal of source voltage, producing a high impedance for rapid current drain from the coil, thereby eliminating the regeneration effect typically present in economizer circuits.

Brief Description of the Drawings

The present invention will be more clearly understood in reference to the following drawings and detailed description.

FIG. 1 is a block diagram of a first embodiment of the invention employing a voltage- controlled oscillator;

FIG. 2 is a circuit schematic for the embodiment of FIG. 1; FIG. 3 is a graph demonstrating armature and contact position in relation to coil current controlled by the embodiment of the invention disclosed in FIG. 2;

FIGS. 4a, 4b and 4c graphically depict variation of the duty cycle for the oscillator of the embodiment shown in FIG. 2 based on a reduction in source voltage;

FIG. 5 is a block diagram of a second embodiment of the invention employing a current-controlled oscillator;

FIG. 6 is an electrical circuit schematic of the embodiment shown in FIG. 5; FIG. 7 is a graph demonstrating position of the armature and contacts in relation to coil current for the embodiment of the invention shown in FIG. 6;

FIGS. 8a, 8b and 8c graphically depict the duty cycle for the embodiment of the invention shown in FIG. 6 based on a reduction in source voltage;

FIGS. 9a and 9b graphically represent armature position and the effects of regeneration based on residual current in the coil of an economizer system without the dropout circuit of the present invention;

FIGS. 10a and 10b graphically represent the armature position during deactivation of the actuator with the dropout circuit of the present invention;

FIG. 11 is an electrical circuit schematic for an embodiment of the invention substantially similar to FIG. 2 employing MOSFET switches and demonstrating component values for a typical application.

Detailed Description

Referring to the drawings, FIG. 1 shows a first embodiment of the invention in block diagram form. As exemplary of the application of the invention, a high-power contactor 100 constitutes the application in which the invention is used. High-power contactors require rapid armature acceleration for proper contact closure and rapid drop-out of the contactor to preclude arcing and to break open welded contacts. Significant armature power is required to minimize contact bounce to limit signal spikes and minimize weld strength caused by arcing during contact bounce during closure under electric load, and again preclude contactor damage by arcing. In FIG. 1, the contactor 100 is driven by coil 102 which is schematically represented by inductor L_c representing a 1600-turn coil having an inductance of approximately 0.2 H and a resistance R_c of 21 ohms. Power is supplied by a voltage source V_s which, in the applications shown, comprises a 28 VDC source, and opening and closing the contactor is accomplished by switch S₀. The present invention is embodied in the electronic controller 104 which comprises a chopper for reducing power from the voltage source to the coil. A switch S, opens the coil circuit, and sustaining current for the coil flows through a sustaining circuit incorporating diode D₅.

Switch S_j is controlled by an oscillator 106, a threshold voltage amplifier

and a gate amplifier V_GATE. The threshold voltage amplifier or threshold voltage detector determines if the source V_s provides sufficient potential for pick-up to occur in the coil. Once the voltage threshold has been exceeded, the gate amplifier provides a gate signal which closes switch S< allowing the coil to receive full source voltage during the pick-up interval. After pick-up, the oscillator 106 begins chopping the source current by periodically opening switch Sj, allowing the holding current to decay to a steady state level based on the duty cycle of the oscillator. The oscillator is voltage-controlled to compensate for source voltage variations. Specifically, the duty cycle is increased when voltage sag occurs in the power supply, thus maintaining essentially constant average current in the coil. A drop-out switch S is provided in the current return to accommodate the second feature of the invention, rapid drop-out. Switch S₂ is maintained in the closed position by the source voltage and, when source voltage is removed by opening of switch S₀, switch S₂ opens which substantially increases the impedance in the coil circuit allowing rapid current drain.

FIG. 2 provides a detailed schematic for the embodiment of the invention shown in FIG. 1. FIG. 3 graphically demonstrates the coil current for the circuit of FIG. 2 and corresponding armature and contact position. When source voltage is applied through switch S₀, the threshold detector amplifier A_t establishes a reference voltage at the inverting input through resistor R., and zener diode Dj. Initially, source voltage is less than the breakdown voltage of D,, hence D< is a high impedance and the source voltage is initially impressed at the inverting input of A^ At the noninverting input of A,, the source voltage is divided by resistors R₂, R₃ and R , impressing a lower voltage. Therefore, the inverting input prevails, maintaining the output of A_! in an initial low condition. The low state of the output of amplifier A, prevents the gate generator, which comprises amplifier A₂, from producing an interval gate while the source voltage is too low to provide adequate pick-up MMF. When the source voltage exceeds the zener reference voltage for D₁₍ a constant voltage V_j^p is provided to the inverting input of the threshold detector amplifier A_j. Rising source voltage above V^_EF increases the voltage at the noninverting input for Aj. Resistors R₂, R₃ and R₄ are calculated with reference to diode D_j such that the noninverting input for A_j exceeds the reference voltage when the source voltage reaches a desired pick-up voltage, nominally 23 volts DC for the 28 volts DC source shown in the embodiment in the drawings. When the noninverting input has sufficient voltage to switch amplifier A_j, the output of A_j goes high and is instantly pulled up to the source voltage through resistor R₅. In the embodiment shown in the drawings, substantial positive feedback through resistors R₅ and R₄ to the noninverting input of amplifier A_j introduces significant hysteresis in the threshold detector cycle. Consequently, the source voltage must be reduced to a voltage much lower than the pick-up voltage in order to reset the threshold detector.

When the threshold detector voltage is pulled up to source voltage, the output of amplifier A, provides a trigger for the monostable gate generator amplifier A₂. A₂ single shots a rectangular output which drives through the inverting input of amplifier A₃, which comprises the oscillator as will be described in greater detail subsequently, closing switch S_j of FIG. 1 which is embodied in the circuit of FIG. 2 as a darlington transistor Q The time duration for the gate pulse is determined by charging of capacitor C_j through resistor R₇, resulting in a decaying voltage impressed at the noninverting input to gate generator A₂. The inverting input of A₂ allows the darlington switch Q_j to be biased on through resistors R and R₁₀, initiating the pick-up cycle of the contactor. During the gate signal, the output of A₂ is low providing a sink preventing charging of capacitor C₃ to maintain the output of amplifier A₃ in a high condition driving switch Q_i through resistors R₁₇ and R₁₈. The gate generator interval ends when voltage developed across Cj exceeds the voltage developed at the inverting inputs of A₂ via the source V_s.

The voltage controlled oscillator 106 of FIG. 1 comprises amplifier A₃ and its associated circuitry in FIG. 2. The oscillator is free running or self-sustaining.

With the oscillator output high and the inverting input low, the noninverting input of A₃ is switched to high as an initial condition at t„ as shown in FIG. 4b. With the output of A₃ high, capacitor C₃ is charged through resistors R_ϊ4 and R₁₅. The voltage at the noninverting input to A₃ increases exponentially toward the voltage value of the output of A₃. The resulting cycle is shown in FIG. 4b, with the noninverting input signal V+ and the inverting input signal V- shown in relation to the reference voltage V_REF. Rise time for the inverting input is governed by the time constant T_RJSE which is equal to (R₁₄//R₁₅)C₃. The asymptote for the exponential increase of the inverting input is the output voltage for amplifier A₃, Vosc.

At time t_v, the inverting input exceeds the noninverting input, driving the oscillator output low which completes an on-cycle for darlington switch Q₁ With darlington Q_j switched off, sustaining current for the coil flows through the free-wheeling diode D₅. Substantial positive feedback for the oscillator amplifier A₃ insures rapid transition in driving the noninverting input V- low. With the oscillator output signal V_osc low, capacitor C₃ begins discharging into the low potential sink of the output of A₃. For the embodiment shown in the drawings, discharge of C₃ is accomplished through resistor R₁₅ only due to diode D₇ blocking the path through resistor R₁₄. The discharge rate for capacitor Cg is, therefore, different than the charging rate with diode D₇ forming a pass/blocking circuit regulating the duty cycle of the oscillator. At time t-w, the inverting input signal V-, decaying exponentially toward the ground potential asymptote with a time-constant τ_FALL governed by (R₁₅)C₃, is discharged below the noninverting input signal V+, the oscillator is again driven into a high state and the cycle is repeated. The oscillating frequency is entirely determined by selection of resistors R_H, R₁₅ and capacitor C₃.

As previously discussed, the duty cycle of the oscillator in the present circuit is voltage controlled. Referring to FIG. 4a, if the source voltage V_s drops from 28 volts to 16 volts, the behavior of the oscillator can be shown in FIG. 4b. Reduction in the source voltage results in a reduction in the oscillator output voltage V_osc, affecting the upper asymptote for the charging cycle of the inverting input signal V-. T_RJSE has a significantly lower slope, increasing the time that the oscillator remains on, hence maintaining darlington Q_j on, providing current from the source to the coil. Since the ground potential provides the lower asymptote for the discharging cycle of the inverting input signal V- capacitor C^ discharges through resistor R₁₅ starting from a lower potential, and since the low state of the noninverting input signal V+ is not shifted down as wholly determined by the constant voltage V_RE , the inverting input signal V- reaches the noninverting signal V+ in a shorter time. The cooperation of these two factors increases the duty cycle of the amplifier. This results in a fairly constant average voltage to the coil over a range of 36 volts to 14 volts for a nominal 28 volt condition with the embodiment shown in the drawings. The relative oscillator output signal V_osc is shown in FIG. 4c for both the nominal condition of source voltage at 28 volts and the sag condition of 16 volts as shown in FIGS. 4a and 4b described previously.

The drop-out feature of the present invention is provided by darlington transistor Q₂ which acts as switch S₂ of FIG. 1. When source voltage is applied to the circuit, switch k is fully on, allowing operation of the circuit in a powered mode through switch Qi and in a current return mode through free-wheeling diode D₅. Voltage from the source divided through resistors R₂₀ and R₂₁ provides the current bias to turn on switch Q₂. Capacitor C₄ provides capacity for sustaining Q₂ on which Q₁ is in the off interval of the oscillator cycle. The desired fast drop-out to rapidly drain current from the coil when the actuator is turned off is accomplished by the drop-out switch Q₂. Removal of source power causes Q₂ to open, breaking the low impedance current-return circuit. In the embodiment shown in the drawings, reverse-biased zener diode D₄ is then in the coil discharge loop in series with free- wheeling diode D₅. The dynamic resistance of diode D₄ immediately and substantially raises the discharge time constant of the coil through D₄. This results in rapid coil-current decay. The operation of the present circuit without and with the fast drop-out switch is shown in FIGS. 9a and 9b and 10a and 10b, respectively. Without the rapid drop-out capability, the current in the coil decays slowly through free-wheeling diode D₅, as shown by curve 900. The armature position is shown in curve 910, with the contact state shown in curve 920. As the armature begins to move, the remaining current in the coil causes regeneration in the coil due to the continued magnetization of the armature. This in turn results in a reduction of velocity for the armature, as is clearly seen in FIG. 9b wherein the slope of curve 910 changes as the regeneration bump occurs. This slowing of velocity, which occurs substantially simultaneously with opening of the contacts, impedes rapid displacement of the contacts from the closed position, thereby increasing the opportunity for arcing as the contacts open and increasing the probability that welded contacts will not be opened by the less energetic armature action.

With the drop-out switch in place, removing source power results in an immediate decay of the coil current, as shown in FIG. 10a, with coil current represented by signal 1000 which immediately drops to zero. Armature position, shown by curve 1010, has significantly less velocity change since no regeneration occurs due to residual coil current. As shown in an expanded scale in FIG. 10b, the armature attains a full open position significantly sooner than shown in FIGS. 9a and 9b. The overall operation of the present invention is shown in FIG. 3 in a time profile comparing coil current, armature position and contact position. The coil current is shown as a function of time by trace 300, while the armature position is shown by trace 310 and contact position is shown by trace 320. The initial power pull-in interval shown in FIG. 3 includes the pick-up time t_PICKUP during which the gate signal amplifier provides the gate signal initiating power to the coil for closing of the armature. The time for initiation of operation of the oscillator to begin economizing by chopping the coil current is shown as t_delta. During the pick-up interval, the MMF for the coil builds to a total 1920 AT with a coil current of 1.2 amps and a total power supplied of 34 watts. When t_delta is reached and the economizer circuit is engaged, the chopped source voltage is readily apparent in the saw- tooth shape of the coil-current signal. In a few coil time constants, the holding current decays to a steady state holding level, resulting in an MMF of 610 AT with a coil power dissipation of only 3.4 watts. The drop-out portion of the cycle after removal of the source voltage from the coil is shown corresponding to FIGS. 9a and 9b for the case without the drop-out switch of the present invention, identified as corresponding curves 900 and 910.

A second embodiment for the invention is shown in FIG. 5 in block diagram form. The basic conception of the invention is the same, however, the voltage-controlled oscillator is replaced with a current-controlled oscillator 106'. A detailed schematic for the current-controlled oscillator controller is shown in FIG. 6. Operation of the threshold detector and gate detector are substantially identical to that previously described for the voltage-controlled chopper, however, the circuit arrangements have been inverted to accommodate a PNP darlington Q_r for switch S_j. Overall concept of operation for the current-controlled circuit is similar to that disclosed for the voltage-controlled circuit, however, the oscillator amplifier A₃ reacts to small voltage fluctuations in a sense resistor R_s connected to the noninverting input of the amplifier. The coil current rises and falls with power supplied and cut off from the voltage source through the darlington switch Q_j which is in turn controlled by the oscillator. Hence, since resistor R_s is in series with the coil and has the same current flow, the voltage signal across R_s is exactly proportional to the coil current and a closed loop for current control is provided. The inverting input for amplifier A₃ maintains an exact reference derived by voltage divider from the reference voltage V ^ . The noninverting input of amplifier A₃ tracks, with the exception of a small amount of positive feedback, the voltage developed across the current-sense resistor R_s. Without current control, the voltage across resistor R_s would rise in several coil time constants to a value of V_S(R_S/)R_S+R_C)) where R_c is the coil resistance. Setting the reference voltage applied to the inverting input of amplifier A₃ anywhere below the maximum value achievable by the signal provided to the noninverting input will force current control because the oscillator will begin switching the voltage from the source through darlington Q_j, thereby limiting coil current. The current limit realized is dependent on the duty cycle of the oscillator and the duty cycle is defined by resistors R₁₄, R_n, R_s and the coil resistance R_c.

A calculation of the duty cycle is shown in equation 1.

Duty Cycle = Vref (R14)(Rs+Rc) EQ 1 Vs(R14+Rll)(Rs)

From equation 1, it is clear that the duty cycle increases as the coil resistance Rc increases and decreases as the source voltage V_s increases, both representing compensation in the correct direction. Operation of the current-controlled oscillator is best described with regard to FIGS. 8a, 8b and 8c. If the noninverting input signal V+ is lower than the inverting input signal V-, as indicated at time t_υ in FIG. 8b, the output of amplifier A₃ is low, resulting in closure of switch Q,. Coil current begins to rise, resulting in a change in V+ sensed from resistor R_s. Subsequently, V+ rises exponentially with coil current toward the upper asymptote. At time t_v, the noninverting input signal V- exceeds the inverting signal input V-, triggering amplifier A₃ to shut off, pulling up the output signal V_osc through resistor R₁₇. The speed of the switching action is increased by positive feedback leading through capacitor C₅ and sustained through resistor R₁₆. The sustained feedback shifts the noninverting input signal V+ above the inverting input signal V- by the ratio [V_s][Rι₅/(R₁₅+R₁₆)]. where R₁₆ > > R₁₅. The output of the amplifier V_osc being pulled high shuts off transistor Q_{l 5} opening the switch which ends the to_N interval. At time t_v, the coil current begins to decay through diode D₅. The coil current decays according to its time constant toward the lower asymptote, and the noninverting input of amplifier A₃ receives the analog of this signal through resistor R₁₅ as sensed from resistor R_s. At time t , the noninverting input signal V+ recedes below the inverting input signal V- causing amplifier A₃ to go low (V_osc low), again turning on the darlington Q, , closing the switch and ending the to interval. Pulling the output of the amplifier low is again aided by leading the positive feedback through capacitor C₅ sustained through resistor R₁₆. The present invention, as disclosed in this embodiment, maintains constant coil current independent of source voltage or temperature variations because feedback is realized as a direct analog of coil current. The operating frequency of the oscillator, as well as its duty cycle, varies with changing conditions in order to maintain constant average current in the coil. Coil current fluctuates somewhat as switching transistor Q, cycles on and off. Effects of varying source voltage from nominal 28 volts to 16 volts, as shown in FIG. 8a, is reflected by the changing duty cycle of the oscillator in FIG. 8b. Output signal from the oscillator Vos_C is shown in FIG. 8c, which clarifies the changing duty cycle. Note that in the PNP arrangement for darlington Q_j, V_osc low drives Q_j on.

Implementation of the fast drop-out switch through transistor Q₂ is substantially identical to that described previously for the voltage-controlled chopper of FIG. 2.

Overall operation of the current-controlled embodiment of the invention is shown in FIG. 7, with coil current shown in trace 700, armature position shown in trace 710, and contact position shown in trace 720. Operation of the embodiment for the pick-up interval and economizing interval is substantially identical to that previously described with regard to FIG. 3.

A third embodiment of the invention is demonstrated in FIG. 11 for an additional voltage-controlled chopper. This embodiment of the invention is substantially identical to that disclosed and described with regard to FIG. 2, however, MOSFET's have been substituted for switching transistors Q_j and Q₂. Resistors Rg, R_u and R₁₃ have been eliminated in this embodiment as unnecessary, and the inverting signal input for the gate amplifier A₂ is provided directly from the reference voltage ^_EF, eUminating resistors Ro, and R₁₀. An additional zener diode D₇ has been added to the circuit to accommodate voltage regulation for the MOSFET operating as switch Q₂. Values for the various components in the circuit are identified on the drawing, and amplifiers implemented for A_1? A₂ and A₃ are 1/4 LM339's with pin inputs as identified in the drawing. Having now described the invention in detail as required by the patent statutes, those skilled in the art will recognize modifications and substitutions to the specific embodiments disclosed to accommodate particular applications. Such substitutions and modifications are within the scope and intent of the present invention as defined in the following claims.

Claims

WHAT IS CLAIMED IS:

1. A control circuit for coil current in a spring biased DC actuator having a coil circuit receiving power from a voltage source, the control circuit comprising: a switch in the coil circuit and controllable by a switch control signal to close the coil circuit; a power switching circuit connected to the switch to provide the switch control signal, the power switching circuit having a threshold detector providing a triggering signal for initiating operation of the control circuit responsive to a predetermined source voltage, a gate signal generator responsive to the triggering signal and having a time constant sufficient to drive the coil through a pickup interval, and means for generating the switch control signal responsive to the gate signal and having a duty cycle responsive to the voltage of the voltage source.

2. A control circuit as defined in claim 1 wherein the generating means comprises: a voltage controlled oscillator connected to sense the source voltage; and a capacitor charged by feedback from the oscillator, the capacitor connected to an input of the oscillator to control the duty cycle.

3. A control circuit as defined in claim 1 wherein the generating means comprises: a current controlled oscillator connected to sense the coil current, the oscillator modifying its duty cycle responsive to the current.

4. A control circuit for coil current in a spring biased DC actuator having a coil circuit receiving power from a voltage source, the control circuit comprising: a chopper circuit for periodically interrupting power to the coil; a sustaining current return circuit providing a current return for the coil when power is interrupted by the chopper; and a dropout switch for opening the current return circuit responsive to removing power to the coil circuit.

5. A control circuit as defined in claim 4 wherein the dropout switch comprises a transistor switched by the voltage source.

6. A control circuit as defined in claim 5 further comprising a zener diode in parallel with the dropout switch for power dissipation.

7. A control circuit for coil current in a spring biased DC actuator having a coil circuit receiving power from a voltage source, the control circuit comprising: a switch in the coil circuit and controllable by a switch control signal to close the coil circuit; a power switching circuit connected to the first switch to provide the switch control signal, the power switching circuit having a threshold detector providing a triggering signal for initiating operation of the control circuit responsive to a predetermined source voltage, a gate signal generator responsive to the triggering signal and having a time constant sufficient to drive the coil through a pickup interval, means for generating the switch control signal responsive to the gate signal and having a duty cycle responsive to the voltage of the voltage source; a sustaining current return circuit providing a current return for the coil when power in interrupted by the chopper; and a dropout switch for opening the current return circuit responsive to removing power to the coil circuit.

8. A control circuit as defined in claim 7 wherein the dropout switch comprises a transistor switched by the voltage source.

9. A control circuit as defined in claim 9 further comprising a zener diode in parallel with the dropout switch for power dissipation.